JP2007082380A - Synchronous motor control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a synchronous motor control device capable of reducing an amount of computation needed for the estimation of speed. <P>SOLUTION: A first estimated pole position θ<SB>es1</SB>is computed according to impressed voltage command values to synchronous motor V<SP>*</SP><SB>α</SB>, V<SP>*</SP><SB>β</SB>and current detected values i<SB>α</SB>, i<SB>β</SB>. In a loop consisting of a deviation calculator 11, a compensator 12, and an integrator 13; a speed estimated value ω<SB>es</SB>is determined by multiplying a deviation between the first estimated pole position θ<SB>es1</SB>and the second estimated pole position θ<SB>es2</SB>by a predetermined gain, and the result of integrating the speed estimated value ω<SB>es</SB>is let to be the second estimated pole position θ<SB>es2</SB>. As a result, a rotor speed for making a deviation between a first rotor pole position and a second rotor pole position to be zero can be computed as a speed estimated value ω<SB>es</SB>. Therefore, the amount of computation to determine the speed estimated value ω<SB>es</SB>can be substantially reduced. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a synchronous motor control device that controls a synchronous motor having a permanent magnet as a rotor.

  For example, when operating a synchronous motor (SPMSM, IPMSM) using a permanent magnet as a rotor, the energization phase to the stator winding is controlled in synchronization with the position of the rotor (magnetic pole position). A position detector is needed to detect the position of the child. As this position detector, a Hall element, an encoder, a resolver, or the like can be used. However, when such a position detector is used, it is difficult to reduce the size of the motor, and it is necessary to perform connection wiring with a control device that controls the motor.

  For this reason, in recent years, various sensorless control devices for obtaining the magnetic pole position and speed of the rotor necessary for controlling the motor without using a position detector for detecting the magnetic pole position of the rotor have been proposed. For example, in Non-Patent Document 1, an extended induced voltage model shown in the following Equation 3 is used as a model of a synchronous motor.

上記の数式１において、右辺第２項が拡張誘起電圧として定義される。この拡張誘起電圧モデルは、永久磁石による誘起電圧に加え、リラクタンストルクを生じるインダクタンスの差を新たに磁束成分として扱うものである。このような拡張誘起電圧モデルを用いると、磁極位置の推定演算を容易にするメリットがある。なお、ＳＰＭＳＭの場合、Ｌ_ｄ＝Ｌ_ｑとなり、拡張誘起電圧は、永久磁石の誘起電圧項のみによって決まる。そして、上記数式１に示されるモデルに対して、拡張誘起電圧を外乱とみなした外乱オブザーバを構成することにより、拡張誘起電圧を求めることができる。

In the above mathematical formula 1, the second term on the right side is defined as the extended induced voltage. In this extended induced voltage model, a difference in inductance that generates reluctance torque is newly treated as a magnetic flux component in addition to an induced voltage caused by a permanent magnet. Use of such an extended induced voltage model has an advantage of facilitating the calculation of the magnetic pole position. In the case of SPMSM, L _d = L _q , and the expansion induced voltage is determined only by the induced voltage term of the permanent magnet. Then, an extended induced voltage can be obtained by configuring a disturbance observer in which the extended induced voltage is regarded as a disturbance with respect to the model expressed by Equation 1 above.

Further, the rotor speed ω _es is estimated by performing adaptive identification of the speed from the model of the synchronous motor related to the expansion induced voltage. This adaptive identification of speed is given by Equation 4 below.

“Position / Speed Sensorless Control of Cylindrical Brushless DC Motor by Disturbance Observer and Velocity Adaptive Identification” 1998 Industrial Society of Japan Vol.118-D7 / 8, page 828-835

  As described above, when the rotor speed is estimated by adaptive identification, it is necessary to perform a complicated calculation as shown in Formula 4, which causes a problem that the calculation amount is enormous.

  The present invention has been made in view of the above points, and an object of the present invention is to provide a synchronous motor control device capable of reducing the amount of calculation required for speed estimation.

In order to achieve the above object, a synchronous motor control device according to claim 1 controls a drive signal output to a synchronous motor based on a deviation between a speed command value and a speed estimation value,
Motor current detection means for detecting the motor current flowing in the stator winding of the synchronous motor;
First estimation calculation means for estimating and calculating the first rotor magnetic pole position based on the command value of the voltage applied to the synchronous motor and the detected value of the motor current;
Second estimation calculation means for estimating and calculating the second rotor magnetic pole position by integrating the speed estimation value of the motor magnetic pole calculated by the following speed estimation value calculation means;
Based on a deviation between the first rotor magnetic pole position and the second rotor magnetic pole position, a speed estimated value calculating means for calculating a speed of the rotor for setting the deviation to zero as a speed estimated value is provided. It is characterized by that.

  According to the above-described configuration, it is only necessary to obtain the rotor speed for making the deviation between the first rotor magnetic pole position and the second rotor magnetic pole position zero as the estimated speed value. Therefore, the amount of calculation for obtaining the speed estimated value can be significantly reduced as compared with the conventional case.

  Since the second rotor magnetic pole position is obtained by integrating the estimated speed value, the position is different (delayed) in time from the first rotor magnetic pole position by the period of the integration calculation. Will be shown. In other words, the deviation between the first rotor magnetic pole position and the second rotor magnetic pole position corresponds to the amount of change in the rotor magnetic pole position during the above-described integration calculation processing period. Accordingly, by calculating the estimated speed value so that the deviation between the first rotor magnetic pole position and the second rotor magnetic pole position is zero, the accurate rotor speed can be obtained.

  According to a second aspect of the present invention, the speed estimated value calculating means calculates the speed estimated value by multiplying a deviation between the first rotor magnetic pole position and the second rotor magnetic pole position by a predetermined gain. Can be configured. This is because the estimated speed value has a relationship that increases or decreases depending on the deviation between the first rotor magnetic pole position and the second rotor magnetic pole position.

  In addition, as described in claim 3, the first first estimation calculation means calculates an induced voltage using an induced voltage model represented by Formula 1 in the α-β coordinate system, and this calculation is performed. The first rotor magnetic pole position may be estimated and calculated from the induced voltage, or, as described in claim 4, the wave of the motor current represented by Formula 2 in the α-β coordinate system. The induced voltage may be calculated by an approximate induced voltage model assuming that the time derivative of the high value is zero, and the first rotor magnetic pole position may be estimated and calculated from the calculated induced voltage.

  According to the configuration of the third aspect, since the induced voltage is calculated based on the detailed motor model, the speed responsiveness and the load fluctuation responsiveness can be kept good for the magnetic pole position calculated from the induced voltage. it can. However, in this case, the amount of calculation increases because the motor model includes a differential term. In this regard, if the configuration described in claim 4 is adopted, the induced voltage can be calculated using an approximate model in which the differential term is omitted, so the amount of calculation for obtaining the first rotor magnetic pole position can be reduced. .

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing the overall configuration of the control device for the synchronous motor 8 according to the present embodiment. The synchronous motor 8 in the present embodiment has a three-phase (U-phase, V-phase, W-phase) stator winding and a rotor composed of permanent magnets.

In FIG. 1, the controller 2 receives a deviation between the speed command ω ^* and the estimated speed value ω _es from the deviation calculation unit 1 and a d-axis current value i _d and a q-axis current value i _q from an αβ / dq conversion unit 14 described later. Are input, and based on these input values, the d-axis voltage command value V ^* _d and the q-axis voltage command value V ^* _q are calculated and output so that the speed estimation value ω _es approaches the speed command ω ^*. . The d-axis voltage command value V ^* _d and the q-axis voltage command value V ^* _q output from the controller 2 are output to the dq / αβ coordinate conversion unit 3. Here, as is well known, the dq coordinate is, for example, a rotational coordinate defined by the q axis perpendicular to the d axis, where the direction from the S pole to the N pole of the rotor is the d axis, and the αβ coordinate is The stationary coordinates are defined by the α axis and the β axis orthogonal to each other with the rotation axis of the rotor as the origin.

The coordinate conversion from the dq rotation coordinate to the αβ static coordinate is executed based on the second estimated magnetic pole position θ _es2 of the rotor output from the integrator 13 described later. If the magnetic pole position of the rotor can be estimated, the relative phase relationship between the dq rotation coordinates and the αβ stationary coordinates can be specified, so that coordinate conversion can be performed. By this coordinate conversion, the dq / αβ coordinate conversion unit 3 outputs the α-axis voltage command value V ^* _α and the β-axis voltage command value V ^* _β .

The α-axis voltage command value V ^* _α and β-axis voltage command value V ^* _β from the dq / αβ coordinate conversion unit 3 are supplied to the αβ / uvw conversion unit 4 and the magnetic pole position estimator 10 which are two-phase to three-phase conversion units. Given. The αβ / uvw conversion unit 4 determines each stator winding (U phase, V phase, W) of the three-phase synchronous motor 8 from the α axis voltage command value V ^* _α and β axis voltage command value V ^* _β in αβ static coordinates. U-phase voltage command value V ^* _u , V-phase voltage command value V ^* _v , and W-phase voltage command value V ^* _w that are to be output to (phase).

The PWM signal generator 5 is configured to output each stator winding of the inverter in the power converter 6 based on the U-phase voltage command value V ^* _u , the V-phase voltage command value V ^* _v , and the W-phase voltage command value V ^* _w . The PWM signal for driving the switching element corresponding to is generated. By driving each switching element constituting the inverter by this PWM signal, voltages corresponding to the U-phase voltage command value V ^* _u , the V-phase voltage command value V ^* _v, and the W-phase voltage command value V ^* _w are set. Can be applied to the stator windings.

The current sensor 7 detects a current flowing through each stator winding. The detected current value of each stator winding detected by the current sensor 7 is input to the uvw / αβ converter 9 and converted into an α-axis current i _α and a β-axis current i _β in αβ coordinates. The current sensor 7 does not need to detect the currents of all the three-phase stator windings, but detects the two-phase currents, and the remaining one-phase current is estimated from the two-phase currents. Also good.

The magnetic pole position estimator 10 includes α-axis and β-axis voltage command values V ^* _α and V ^* _β input from the dq / αβ coordinate converter 3 and α-axis and β-axis input from the uvw / αβ converter 9. The rotor magnetic pole position is estimated based on the detected current values i _α and i _β . As this magnetic pole position estimator 10, for example, “IPMSM position / velocity sensorless control using an extended induced voltage disturbance observer”, Ichikawa, Chen, Tomita, Michiki, Okuma, 2000 The described disturbance observer can be applied. Here, the disturbance observer will be briefly described.

The extended induced voltage model of the synchronous motor is expressed as Equation 3 described above. In Equation 3, it is possible to obtain the extended induced voltage by configuring a disturbance observer that regards the extended induced voltage as a disturbance. In this case, the expansion induced voltages e _α and e _β are expressed by the following formula 5.

上記の数式５によって得られた拡張誘起電圧ｅ_α、ｅ_βに対して、以下の数式６に示すように逆正接演算を行うことにより、回転子の第１の推定磁極位置θ_ｅｓ１を求めることができる。

The first estimated magnetic pole position θ _es1 of the rotor is obtained by performing an arctangent calculation on the extended induced voltages e _α and e _β obtained by the above mathematical formula 5 as shown in the following mathematical formula 6. Can do.

上述した例では、詳細なモータモデルを基に誘起電圧を演算するため、その誘起電圧から演算される磁極位置について、速度応答性や負荷変動応答性を良好に保つことができる。ただし、この場合、モータモデルには微分項が含まれるので、演算量が増加する。この点を考慮し、微分項を省略した近似モデルにより誘起電圧を演算するようにしても良い。これにより、第１の回転子磁極位置を求めるための演算量を低減することができる。微分項を省略する場合、電流の波高値Ｉ_ａは一定であり、その時間微分はゼロであるとみなす。この仮定により、例えばα軸電流ｉ_αの微分項は以下の数式７のように近似できる。

In the above-described example, the induced voltage is calculated based on the detailed motor model, so that the speed responsiveness and the load fluctuation responsiveness can be kept good for the magnetic pole position calculated from the induced voltage. In this case, however, the amount of calculation increases because the motor model includes a differential term. In consideration of this point, the induced voltage may be calculated by an approximate model in which the differential term is omitted. As a result, the amount of calculation for obtaining the first rotor magnetic pole position can be reduced. When omitting the differential term, the peak value I _a current is constant, the time derivative is regarded as being zero. With this assumption, for example, the differential term of the α-axis current i _α can be approximated as shown in Equation 7 below.

同様にして、β軸電流ｉ_βの微分項の近似も行うことにより、上記した数式５は、以下の数式８のように簡略化できる。

Similarly, by performing approximation of the differential term of the β-axis current i _β , Equation 5 described above can be simplified as Equation 8 below.

上述したようにして、磁極位置推定器１０において、拡張誘起電圧ｅ_α、ｅ_β及び回転子の第１の推定磁極位置θ_ｅｓ１が算出される。この算出された第１の推定磁極位置θ_ｅｓ１は、偏差演算部１１に与えられる。この偏差演算部１１は、磁極位置推定器１０から出力された第１の推定磁極位置θ_ｅｓ１と後述する積分器１３から出力される第２の推定磁極位置θ_ｅｓ２との偏差を算出して出力するものである。この第１の推定磁極位置θ_ｅｓ１と第２の推定磁極位置θ_ｅｓ２との偏差は、補償器１２に出力される。

As described above, the magnetic pole position estimator 10 calculates the expansion induced voltages e _α and e _β and the first estimated magnetic pole position θ _es1 of the rotor. The calculated first estimated magnetic pole position θ _es1 is given to the deviation calculator 11. The deviation calculation unit 11 calculates the deviation between the second estimated magnetic pole position theta _es2 output from the integrator 13 to be described later to the first estimated magnetic pole position theta _es1 output from the magnetic pole position estimator 10 outputs To do. The deviation between the first estimated magnetic pole position θ _es1 and the second estimated magnetic pole position θ _es2 is output to the compensator 12.

The compensator 12 is configured to calculate the estimated speed value ω _es by multiplying the input deviation by a predetermined gain. For example, the compensator 11 calculates the estimated speed value ω _es by multiplying the deviation by a predetermined PI gain (Kp, Ki) as shown in Equation 9 below.

このようにして算出された速度推定値ω_ｅｓは、上述した偏差演算部１に出力されるとともに、積分器１３に出力される。積分器１３は、速度推定値ω_ｅｓを積分することによって、回転子の第２の推定磁極位置θ_ｅｓ２を算出して、ｄｑ／αβ座標変換部３、偏差演算部１１、及びαβ／ｄｑ座標変換部１４へ出力する。

The speed estimated value ω _es calculated in this way is output to the above-described deviation calculating unit 1 and also output to the integrator 13. The integrator 13 integrates the speed estimation value ω _es to calculate the second estimated magnetic pole position θ _es2 of the rotor, and the dq / αβ coordinate conversion unit 3, the deviation calculation unit 11, and the αβ / dq coordinate. The data is output to the conversion unit 14.

The αβ / dq coordinate converter 14 is based on the second estimated magnetic pole position θ _es2 given from the integrator 13 and detects the α axis and β axis current detection values i _α , i input from the uvw / αβ converter 9. _β is converted into d-axis and q-axis current values i _d and i _q and output to the controller 2.

Here, in the present embodiment, the second estimated magnetic pole position θ _es2 is obtained by integrating the speed estimated value ω _es in the loop including the deviation calculating unit 11, the compensator 12, and the integrator 13. For this reason, the second estimated magnetic pole position θ _es2 indicates a position _that is temporally different (delayed) from the first estimated magnetic pole position θ _es1 by the period for performing the integral calculation. In other words, the deviation between the first estimated magnetic pole position θ _es1 and the second estimated magnetic pole position θ _es2 corresponds to the amount of change in the magnetic pole position of the rotor during the above-described integration calculation processing period. Therefore, if a speed estimation value is obtained in which the deviation between the first rotor magnetic pole position and the second rotor magnetic pole position is zero during the integration calculation processing cycle, an accurate rotor speed can be obtained. it can.

As described above, the calculation for _{obtaining the} estimated speed value ω _es is executed in a loop including the deviation calculation unit 11, the compensator 12, and the integrator 13. In this loop, the calculation for _obtaining the rotor speed for _setting the deviation between the first estimated magnetic pole position θ _es1 and the second estimated magnetic pole position θ _es2 to zero is performed by adaptive identification of the speed as in the prior art. Since the calculation is simpler than the calculation to be performed, the calculation amount for _obtaining the speed estimated value ω _es can be greatly reduced.

Note that the estimated speed value ω _es has a relationship that is increased or decreased according to the deviation between the first estimated magnetic pole position θ _es1 and the second estimated magnetic pole position θ _es2 . For this reason, the estimated value ω _es can be obtained by multiplying the deviation by a predetermined gain in the compensator 12.

  As mentioned above, although preferable embodiment of this invention was described, this invention can be implemented in various deformation | transformation, without being restrict | limited to the embodiment mentioned above at all.

For example, in the above-described embodiment, the rotor speed for making the deviation between the first estimated magnetic pole position θ _es1 and the second estimated magnetic pole position θ _es2 zero is obtained as the estimated speed value ωes. However, the first estimated magnetic pole position theta _es1 results obtained by multiplying a predetermined gain a deviation between the second estimated magnetic pole position theta _es2, is defined as the estimated speed error [Delta] [omega _es, the estimated speed error [Delta] [omega _es and the speed command value A result obtained by adding ω ^* may be used as the speed estimated value ω _es .

In the above-described embodiment, the coordinate conversion from the dq coordinate system to the αβ coordinate system is performed, but the voltage command value and the current detection value in the dq coordinate system are used without performing the coordinate conversion to the αβ coordinate system. The first estimated magnetic pole position _es1 may be obtained. Conversely, the voltage command value in the αβ coordinate system may be directly calculated from the deviation between the speed command value ω ^* and the estimated speed value ω _es instead of the voltage command value in the dq coordinate system.

It is a block diagram which shows the whole structure of the control apparatus of the synchronous motor by embodiment.

Explanation of symbols

2 ... Controller 3 ... dq / αβ coordinate converter 4 ... αβ / uvw converter 5 ... PWM signal generator 6 ... Power converter 7 ... Current sensor 8 ... Synchronous motor 9 ... uvw / αβ converter 10 ... Magnetic pole position estimation 11: Deviation calculation unit 12 ... Compensator 13 ... Integrator

Claims (4)

In the synchronous motor control device that controls the drive signal output to the synchronous motor based on the deviation between the speed command value and the speed estimated value,
Motor current detecting means for detecting a motor current flowing in the stator winding of the synchronous motor;
First estimation calculation means for estimating and calculating a first rotor magnetic pole position based on a command value of a voltage applied to the synchronous motor and a detection value of the motor current;
Second estimation calculation means for estimating and calculating a second rotor magnetic pole position by integrating the speed estimation value of the motor magnetic pole calculated by the following speed estimation value calculation means;
Based on a deviation between the first rotor magnetic pole position and the second rotor magnetic pole position, speed estimated value calculating means for calculating the speed of the rotor for making the deviation zero as the speed estimated value; A synchronous motor control device comprising:   2. The speed estimated value calculating means calculates the speed estimated value by multiplying a deviation between the first rotor magnetic pole position and the second rotor magnetic pole position by a predetermined gain. The synchronous motor control device described in 1. The first estimation calculation means calculates an induced voltage by an induced voltage model represented by the following formula 1 in the α-β coordinate system, and estimates the first rotor magnetic pole position from the calculated induced voltage. The synchronous motor control device according to claim 1, wherein the synchronous motor control device performs calculation.

The first estimation calculation means calculates an induced voltage by an approximate induced voltage model in which the time derivative of the peak value of the motor current is assumed to be zero, represented by the following formula 2 in the α-β coordinate system. The synchronous motor control device according to claim 1, wherein the first rotor magnetic pole position is estimated and calculated from the induced voltage.

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