JP5273450B2 - Motor control device - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller that makes it possible to continue to drive a motor even when a rotation sensor becomes faulty. <P>SOLUTION: A sensor failure determination unit 25 determines whether or not a resolver 2 is faulty. Under normal conditions under which the resolver 2 is not faulty, a motor 1 is driven by a sine wave by a d- and q-axis target current value computation unit 16, a d- and q-axis voltage command value computation unit 19, a voltage command value coordinate transformation unit 20, a PWM control unit 21, and the like. When the resolver 2 is faulty, the motor 1 is driven by a rectangular wave in a predetermined energization pattern by an energization pattern drive unit 24. When the operating torque detected by a torque sensor 7 is large, driving of the motor 1 by the energization pattern drive unit 24 is stopped so as to avoid step-out. <P>COPYRIGHT: (C)2010,JPO&amp;INPIT

Description

  The present invention relates to a motor control device for driving a brushless motor. The brushless motor is used, for example, as a source for generating a steering assist force in an electric power steering apparatus.

A motor control device for driving and controlling a brushless motor is generally configured to control the supply of motor current in accordance with the output of a rotation sensor for detecting the rotational position of the rotor. As the rotation sensor, for example, a resolver that outputs a sine wave signal and a cosine wave signal corresponding to the rotor rotational position (electrical angle) is used.
If a failure occurs in the rotation sensor, it becomes impossible to uniquely specify the rotor rotation position, so that it is impossible to continue the drive control of the brushless motor. Therefore, the brushless motor is stopped when the rotation sensor fails.

The failure of the rotation sensor includes a failure of the rotation sensor itself and a disconnection failure of the signal line or the feed line of the rotation sensor.
JP 2001-112282 A

In the electric power steering apparatus, if the motor stops, the steering assist force cannot be obtained, so that the steering feeling is inevitably deteriorated. Therefore, it is preferable that the drive of the motor can be continued as much as possible even when the rotation sensor fails.
Not only the motor as the drive source of the electric power steering apparatus but also the brushless motor for other uses, it is preferable to continue the motor drive as much as possible even after the failure of the rotation sensor.

  SUMMARY OF THE INVENTION An object of the present invention is to provide a motor control device that can continue to drive a motor even when a failure occurs in a rotation sensor.

This invention includes a rotor (50), a stator that faces the rotor (55) a motor control device for controlling the motor (1) with a (10, 10A), the rotational position of the rotor A rotation sensor (2) for detecting the motor, and when the rotation sensor is normal, the motor is controlled on the basis of an output signal of the rotation sensor. A motor control apparatus including control means (12, 12A) for controlling the motor without using the output signal of the rotation sensor in the added control mode is provided . The alphanumeric characters in parentheses indicate corresponding components in the embodiments described later. The same applies hereinafter.

According to this configuration, when a failure occurs in the rotation sensor, the motor is controlled in a control mode in which a limitation is imposed on the control mode when the rotation sensor is normal. This enables so-called life extension control when the rotation sensor is out of order, and the rotation of the motor can be continued.
More specifically, the invention of this, the motor is a steering motor which imparts steering force to a steering mechanism of the vehicle (1), wherein the control means of the rotation sensor at the time of failure, the vehicle it on condition operation torque applied to the operation member for steering is below a predetermined torque threshold state, and are not to drive the motor in a predetermined energization pattern, the torque threshold at the predetermined energization pattern unless that have been established to a value that is possible is no possibility of causing step-out when driving the motor (24, S5~S8).
The steering motor imparts a steering force according to the operation torque applied to the operation member to the steering mechanism. Therefore, when a large operating torque is applied to the operating member, a large current is supplied to the steering motor, and a large steering torque is generated from the steering motor. When an accurate rotational position by the rotation sensor is not detected, if the motor is driven by a predetermined energization pattern to generate a large torque, the motor may step out. Therefore, in the present invention, the motor is driven by a predetermined energization pattern on the condition that the operation torque applied to the operation member is a predetermined value or less. Thus, the motor is continuously driven using the predetermined energization pattern under the condition that no step-out occurs when the rotation sensor fails.

The failure of the rotation sensor can be detected by known means. For example, when the signal line for transmitting the output signal of the rotation sensor is pulled up to the power supply voltage or pulled down to the ground potential, and the output signal of the rotation sensor is not derived, the power supply voltage or the ground potential is applied to the signal line. You should make it appear. Thereby, based on the fact that the potential of the signal line is fixed to the power supply voltage or the ground potential, it is possible to detect a failure of the rotation sensor (including disconnection of the signal line and short-circuit failure).

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
FIG. 1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to an embodiment of the present invention is applied. This electric power steering apparatus includes a torque sensor 7 that detects an operation torque applied to a steering wheel of a vehicle, a vehicle speed sensor 8 that detects the speed of the vehicle, a motor 1 that applies a steering assist force to the steering mechanism 3 of the vehicle, And a motor control device 10 for driving and controlling the motor 1. The motor control device 10 drives the motor 1 in accordance with the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8, thereby realizing appropriate steering assistance according to the steering situation. The motor 1 is, for example, a three-phase brushless motor, and, as schematically shown in FIG. 2A, a rotor 50 as a field and U-phase, V-phase, and W-phase stator windings 51, 52, And a stator 55 including 53. The motor 1 may be an inner rotor type in which a stator is arranged outside the rotor, or may be an outer rotor type in which a stator is arranged inside a cylindrical rotor.

The motor control device 10 includes a current detection unit 11, a microcomputer 12 as a signal processing unit, and a drive circuit 13. The above-described torque sensor 7 and vehicle speed sensor 8 are connected to the motor control device 10 together with the resolver 2 (rotation sensor) that detects the rotational position of the rotor in the motor 1.
The current detector 11 detects the current flowing through the stator windings 51, 52, 53 of the motor 1. More specifically, the current detection unit 11 includes current detectors 11u, 11v, and 11w that detect phase currents in three-phase (U-phase, V-phase, and W-phase) stator windings 51, 52, and 53, respectively. .

  The microcomputer 12 includes a plurality of function processing units realized by program processing (software processing). The plurality of function processing units include a basic target current value calculation unit 15, a dq axis target current value calculation unit 16, a dq axis current calculation unit 17, a d axis deviation calculation unit 18d, a q axis deviation calculation unit 18q, and a dq axis. Voltage command value calculation unit 19, voltage command value coordinate conversion unit 20, PWM (pulse width modulation) control unit 21, position calculation unit 22, energization pattern drive unit 24, sensor failure determination unit 25, switching unit 26, torque determination unit 27 The torque direction determination unit 28, the deviation calculation unit 29, the PI control unit 30, and the like are included.

  The drive circuit 13 is configured by an inverter circuit, and a control signal from the PWM control unit 21 or the energization pattern drive unit 24 is supplied from the switching unit 26. By being controlled by this control signal, the drive circuit 13 uses the power from a power source (not shown) such as an in-vehicle battery to supply the U-phase, V-phase, and W-phase stator windings 51, 52, 53 of the motor 1. To supply.

The position calculation unit 22 calculates the rotational position (electrical angle) θ of the rotor 50 of the motor 1 based on the output signal of the resolver 2.
The basic target current value calculation unit 15 calculates the basic target current value I ^* of the motor 1 based on the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8. For example, the basic target current value I ^* is determined so as to increase as the operating torque increases and to increase as the vehicle speed decreases.

The basic target current value I ^* calculated by the basic target current value calculation unit 15 is input to the dq-axis target current value calculation unit 16. The dq-axis target current value calculation unit 16 calculates a d-axis target current value I _d ^* for generating a magnetic field in the d-axis direction and a q-axis target current value I _q ^* for generating a magnetic field in the q-axis direction. Calculate. However, the d-axis is an axis along the magnetic flux direction of the field of the rotor 50 of the motor 1, and the q-axis is an axis orthogonal to the d-axis and the rotor rotation axis. The dq coordinate plane is a plane along the rotation direction of the rotor 50, and the d axis and the q axis define a rotation coordinate system that rotates together with the rotor 50 (see FIG. 2). The calculation in the dq-axis target current value calculation unit 16 can be performed using a known calculation formula.

Phase currents I _u , I _v , I _w (hereinafter, collectively referred to as “three-phase detection current I _uvw ”) output from the current detection unit 11 are input to the dq-axis current calculation unit 17. The dq-axis current calculation unit 17 performs coordinate conversion of the phase currents I _u , I _v , and I _w based on the rotational position θ calculated by the position calculation unit 22, so that the d-axis current value I _d and the q-axis current are converted. The value I _q (the detected current value as the motor current value) is calculated. The calculation in the dq axis current calculation unit 17 can be performed using a known calculation formula.

The d-axis deviation calculating unit 18d obtains a deviation (d-axis deviation) δI _d between the d-axis target current value I _d ^* and the d-axis current I _d . Similarly, the q-axis deviation calculating unit 18q obtains a deviation (q-axis deviation) δI _q between the q-axis target current value I _q ^* and the q-axis current I _q .
The dq-axis voltage command value calculation unit 19 obtains a d-axis voltage command value V _d ^* corresponding to the d-axis deviation δI _d and a q-axis voltage command value V _q ^* corresponding to the q-axis deviation δI _q .

The voltage command value coordinate conversion unit 20 performs coordinate conversion of the d-axis voltage command value V _d ^* and the q-axis voltage command value V _q ^* on the basis of the rotational position θ calculated by the position calculation unit 22, and performs a U-phase stator. Applied voltage command values V _u ^* , V _v ^* , and V _w ^* to be applied to the winding 51, the V-phase stator winding 52, and the W-phase stator winding 53 are calculated. The calculation in the voltage command value coordinate conversion unit 20 may be performed using a known calculation formula.

The PWM control unit 21 generates a PWM control signal for each phase, which is a pulse signal having a duty ratio corresponding to the applied voltage command values V _u ^* , V _v ^* , and V _w ^* . By applying this PWM signal to the drive circuit 13 via the switching unit 26, voltages corresponding to the d-axis voltage command value V _d ^* and the q-axis voltage command value V _q ^* are transferred from the drive circuit 13 to the stator windings of each phase. Applied to the lines 51, 52, 53, a rotational force of the rotor 50 is generated.

The energization pattern driving unit 24 generates a control signal for driving the motor 1 in accordance with a predetermined energization pattern by the 120-degree energization method. The energization pattern driving unit 24 includes a PWM duty setting unit 24a. The energization pattern driving unit 24 generates a rectangular wave driving signal that is pulse-width modulated with the duty set by the PWM duty setting unit 24a.
The sensor failure determination unit 25 determines whether or not there is a failure in the resolver 2. For example, the sensor failure determination unit 25 can detect a failure of the resolver 2, a disconnection failure of the signal line 2a, and a grounding failure of the signal line 2a by monitoring a signal derived to the signal line 2a of the resolver 2. . More specifically, the signal line 2a between the resolver 2 and the motor control device 10 is connected to the power supply potential via a pull-up resistor or connected to the ground potential via a pull-down resistor. Can do. In this case, when the signal line 2a is disconnected, the signal (sine signal or cosine signal) from the resolver 2 is not derived to the signal line 2a. Instead, the signal line 2a is fixed to the power supply potential or the ground potential. Is done. Therefore, the sensor failure determination unit 25 can determine whether or not the resolver 2 has failed (including signal line failure) by determining whether or not the signal line 2a is fixed at the power supply potential or the ground potential. . Of course, other known methods may be applied to the failure detection of the resolver 2.

The switching unit 26 supplies a control signal from the PWM control unit 21 to the drive circuit 13 in a normal state in which the sensor failure determination unit 25 determines that there is no failure. On the other hand, in a failure state in which the sensor failure determination unit 25 determines that there is a failure, the switching unit 26 supplies the drive circuit 13 with a control signal generated by the energization pattern drive unit 24.
The torque determination unit 27 compares the magnitude of the operation torque detected by the torque sensor 7 with a predetermined torque threshold (for example, 7 Nm), and gives the comparison result to the energization pattern driving unit 24. The torque threshold is determined to be as large as possible within a range where there is no possibility of stepping out when the motor 1 is driven with a predetermined energization pattern by the energization pattern driving unit 24.

The torque direction determination unit 28 determines whether the direction of the operation torque detected by the torque sensor 7 is clockwise torque or counterclockwise torque, and gives the determination result to the energization pattern driving unit 24. However, the clockwise direction corresponds to the direction in which the steering wheel is steered in the right direction, and the counterclockwise direction corresponds to the direction in which the steered wheel is steered in the left direction.
The deviation calculation unit 29 is obtained by converting the basic target current value I ^* set by the basic target current value calculation unit 15 and the q-axis current value I _q generated by the UVW / dq coordinate conversion unit 17 by the conversion unit 31. The deviation δI (= I ^* −I) from the phase current amplitude (detected current value) I (refer to equation (2) described later, I = −√ (2/3) I _q ) is obtained.

The PI control unit 30 obtains the applied voltage command values V _u ^* , V _v ^* , and V _w ^* by performing proportional-integral control on the deviation δI.
The PWM duty setting unit 24 a provided in the energization pattern driving unit 24 sets the PWM duty corresponding to the applied voltage command values V _u ^* , V _v ^* , and V _w ^* calculated by the PI control unit 30.

FIG. 3 is an electric circuit diagram for explaining a specific configuration of the drive circuit 13. A U-phase current I _u , a V-phase current I _v and a W-phase current I _w are given from the drive circuit 13 to the U-phase stator winding 51, the V-phase stator winding 52 and the W-phase stator winding 53 of the motor 1. It is supposed to be.
The drive circuit 13 includes a series circuit 40U corresponding to the U phase, a series circuit 40V corresponding to the V phase, and a series circuit 40W corresponding to the W phase, which are for a power supply (on-vehicle battery, not shown). Connected in parallel. The series circuit 40U is configured by connecting a high-side power switching element 41U and a low-side power switching element 42U in series, and connecting a connection point 43U between them to the U-phase stator winding 51 of the motor 1. . Similarly, the series circuit 40V includes a high-side power switching element 41V and a low-side power switching element 42V connected in series, and a connection point 43V between them is connected to the V-phase stator winding 52 of the motor 1. Has been. The series circuit 40W is configured by connecting a high-side power switching element 41W and a low-side power switching element 42W in series, and connecting a connection point 43W between them to the W-phase stator winding 53 of the motor 1. ing. The PWM control signal from the switching unit 26 is amplified as necessary and is supplied to the gates of the power switching elements 41U and 42U; 41V and 42V; 41W and 42W.

The dq axis voltage command value calculation unit 19 generates voltage command values V _d ^* and V _q ^* for sine wave drive, and a control signal corresponding to the voltage command values V _d ^* and V _q ^* is generated from the PWM control unit 21. Therefore, when the control signal generated by the PWM control unit 21 is supplied from the switching unit 26 to the drive circuit 13, sinusoidal phase currents are supplied to the U-phase, V-phase, and W-phase stator windings 51, 52, 53 of the motor 1. The motor 1 is driven in a sine wave so that I _u , I _v , and I _w flow.

4A and 4B are waveform diagrams illustrating an example of a control signal generated by the energization pattern driving unit 24. FIG. 4A illustrates a waveform (energization pattern) when torque in the clockwise direction from the motor 1 should be generated. 4B shows a waveform (energization pattern) when torque in the counterclockwise direction from the motor 1 should be generated.
The energization pattern driving unit 24 outputs a control signal including a PWM pulse to the high-side power switching elements 41U, 41V, and 41W of each phase over a period of 120 degrees electrical angle. During the 120-degree period in which the PWM pulse is applied to the high-side power switching elements 41U, 41V, 41W of the respective phases, the phases are shifted by 120 degrees. That is, the high-side power switching elements 41U, 41V, and 41W are cyclically PWM-controlled during each 120-degree period whose phase is shifted by 120 degrees. A voltage corresponding to the PWM duty is applied to each phase of the motor 1 during each 120-degree period under PWM control.

  On the other hand, the low-side power switching elements 42U, 42V, and 42W of each phase extend over a period of 120 degrees that is shifted by 60 degrees in electrical angle from the period in which the high-side power switching elements 41U, 41V, and 41W of each phase are PWM-controlled. Are turned on, and the remaining period is turned off. Accordingly, the phase is shifted by 120 degrees during the 120-degree period in which the low-side power switching elements 42U, 42V, and 42W of the respective phases are turned on. That is, the low-side power switching elements 42U, 42V, and 42W are cyclically turned on during each 120-degree period whose phase is shifted by 120 degrees.

In each phase, during the period in which all of the high-side power switching elements 41U, 41V, 41W and the low-side power switching elements 42U, 42V, 42W are in the OFF state, only the induced voltage is applied to the stator winding of the corresponding phase. appear.
In the clockwise energization pattern waveform shown in FIG. 4A, the phase of the V-phase drive waveform is delayed by 120 degrees with respect to the U-phase drive waveform, and the phase of the W-phase drive waveform is delayed by 120 degrees with respect to the V-phase drive waveform. I understand. On the other hand, in the counterclockwise energization pattern waveform shown in FIG. 4B, the phase of the V-phase drive waveform is delayed by 120 with respect to the W-phase drive waveform, and the phase of the U-phase drive waveform by 120 degrees with respect to the V-phase drive waveform. You can see that is behind.

FIG. 5 is a flowchart for explaining the control contents that the microcomputer 12 repeatedly executes at predetermined control cycles. The microcomputer 12 takes in the output signals of the torque sensor 7, the vehicle speed sensor 8, the resolver 2, and the current detector 11 (step S1). The basic target current value calculation unit 15 calculates the basic target current value I ^* based on the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8 (step S2). On the other hand, the sensor failure determination unit 25 determines the presence or absence of a failure of the resolver 2 based on a signal derived to the signal line 2a (step S3).

If there is no failure in the resolver 2 (step S3: NO), normal control is executed (step S4). More specifically, the d-axis target current value I _d ^* and the q-axis target current value I _q ^* are set by the dq-axis target current value calculation unit 16. Further, the three-phase detection current I _uvw detected by the current detection unit 11 is coordinate-converted by the dq-axis current calculation unit 17 to _obtain the d-axis detection current I _d and the q-axis detection current I _q . The d-axis deviation calculating unit 18d and the q-axis deviation calculating unit 18q obtain a d-axis current deviation δI _d (= I _d ^* −I _d ) and a q-axis current deviation δI _q (= I _q ^* −I _q ), respectively. The dq-axis voltage command value calculation unit 19 generates a d-axis voltage command value V _d ^* and a q-axis voltage command value V _q ^* by performing PI (proportional integration) calculation and the like on the current deviations δI _d and δI _q . . These are subjected to coordinate conversion by the voltage command value coordinate conversion unit 20 to generate UVW phase voltage command values V _u ^* , V _v ^* , V _w ^* . A PWM control signal corresponding to these voltage command values V _u ^* , V _v ^* , V _w ^* is generated by the PWM control unit 21. The switching unit 26 supplies this PWM control signal to the drive circuit 13. The rotation position θ calculated by the position calculation unit 22 is used for the coordinate conversion calculation in the dq axis current calculation unit 17 and the voltage command value coordinate conversion unit 20.

  On the other hand, when the sensor failure determination unit 25 determines that a failure has occurred in the resolver 2 (step S3: YES), the torque determination unit 27 determines whether the operation torque is equal to or less than a predetermined torque threshold (for example, 7 Nm). Is determined (step S5). If the operation torque is equal to or less than the torque threshold (step S5: YES), the torque direction determination unit 28 determines whether the direction of the operation torque is clockwise or counterclockwise based on the output signal of the torque sensor 7 ( Step S6). If the operation torque direction is clockwise, the energization pattern drive unit 24 generates a rectangular wave drive signal according to the clockwise energization pattern (see FIG. 4A) (step S7), and if the operation torque direction is counterclockwise. The energization pattern drive unit 24 generates a rectangular wave drive signal according to the counterclockwise energization pattern (see FIG. 4B) (step S8).

At this time, the PWM duty setting unit 24a sets the duty corresponding to the voltage command values V _u ^* , V _v ^* , and V _w ^* obtained by the PI control unit 30. A PWM control signal that is pulse-width modulated with the set duty is generated from the energization pattern driving unit 24. The switching unit 26 supplies this PWM control signal to the drive circuit 13. That is, as shown in FIG. 4A and FIG. 4B, the high-side power switching elements 41U, 41V, and 41W of each phase are cyclically PWM-controlled during each 120-degree period that is 120 degrees out of phase. In each 120 degree period, a voltage corresponding to the set duty is applied to each phase of the motor 1.

  On the other hand, when the operating torque exceeds the torque threshold (step S5: NO), the energization pattern driving unit 24 does not generate a control signal. Thereby, the motor 1 is stopped. When the operation torque is large enough to exceed the torque threshold, the motor 1 may step out in driving with a constant energization pattern. Therefore, in this embodiment, when the motor 1 is driven with a predetermined energization pattern by the energization pattern driving unit 24, it is a condition that the operation torque is equal to or less than the torque threshold value. In other words, the torque threshold value is determined to be as large as possible within a range in which the motor 1 can be driven normally without causing the step-out of the motor 1 by the driving with the predetermined energization pattern shown in FIGS. 4A and 4B. Yes.

As described above, according to this embodiment, when a failure occurs in the resolver 2, the rectangular wave drive by the 120-degree energization method is performed by the rectangular wave drive signal of the predetermined energization pattern generated by the energization pattern drive unit 24. Is done. Thereby, since the rotation of the motor 1 can be continued, a steering assist force can be applied to the steering mechanism 3.
Further, when the resolver 2 is normal, sinusoidal wave driving is performed, whereas when the resolver 2 has a failure, rectangular wave driving is performed, so that torque ripple occurs at the time of failure. As a result, the driver can be notified of the occurrence of abnormality and prompt early repair.

FIG. 6 is a block diagram showing an electrical configuration of the electric power steering apparatus according to the reference embodiment. In FIG. 6, parts corresponding to those shown in FIG. 1 are given the same reference numerals. This embodiment, the induced voltage detection unit 35 is provided with rotation speed estimation unit 36 and the phase estimator 37, the point that the torque determination unit 27 and the torque direction determining unit 28 is not provided, the implementation form described above Is different.

The induced voltage detection unit 35 measures the induced voltage of each phase of the motor 1 for each predetermined sampling period. More specifically, a period in which no voltage is applied to the stator windings 51, 52, 53 of each phase (a period in which both the high-side and low-side power switching elements of each phase are off. See FIGS. 4A and 4B). In addition, a voltage induced in the winding is detected.
The rotation speed estimation unit 36 estimates the rotation speed of the rotor from the induced voltage detected by the induced voltage detection unit 35. For example, the rotor rotational speed can be estimated by fitting a change in the induced voltage at two adjacent sampling points into a sine wave.

  The phase estimation unit 37 estimates the rotor phase (electrical angle) based on the induced voltage detected by the induced voltage detection unit 35 and the rotor rotation speed estimated by the rotation speed estimation unit 36. Specifically, by fitting an induced voltage detected at a predetermined sampling period to a sine wave, it is possible to estimate a zero cross point at which the polarity of the induced voltage switches. The phase of the rotor is estimated based on the elapsed time from the zero cross point estimated in this way and the rotor rotational speed. Based on the estimated phase, the energization pattern driving unit 24 obtains the phase switching timing. The energization pattern drive unit 24 raises and lowers the rectangular wave drive signal of each phase based on the obtained phase switching timing. More specifically, as shown in FIGS. 4A and 4B, the on / off pattern of the power switching element of the drive circuit 13 is switched at a phase of every 60 degrees.

Thus, according to this reference mode, when the resolver 2 fails, the rotor phase is estimated based on the zero cross point estimated from the off-phase induced voltage and the rotor rotational speed, and the estimated rotor phase is Based on this, the motor 1 can be driven by a 120-degree energization type rectangular wave drive. Therefore, even after the resolver 2 breaks down, the motor 1 can be continuously driven and the steering assist can be continued. Moreover, unlike the first embodiment, there is no limit due to the operating torque.

When the resolver 2 is out of order and the motor 1 is not rotating, the induced voltage detector 35 cannot detect the induced voltage. In this case, the switching unit 26 may be switched to the PWM control unit 21 side, and the basic target current value calculation unit 15 may set the basic target current value I ^* = 0. When the steering wheel is operated while the basic target current value I ^* is maintained at zero, an induced voltage is generated as a result of the motor 1 being rotated by an external force. Therefore, the switching unit 26 may be switched to the energization pattern driving unit 24 side in response to detection of an induced voltage of a predetermined value or more by the induced voltage detection unit 35. Thereby, after that, the motor 1 can be controlled by the PWM control signal generated by the energization pattern driving unit 24.

FIG. 7 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to another reference embodiment is applied. In FIG. 7, portions corresponding to the respective portions shown in FIG. 1 are given the same reference numerals.
10 A of motor control apparatuses are the microcomputer 12A, the drive circuit (inverter circuit) 13 which is controlled by this microcomputer 12A, and supplies electric power to the motor 1, and the electric current which flows into the stator windings 51-53 of each phase of the motor 1 And a current detection unit 11 for detecting.

  The microcomputer 12A includes a CPU and a memory (such as a ROM and a RAM), and functions as a plurality of function processing units by executing a predetermined program. The plurality of function processing units include a basic target current value calculation unit 15, a dq-axis target current value calculation unit 16, a PI (proportional integration) control unit 19A, a voltage command value generation unit 19B, and γδ / αβ coordinates. Conversion unit 20A, αβ / UVW coordinate conversion unit 20B, PWM control unit 21, UVW / αβ coordinate conversion unit 17A, αβ / γδ coordinate conversion unit 17B, deviation calculation unit 18, position calculation unit 22, A rotation speed calculation unit 23, a sensor failure determination unit 25, a restriction unit 33, a position estimation unit 60, a sensing signal generation unit 65, and a switching unit 66 are provided.

The basic target current value calculation unit 15 calculates the basic target current value I ^* of the motor 1 based on the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8. For example, the basic target current value I ^* is determined so as to increase as the operating torque increases and to increase as the vehicle speed decreases.
The limiter 33 limits the basic target current value I ^* when the sensor failure determination unit 25 determines that the resolver 2 has failed.

Based on the basic target current value I ^* , the dq-axis target current value calculation unit 16 applies the d-axis current component target value (d-axis target current value I _d ^* ) along the rotor magnetic pole direction of the motor 1 to the d-axis. A target value (q-axis target current value I _q ^* ) of the orthogonal q-axis current component is generated. Hereinafter, these are collectively referred to as “target current value I _dq ”.
When the basic target current value I ^* representing the amplitude of the current (sine wave current) to be applied to the U phase, V phase and W phase of the motor 1 is used, the d axis target current value I _d ^* and the q axis target current value I _q ^* Is expressed by the following equations (1) and (2).

したがって、ｄｑ軸目標電流値演算部１６は、ｄ軸目標電流値Ｉ_d ^*＝０を生成する一方で、トルクセンサ７によって検出される操作トルクに応じたｑ軸目標電流値Ｉ_q ^*を生成する。

Therefore, the dq-axis target current value calculation unit 16 generates the d-axis target current value I _d ^* = 0, while generating the q-axis target current value I _q ^* corresponding to the operation torque detected by the torque sensor 7. To do.

The current detector 11 detects the U-phase current I _u , the V-phase current I _v and the W-phase current I _w of the motor 1 (hereinafter referred to as “three-phase detection current I _uvw ” when collectively referred to). The detected value is given to the UVW / αβ coordinate converter 17A.
The UVW / αβ coordinate converter 17A converts the three-phase detection current I _uvw into the currents I _α and I _β on the two-phase fixed coordinate system (α-β) (hereinafter referred to as “two-phase detection current I”). _The coordinates are converted to “ _αβ ”. The two-phase fixed coordinate system (α-β) is a fixed coordinate system in which the rotation center of the rotor 50 is the origin and the α axis and the β axis perpendicular to the rotation axis of the rotor 50 are defined (see FIG. 2). ). The coordinate-converted two-phase detection current I _αβ is given to the αβ / γδ coordinate conversion unit 17B.

The αβ / γδ coordinate converter 17B converts the two-phase detection current I _αβ on the two-phase rotational coordinate system (γ-δ) according to the rotor rotation position θ ^ (hereinafter referred to as “control rotation position θ ^”) in control. The coordinates are converted into currents I _γ and I _δ (hereinafter referred to as “two-phase detection current I _γδ ”). The two-phase rotational coordinate system (γ−δ) is a rotational coordinate system defined by a γ axis along the rotor magnetic pole direction and a δ axis orthogonal to the γ axis when the rotor 50 is at the control rotational position θ ^. It is. When the control rotational position θ ^ has no error and coincides with the actual rotor rotational position, the two-phase rotational coordinate system (dq) and the two-phase rotational coordinate system (γ-δ) coincide. The control rotation position θ ^ is a rotor rotation position calculated by the position calculation unit 22 or the position estimation unit 60 and selected by the switching unit 66.

The two-phase detection current I _γδ is supplied to the deviation calculation unit 18. The deviation calculator 18 calculates a deviation of the γ-axis current I _{γ with} respect to the d-axis target current value I _d ^* and a deviation of the δ-axis current I _{δ with} respect to the q-axis target current value I _q ^* . These deviations are given to the PI control unit 19A and undergo PI processing. Then, according to these calculation results, the voltage command value generation unit 19B causes the γ-axis voltage command value V _γ ^* and δ-axis voltage command value V _δ ^* (hereinafter referred to as “two-phase voltage command value V _{the ??} "hereinafter.) is produced, provided to the ?? / .alpha..beta coordinate conversion unit 20A.

The γδ / αβ coordinate conversion unit 20A converts the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^* into an α-axis voltage command value V that is a voltage command value of the two-phase fixed coordinate system (α-β). Coordinates are converted to _α ^* and β-axis voltage command value V _β ^* (hereinafter, collectively referred to as “two-phase voltage command value V _αβ ”). The two-phase voltage command value V _αβ is given to the αβ / UVW coordinate conversion unit 20B.

The αβ / UVW coordinate conversion unit 20B converts the α-axis voltage command value V _α ^* and the β-axis voltage command value V _β ^* into voltage command values of a three-phase fixed coordinate system, that is, voltage commands for the U phase, V phase, and W phase. The values are converted into values V _u ^* , V _v ^* , V _w ^* (hereinafter, collectively referred to as “three-phase voltage command value V _uvw ”).
The PWM control unit 21 generates a drive signal having a duty ratio controlled according to the three-phase voltage command values V _u ^* , V _v ^* , and V _w ^* and supplies the drive signal to the drive circuit 13. As a result, a voltage is applied to each phase of the motor 1 with a duty ratio corresponding to the voltage command values V _u ^* , V _v ^* , V _w ^* of the corresponding phase.

With such a configuration, when an operation torque is applied to a steering wheel (not shown) as an operation member coupled to the steering mechanism 3, this is detected by the torque sensor 7. Then, a target current value I _dq corresponding to the detected operation torque and vehicle speed is generated by the dq axis target current value calculation unit 16. A deviation between the target current value I _dq and the two-phase detection current I _γδ is obtained by the deviation calculation unit 18, and PI calculation is performed by the PI control unit 19 A so as to lead this deviation to zero. A two-phase voltage command value V _γδ corresponding to the calculation result is generated by the voltage command value generation unit 19B, and this is converted into a three-phase voltage command value V _uvw through the coordinate conversion units 20A and 20B. The drive circuit 13 operates with a duty ratio corresponding to the three-phase voltage command value V _uvw by the action of the PWM control unit 21, so that the motor 1 is driven and an assist torque corresponding to the target current value I _dq is generated. The steering mechanism 3 will be given. Thus, steering assistance can be performed according to the operation torque and the vehicle speed. The three-phase detection current I _uvw detected by the current detection unit 11 passes through the coordinate conversion units 17A and 17B, and is expressed by a two-phase rotational coordinate system (γ−δ) so as to correspond to the target current value I _dq. After being converted to the phase detection current I _γδ , it is given to the deviation calculating section 18.

  In order to perform coordinate conversion between the rotating coordinate system and the fixed coordinate system, the rotational position (phase angle, ie, electrical angle) θ of the rotor 50 is required. The control rotational position θ ^ representing this rotational position is generated by the position calculation unit 22 using the output of the resolver 2 or estimated by the estimation calculation by the position estimation unit 60. Then, the control rotation position θ ^ calculated by any of the above is given from the switching unit 66 to the αβ / γδ coordinate conversion unit 17B and the γδ / αβ coordinate conversion unit 20A.

The rotational speed calculation unit 23 calculates the rotational speed ω ^ (= Δθ ^) of the rotor 50 by obtaining the difference Δθ ^ of the control rotational position θ ^ given every predetermined control period from the switching unit 26. This rotational speed ω is used in the restriction unit 33 and the position estimation unit 60.
The sensing signal generator 65 applies a search signal (sensing signal) to the motor 1 in order to estimate the rotational position θ of the rotor 50 when the rotor 50 is stopped and when rotating at a very low speed (250 rpm or less). Function as. The sensing signal generator 65 uses a high frequency sine voltage (see FIG. 8B) having a sufficiently high frequency (for example, 200 Hz) as compared with the rated frequency of the motor 1 as a sensing signal, Voltage command values to be applied to the V-phase and W-phase stator windings 51, 52, 53 are generated and provided to the PWM control unit 21. More specifically, V-W phase energization, W-U phase energization, and U-V phase energization are sequentially repeated by applying a high-frequency voltage having a duty ratio that does not cause the rotor 50 to rotate. A high frequency voltage vector that spatially rotates around the rotation center of the rotor 50 is applied. This high-frequency voltage vector is a voltage vector having a constant magnitude that rotates at a constant speed around the origin of αβ coordinates, which are fixed coordinates with the rotation center of the rotor 50 as the origin (see FIG. 2A).

The sensing signal generation unit 65 generates a command value for applying a high-frequency voltage as described above and applies it to the PWM control unit 21 when the rotor 50 is stopped and when the rotor 50 is rotating at an extremely low speed. When the rotation of the rotor 50 becomes sufficiently fast (for example, exceeding 250 rpm), the sensing signal generator 65 stops generating the high-frequency voltage command.
FIG. 9 is a block diagram for explaining the configuration of the position estimation unit 60. The position estimation unit 60 estimates the rotational position of the rotor 50 based on the motor current flowing through the motor 1 and the motor voltage applied to the motor 1. The position estimation unit 60 includes a low speed region position estimation unit 61, a high speed region position estimation unit 62, and an estimated position switching unit 63.

The low speed range position estimation unit 61 is designed to be adapted to position estimation of the rotor 50 when the motor 1 is stopped and when rotating at a very low speed (for example, 0 to 100 rpm). The UVW / αβ coordinate conversion unit 17A The rotational position of the rotor 50 is estimated based on the output two-phase detection current I _αβ and the two-phase voltage command value V _αβ generated by the γδ / αβ coordinate conversion unit 20A. Hereinafter, the rotor rotational position estimated by the low speed region position estimating unit 61 is referred to as “low speed estimated rotational position θ _L ”.

The high-speed range position estimation unit 62 is designed to match the position estimation of the rotor 50 during high-speed rotation of the motor 1 (for example, 200 rpm or more), and the two-phase detection output from the UVW / αβ coordinate conversion unit 17A. The rotational position of the rotor 50 is estimated based on the current I _αβ and the two-phase voltage command value V _αβ generated by the γδ / αβ coordinate conversion unit 20A. Hereinafter, the rotor rotational position estimated by the high speed region position estimating unit 62 is referred to as a “high speed estimated rotational position θ _H ”.

The estimated position switching unit 63 selects either the low speed estimated rotational position θ _L or the high speed estimated rotational position θ _H based on the rotational speed ω obtained by the rotational speed calculating unit 23, and outputs it as the control rotational position θ ^. . The position estimator 60 uses the low speed estimated rotational position θ _L obtained by the low speed region position estimator 61 and the high speed estimated rotational position θ _H obtained by the high speed region position estimator 62 as the rotor rotational speed ω. Accordingly, an internal division processing unit that internally divides and calculates the estimated internal rotation position θ _M may be further provided. In this case, the estimated position switching unit 63 selects one of the low speed estimated rotational position θ _L , the high speed estimated rotational position θ _H, and the internal component estimated rotational position θ _M based on the rotational speed ω. Output.

FIG. 10 is a block diagram for explaining a configuration example of the low-speed range position estimation unit 61. The low speed range position estimation unit 61 includes a high frequency response extraction unit 68 and a rotor position estimation unit 69. The high-frequency response extraction unit 68 is supplied with the two-phase detection current I _αβ output from the UVW / αβ coordinate conversion unit 17A. The high-frequency response extraction unit 68 is, for example, a high-pass filter, and extracts a frequency component corresponding to the frequency of the sensing signal (high-frequency voltage) generated by the sensing signal generation unit 65 from the output signal of the UVW / αβ coordinate conversion unit 17A. Execute the process. The rotor position estimation unit 69 estimates the rotor rotation position based on the high frequency component extracted by the high frequency response extraction unit 68 and the two-phase voltage command value V _αβ .

  Specifically, by applying a sensing signal from the sensing signal generator 65, as shown in FIG. 2 (a), a high-frequency voltage vector (magnitude) that rotates along the rotation direction of the rotor 50 around the origin of the αβ coordinate. Is constant). The high-frequency voltage vector is a voltage vector that rotates at a sufficiently high speed with respect to the rotational speed of the rotor 50. Along with the application of the high-frequency voltage vector, current flows through the U, V, and W-phase stator windings 51, 52, and 53. A current vector representing the magnitude and direction of the three-phase current on the αβ coordinate rotates around the origin.

The inductance of the rotor 50 takes different values in the d-axis direction and the q-axis direction. For this reason, the magnitude of the current vector is large in the direction close to the d-axis, and is small in the direction close to the q-axis. As a result, as shown in FIG. 2B, the end point of the current vector draws an elliptical locus 54 having the major axis in the d-axis direction of the rotor 50 on the αβ coordinates.
Therefore, the magnitude of the current vector has a maximum value in the N-pole direction and the S-pole direction of the rotor 50. That is, the magnitude of the voltage vector changes as shown in FIG. 8 (b), whereas the magnitude of the current vector, as shown in FIG. 8 (a), has two maximum values during one period. Have In this case, if the magnitude of the voltage vector is sufficiently large, the inductance of the N pole side of the rotor 50 becomes smaller than that of the S pole side due to the influence of the magnetic saturation of the stator, and the magnitude of the current vector in the N pole direction. Takes the maximum value (see curve L1).

Therefore, a sufficiently large high-frequency voltage vector is applied to specify the maximum of the current vector corresponding to the N pole, and thereafter, a high-frequency voltage vector having a reduced size is applied, based on the maximum value of the current vector. The phase of the rotor 50 can be estimated. More specifically, the low-speed estimated rotational position θ _L is expressed as θ by the α-axis component I _α and β-axis component I _β (that is, the two-phase detection current I _αβ ) of the current vector when the magnitude is maximum. = Tan ^-1 (I _β / I _α ).

However, in this reference form, the low-speed estimated rotational position θ _L is determined according to the following equation (3) using the two-phase voltage command values V _α ^* and V _β ^* when the magnitude of the current vector takes a maximum value. It has come to be required. This is because the high-frequency voltage vector has a constant magnitude, so that the calculation is easy and the calculation result is accurate.
θ _L = Tan ⁻¹ (V _β ^* / V _α ^* ) (3)
FIG. 11 is a block diagram illustrating a configuration example of the high speed region position estimation unit 62. The high speed range position estimation unit 62 includes a signal processing unit 70 and a rotor position estimation unit 73.

The signal processing unit 70 includes a voltage filter 71 configured by a low-pass filter that removes high-frequency components of the two-phase voltage command value V _αβ and a low-pass filter that removes high-frequency components of the two-phase detection current I _αβ. Current filter 72.
The rotor position estimation unit 73 is provided with the two-phase voltage command value V _αβ and the two-phase detection current I _αβ after the signal processing (filtering) by the signal processing unit 70. The rotor position estimation unit 73 removes a high-frequency component from the disturbance observer 75 that estimates the induced voltage of the motor 1 as a disturbance and the estimated induced voltage output by the disturbance observer 75 based on a motor model that is a mathematical model of the motor 1. Estimated position generation for generating a high-speed estimated rotational position θ _H of the rotor 50 based on an estimated value filter 76 constituted by a low-pass filter and an estimated induced voltage (value after filtering) output from the estimated value filter 76 Part 77. Then, the two-phase voltage command value V _αβ filtered by the voltage filter 71 of the signal processing unit 70 and the two-phase detection current I _αβ filtered by the current filter 72 are respectively transmitted to the disturbance observer 75 of the rotor position estimation unit 73. It is designed to be entered.

FIG. 12 is a block diagram for explaining an example of the disturbance observer 75 and a configuration related thereto. A motor model that is a mathematical model of the motor ¹ can be expressed as, for example, (R + pL) ⁻¹ . Here, R is an armature winding resistance, L is an αβ axis inductance, and p is a differential operator. It can be considered that the two-phase voltage command value V _αβ and the induced voltage E _αβ (α-axis induced voltage E _α and β-axis induced voltage E _β ) are applied to the motor 1.

The disturbance observer 75 receives a two-phase detection current I _αβ as an input and estimates a motor voltage, which is an inverse motor model (inverse model of the motor model) 78, a motor voltage estimated by the inverse motor model 78, and a two-phase voltage command value V A voltage deviation calculation unit 79 for _obtaining a deviation from _αβ can be used. The voltage deviation calculation unit 79 obtains a disturbance with respect to the two-phase voltage command value V _αβ , and as is apparent from FIG. 12, this disturbance is an estimated value E ^ _αβ (α-axis induced voltage corresponding to the induced voltage E _αβ. The estimated value E ^ _α and the β-axis induced voltage estimated value E ^ _β (hereinafter collectively referred to as “estimated induced voltage E ^ _αβ ”) The reverse motor model 78 is represented by, for example, R + pL.

Thus, in this reference embodiment, the disturbance observer 75 is configured to obtain the estimated induced voltage E ^ _αβ using the voltage command value V _αβ and the detected current I _αβ of the two-phase fixed coordinate system. The induced voltage can be estimated without being affected by the speed. Thereby, it can contribute to the improvement of the rotor rotational position estimation accuracy of the motor 1 used in the electric power steering apparatus in which the rotational speed fluctuation is likely to occur.

The estimated value filter 76 can be constituted by, for example, a low-pass filter represented by a / (s + a). a is a design parameter, and the cut-off frequency ω _c of the estimated value filter 76 is determined by the design parameter a.
The induced voltage E _αβ can be expressed by the following equation (4). However, K _E is an induced voltage constant, θ is a rotor rotational position, and ω is a rotor rotational speed.

したがって、推定誘起電圧Ｅ^_αβが求まれば、次の(5)式に従って、高速推定回転位置θ_Hが求まる。この演算が、推定位置生成部７７によって行われるようになっている。

Therefore, if the estimated induced voltage E ^ _.alpha..beta is determined, according to the following equation (5), a high speed estimated rotational position theta _H is obtained. This calculation is performed by the estimated position generation unit 77.

低速域用位置推定部６１による位置推定のためには、モータ電流の応答からセンシング信号成分が抽出される必要がある。しかし、モータ電流には、トルクを発生させるための電流成分（トルク電流成分）が含まれているから、トルク電流成分が大きいときには、センシング信号成分の抽出効率が悪くなる。その結果、位置推定精度が悪くなる。

In order to estimate the position by the low speed range position estimation unit 61, it is necessary to extract a sensing signal component from the response of the motor current. However, since the motor current includes a current component (torque current component) for generating torque, when the torque current component is large, the extraction efficiency of the sensing signal component is deteriorated. As a result, the position estimation accuracy deteriorates.

Therefore, in this reference embodiment, when a failure occurs in the resolver 2 and the rotor position estimation is performed by the position estimation unit 60, the target current value is set in the low rotation speed region where the position estimation by the low speed region position estimation unit 61 is performed. I try to limit it.
Specifically, when it is determined by the sensor failure determination unit 25 that the resolver 2 has failed, the upper limit value of the target current value (hereinafter referred to as “target current upper limit value”) is acted by the function of the limiting unit 33 as shown in FIG. As shown in In the low speed region where the rotational speed ω of the motor 1 is equal to or lower than the predetermined threshold ω1, the target current upper limit value is limited to the first upper limit value I1. In a high-speed region that is greater than or equal to the threshold value ω2 that is greater than the threshold value ω1, the target current upper limit value is a second upper limit value I2 that is greater than the first upper limit value I1. However, in the rotational speed region of the predetermined speed ω3 that is greater than the threshold ω2 and higher than the threshold speed ω2, the motor current decreases due to the counter electromotive force of the motor 1, and accordingly, the characteristic that linearly decreases as the rotational speed ω increases. The basic target current value I ^* is set in the basic target current value calculation unit 15 according to this characteristic.

In the transition region between the threshold values ω1 and ω2, the target current upper limit value is determined so as to increase linearly from the first upper limit value I1 to the second upper limit value I2 as the rotational speed ω increases. The limiting unit 33 determines a target current upper limit value lower than that in the normal state (see the broken line in FIG. 13) in the rotation speed region below the threshold ω2.
For example, the estimation of the rotor position in the low speed region below the threshold value ω1 is performed by the low speed region position estimation unit 61, and the estimation of the rotor position in the speed region (medium / high speed region) exceeding the threshold value ω2 is performed. 62.

Accordingly, when position estimation is performed by the low speed range position estimation unit 61, the target current upper limit value is set to the first upper limit value I1. As a result, it is possible to maintain a state in which the sensing signal component can be efficiently extracted from the response of the motor current, and therefore it is possible to ensure position estimation accuracy in the low speed region.
At normal times when the resolver 2 has not failed, the target current upper limit value in the region below the threshold ω2 is set to the second upper limit value I2 as indicated by a broken line in FIG. That is, the restriction by the restriction unit 33 is not performed, and the basic target current value calculation unit 15 determines the basic target current value I ^* with the second upper limit value I2 as an upper limit.

Note that the limitation of the target current upper limit value by the limiting unit 33 may be performed according to the characteristics indicated by the two-dot chain line L11 in FIG. That is, the target current upper limit value may be limited to the first upper limit value I1 or less at any rotational speed.
FIG. 14 is a flowchart for explaining processing that the microcomputer 12A repeatedly executes at predetermined control cycles. The microcomputer 12A takes in the output signals of the torque sensor 7, the vehicle speed sensor 8, the resolver 2, and the current detection unit 11 (step S11). The basic target current value calculation unit 15 calculates the basic target current value I ^* based on the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8 (step S12). On the other hand, the sensor failure determination unit 25 determines the presence or absence of a failure of the resolver 2 based on the signal derived to the signal line 2a (step S13).

If there is no failure in the resolver 2 (step S13: NO), the switching unit 66 selects the control rotation position θ ^ (control rotation position based on the resolver output) calculated by the position calculation unit 22 (step S14). The normal control for driving the motor 1 is executed based on the basic target current value I ^* using the output signal (step S15). More specifically, the d-axis target current value I _d ^* and the q-axis target current value I _q ^* are set by the dq-axis target current value calculation unit 16. Further, the phase current I _uvw detected by the current detection unit 11 is _subjected to coordinate conversion by the coordinate conversion units 17A and 17B, and the γ-axis current I _γ and the δ-axis current I _δ are obtained. The deviation calculation unit 18 calculates a d-axis current deviation δI _d (= I _d ^* −I _γ ) and a q-axis current deviation δI _q (= I _q ^* −I _δ ). The PI control unit 19A performs a PI (proportional integration) calculation or the like on the current deviations δI _d and δI _q , and based on this PI calculation, the voltage command value generation unit 19B performs the d-axis voltage command value V _d ^* and the q-axis. A voltage command value V _q ^* is generated. These are subjected to coordinate conversion by the coordinate conversion units 20A and 20B, thereby generating UVW phase voltage command values V _u ^* , V _v ^* and V _w ^* . A PWM control signal corresponding to these voltage command values V _u ^* , V _v ^* , V _w ^* is generated by the PWM control unit 21. For the coordinate conversion calculation in the αβ / γδ coordinate conversion unit 17B and the γδ / αβ coordinate conversion unit 20A, the control rotational position θ ^ calculated by the position calculation unit 22 based on the output signal of the resolver 2 is used (step S14). .

On the other hand, when the sensor failure determination unit 25 determines that the resolver 2 has a failure (step S13: YES), the switching unit 66 calculates the control rotation position θ ^ (estimated rotation position) calculated by the position estimation unit 60. ) Is selected (step S16), and restriction processing by the restriction unit 33 is further performed (steps S17 and S18). That is, it is determined whether or not the basic target current value I ^* calculated by the basic target current value calculation unit 15 is equal to or greater than the target current upper limit value (see FIG. 13, a value corresponding to the rotational speed ω) (step S17). If the basic target current value I ^* is less than the target current upper limit value (step S17: NO), the basic target current value I ^* is used as it is. That is, the motor 1 is controlled based on the basic target current value I ^* while using the control rotation position θ ^ calculated by the position estimation unit 60 (step S15). On the other hand, when the basic target current value I ^* is greater than or equal to the target current upper limit value (step S17: YES), the target current upper limit value is substituted into the basic target current value I ^* by the action of the limiting unit 33 ( Step S18). Therefore, based on the target current upper limit value, the motor 1 is controlled using the control rotation position θ ^ calculated by the position estimation unit 60.

The estimated position switching unit 63 of the position estimating unit 60 determines that the estimated position switching unit 63 estimates the low speed estimated by the low speed range position estimating unit 61 when the motor rotational speed ω obtained by the rotational speed calculating unit 23 is equal to or less than the threshold ω1. The rotational position θ _L is output. When the motor rotation speed exceeds the threshold value ω1, the estimated position switching unit 63 outputs the high-speed estimated rotation position θ _H estimated by the high-speed range position estimating unit 62.

As described above, in this reference embodiment, when a failure occurs in the resolver 2, sensorless control using the estimated rotational position obtained by the position estimating unit 60 as the control rotational position θ ^ is performed without using the output signal of the resolver 2. Switch to motor control. Thereby, even after the resolver 2 breaks down, the drive of the motor 1 can be continued and the steering assist force can be applied to the steering mechanism 3. In addition, when sensorless control is performed, the target current upper limit value is limited to a value I1 that provides a certain or higher rotor position estimation accuracy in the low speed range where the rotor rotational position is estimated by the low speed range position estimation unit 61. It has come to be. Thereby, since the rotor rotational position can be accurately estimated even in the low speed region, the motor 1 can be smoothly rotated. Thereby, a favorable steering feeling can be realized.

  A certain amount of time is required for the estimation of the rotor rotational position by the low-speed range position estimation unit 61. That is, for sensorless control using the low-speed range position estimation unit 61, a sensing signal is generated from the sensing signal generation unit 65, a response of the motor current to the detection signal is detected, and the rotor rotational position is calculated based thereon. A series of processing is required. Because of these processes, a position indefinite period occurs at the beginning of control. This problem is that the control rotational position θ ^ obtained from the output signal of the resolver 2 immediately before that is used as the initial value for the sensorless control in the period immediately after the failure of the resolver 2 is detected and shifts to the sensorless control. Can be solved.

  Since the induced voltage of the motor 1 is used in the position estimation process by the high speed range position estimation unit 62, the rotor rotational position can be estimated quickly. Therefore, only when the rotor rotational speed ω immediately after the shift to the sensorless control is equal to or less than the threshold ω1 (when the calculation by the low-speed range position estimation unit 61 is performed), based on the output of the resolver 2 (just before the failure determination). The control rotation position θ ^ calculated in () may be used as an initial value for sensorless control.

FIG. 15 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to still another reference embodiment is applied. In FIG. 15, parts corresponding to those shown in FIG. 7 are given the same reference numerals.
In this reference embodiment, the position estimation unit 60A is not provided with the low speed region position estimation unit, and is provided with only the high speed region position estimation unit 62 that estimates the rotor rotational position based on the induced voltage. Further, a control law determiner 80 is provided for determining a control mode when the resolver 2 fails.

When the rotor is stopped or the rotor rotational speed is low, the induced voltage is small, so the rotor position cannot be estimated. Therefore, in this reference embodiment, when the induced voltage is small, the target current value is set to zero, and when the induced voltage is generated from the motor 1 by the rotation of the steering wheel, the rotor rotational position is estimated based on the induced voltage. ing.

FIG. 16 is a flowchart for explaining the operation of the microcomputer 12A. In FIG. 16, steps in which processing similar to the steps shown in FIG. 14 is performed are denoted by the same reference numerals as those in FIG. 14.
The microcomputer 12A takes in the output signals of the torque sensor 7, the vehicle speed sensor 8, the resolver 2, and the current detection unit 11 (step S11). The basic target current value calculation unit 15 calculates the basic target current value I ^* based on the operation torque detected by the torque sensor 7 and the vehicle speed detected by the vehicle speed sensor 8 (step S12). On the other hand, the sensor failure determination unit 25 determines the presence or absence of a failure of the resolver 2 based on the signal derived to the signal line 2a (step S13).

At normal time when the resolver 2 has no failure (step S13: NO), the control rotational position θ ^ calculated by the position calculation unit 22 using the output signal of the resolver 2 is selected by the switching unit 66 (step S14). Then, the control of the motor 1 is executed based on the basic target current value I ^* using the control rotational position θ ^ (step S15).
On the other hand, when the sensor failure determination unit 25 determines that a failure has occurred in the resolver 2 (step S13: YES), the switching unit 66 determines the control rotational position θ ^ ( (Estimated rotational position) is selected (step S16).

Further, the control law determiner 80 sets the basic target current value I ^* generated from the basic target current value calculation unit 15 to zero (step S20). As a result, the voltage command values V _u ^* , V _v ^* , and V _w ^* are equal to the induced voltage of the motor 1. Therefore, the control law determiner 80 obtains the induced voltage of the motor 1 from the voltage command values V _u ^* , V _v ^* , V _w ^* (induced voltage) (step S21). Then, the control law determiner 80 compares the obtained induced voltage with a predetermined threshold value V _th (step S22). This threshold value V _th is a value close to the lower limit value at which the induced voltage can be estimated by the disturbance observer 75 (see FIG. 11).

When the induced voltage is larger than the threshold value V _th (step S22: YES), the control law determiner 80 causes the high-speed region position estimation unit 62 to estimate the rotor rotational position (step S23), and the basic target current value. The calculation unit 15 is caused to calculate a basic target current value I ^* corresponding to the operation torque and the vehicle speed (step S24). This calculation may be omitted, and the basic target current value I ^* obtained in step S12 may be used as it is. Based on the basic target current value I ^* thus obtained, the motor 1 is controlled using the control rotational position θ ^ calculated by the high speed range position estimation unit 62 (step S15).

When the induced voltage is less than or equal to the threshold V _th (step S22: NO), the rotor rotational position estimation calculation by the high speed region position estimating unit 62 is not performed, and the basic target current value I ^* is maintained at zero. That is, the motor 1 is driven with the basic target current value I ^{* set} to zero (step S15). Thereby, it can avoid that the motor 1 becomes steering resistance.
When the steering wheel is operated while the basic target current value I ^* is maintained at zero, an induced voltage is generated as a result of the motor 1 being rotated by an external force. When the induced voltage exceeds the threshold value V _th (step S22: YES), the high speed region position estimation unit 62 calculates the control rotation position θ ^ (step S23), and based on the calculated control rotation position θ ^. The motor 1 is driven (step S15). At this time, the basic target current value I ^* is set to a significant value corresponding to the operation torque and the vehicle speed (step S24).

Thus, according to this reference embodiment, even after the resolver 2 has failed, to the extent that the induced voltage of the position estimation can be sized according to the high speed region for the position estimator 62 is caused to drive the motor 1 And a steering assist force can be applied to the steering mechanism 3.
Having described implementation form and reference embodiment of the present invention, it is also possible implementation in other forms further. For example, in the reference shape state of FIG. 7 described above, but so as to limit the basic target current value, the same effect can be obtained so as to limit the actual current supplied to the motor 1.

Further, in the reference form of FIG. 7 described above, the sensing signal is superimposed on the three-phase voltage command value V _UVW , but the voltage command value V _γδ or γδ / αβ coordinate conversion unit 20A generated by the voltage command value generation unit 19B. The sensing signal may be superposed on the two-phase voltage command value V _αβ generated by.
Furthermore, in the embodiment and reference embodiment described above has described the case that control the motor 1 as a drive source of an electric power steering apparatus, reference embodiment described above, the application of the motor control other than the electric power steering device It can also be applied to.

  In addition, various design changes can be made within the scope of matters described in the claims.

1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control device according to an embodiment of the present invention is applied. FIG. It is a figure for demonstrating the structure of a motor, and rotation of a high frequency voltage vector and an electric current vector. It is an electric circuit diagram for demonstrating the structure of a drive circuit. 4A and 4B are waveform diagrams illustrating an example of a control signal generated by the energization pattern driving unit. It is a flowchart for explaining the operation before you facilities embodiment. It is a block diagram which shows the electric constitution of the electric power steering apparatus which concerns on a reference form. It is a block diagram for demonstrating the electric constitution of the electric power steering apparatus to which the motor control apparatus which concerns on another reference form is applied. It is a figure for demonstrating the rotor phase angle estimation operation | movement by application of a high frequency voltage vector. It is a block diagram for demonstrating the structure of a position estimation part. It is a block diagram for demonstrating the structural example of the position estimation part for low speed areas. It is a block diagram which shows the structural example of the position estimation part for high speed areas. It is a block diagram for demonstrating an example of a disturbance observer and the structure relevant to this. It is a characteristic view for demonstrating the restriction | limiting of a basic target electric current value. It is a flowchart for demonstrating operation | movement of the reference form of the said FIG . It is a block diagram for demonstrating the electrical structure of the electric power steering apparatus to which the motor control apparatus which concerns on another reference form is applied. 16 is a flowchart for explaining the operation of the reference embodiment of FIG.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Motor, 2 ... Resolver 10, 10A ... Motor controller, 12, 12A ... Microcomputer, 50 ... Rotor, 55 ... Stator

Claims (1)

A motor control device for controlling a motor including a rotor and a stator facing the rotor,
A rotation sensor for detecting a rotation position of the rotor;
When the rotation sensor is normal, the motor is controlled based on the output signal of the rotation sensor. On the other hand, when the rotation sensor is out of order, the control mode is limited to the normal control mode. Control means for controlling the motor without using an output signal,
The motor is a steering motor that applies a steering force to a steering mechanism of a vehicle;
The control means drives the motor in a predetermined energization pattern on condition that an operation torque applied to an operation member for steering the vehicle is equal to or less than a predetermined torque threshold when the rotation sensor fails. der is, that have been established to a larger value as possible within a range with no possibility of step-out occurs when the torque threshold is to drive the motor at the predetermined energization pattern, the motor control device.

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