JP2009261102A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor controller for improving the reliability of motor control by verifying the reliability of a rotor rotation position estimated using a sensing signal. <P>SOLUTION: A sensing signal generating section 21 generates a sensing signal of a predetermined sensing frequency. The sensing signal is injected into a stator of a motor 3. A response of the motor 3 to the sensing signal is detected by a current sensor 9. A position estimating section 20 obtains an estimated rotation position θ<SP>^</SP>based on the peak of a motor current waveform. A rotation speed calculating section 25 calculates a rotation speed ω<SB>1</SB>of the rotor using the estimated rotation position θ<SP>^</SP>. Meanwhile, a frequency analyzing section 26 analyzes a frequency of a three-phase detection current I<SB>UVW</SB>to obtain a rotation speed ω<SB>2</SB>of the rotor. A reliability verifying section 27 determines that the estimated rotation position θ<SP>^</SP>has no reliability when the difference between the two rotating speeds ω<SB>1</SB>and ω<SB>2</SB>obtained independently from each other exceeds a threshold of speed difference. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor control device for sensorless driving of a brushless motor. The brushless motor is used, for example, as a source for generating a steering assist force in an electric power steering apparatus.

  A motor control device for driving and controlling a brushless motor is generally configured to control the supply of motor current in accordance with the output of a position sensor for detecting the rotational position of the rotor. However, the environmental resistance of the position sensor becomes a problem, and the expensive position sensor and the wiring associated therewith hinder the cost reduction and the size reduction. Therefore, a sensorless driving system that drives a brushless motor without using a position sensor has been proposed. The sensorless driving method is a method for estimating the phase of the magnetic pole (electrical angle of the rotor) by estimating the induced voltage accompanying the rotation of the rotor.

  Since the induced voltage cannot be estimated when the rotor is stopped and when rotating at a very low speed, the phase of the magnetic pole is estimated by another method. Specifically, as shown in FIG. 4A, a high-frequency voltage vector that rotates along the rotation direction of the rotor 50 around the origin of αβ coordinates, which are two-phase fixed coordinates with the rotation center of the rotor 50 as the origin. The size is constant). More specifically, a sensing signal that is a sinusoidal high-frequency exploration voltage is applied to the U, V, and W-phase stator windings 51, 52, and 53 so that such a high-frequency voltage vector is formed. The high-frequency voltage vector is a voltage vector that rotates at a sufficiently high speed with respect to the rotation speed of the rotor 50. With the application of the high-frequency voltage vector, a current flows through the U, V, and W-phase stator windings 51, 52, and 53. A current vector representing the magnitude and direction of the three-phase current on the αβ coordinate rotates around the origin.

  The inductance of the rotor 50 takes different values for the d-axis, which is the magnetic pole axis along the magnetic flux direction, and the q-axis (axis along the torque direction) orthogonal thereto. For this reason, the magnitude of the current vector is large in the direction close to the d-axis, and is small in the direction close to the q-axis. As a result, as shown in FIG. 4B, the end point of the current vector draws an elliptical locus 55 having the major axis in the d-axis direction of the rotor 50 on the αβ coordinates.

  Therefore, the magnitude of the current vector has a maximum value in the N-pole direction and the S-pole direction of the rotor 50. That is, as shown by a curve L2 in FIG. 5A, the magnitude of the current vector has two maximum values in one cycle. In this case, if the magnitude of the voltage vector is sufficiently large, the inductance of the N pole side of the rotor 50 becomes smaller than that of the S pole side due to the influence of the magnetic saturation of the stator, and the magnitude of the current vector in the N pole direction. Takes the maximum value (see curve L1).

Therefore, a sufficiently large high-frequency voltage vector is applied to specify the maximum of the current vector corresponding to the N pole, and thereafter, a high-frequency voltage vector having a reduced size is applied, based on the maximum value of the current vector. The rotational position of the rotor 50 can be estimated. More specifically, the phase angle (electrical angle) θ of the rotor 50 becomes θ = Tan ⁻¹ (I by the α-axis component I _α and the β-axis component I _β of the current vector when the magnitude is maximum. _β / I _α ).
JP 2004-343963 A

  In the position estimation algorithm using the sensing signal as described above, the motor current detected by the current sensor is filtered to extract the current response component corresponding to the sensing signal and to estimate the rotor rotational position. It has become. However, if the surrounding environment such as the motor temperature changes, the motor parameters (resistance and inductance) change, which may make it impossible to extract necessary signals. Thereby, there is a possibility that an accurate rotor rotational position cannot be detected, and there is a concern that the control performance of the motor is deteriorated.

  Accordingly, an object of the present invention is to provide a motor control device capable of improving the reliability of motor control by verifying the reliability of the rotor rotational position estimated using the sensing signal.

  In order to achieve the above object, a first aspect of the present invention provides a motor control device for controlling a motor (3) including a rotor (50) and a stator (51-53) facing the rotor (50). 5) a sensing signal injection means (21) for injecting a sinusoidal sensing signal into the stator, a motor current detection means (9) for detecting the motor current flowing in the motor, and the motor current detection means Position estimation means (20) for estimating the rotor rotational position based on the detected motor current, and verification means (25-27; 40, 41) for verifying the reliability of the rotor rotational position estimated by this position estimation means And a motor control device. The alphanumeric characters in parentheses indicate corresponding components in the embodiments described later. The same applies hereinafter.

According to this configuration, by detecting the motor current response when the sinusoidal sensing signal is injected into the stator by the motor current detection means, the rotor rotational position can be estimated based on the motor current response. The reliability of the estimated rotor rotational position is verified by the verification means. Thereby, the reliability of motor control can be improved.
The invention according to claim 2 further includes motor stop control means (28) for stopping the motor when the verification means determines that the rotor rotational position estimated by the position estimation means is not reliable. A motor control device according to claim 1.

According to this configuration, since the motor is stopped when the rotor rotational position is not reliable, when the calculation result by the position estimation means is inaccurate, improper motor control based on the inaccurate calculation result is performed. There is nothing. Thereby, the reliability of motor control can be further improved.
According to a third aspect of the present invention, the verification means includes first rotor rotation speed calculation means (25) for calculating a rotor rotation speed based on the rotor rotation position estimated by the position estimation means, and the motor current detection means. The second rotor rotation speed calculation means (26) for calculating the rotor rotation speed by analyzing the frequency of the motor current detected by the rotor, the rotor rotation speed obtained by the first rotor rotation speed calculation means, and the second 3. The motor control device according to claim 1, further comprising comparison means (27, S5) for comparing the rotor rotation speed obtained by the rotor rotation speed calculation means.

  According to this configuration, the rotor rotational speed (first rotor rotational speed) is obtained based on the rotor rotational position estimated by the position estimating means. On the other hand, the rotor rotational speed (second rotor rotational speed) is calculated by performing a frequency analysis of the motor current detected by the motor current detecting means. The first rotor rotation speed reflects the calculation result by the position estimation means, while the second rotor rotation speed is obtained independently from the calculation by the position estimation means. Therefore, the first and second rotor rotational speeds are compared. As a result of the comparison, for example, if the difference between the first and second rotor rotational speeds exceeds a predetermined speed difference threshold, it can be determined that the reliability of the rotor rotational position obtained by the position estimating means is low. On the other hand, if the difference between the first and second rotor rotational speeds is equal to or smaller than the speed difference threshold value, it can be determined that the rotor rotational position obtained by the position estimating means has sufficient reliability. In this way, the reliability of the rotor rotational position required by the position estimating means can be verified.

  In addition, as the verification means, second position estimation means (40) for obtaining the rotor rotational position by a method different from the position estimation means (first position estimation means) is provided, and the first and second positions are provided. It is also possible to take a configuration for comparing the calculation results by the estimation means. That is, if the difference between the calculation results by the first and second position estimation means exceeds a predetermined position difference threshold, it can be determined that the estimated position is not reliable, and the difference between the calculation results is the position difference threshold. If it is below, it can be judged that the estimated position is reliable.

For example, the second position estimating means may obtain an induced voltage of the motor from the motor current and the motor voltage, and estimate the rotor rotational position based on the induced voltage.
Furthermore, the second position estimating means may estimate (detect) the rotor rotational position using a simple sensor.

Hereinafter, embodiments of the present invention will be described in detail with reference to the accompanying drawings.
FIG. 1 is a block diagram for explaining an electrical configuration of an electric power steering apparatus to which a motor control apparatus according to an embodiment of the present invention is applied. The electric power steering apparatus includes a torque sensor 1 that detects a steering torque applied to a steering wheel of a vehicle, a motor 3 (electric motor) that applies a steering assist force to a steering mechanism 2 of the vehicle, and drive control of the motor 3. And a motor control device 5. The motor control device 5 drives the motor 3 according to the steering torque detected by the torque sensor 1, thereby realizing appropriate steering assistance according to the steering situation.

  In this embodiment, the motor 3 is a three-phase brushless motor, and, as schematically shown in FIG. 2, the rotor 50 as a field and the U-phase, V-phase, and W-phase disposed in the stator facing it. Phase stator windings 51, 52, 53. The motor 3 may be of an inner rotor type in which a stator is disposed facing the outside of the rotor, or may be of an outer rotor type in which a stator is disposed facing the inside of a cylindrical rotor.

The motor control device 5 detects a current that flows through the microcomputer 7, a drive circuit (inverter circuit) 8 that is controlled by the microcomputer 7 and supplies power to the motor 3, and a stator winding of each phase of the motor 3. And a current sensor 9.
The microcomputer 7 includes a CPU and a memory (such as a ROM and a RAM), and functions as a plurality of function processing units by executing a predetermined program. The plurality of function processing units include a current command value generation unit 11, a PI (proportional integration) control unit 12, an instruction voltage generation unit 13, a γδ / αβ coordinate conversion unit 14, and an αβ / UVW coordinate conversion unit 15. A PWM control unit 16, a UVW / αβ coordinate conversion unit 17, an αβ / γδ coordinate conversion unit 18, a deviation calculation unit 19, a position estimation unit 20, a sensing signal generation unit 21, an addition unit 22, A rotation speed calculation unit 25, a frequency analysis unit 26, a reliability verification unit 27, and a stop control unit 28 are provided.

The current command value generation unit 11 generates a command value I _d ^* of a d-axis current component along the rotor magnetic pole direction of the motor 3 and a command value I _q ^* of a q-axis current component orthogonal to the d axis. Hereinafter, these are collectively referred to as “current command value I _dq ^* ”. However, the dq coordinate plane is a plane along the rotation direction of the rotor 50, and the d axis and the q axis define a two-phase rotation coordinate system (dq) that rotates together with the rotor 50 (see FIG. 2).

When the current command value I ^* representing the amplitude of the current (sine wave current) to be applied to the U phase, V phase and W phase of the motor 3 is used, the d axis current command value I _d ^* and the q axis current command value I _q ^*. Is expressed by the following equations (1) and (2).
I _d ^* = 0 (1)
I _q ^* = − (3/2) ^1/2・ I ^* (2)
Therefore, the current command value generation unit 11 generates the d-axis current command value I _d ^* = 0, while generating the q-axis current command value I _q ^* corresponding to the steering torque detected by the torque sensor 1. More specifically, the q-axis current command value I _q ^* may be generated using a map (table) that stores the q-axis current command value I _q ^* corresponding to the steering torque. Since the torque generated by the motor 3 corresponds to the motor current, the current command value I _dq ^* can be rephrased as a “torque command value” for commanding the torque to be generated from the motor 3.

The current sensor 9 detects the U-phase current I _U , the V-phase current I _V and the W-phase current I _w of the motor 3 (hereinafter referred to as “three-phase detection current I _UVW ” when collectively referred to). The detected value is sampled at a predetermined sampling frequency f _s , taken into the microcomputer 7, and given to the UVW / αβ coordinate converter 17.
The UVW / αβ coordinate converter 17 converts the three-phase detection current I _UVW into the currents I _α and I _β on the two-phase fixed coordinate system (α-β) (hereinafter referred to as “two-phase detection current I _The coordinates are converted to “ _αβ ”. The two-phase fixed coordinate system (α-β) is a fixed coordinate system in which the rotation center of the rotor 50 is the origin and the α axis and the β axis orthogonal to the rotation axis of the rotor 50 are defined (see FIG. 2). ). The coordinate-converted two-phase detection current I _αβ is given to the αβ / γδ coordinate conversion unit 18.

The αβ / γδ coordinate conversion unit 18 converts the two-phase detection current I _αβ into a two-phase rotation coordinate system (hereinafter referred to as “estimated rotation position θ ^”) estimated by the position estimation unit 20. Coordinates are converted into currents I _γ and I _δ on γ−δ) (hereinafter, collectively referred to as “two-phase detection current I _γδ ”). The two-phase rotational coordinate system (γ−δ) is a rotational coordinate system defined by a γ axis along the rotor magnetic pole direction and a δ axis orthogonal to the γ axis when the rotor 50 is at the estimated rotational position θ ^. (See FIG. 2). When the estimated rotational position θ ^ has no error and coincides with the actual rotor rotational position, the two-phase rotational coordinate system (dq) and the two-phase rotational coordinate system (γ-δ) coincide.

The two-phase detection current I _γδ is supplied to the deviation calculation unit 19. The deviation calculator 19 calculates a deviation of the γ-axis current I _{γ with} respect to the d-axis current command value I _d ^* and a deviation of the δ-axis current I _{δ with} respect to the q-axis current command value I _q ^* . These deviations are given to the PI control unit 12 and are each subjected to PI calculation processing. Then, in accordance with these calculation results, the command voltage generation unit 13 causes the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* (hereinafter referred to as “two-phase command voltage V _γδ ^* ” when collectively referred to). .) Is generated and provided to the γδ / αβ coordinate converter 14.

The γδ / αβ coordinate converter 14 converts the γ-axis command voltage V _γ ^* and the δ-axis command voltage V _δ ^* into α-axis command voltages V _α ^* and β that are command voltages of the two-phase fixed coordinate system (α-β). Coordinates are converted to an axis command voltage V _β ^* (hereinafter referred to as “two-phase command voltage V _αβ ^* ” when these are collectively referred to). The two-phase instruction voltage V _αβ ^* is given to the αβ / UVW coordinate conversion unit 15.
The αβ / UVW coordinate converter 15 converts the α-axis command voltage V _α ^* and the β-axis command voltage V _β ^* into the command voltages of the three-phase fixed coordinate system, that is, the command voltages V _U ^{* of the} U phase, V phase, and W phase ^. , V _V ^* , V _W ^* (hereinafter collectively referred to as “three-phase indicating voltage V _UVW ^* ”).

The PWM control unit 16 generates a drive signal having a duty ratio controlled according to the three-phase instruction voltages V _U ^* , V _V ^* , and V _W ^* and supplies the drive signal to the drive circuit 8. As a result, a voltage is applied to each phase of the motor 3 with a duty ratio corresponding to the instruction voltages V _U ^* , V _V ^* , and V _W ^* of the corresponding phase.
With such a configuration, when a steering torque is applied to a steering wheel (not shown) as an operation member coupled to the steering mechanism 2, this is detected by the torque sensor 1. Then, the current command value generation unit 11 generates a current command value I _dq ^* corresponding to the detected steering torque. A deviation between the current command value I _dq ^* and the two-phase detection current I _γδ is obtained by the deviation calculating unit 19, and PI calculation is performed by the PI control unit 12 so as to lead this deviation to zero. A two-phase command voltage V _γδ ^* corresponding to this calculation result is generated by the command voltage generation unit 13, and this is converted into a three-phase command voltage V _UVW ^* via the coordinate conversion units 14 and 15. The drive circuit 8 operates at a duty ratio corresponding to the three-phase instruction voltage V _UVW ^* by the action of the PWM control unit 16, thereby driving the motor 3 and assist torque corresponding to the current command value I _dq ^*. Is given to the steering mechanism 2. Thus, steering assistance can be performed according to the steering torque. The three-phase detection current I _UVW detected by the current sensor 9 passes through the coordinate conversion units 17 and 18 and is expressed in two-phase rotational coordinate system (γ−δ) so as to correspond to the current command value I _dq ^*. After being converted to the phase detection current I _γδ , it is given to the deviation calculating unit 19.

In order to perform coordinate conversion between the rotating coordinate system and the fixed coordinate system, a phase angle (electrical angle) θ representing the rotational position of the rotor 50 is required. An estimated rotational position θ ^ representing this phase angle is generated by the position estimation unit 20 and is provided to the γδ / αβ coordinate conversion unit 14 and the αβ / γδ coordinate conversion unit 18.
The sensing signal generator 21 functions as sensing signal injection means for injecting a sinusoidal sensing signal into the stator windings 51 to 53 in order to estimate the phase angle θ of the rotor 50. The sensing signal generator 21 uses a high-frequency sine voltage (see FIG. 5B) having a sufficiently high sensing frequency f _a (for example, f _a = 400 Hz) as compared with the rated frequency of the motor 3 as a search voltage. Sensing signals to be applied to the U-phase, V-phase and W-phase stator windings 51, 52 and 53 of the motor 3 are generated. More specifically, V-W phase energization, W-U phase energization, and U-V phase energization are sequentially repeated by applying a high-frequency voltage having a duty ratio that does not cause the rotor 50 to rotate. A sensing signal that can apply a high-frequency voltage vector that rotates spatially around the rotation center of the rotor 50 is generated. This high-frequency voltage vector is a voltage vector having a constant magnitude that rotates at a constant speed around the origin of αβ coordinates, which are fixed coordinates with the rotation center of the rotor 50 as the origin (see FIG. 4A).

In this embodiment, the sensing signal generation unit 21 generates a sensing signal to be superimposed on the two-phase instruction voltage V _αβ ^* . This sensing signal is superimposed on the two-phase instruction voltage V _αβ ^* by the adding unit 22.
The position estimation unit 20 estimates the rotational position of the rotor 50 based on the two-phase detection current I _αβ provided from the UVW / αβ coordinate conversion unit 17 at each sampling period Ts (= 1 / f _s ), and the estimated rotational position. Generate θ ^.

Then, the rotation speed calculation unit 25 calculates the rotation speed ω ₁ (angular speed) of the rotor 50 based on the estimated rotation position θ ^ generated by the position estimation unit 20. That is, when the estimated rotational position θ ^ is generated every control period (for example, equal to the sampling period), the rotational speed ω ₁ is calculated based on the difference between the estimated rotational position θ ^ in the previous control period and the current control period. Calculated. The rotational speed ω ₁ is used for position estimation calculation in the position estimation unit 20.

The frequency analysis unit 26 performs frequency analysis of the three-phase detection current I _UVW by performing, for example, fast Fourier transform (FFT) calculation. The frequency spectrum of the three-phase detected current I _UVW, the double frequency 2f _a sensing frequency f _a (e.g., 2f _a = 800Hz) peak appears at the position of. This is because the three-phase detection current I _UVW has two peaks corresponding to the N pole and S pole of the rotor 50 in one cycle. In the frequency spectrum of the three-phase detection current I _UVW , a peak corresponding to a “beat component” generated by superimposing a signal corresponding to the rotation of the rotor 50 on the sensing signal appears. For example, when the rotor 50 is rotating at a rotational speed omega ₂ (angular velocity) in the counterclockwise direction in FIG. 2, the frequency f _b of the peak corresponding to the beat _{components, f b = 2f a -2 (} ω 2 / 2π). Conversely, when the rotor 50 is rotating at a rotational speed omega ₂ in a clockwise direction in FIG. 2, the frequency f _b of the peak corresponding to the beat component becomes _{f b = 2f a +2 (ω} 2 / 2π) . Therefore, the frequency analyzing unit 26, the difference between the frequency f _b of the peak frequency 2f _a shaking becomes component corresponding to the sensing signal δf seeking _{(= | | 2f a -f b} = 2 (ω 2 / 2π)), now A rotational speed ω ₂ (= πδf) is obtained.

The reliability verification unit 27 compares the rotation speed ω ₁ obtained by the rotation speed calculation unit 25 with the rotation speed ω ₂ obtained by the frequency analysis unit 26. More specifically, the difference Δω (= | ω ₁ −ω ₂ |) between these two rotational speeds ω ₁ and ω ₂ is compared with a predetermined speed difference threshold Thω. When the rotational speed difference Δω is equal to or smaller than the speed difference threshold Thω, that is, when the two rotational speeds ω ₁ and ω ₂ are in good agreement, the reliability verification unit 27 estimates the rotor rotational position by the position estimation unit 20. Therefore, it is determined that the estimated rotational position θ ^ is reliable. On the other hand, when the rotational speed difference Δω exceeds the speed difference threshold Thω, that is, when the two rotational speeds ω ₁ and ω ₂ are significantly different, the estimation accuracy of the rotor rotational position by the position estimating unit 20 is high. Therefore, it is determined that the estimated rotational position θ ^ is not reliable.

The stop control unit 28 stops the control of the motor 3 in response to the reliability verifying unit 27 determining that the estimated rotational position θ ^ is not reliable. This prevents the motor 3 from being improperly controlled based on the inaccurate estimated rotational position θ ^.
FIG. 3 is a block diagram for explaining the configuration of the position estimation unit 20. The position estimation unit 20 includes a current value calculation unit 31, an inverse notch filter 32, a current peak timing extraction unit 33, a current synchronization extraction unit 34, and a rotor position calculation unit 35.

The current value calculation unit 31 obtains the magnitude of the current vector, that is, the magnitude I of the current. More specifically, the current value calculation unit 31 calculates the current magnitude I based on the two-phase detection current I _αβ provided from the UVW / αβ coordinate conversion unit 17. For example, the current magnitude I is calculated according to the following equation (3).
I = {I _α ² + I _β ² } ^1/2 (3)
The inverse notch filter 32 removes a noise component from the output of the current value calculation unit 31. The current magnitude I has two local maxima during one period of the high-frequency voltage vector. Therefore, the frequency of the significant signal component in the output of the current value calculation unit 31 has a frequency about twice the frequency f _a of the sensing signal generated by the sensing signal generation unit 21. Therefore, inverse notch filter 32, a frequency twice the frequency f _a of the sensing signal as a center frequency fc, corresponding to the rotational speed omega ₁ that is calculated by the rotational speed calculating section _{25 α (= ω 1 / 2π} ) 2 has a characteristic (reverse notch filter characteristic) that allows signals in the range fc ± α to pass through and remove signals outside this range.

  The current peak timing extraction unit 33 detects the timing (maximum value timing) at which the current magnitude I obtained by the current value calculation unit 31 takes a maximum value (peak value) based on the output of the inverse notch filter 32. More specifically, the current peak timing extraction unit 33 performs peak hold processing for each sensing signal period, and extracts a maximum value timing at which the current magnitude I takes a maximum value during the sensing signal period. This local maximum timing corresponds to the rotational position (phase) of the rotor 50.

When the current peak timing extraction unit 33 generates the maximum value timing t, the current synchronization extraction unit 34 responds to the two-phase detection current I _αβ (t) at the timing t from the UVW / αβ coordinate conversion unit 17. Import and output.
The rotor position calculation unit 35 calculates the estimated rotational position θ ^ according to the following equation (4) using the local maximum timing two-phase detection current I _αβ (t) given from the current synchronization extraction unit 34.

θ ^ = Tan ⁻¹ (I _β / I _α ) (4)
FIG. 4A shows a high-frequency voltage vector corresponding to a sensing signal (rotational exploration voltage) generated by the sensing signal generator 21 and applied to the stator windings 51, 52, and 53. FIG. The current vector response to the high frequency voltage vector is shown. The high-frequency voltage vector formed by applying the rotation exploration voltage has a constant magnitude and rotates at a constant speed around the origin of the αβ coordinate. At this time, the magnitude of the current vector changes according to the pole position of the rotor 50. More specifically, the magnitude of the current vector, that is, the magnitude I of the current takes a maximum value at positions corresponding to the N pole and the S pole of the rotor 50, and the electrical angle with respect to them differs by two by 90 degrees. The current magnitude I takes a minimum value at the position. As a result, the end point of the current vector forms an elliptical locus 55 around the origin of the αβ coordinate. The ellipse has a major axis direction corresponding to the N pole and S pole of the rotor 50.

FIG. 5A is a waveform diagram showing an example of a current waveform. That is, an example of a time change of the current magnitude I is shown. In the current waveform, maximum points P1 and P2 corresponding to two major axis directions of an elliptical locus 55 (see FIG. 4B) formed by the end point of the current vector appear.
When the rotational speed ω of the rotor 50 is sufficiently small relative to the rotational speed of the high-frequency voltage vector, the current magnitude I takes two maximum points P1 and P2 during one cycle of the high-frequency voltage vector. This state is shown by a curve L2 in FIG.

  FIG. 6 is a flowchart for explaining the flow of processing that is repeatedly executed by the microcomputer 7 every predetermined control period. However, a process (N pole determination process) for specifying a maximum corresponding to the N pole of the rotor 50 among the maximums of the current signal that appears twice in one period (sensing signal period) of the high-frequency voltage vector is performed in advance. It shall be. A known method can be applied to the N pole determination process. For example, when a sufficiently large high-frequency voltage vector is applied, due to the magnetic saturation of the stator, the inductance on the N pole side of the rotor 50 is smaller than that on the S pole side, and the magnitude of the current vector in the N pole direction is maximum. The value is taken (see curve L1 in FIG. 5 (a)). Using this, one of the two maximum points of the current waveform can be specified as the maximum point corresponding to the N pole.

First, a sensing signal is injected from the sensing signal generator 21 into the stator windings 51 to 53 (step S1). By applying the sensing signal, a current response as shown in FIGS. 4B and 5A is obtained from the output of the current sensor 9.
Based on this current response, more specifically, based on the two-phase detection current I _αβ obtained by coordinate conversion of the three-phase detection current I _UVW by the UVW / αβ coordinate conversion unit 17, the position estimation unit 20 The estimated rotational position θ ^ is obtained (step S2). Further, the rotational speed ω ₁ is obtained by the rotational speed calculator 25 based on the estimated rotational position θ ^ (step S3).

On the other hand, the frequency analysis unit 26 performs a frequency analysis process on the three-phase detection current I _UVW to obtain the rotation speed ω ₂ of the rotor 50 without using the estimated rotation position θ ^ obtained by the position estimation unit 20. (Step S4).
The reliability verification unit 27 compares the rotational speed difference Δω (= | ω ₁ −ω ₂ |) with the speed difference threshold Thω (step S5). If the rotational speed difference Δω is equal to or smaller than the speed difference threshold Thω (step S5: YES), the reliability verification unit 27 determines that the estimated rotational position θ ^ is reliable (step S6). On the other hand, when the rotational speed difference Δω exceeds the speed difference threshold Thω (step S5: NO), the reliability verification unit 27 determines that the estimated rotational position θ ^ is not reliable (step S7).

If the reliability verification unit 27 determines that the estimated rotational position θ ^ is not reliable, the stop control unit 28 stops the motor 3 (step S8).
As described above, according to this embodiment, the difference between the rotational speed ω ₁ obtained based on the estimated rotational position θ ^ and the rotational speed ω ₂ obtained by frequency analysis and the speed difference threshold Thω are increased or decreased. Based on this, the reliability of the estimated rotational position θ ^ is evaluated. Then, for example, in a situation where the position estimation accuracy deteriorates and the reliability of the estimated rotational position θ ^ is impaired, as in the case where the motor parameters change due to changes in the surrounding environment such as the motor temperature, the motor 3 Control is stopped. Thereby, since it can suppress that the motor 3 is controlled improperly, the reliability of motor control can be improved and steering assistance can be performed more appropriately.

FIG. 7 is a block diagram for explaining the configuration of an electric power steering apparatus according to another embodiment of the present invention. In FIG. 7, portions corresponding to the respective portions shown in FIG. 1 are denoted by the same reference numerals as in FIG.
In this embodiment, in addition to the position estimation unit 20, the verification position estimation unit 40 for estimating the rotational position of the rotor 50 by an algorithm different from the position estimation unit 20 and the verification position estimation unit 40 using the estimated rotational position theta _V ^ estimated, and reliability verification unit 41 is provided to verify the estimated rotational position theta ^ reliability generated by the position estimation unit 20. Reliability verification unit 41 compares the estimated rotational position theta ^ obtained by the position estimation unit 20, the estimated rotational position theta _V ^ found by the verification position estimation unit 40. More specifically, the difference Δθ ^ (= | θ ^ −θ _V ^ |) between these estimated rotational positions θ ^ and θ _V ^ is compared with a predetermined position difference threshold Thθ. When the positional difference Δθ ^ is equal to or smaller than the positional difference threshold Thθ, that is, when the two estimated rotational positions θ ^ and θ _V ^ are in good agreement, the reliability verification unit 41 determines the rotor rotational position by the position estimating unit 20. It is determined that the estimation accuracy is high, and therefore the estimated rotational position θ ^ is reliable. Meanwhile, when the position difference [Delta] [theta] ^ exceeds the position difference threshold Thshita, i.e., two estimated rotational position theta ^, when theta _V ^ is significantly different from the estimated rotor rotational position by the position estimation unit 20 Therefore, it is determined that the estimated rotational position θ ^ is not reliable.

The stop control unit 28 stops the motor 3 in response to the reliability verification unit 27 determining that the estimated rotational position θ ^ is not reliable. Thereby, it is possible to prevent the motor 3 from being inappropriately controlled based on the inaccurate estimated rotational position θ ^.
FIG. 8 is a block diagram illustrating a configuration example of the verification position estimation unit 40. The verification position estimation unit 40 includes a signal processing unit 48 and a rotor position estimation unit 49. The signal processing unit 48 includes a voltage filter 61 configured by a low-pass filter that removes a high-frequency component of the two-phase indication voltage V _αβ ^* and a low-pass filter that removes a high-frequency component of the two-phase detection current I _αβ. Current filter 62. The rotor position estimation unit 49 is supplied with the two-phase command voltage V _αβ ^* and the two-phase detection current I _αβ after the signal processing (filtering) by the signal processing unit 48. The rotor position estimation unit 49 removes a high-frequency component from the disturbance observer 65 that estimates the induced voltage of the motor 3 as a disturbance and the estimated induced voltage output by the disturbance observer 65 based on a motor model that is a mathematical model of the motor 3. Estimated position generation for generating an estimated rotational position θ _V ^ of the rotor 50 based on an estimated value filter 66 composed of a low-pass filter and an estimated induced voltage (value after filtering) output from the estimated value filter 66 Part 67. Then, the two-phase command voltage V _αβ ^* filtered by the voltage filter 61 and the two-phase detection current I _αβ filtered by the current filter 62 are input to the disturbance observer 65 of the rotor position estimation unit 49, respectively. It has become.

FIG. 9 is a block diagram for explaining an example of the disturbance observer 65 and a configuration related thereto. A motor model that is a mathematical model of the motor 3 can be expressed as, for example, (R + pL) ⁻¹ . Here, R is an armature winding resistance, L is an αβ axis inductance, and p is a differential operator. It can be considered that the two-phase command voltage V _αβ ^* and the induced voltage E _αβ (α-axis induced voltage E _α and β-axis induced voltage E _β ) are applied to the motor 3.

The disturbance observer 65 receives a two-phase detection current I _αβ as an input and estimates a motor voltage, which is an inverse motor model (inverse model of the motor model) 68, a motor voltage estimated by the inverse motor model 68, and a two-phase indication voltage V _αβ. And a voltage deviation calculating unit 69 for obtaining a deviation from ^* . The voltage deviation calculator 69 obtains a disturbance with respect to the two-phase indicating voltage V _αβ ^* . As is apparent from FIG. 9, this disturbance is an estimated value E ^ _αβ (α-axis induced voltage corresponding to the induced voltage E _αβ. The estimated value E ^ _α and the β-axis induced voltage estimated value E ^ _β (hereinafter collectively referred to as “estimated induced voltage E ^ _αβ ”) The reverse motor model 65 is represented by R + pL, for example.

The estimated value filter 66 can be configured by, for example, a low-pass filter represented by a / (s + a). a is a design parameter, and the cut-off frequency ω _c of the estimated value filter 66 is determined by the design parameter a.
The induced voltage E _αβ can be expressed by the following equation (5). However, K _E is an induced voltage constant, θ is a rotor rotational position, and ω is a rotor rotational speed.

したがって、推定誘起電圧Ｅ^_αβが求まれば、次の(6)に従って、推定回転位置θ_V^が求まる。この演算が、推定位置生成部６７によって行われるようになっている。

Therefore, if the estimated induced voltage E ^ _.alpha..beta is determined, according to the following (6), the estimated rotational position theta _V ^ is obtained. This calculation is performed by the estimated position generator 67.

以上のとおり、この実施形態では、位置推定部２０によって求められる推定回転位置θ＾から独立し、かつ別のアルゴリズムで推定回転位置θ_V＾を求める検証用位置推定部４０が設けられている。そして、２つの推定回転位置θ＾，θ_V＾の差と位置差閾値Ｔｈθとの大小に基づいて、位置推定部２０による位置推定演算の信頼性が評価される。推定回転位置θ＾に信頼性がないときには、モータ３の制御が停止されるので、モータ制御の信頼性を向上することができる。

As described above, in this embodiment, the position estimation unit 20 independent of the estimated rotational position theta ^ as determined by, and validated for position estimation unit 40 for determining the estimated rotational position theta _V ^ at different algorithm is provided. The reliability of the position estimation calculation by the position estimation unit 20 is evaluated based on the difference between the two estimated rotational positions θ ^ and θ _V ^ and the position difference threshold Thθ. When the estimated rotational position θ ^ is not reliable, the control of the motor 3 is stopped, so that the reliability of the motor control can be improved.

  Note that the position estimation by the position estimation unit 20 using the current response to the sensing signal can achieve a certain level of accuracy when the rotational speed of the rotor 50 is in the low to medium speed range. On the other hand, the position estimation by the verification position estimation unit 40 using the induced voltage of the motor 3 has a certain level of accuracy in the medium / low speed range. Therefore, the determination by the reliability verification unit 41 may be performed in the medium speed range where the rotational speed of the rotor 50 is high in the position estimation accuracy of both the position estimation unit 20 and the verification position estimation unit 40.

As mentioned above, although two embodiment of this invention was described, this invention can also be implemented with another form.
For example, as a reliability verification means for verifying the reliability of the estimated rotational position θ ^, the output torque of the motor 3 is detected, and the output torque is compared with the current command value I _dq ^* (torque command value). You may do it. If the output torque of the motor 3 and the torque command value are greatly different, it is considered that the motor control is defective due to the deterioration of the position estimation accuracy, and therefore it is determined that the reliability of the estimated rotational position θ ^ is low. Can do.

In addition, a simple sensor for detecting the rotational position of the rotor 50 (simple sensor that is not as accurate as the resolver) is provided, and by comparing the output of this sensor with the estimated rotational position θ ^, the reliability of the estimated rotational position θ ^ It is good also as a structure which evaluates property.
Further, in the configuration of FIG. 1, the amount of change (movement distance) of the rotor rotational position over a predetermined time may be obtained by time-integrating the rotational speed ω ₂ obtained by the frequency analysis unit 26. . In this case, the amount of change of the estimated rotational position θ ^ obtained by the position estimation unit 20 at the predetermined time is compared with the amount of change of the rotor rotational position obtained by frequency analysis and integration processing, and the estimated rotational position θ ^ Reliability may be evaluated.

Further, for example, in the above-described embodiment, the rotational position θ of the rotor 50 is obtained using the two-phase detection current I _αβ when the current magnitude I takes the maximum value. However, the current magnitude I is the maximum. The estimated rotational position θ ^ may be obtained according to the following equation (7) using the two-phase indicating voltages V _α ^* and V _β ^* when taking the values (see the line 23 in FIGS. 1 and 7). .
θ ^ = Tan ⁻¹ (V _β ^* / V _α ^* ) (7)
Since the magnitude of the voltage vector is constant, the phase can be easily calculated as compared with the current vector in which distortion occurs. Therefore, the calculation process can be simplified by applying Expression (7).

  In order to obtain the phase of the voltage vector when the current magnitude I takes the maximum value, a counter that performs a counting operation in synchronization with the application of the sensing signal is used instead of performing the calculation according to the above equation (7). It may be. Specifically, a counter that is repeatedly operated so as to be initialized and to start a counting operation when the high-frequency voltage vector is along the α-axis (coincident with the U-phase direction) (that is, when the phase of the high-frequency voltage vector is zero). Is provided. For example, this counter counts up every period Ta / i obtained by dividing the period of the high-frequency voltage vector (sensing signal period) Ta into i equal parts (i is the number of samplings per period. For example, i = 360). The output represents the phase of the high frequency voltage vector. Therefore, as shown in FIG. 5 (c), if the count value of the counter is referred to when the maximum value of the current magnitude I is detected, the count value is obtained as the magnetic pole position of the rotor 50 (the magnitude of the current vector) Represents the phase angle of the high-frequency voltage vector when is maximum.

In the above-described embodiment, the example in which the present invention is applied to the motor 3 as the drive source of the electric power steering apparatus has been described. However, the present invention is not limited to the control of the motor for applications other than the electric power steering apparatus. Is also applicable.
In addition, various design changes can be made within the scope of matters described in the claims.

1 is a block diagram for explaining an electrical configuration of an electric power steering device to which a motor control device according to an embodiment of the present invention is applied. FIG. It is an illustration figure for demonstrating the structure of a motor. It is a block diagram for demonstrating the structure of a position estimation part. FIG. 4A shows a high-frequency voltage vector corresponding to the sensing signal (rotational exploration voltage), and FIG. 4B shows a response of the current vector to the high-frequency voltage vector. FIG. 5A shows an example of a current waveform, FIG. 5B shows an example of a voltage waveform, and FIG. 5C shows a change in the count value of the counter for detecting the phase of the voltage vector. It is a flowchart for demonstrating the flow of the process repeatedly performed by a microcomputer for every predetermined control period. It is a block diagram for demonstrating the structure of the electric power steering apparatus which concerns on other embodiment of this invention. It is a block diagram which shows the structural example of the position estimation part for verification. It is a block diagram for demonstrating an example of a disturbance observer and the structure relevant to this.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 5 ... Motor control apparatus, 7 ... Microcomputer, 9 ... Current sensor, 50 ... Rotor, 51, 52, 53 ... Stator winding

Claims (3)

A motor control device for controlling a motor including a rotor and a stator facing the rotor,
Sensing signal injection means for injecting a sinusoidal sensing signal into the stator;
Motor current detecting means for detecting a motor current flowing through the motor;
Position estimating means for estimating the rotor rotational position based on the motor current detected by the motor current detecting means;
And a verification unit that verifies the reliability of the rotor rotational position estimated by the position estimation unit.   The motor control device according to claim 1, further comprising: a motor stop control unit that stops the motor when the verification unit determines that the rotor rotational position estimated by the position estimation unit is not reliable. The verification means includes
First rotor rotation speed calculation means for calculating the rotor rotation speed based on the rotor rotation position estimated by the position estimation means;
Second rotor rotation speed calculation means for calculating the rotor rotation speed by analyzing the frequency of the motor current detected by the motor current detection means;
The rotor rotation speed calculated | required by the said 1st rotor rotation speed calculating means and the comparison means which compares the rotor rotation speed calculated | required by the said 2nd rotor rotation speed calculation means are included. Motor control device.

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