JP6768594B2 - AC motor control device - Google Patents

Description

The present invention relates to a control device that controls an AC motor without using a speed sensor.

There are various methods for controlling an AC motor, and they are used properly depending on the application. Especially when performing high-precision torque control or speed control, current control is performed as a minor loop of control.
The speed control methods of conventional induction motors are as follows. The control device outputs a three-phase voltage command to the PWM inverter. The control device generates a torque current command so that the detection speed follows the speed command. In addition, the exciting current command is generated so that the magnetic flux calculated from the detected current follows the magnetic flux command. Furthermore, the slip frequency and the output frequency are calculated, and the phase for coordinate conversion is obtained based on the output frequency. A voltage command on the two rotating axes is generated so that the detected current on the two rotating axes follows the torque current command and the exciting current command, and coordinate conversion is performed to generate a three-phase voltage command. Further, in the speed control method of the conventional synchronous motor, the phase for coordinate conversion is set as the magnetic pole position of the synchronous motor (see, for example, Non-Patent Document 1).

The conventional speed sensorless control method for controlling the speed of such an AC motor without the need for speed detection is shown below.
Using a state observer called an observer, which is constructed using a mathematical model of an induction motor, the observer inputs voltage information applied to the induction motor to estimate magnetic flux and current. Further, it has a configuration in which a predetermined gain is applied to the difference between the detected current and the estimated current of the induction motor and fed back. Then, by devising the feedback gain of the observer and adding the speed estimator of the induction motor, the speed control is performed without detecting the rotation speed of the induction motor. The mathematical model of the induction motor has built-in rotation speed information of the induction motor, and the same applies to the observer. Based on the concept of adaptive identification, if there is a difference between the estimated current output by the observer and the actual detected current (current estimation error), it is assumed that there is an error in the rotation speed information built into the observer, and based on the above current estimation error. This is a method to obtain the estimated speed by modifying the rotation speed information built into the observer. If speed control is performed based on this estimated speed, the rotation speed of the induction motor can be controlled to a desired value (see, for example, Non-Patent Document 2).
Further, even in the speed sensorless control method in the conventional synchronous motor, the speed is estimated by using an observer (see, for example, Patent Document 1).

Japanese Patent No. 4672236

Practice of theoretical design of AC servo system, General Electronic Publishing (1997), p. 122, p. 86 Hidehiko Sugimoto, et al. "Improvement of Stability of Speed Sensorless Vector Control of Induction Motors" Proceedings of the 2001 IEEJ National Convention, 4-109

In the speed sensorless control method of the conventional induction motor, the observer of the induction motor incorporates the slip frequency as a parameter in addition to the rotation speed information. Due to the difference in voltage-to-current transmission characteristics between the induction motor and the observer, there is a difference between the slip frequency and the estimated slip frequency obtained based on the estimated current output by the observer and the estimated magnetic flux. There is. This difference becomes particularly remarkable as the current of the induction motor is changed transiently, and the current estimation error by the observer includes not only the error due to the rotation speed information but also the error due to the slip frequency. For this reason, the transient accuracy of the estimated speed is lowered, and the speed control response is deteriorated.
When increasing the speed control response of an induction motor, the gain of the speed controller is usually increased to increase the torque swing width of the induction motor. However, as a result, the torque current of the induction motor also fluctuates greatly, and the difference in slip frequency also widens, so that the accuracy of the estimated speed decreases. Therefore, there is a problem that the improvement of the speed control response is limited.

In the speed sensorless control method in the conventional synchronous motor, the frequency of the coordinate conversion phase is track-controlled so as to synchronize the two-axis rotating coordinates in which the observer is constructed with the rotor magnetic flux of the synchronous motor. At that time, there is a problem that the transient accuracy of the estimated speed is lowered due to the error due to the transient fluctuation of the frequency for tracking control, and the improvement of the speed control response is limited as in the case of the induction motor.

The present invention has been made to solve the above-mentioned problems, and in a control device that performs speed sensorless control of an AC motor, improves the transient accuracy of the estimated speed and improves the speed control response. The purpose is to plan.

The control device for an AC motor according to the present invention includes a command generator that generates a voltage command on the dq axis for driving the AC motor, a current detection unit that detects a current on the dq axis of the AC motor, and the above. A state observation unit that inputs the detected current and the voltage command and outputs the estimated current and the estimated magnetic flux of the AC motor, and the estimated speed of the AC motor based on the estimated current, the estimated magnetic flux, and the detected current. A speed estimation unit that outputs the above, a first frequency calculation unit that calculates the first frequency so that the q-axis component of the estimated magnetic flux becomes 0 based on the estimated current, the estimated magnetic flux, and the detected current, and the above. It is generated in the output frequency calculation unit that calculates the output frequency, which is the frequency of the dq-axis rotation coordinate phase by adding the first frequency to the estimated speed of the AC motor, and in the state observation unit due to the first frequency. A compensation unit for generating a compensation signal for compensating for disturbance is provided. The state observation unit inputs the detected current, the voltage command, the compensation signal, the output frequency, and the first frequency, and calculates the estimated current and the estimated magnetic flux of the AC motor. It is a thing.

According to the control device of the AC motor according to the present invention, the disturbance in the state observation unit can be compensated by the compensation signal, the influence of the disturbance is suppressed, the transient accuracy of the estimated speed is improved, and the speed control response is improved. Can be planned.

It is a figure which shows the structure of the control device of the induction motor according to Embodiment 1 of this invention. It is a figure which shows the structure which realized the function of each part in the control device of the induction motor by Embodiment 1 of this invention by using software. It is a figure which shows the structure of the control device of the synchronous motor by Embodiment 2 of this invention.

Embodiment 1.
Hereinafter, as the control device for the AC motor according to the first embodiment of the present invention, the control device for the induction motor that performs speed sensorless control will be described.
FIG. 1 is a diagram showing a configuration of a control device for an induction motor according to the first embodiment.
As shown in FIG. 1, the control device 1 controls the induction motor 4 by driving and controlling an inverter 2 having a switching element. Hereinafter, the subscript * indicates the command value, and the subscript # indicates the estimated value. Further, the subscript s indicates the state quantity of the stator of the induction motor 4, and the subscript r indicates the state quantity of the rotor of the induction motor 4.
The control device 1 performs control calculation processing and outputs a voltage command Vuvw * (Vu *, Vv *, Vw *) to the inverter 2. In the inverter 2, the PWM process is performed, the switching element is driven based on the PWM process, and the output voltage in accordance with the voltage command Vuvw * is supplied to the induction motor 4. The current sensor 3 detects the current iuvw (iu, iv, iw) of the induction motor 4 and transmits a signal of the detected current to the control device 1.

The control device 1 includes coordinate conversion units 11a and 11b that perform coordinate conversion between three-phase stationary coordinates and dq-axis rotating coordinates, a current control unit 12, a speed control unit 14, a magnetic flux control unit 15, and a state observation. Observer 17 as a unit, velocity estimation unit 18, estimated slip frequency calculation unit 19 as a first frequency calculation unit that calculates the estimated slip frequency ωs # as the first frequency, and adder 13 as an output frequency calculation unit. And an adder 16 and a compensating unit 20. Further, the current detection unit 21 includes a current sensor 3 and a coordinate conversion unit 11a, and the command generation unit 22 includes a current control unit 12 and a speed control unit 14.
The current sensor 3 may be arranged outside the current detection unit. In that case, the current detection unit detects the current on the dq axis of the AC motor based on the current detected from the current sensor 3 iuvw. Further, the control device 1 of the induction motor may include the inverter 2.

The control device 1 performs the following processing to control the rotation speed of the induction motor 4 to a desired value.
The speed control unit 14 inputs the speed command ωr * and the estimated speed ωr #, performs arithmetic processing so that they match, and outputs the torque current command iq *. The magnetic flux control unit 15 inputs the magnetic flux command Φdr * and the estimated magnetic flux Φdr #, performs arithmetic processing so that they match, and outputs the exciting current command id *.
The detected current iuvw from the current sensor 3 is converted into a current idq (id, iq) on the dq-axis rotating coordinates by the coordinate conversion unit 11a and input to the current control unit 12. The current control unit 12 inputs the detected current idq (id, iq) on the dq axis, the torque current command iq *, and the exciting current command id *, performs arithmetic processing so that they match, and performs arithmetic processing on the dq axis. Outputs the voltage command Vdq * (Vd *, Vq *) of. The voltage command Vdq * is converted into a three-phase voltage command Vuvw * by the coordinate conversion unit 11b and output to the inverter 2.

The observer 17 outputs the estimated current idq # and the estimated magnetic flux Φdr # of the induction motor 4 based on the detected current idq and the voltage command Vdq *. The velocity estimation unit 18 outputs the estimated velocity ωr # based on the estimated current idq #, the estimated magnetic flux Φdr #, and the detected current idq. The estimated slip frequency calculation unit 19 calculates the estimated slip frequency ωs # based on the estimated current idq #, the estimated magnetic flux Φdr #, and the detected current idq. The adder 13 calculates the output frequency ω # by adding the estimated speed ωr # and the estimated slip frequency ωs #. The integrator 16 integrates the output frequency ω # to obtain the phase θ for coordinate conversion, and outputs the phase θ to the coordinate conversion units 11a and 11b. The compensation unit 20 generates a compensation signal ADD that compensates for the disturbance generated by the observer 17 based on the estimated current idq #, the estimated magnetic flux Φdr #, the detected current idq, and the estimated slip frequency ωs #. This compensation signal ADD is additionally input to the observer 17 together with the detection current idq and the voltage command Vdq *. The estimated slip frequency ωs # and the output frequency ω # are also input to the observer 17.

Next, the process of estimating the rotation speed of the induction motor 4 will be described. In this embodiment, the compensation signal ADD for compensating for the disturbance generated by the observer 17 is input to the observer 17, but here, the basic arithmetic processing will be described first.
Based on the arithmetic processing described in Non-Patent Document 2, the case where there is no compensation signal ADD will be described below. However, various methods have been studied for the configuration of the observer and the calculation of the estimated speed, and the method described below is an example, and this embodiment can be applied to a device in which a disturbance occurs inside the observer.

In the observer, the estimated current idqs # (ids #, iqs #) and the estimated magnetic flux Φdqr # (Φdr #, Φqr #) are set as state variables, and the following equation (1), which is the observer's equation of state, is calculated. The estimated current idqs # and the estimated magnetic flux Φdqr # are output. Here, since the known method described in Non-Patent Document 2 is used, detailed description thereof will be omitted.
In the equation (1), the input signal to the observer is an input signal to the state variable to be integrated. That is, in the equation (1), the second term on the right side based on the voltage information and the third term on the right side feeding back the current estimation error are input signals. Since the estimated current idqs #, which is the calculation result of the equation (1), is used for the calculation of the third term, the signals input to the observer from the outside are the detection current idqs (ids, iqs) and the voltage command Vds *.

However, Rs: primary resistance, Rr: secondary resistance, Ls: primary inductance, Lr: secondary inductance, M: mutual inductance, ωre #: estimated electrical angular frequency (pole pair multiple of estimated speed ωr #), g11 ~ G42: Observer feedback gain, ωs #: Estimated slip frequency, ω #: Output frequency (addition value of estimated electrical angular frequency and estimated slip frequency).
Since A11 #, A12 #, and A22 # in the above formula (1) contain ω #, ωre #, and ωs # inside, a subscript # indicating an estimated value is added.
The speed estimation unit 18 estimates the speed by the PI (proportional integration) operation shown in the following equation (2). However, Kp and Ki are speed estimation gains.

The estimated slip frequency calculation unit 19 performs the calculation shown in the following equation (3). This is for controlling the q-axis estimated magnetic flux Φqr # to be zero, and an object is to set the d-axis of the dq-axis rotating coordinate to the rotor magnetic flux of the induction motor 4.

The output frequency ω # obtained by adding the estimated speed ωr # and the estimated slip frequency ωs # is fed back and used as an observer parameter described in the above equation (1).
As described above, in the basic arithmetic processing, the estimated speed ωr # is obtained by calculating the above equations (1) to (3), and the estimated speed ωr # is fed back to the speed control unit 14. , Sensorless speed control of the induction motor 4 can be realized.

Next, the following equation (4) is a mathematical model of the induction motor 4 that rotates at a speed of ωr. However, Φdqr (Φdr, Φqr): magnetic flux, Vds: d-axis voltage, ωre: electric angular frequency (pole pairs of speed ωr), ωs: slip frequency, ω: output frequency (addition of electric angular frequency and slip frequency) value). The symbols not particularly explained are the same as those in the above equation (1).

As described above, a difference may occur between the slip frequency ωs and the estimated slip frequency ωs #, and the difference causes a disturbance signal DI in the observer. As shown in the above equation (1), the integrated value of the estimated slip frequency ωs # and the state variable (estimated current idqs #, estimated magnetic flux Φdqr #) is in the integrand on the right side. Similarly, in the mathematical model of the induction motor 4 shown in the above equation (4), the integrated value of the slip frequency ωs and the state variables (current idqs, magnetic flux Φdqr) is in the integrand on the right side. Therefore, the disturbance signal DI caused by the difference between the slip frequency ωs and the estimated slip frequency ωs # is also in the form of a product signal with the state variable of the observer.

Here, assuming that the voltage according to the voltage command is applied to the induction motor 4 and there is no difference between the estimated speed ωr # and the speed ωr, the disturbance signal DI is obtained from the right side of the above equation (1) and the above equation (4). It can be calculated by the difference from the right side and is expressed by the following equation (5).
If the observer performs the calculation of the above equation (1) when the estimated slip frequency ωs # is different from the slip frequency ωs, the disturbance signal DI is automatically generated inside the right side of the equation (1).

In this embodiment, in order to cancel the disturbance signal DI, the compensation unit 20 generates a compensation signal ADD whose polarity is inverted by applying a predetermined compensation feedback gain to the above equation (5) and adds it to the observer 17. input. The compensation signal ADD is represented by the following equation (6). k1 to k4 are compensating feedback gains.
Further, the secondary magnetic flux Φdr and the slip frequency ωs of the induction motor 4 are obtained based on the detected current idqs. The secondary magnetic flux Φdr is calculated by the following formula (6a), and the slip frequency ωs is calculated by the following formula (6b).

In the observer 17, the disturbance signal DI is compensated by adding the compensation signal ADD to the integrand calculation term on the right side of the above equation (1). That is, the observer 17 calculates the following equation (7), which is an equation of state, and outputs the estimated current idqs # and the estimated magnetic flux Φdqr #.

As shown in the above equation (6), the compensation signal ADD is a first signal composed of the product of the estimated slip frequency ωs # calculated using the estimated current idqs #, the estimated current idqs #, and the estimated magnetic flux Φdqr #, respectively. It is composed of a group and a difference signal group based on the difference between the group and the second signal group formed by multiplying the slip frequency ωs calculated by using the detected current idqs by the detected current idqs and the magnetic flux Φdqr.

As described above, in this embodiment, the observer 17 calculates the above equation (7), which is a state equation, and outputs the estimated current idqs # and the estimated magnetic flux Φdqr #, and the speed estimation unit 18 calculates the above equation (7). (2) is calculated to calculate the estimated slip frequency ωr #, and the estimated slip frequency calculation unit 19 calculates the above equation (3) to calculate the estimated slip frequency ωs #. An estimated speed ωr # is obtained by such an operation, and by feeding back the estimated speed ωr # to the speed control unit 14, sensorless speed control of the induction motor 4 can be realized.

In the observer 17, the disturbance signal DI generated internally due to the difference between the estimated slip frequency ωs # and the slip frequency ωs due to the estimated slip frequency ωs # is additionally input as the compensation signal ADD. Compensate by. Therefore, in the control device 1 of the induction motor 4, the disturbance signal DI generated inside the observer 17 can be suppressed, and the accuracy of the estimated current idqs # and the estimated magnetic flux Φdqr # is improved. Therefore, the transient accuracy of the estimated speed ωr # is improved, and the speed control response is improved.

As the compensation signal ADD, the priority may be set from the first to fourth rows of the above equation (6), and only the highly effective row may be used, and the calculation load can be reduced. The priority order is determined based on the characteristics of the induction motor 4 itself and the configuration of the entire control system.
For example, in this embodiment in which the observer 17 and the velocity estimation system are configured based on the non-patent document 2, the q-axis estimated magnetic flux Φqr # and the q-axis magnetic flux Φqr become zero by the action of the above equation (3). Is controlled by. Therefore, the magnitude of the signal in the third line of the above equation (6) becomes very small and can be omitted. That is, the compensation signal ADD may be represented by the following equation (8), in which case the equation of state of the observer 17 is represented by the following equation (8a).

By the way, generally, when operating an induction motor, the secondary magnetic flux is always controlled to be constant regardless of the load torque and speed. This is to ensure the linearity of the torque current and the output torque of the induction motor, but the d-axis current ids always flow along with it, and the d-axis current ids are used for integration. Therefore, the signal of the second line related to the d-axis current ids in the compensation signal ADD has a relatively large signal as compared with the signals of the other lines (first line, third line, fourth line). The second line of the equation of state of the observer 17 is a line in which the q-axis current iqs is a state variable, and the velocity is estimated using the estimation error of the q-axis current as described in the above equation (2). Therefore, the sensitivity to the disturbance signal DI is high as compared with other lines.
Therefore, the signal of the second line of the compensation signal ADD contributes most to the suppression operation of the disturbance signal DI. That is, the compensation signal ADD may be represented by the following equation (9), in which case the equation of state of the observer 17 is represented by the following equation (9a).

Further, when limiting to operating conditions in which the secondary magnetic flux is controlled to be constant and the d-axis current ids are controlled to be substantially constant, the d-axis current ids and the d-axis estimated current ids # are treated as fixed values. Can be done. At this time, the compensation signal ADD of the above equation (9) can be simplified as shown by the following equation (10), and in that case, the equation of state of the observer 17 is expressed by the following equation (10a). Here, both the secondary magnetic flux Φdr and the estimated magnetic flux Φdr # can be expressed by the product of the mutual inductance M and the fixed value of the d-axis current ids.

As described above, when the priority order is set from the first to fourth lines of the above equation (6) and the signals of some lines are omitted and used, compared with the case where all the signals are used. Although the effect of suppressing the disturbance signal DI is inferior to some extent, the number of compensation feedback gains k1 to k4 can be reduced, and the design can be simplified.

In this embodiment, a four-dimensional observer having a current and a magnetic flux as state variables is taken as an example, but if the observer has a structure having the product of the slip frequency and the state variables inside, the above-mentioned disturbance signal generation principle is the same. Therefore, it can be applied regardless of the type of observer.
The estimated current, estimated magnetic flux, and estimated slip frequency include both a time-varying component and a non-time-changing component. This also applies to the current, magnetic flux, and slip frequency of the induction motor. In the calculation of the compensation signal ADD, integration is performed, but the result of the integration processing includes a component that can be regarded as integration of a time-varying signal and a time-non-changing signal. That is, in the observer's equation of state, the addition of the compensation signal ADD can be regarded as equivalently manipulating the observer feedback gain. Therefore, it is necessary to design the compensation feedback gains k1 to k4 in the compensation signal ADD in consideration of the design of the existing observer feedback gain.

On the other hand, the result of the integration processing of the compensation signal ADD also includes a component that can be regarded as a product of signals including a time-changing component, which is different from a linear operation such as an observer feedback term. Therefore, it is necessary to consider this non-linear calculation result in the design of the compensation feedback gains k1 to k4 in the compensation signal ADD.
If the compensation feedback gains k1 to k4 of the compensation signal ADD are simply set to, for example, 1 in order to suppress the disturbance signal DI described in the above equation (5), the observer brought about by the observer feedback gain for the above reason It may affect the dynamic characteristics. Therefore, the compensation feedback gains k1 to k4 are designed in consideration of the circumstances including the above-mentioned linear operation and non-linear operation in the calculation of the compensation signal ADD, and the disturbance signal DI is compensated through the design. As a result, the effect of suppressing the influence of the disturbance signal DI and improving the speed control response in the speed sensorless control can be obtained while maintaining the characteristics of the original observer and ensuring the stability.

The control device 1 shown in the above embodiment can also realize the functions of each part by using software, and will be described below with reference to FIG.
As shown in FIG. 2, the control device 1 includes a storage device 6 and a processor 7, and the result of signal processing of a program executed on the processor 7 and signal processing in a logic circuit provided on the processor 7 The configuration is such that the functions are realized. The processor 7 exchanges read / write signals 8 with the storage device 6, reads a program from the storage device 6, and executes the program. Further, the processor 7 writes the information to be temporarily stored in the storage device 6 in the process of the processing and reads the information from the storage device 6.
For example, the functions of the coordinate conversion units 11a and 11b, the current control unit 12, the speed control unit 14, the magnetic flux control unit 15, the integrator 16, the observer 17, the speed estimation unit 18, the estimated slip frequency calculation unit 19, and the compensation unit 20 are It can be realized by executing a program on the processor 7. In this way, even when the functions of each part are realized by the processing by the processor 7, the same effect can be obtained by the same operation as that of the first embodiment.

Embodiment 2.
Next, as a control device for the AC motor according to the second embodiment of the present invention, a control device for a synchronous motor that performs speed sensorless control will be described.
FIG. 3 is a diagram showing a configuration of a control device for a synchronous motor according to a second embodiment.
As shown in FIG. 3, the control device 1 controls the synchronous motor 5 by driving and controlling the inverter 2 having a switching element.
The control device 1 performs control calculation processing and outputs a voltage command Vuvw * (Vu *, Vv *, Vw *) to the inverter 2. In the inverter 2, the PWM process is performed, the switching element is driven based on the PWM process, and the output voltage according to the voltage command Vuvw * is supplied to the synchronous motor 5. The current sensor 3 detects the current iuvw (iu, iv, iw) of the synchronous motor 5 and transmits a signal of the detected current to the control device 1.

The control device 1 includes coordinate conversion units 11a and 11b that perform coordinate conversion between three-phase stationary coordinates and dq-axis rotating coordinates, a current control unit 12, a speed control unit 14, and an observer 17A as a state observation unit. A speed estimation unit 18, a tracking frequency calculation unit 19A as a first frequency calculation unit that calculates a tracking frequency ωt # as a first frequency, an adder 13, an integrator 16, and a compensation unit 20A are provided. Further, the current detection unit 21 includes a current sensor 3 and a coordinate conversion unit 11a, and the command generation unit 22 includes a current control unit 12 and a speed control unit 14.
The current sensor 3 may be arranged outside the current detection unit. In that case, the current detection unit detects the current on the dq axis of the AC motor based on the current detected from the current sensor 3 iuvw. Further, the control device 1 of the synchronous motor may include the inverter 2.

The control device 1 performs the following processing to control the rotation speed of the synchronous motor 5 to a desired value.
The speed control unit 14 inputs the speed command ωr * and the estimated speed ωr #, performs arithmetic processing so that they match, and outputs the torque current command iq *. The exciting current command id * is usually set to zero, but it may be set to a negative value when the output voltage of the inverter 2 is saturated. The detected current iuvw from the current sensor 3 is converted into a current idq (id, iq) on the dq-axis rotating coordinates by the coordinate conversion unit 11a and input to the current control unit 12. The current control unit 12 inputs the detected current idq (id, iq) on the dq axis, the torque current command iq *, and the exciting current command id *, performs arithmetic processing so that they match, and performs arithmetic processing on the dq axis. Outputs the voltage command Vdq * (Vd *, Vq *) of. The voltage command Vdq * is converted into a three-phase voltage command Vuvw * by the coordinate conversion unit 11b and output to the inverter 2.

The observer 17A outputs the estimated current idq # and the estimated magnetic flux Φdr # of the synchronous motor 5 based on the detected current idq and the voltage command Vdq *. The velocity estimation unit 18 outputs the estimated velocity ωr # based on the estimated current idq #, the estimated magnetic flux Φdr #, and the detected current idq. The tracking frequency calculation unit 19A calculates the tracking frequency ωt # based on the estimated current idq #, the estimated magnetic flux Φdr #, and the detected current idq. The follow-up frequency ωt # is a correction frequency for synchronizing the d-axis of the dq-axis rotating coordinate with the rotor magnetic flux of the synchronous motor 5.
The adder 13 calculates the output frequency ω # by adding the estimated speed ωr # and the tracking frequency ωt #. The integrator 16 integrates the output frequency ω # to obtain the phase θ for coordinate conversion, and outputs the phase θ to the coordinate conversion units 11a and 11b. The compensation unit 20A generates a compensation signal ADDa that compensates for the disturbance generated in the observer 17A based on the estimated current idq #, the estimated magnetic flux Φdr #, the detected current idq, and the tracking frequency ωt #. This compensation signal ADDa is additionally input to the observer 17A together with the detection current idq and the voltage command Vdq *. The tracking frequency ωt # and the output frequency ω # are also input to the observer 17A.

Next, the processing for estimating the rotation speed of the synchronous motor 5 will be described. In this embodiment, the compensation signal ADDa for compensating for the disturbance generated in the observer 17A is input to the observer 17A, but here, the basic arithmetic processing will be described first.
A case where there is no compensation signal ADDa will be described below based on the arithmetic processing described in Patent Document 1. However, various methods have been studied for the configuration of the observer and the calculation of the estimated speed, and the method described below is an example, and this embodiment can be applied to a device in which a disturbance occurs inside the observer.

In the observer, the estimated current idqs # (ids #, iqs #) and the estimated magnetic flux Φdqr # (Φdr #, Φqr #) are set as state variables, and the following equation (11), which is the observer's equation of state, is calculated. The estimated current idqs # and the estimated magnetic flux Φdqr # are output. Since a known method is used here, detailed description thereof will be omitted.
In the equation (11), the input signal to the observer is an input signal to the state variable to be integrated. That is, in the equation (11), the second term on the right side based on the voltage information and the third term on the right side feeding back the current estimation error are input signals. Since the estimated current idqs #, which is the calculation result of the equation (11), is used for the calculation of the third term, the signals input to the observer from the outside are the detection current idqs (ids, iqs) and the voltage command Vds *.

However, Rs: resistance, L: inductance, ωre #: estimated electric angular frequency (pole logarithmic multiple of estimated speed ωr #), g11 to g42: observer feedback gain, ω #: output frequency (estimated electric angular frequency and tracking frequency) Addition value with).
The speed estimation unit 18 estimates the speed by the PI calculation shown in the following equation (12). However, Kp and Ki are speed estimation gains.

The tracking frequency calculation unit 19A performs the calculation shown in the following equation (13). This calculation controls the q-axis estimated magnetic flux Φqr # to be zero, and the tracking frequency ωt # is the coordinate conversion phase so that the d-axis of the dq-axis rotation coordinate is synchronized with the rotor magnetic flux of the synchronous motor 5. It is a frequency that corrects the frequency of θ (output frequency).

The output frequency ω # obtained by adding the estimated speed ωr # and the tracking frequency ωt # is fed back and used as an observer parameter described in the above equation (11).
As described above, in the basic arithmetic processing, the estimated speed ωr # is obtained by calculating the above equations (11) to (13), and the estimated speed ωr # is fed back to the speed control unit 14. , Sensorless speed control of the synchronous motor 5 can be realized.

Next, the following equation (14) is a mathematical model of the synchronous motor 5 that rotates at a speed of ωr. However, Φdqr (Φdr, Φqr): magnetic flux, Vds: d-axis voltage, ωre: electric angular frequency (pole logarithmic multiple of velocity ωr). The symbols not particularly explained are the same as those in the above equation (11).

As shown by the above equation (14), the mathematical model itself of the synchronous motor 5 does not have a parameter of the frequency ωt which is the basis of the tracking frequency ωt #. In the first embodiment, the disturbance signal DI is generated in the observer due to the difference between the slip frequency ωs and the estimated slip frequency ωs #, but in the second embodiment, the tracking frequency ωt # itself. Disturbance occurs due to.
The tracking frequency ωt # is calculated so as to synchronize the d-axis of the dq-axis rotating coordinate in which the observer shown in the above equation (11) is installed with the rotor magnetic flux of the synchronous motor 5. Since this tracking frequency ωt # is calculated based on the current estimation error, when the torque current iqs of the synchronous motor 5 is transiently operated, disturbance may occur due to the difference in transmission characteristics between the observer and the synchronous motor. appear.

Here, assuming that the voltage according to the voltage command is applied to the synchronous motor 5 and there is no difference between the estimated speed ωr # and the speed ωr, the disturbance signal DIa is obtained from the right side of the above equation (11) and the above equation (14). It can be calculated by the difference from the right side and is expressed by the following equation (15). The frequency corresponding to the frequency ωt is expressed as 0. When the observer performs the calculation of the above equation (11), the disturbance signal DIa is automatically generated inside the right side of the equation (11).

In this embodiment, in order to cancel the disturbance signal DIa, the compensation unit 20A generates a compensation signal ADDa whose polarity is inverted by applying a predetermined compensation feedback gain to the above equation (15) and adds it to the observer 17A. input. The compensation signal ADDa is represented by the following equation (16). k1 to k4 are compensating feedback gains. Further, Φdr # is a kind of motor constant indicating the magnitude of the rotor magnetic flux called the induced voltage constant, and is a value set in advance like the resistance value and the inductance value of the motor.

In the observer 17A, the disturbance signal DIa is compensated by adding the compensation signal ADDa to the integrand calculation term on the right side of the above equation (11). That is, the observer 17A calculates the following equation (17), which is an equation of state, and outputs the estimated current idqs # and the estimated magnetic flux Φdqr #.

As shown in the above equation (16), the compensation signal ADDa is a first signal group composed of the product of the tracking frequency ωt # calculated using the estimated current idqs # multiplied by the estimated current idqs # and the estimated magnetic flux Φdqr #, respectively. Consists of.
As described above, in this embodiment, the observer 17A calculates the above equation (17), which is a state equation, and outputs the estimated current idqs # and the estimated magnetic flux Φdqr #, and the speed estimation unit 18 calculates the above equation (17). (12) is calculated to calculate the estimated speed ωr #, and the tracking frequency calculation unit 19A calculates the above equation (13) to calculate the tracking frequency ωt #. An estimated speed ωr # is obtained by such an operation, and by feeding back the estimated speed ωr # to the speed control unit 14, sensorless speed control of the synchronous motor 5 can be realized.

In the observer 17A, the disturbance signal DIa generated internally due to the tracking frequency ωt # is compensated by additionally inputting the compensation signal ADDa. Therefore, in the control device 1 of the synchronous motor 5, the disturbance signal DIa generated inside the observer 17A can be suppressed, and the accuracy of the estimated current idqs # and the estimated magnetic flux Φdqr # is improved. Therefore, the transient accuracy of the estimated speed ωr # is improved, and the speed control response is improved.

As in the first embodiment, the compensation signal ADDa may be used only in the highly effective line by setting the priority order among the first to fourth lines of the above equation (16). The load can be reduced. The priority order is determined based on the characteristics of the induction motor 4 itself and the configuration of the entire control system.
For example, in this embodiment in which the observer 17A and the velocity estimation system are configured based on the above Patent Document 1, the q-axis estimated magnetic flux Φqr # and the q-axis magnetic flux Φqr are set to zero by the action of the above equation (13). Be controlled. Therefore, the magnitude of the signal in the third line of the above equation (16) becomes very small and can be omitted. That is, the compensation signal ADDa may be represented by the following equation (18), in which case the equation of state of the observer 17A is expressed by the following equation (18a).

As described above, the compensation signal ADDa is inferior in the effect of suppressing the disturbance signal DI to some extent as compared with the case where all the signals are used by omitting the signals of some lines, but the number of compensation feedback gains k1 to k4. Can be reduced and the design can be simplified.

In this embodiment, a four-dimensional observer having a current and a magnetic flux as state variables is taken as an example, but if the observer is constructed on the dq-axis rotational coordinates, the d-axis is synchronized with the rotor magnetic flux of the synchronous motor 5. Since processing or similar processing is performed to include a component corresponding to the tracking frequency ωt # and a similar disturbance signal DIa is generated, it can be applied.
Further, the design of the compensation feedback gains k1 to k4 in the compensation signal ADDa is designed in consideration of the existing observer feedback gain and the non-linear arithmetic processing as in the first embodiment. The compensation feedback gains k1 to k4 are designed in this way, and the disturbance signal DIa is compensated through the design. As a result, it is possible to obtain the effect of suppressing the influence of the disturbance signal DIa and improving the speed control response in the speed sensorless control while maintaining the characteristics of the original observer and ensuring the stability.

In the above embodiment as well, the configuration shown in FIG. 2 may be used, and the functions of each part may be realized by processing by the processor 7.

Further, in the present invention, each embodiment can be freely combined, and each embodiment can be appropriately modified or omitted within the scope of the invention.

Claims (6)

A command generator that generates a voltage command on the dq axis to drive an AC motor,
A current detector that detects the current on the dq axis of the AC motor,
A state observation unit that inputs the detected current and the voltage command and outputs the estimated current and the estimated magnetic flux of the AC motor.
A speed estimation unit that outputs the estimated speed of the AC motor based on the estimated current, the estimated magnetic flux, and the detected current.
A first frequency calculation unit that calculates the first frequency so that the q-axis component of the estimated magnetic flux becomes 0 based on the estimated current, the estimated magnetic flux, and the detected current.
An output frequency calculation unit that calculates the output frequency, which is the frequency of the dq-axis rotating coordinate phase, by adding the first frequency to the estimated speed of the AC motor.
It is provided with a compensation unit that generates a compensation signal that compensates for the disturbance generated in the state observation unit due to the first frequency.
The state observation unit inputs the detected current, the voltage command, the compensation signal, the output frequency, and the first frequency, and calculates the estimated current and the estimated magnetic flux of the AC motor. ,
Control device for AC motors. The compensation unit generates the compensation signal by using a signal in the first signal group obtained by multiplying the first frequency by the estimated current and the estimated magnetic flux, respectively.
The control device for an AC motor according to claim 1. The AC motor is an induction motor, and the first frequency is an estimated slip frequency.
The compensator uses a slip frequency and magnetic flux calculated based on the detected current, and is a difference signal between the second signal group obtained by multiplying the slip frequency by the detected current and the magnetic flux, respectively, and the first signal group. At least one of the groups is the compensation signal.
The control device for an AC motor according to claim 2. The AC motor is a synchronous motor, and the first frequency is a follow-up frequency for synchronizing the d-axis with the rotor magnetic flux.
The control device for an AC motor according to claim 2. The dq-axis rotating coordinate phase of the AC motor, which is calculated based on the output frequency, is used for coordinate conversion between the dq-axis rotating coordinate and the three-phase stationary coordinate in both the detected current and the voltage command.
The control device for an AC electric motor according to any one of claims 1 to 4. The command generation unit includes a speed control unit that generates a current command so that the estimated speed follows the speed command, and a current control unit that generates a voltage command so that the detected current follows the current command. Prepared
The control device for an AC electric motor according to any one of claims 1 to 5.

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