JPH07123798A - Speed sensorless vector control system for induction motor - Google Patents

Abstract

PURPOSE:To vector-control under complete non-interference control by preventing a deviation of a vector control coordinate axis to become a 'delay' of estimating step of an induction motor real speed. CONSTITUTION:In order to correct a deviation of a power source angular frequency omega0 generated by a 'delay' of an estimated value omegar of a motor speed, motor secondary magnetic flux estimated values lambda2d, lambda2q in stator coordinates estimated by the same dimensional magnetic flux observer 4 are coordinate- converted to motor secondary magnetic flux lambda2 in a synchronous rotary coordinate axis of a coordinate converter 10. A torque axis component lambda2b of the flux lambda2 is integrated in terms of a slip angle frequency correcting integrator 16 to obtain a slip angle frequency correction value omegasc, and added to a power source angle frequency command value omegas by an adder 17. Thus, the deviation of a motor real speed estimated value omegar is corrected by the value omegasc thereby to prevent a deviation of a basic phase angle theta0 due to the deviation of the frequency omega0 thereby to prevent the deviation of a coordinate axis, thereby providing a compete vector control.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor vector control system, and more particularly to an induction motor vector control system which does not use a speed sensor.

[0002]

2. Description of the Related Art A vector control method has become popular as a high-performance speed control method for an induction motor, and a speed sensorless vector control method for controlling this without a speed sensor is known.

FIG. 2 shows a conventional speed sensorless vector control system of an induction motor which estimates the actual speed of the induction motor using a speed adaptive secondary magnetic flux observer.

First, the estimation of the actual speed (rotational angular frequency ω _r ) of the induction motor 1 will be described with reference to FIG.

The voltage equation of the induction motor 1 is given by the following equation (1) when it is represented by the ab axis for observing various quantities from the synchronous rotation coordinate system rotating at the power source angular frequency (ω ₀ ).

[0006]

[Equation 1]

However, v ₁ a, v ₁ b ... primary excitation axis voltage on the synchronous rotation coordinate (ab axis), primary torque axis voltage (V) i ₁ a, i ₁ b ... synchronous rotation coordinate (ab axis) ) Above primary excitation axis current, primary torque axis current (A) λ ₂ a, λ ₂ b ... secondary excitation axis magnetic flux on synchronous rotation coordinate (ab axis), secondary torque axis magnetic flux (Wb) ω ₀ ‥‥‥‥ Power supply angular frequency (rad / sec) ω _r ‥‥‥‥ Motor speed (rotational angular frequency, rad / sec) ω _s ‥‥‥‥ Slip angular frequency (rad / sec) R ₁ , R ₂ ‥‥‥ Primary and secondary resistance (Ω) L ₁ , L ₂ …… Primary and secondary inductance (H) M ‥‥‥‥ Mutual (excitation) inductance (H) Lσ ‥‥‥ Equivalent leakage inductance (H) ( Lσ = (L
₁ L ₂ −M ² ) / L ₂ ) s ·········· Time fine molecule (d / dt) and the power source angular frequency (ω ₀ ), motor speed (ω _r ), slip angular frequency (ω _s *) The relationship and the calculation of the slip angular frequency (ω _s *) are expressed by the following equation (2). ω ₀ = ω _r + ω ₃ * ω _s * = i ₁ b * / i ₁ a ・ τ ₂・ ・ ・ ・ ・ ・ ・ ・ ・ ・ (2) However, τ ₂ ‥‥‥‥‥‥‥‥‥‥‥‥‥ Secondary time constant (τ ₂ = L ₂ / R ₂ ) Subscript (*) ..... indicates command value or set value.
Now, i ₁ a * = constant ‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥ (3), and in the above equation (1), (2), (3 ), The primary voltage command value (v ₁
a *, v ₂ b *) is given by the following equation (4) which is a constitutive equation of the digital current controller 3,

[0008]

[Equation 2]

[0009] synchronous rotating coordinate primary current i1 on (ab axis) the command value _{i 1 * (i 1 a *} , i 1 b *) as expected current flow,
Secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on synchronous rotation coordinates (ab axis)
Is kept at λ ₂ a = Mi ₁ a (constant), λ ₂ b = 0 .................................. (5).

As a result, the torque (T) of the induction motor 1
Is T = M / L ₂ · (λ ₂ a · i ₁ b−λ ₂ b · i ₁ a) = M ² / L ₂ · (i ₁ a · i ₁ b) (6) Secondary magnetic flux λ ₂ (λ
₂ a, λ ₂ b) and the secondary current i ₂ (i ₂ a, i ₂ b) are irrelevant, and vector control of decoupling control is established.

By the way, as is apparent from the equation (2), when the speed sensor is not used, the slip angular frequency ω _s *
Even if is set, the motor speed ω _r is unknown,
Although the power source angular frequency ω ₀ cannot be determined, the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on the synchronous rotation coordinate (ab axis) that rotates at the power source angular frequency ω ₀ is (5) above. By controlling the power source angular frequency ω ₀ so as to satisfy the expression, similarly, vector control of decoupling control can be realized. That is, by using the velocity adaptive secondary magnetic flux observer composed of the same-dimensional magnetic flux observer 4 and the velocity adaptive mechanism 7, the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) is estimated, and the actual motor speed ω _r is estimated (ω _r #) based on the secondary magnetic flux estimated value λ ₂ # (λ ₂ a #, λ ₂ b #). ) Expression (ω ₀ =
By obtaining the power source angular frequency ω ₀ from ω _r # + ω _s *) and controlling the digital current controller 3 with the power source angular frequency ω ₀ , vector control of decoupling control can be realized.

In the conventional induction motor speed sensorless vector control system in the vector control system shown in FIG. 2, in order to detect the actual speed of the induction motor 1 without using the speed sensor, the primary current of the induction motor 1 ( Phase current) iu, iv, iw is detected and the number of phases is converted by the three-phase to two-phase converter 14 on the stator coordinate (dq axis) primary current detection value i ₁ (i ₁ d, i ₁ q), and the primary current detection value i ₁ and PW
Same as the input of the motor primary voltage command value v ₁ * (v ₁ d *, v ₁ q *) on the stator coordinates (dq axes) to the M control inverter 2 and the motor actual speed estimated value ω _r # Dimensional magnetic flux observer 4
Causes the secondary magnetic flux of the motor on the stator coordinates (dq axes) λ ₂ (λ
₂ d, λ ₂ q) and the motor primary current i ₁ (i ₁ d, i ₁ q) are estimated, and the speed adaptation mechanism 7 estimates the primary current i ₁ # (i ₁ d
#, I ₁ q #) and the primary current detection value i ₁ (i ₁ d, i ₁ q) are compared based on the estimated error signal (i ₁ −i ₁ #)
The actual speed (ω _r #) of the electric motor is calculated and estimated by the adaptive adjustment rule represented by the following, and the speed of the induction motor 1 is detected.

Ω _r # = K _p (e _i dλ ₂ q # −e _i qλ ₂ d #) + K _i ∫ (e _i dλ ₂ q # −e _i qλ ₂ d #) dt (7) However, , E _i d = i ₁ d−i ₁ d #: estimation error e _i q = i ₁ q−i ₁ q #: estimation error K _p : velocity estimation part proportional gain K _i : velocity estimation part integration gain Dimensional secondary magnetic flux observer 4 and speed adaptation mechanism 7
For the speed sensorless vector control method of the induction motor 1 that estimates the actual speed of the induction motor by the speed adaptive secondary magnetic flux observer consisting of and, see "Journal of the Institute of Electrical Engineers of Japan, Volume 11, No. 11, 1991" (Kubota, Ozaki, Matsuse, Nakano: "Velocity sensorless direct vector control of induction motor using adaptive secondary magnetic flux observer").

A conventional speed sensorless vector control system for estimating the actual speed of the induction motor using the speed adaptive secondary magnetic flux observer will be described below.

Explaining the operation in FIG. 2 (control system configuration), when the motor speed command (ω _r *) is given to the speed controller (ASR), the speed command value (ω _r *) and the motor actual speed are given. The estimated value (ω _r #) is compared, the comparison error signal is proportional-integral (PI) controlled by the speed controller 6, and the primary current command value i on the synchronous rotation coordinate (ab axis) of the induction motor 1 is calculated. is converted to ₁ * primary torque axis current command value is the torque axis component of (i ₁ b *). Next, the current controller (AC
In the digital current controller 3 in R), the synchronous rotation coordinate of the induction motor 1 that controls the PWM control inverter 2
The primary voltage command value v ₁ * (v ₁ a *, v ₁ b *) on the (ab axis) is
The primary current command value i ₁ * (i ₁ a *, i ₁ b *) and the primary current detection value i ₁ (i ₁ a, i ₁ b) are equal (i ₁ a * = i ₁ a, i ₁ b *
= I ₁ b) is calculated by the above-mentioned expression (4) which is a conditional expression that enables decoupling control.

The primary voltage command value v ₁ * (v ₁ a *, v ₁ b *) on the synchronous rotation coordinate (ab axis) is converted by the coordinate converter 9 into the primary voltage on the stator coordinate (dq axis). Command value v ₁ * (v ₁ d *, v ₁ q *)
Is converted to the output voltage control command voltage Vu, Vv, Vw of each phase of the PWM control inverter 2, and the PWM control is performed. The output voltage of each phase of the inverter 2 is controlled. As a result, the induction motor 1 is speed-controlled according to the desired speed command value (ω _r *). Also used for coordinate converters 8 and 9 for converting between synchronous rotation coordinates (ab axis) rotating at the power supply angular frequency ω ₀ and stator coordinates (dq axes) fixed to the stator of the induction motor 1. Unit vector (Sin θ ₀ , Co
basic phase angle θ ₀ (θ ₀ = ω ₀ t) for producing sθ ₀ ).
Is calculated by the slip calculator 5 as shown in the above equation (2),
Primary excitation axis current command value i ₁ a * on synchronous rotation coordinates (ab axis)
Primary torque axis current command value i ₁ b * and induction motor 1
Of the slip angular frequency command value (ω _s *) obtained by the secondary time constant τ ₂ (= L ₂ / R ₂ ) of the motor and the estimated value of the actual motor speed estimated by the speed adaptive secondary magnetic flux observer (4,7) ( It can be obtained by integrating the power source angular frequency (ω ₀ ) obtained from ω _r #) by the basic phase angle calculating integrator 11.

The delay compensating element 12 is provided to moderate the rising and falling of the slip angular frequency command value (ω _s *) in order to match the control delay in the digital current controller 3.

[0018]

As described above, since the conventional speed sensorless vector control system has a delay in the calculation process for estimating the actual motor speed value (ω _r ),
Actual velocity estimate (ω _r #) and slip angular frequency estimate (ω _s *)
Power source angular frequency ω ₀ (ω ₀ = ω _r # +
ω _s *) deviates from the true value, and as a result, the motor primary voltage command value v ₁ * for decoupling control calculated based on the power supply angular frequency ω ₀ in the digital current controller 3 The shift and the basic phase angle θ ₀ used for coordinate conversion in the coordinate converters 8 and 9 may also shift, and eventually the coordinate conversion axis may shift and the vector control of the decoupling control may not be established.

This means that the induction motor 1
In the speed control of 1), a serious problem that the torque according to the torque command value cannot be obtained occurs.

The present invention has been made in view of the above points, and complete vector control can be performed by preventing the deviation of the vector control coordinate axis which is a "delay" in the process of estimating the actual speed of the induction motor. The purpose is to obtain a speed sensorless vector control method.

[0021]

A speed sensorless vector for performing vector control of decoupling control of an induction motor by a digital current controller controlled based on a comparison error signal comparing a speed command value and an actual speed estimated value. In the control method, the actual speed value of the induction motor is estimated by the speed adaptive secondary magnetic flux observer including the same-dimensional magnetic flux observer and the speed adaptive mechanism, and the secondary magnetic flux estimation of the induction motor is estimated by the same-dimensional magnetic flux observer. A slip angular frequency correction value is obtained based on the value, and the slip angular frequency command value calculated based on the primary current command value of the induction motor is corrected by the slip angular frequency correction value and added to the actual speed estimated value of the induction motor. By doing so, a power source angular frequency for controlling the digital controller and the coordinate converter is obtained.

[0022]

The "delay" in the process of estimating the actual speed estimated value (ω _r #) of the induction motor 1 is the slip angular frequency command value (ω) calculated by the slip calculator 5 when viewed from the induction motor 1 side.
_s *) is shifted.

(Equation (2) above, ω ₀ = ω _r + ω _s *, ω _s * = i
₁ b * / i ₁ a * -see τ ₂ ). Therefore, the requirement for determining the power supply angular frequency ω ₀ when the vector control is established, that is, the secondary magnetic flux estimated value λ ₂ # on the synchronous rotation coordinate (ab axis)
Torque axis component of (lambda ₂ b #) zero (the above equation (5), (6) see formula) in order to, the secondary magnetic flux torque axis component estimate .lambda.2 _b
The slip angular frequency correction value ω _sc obtained by integrating # (ω _sc = Kω _i ∫λ ₂ b # · dt Kω _i : integral gain) is added to the slip angular frequency command value ω _s * (ω _s * + ω _sc )
The deviation Δω ₀ of the power source angular frequency ω ₀ due to the “lag” of the estimated speed value ω _r # of the electric motor 1 can be corrected, the deviation of the coordinate axes can be prevented, and complete vector control can be performed. (See the following equation) ω ₀ + Δω ₀ = ω _r # + ω _s * + ω _sc Δω ₀ = ω _sc

[0024]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 shows a control system according to an embodiment of the present invention.

In the illustrated control system, when the motor speed command (ω _r *) is given to the speed controller (ASR), the speed command value (ω _r *) and the actual speed estimated value (negative feedback signal) ( ω _r #) and the comparison error signal is proportional-integral (PI) controlled by the speed controller 6, and the torque axis component (1) of the motor primary current command value i ₁ * on the synchronous rotation coordinate (ab axis) ( i ₁ b *). Next, the current controller (A
In the digital current controller 3 in CR), the torque axis component (i ₁ b *) and the excitation axis component (i ₁ a *) of the motor primary current command value i ₁ * and the motor primary current detection value i ₁ Of the torque axis component (i ₁ b) and the excitation axis component (i ₁ a) are compared, and PWM control is performed so that i ₁ b * = i ₁ b and i ₁ a * = i ₁ a are controlled. The primary voltage command value v ₁ * (v ₁ a *, v ₁ b *) on the synchronous rotation coordinate axis (ab axis) that controls the inverter 2 is calculated by the above equation (4).

The primary voltage command value v ₁ * (v ₁ a *, v ₁ b *), which is the output of the digital current controller 3, is converted by the coordinate converter 9 into the motor primary on the stator coordinates (dq axes). Voltage command value v ₁ *
After being converted into (v ₁ d *, v ₁ q *), the number of phases is converted by the 2-phase to 3-phase phase converter 15 and the primary voltage control command voltage Vu of each of the three phases of the PWM control inverter 2 is output. , Vv, Vw, and controlling the output voltage of the PWM control inverter 2, the induction motor 1 is speed-controlled according to a desired speed command value (ω _r *).

The actual speed value (ω _r ) of the induction motor 1 is the primary current detection value i on the stator coordinates (dq axes) of the induction motor 1.
₁ (i ₁ d, i ₁ q), primary voltage command value v ₁ * (v ₁ d *, v ₁ q *)
And the same-dimensional magnetic flux observer 4 that receives the actual velocity estimated value (ω _r #), and the primary current estimated value i ₁ # (i ₁ d #, i ₁ q #) estimated by the same-dimensional magnetic flux observer 4. Secondary magnetic flux estimated value λ ₂ # (λ ₂ d #, λ ₂ q #) and primary current detection value i ₁ (i
The actual speed estimation value ω _r # is estimated by using the speed adaptive secondary magnetic flux observer including the speed adaptive mechanism 7 which is calculated by the above equation (7) based on ₁ d, i ₁ q).

Then, the deviation of the power source angular frequency ω ₀ (ω ₀ = ω _r # + ω _s *) caused by the “delay” of the estimated value (ω _r #) in the process of estimating the actual motor speed (ω _r ). For correction, the motor secondary magnetic flux estimated value λ ₂ # (λ ₂ on the stator coordinates (dq axes) estimated by the same-dimensional magnetic flux observer 4
d #, λ ₂ q #) is coordinate-converted by the coordinate converter 10 into the motor secondary magnetic flux λ 2 # on the synchronous rotation coordinate axis (ab axis), and the torque axis component (λ _{2 2} ) of the motor secondary magnetic flux estimated value λ ₂ # is calculated. b #) is integrated by the slip angular frequency correction integrator 16 and the slip angular frequency correction value ω
_sc (ω _sc = Kω _s ∫λ ₂ b # · dt) is obtained and added by the adder 17 to the power supply angular frequency command value ω _s *.

The addition of the angular frequency correction value ω _sc to the power source angular frequency command value ω _s * is performed by correcting the deviation of the motor actual speed estimated value (ω _r #) with the slip angular frequency correction value ω _sc . , The deviation of the basic phase angle (θ ₀ ) due to the deviation of the power source angular frequency (ω ₀ ) determined by the equation (2) is prevented, the deviation of the coordinate axis is prevented, and complete vector control is established. Become.

[0030]

As described above, according to the present invention, the slippage of the coordinate conversion axis due to the "lag" for estimating the actual speed of the induction motor can be prevented only by adding the slip angular frequency correcting integrator. Therefore, the vector control of complete decoupling control can be realized.

[Brief description of drawings]

FIG. 1 is a configuration diagram of a control system that is an embodiment of the present invention.

FIG. 2 is a block diagram of a conventional control system

[Explanation of symbols]

 1: Induction motor 2: PWM control inverter 3: Digital current controller 4: Same-dimensional magnetic flux observer 5: Slip calculator 6: Speed controller 7: Speed adaptation mechanism 8: Coordinate converter 9: Coordinate converter 10: Coordinate conversion Device 11: Integrator for calculating basic phase angle 16: Integrator for correcting slip angle frequency

Claims (2)

[Claims] 1. A synchronous rotation coordinate of the induction motor (1), wherein speed control is performed based on a comparison error signal comparing a speed command value (ω _r *) of the induction motor (1) and an actual speed estimation value (ω _r #). A speed controller (6) for obtaining the above primary torque axis current command value (i ₁ b *), and a primary excitation axis current command value (i ₁ a *) on the synchronous rotation coordinates of the induction motor (1). And the primary torque axis current command value (i ₁ b *),
Primary excitation axis current detection value (i ₁ d) and primary torque axis current detection value
(i ₁ q) and the power source angular frequency (ω ₀ ) are input to perform current decoupling control of the induction motor, and the primary excitation axis voltage command value (via *) and the primary torque axis voltage on the synchronous rotation coordinate are input. A digital current controller (3) that outputs a command value (vib *), and a primary excitation axis voltage command value (v ₁ a *) on the synchronous rotation coordinate axis that is the output of the digital current controller (3) and The primary excitation axis voltage command value (v ₁ d *) and the primary torque axis voltage command value (v ₁ d *) on the stator coordinates of the induction motor (1), which is the coordinate conversion of the primary torque axis voltage command value (v ₁ b *). v ₁ q *) a power converter (2) that controls the speed of the induction motor, a primary excitation axis current detection value (i ₁ d) on the stator coordinates of the induction motor (1), and a primary torque axis current. The detected value (i ₁ q), the primary excitation axis voltage command value (v ₁ d *) and the primary torque axis voltage command value (v ₁ q *) on the stator coordinate, and the actual speed estimated value (ω _r #) and enter the stator Primary excitation axis current (i ₁ d) and primary torque axis current (i ₁ q) on the coordinates, and secondary excitation axis magnetic flux (λ ₂ d) and secondary torque axis magnetic flux (λ ₂ ) on the stator coordinates. q) the same dimension magnetic flux observer (4) and the primary current estimated value (i ₁ # (i ₁ d #, i ₁ q #) on the stator coordinate which is the output of the same dimension magnetic flux observer (4). ) Secondary magnetic flux estimation value (λ ₂ # (λ ₂ d #, λ ₂ q #)), primary excitation axis current detection value (i ₁ d) and primary torque axis current detection value (i
₁ q) is input to each of them, and the actual velocity estimation value (ω _r #) is output, and the secondary magnetic flux estimation on the stator coordinate which is the output of the same-dimensional magnetic flux obvard (4). The estimated value of the secondary torque axis magnetic flux on the synchronous rotation coordinates, which is the coordinate conversion of the value (λ ₂ # (λ ₂ d #, λ ₂ q #)) λ ₂ b #
And an integrator (16) that outputs a slip angular frequency correction value (ω _sc ) by integrating control, and a primary excitation axis current command value (i ₁ a *) on the synchronous rotation coordinate of the induction motor (1). And a primary torque axis current command value (i ₁ b *) are input to calculate and output a slip angular frequency command value (ω _s *) of the motor, and a slip calculator (5) 5) The slip angular frequency command value (ω _s *) which is the output of 5) is added with the slip angular frequency correction value (ω _sc ) which is the output of the integrator (16), and the actual speed of the induction motor (1) is increased. Estimated value (ω _r #) is added and the power supply angular frequency which is the control input of the digital current controller (3)
A speed sensorless vector control system for an induction motor, comprising: an adder (17) that outputs (ω ₀ ). 2. A basic phase angle (θ) for coordinate conversion between the synchronous rotation coordinate and the stator coordinate of the induction motor by time-integrating the power source angular frequency (ω ₀ ) which is the output of the adder (17). The speed sensorless vector control system for an induction motor according to claim 1, further comprising an integrator (11) for obtaining ₀ ).