JPH07123799A - Speed sensorless vector control system for induction motor - Google Patents

Abstract

PURPOSE:To control an induction motor to a speed coincident with its speed command value by estimating a speed of the motor by using a torque axis component of a secondary magnetic flux estimated value. CONSTITUTION:The speed sensorless vector control system for an induction motor comprises a minimum order magnetic flux observer 4 which inputs primary voltage command values V1d, V1q of stator coordinates of the motor 1, primary current detection values i1d, i1q and motor real speed estimated value omegar. The observer 4 estimates secondary magnetic fluxes lambda2d, lambda2q of stator coordinates, and estimates a motor real speed estimated value omegar by using secondary magnetic flux torque axis component lambda2b in synchronous rotary coordinates after coordinate-converting by a coordinate converter 10. Thus, even if the secondary magnetic flux estimated value has an initial error, the estimated value is converged to a correct value, and hence a real speed estimated value of the motor obtained from the magnetic flux estimated value can always estimate a real speed (true value).

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor vector control system, and more particularly to an induction motor vector control system which does not use a speed sensor.

[0002]

2. Description of the Related Art A vector control method has become popular as a high-performance speed control method for an induction motor, and a speed sensorless vector control method for controlling this without a speed sensor is known.

FIG. 3 shows the structure of a conventional induction motor speed sensorless vector control system control system.

In the conventional induction motor speed sensorless vector control system, in order to detect the speed of the induction motor without using the speed sensor, as shown in FIG. 3, the primary current (phase current) iu of the induction motor 1 is increased. , iv, iw are detected, the number of phases is converted by the three-phase to two-phase phase converter 14 to obtain the primary current i ₁ (i ₁ d, i ₁ q) on the stator coordinates (dq axes), and the magnetic flux is calculated. In the transformer 4, the secondary magnetic flux λ ₂ (λ ₂ d, λ ₂ q) of the induction motor 1 is calculated by the primary current i ₁ and the motor primary voltage command value v ₁ * (v ₁ d *, v ₁ q *). And the secondary magnetic flux λ ₂ (λ ₂ d, λ
The torque axis component λ ₂ b of the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) obtained by coordinate conversion of ₂ q) onto the synchronous rotation coordinate (ab axis) by the coordinate converter 10 is calculated by the proportional integrator 7 ( (See formula (7) below)
Then, the rotational angular frequency ω _r # is obtained to detect the speed ω _r of the induction motor 1.

Here, the estimation of the speed (rotational angular frequency ω _r ) of the induction motor 1 will be described. The voltage equation of the induction motor 1 is given by the following equation (1) when it is represented by the ab axis that observes various quantities from the synchronous rotation coordinate system that rotates at the power supply angular frequency (ω ₀ ).

[0006]

[Equation 2]

However, v ₁ a, v ₁ b ・ ・ ・ Primary excitation axis voltage and primary torque axis voltage (V) i ₁ a, i ₁ b on synchronous rotation coordinates (ab axis) Synchronous rotation coordinates ( primary excitation axis current on the ab axis), primary torque axis current (A) λ ₂ a, λ ₂ b ..... secondary excitation axis magnetic flux on the synchronous rotation coordinate (ab axis), secondary torque axis magnetic flux (Wb ) ω ₀ ‥‥‥‥‥‥ Power supply angular frequency (rad / sec) ω _r ‥‥‥‥‥‥ Motor speed (rotational angular frequency, rad / sec) ω _s ‥‥‥‥‥ Slip angular frequency (rad / sec) R ₁ 、 R ₂・ ・ ・ Primary and secondary resistance (Ω) L ₁ , L ₂ ‥‥‥ Primary and secondary inductance (H) M ‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥ ‥‥‥ Equivalent leakage inductance (H) (Lσ =
(L ₁ L ₂ −M ² ) / L ₂ ) s ············· Small molecule (d / dt) And power source angular frequency (ω ₀ ), motor speed (ω _r ), slip angular frequency (ω _s *) And slip angular frequency (ω
The calculation of _s *) is expressed by the following equation (2). ω ₀ = ω _r + ω _s * ω _s * = i ₁ b * / (i ₁ a * ・ τ ₂ ) ‥‥‥‥‥‥ (2) However, τ ₂ ‥‥‥‥‥‥‥‥‥‥ (Τ ₂ = L ₂ /
R ₂ ) Subscript (*) ..................... Indicates a command value or set value.
Now, i ₁ a * = constant ‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥ (3), and in the above equation (1), (2), (3 ), The primary voltage command values (v ₁ a *, v ₂ b *) of the synchronous rotation coordinate system (ab axis) are the constitutive expressions of the digital current controller 3 below (4). Given by the formula,

[0008]

[Equation 3]

As the primary current i ₁ on the synchronous rotation coordinate (ab axis), the current according to the command value i ₁ * (i ₁ a *, i ₁ b *) flows,
Secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on synchronous rotation coordinates (ab axis)
Is kept at λ ₂ a = Mi ₁ a (constant) λ ₂ b = 0 .................................. (5).

As a result, the torque (T) of the induction motor 1
Is

[0011]

[Equation 4]

Therefore, the decoupling irrelevant to the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) and the secondary current i ₂ (i ₂ a, i ₂ b) on the synchronous rotation coordinate (ab axis). Vector control of control is established.

By the way, as is apparent from the equation (2), when the velocity sensor is not used, the slip angular frequency ω _s
Even if is set, the motor speed ω _r is unknown,
Although the power source angular frequency ω ₀ cannot be determined, the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on the synchronous rotation coordinate (ab axis) that rotates at the power source angular frequency ω ₀ is (5) above. By controlling the power source angular frequency ω ₀ so as to satisfy the expression, similarly, vector control of decoupling control can be realized. That is, the secondary magnetic flux λ ₂ on the synchronous rotation coordinate (ab axis)
Estimate the actual motor speed ω _r based on the following equation (7) by the proportional integrator 7 that receives the torque axis component (λ ₂ b) of
_r #), the power source angular frequency ω ₀ is obtained by the above equation (ω ₀ = ω _r # + ω _s *), and the digital current controller 3 is controlled by the power source angular frequency ω ₀ to cause non-interference. It is possible to realize the vector control of the computerized control.

[0014] _{ω r # = K rp λ 2} b + K ri ∫λ 2 b · dt ‥‥‥‥‥‥‥‥‥‥‥‥ (7) K rp: proportional constant K _ri: integration constant subscript #: Estimated Represents a value. Then, 2 on the synchronous rotation coordinate (ab axis) in the above equation (7) for obtaining the actual motor speed (ω _r #)
The torque axis component (λ ₂ b) of the next magnetic flux λ ₂ is first transformed into the following equation (8) by modifying the first and second lines of the equation (1) which is the voltage equation.

[0015]

[Equation 5]

As shown in the following equation (9)

[0017]

[Equation 6]

Primary current i ₁ (i on the stator coordinates (dq axes)
₁ d, i ₁ q) and the secondary voltage λ _{2 of} the stator coordinates (dq axes) fixed to the motor stator from the primary voltage v ₁ (v ₁ d, v ₁ q)
(Λ ₂ d, λ ₂ q) can be obtained.

[0019] As described above, if not using the speed sensor, the stator coordinates for estimating the motor actual speed ω _r # (d-
For the magnetic flux calculation (magnetic flux calculator 4) of the secondary magnetic flux λ ₂ (λ ₂ d, λ ₂ q) on the q-axis), the primary current detection value i ₁ (i ₁ d, i ₁ q) and the primary By using the voltage command value v ₁ * (v ₁ d *, v ₁ q *) and considering the offset of the primary current detection value (i ₁ d, i ₁ q) as shown in the above equation (9), “ It is approximated using "first-order delay".

A conventional speed sensorless vector control system for an induction motor, in which the secondary magnetic flux calculator 4 is approximated by "first-order lag" as shown in the equation (9), will be described below. In FIG. 3 (control system configuration), the speed control unit (AS
R) is the speed command value ω _r * of the induction motor 1 and the estimated actual speed value ω
comparison error signal proportional integral control to synchronous rotating coordinate with _r # (a
-The speed controller 6 that obtains the primary torque axis current command value i ₁ b # on the (b-axis), and the primary torque axis current command value i ₁ b # and the primary excitation axis current command on the synchronous rotation coordinates (ab axis) It is composed of a slip calculator 5 which obtains a slip angular frequency command value ω _s * from the value i ₁ a *, and the current control unit (ACR) controls the PWM control inverter 2 to control the speed command value ω of the induction motor 1. It is composed of a digital current controller 3 that obtains a primary voltage command value v ₁ * (v ₁ a *, v ₁ b *) on the synchronous rotation coordinates (ab axis) for speed control to _r *. The speed control unit (ASR) and the current control unit (ACR) are controlled with different calculation cycles (sampling cycles).

FIG. 4 shows the internal structure of the digital current controller 3 in the current control unit (ACR). The digital current controller 3 interferes with the induction motor 1 for controlling the current decoupling. Compensation means (interference compensation term ω ₀ L ₁ , ω
₀ Lσ), the same detection value (i ₁ a) as the primary excitation axis current command value (i ₁ a *) on the synchronous rotation coordinate (ab axis), the motor primary torque axis current command value (i ₁ b) *) And the same detection value (i ₁ b) are compared and controlled so that they are equal (i ₁ a * = i ₁ a, i ₁ b * = i
₁ b) The primary voltage command value v ₁ * (v ₁ a *, v ₁ b on the synchronous rotation coordinate (ab axis) for controlling the PWM control inverter 2
*) Is calculated.

The operation in FIG. 3 (control system) will now be described. Now, when the motor speed command (ω _r *) is given to the speed controller (ASR), the speed command value (ω _r *) is given.
And the actual motor speed estimated value (ω _r #) are compared, the comparison error signal is proportional-integral (PI) controlled by the speed controller 6, and the primary current on the synchronous rotation coordinate (ab axis) of the induction motor 1 is calculated. It is converted to a primary torque axis current command value (i ₁ b *) which is a torque axis component of the command value i ₁ *. Next, the current controller (AC
In the digital current controller 3 (see FIG. 4) in R), the induction motor 1 that controls the PWM control inverter 2
Primary voltage command value v ₁ (v ₁ a on the synchronous rotation coordinate (ab axis) of
*, V ₁ b *) so that the primary current command value i ₁ * and the primary current detection value i ₁ are equal (i ₁ a * = i ₁ a, i ₁ b * = i ₁ b) , Which is a conditional expression that enables decoupling control (4)
It is calculated based on the formula. Then, the primary voltage command value v ₁ *
(V ₁ a *, v ₁ b *) was converted into the primary voltage command value v ₁ * (v ₁ d *, v ₁ q *) on the stator coordinates (dq axes) by the coordinate converter 9. After that, the number of phases is converted by the two-phase / three-phase converter 15 to be converted into the output voltage control command voltages Vu, Vv, Vw of each phase of the PWM control inverter 2 and the PWM control inverter 2 Controls the output voltage of each phase. As a result, the induction motor 1 is speed-controlled according to the desired speed setting value (ω _r *).

Further, the coordinate converter 8 for converting between the synchronous rotation coordinates (ab axis) rotating at the power source angular frequency ω ₀ and the stator coordinates (dq axes) fixed to the stator of the induction motor 1. The unit vector (Sin θ ₀ , Co used for 9 and 10)
The phase angle θ ₀ (θ ₀ = ω ₀ t) for creating sθ ₀ ) is
As shown in the above equation (2), the slip calculator 5 calculates the primary excitation current i ₁ a, the primary torque current i ₁ b, and the secondary time constant τ _{2 of the} induction motor ₁ on the synchronous rotation coordinate (ab axis). (= L ₂ / R
₂ ) Slip angular frequency command value (ω _s #)
And a motor actual speed estimated value (ω _r #) obtained by the above equation (7) by the secondary magnetic flux λ ₂ which is the calculation output of the magnetic flux calculator 4 approximated by “first-order delay” It can be obtained by integrating the angular frequency (ω ₀ ) by the integrator 11.

[0024]

As described above, in the conventional speed sensorless vector control system, since the magnetic flux calculator 4 is approximated by "first-order delay", the magnetic flux in the low speed range of the induction motor 1 is reduced. The integral calculation in the calculator 4 cannot be performed accurately, and the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on the synchronous rotation coordinate (ab axis) is estimated, that is, the estimated value of the actual motor speed (ω there is a problem that _r #) to the speed command value of the electric motor 1 will an error occurs (ω _r *) to not be able to control the matched speed.

The present invention provides a speed sensorless vector control system having an estimation unit for accurately estimating the speed of an induction motor.

[0026]

In order to solve the above problems, the present invention provides a method for estimating the speed (ω _r ) of an induction motor in a sensorless vector control system in which the induction motor is vector-controlled. Two-phase primary voltage command value on the stator coordinates (dq axes) which is the output of the digital current controller 3 in the current control unit (ACR) that performs vector control of decoupling control in order to estimate the secondary magnetic flux. v ₁ * (v ₁
d *, v ₁ q *), primary current detection value i ₁ (i ₁ d, i ₁ q), and the actual speed estimated value of the induction motor (ω _r #) Secondary magnetic flux λ on rotating coordinates
₂ (λ ₂ a, λ ₂ b) is estimated, and the speed (ω _r ) of the induction motor is calculated using the torque axis component (λ ₂ b #) of the secondary magnetic flux estimated value λ ₂ #.
Is estimated. The subscript “#” represents the estimated value.

The minimum dimension magnetic flux observer in the present invention will be described below.

The equation of state of the induction motor is given by the following equation (10) when various quantities are observed from the stator coordinates (dq axes).

[0029]

[Equation 7]

However,

[0031]

[Equation 8]

I ₁ (i ₁ d, i ₁ q): Primary current on the stator coordinates (dq axes)
(A) λ ₂ (λ ₂ d, λ ₂ q): Secondary magnetic flux on the stator coordinates (dq axes)
(wd) v ₁ (v ₁ d, v ₁ q): Primary voltage on the stator coordinates (dq axes)
_{(v) R 1, R 2} : 1 , second resistance _{(Ω) L 1, L 2} : 1 , second inductance (H) M: mutual (energized) inductance (H) Erushiguma: Equivalent leakage inductance (H ) s: Time fine molecule (d / dt) The output equation is the following equation (11).

[0033]

[Equation 9]

From this equation (11), the stator coordinates (dq
Since the primary current i ₁ on the (axis) can be directly observed, if only the secondary magnetic flux λ _{2 of} the stator coordinates (dq axes) is estimated by the observer, the estimated value of the secondary magnetic flux λ ₂ is “λ ₂
The observer is composed of the following equation.

Sλ ₂ # = A ₂₁ i ₁ + A ₂₂ λ ₂ # + G [si ₁ − (A ₁₁ i ₁ + A ₁₂ λ ₂ # + B ₁ v ₁ )] (12) However, observer gain (G) Is adjusted by the estimated actual speed (ω _r #) of the induction motor. Further, the above equation (12) is modified to obtain the following equation (13).

Sλ ₂ # = (A ₂₂ −GA ₁₂ ) λ ₂ # + (A ₂₁ −GA ₁₁ ) i ₁ −GB ₁ v ₁ + Gsi ₁ (13) However, s: time fine molecule (d / dt) Then, the equation (13) is expressed in a block diagram as shown in the configuration diagram of FIG. 5, and this is the configuration of the minimum-dimensional magnetic flux observer.

[0037]

EXAMPLES Examples of the present invention will be described below.

1 and 2 show the configuration of a control system which is an embodiment of the present invention. Of the configuration of the control system shown in FIGS. 1 and 2, the same parts as the configuration of the conventional control system shown in FIG. 3 are the same in the operation, and therefore the description thereof will be omitted.

[Embodiment 1] -See FIG. 1 When the induction motor 1 is controlled by the PWM control inverter 2, when the vector control of the current decoupling control is established,
As is clear from the equations (5) and (6), the synchronous rotation coordinate (a
The torque axis component (λ ₂ b) of the secondary magnetic flux λ 2 on the −b axis) becomes zero.

Therefore, in order to establish the vector control of the current decoupling control, the secondary magnetic flux torque axis component (λ ₂ b) on the synchronous rotation coordinate (ab axis) based on the above equation (7).
Is calculated and the actual motor speed value (ω _r ) is estimated so that the secondary magnetic flux torque axis component (λ ₂ b) becomes zero. However, as is clear from the equation (7), Since the speed estimation value (ω _r #) is obtained by proportional integration (PI) of the secondary magnetic flux torque axis component (λ ₂ b), the motor actual speed estimation value (ω _r #) is calculated as the actual speed value ( In order to make it equal to ω _r ), the secondary magnetic flux torque axis component (λ ₂ b) must be accurate.

In this embodiment, the "minimum dimension magnetic flux observer" shown in FIG. 5, that is, the primary voltage command value v ₁ * (v ₁ d *, v) on the stator coordinates (dq axes) of the induction motor 1 is used. ₁ q
*), Primary current detector i ₁ (i ₁ d, i ₁ q) and motor actual speed estimated value (ω _r #) are used as inputs, and fixed by the minimum dimension observer 4. The secondary magnetic flux λ ₂ # (λ ₂ d #, λ ₂ q #) on the child coordinates (dq axes) is estimated, and the synchronous rotation coordinates (ab
The secondary magnetic flux torque axis component (λ ₂ b) on the shaft) is substituted into the above equation (7) to estimate the actual motor speed value (ω _r ).

[Embodiment 2] -See Fig. 2, The voltage equation of the induction motor 1 is obtained by observing various quantities from the synchronous rotation coordinates (ab axis) rotating at the power supply angular frequency (ω ₀ ). Given in.

Further, the slip angular frequency of the induction motor 1 (ω
_s ), the power source angular frequency (ω ₀ ) and the actual speed (ω _r ) of the induction motor 1 are as shown in the above equation (2).

The lower two equations of the above equation (1) and the above equation
When the slip angular frequency (ω _s ) is obtained from the equation (2), the following equation is obtained.

[0045]

[Equation 10]

However, s: represents a time fine molecule. As is clear from this equation, the synchronous rotation coordinate (a-
The secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on the b axis), their differential values (sλ ₂ a, sλ ₂ b) and the primary current i ₁ (i on the same coordinate (ab axis) ₁ a, i ₁ b) is known, the slip angular frequency (ω _s )
Is required.

When the slip angular frequency (ω _s ) is obtained, the actual speed of the induction motor 1 is calculated by subtracting the slip angular frequency (ω _s ) from the power source angular frequency (ω ₀ ) based on the equation (2). The value (ω _r ) can be obtained.

However, in order to obtain the slip angular frequency (ω _s ) by the above equation, the secondary magnetic flux λ ₂ (λ ₂ a, on the synchronous rotation coordinate (ab axis) of the induction motor 1 that cannot be detected is detected. We must know λ ₂ b).

In this embodiment, the "minimum dimension magnetic flux observer" described above, that is, the primary voltage command value v ₁ * (v ₁ d *, on the stator coordinate (dq axis) of the induction motor 1 is used. v ₁
q *), the primary current detection value i ₁ (i ₁ d, i ₁ q), and the actual motor speed estimation value (ω _r #) are used as input, and the minimum dimension magnetic flux observer 4 is used. The secondary magnetic flux λ on the stator coordinate (dq axis) of the induction motor 1 by the observer 4.
₂ (λ ₂ d, λ ₂ q) is estimated, coordinate transformation (10) is performed, and the secondary magnetic flux λ ₂ (λ ₂ a, λ ₂ b) on the synchronous rotation coordinate (ab axis) and its differential value sλ ₂ (Sλ ₂ a, sλ ₂ b) is obtained.

The secondary magnetic flux differential value sλ ₂ (sλ ₂ a,
sλ ₂ b) is the secondary magnetic flux estimated value λ ₂ # on the synchronous rotation coordinate (ab axis) in the differentiator 7, as shown in the following equation.
Sampling time T _s which is the calculation cycle of the current control unit (ACR) by subtracting the previous value from the current value of (λ ₂ a #, λ ₂ b #)
It can be obtained by dividing by.

Sλ ₂ # = [λ ₂ # (n) −λ ₂ # (n-1)] / T _s ‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥‥ These estimated synchronous rotation coordinates (ab axis) Slip the above secondary magnetic flux estimation value (λ ₂ #) and its differential value (sλ ₂ #), and the primary current detection value i ₁ (i ₁ a, i ₁ b) on the synchronous rotation coordinate (ab axis). in calculator 16, as shown in the above equation to calculate the estimated value omega _s # slip angular frequency (omega _s), slip angular frequency estimate value (ω _s #) is subtracted from the power source angular frequency omega ₀ of the time To estimate the actual motor speed (ω _r ).

[0052]

As described above, according to the present invention, the induction motor is used as a magnetic flux observer for estimating the secondary magnetic flux of the induction motor that cannot be detected in order to estimate the actual speed of the induction motor without using the speed sensor. Stator coordinates of 1 (dq
Axis) primary voltage command value v ₁ * (v ₁ d *, v ₁ q *) and primary current detection value i ₁ (i ₁ d, i ₁ q), and estimated motor speed (ω _r By inputting #) and using the "minimum dimension magnetic flux observer" expressed by the above equation (13),
Even if the secondary magnetic flux estimated value has an initial error, the secondary magnetic flux estimated value converges to a correct value. Therefore, the actual speed estimated value of the induction motor obtained from the secondary magnetic flux estimated value is always the actual speed (true Value) can be estimated.

[Brief description of drawings]

FIG. 1 is a configuration diagram of a control system that is an embodiment of the present invention.

FIG. 2 is a configuration diagram of a control system that is another embodiment of the present invention.

FIG. 3 is a block diagram of a conventional control system

FIG. 4 is a block diagram of a digital current controller in the control system.

[Fig. 5] Configuration diagram of the minimum-dimensional magnetic flux observer

[Explanation of symbols]

1: Induction motor 2: PWM control inverter 3: Digital current controller 4: Magnetic flux calculator, minimum dimension magnetic flux observer 5: Slip calculator 6: Speed controller 7: Proportional integrator ω ₀ : Power source angular frequency ω _r : Electric motor Actual speed (rotational angular speed) ω _s : Motor slip angular frequency

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[Procedure amendment]

[Submission date] January 6, 1994

[Procedure Amendment 1]

[Document name to be corrected] Drawing

[Name of item to be corrected] Figure 2

[Correction method] Change

[Correction content]

[Fig. 2]

Claims (3)

[Claims] 1. A synchronous rotation coordinate of the induction motor (1), wherein speed control is performed based on a comparison error signal comparing a speed command value (ω _r *) of the induction motor (1) and an actual speed estimation value (ω _r #). A speed controller (6) for obtaining the above primary torque axis current command value (i ₁ b *), and a primary excitation axis current command value (i ₁ a *) on the synchronous rotation coordinates of the induction motor (1). And the primary torque axis current command value (i ₁ b *),
Primary excitation axis current detection value (i ₁ d) and primary torque axis current detection value
(i ₁ q) and the power source angular frequency (ω ₀ ) are input to perform current decoupling control of the induction motor, and the primary excitation axis voltage command value (v ₁ a *) on the synchronous rotation coordinate and the primary torque Axis voltage command value (v ₁ b
Digital current controller (3) that outputs *), the primary excitation axis voltage command value (v ₁ a *) on the synchronous rotation coordinates that is the output of the digital current controller (3), and the primary torque axis voltage The induction motor is converted by the primary excitation axis voltage command value (v ₁ d *) and the primary torque axis voltage command value (v ₁ q *) on the stator coordinates of the induction motor, which is the coordinate conversion of the command value (v ₁ b *). A power converter (2) for controlling the speed of the induction motor, a primary excitation axis current detection value (i ₁ d) and a primary torque axis current detection value (i ₁ q) on the stator coordinates of the induction motor (1), Input the primary excitation axis voltage command value (v ₁ d *) and primary torque axis voltage command value (v ₁ q *) on the stator coordinates, and the estimated actual speed value (ω _r #) to the induction motor ( Secondary magnetic flux (λ ₂ ) on the stator coordinate of 1)
Of the secondary magnetic flux observer (4) for estimating the value of the secondary magnetic flux estimated value (λ ₂ #) on the stator coordinate of the induction motor (1), which is the output of the minimum dimension magnetic flux observer (4). A proportional integrator (7) that obtains the actual speed estimated value (ω _r #) by proportionally integrating the secondary torque axis magnetic flux estimated value (λ ₂ b #) on the synchronous rotation coordinate of the electric motor (1); A speed sensorless vector control method for an induction motor, which is characterized in that 2. The synchronous rotation coordinates of the induction motor (1) are controlled on the basis of a comparison error signal comparing a speed command value (ω _r *) of the induction motor (1) and an actual speed estimation value (ω _r #). Top one
Speed controller (6) to obtain the next torque axis current command value (i ₁ b *)
And a primary excitation axis current command value (i ₁ a *) and a primary torque axis current command value (i ₁ b *) on the synchronous rotation coordinate of the induction motor (1), and a power source angular frequency (ω ₀ ). To output the primary excitation axis voltage command value (v ₁ a *) and the primary torque axis voltage command value (v ₁ b *) on the synchronous rotation coordinate by performing current decoupling control of the induction motor. A current controller (3), a primary excitation axis voltage command value (v ₁ a *) and a primary torque axis voltage command value (v ₁ on the synchronous rotation coordinates which are outputs of the digital current controller (3). (b *) coordinate-converted induction motor on the stator coordinates primary excitation axis voltage command value (v ₁ d *) and primary torque axis voltage command value (v ₁ q *) to perform electric power control
A power converter (2) for speed control of (1), a primary excitation axis current detection value (i ₁ d) and a primary torque axis current detection value (i ₁ ) on the stator coordinates of the induction motor (1). q), primary excitation axis voltage command value (v ₁ d *) and primary torque axis voltage command value (v ₁ q *) on the stator coordinates, and actual speed estimated value (ω _r #) are input respectively. Secondary magnetic flux (λ ₂ ) on the stator coordinate of the induction motor (1)
Of the secondary magnetic flux observer (4) for estimating the value of the secondary magnetic flux estimated value (λ ₂ #) on the stator coordinate of the induction motor (1), which is the output of the minimum dimension magnetic flux observer (4). Secondary magnetic flux (λ ₂ on the synchronous rotation coordinate of the motor (1)
Differentiator (7) for differentially controlling #) to obtain a secondary magnetic flux estimation differential value (sλ ₂ #) on the synchronous rotation coordinate axis of the induction motor (1), and a secondary magnetic flux estimated value (s) on the synchronous rotation coordinate ( λ ₂ #), the secondary magnetic flux differential estimated value (sλ ₂ #) output from the differentiator (7), and the primary current detection value on the synchronous rotation coordinate of the induction motor (1).
(i ₁ ) as an input, a slip calculator (16) for calculating the slip angular frequency (ω _s ) of the motor, and the slip angular frequency (ω _s ) and the power supply angular frequency (ω ₀ ) are added to obtain the actual value. A speed sensorless vector control system for an induction motor, comprising: an adder that obtains an estimated speed value (ω _r #). 3. The minimum-dimensional magnetic flux observer (4) is an observer gain controlled by an actual speed estimation value (ω _r #).
The speed sensorless vector control system for an induction motor according to claim 1 or 2, which comprises the following constitutive equation including (G). sλ ₂ # = (A ₂₂ −GA ₁₂ ) λ ₂ # + (A ₂₁ −GA ₁₁ ) i ₁ −GB ₁ v ₁ + Gsi ₁ where, i ₁ (i ₁ d, i ₁ q): Primary current on the stator coordinates (dq axes)
(A) λ ₂ (λ ₂ d, λ ₂ q): Secondary magnetic flux on the stator coordinates (dq axes)
(wd) v ₁ (v ₁ d, v ₁ q): Primary voltage on the stator coordinates (dq axes)
_{(v) R 1, R 2} : 1 , second resistance _{(Ω) L 1, L 2} : 1 , second inductance (H) M: mutual (energized) inductance (H) Erushiguma: Equivalent leakage inductance (H ) s: Time fine molecule (d / dt) G: Observa bagin Subscript #: Represents an estimated value.