JPH0570394B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to an induction motor control device that performs variable speed control of an induction motor by vector control.

(Prior Art) As power converters such as inverters and cycloconverters have become easier to construct due to the remarkable development of power semiconductor devices, induction motors, which were often used as constant-speed motors in the past, In recent years, electric motors have been increasingly used as variable speed electric motors.

As a variable speed control method for an induction motor, a vector control method is often adopted because it provides excellent response characteristics. This vector control method includes a magnetic flux detection vector control method that detects the secondary magnetic flux as a vector quantity and uses it as a control signal for the primary current, and a slip frequency vector control method that calculates and controls the magnetic flux vector based on motor constants. It has been known.

The conventional slip frequency type vector control device based on the latter, as illustrated in FIG. 4, calculates a vector control command by inputting a secondary magnetic flux command Φ ^* ₂ and a torque command τ ^* and outputting a primary current command I ^* ₁ . A circuit 10, a power converter 2 that controls the primary current I1 so that the induction motor ₃ rotates at a predetermined speed and torque based on the primary current command I ^* ₁ , and a power converter 2 that controls the rotational speed ω _R of the induction motor 3. It is composed of a speed detector 4 for detecting the speed.

The vector control command calculation circuit 10 calculates the secondary magnetic flux command Φ ^* ₂ and the torque command τ ^* according to the transfer characteristics of the induction machine 3, and calculates the real component command of the primary current, that is, the magnetizing current component command i ^* _1R , and the imaginary number. It includes a division element, a multiplication element, a constant element, a differential element, an addition element, etc. so as to output a component command, that is, a torque current component command i ^* _1I , and a slip frequency command _ω ^* S.
In order to obtain the magnetizing current component command i ^* _1R , a constant element 11, a constant element 12, a differential element 13, and an addition element 14 are provided. In order to obtain the torque current component command i ^* _1I , a division element 15 and a constant element 16 are used.
It is equipped with Further, in order to obtain the slip frequency command ω ^* _S , a division element 17, a constant element 18, and an addition element 19 are provided. Addition element 19 adds slip frequency command ω ^* _S from constant element 18 and rotational frequency ω _R obtained by speed detector 4 to obtain slip frequency command ω ^* _S . This slip frequency command ω ^* _S is generated by the vector generator 20,
It is converted into a unit vector indicating the predicted position of the secondary magnetic flux. The secondary self-inductance L ₂ , secondary resistance R ₂ , and mutual inductance M of the induction motor 3 are used as the constant elements, and these are set before the device is operated.

The magnetizing current component command i ^* _1R and the torque current component command i ^* _1I are vector-synthesized in a vector composition circuit 21 to form a current command i ^* ₁ =i ^* _1R +ji ^* _1I . Multiplier 22 multiplies this current command i ^* ₁ and the above unit vector.
A primary current command I ^* ₁ is formed by multiplying by , and the output current of the power converter 2, that is, the primary current of the induction motor 3 is controlled according to this primary current command I ^* ₁ .

FIG. 5 illustrates a magnetic flux detection type vector control device. In this method, the vector control command calculation circuit 30 calculates the voltages based on the phase voltages v _al , v _bl , v _cl and the primary currents i _al , i _bl , i _cl of the induction motor 3 without providing a magnetic flux detector. It is calculated by the magnetic flux calculator 31. The output of the magnetic flux calculator 31 gives the stationary two-axis components Φ〓 _l , Φ〓 ₁ of the secondary magnetic flux vector, and the vector analyzer 32 generates the absolute value component |Φ ₂ | and the signal sinφ, which indicates the phase φ.
Converted to cosφ.

On the other hand, the secondary magnetic flux command Φ ^* ₂ is obtained by the magnetizing current calculator 33
The torque command τ ^* is converted into a torque current command i ^* _q1 by the torque current calculator ₃₄ using also the secondary magnetic flux command ^{Φ *} ₂ . Magnetizing current command i ^* _d1 and torque current command
i ^* _q1 are the magnetizing current command and torque current command in the rotating coordinate system, respectively, with the rotating magnetic flux aligned with the d-axis, and the current command in the stator coordinate system is generated by the vector rotator 35 based on the phase Φ. Converted to i〓 ^* ₁ , i〓 ^* ₁ . The current commands i〓 ^* ₁ , i〓 ^* ₁ obtained as the output of the vector rotator 35 are the current commands for creating the secondary magnetic flux components Φ〓 ₁ , Φ〓 ₁ , and this is the current command for the two-phase to three-phase converter. 36, the three-phase current command i ^* _q1 ,
Converted to i ^* _b1 , i ^* _c1 . The output current of the power converter 2, that is, the three-phase primary current of the induction motor 3 i _al , i _bl , i _cl
are added to the three-phase current commands i ^* _a1 , i ^* _B1 , i ^* _C1 outputted from the two-phase to three-phase converter 36 in adders 37, 38, and 39, so that the deviation between the two becomes zero. Power converter 2 is controlled.

(Problems to be Solved by the Invention) In the slip frequency type vector control device shown in Fig. 4, the magnetizing current component command i ^* _1R , the torque current component command i ^* _1I , and the slip frequency command ω ^* _S are calculated. In this process, the secondary resistance R ₂ is directly involved. However, since the secondary resistance R ₂ changes as the rotor temperature rises, a corresponding difference occurs between the calculated value and the actual value, and the response of the main magnetic flux changes, causing the calculated result of the slip frequency command ω ^* _S to change. This has the disadvantage that the vector control characteristics cannot be maintained.

In the magnetic flux detection type vector control device shown in Fig. 5, it is not affected by the secondary resistance, but what is obtained by calculation is the secondary magnetic flux when the secondary leakage inductance is ignored, that is, the air gap magnetic flux. However, it has the disadvantage that it involves an error corresponding to the secondary leakage inductance. However, in the case of a magnetic flux detection type vector control device, even if the constants of the primary circuit and secondary circuit of the induction motor change,
This is interpreted as a change in the input voltage v _al , v _bl , v _cl and input current i _al , i _bl , i _cl to the magnetic flux calculator, and the magnetic flux calculation result changes accordingly, so vector control based on parameter changes is possible. There is little deterioration in characteristics.

From a mounting perspective, installing a magnetic flux detector has many problems with the accuracy and resolution of the sensor.
When a magnetic flux calculator is used, there is a problem in that calculation accuracy decreases due to voltage distortion, especially at low speeds, and as a result, it is difficult to obtain sufficient starting torque.

Therefore, the present invention provides a control device for an induction motor that is minimally affected by changes in the secondary resistance R ₂ and can generate torque according to the command value even during starting. The purpose is to provide.

[Structure of the invention]

(Means for Solving the Problems) The present invention provides magnetizing current calculation means for obtaining a direct axis magnetizing current command by dividing a given secondary magnetic flux command by the mutual inductance of an induction machine, and a torque current calculation means for calculating a torque current command on a horizontal axis perpendicular to the direct axis based on the quotient obtained by dividing the torque command obtained by dividing the torque command by the secondary magnetic flux command; a slip frequency calculation means for calculating a slip frequency command based on a quotient obtained by dividing the product of In an induction machine control device comprising means for forming a primary current command based on a phase corresponding to a frequency command, and means for controlling the primary current of the induction machine in accordance with the primary current command, the actual A detection calculation means for calculating the actual direct axis secondary magnetic flux and horizontal axis secondary magnetic flux based on the voltage, current and induction machine constant, and a secondary magnetic flux command and the direct axis secondary magnetic flux detected by the detection calculation means. means for determining a deviation from the magnetizing current, means for determining a magnetizing current correction signal according to the magnitude of the deviation, and means for reducing a direct axis secondary magnetic flux error by adding the magnetizing current correction signal to the magnetizing current command; means for obtaining a deviation between the horizontal axis magnetic flux command set to approximately zero and the horizontal axis secondary magnetic flux detected by the detection calculation means; and means for obtaining a slip frequency correction signal according to the magnitude of the deviation; The present invention is characterized in that it includes means for reducing a slip frequency error by adding a slip frequency correction signal to a slip frequency command.

(Function) According to the control device of the present invention, instead of simply performing predictive control based on motor constants as in conventional slip frequency vector control, each magnetic flux on the vertical axis and the horizontal axis is maintained at a predetermined value. By configuring a control loop that does this, even if the motor constants, i.e., secondary resistance, etc. fluctuate, the magnetic flux control loop maintains normal vector control and maintains torque as per the command value, including during startup. can be generated.

(Example) FIG. 1 shows an embodiment of the present invention.

The main circuit includes a power converter 2 and an induction motor 3. Although a cycloconverter (C/C) is symbolically shown here as the power converter 2, this can also be replaced by a rectifier and an inverter. The rotational speed of the induction motor 3 is detected by a speed detector 4. Further, current detectors 40, 41, 42 are provided to detect the primary currents i _al , i _bl , i _cl of the induction motor 3, and the detected primary currents i _al , i _bl , i _cl
and the input voltage v _al of the induction motor 3 which is separately derived,
v _bl and v _cl are introduced into the control device 50 as detected values. A secondary magnetic flux command, that is, a direct axis magnetic flux command Φ ^* ₂ and a torque command τ ^* are also introduced into the control device 50.

The major difference between the present invention and the conventional concept lies in how the magnetizing current command i ^* _d1 and the slip frequency command ω ^* _S are created, and as described below, both use a combination of feedback control and feedback control. It's where you are.

Regarding the magnetizing current command i ^* _d1 , it has conventionally been thought that once the direct axis magnetic flux command Φ ^* _d2 is determined, the corresponding magnetizing current command i ^* _d1 is also uniquely determined. However, as a practical matter, when the mutual inductance M changes due to magnetic saturation, it directly affects the calculation of the magnetizing current command i ^* _d1 from the relationship M・i ^* _d1 = Φ ^* _d2 ... (1) . Therefore, even if calculations such as equation (1) are simply performed, there is no guarantee that the actual magnetic flux Φ _d2 of the motor is generated according to the command value. Therefore
Subtractor 51 adds magnetizing current command i ^* _d10 obtained from equation (1).
A corrected magnetizing current command i ^* _d1 is obtained by adding, by an adder 53, a magnetizing current correction signal Δi ^* _d1 formed by passing the magnetic flux deviation Φ ^* _d2 −Φ _d2 obtained in the above through a controller 52. That is, i ^* _d1 = i ^* _d10 + Δi ^* _d1 (2) The same can be said about the slip frequency command ω ^* _S. According to the conventional way of thinking, the torque current is calculated from the torque command τ ^* by i ^* _q1 = (L ₂ /M)・(τ ^* /Φ ^* _d2 ) ...(3) in the torque current calculator 34, and then the The frequency command ω ^* _S is given by ω ^* _S = (M/L ₂ )・(R ₂・Φ ^* _d2 )・(i ^* _q1 ) (4). Assuming that as a result of the above direct axis magnetic flux control, the actual magnetic flux Φ _d2 also matches the magnetic flux command Φ ^* _d2 .
From equation (3), torque current command i ^* _q1 does not include any error. However, if the secondary resistance R ₂ fluctuates,
From equation (4), the slip frequency command ω ^* _S includes an error corresponding to the variation in the secondary resistance R ₂ , making it impossible to give a correct slip frequency command. The reason for this is that even though we defined the position of the secondary magnetic flux as the vertical axis and the position orthogonal to it, that is, the position where the magnetic flux is zero, as the horizontal axis, and derived the relationship in equation (4), This is not guaranteed in most electric motors. In order to solve this problem, in the present invention, the slip frequency given by equation (4) is calculated by the slip frequency calculator 55, and its output is set as ω ^* _S0 , and further, the horizontal axis magnetic flux command Φ ^* _q2 is set as A slip frequency correction signal Δω ^* _S is obtained via the controller 56 so that the actual horizontal axis magnetic flux Φ _q2 follows the command value, and this is added to the above-mentioned ω ^* _S0 by the adder 57 to calculate the actual value. Create the slip frequency command ω ^* _S. That is, ω ^* _S = ω ^* _S0 + Δω ^* _S (5) The control device shown in FIG. 1 is constructed based on the above idea.

In the control device shown in Fig. 1, the slip frequency command ω ^* _S is finely adjusted by the feedback system so that the position of the secondary magnetic flux is on the vertical axis and the position orthogonal to it is on the horizontal axis, and the secondary magnetic flux, that is, the vertical axis magnetic flux is A closed loop system is configured such that the size Φ _d2 also matches the command value Φ ^* _d2 . Although basic vector control can be realized with such a feedback system, the magnetizing current calculator 33 and the slip frequency calculator 55 are used together in order to improve the quick response during transient times.

The magnetic flux Φ _d2 used to obtain the magnetizing current correction signal Δi ^* _d1 and the magnetic flux Φ _q2 used to obtain the slip frequency correction signal Δω ^* _S are calculated by the magnetic flux calculator 31. This magnetic flux calculator 31 will be described later.

When the primary current commands i ^* _d1 , i ^* _q1 and slip frequency command ω ^* _S are given by equations (2), (3), and (5), the subsequent vector control calculations are the same as the conventional concept. That is, the vector converter 54 converts the direct axis current command i ^* _d1 and the horizontal axis current command i ^* _q1 into the absolute value i ^* ₁ and phase θ ^* of the primary current on the rotation coordinate. On the other hand, an adder 58 adds the slip frequency command ω ^* _S and the rotor angular frequency _ωr , and an integrator 59 calculates the phase Φ ^* . This phase Φ ^* indicates the position of the magnetic flux (d-axis position) as seen on the two stationary axes, and the sum of the phase Φ ^* and the phase θ ^* , that is, θ ^* ₁ = Φ ^* + θ ^*, is the position of the magnetic flux on the two stationary axes. is the phase of the primary current i ^* ₁ seen. The primary current i ^* ₁ e ^j 〓 ^* ₁ displayed in polar coordinates is converted into three phases by the two-phase/three-phase converter 36, and the phase current commands for each phase of a, b, and c are obtained as i ^* _a1 , i ^* _b1 , i ^* Make _c1 , current detector 40~4
Subtractors 37, 38, and 39 obtain the deviations from the primary currents i _al , i _bl , and i _cl detected in step 2, and control power converter 2 so as to make the deviations zero.

Next, a method of calculating or estimating the magnetic fluxes Φ _d2 and Φ _q2 will be explained.

Although it is possible to attach a magnetic flux sensor to the stator of the electric motor to detect the magnetic flux of the two stator axes (α-axis and β-axis), the magnetic flux can be obtained by calculation or estimation without attaching a magnetic flux sensor. The magnetic flux calculator 31 shown in FIG. 1 is provided to perform such a function. This magnetic flux calculator 31 will be explained in more detail with reference to FIGS. 2 and 3.

FIG. 2 shows the internal configuration of the magnetic flux calculator 31. The three-phase voltage v _al supplied to the induction motor 3,
Signals representing v _bl and v _cl are sent to the three-phase/two-phase converter 60,
Further, signals representing three-phase currents i _al , i _bl , and i _cl are respectively introduced into a three-phase/two-phase converter 61 and converted into a stator two-axis (α-axis, β-axis) system using a well-known method. The voltage and current on the α-axis and β-axis after conversion are respectively v〓 _l ,
Expressed as v〓 _l and i〓 _l , i〓 _l , the α-axis component of the secondary magnetic flux Φ
〓 _l
and β-axis component Φ〓 _l are calculated as follows.

Φ〓 _l =∫(V〓 ₁ −i〓 ₁・R ₁ − ₁ (di〓 ₁ /dt)) dt
……(6) Φ〓 ₁ =∫(v〓 ₁ −i〓 ₁・R ₁ − ₁ (di〓 ₁ /dt)) dt
...(7) The calculations of equations (6) and (7) are performed by the integrator 62. Note that R ₁ is the primary resistance, and ₁ is the primary leakage inductance.

Furthermore, the output of the integrator 62 is
3 to obtain the composite magnetic flux Φ ₂ , and on the other hand, it is applied to the two-phase (αβ-axis)/two-phase (dq-axis) converter 64 together with the composite magnetic flux Φ ₂ . The converter 64 converts the magnetic flux onto two rotating axes (dq axes) that rotate at a synchronous speed with the magnetic flux on the two stator axes, and performs matrix calculations in the following form.

Φ _d2 Φ _q2 = cosΦ ^* sinΦ ^* −sinΦ ^* cosΦ ^* Φ〓 ₁ Φ〓 ₁ (8) where Φ ^* is the output phase from the integrator 59.

The direct axis magnetic flux Φ _d2 obtained in this way is the first
It is used as a feedback signal for the direct axis magnetic flux control system in the figure, and is compared with the direct axis magnetic flux command Φ ^* _d2 in the subtracter 51 to perform the predetermined control operation described above. Further, the horizontal axis magnetic flux Φ _q2 is combined with the horizontal axis magnetic flux command (usually zero) Φ ^* _q2 by a subtracter 48 in FIG. 1, and the slip frequency command is corrected according to the deviation.

FIG. 3 is an explanatory diagram when the motor system 70 is expressed by a state equation and the magnetic fluxes Φ _d2 and Φ _q2 are estimated using a state observation device. It is assumed that the electric motor system creates an equation of state on the rotating coordinate (d, q) axes. The inputs are voltages u ₃ = v _dl and u ₄ = v _ql ,
The state variables are direct axis magnetic flux x ₁ = Φ _d2 , horizontal axis magnetic flux x ₂ = Φ _q2 ,
It is assumed that the direct axis component of the primary current x ₃ = i _dl and the horizontal axis component x ₄ = i _ql . At this time, the state equation and output equation are expressed as follows.

x・₁ x・₂ x・₃ x・₄ =A ₁₁ A ₁₂ A _{21 A 22 x 1} x ₂ x ₃ x ₄ +1/σL ₁ o o u ₃ u ₄ ...(9) However, σ is the leakage count , L ₁ is the primary self-inductance.

y=0010 0001x ₁ x ₂ x ₃ x ₄ ...(10) Elements of this matrix A ₁₁ , A ₁₂ , A ₂₁ , A ₂₂
is given by the constant of the motor. As can be seen from the output equation expressed by equation (10), the currents i _dl (=x ₃ ), i _ql (=
x ₄ ) can be detected, but the secondary magnetic flux Φ _d2
(=x ₁ ) and Φ _q2 (=x ₂ ) cannot be detected, so they are estimated by a state observation device having the configuration shown in FIG. The state observer is a method often applied in the field of modern control theory, and it is roughly composed of the following ideas.

In equation (9), x ₃ and x ₄ are detected, so if we consider these as inputs and rewrite the state equation regarding x ₁ and x ₂ , we get x・₁ x・₂ = [A ₁₁ ] x ₁ x ₂ + _[ A _{12 ] X 3} The right half of Figure 3 shows the state observer configured with the new variable for x ₁ as Z ₁ and the new variable for x ₂ as Z ₂ , and the integrator 7
3. It is equipped with a feedback matrix A^ and a coefficient matrix B^. (11) Instead of using equation (11) as is, we create a new variable Z.
The reason for introducing constants A^ and B^ using an integrator with respect to Just think that you did. If Z converges, it means that Z ₁ and Z ₂ have estimated the quantities related to the variables x ₁ and x ₂ , and when passed through the constant matrix C^, the estimated value of the direct axis magnetic flux x^ ₁ and the horizontal axis magnetic flux The estimated value x^ ₂ is found. The d-axis and q-axis components of the primary current i _dl ,
Since i _ql has been detected, it is output as is after level adjustment using matrix D^.

Since the direct axis component Φ _d2 (=＾x ₁ ) and the horizontal axis component Φ _q2 (=＾x ₂ ) of the secondary magnetic flux can be calculated by using the state observation device shown in Fig. 3, as already explained, It can be used as a feedback signal for magnetic flux control in FIG.

〔Effect of the invention〕

As described above, the present invention provides rotating coordinates (d, q
A feedback loop of direct-axis magnetic flux and horizontal-axis magnetic flux is formed on the shaft), and a slip frequency command determined from the direct-axis magnetic flux command and torque command is given in a feedback manner to improve quick response. With this feature, it is possible to provide an induction machine control device that can automatically correct fluctuations in parameters, especially secondary resistance, and realize highly accurate vector control.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing one embodiment of the present invention;
2 is a block diagram showing an example of the configuration of the magnetic flux calculator in FIG. 1; FIG. 3 is a block diagram of a magnetic flux estimator used as an alternative to the magnetic flux calculator in FIG. 2;
FIG. 4 is a block diagram showing an example of the configuration of a conventional control device, and FIG. 5 is a block diagram showing an example of the configuration of another conventional control device. 2...Power converter, 3...Induction motor, 4...
Speed detector, 31... Magnetic flux calculator, 33... Magnetizing current calculator, 34... Torque current calculator, 35...
...vector rotator, 36...2-phase/3-phase converter,
52, 56...Controller, 59...Integrator.

Claims (1)

[Scope of Claims] 1. Magnetizing current calculation means for obtaining a direct axis magnetizing current command by dividing a given secondary magnetic flux command by the mutual inductance of an induction machine; a torque current calculating means for calculating a torque current command on a horizontal axis orthogonal to the direct axis based on the quotient obtained by dividing by the magnetic flux command; a slip frequency calculation means for calculating a slip frequency command based on a quotient obtained by dividing by a secondary magnetic flux command, and a magnitude corresponding to a vector combination of the magnetizing current command and the torque current command and the slip frequency An induction machine control device comprising: means for forming a primary current command based on a phase corresponding to the command; and means for controlling a primary current of the induction machine in accordance with the primary current command. detection calculation means for calculating actual direct axis secondary magnetic flux and horizontal axis secondary magnetic flux based on the voltage, current and induction machine constant; means for determining the deviation from the secondary magnetic flux; means for determining a magnetizing current correction signal according to the magnitude of the deviation; and reducing the direct axis secondary magnetic flux error by adding the magnetizing current correction signal to the magnetizing current command. means for determining the deviation between the horizontal axis magnetic flux command set to approximately zero and the horizontal axis secondary magnetic flux detected by the detection calculation means, and a slip frequency correction signal corresponding to the magnitude of the deviation. and means for reducing a slip frequency error by adding the slip frequency correction signal to the slip frequency command.