JPH01243870A - Induction machine controller - Google Patents

Abstract

PURPOSE:To maintain torque coefficient constant, by correcting the secondary resistance employed for slip frequency calculation such that a true value is provided at all times. CONSTITUTION:Controller for induction motor 3 comprises a vector control command operating circuit 1 and a power converter 2. The operating circuit 1 comprises a magnetizing circuit operating unit 20, a torque current operating unit 21, a two phase-three phase converter 23, comparators 26-28, a vector transforming unit 30, a slip frequency operating unit 31, an integrator 32, an adder 34 and the like, where the comparators 26-28 control output current of the power converter 3. Furthermore, a resistance compensating circuit 40 is provided in order to correct the secondary resistance R2 employed for slip frequency operation. A coordinate conversion circuit 25 converts the flux on a fixed shaft into the flux on a rotary coordinates, and the secondary resistance employed for slip frequency operation is corrected upon provision of a true value. By such arrangement, the secondary resistance is corrected to provide a true value at all times thus maintaining a normal vector control performance.

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to an induction motor control device that performs variable speed control of an induction motor by vector control.

(Conventional technology) Dielectric core! ! In the past, IJ machines were often used as constant-speed motors, but as power converters such as inverters and cycloconverters have become easier to configure in recent years due to advances in power semiconductor devices, IJ machines have been used as variable-speed motors. Its uses have expanded significantly.

A vector control method is often adopted as a variable speed control method for induction motors because it provides excellent response characteristics.This vector control method detects the secondary magnetic flux as a vector quantity and controls the primary current. A magnetic flux detection type vector control method used for signals and a slip frequency type vector control method in which a magnetic flux vector is calculated and controlled based on a motor constant are known.

As shown in Fig. 4, the conventional slip frequency type vector control method is a vector control command calculation that inputs the secondary magnetic flux command value Φ and the generated torque command value τ and outputs the primary current command value. A power converter 2 and 1. which controls a circuit 1 and a primary current value controller based on the primary current command value I□. It consists of an induction motor 3 that rotates at a predetermined speed and torque using this primary current value 11, and a speed detector 4 that detects the rotational speed ωR of this motor.

The vector control command value calculation circuit 1 calculates the secondary magnetic flux command value Φ2.
and the generated torque command value τ are calculated according to the transfer characteristics of the induction motor 3, and a constant is set so as to output the real component command value 11 of the primary current, the imaginary component command value 1tI, and the slip frequency command value ωS. 9.10°11, 12, division element 13
．． 13', multiplier 19. Differentiator 14, adder 15.17
It is equipped with such things as In the figure, I2 is the secondary self-inductance of the induction motor, R2 is the secondary resistance value, and M is the mutual inductance, which are set before operation.

The real component command value 11R (which gives the magnitude of the magnetic flux and also serves as the primary current) and the imaginary component command value (which gives the torque magnitude) lxr are led to the arithmetic circuit 16, and the primary current index
, is output as 1x=ltn+Jix. on the other hand,
The slip frequency command value ωS is led to an adder 17 and added together with the rotational frequency signal ω from the speed detector 4, and then converted by a vector generator 18 into a unit current vector that determines the position of the secondary magnetic flux. 1 multiplier 19
multiplies this unit current vector by the primary current command i□, and outputs the obtained primary current command common value 11 to the power converter 2.

Furthermore, Fig. 5 shows a conventional example of a magnetic flux detection type vector control system, in which the magnetic flux is determined by the induction motor voltage 111a t V t b * Z' t e and the primary current 11at l lbe l without installing a detector. It is calculated by a magnetic flux calculator based on 1°. The outputs of the magnetic flux calculator are stationary two-axis components φ2α and φ2β of the secondary magnetic flux vector.

The vector analyzer converts it into an absolute value component 1φ21 and a position angle sin and cosf.

On the other hand, the magnetic flux command value Φ2 and the torque command value τ^ are the magnetizing current Ltd, torque current 11q, and member corresponding to each other.
, @lid+ 11q is the magnetizing current and torque current in the rotating coordinate system when the rotating magnetic flux is aligned with the d-axis, and is converted to the current 11αel, a in the stator coordinate system based on the position angle a. do. That is, Pect, II [combinant output 1x (1*z1β is the secondary magnetic flux component
α.

2-phase to 3-phase conversion is performed using the current to create φ2β. I4 Current command of power converter 11ap l tb*
c and control the output current of the power converter by conventional and usual means.

(Problem to be solved by the invention) However, the above conventional slip frequency type vector constraint
,^ When calculating the command value 1x)l of the primary current component and the slip frequency command value ωS, 2
Since the secondary resistance R2 is directly involved, the conventional vector control method has the disadvantage that the vector control characteristics deteriorate as the temperature of the rotor increases.

In addition, in the magnetic flux detection type vector control method, although it is not affected by the secondary resistance, it has the disadvantage that the secondary magnetic flux, that is, the air gap magnetic flux force, cannot be extracted when secondary leakage inductance is ignored. . However, even if the constants of the primary circuit and secondary circuit of the induction motor change, the input voltage of the magnetic flux calculator remains unchanged. , input current 1 za
e l 1be l l. It is perceived as a change in
Since the magnetic flux calculation result changes accordingly, there is little deterioration of vector control characteristics due to parameter changes.

In terms of implementation, there are many problems with the accuracy and resolution of the sensor when providing a magnetic flux detector, and when using a magnetic flux calculator, there are gaps in calculation accuracy due to voltage distortion, especially at low speeds. The current situation is that sufficient starting torque cannot be obtained.

In other words, in conventional slip frequency type vector control, when the secondary resistance changes due to temperature, the response of the main magnetic flux deteriorates, which has a serious effect on the calculation result of the slip frequency command value ω8, and the original vector control characteristics are deteriorated. There are drawbacks that make it unsustainable. Further, even if magnetic flux detection or calculation is used, it is necessary to eliminate the above-mentioned factors that deteriorate the starting characteristics.

SUMMARY OF THE INVENTION Therefore, the present invention aims to provide an induction machine control device that compensates for fluctuations in the secondary resistance R2 that have a significant effect on slip frequency calculations and keeps the torque coefficient constant in slip frequency type vector control. purpose.

[Structure of the invention]

(Means for Solving the Problems and Their Effects) Therefore, in order to achieve the above object, the present invention provides a magnetizing current calculation means for calculating a magnetizing current command based on a secondary magnetic flux command, and a magnetizing current calculation means based on a torque command. a torque current calculation means for calculating a torque current command based on the secondary magnetic flux command and the torque command; a slip frequency calculation means for calculating a slip frequency command based on the secondary magnetic flux command and the torque command; , a primary current forming means for forming a primary current based on the phase corresponding to the slip frequency command, a primary current control means for controlling the primary current of the induction machine according to the primary current command, and a horizontal axis magnetic flux. 2 so that the integral value of becomes zero.
A control device for an induction machine comprising a correction means for correcting a secondary resistance value, a magnetizing current calculation means for calculating a magnetizing current command based on a secondary magnetic flux command, and a magnetizing current calculation means for calculating a torque current command based on a torque command. a torque current calculation means for calculating a slip frequency command based on a secondary magnetic flux command and a torque command, a magnitude corresponding to a vector composition of a magnetizing current command and a torque current command, and a slip frequency command for A primary current forming means that forms a primary current based on the corresponding phase, a primary current control means that controls the primary current of the induction machine according to this primary current command, and a torque current command value and an actual torque current. and a correction means for correcting the secondary resistance value so that the integral value of the deviation from the secondary resistance value becomes zero.

(Example) An embodiment of the present invention will be described below with reference to the drawings.

As shown in FIG. 1, the secondary magnetic flux command Φ2 introduced into the vector control command calculation circuit 1 in this embodiment is transmitted to the magnetization current command operator i-d in the magnetization current calculation unit 20, and also to the torque command τ The true value is converted into a torque current, also referred to as 11Q, by the torque current calculator 21 with the help of the magnetic flux command Φ2. The magnetizing current command Ltd and the torque current command 11Q are converted by the vector converter 30 into a current 11 and a phase angle θ on the rotating coordinate axis (dtq axis). On the other hand, using the torque current command itq and the secondary magnetic flux command Φ2, the slip frequency calculator 31 generates the slip frequency command ωS, and the adder 34 generates the rotational frequency ωR.
and ωS, that is, a single frequency command ω1 is created. Furthermore, the primary station wave number command ω is integrated by the integrator 32 to obtain the phase angle y of the coordinate axis for one rotation, and the sum of the 9 phase angle θ slope with the dq axis, that is, θ, = o + 'f, is obtained. And this θ1 is the phase of the current seen on the stationary axis. The absolute value 11 of the primary current is the magnetization current command lad and the torque current command 1.
It is a composite current of 1q, and 1 phase angle θ□ indicates its phase angle on the two stationary axes. Therefore, this absolute value 11 and the phase angle θ,
is converted from 2-phase to 3-phase by 2-phase to 3-phase conversion 23 to create current command 1 ua * l sbe l 1°, and in comparators 26 and 27. Primary current of motor 2 i tap l
xbp 1 x.

are compared for each phase, and current control is performed to reduce the deviation to zero.

Basically, vector control is possible using the above-mentioned slip frequency type control method, but the calculation is performed in accordance with the calculation of the slip frequency command ωS. However, R2 is the secondary resistance, R2 is 2
The self-inductance, M is the mutual inductance. Here, if the secondary resistance R2 changes depending on the temperature,
The slip frequency command will include an error. As a result, the primary frequency ω1 also includes an error, and the phase angle θ of the primary current also includes an error. Therefore, the orthogonal relationship between the magnetizing current and the torque current cannot be maintained, and the vector control characteristics deteriorate. Therefore, by providing the resistance compensation circuit 40 according to the present invention, the
Next, correction of resistance R2 is performed.

The conventional vector control device has a secondary magnetic flux command Φ2 (dq
It was assumed that a magnetic flux equal to the d-axis magnetic flux Φ2d) in the axial coordinate system was generated, and that an actual torque current equal to the torque current command xxq was flowing, but the actual magnetic flux and torque current were not known. That is, the torque τ of the induction motor is
゛τ= (11Qφzd Ltdφ29)
(2) The L angles φ2d and φ29 are expressed by the d-axis and q-axis components of the secondary magnetic flux. Conventionally, Φzd=Φ2=constant, Φ29=0
Since the discussion is based on this assumption, it is assumed that torque does not occur spontaneously. There's a big problem here. If the secondary resistance changes due to temperature changes, etc., it is not possible to give an appropriate slip frequency command, so the second term 1idφ of equation 0
, 9≠0, and a phenomenon of torque reduction appears. The following two points can be considered as countermeasures.

■ Add correction to the secondary resistance R2 so that Φ2q=0 can always be maintained.

■ Control so that the torque current 11111 always matches the torque current command 11q, and as a result, Φ29=0
A correction is added to the secondary resistance R2 so as to bring it closer to .

In realizing the above ■■, it is assumed that Φzd”Φ2=constant can be maintained.If there is no magnetic saturation, Φ
Achieving 29=0 also means keeping Φ2d constant. When there is magnetic saturation, the actual direct-axis magnetic flux Φ2d is determined by calculation and feedback control is performed. The embodiment shown in FIG. 1 implements the above-mentioned countermeasure (2), and the embodiment shown in FIG. 3 implements the countermeasure (2). FIG. 2 is a detailed diagram of the magnetic flux calculator 24 in the embodiment shown in FIG.

Before explaining the above two embodiments, first the secondary magnetic flux Φ2 (=
The calculation method for ΦZa) will be explained.

A magnetic flux sensor is attached to the stator of the electric motor, and the stator has two axes (
It is also possible to detect the magnetic flux of the αβ axis), but it is also possible to calculate or estimate it without installing a sensor. The magnetic flux calculator (estimator) 24 in FIG. 1 performs this function, and its contents will be explained using FIG. 2.

FIG. 2 shows an embodiment of the magnetic flux calculator shown in FIG. Three-phase voltage Z'xat supplied to induction motor IM?
xb* Vxc and 3-phase current 1 map 1 ib* l
xc is detected, and using a well-known method, the arithmetic unit 60.61
, the stator is converted into a two-axis (αβ-axis) system. The voltage and current on the α-axis and β-axis after conversion are V1αl 11°, V2
β.

When expressed as i and β, the α-axis component, φ2α, and the β-axis component of the secondary magnetic flux are calculated as linear.

(2), Ω) are performed by an integrator 60. However, R1 and Q are a primary resistance and a primary leakage inductance. The magnitude of the secondary magnetic flux φ2 is determined by the square root operator 6.
Obtained through 3.

To convert the magnetic fluxes φ2α and φ2β on the fixed axis (αβ axis) into magnetic fluxes φ2dt and φ2q on the rotating coordinate system (d and q axes), the following calculation is performed in the coordinate conversion circuit 25.

When ωS is given the true value, φzd=constant=φ2゜φ
29=0, but when the secondary resistance R2 fluctuates, φ2q occurs. Therefore, the 2 used for the slip frequency calculation value is
Modify the secondary resistance according to the following formula: where R2 is the secondary resistance value measured before operation and k is a constant. By correcting the secondary resistance value using the equation 0, the two blown fiber bundles φ and q on the q-axis can be made zero on average, and the torque of the second term in the equation (2) can be made zero. This is the principle of the embodiment shown in FIG. Integration of equation 0 is performed by an integrator 33.

On the other hand, in the embodiment shown in FIG. 3, the secondary resistance is corrected to make the actual torque current i□9 match the torque current command ixq. The difference from FIG. 1 is the content of the resistance correction circuit 40. Voltage u ta, V tbt
'l/te and current 1 sat l ibt l
16 is converted into a two-phase signal by a three-phase/two-phase converter 37 and converted to G3 by a method similar to the formula (2) blown fiber bundles φ2α, φ, β
is calculated. Now cospu = φ2α/φ,, sin,
If l/'=φ2β/φ2, the current on the two fixed axes is 11
αe 11β is the current 1 on the rotating coordinate axis as follows.
xd* l 1q.

■The current Ltd obtained from the formula is the actual magnetizing current,
IIQ is the actual torque current. The fact that the actual torque current 11Q matches the torque current command value means that the generated torque is determined only by the first term of equation (2), and the torque due to the second term is not generated. Therefore, 2
When the resistance changes due to temperature, the same effect as shown in FIG. 1 can be obtained even if the deviation of the torque current is corrected.

That is.

The secondary resistance is given as follows.

In FIG. 3, the coordinate converter 38 executes the coordinate transformation of equation (2), and the integrator 33 executes the integration operation of equation (2).

〔Effect of the invention〕

As described above, according to the present invention, the secondary resistance used for slip frequency calculation is always corrected to give the true value, and even if the temperature rises, the normal vector control characteristic is maintained, that is, the torque coefficient is constant. Control can be maintained.

[Brief explanation of the drawing]

FIG. 1 is a block diagram showing one embodiment of the present invention, FIG. 2 is a detailed diagram of the magnetic flux calculator shown in FIG. 1, and FIG. 3 is a block diagram showing another embodiment of the present invention. . FIGS. 4 and 5 are block diagrams showing the configuration of a conventional induction machine control device. 1... Vector control command calculation circuit 2... Power converter 3... Induction machine 4... Speed detector 20... Magnetizing current calculator 21... Torque current calculator 23... 2 Phase-3 phase converter 24... Magnetic flux calculator 25, 38... Coordinate converter 30... Vector converter 31... Slip frequency calculator 32.33...
Integrator 34, 35... Adder 36... Comparator 37... 3-phase to 2-phase converter 60, 61... 2-phase to 3-phase converter 62... Integrator 63... Square Root Calculator Agent Patent Attorney Nori Ken Yudo Chika Ken Daishimaru Akira

Claims (2)

[Claims] (1) Magnetizing current calculation means for calculating a magnetizing current command based on a secondary magnetic flux command, torque current calculation means for calculating a torque current command based on a torque command, and a magnetizing current calculation means for calculating a torque current command based on a torque command; 1 based on the magnitude corresponding to the vector combination of the magnetizing current command and the torque current command, and the phase corresponding to the slip frequency command.
A primary current forming means for forming a secondary current, a primary current control means for controlling the primary current of the induction machine according to the primary current command, and a secondary resistance value such that the integral value of the horizontal axis magnetic flux becomes zero. 1. A control device for an induction machine, comprising: correction means for correcting. (2) magnetizing current calculation means for calculating a magnetizing current command based on a secondary magnetic flux command; torque current calculation means for calculating a torque current command based on a torque command; and a magnetizing current calculation means for calculating a torque current command based on a torque command; 1 based on the magnitude corresponding to the vector combination of the magnetizing current command and the torque current command, and the phase corresponding to the slip frequency command.
A primary current forming means that forms a secondary current, a primary current control means that controls the primary current of the induction machine according to the primary current command, and an integral value of the deviation between the torque current command value and the actual torque current. 1. A control device for an induction machine, comprising: a correction means for correcting a secondary resistance value so that it becomes zero.