JP2005304175A - Speed controller of motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a speed controller of a motor which perform a speed control stable in all regions including a zero speed, a low speed, a regenerating, and a high speed even when the motor is connected to a load through a power transmission means. <P>SOLUTION: The speed controller includes a speed detection means for detecting a motor speed based on a position or speed information inputted from an external detector, an observer for calculating to estimate the motor speed based on a state variable and motor speed composite value which is outputted and controlled by a controller, a current controller for determining an instruction current based on the amount of feedbacks and the speed command value of the motor by setting either the calculated and estimated value of the motor speed by the observer or the motor speed composite value as the amount of the speed feedbacks, and a speed composite means for outputting the motor speed composite value based on the speed detection value by the speed detecting means in a low speed zone, based on the calculated estimated value of the motor speed by the observer in a high speed zone, and based on the calculated estimated values of the speed detection value and the motor speed in an intermediate speed zone. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

  The present invention relates to a motor speed control device that enables speed control over a wide range in combination with an external detector, and controls the rotational speed of motors connected via power transmission means such as gears and belts. The present invention relates to a speed control device for an electric motor that controls a stacker crane, a spindle of a machine tool, a printing machine, and the like, and more particularly to a speed control device that uses an induction motor for the electric motor.

但し、
Ｖｄ，Ｖｑ ：回転座標系（ｄ−ｑ軸）の励磁軸電圧，トルク軸電圧
Ｉｄ，Ｉｑ ：回転座標系（ｄ−ｑ軸）の励磁軸電流，トルク軸電流
Φｄ，Φｑ ：回転座標系（ｄ−ｑ軸）の励磁軸電動機磁束，トルク軸電動機磁束
ω1 ：１次周波数
ωr ：電動機速度
ωs* ：滑り周波数指令値
Ｒ1 ，Ｒ2 ：１次，２次抵抗
Ｌ1 ，Ｌ2，Ｍ ：１次，２次，相互（励磁）インダクタンス
LL ：漏れインダクタンス （LL＝（Ｌ1・ Ｌ2 −Ｍ² ）／Ｌ2 ）
ｐ ：微分演算子
また、１次周波数ω1、電動機速度ωr、滑り周波数指令値ωs*の関係、及び滑り周波数指令値ωs*の算出は以下の（２）式で表わされる。
ω1＝ωr＋ωs*、
ωs* ＝Ｉｑ* ／（Ｉｄ* ・T2） ＝ Ｉｑ* ・Ｒ2／Φ* …（２）
但し、T2 は２次回路時定数（T2 ＝Ｌ2 ／Ｒ2 ）であり、添字（*）は指令値を表わす。 As a high-performance speed control method for induction motors, a vector control method of a slip frequency control type is widely used, and a speed sensorless vector control method for controlling this without a speed detector is known.
First, an example of estimating the motor speed of the induction motor will be briefly described.
The voltage equation of the induction motor represented by the rotating coordinate system (dq axes) rotating at the primary frequency ω1 is given by the following equation (1).

However,
Vd, Vq: Excitation axis voltage of rotating coordinate system (dq axis), Torque axis voltage Id, Iq: Exciting axis current of rotating coordinate system (dq axis), Torque axis current Φd, Φq: Rotating coordinate system ( dq axis) excitation shaft motor magnetic flux, torque shaft motor magnetic flux ω1 : Primary frequency ωr : Motor speed ωs * : Slip frequency command value R1 , R2: primary, secondary resistance L1 , L2, M : Primary, secondary, mutual (excitation) inductance
LL: Leakage inductance (LL = (L1 · L2 -M ²⁾ / L2 )
p: Differential operator The relationship between the primary frequency ω1, the motor speed ωr, the slip frequency command value ωs *, and the calculation of the slip frequency command value ωs * are expressed by the following equation (2).
ω1 = ωr + ωs *,
ωs * = Iq * / (Id * ・ T2) = Iq * ・ R2 / Φ * (2)
T2 Is the secondary circuit time constant (T2 = L2 / R2 ) And the subscript (*) represents the command value.

電動機の速度制御装置が（２）式を満足するように制御すると、回転座標系（ｄ−ｑ軸）上の１次電流検出値Ｉ1 は１次電流指令値Ｉ1*（Ｉｄ*，Ｉｑ*）通りの電流が流れ、電動機磁束Φ（Φd，Φq）は、
Φd＝Ｍ・Id（一定），Φｑ＝０ …（５）
に保たれる。これにより、誘導電動機の発生トルクτは、
τ＝Ｍ／Ｌ2・(Φｄ・Iｑ−Φｑ・Iｄ)
＝Ｍ² ／Ｌ2 ・(Ｉｄ・Iｑ) …（６）
となる。 Now, Id * = constant (3)
And the voltage command values (Vd *, Vq *) expressed in the rotating coordinate system (dq axes) under the conditions of the expressions (2) and (3) are expressed as follows: 4) It becomes like a formula.

When the speed control device of the motor is controlled so as to satisfy the expression (2), the primary current detection value I1 on the rotating coordinate system (dq axes) Current flows according to the primary current command value I1 * (Id *, Iq *), and the motor magnetic flux Φ (Φd, Φq) is
Φd = M · Id (constant), Φq = 0 (5)
To be kept. Thereby, the generated torque τ of the induction motor is
τ = M / L 2 · (Φd · Iq−Φq · Id)
= M ² / L2 ・ (Id ・ Iq) (6)
It becomes.

When the speed detector is not used, for example, an electric motor magnetic flux Φ that satisfies the above equation (3) using a speed adaptive secondary magnetic flux observer is used. (Φd, Φq) is estimated, and the primary current (phase current) Iu of the induction motor , Iv , Iw And the primary current detection value I1 (Ia, Ib), the voltage command value V1 * (Vref, Vbref), and the estimated speed value ωr ^ converted into the stationary coordinate system (ab axis) are input. , Estimated motor flux Φ ^ (Φa ^ , Φb ^ ) And the estimated primary current I1 ^ (Ia ^, Ib ^) and compare the primary current estimated value I1 ^ (Ia ^, Ib ^) with the primary current detection value I1 (Ia, Ib) Based on the estimated error signal (I1-I1 ^), the motor speed is calculated and estimated by the adaptive adjustment law expressed by the following equation (7).
ωr ^ = Ka (eIa / Φb ^ −eIb / Φa ^)
+ Kb ∫ (eIa · Φb ^ −eIb · Φa ^) dt (7)
However, the estimation errors eIa and eIb are as follows.
eIa = Ia-Ia ^
eIb = Ib−Ib ^
Ka: Speed estimation unit proportional gain, Kb : Speed estimation unit integral gain In this way, the speed sensorless vector control that estimates the actual speed of the induction motor by the speed adaptive secondary magnetic flux observer composed of the same-dimensional magnetic flux observer (hereinafter simply referred to as magnetic flux observer) and the speed adaptive mechanism. As a method, for example, there is Non-Patent Document 1.

However, on the other hand, regarding the speed sensorless vector control method that estimates the actual speed of the induction motor with an adaptive secondary magnetic flux observer, it has been reported that the control performance deteriorates at low speed and regenerative state, and there is a problem in stability ( For example, refer to Non-Patent Document 2 and Non-Patent Document 3), and it has also been reported to improve low-speed / regenerative performance at low-speed rotation by applying a voltage including a sufficiently high frequency component to the induction motor (Non-Patent Document). 3).

As a control device using a speed detector on the machine shaft side, for example, there is Patent Literature 1. FIG. 6 is a block diagram described in Patent Document 1. 101 is a three-phase two-phase converter, 102 is an integrator, 103 is a two-phase three-phase converter, 104 is an inverter, 105 is an induction motor, 107 is a main shaft, 108 is a rotational position detector, 109 is a sensorless speed detector, 110 is a position / speed converter, 111 is a high-pass filter, 112 is a low-pass filter, 113 is an adder, 114 is a comparator, 115 is a switch, and 116 is a limiter. , 117 is a current controller, 118 is a magnetic flux density command generator, 119 is an excitation current command generator, 120 is a subtractor, 121 is a PI amplifier, 122 and 123 are dividers, 124 is a multiplier, 125 is an adder, 126 is a DC voltage source, and 127 (a), (b), and (c) are current detectors.
In this control apparatus, the excitation current id and the torque current iq inside the motor are detected as direct current amounts from the instantaneous values iu, iv, iw of the motor current detected by the current detector 127 by the three-phase / two-phase conversion means 101. Yes. The excitation current id and the torque current iq are feedback-controlled by the current controller 117 according to the deviation between the excitation current command value id * and the torque current command value iq *, respectively. If there is an error in the estimated value of the motor speed, the first speed estimation means determines the deviation between the torque current detection value iq and the torque current command value iq * or the excitation current detection value id and the excitation current command value id *. The first speed detection value is calculated by using the sensorless speed detection unit 109 by utilizing the fact that the deviation of. This first speed detection value has a property of including a lot of detection errors in low frequency components, but a relatively high detection value can be obtained for high frequency components. On the other hand, the second motor rotational speed estimated value calculated by the position / velocity converter 110 from the rotational position detector 108 attached to the main shaft, that is, the second speed detected value, has a characteristic that includes a lot of errors in a relatively high frequency component. However, relatively low detection values can be obtained for low frequency components. Therefore, accurate frequency components of both the first speed detection value and the second speed detection value by the detector attached to the spindle are extracted by the high-pass filter 111 and the low-pass filter 112, respectively, and are added and synthesized by the adder 113. When the difference between the first speed detection value and the second speed detection value is equal to or greater than a predetermined value, the first speed detection value is detected as the motor speed. It is used as a value.
Further, for the problem that the torque in the deceleration direction cannot be accurately controlled particularly in the low motor speed region, the limiter 116 allows the q-axis current command value in the low speed / regenerative region to be set to a predetermined limit value. This is solved by providing a sequence processing that is limited to the above.

Kubota, Ozaki, Matsuse, Nakano, “Velocity sensorless direct vector control of induction motor using adaptive secondary magnetic flux observer”, IEEJ Transactions D, Vol. 111, No. 11, p. 954-960 Sugimoto, Ding, “A Study on Stability of Induction Motor System Using Adaptive Secondary Magnetic Flux Observer”, IEEJ Transactions D, 119, 10, p. 1212-1222 Kanehara and Koyama, “Speed Sensorless Vector Control Method for Induction Motors Including Low-Speed / Regenerative Regions”, IEEJ Transactions D, Volume 120, No. 2, 223-229 JP 2002-51594 A

However, in the speed sensorless control method using the conventional speed adaptive secondary magnetic flux observer, the control performance deteriorates in the low speed / regenerative state, there is a problem in stability, and a voltage including a sufficiently high frequency component is applied to the induction motor. In the method of applying, there is a problem that torque ripple is generated due to the superimposed voltage, and that strict gain adjustment is required. Further, the conventional example described in Patent Document 1 has a problem that requires special sequence processing such as limiting the q-axis current command value in a deceleration direction in a low motor speed region, that is, in a low speed / regenerative region. .
Therefore, the present invention has been made in view of such problems, and an object of the present invention is to provide an electric motor speed control device capable of stable operation in all regions including zero speed, low speed / regeneration regions. There is.

In order to solve the above problems, in a motor speed control device, speed detection means for detecting motor speed based on position or speed information input from an external detector, state variables output by the speed control device, and motor speed Based on the feedback amount and the speed command value of the motor, the observer for calculating and estimating the motor speed based on the combined value, and either the calculation estimated value of the motor speed or the motor speed combined value by the observer is a speed feedback amount. A current controller that determines a command current, and based on a speed detection value by the speed detection means when the motor speed is in a low speed range, and based on a calculation estimated value of the motor speed by the observer in a high speed range, In the region, based on the speed detection value and the operation estimated value of the motor speed, It is characterized by comprising a speed synthesis means for outputting a motive speed combined value.
Because of this, speed control can be performed without being affected by backlash of the power transmission means in the high speed range, and in the low speed range where the control performance deteriorates by using an external detector, especially at low speed and regeneration, The control system does not become unstable, and very stable operation can be performed in the entire region, and switching between these can be performed smoothly through the medium speed region.

Further, in the speed control device for the induction motor, based on the speed detection means for detecting the motor speed based on the position or speed information input from the external detector, the state variable controlled by the speed control device and the motor speed composite value. An observer for calculating and estimating the motor speed, and either the motor speed calculation estimated value or the motor speed combined value by the observer is a speed feedback amount, and the command current is calculated based on the feedback amount and the motor speed command value. A current controller for determining, a slip frequency calculator for calculating and outputting a slip frequency, and a motor speed calculation by the observer in a high speed range based on a speed detection value by the speed detection means when the motor speed is a low speed range. Based on the estimated value, the speed detection value and the power Based on the calculated estimated value of the machine speed is characterized by comprising a speed synthesis means for outputting the motor speed combined value.

Further, in the motor speed control device, based on the speed detection means for detecting the motor speed based on the position or speed information inputted from the external detector, the state variable output by the speed control device and the motor speed composite value An observer that computes and estimates the phase and motor speed, a slip frequency calculator that computes and outputs a slip frequency, a differentiator that differentiates the estimated computation value of the phase by the observer, and an output of the differentiator A first subtractor for subtracting the output; a current controller for determining a command current based on the speed command value of the motor and the output of the subtractor; and a speed by the speed detecting means when the motor speed is in a low speed range. Based on the detected value, in the high speed range, based on the estimated value of the motor speed by the observer, in the medium speed range It is characterized by comprising a speed synthesis means for outputting the motor speed combined value based on the calculated estimated value of the speed detected value and the motor speed.

Furthermore, a first weighting function unit that receives a speed detection value obtained by the detection unit and a second weighting function unit that receives an operation estimated value of the motor speed by the observer, and the speed synthesis unit includes: The output of the first weight function unit and the output of the second weight function unit are added, and this added value is used as the motor speed composite value in the medium speed range.
Means for smoothly performing the switching through the medium speed range are specifically shown.

Further, the speed detection means detects the motor speed based on position or speed information inputted from an external detector on the load side connected to the electric motor via a power transmission means.
As a result, even if there is play in the power transmission means, stable speed control can be performed even in the middle and high speed range that is affected by play, and over the entire speed range, including the low speed range that is not affected by play. Stable speed control can be performed.

Furthermore, a second subtracter that subtracts the speed detection value from the speed detection means from the speed command value, an integrator that integrates the output of the second subtractor, an output of the slip frequency calculator, and the integrator And a slip frequency correcting means for outputting a multiplication value obtained by multiplying the output as an output correction amount to the slip frequency device.
Thus, the slip frequency can be corrected even at an extremely low speed or in a regenerative state, and the motor can be driven and controlled according to the speed command value.

  Since the motor speed control is performed as described above, a stable speed control operation can be performed in the entire range from the low speed range to the high speed range, and the control system becomes unstable especially at ultra low speeds and even during regeneration. There is also an effect that a very stable operation can be performed.

  Hereinafter, specific examples of the present invention will be described.

また、座標変換器１１Ａは、Ｉａ、Ｉｂを同一次元磁束オブザーバ２０へ送る。速度演算器１２は、ドラム３２に連結された位置決め用検出器３４からの機械軸速度検出値に動力伝達手段であるギア３１の変速比を乗じて速度検出値ωｒを算出し、係数器２２Ａに送る。同一次元磁束オブザーバ２０、速度適応機構２１は、座標変換器１１ＡからＩａ、Ｉｂと後述する座標変換器１１ＢからのＶａref、Ｖｂrefおよび、後述する加算器２３からの合成速度信号ωcを基に、上記（７）式により速度推定演算してωｒ＾を求め、減算器４３にフィードバックするとともに、係数器２２Ｂへも送る。係数器２２Ａ、２２Ｂは、図５に示す速度推定値ωｒ＾の関数として、予め決められたゲインＫを用いて、それぞれ、Ｋ、（１−Ｋ）で与えられる。加算器２３は、係数器２２Ａ、２２Ｂで重みづけされたそれぞれの信号を加算して合成速度信号ωcとし、同一次元磁束オブザーバ２０に帰還する。減算器４３は、位置決め制御装置（図示せず）からの速度指令値ωｒ*と速度適応機構２１からの速度推定値ωｒ＾の偏差をとって速度制御部１４に送る。速度制御部１４は、減算器４３から与えられた偏差を零とするような、すなわち速度指令値ωｒ*と速度推定値ωｒ＾を一致させるようなトルク電流指令信号Ｉｑ*を求め、減算器１３Ａに送る。減算器１３Ａは、座標変換器１１Ａからのトルク電流フィードバック信号Ｉｑと速度制御部１４からのトルク電流指令信号Ｉｑ*の偏差を求め、トルク電流制御部１６Ａに送る。励磁電流設定器１５は、所定の励磁電流値が設定されており、その設定値を励磁電流指令信号Ｉｄ*として減算器１３Ｂに送る。減算器１３Ｂは、励磁電流設定器１５からの励磁電流指令信号Ｉｄ*と、座標変換器１１Ａからの励磁電流フィードバック信号Ｉｄの偏差を求め、励磁電流制御部１６Ｂに送る。トルク電流制御部１６Ａおよび励磁電流制御部１６Ｂは、それぞれ減算器１３Ａと１３Ｂからの偏差信号と各逆起電力の電圧補償値（図示せず）との和にしたがって、回転磁界座標系における電圧指令信号Ｖｑ*とＶｄ*を生成し、座標変換器１１Ｂに送る。座標変換器１１Ｂは、電圧指令信号Ｖｄ*、Ｖｑ*を位相θで（ａ−ｂ軸）上のＶａref、Ｖｂrefに変換し、更に電動機２の固定子座標系における３相交流出力電圧指令信号Ｖｕ*、Ｖｖ*、Ｖｗ*に変換し、ＰＷＭ制御部７に送る。座標変換器１１Ｂによる変換は以下の（９）式にしたがって行なわれる。なおθについては後述する。

また、座標変換器１１Ｂは、Ｖａref、Ｖｂrefを同一次元磁束オブザーバ２０へ送る。ＰＷＭ制御部７は、これら３相交流出力電圧指令Ｖｕ*、Ｖｖ*、Ｖｗ*を搬送波信号と比較してパルス幅変調信号に変換する。このパルス幅変調信号を図示していないパルス増幅器を介して点弧信号となり、トランジスタのベース電流として逆変換回路５をスイッチング制御する。これにより、平滑コンデンサ４の両端の直流電圧が３相の交流電圧に変換される。滑り周波数指令演算器１７は、励磁電流指令Ｉｄ*、トルク電流指令Ｉｑ*と２次抵抗Ｒ２（図示せず）から滑り周波数指令ωｓ*を求める。加算器１８は、速度適応機構２１で推定されたωｒ＾と滑り周波数指令ωｓ*を加算して、１次角周波数指令信号ω１*を求める。積分器１９は、１次角周波数指令信号ω１*を積分し位相θを求め、座標変換１１Ａ、１１Ｂに送る。
なお、図１では、電動機２により発生する誘起電圧Ｅ、１次抵抗Ｒ1や漏れインダクタンスLLによる逆起電力の電圧を補償する回路は省略している。
また、図５記載の所定値１，２の値は、機械の共振周波数が問題にならない速度域に決定しており、所定値２以上の速度域では、完全な速度センサレス制御に切り替えられる。なお、図５では、速度推定値ωｒ＾で切り替えるように記載しているが、位置決め用検出器３４から速度演算器１２で求めた速度検出値ωrや、一次周波数指令値ω1*、あるいは、速度指令値ωr*等で切り替えることもできる。
また、上記説明で、減算器４３と加算器１８への速度フィードバック信号は、速度適応機構２１からの速度推定値ωｒ＾としたが、加算器２３で演算され同一次元磁束オブザーバ２０に送られる合成速度信号ωcを用いてもよい。
このように、速度制御が行われるので、制御性能が劣化する低速領域では、電動機の実際の速度に基づく位置決め用検出器３４から速度情報が同一次元磁束オブザーバ２０に帰還されるので、低速・回生時に制御系が不安定になることもなく運転が行われる。また、検出器に基づく低速運転とセンサレス制御に基づく高速運転との間の中間領域においては、これら双方を加味した制御を行うので、低速から中速・高速域に移ったとしても、その間もその後もスムーズに安定した速度制御運転が行われることとなる。 FIG. 1 is a block diagram showing a configuration of an electric motor vector control apparatus according to a first embodiment of the present invention. In FIG. 1, the motor control device has a PWM inverter 1 and a vector controller 10 and controls a mechanism system 30. In the PWM inverter 1, there are a converter 3, a smoothing capacitor 4, an inverse conversion circuit 5, current detectors 6 </ b> A, 6 </ b> B, 6 </ b> C, and a PWM control unit 7. In the vector controller 10, coordinate converters 11A and 11B, a speed calculator 12, subtractors 13A, 13B, and 43, a speed controller 14, an excitation current setting unit 15, a current controller 16, and a slip frequency command calculator 17 are provided. , An adder 18, an integrator 19, a one-dimensional magnetic flux observer 20, and a speed adaptation mechanism 21, and the current controller 16 includes a torque current controller 16A and an excitation current controller 16B.
In the mechanical system 30, a positioning detector 34 for controlling the position of a drum 32 and a load 33 is connected to a gear 31 as a power transmission means to the electric motor 2, and the load 33 is a rope 35 via a pulley 36 and a drum 32. It is connected with. The signal from the positioning detector 34 is sent to a positioning controller (not shown), and the positioning controller sends a speed command to the motor control device.
The converter 3 is connected to a three-phase AC power source (R, S, T) and rectifies its output. The smoothing capacitor 4 is connected to the converter 3 and smoothes its output. The inverse conversion circuit 5 is composed of, for example, a transistor whose base current is controlled by the output of the PWM control unit 7. As a result, the DC voltage across the smoothing capacitor 4 is converted into a three-phase AC voltage controlled by the output of the PWM controller 7 and supplied to the electric motor 2. The current detector 6A detects a U-phase current Iu. The current detector 6B detects the V-phase current Iv. The current detector 6C detects the W-phase current Iw. The detected currents Iu, Iv, Iw of each phase detected by the current detectors 6A, 6B, 6C are supplied to the coordinate converter 11A. The coordinate converter 11A converts the three-phase detection currents Iu, Iv, and Iw into Ia and Ib on the coordinate system (ab axis), and further, an excitation current feedback signal Id and a torque current feedback signal in the rotating magnetic field coordinate system. Iq is converted and sent to the subtracters 13A and 13B. The conversion by the coordinate converter 11A is performed according to the following equation (8). Note that θ will be described later.

Also, the coordinate converter 11A sends Ia and Ib to the same-dimensional magnetic flux observer 20. The speed calculator 12 calculates the speed detection value ωr by multiplying the mechanical shaft speed detection value from the positioning detector 34 connected to the drum 32 by the gear ratio of the gear 31 that is the power transmission means, and the coefficient unit 22A send. The one-dimensional magnetic flux observer 20 and the speed adaptation mechanism 21 are based on the coordinate converters 11A to Ia and Ib, the Varref and Vbref from the coordinate converter 11B described later, and the combined speed signal ωc from the adder 23 described later. The speed is estimated by the equation (7) to obtain ωr ^, which is fed back to the subtractor 43 and also sent to the coefficient unit 22B. The coefficient units 22A and 22B are given by K and (1-K), respectively, using a predetermined gain K as a function of the estimated speed value ωr ^ shown in FIG. The adder 23 adds the respective signals weighted by the coefficient units 22 </ b> A and 22 </ b> B to obtain a combined speed signal ωc, and feeds back to the same-dimensional magnetic flux observer 20. The subtractor 43 takes the deviation between the speed command value ωr * from the positioning control device (not shown) and the estimated speed value ωr ^ from the speed adaptation mechanism 21 and sends it to the speed control unit 14. The speed control unit 14 obtains a torque current command signal Iq * that makes the deviation given from the subtractor 43 zero, that is, matches the speed command value ωr * and the estimated speed value ωr ^, and the subtractor 13A. Send to. The subtractor 13A calculates a deviation between the torque current feedback signal Iq from the coordinate converter 11A and the torque current command signal Iq * from the speed control unit 14, and sends it to the torque current control unit 16A. The excitation current setting unit 15 is set with a predetermined excitation current value, and sends the set value to the subtractor 13B as an excitation current command signal Id *. The subtractor 13B obtains a deviation between the excitation current command signal Id * from the excitation current setter 15 and the excitation current feedback signal Id from the coordinate converter 11A, and sends it to the excitation current control unit 16B. The torque current control unit 16A and the excitation current control unit 16B, respectively, are voltage commands in the rotating magnetic field coordinate system according to the sum of the deviation signals from the subtracters 13A and 13B and the voltage compensation values (not shown) of the back electromotive forces. Signals Vq * and Vd * are generated and sent to the coordinate converter 11B. The coordinate converter 11B converts the voltage command signals Vd * and Vq * into Vref and Vbref on the phase ab (axis ab), and further converts the voltage command signals Vd * and Vq * into a three-phase AC output voltage command signal Vu in the stator coordinate system of the motor 2. *, Converted into Vv * and Vw * and sent to the PWM control unit 7. Conversion by the coordinate converter 11B is performed according to the following equation (9). Note that θ will be described later.

Further, the coordinate converter 11 </ b> B sends Varef and Vbref to the same-dimensional magnetic flux observer 20. The PWM controller 7 compares these three-phase AC output voltage commands Vu *, Vv *, and Vw * with a carrier wave signal and converts them into a pulse width modulation signal. This pulse width modulation signal becomes an ignition signal via a pulse amplifier (not shown), and the inverse conversion circuit 5 is subjected to switching control as a base current of the transistor. As a result, the DC voltage across the smoothing capacitor 4 is converted into a three-phase AC voltage. The slip frequency command calculator 17 calculates a slip frequency command ωs * from the excitation current command Id *, the torque current command Iq *, and the secondary resistance R2 (not shown). The adder 18 adds the ωr ^ estimated by the speed adaptation mechanism 21 and the slip frequency command ωs * to obtain a primary angular frequency command signal ω1 *. The integrator 19 integrates the primary angular frequency command signal ω1 * to obtain the phase θ and sends it to the coordinate transformations 11A and 11B.
In FIG. 1, a circuit for compensating for the induced voltage E generated by the electric motor 2, the primary resistance R1, and the back electromotive force voltage due to the leakage inductance LL is omitted.
Further, the predetermined values 1 and 2 shown in FIG. 5 are determined in a speed range in which the resonance frequency of the machine does not cause a problem. In the speed range of the predetermined value 2 or more, the control is switched to complete speed sensorless control. In FIG. 5, it is described that switching is performed with the estimated speed value ωr ^, but the detected speed value ωr obtained from the positioning detector 34 by the speed calculator 12, the primary frequency command value ω1 *, or the speed It can also be switched by the command value ωr * or the like.
In the above description, the speed feedback signal to the subtracter 43 and the adder 18 is the speed estimated value ωr ^ from the speed adaptation mechanism 21, but is synthesized by the adder 23 and sent to the same-dimensional magnetic flux observer 20. The speed signal ωc may be used.
Since speed control is performed in this way, in the low speed region where the control performance deteriorates, the speed information is fed back from the positioning detector 34 based on the actual speed of the motor to the same-dimensional magnetic flux observer 20, so that the low speed / regeneration is performed. Occasionally the operation is carried out without the control system becoming unstable. In addition, in the intermediate region between low-speed operation based on the detector and high-speed operation based on sensorless control, control is performed taking both of these into consideration. As a result, a smooth and stable speed control operation is performed.

Next, a second embodiment of the present invention will be described with reference to FIG. The difference from FIG. 1 is that the calculation processing of the speed adaptation mechanism 21 is changed to a part 21 (a), the adder 18 and the integrator 19 are removed, and a differentiator 24 and a subtractor 25 are added instead. That is.
The speed adaptation mechanism 21 (a) uses the estimated motor magnetic flux Φ ^ (Φa ^) estimated as described above. , Φb ^ ) To change the phase θ ^ to tan ^-1 (Φa ^ / Φb ^) and outputs to the coordinate converters 11A and 11B and the differentiator 24. The differentiator 24 differentiates the phase θ ^ that is the output of the speed adaptation mechanism 21 (a), outputs the primary frequency ω1 ^, and sends it to the subtractor 25. The subtractor 25 subtracts the primary frequency ω1 ^ from the differentiator 24 by the slip frequency command value ωs * from the slip frequency command calculator 17 to obtain the estimated speed value ωr ^ and feeds it back to the subtractor 43. Other operations are the same as those in the first embodiment, and a description thereof will be omitted.
Even in the vector control method configured as in the second embodiment, the basic idea is not changed at all, only the method of using the speed adaptive secondary magnetic flux observer is changed, so that the present invention can be implemented as in the first embodiment. Operation is performed without the control system becoming unstable at low speed and regeneration.

Next, a third embodiment of the present invention will be described with reference to FIG. The third embodiment improves the speed control accuracy by correcting the magnitude of the slip frequency command so that the speed detection value obtained based on the positioning detector 34 matches the speed command value ωr *. is there.
3 is different from FIG. 1 in that a subtracter 26, a switching SW 27, a multiplier 28, and an integrator 19A are added.
The subtracter 26 subtracts the speed command value ωr * and the speed detection value ωr from the speed calculator 12 and sends the result to the switch SW27. The switch SW27 outputs the subtractor 26 output to the integrator 19A if the speed command value ωr * is in a constant speed state where the acceleration / deceleration has not been accelerated and the speed detection value ωr is equal to or less than a predetermined value, and 0 otherwise. send. The integrator 19 </ b> A is an integrator having a long time constant considering the resonance frequency of the mechanical system, and integrates the input value and sends it to the multiplier 28. The multiplier 28 multiplies the slip frequency command ωs *, which is the output of the slip frequency command calculator 17, by the output of the integrator 19 </ b> A to obtain a slip frequency command value correction amount Δωs * and sends it to the adder 18. The adder 18 adds the estimated speed value ωr ^ output from the speed adaptation mechanism 21, the slip frequency command ωs * output from the slip frequency command calculator 17, and the slip frequency command value correction amount Δωs * output from the multiplier 28. Other operations are the same as those in the first embodiment, and a description thereof will be omitted.

Next, a fourth embodiment of the present invention will be described with reference to FIG. 4 differs from FIG. 2 in that a subtracter 26, a switching SW 27, a multiplier 28, an adder 29, and an integrator 19A are added.
The subtracter 26 subtracts the speed command value ωr * and the speed detection value ωr from the speed calculator 12 and sends the result to the switch SW27. The switch SW27 outputs the subtractor 26 output to the integrator 19A if the speed command value ωr * is in a constant speed state where the acceleration / deceleration has not been accelerated and the speed detection value ωr is equal to or less than a predetermined value, and 0 otherwise. send. The integrator 19 </ b> A is an integrator having a long time constant considering the resonance frequency of the mechanical system, and integrates the input value and sends it to the multiplier 28. The multiplier 28 multiplies the slip frequency command ωs *, which is the output of the slip frequency command calculator 17, and the output of the integrator 19 </ b> A to obtain a slip frequency command value correction amount Δωs * and sends it to the adder 29.
In the adder 29, the slip frequency command ωs * output from the slip frequency command calculator 17 and the slip frequency command value correction amount Δωs * output from the multiplier 28 are added and sent to the subtractor 25. The subtractor 25 subtracts the output of the differentiator 24 by the output of the over-counter 29 to obtain a speed estimated value ωr ^ and feeds back to the subtractor 43. Since other operations are exactly the same as those in the second embodiment, a description thereof will be omitted.
In this manner, the slip frequency command value ωs * is corrected in the third embodiment and the fourth embodiment. The predetermined value, which is one condition for switching the switching SW 27, is determined to be about several Hz in consideration of the resonance frequency of the mechanical system. Thereby, the motor speed can be driven according to the speed command value even at an extremely low speed.

  According to the present invention, speed control over a wide range can be achieved by combining with an external detector, and in particular, when controlling the rotational speed of electric motors connected via power transmission means such as gears and belts, the external detector Even when the motor is on the load side, for example, even in a speed control device for an electric motor that controls a stacker crane, a main spindle of a machine tool, a printing machine, etc., a stable speed control operation can be performed in all regions from a low speed region to a high speed region. Even at ultra-low speeds, and even during regeneration, the control system does not become unstable, and extremely stable operation can be performed.

It is a block diagram of one Example of the vector control apparatus which shows the 1st Embodiment of this invention. It is a block diagram of one Example of the vector control apparatus which shows the 2nd Embodiment of this invention. It is a block diagram of one Example of the vector control apparatus which shows the 3rd Embodiment of this invention. It is a block diagram of one Example of the vector control apparatus which shows the 4th Embodiment of this invention. It is an example of a function of the gain K that determines the values of the coefficient units 22A and 22B used in the present invention. It is a block diagram of one Example of the spindle control apparatus which shows the Example of a prior art.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 PWM inverter 2 Electric motor 3 Converter 4 Smoothing capacitor 5 Inverse conversion circuit 6A, 6B, 6C Current detector 7 PWM control part 10 Vector controller 11A, 11B Coordinate converter 12 Speed calculator 43, 13A, 13B Subtractor 14 Speed control Unit 15 Excitation current setter 16 Current control unit 17 Slip frequency command calculator 18 Adder 19, 19A Integrator 20 Same-dimensional magnetic flux observer 21, 21A Speed adaptive mechanism 22A, 22B Coefficient unit 23 Adder 24 Differentiator 25 Subtractor 26 Subtractor 27 switching SW
28 Multiplier 29 Adder 30 Mechanical system 31 Gear 32 Drum 33 Load 34 Positioning detector 35 Rope 36 Pulley 101 Three-phase two-phase converter 102 Integrator 103 Two-phase three-phase converter 104 Inverter 105 Induction motor 107 Main shaft 108 Rotation Position detector 109 Sensorless velocity detector 110 Position / velocity converter 111 High pass filter 112 Low pass filter 113 Adder 114 Comparator 115 Switcher 116 Limiter 117 Current controller 118 Magnetic flux density command generator 119 Excitation current command generator 120 Subtractor 121 PI amplifier 122, 123 Divider 124 Multiplier 125 Adder 126 DC voltage source 127 Current detector

Claims (6)

In the motor speed control device,
Speed detecting means for detecting the motor speed based on position or speed information input from an external detector;
An observer that computes and estimates the motor speed based on the state variable and the motor speed composite value output by the speed control device;
A current controller that determines a command current based on the speed feedback amount and the speed command value of the motor, using either one of a motor speed calculation estimated value or a motor speed composite value by the observer as a speed feedback amount,
When the motor speed is low, based on the speed detection value by the speed detection means, in the high speed range, based on a calculation estimated value of the motor speed by the observer, in the middle speed range, the speed detection value and the motor speed. A speed control device comprising speed synthesis means for outputting the motor speed synthesized value based on the calculated estimated value of the motor. In the induction motor speed control device,
Speed detecting means for detecting the motor speed based on position or speed information input from an external detector;
An observer that computes and estimates the motor speed based on the state variable and the motor speed composite value output by the speed control device;
A current controller that determines a command current based on the speed feedback amount and the speed command value of the motor, using either one of a motor speed calculation estimated value or a motor speed composite value by the observer as a speed feedback amount,
A slip frequency calculator for calculating and outputting the slip frequency;
When the motor speed is low, based on the speed detection value by the speed detection means, in the high speed range, based on a calculation estimated value of the motor speed by the observer, in the middle speed range, the speed detection value and the motor speed. A speed control device comprising speed synthesis means for outputting the motor speed synthesized value based on the calculated estimated value of the motor. In the motor speed control device,
Speed detecting means for detecting the motor speed based on position or speed information input from an external detector;
An observer that computes and estimates the phase and motor speed based on the state variable and the motor speed composite value output by the speed control device;
A slip frequency calculator for calculating and outputting the slip frequency;
A differentiator for differentiating the operation estimated value of the phase by the observer;
A first subtractor for subtracting the output of the slip frequency calculator from the output of the differentiator;
A current controller that determines a command current based on a speed command value of the motor and an output of the subtractor;
When the motor speed is low, based on the speed detection value by the speed detection means, in the high speed range, based on a calculation estimated value of the motor speed by the observer, in the middle speed range, the speed detection value and the motor speed. A speed control device comprising speed synthesis means for outputting the motor speed synthesized value based on the calculated estimated value of the motor. A first weight function unit that receives a speed detection value obtained by the detection means;
A second weight function unit that receives an operation estimated value of the motor speed by the observer as an input;
The speed synthesizing unit adds the output of the first weight function unit and the output of the second weight function unit, and uses the added value as a motor speed synthesis value in a medium speed range. The speed control device according to claim 1. The speed detection means detects a motor speed based on position or speed information inputted from an external detector on a load side connected to the motor via a power transmission means. The speed control device according to claim 4. A second subtracter for subtracting a speed detection value from the speed detection means from a speed command value;
An integrator for integrating the second subtractor output;
3. A slip frequency correction means for outputting a multiplication value obtained by multiplying the output of the slip frequency calculator by the output of the integrator as an output correction amount to the slip frequency unit. 3. The speed control device according to 3.

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