JP4680754B2 - DC brushless motor rotor angle estimation method and DC brushless motor control device - Google Patents

Description

  The present invention relates to a method and apparatus for estimating a rotor angle of a DC brushless motor by superimposing a high-frequency voltage on a driving voltage of the motor.

  In order to obtain a desired output torque from the DC brushless motor, it is necessary to apply a drive voltage to the armature at an appropriate phase corresponding to the electrical angle of the rotor having the magnetic pole (hereinafter referred to as the rotor angle). A method for detecting a rotor angle from a change in phase current when a high-frequency voltage is superimposed on a driving voltage of the motor so as to reduce the cost of the DC brushless motor and the driving device by omitting a position detection sensor for detecting the rotor angle. Has been proposed (see, for example, Patent Document 1 and Patent Document 2).

  In the previous application (Japanese Patent Application No. 2004-304492), the inventors of the present application also perform a predetermined control cycle including two or more control cycles when performing feedback control of the energization current of the DC brushless motor by a so-called dq vector conversion. A first-order difference of the motor d-axis current and q-axis current between adjacent control cycles is calculated by superimposing a periodic signal with the sum of output voltages in the cycle being zero on the motor drive voltage, and the first-order difference is calculated. Has proposed a method for calculating an estimated value of the rotor angle.

In this method, a sine reference value and a cosine reference value corresponding to a double angle of the phase difference between the actual value of the rotor angle and the estimated value are calculated using the calculated value of the first-order difference between the d-axis current and the q-axis current. The phase difference estimated value corresponding to the phase difference between the actual value of the rotor angle and the estimated value is calculated using the sine reference value. Then, based on the estimated phase difference value and the estimated value of the angular velocity of the previous motor, the estimated value of the rotor angle and the estimated angular velocity of the motor so as to eliminate the phase difference between the actual value and estimated value of the rotor angle. The estimated value of the rotor angle is updated by an observer that calculates the values while sequentially updating them.
JP 2000-125589 A JP 2002-262592 A

  When the estimated value of the rotor angle is calculated by using the above-described observer, the inventors of the present application may increase the phase difference between the actual value of the rotor angle and the estimated value, thereby increasing the estimation error of the rotor angle. I found out that there was. If the estimation error of the rotor angle becomes large in this way, the motor drive voltage set based on the estimated value of the rotor angle becomes inappropriate, and there is a disadvantage that the motor step-out occurs.

  SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide a DC brushless motor rotor angle estimation method and a DC brushless motor control device that solve the above problems and suppress an increase in estimation error of the rotor angle.

  The present invention has been made to achieve the above object, and a first aspect of a rotor angle estimation method for a DC brushless motor according to the present invention is a q brush which is a magnetic flux direction of a field of the motor. A current detection means for detecting a phase current flowing in the armature of the motor by converting it into an equivalent circuit having a q-axis armature on the axis and a d-axis armature on the d-axis orthogonal to the q-axis And a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the phase current detected by the current detection means and the estimated value of the rotor angle of the motor. A dq current calculation means that performs a deviation between a d-axis command current and a d-axis current that is a command value of a current flowing through the d-axis armature, and a q-axis command that is a command value of a current that flows through the q-axis armature Reduce the deviation between current and q-axis current In the motor control device that determines the d-axis voltage applied to the d-axis armature and the q-axis voltage applied to the q-axis armature for each predetermined control cycle and performs energization control of the motor, The present invention relates to an improvement in a method for estimating a rotor angle.

  Then, a first step of superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage, and a calculation based on changes in the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. , Using at least one of a sine reference value and a cosine reference value corresponding to a double angle of the phase difference between the estimated value and the actual value of the rotor angle of the motor, and the estimated value and the estimated value of the rotor angle of the motor A second step of calculating a phase difference estimated value corresponding to the phase difference from the value, and a second step of calculating an angular acceleration estimated value corresponding to the angular acceleration of the motor based on the d-axis command current or the d-axis current. 3, and the phase difference estimated value, the previous estimated value of the rotor angle of the motor, and the previous angular velocity of the motor so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. Based on the estimated value of the motor The estimated value of the motor angle is sequentially updated, and the estimated value of the angular velocity of the motor is sequentially calculated based on the estimated value of the phase difference, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor. And a fourth step of updating the estimated value of the rotor angle of the motor by an observer that is calculated while updating.

  According to the present invention, the output torque of the motor is proportional to the magnitude of the d-axis current, and it is assumed that the angular acceleration of the motor increases as the output torque of the motor increases. Then, the d-axis voltage and the q-axis voltage are determined by the energization control so as to reduce the deviation between the d-axis command current and the d-axis current. Therefore, in the third step, the estimated angular acceleration value can be calculated based on the d-axis command current or the d-axis current.

  Then, in the fourth step, the estimated value of the angular velocity of the motor is updated by the observer based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor. Thus, the angular velocity of the motor can be calculated by reflecting the influence of the angular acceleration. Therefore, by updating the estimated value of the rotor angle of the motor based on the estimated value of the phase difference and the estimated value of the previous angular velocity by the observer, the rotor of the motor is caused by the change in the angular velocity of the motor. It is possible to increase the estimation accuracy of the rotor angle by suppressing an increase in the phase difference between the actual value of the angle and the estimated value.

  Further, according to a first aspect of the control device for a DC brushless motor of the present invention, a DC brushless motor includes a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor, and d orthogonal to the q-axis. An equivalent circuit having a d-axis armature on the shaft is converted into an equivalent circuit, current detection means for detecting a phase current flowing in the armature of the motor, and rotor angle estimation for calculating an estimated value of the rotor angle of the motor And a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the phase current detected by the current detection means and the estimated value of the rotor angle of the motor. Dq current calculation means for calculating, a deviation between a d-axis command current and a d-axis current which is a command value of a current flowing through the d-axis armature, and a q-axis command current which is a command value of a current flowing through the q-axis armature To reduce the deviation between the q-axis current and the q-axis current And a DC power supply control means for determining a d-axis voltage applied to the d-axis armature and a q-axis voltage applied to the q-axis armature every predetermined control cycle, and for controlling the power supply of the motor. The present invention relates to an improvement of a brushless motor control device.

And high-frequency superimposing means for superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage, and a rotor of the motor based on changes in the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. Using at least one of a sine reference value and a cosine reference value corresponding to a double angle of the phase difference between the estimated value of the angle and the actual value, the phase difference between the actual value and the estimated value of the rotor angle of the motor An angle corresponding to the angular acceleration of the motor using a phase difference estimated value calculating means for calculating a phase difference estimated value corresponding to the motor and an estimated value of the motor output torque based on the d-axis command current or the d-axis current. Angular acceleration estimated value calculating means for calculating an acceleration estimated value, wherein the rotor angle estimating means is configured to cancel the phase difference between the actual value of the rotor angle of the motor and the estimated value, The previous mode Based on the estimated value of the rotor angle and the previous estimated value of the angular velocity of the motor, the estimated value of the rotor angle of the motor is calculated while being sequentially updated, and the estimated phase difference value and the estimated angular acceleration value Based on the previous estimated value of the angular velocity of the motor, the estimated value of the rotor angle of the motor is updated by an observer that sequentially calculates and updates the angular velocity of the motor.

  According to the present invention, the output torque of the motor is proportional to the magnitude of the d-axis current, and it is assumed that the angular acceleration of the motor increases as the output torque of the motor increases. Then, the d-axis voltage and the q-axis voltage are determined by the energization control means so as to reduce the deviation between the d-axis command current and the d-axis current. Therefore, the angular acceleration estimation means can calculate the angular acceleration estimated value based on the d-axis command current or the d-axis current.

  Then, the rotor angle estimation means updates the angular velocity of the motor based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor by the observer. The angular velocity of the motor can be calculated by reflecting the influence of acceleration. Therefore, the rotor angle estimating means updates the estimated value of the rotor angle of the motor based on the estimated value of the phase difference and the estimated value of the previous angular velocity by the observer, thereby changing the angular velocity of the motor. As a result, it is possible to suppress an increase in the phase difference between the actual value and the estimated value of the rotor angle of the motor, and to increase the estimation accuracy of the rotor angle.

  3. The control of a DC brushless motor according to claim 2, wherein the rotor angle estimation means updates the estimated value of the rotor angle of the motor and the angular velocity of the motor by an observer of the following formula (4). apparatus.

Where θ ^ (n + 1) is the updated value of the estimated value of the rotor angle, ω ^ (n + 1) is the updated value of the estimated value of the angular velocity of the motor, Δt is the elapsed time since the previous estimated value was calculated, θ ^ (n): Estimated value of previous rotor angle, ω ^ (n): Estimated value of angular velocity of previous motor, K1, K2: Calculation gain, θ _E1 (n): Estimated value of phase difference, A _ω ( n): Estimated angular acceleration.

According to the present invention, by the observer of the above equation (4), the previous estimated value of angular velocity ω ^ (n) is multiplied by the product of the phase difference estimated value θ _E1 (n) and the calculation gain K2, and the angle An updated value ω ^ (n + 1) of the estimated angular velocity value is calculated by adding the multiplication value of the estimated acceleration value A _ω (n) and the elapsed time Δt from the previous estimation of the rotor angle and angular velocity. In this case, the estimated value of the rotor angle of the motor is updated by eliminating the phase difference between the actual value and the estimated value of the rotor angle of the motor, and further adding the increase in angular velocity due to the angular acceleration of the motor. can do.

Then, the rotor angle estimation means uses the observer of the above formula (4) to multiply the estimated value θ ^ (n) of the previous rotor angle by the elapsed time Δt from the previous estimation of the rotor angle and angular velocity, The updated value θ ^ (n + 1) of the estimated value of the rotor angle is calculated by adding the multiplication value of the phase difference estimated value θ _E1 (n) and the calculation gain K1. In this case, since the previous angular velocity ω ^ (n) is calculated by reflecting the angular acceleration, the updated estimated rotor angle θ ^ (n + 1) includes the variation of the angular velocity due to the angular acceleration. Will be. Therefore, it is possible to increase the estimation accuracy of the rotor angle by suppressing an increase in the phase difference between the actual value and the estimated value of the rotor angle of the motor due to the influence of the angular acceleration of the motor.

  Further, the angular acceleration estimation means calculates the angular acceleration estimated value by the following equation (5).

Where A _ω (n): Estimated angular acceleration, T _t (n): Estimated value of motor output torque, I _e : Inertia of motor load, Id_c (n): d-axis command current, K _t : Motor Torque coefficient of

According to the present invention, when the output torque of the motor is T _t (n) and the inertia of the load of the motor is I _e , the operation of the motor is angular acceleration dω / dt = T _t / I _e. It is thought that it becomes the equiangular acceleration motion. Therefore, the observer of formula (4) can be configured by calculating the angular acceleration estimated value A _ω (n) by formula (5).

  Further, the angular acceleration estimation means calculates the angular acceleration estimated value by the following equation (6).

Where A _ω (n): angular acceleration estimated value, T _t (n): motor output torque estimated value, F _t (n): motor load friction torque estimated value, I _e : motor load Inertia, K _t : Motor torque coefficient, Id_c: d-axis command current, K _f : Motor load friction coefficient, ω ^ (n): Estimated value of previous motor angular velocity.

According to the present invention, the estimated angular acceleration value A _ω (n) is obtained by subtracting the friction torque F _t (n) of the motor load from the output torque T _t (n) of the motor according to the above equation (6). By calculating, the estimation accuracy of angular acceleration can be increased.

Further, characterized by comprising temperature detecting means for detecting the ambient temperature of the motor, and a torque coefficient correction means said correcting the torque coefficient K _t according to the detected temperature of the temperature detection means.

According to the present invention, in the above formula (5) or the formula (6), and correcting the torque coefficient K _t of the motor which change according to the temperature to the detected temperature of said temperature detecting means, said angle By calculating the acceleration estimated value A _ω (n), the estimation accuracy of the angular velocity estimated value ω ^ (n + 1) calculated by the above equation (4) can be increased. As a result, the estimation accuracy of the angular acceleration of the rotor angle calculated using the angular velocity ω ^ (n) in the above equation (4) can be improved.

Further, characterized by comprising temperature detecting means for detecting the ambient temperature of the motor, and a friction coefficient correcting means for correcting the friction coefficient K _f in accordance with the detected temperature of the temperature detection means.

According to the present invention, in the above equation (6), the friction coefficient K _f of the motor load that changes according to the temperature is corrected according to the temperature detected by the temperature detecting means, and the angular acceleration estimated value A By calculating _ω (n), it is possible to improve the estimation accuracy of the estimated angular velocity value ω ^ (n + 1) calculated by the above equation (4). Thus, the estimation accuracy of the estimated value θ ^ (n + 1) of the rotor angle calculated using the angular velocity ω ^ (n) in the above equation (4) can be improved.

  The motor is connected to an engine, and the engine is rotationally driven to start.

  According to the present invention, when the engine is rotationally driven by the motor and started, the angular velocity of the motor rapidly increases. Therefore, by updating the angular velocity of the motor using the estimated angular acceleration value, the phase difference between the actual value and the estimated value of the rotor angle of the motor is increased when the engine is started, and the motor step-out is reduced. Or the like can be prevented.

  The second aspect of the rotor angle estimation method for a DC brushless motor according to the present invention is such that the deviation between the phase current flowing through the armature of the DC brushless motor and the target current is reduced every predetermined control cycle. The present invention relates to an improvement in a method for estimating a rotor angle of a motor in a motor control apparatus that controls energization of the motor by determining a drive voltage to be applied to an armature of the motor using an estimated value of a rotor angle of the motor.

A first step of superimposing a high frequency voltage on the drive voltage; and a rotor angle of the motor calculated based on a change in phase current of the motor when the high frequency voltage is superimposed on the drive voltage. Using at least one of a sine reference value and a cosine reference value corresponding to a double angle of the actual value, a phase difference estimated value corresponding to a phase difference between the actual value and estimated value of the rotor angle of the motor is calculated. A phase difference between an actual value and an estimated value of the rotor angle of the motor, a second step of calculating an estimated angular acceleration value corresponding to the angular acceleration of the motor based on the target current, The estimated value of the rotor angle of the motor is sequentially updated based on the estimated value of the phase difference, the estimated value of the rotor angle of the previous motor, and the estimated value of the angular velocity of the previous motor. When calculated , On the basis of the phase difference estimated value and the estimated value of the angular acceleration estimated value and the previous said motor angular velocity, the observer for calculating while sequentially updating the estimated value of the angular velocity of the motor, the rotor angle of the motor And a fourth step of updating the estimated value.

  According to the present invention, the output torque of the motor is proportional to the magnitude of the phase current, and it is assumed that the angular acceleration of the motor increases as the output torque of the motor increases. Then, the drive voltage is determined by the energization control so as to reduce the deviation between the target current and the phase current. Therefore, in the third step, the angular velocity can be calculated based on the target current or phase current.

  Then, in the fourth step, the estimated value of the angular velocity of the motor is updated by the observer based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor. Thus, the angular velocity of the motor can be calculated by reflecting the influence of the angular acceleration. Therefore, by updating the estimated value of the rotor angle of the motor based on the estimated value of the phase difference and the estimated value of the previous angular velocity by the observer, the rotor of the motor is caused by the change in the angular velocity of the motor. It is possible to increase the estimation accuracy of the rotor angle by suppressing an increase in the phase difference between the actual value of the angle and the estimated value.

  According to a second aspect of the control device for a DC brushless motor of the present invention, there is provided current detection means for detecting a phase current flowing in the armature of the DC brushless motor, and rotor angle estimation for calculating an estimated value of the rotor angle of the motor. And an estimated value of the rotor angle so as to reduce the deviation between the phase current detected by the current detecting means and the predetermined target current for each predetermined control cycle. The present invention relates to an improvement of a control device for a DC brushless motor, which includes an energization control unit that determines an applied drive voltage and controls energization of the motor.

Then, a high frequency superimposing means for superimposing a high frequency voltage to the drive voltage, based on the change in the phase current of the motor when the front Symbol high frequency voltage is superimposed, depending on the double angle of the actual value of the rotor angle of the motor A phase difference estimated value calculating means for calculating a phase difference estimated value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor using at least one of the sine reference value and the cosine reference value And an angular acceleration estimated value calculating means for calculating an angular acceleration estimated value corresponding to the angular acceleration of the motor using an estimated value of the output torque of the motor based on the target current or phase current, and the rotor angle The estimation means is configured to eliminate the phase difference estimated value, the previous estimated value of the rotor angle of the motor, and the previous angular velocity of the motor so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. Based on the constant value, the estimated value of the rotor angle of the motor is calculated while being sequentially updated, and based on the estimated value of the phase difference, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor, The estimated value of the rotor angle of the motor is updated by an observer that calculates the angular velocity of the motor while sequentially updating it.

  According to the present invention, the output torque of the motor is proportional to the magnitude of the phase current, and it is assumed that the angular acceleration of the motor increases as the output torque of the motor increases. Then, the drive voltage is determined by the energization control means so as to reduce the deviation between the target current and the phase current. Therefore, the angular acceleration estimation means can calculate the estimated angular acceleration value based on the target current or phase current.

  Then, the rotor angle estimation means updates the angular velocity of the motor based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor by the observer. The angular velocity of the motor can be calculated by reflecting the influence of acceleration. Therefore, the rotor angle estimating means updates the estimated value of the rotor angle of the motor based on the estimated value of the phase difference based on the estimated value of the previous angular velocity by the observer, thereby changing the angular velocity of the motor. As a result, it is possible to suppress an increase in the phase difference between the actual value and the estimated value of the rotor angle of the motor, and to increase the estimation accuracy of the rotor angle.

  Embodiments of the present invention will be described with reference to FIGS. FIG. 1 is a configuration diagram of a DC brushless motor, FIG. 2 is a configuration diagram of a motor control device in the first embodiment, FIG. 3 is an explanatory diagram of an output mode of a periodic signal in the motor control device shown in FIG. Is a flowchart of the rotor angle estimation process in the motor control device shown in FIG. 2, FIG. 5 is a timing chart corresponding to the flowchart of FIG. 4, FIG. 6 is a block diagram of the motor control device in the second embodiment, and FIG. 7 is a flowchart of a rotor angle estimation process in the motor control device shown in FIG. 6.

  [First Embodiment] First, the first embodiment of the present invention will be described with reference to FIGS. The DC brushless motor control device 40 (hereinafter simply referred to as the control device 40) shown in FIG. 2 is a three-phase (U, V, W) of the DC brushless motor 1 (hereinafter simply referred to as the motor 1) shown in FIG. Feedback control of the current flowing through the armatures 3, 4 and 5 of the motor 1, the motor 1 includes a q-axis armature on the q-axis which is the magnetic flux direction of the field pole of the rotor 2, and a d-axis orthogonal to the q-axis. It is converted into an equivalent circuit based on a dq coordinate system having a d-axis armature located above.

  The control device 40 is provided in a hybrid vehicle that rotationally drives drive wheels (not shown) by the motor 1 and the engine 6 connected to the motor 1, and the motor 1 rotationally drives the engine when the engine 6 is started. It also functions as a starter. Further, the control device 40 determines a current flowing in the d-axis armature (hereinafter referred to as a d-axis current) and a current flowing in the q-axis armature according to the d-axis command current Id_c and the q-axis command current Iq_c given from the outside. (Hereinafter referred to as q-axis current) is feedback controlled.

The control device 40 determines the rotor angle based on the voltage Vfb _d applied to the d-axis armature (hereinafter referred to as d-axis voltage Vfb _d ) and the voltage Vfb _q applied to the q-axis armature (hereinafter referred to as q-axis voltage Vfb _q ). A periodic signal superimposing unit 51 (corresponding to the high-frequency superimposing means of the present invention) for superimposing the estimation periodic signals Vh _d and Vh _q (corresponding to the high-frequency voltage of the present invention), a d-axis voltage Vfb _d and a q-axis voltage Vfb. _q (when periodic signals Vh _d and Vh _q is superimposed, Vfb _d + Vh _d and Vfb _q + Vh _q) a, U of the motor 1, V, drive voltage applied to the 3-phase armatures of W Vfb _u , Vfb _v , Vfb _w , dq / 3-phase converter 20, and voltages Vu, Vv, Vw corresponding to Vfb _u , Vfb _v , Vfb _w are the armatures of the U, V, W phases of the motor 1. The power drive unit is composed of an inverter circuit configured by bridge-connecting a plurality of switching elements. A knit 22 is provided.

The control device 40 further includes a U-phase current sensor 23 (corresponding to current detection means of the present invention) that detects a current flowing through the U-phase armature of the motor 1, and a current flowing through the W-phase armature of the motor 1. W-phase current sensor 24 for detecting (corresponding to the current detecting means of the present invention), U-phase current detected current value I _u and W detected current value of the phase current sensor 24 I _w and d-axis current I in response to the sensor 23 A three-phase / dq converter 26 (corresponding to the dq current calculator of the present invention) for calculating _d and the q-axis current I _q, and an angle estimator 50 for calculating an estimated value θ ^{^} of the rotor angle θ of the motor 1 ( Including the functions of the rotor angle estimating means, phase difference estimated value calculating means, and angular acceleration estimated value calculating means of the present invention), and non-performing processing for canceling the influence of the speed electromotive force that interferes between the d axis and the q axis. An interference calculation unit 27 is provided.

Then, the control unit 40 subtracts the d-axis command current Id_c and the d-axis current I _d at first subtractor 28 performs PI (proportional integral) processing by the first PI calculation portion 29 on the subtraction result, the by adding the non-interference component α 1 adder 30, the following equation (7) for generating a d-axis voltage Vfb _d corresponding to the deviation of the d-axis command current Id_c and the d-axis current I _d.

  However, Kp: proportional gain, Ki: integral gain, α: non-interference component.

Similarly, the control device 40 subtracts the q-axis command current Iq_c and the q-axis current _Iq by the second subtractor 31, performs a PI process on the subtraction result by the second PI calculator 32, by adding the non-interference component β 2 adder 33 generates the q-axis voltage Vfb _q corresponding to a deviation between the q-axis command current Iq_c and the q-axis current I _q by the following equation (8).

  However, Kp: proportional gain, Ki: integral gain, β: non-interference component.

Then, the control device 40 inputs the d-axis voltage Vfb _d and the q-axis voltage Vfb _q to the dq / 3-phase conversion unit 20. The dq / 3-phase conversion unit 20 calculates three-phase drive voltages Vfb _u , Vfb _v , and Vfb _{w according} to the following equation (9) and outputs them to the power drive unit 22.

Thus, via the power drive unit 22, the d-axis command current Id_c and the d-axis current I _d and deviation, and q-axis command current Iq_c and the q-axis current I _q and 3-phase driving voltage Vfb _u of the deviation to reduce the , Vfb _v , Vfb _w are applied to the armature of the motor 1, and the current flowing through the armature of the motor 1 is feedback controlled.

In this manner, so as to reduce the deviation between the d-axis command current Id_c and the d-axis current I _d, and the deviation between the q-axis command current Iq_c and the q-axis current I _q, the 3-phase drive voltages Vfb _u , Vfb _v , Vfb _w is determined to correspond to the energization control means of the present invention.

Here, when the _d- q / 3-phase converter 20 converts the d-axis driving pressure Vfb _d and the q-axis voltage Vfb _q into three-phase driving voltages Vfb _u , Vfb _v , Vfb _w , the rotor angle θ of the motor 1 Is required. Further, even when converting the detected current value Iw of the detected current values Iu and the W-phase current sensor 24 of the U-phase current sensor 23 to d-axis current I _d and the q-axis current I _q by 3-phase / dq converter 26, The rotor angle θ of the motor 1 is required.

Then, the control device 10 does not use a position detection sensor such as a resolver, but in the third adder 52 and the fourth adder 53, the periodic signal superimposing unit 51 converts the periodic signals Vh _d to the d-axis voltages Vfb _d and Vfb _q. , Vh _q are superimposed on each other to execute the rotor angle θ estimation process.

Note that the periodic signals Vh _d and Vh _q in the present embodiment are as follows: 3 control cycles (T ₁₁ , T ₁₂ , T ₁₃ ) of the control device 10 are 1 period as shown in FIG. The output pattern is set so that the sum of the output voltages of each phase (Vh _d , Vh _q ) becomes zero in a cycle. As another output pattern, for example, as shown in FIG. 3B, four control cycles (T ₂₁ , T ₂₂ , T ₂₃ , T ₂₄ ) are defined as one period, and two control cycles ( sum of T ₂₁ and T _22, T ₂₃ and T ₂₄₎ the output voltage for each may be set to be zero. Further, the length of one cycle of the periodic signals Vh _d and Vh _q may be two or more control cycles.

Here, when the length of one cycle of the periodic signals Vh _d and Vh _q is n control cycles, a sine reference value corresponding to the double angle of the phase difference θ _e between the actual value θ of the rotor angle and the estimated value θ ^. Vs _dq, cosine reference value Vc _dq and inductance reference value L0 can be calculated by the following equations (10) to (14).

Where Vs _{dq is a} sine reference value, Vc _{dq is a} cosine reference value, L0 is an inductance reference value, dI _dq (i + 1) to dI _dq (i + n), and the sum of output voltages of periodic signals Vh _d and Vh _q. When n control cycles are included in a period in which zero is zero, a first-order difference between d-axis current and q-axis current between adjacent control cycles, i: sine reference value Vs _dq and cosine reference value Vc _dq Number of the calculation cycle of the inductance reference value L0.

Where L _{d is} the inductance of the d-axis armature, and L _{q is} the inductance of the q-axis armature.

However, j = 1,2, ..., n , dI d (i + j): 1 -order difference of the current flowing in the d-axis _{armature, dI q (i + j)} : 1 Floor of the current flowing in the q-axis armature Difference.

Where Vh _d (i + j): the output voltage to the d-axis armature of the j-th control cycle during the period when the sum of the output voltages of the periodic signals Vh _d and Vh _q is zero, Vh _q (i + j) : The output voltage to the q-axis armature of the j-th control cycle during the period when the sum of the output voltages of the periodic signals Vh _d and Vh _q is zero.

Hereinafter, the process of estimating the rotor angle of the motor 1 in the control device 40 will be described with reference to the flowchart shown in FIG. The control device 40 sets the counter variable ptr to 0, and the periodic signal superimposing unit 51 superimposes the periodic signals Vh _d and Vh _q on the d-axis voltage Vfb _d and the q-axis voltage Vfb _q , and repeatedly executes the flowchart of FIG. To do.

The angle estimation unit 50 takes in the phase currents I _u and I _w detected by the U-phase current sensor 23 and the W-phase current sensor 24 in STEP 30, and in STEP 31, 1 of the phase currents I _u and I _w in the previous control cycle. The floor differences dI _u and dI _w are calculated. Then, in the next STEP 32, dI _d and dI _q obtained by three-phase / dq conversion of the first-order differences dI _u and dI _w are calculated and held.

  If the counter variable ptr is 1 in the next STEP 33, the process proceeds to STEP 34 and the processes of STEP 34 to STEP 37 are performed. If the counter variable ptr is not 1 in STEP 33, the process branches to STEP 38.

Here, the counter variable ptr is incremented for each control cycle in STEP43, and cleared in STEP45 (ptr = 0) when ptr = 3 in STEP44. Therefore, every time three control cycles corresponding to one period of the periodic signals Vh _d and Vh _q elapse, ptr = 1 is set at STEP33, and the processing of STEP34 to STEP37 is executed.

In STEP 34, the angle estimation unit 50 calculates the sine reference value Vs _dq and the cosine reference value Vc _dq using the first-order differences dI _d and dI _q calculated in STEP 32 of the current, previous and previous control cycles. Specifically, the angle estimator 50 is expressed by the following equation (15) in which n is replaced with 3 which is the number of control cycles included in one cycle of the periodic signals Vh _d and Vh _{q in} the above equation (10). Calculates a sine reference value Vs _{dq, a} cosine reference value Vc _dq, and an inductance reference value L0.

However, in a control cycle in which Vs _{dq is a} sine reference value, Vc _{dq is a} cosine reference value, L0 is an inductance reference value, and dI _dq (i + 1) to dI _dq (i + 3) are ptr = 2, 0, 1. First-order difference between the calculated d-axis current and q-axis current.

Here, by calculating the sine reference value Vs _{dq, the} cosine reference value Vc _dq and the inductance reference value L0 using the first-order difference between the d-axis current and the q-axis current according to the above equation (15), The effect of reducing the influence of detection errors of the U-phase current sensor 23 and the W-phase current sensor 24 can be expected as compared with the case where the second-order difference of the q-axis current is used.

Then, in the next STEP 35, the angle estimation unit 50 determines the actual value θ of the rotor angle and the estimated value θ ^ of the rotor angle used for the conversion processing in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26. The phase difference estimated value θ _E1 corresponding to the phase difference θ _e is calculated by the following equation (16). The configuration in which the angle estimation unit 50 calculates the phase difference estimated value θ _E1 by the equation (16) corresponds to the phase difference estimated value calculation means of the present invention.

Where θ _E1 : phase difference estimated value, Vs _dq : sine reference value, L 0: inductance reference value, θ _e : phase difference between the actual value θ of the rotor angle and the estimated value θ ^.

The above equation (16), a normalization process of dividing the sine reference value Vs _dq inductance reference value L0, by suppressing fluctuation in level due to positive and negative phase difference theta _e, in accordance with the phase difference theta _e A phase difference estimated value θ _E1 that changes within a predetermined range can be calculated. Then, in STEP 40, the angle estimation unit 50 updates and calculates the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 by the follow-up calculation by the observer of the following equation (17).

Where θ ^ (n + 1) is the updated value of the estimated value of the rotor angle, ω ^ (n + 1) is the updated value of the estimated value of the angular velocity of the motor, Δt is the elapsed time since the previous estimated value was calculated, θ ^ (n): Estimated value of previous rotor angle, ω ^ (n): Estimated value of angular velocity of previous motor, K1, K2: Calculation gain, θ _E1 (n): Estimated value of phase difference, A _ω ( n): Estimated angular acceleration.

In the above equation (17), by correcting the phase difference estimated value θ _E1 by adding an offset value proportional to the d-axis command current Id_c or the d-axis current Id, the estimated value of the rotor angle by the above equation (17). , The convergence point of the phase difference between the actual value and estimated value of the rotor angle can be shifted to a point closer to zero. Thereby, the estimation accuracy of the rotor angle can be increased.

Here, the output torque of the motor 1 is proportional to the d-axis current Id, and the voltage applied to the d-axis armature so that the deviation between the d-axis command current Id_c and the d-axis current Id is reduced by the energization control described above. Is controlled. As the output torque of the motor 1 increases, the angular acceleration of the motor 1 increases, and the load on the motor 1 (the motor 1 itself, the engine 6 connected to the motor 1, the power transmission mechanism connected to the motor 1 (not shown)). The greater the inertia of the total load is, the smaller the angular acceleration of the motor 1 is. Therefore, the rotor angle estimation unit 50 calculates an angular acceleration estimated value A _ω (n) corresponding to the angular acceleration of the motor 1 by the following equation (18).

Where A _ω (n): Estimated angular acceleration, T _t (n): Estimated value of motor output torque, I _e : Inertia of motor load, Id_c (n): d-axis command current, K _t : Motor Torque coefficient of

Then, the estimated value ω ^ (n + 1) of the angular velocity of the motor 1 is updated using the estimated angular acceleration value A _ω (n) by the observer of the above equation (17), and the estimated value ω ^ (n) of the angular velocity is updated. By updating the estimated value θ ^ (n + 1) of the rotor angle using, the influence of the change in the angular velocity of the motor 1 can be suppressed and the rotor angle can be estimated. Thereby, it is possible to suppress an increase in the phase difference between the actual value and the estimated value of the rotor angle due to a change in the angular velocity of the motor 1, and to increase the estimation accuracy of the rotor angle.

  As another method for suppressing the increase in the phase difference between the actual value and the estimated value of the rotor angle, it is conceivable to increase the calculation gains K1 and K2 in the above equation (17). The influence of an error when the detection signals of the U-phase current sensor 23 and the W-phase current sensor 24 are taken into the three-phase / dq converter 26 by A / D conversion becomes large, and the calculation result of the estimated value of the rotor angle is vibrated. There is an inconvenience of becoming.

Further, instead of the above equation (18), the angular acceleration estimated value A _ω (n) may be calculated by the following equation (19). Equation (19) takes into account the frictional resistance of the load of the motor 1 (the total load of the motor 1 itself, the engine 6 connected to the motor 1, the power transmission mechanism (not shown) connected to the motor 1, etc.). Compared with the case of the above equation (18), the estimation accuracy of the rotor angle can be further improved.

Where A _ω (n): estimated angular acceleration value, T _t : estimated value of motor output torque, F _t : estimated value of friction torque of motor load, I _e : inertia of motor load, K _t : motor Torque coefficient, Id_c: d-axis command current, K _f : friction coefficient of motor load, ω ^ (n): estimated value of the previous motor angular velocity.

Next, in STEP 36, I _d and I _q calculated by the three-phase / dq converter 26 are held for current feedback control. In STEP 37, the estimated value θ ^ of the rotor angle used in the three-phase / dq conversion unit 26 and the dq / 3-phase conversion unit 20 is updated.

When the counter variable ptr becomes 0 in the next STEP 38, the process proceeds to STEP 39, and the d-axis voltage Vfb _d and the q-axis voltage Vfb _q are calculated by the above formulas (7) and (8). In STEP 40, the angle estimating unit 50 updates the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 by the above equation (17). On the other hand, if ptr is not zero in STEP38, the process branches to STEP41, and the processing of STEP39 to STEP40 is not executed.

Therefore, the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _q and the estimated value θ ^ of the rotor angle are updated every period (every three control cycles) of the periodic signals Vh _d and Vh _q , and the periodic signals Vh _d and Vh. During one cycle of _q , the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _q, and the estimated value θ ^ of the rotor angle are kept constant. Further, as described above, the estimated value θ ^ of the rotor angle used in the three-phase / dq converter 26 and the dq / 3-phase converter is updated in STEP 37 when ptr = 1. Therefore, the estimated value θ ^ of the rotor angle updated in STEP 40 is used in the dq / 3-phase converter 20 and the 3-phase / dq converter 26 from the next control cycle.

In this way, the update of the estimated value θ ^ of the rotor angle for dq conversion is delayed because the estimated value θ ^ of the rotor angle at the time when the d-axis voltages Vfb _d and Vfb _q are calculated in STEP 39, and STEP 32 it is to match the estimated value of the rotor angle theta ^ at the time of calculating the first-order difference dI _q of the first-order difference dI _d and q-axis current of the d-axis current. By matching the estimated value θ ^ of the rotor angle in STEP39 and STEP32, it is possible to prevent an error due to a change in the estimated value θ ^ of the rotor angle when the differential currents dI _d and dI _{q are} calculated.

In subsequent STEP 41, the periodic signal superimposing unit 51 calculates the periodic signals Vh _d and Vh _q to the d-axis voltage Vfb _d and the q-axis voltage Vfb _q , and in STEP 42, the dq / three-phase conversion unit 20 calculates the d-axis voltages Vfb _d and q The shaft voltage Vfb _q is converted into a three-phase drive voltage Vfb _u , Vfb _v , Vfb _w and output to the power drive unit 22. In subsequent STEP44 to STEP46, the counter variable ptr described above is incremented or cleared, and one control cycle is completed.

The process of superimposing the periodic signals Vh _d and Vh _q by the periodic signal superimposing unit 21 in STEP 41 corresponds to the first step of the present invention. In STEP 34, the sine reference value Vs _{dq, the} cosine reference value Vc _dq, and the inductance reference value The process of calculating L0 and calculating the phase difference estimated value θ _E1 in STEP 35 corresponds to the second step of the present invention. In STEP 40, the process of calculating the angular acceleration estimated value A _ω (n) of the above equation (17) corresponds to the third step of the present invention, and the estimated value θ ^ of the rotor angle is calculated by the above equation (17). The process of updating (θ ^ (n) → θ ^ (n + 1)) and updating the estimated angular velocity ω ^ of the motor 1 (ω ^ (n) → ω ^ (n + 1)) is the present invention. This corresponds to the fourth step.

FIG. 5 is a timing chart when the estimated value θ ^ of the rotor angle is calculated by repeatedly executing the process according to the flowchart of FIG. 4 described above. Processing is started in cycle 50, and three control cycles (cycles 51, 52, 60, cycles 61, 62, 70,...) Corresponding to ptr = 1, 2, 0 are generated by periodic signals Vh _dq (Vh _d , Vh _q ) And the periodic signal is Vh _dq (0) (when ptr = 1) → Vh _dq (1) (when ptr = 2) → Vh _dq (2) (when ptr = 0) It has been switched.

Since the d-axis voltage Vfb _d and the q-axis voltage Vfb _q (hereinafter collectively referred to as dq voltage Vfb _dq ) are calculated in the control cycle in which ptr = 0 (step 39 in FIG. 4), the period While the signal Vh _dq switches from Vh _dq (0) → Vh _dq (1) → Vh _dq (2), the dq voltage Vfb _dq is constant (Vfb _dq (0) during cycles 51, 52 and 53, cycle 61). , 62 and 63 are maintained at Vfb _dq (1)).

Then, in control cycle 61 and control cycle 71 in which ptr = 1, a phase difference estimated value θ _E1 corresponding to the phase difference θ _e between the actual value and the estimated value of the rotor angle is calculated (processing of STEP 35 in FIG. 4). At the same time, the estimated value θ ^ of the rotor angle used in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26 is updated (processing of STEP 37 in FIG. 4). In addition, in the control cycle 61 and the control cycle 71 where ptr = 0, the dq voltage Vfb _dq and the estimated value θ ^ of the rotor angle are updated (processing of STEP39 to STEP40 in FIG. 4). In this way, the updating of the dq voltage Vfb _{dq and} the updating of the estimated value θ ^ of the rotor angle are performed at the same timing, thereby eliminating the rotor angle deviation when updating the dq voltage Vfb _dq .

Further, in the figure, c indicates changes in the d-axis current and the q-axis current, and d indicates the response of feedback control of the d-axis current and the q-axis current. The influence of feedback control of the d-axis current and the q-axis current due to the superposition of the periodic signal Vh _dq is canceled by the dq voltage Vfb _dq calculated at the timings P ₁₂ and P _{13 in} the figure.

In the first embodiment, the sine reference value Vs _dq and the cosine reference value Vc _dq and the inductance reference value L0 are calculated by the above equation (15), but the sine reference value Vs _dq and the cosine reference value Vc _dq The calculation formula of the inductance reference value L0 is determined according to the mode of the periodic signal Vh _dq .

In the first embodiment, the angular acceleration estimated value A _ω (n) is calculated using the d-axis command current Id_c (n) by the above formulas (18) and (19). The angular acceleration estimated value A _ω (n) may be calculated using the d-axis current Id instead of the command current Id_c.

In the first embodiment, the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 are updated by the observer of the above equation (17), but the d-axis command current Id_c or the d-axis current is updated. Using the observer that updates the angular velocity estimate ω ^ using the angular acceleration estimate A _ω (n) calculated using, and updates the rotor angle estimate θ ^ using the angular velocity estimate ω ^, The present invention can be applied if the estimated value of the rotor angle is updated.

  [Second Embodiment] Next, a second embodiment of the present invention will be described with reference to FIGS. The DC brushless motor control device 10 (hereinafter simply referred to as the control device 10) shown in FIG. 12 performs feedback control of the phase current of the motor 1 in the same manner as the control device 40 shown in FIG. The same components as those in FIG.

In the control apparatus 10, a method of estimating the rotor angle is different from the controller 40, the driving voltage Vfb _u and Vfb _w to the rotor angle estimation of a periodic signal Vh _u and Vh _w (corresponding to a high frequency voltage of the present invention) Functions of superimposing periodic signal superimposing unit 21 (corresponding to high frequency superimposing unit of the present invention) and angle estimating unit 25 (rotor angle estimating unit, phase difference estimated value calculating unit, and angular acceleration estimated value calculating unit of the present invention) Including).

Then, the control unit 10, in a fifth adder 34 sixth adder 36, the periodic signal superimposing section 21 superimposes the periodic signal Vh _u to the drive voltage Vfb _u, a periodic signal Vh _w to the drive voltage Vfb _w The estimated value θ ^ of the rotor angle is calculated by superimposing. The output signals of the periodic signals Vh _u and Vh _w are set so that the total sum of the output voltages of the respective phases is zero in one period, with three control cycles of the motor control device 10 as one period. The length of one cycle of the periodic signals Vh _u and Vh _w may be two control cycles or more.

Here, when the length of one cycle of the periodic signals Vh _u and Vh _w is n control cycles, the sine reference value Vs, the cosine reference value, and the inductance reference value corresponding to the double angle of the rotor angle θ (actual value). L _lm can be calculated by the following equations (20) to (24).

Where Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, l: average value of self-inductance of each phase of motor, m: average value of mutual inductance between each phase of motor, dI (i +1) to dI (i + n) ... of phase currents between adjacent control cycles when n control cycles are included in a period in which the sum of the output voltages of the periodic signals Vh _u and Vh _w is zero First-order difference, i: number of calculation cycle of sine reference value Vs and cosine reference value Vc and inductance reference value _Llm .

  Where θ is the actual value of the rotor angle of the motor.

Here, j = 1, 2,..., N, dI _u (i + j): first-order difference of U-phase phase current, dI _w (i + j): first-order difference of W-phase phase current.

Where Vh _uv (i + j) is the voltage between the U-phase and V-phase of the j-th control cycle during the period when the sum of the output voltages of the periodic signal is zero, and Vh _wv (i + j) is the output of the periodic signal. The interphase voltage between the W phase and the V phase of the jth control cycle in a period in which the sum of the voltages is zero.

Hereinafter, the process of estimating the rotor angle of the motor 1 in the control device 10 will be described with reference to the flowchart shown in FIG. The control device 10 sets the initial value of the counter variable ptr to 0, superimposes the periodic signals Vh _u and Vh _w on the drive voltages Vfb _u and Vfb _w by the periodic signal superimposing unit 21, and repeatedly executes the flowchart of FIG.

Angle estimator 25, phase currents I _u detected by the U-phase current sensor 23 and the W-phase current sensor 24 in STEP1, captures I _w, the phase current I _u in the previous control cycle in STEP2, 1 and I _w Calculate and hold the floor differences dI _u and dI _w . If the counter variable ptr is 1 in the subsequent STEP3, the process proceeds to STEP4 and the processes of STEP4 to STEP6 are performed. If the counter variable ptr is not 1 in STEP3, the process branches to STEP7.

Here, the counter variable ptr is incremented for each control cycle in STEP12, and cleared in STEP14 (ptr = 0) when ptr = 3 in STEP13. Therefore, every time three control cycles corresponding to one cycle of the periodic signals Vh _u and Vh _w elapse, ptr = 1 is set at STEP 3 and the processing of STEP 4 to STEP 6 is executed.

In STEP 4, the angle estimation unit 25 uses the first-order differences dI _u and dI _w calculated in STEP 2 in the current, previous and previous control cycles, and uses the sine reference value Vs, the cosine reference value Vc, and the inductance reference value. _Llm is calculated. Specifically, the angle estimator 25 is obtained by the following equation (25) in which n is replaced by 3 in the above equation (20), which is the number of control cycles included in one cycle of the periodic signals Vh _u and Vh _w. Calculates a sine reference value Vs, a cosine reference value Vc, and an inductance reference value _Llm .

Where Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, l: average value of self-inductance of each phase of motor, m: average value of mutual inductance between angular phases of motor, dI (i +1) to dI (i + 3): the first-order difference of the phase currents calculated in the control cycle where ptr = 2, 0, 1.

Here, when the second-order difference of the phase current is used by calculating the sine reference value Vs, the cosine reference value Vc, and the inductance reference value L _lm using the first-order difference of the phase current by the above equation (25). In comparison, an effect of reducing the influence of detection errors of the U-phase current sensor 23 and the W-phase current sensor 24 can be expected.

The angle estimation unit 25 then compares the phase difference θ _e between the actual value θ of the rotor angle and the estimated value θ ^ of the rotor angle used for the conversion in the dq / 3-phase conversion unit 20 and the 3-phase / dq conversion unit 26. The phase difference estimated value θ _E2 corresponding to the above is calculated by the following equation (26). The configuration in which the angle estimation unit 25 calculates the phase difference estimated value θ _E2 by the equation (26) corresponds to the phase difference estimated value calculation means of the present invention.

However, θ _E2 : phase difference estimated value, Vs: sine reference value, Vc: cosine reference value, L _lm : inductance reference value, θ: actual value of rotor angle, θ ^: estimated value of rotor angle, l: motor value Average value of self-inductance of each phase, m: average value of mutual inductance between phases of motor, θ _e : phase difference between actual value θ of rotor angle and estimated value θ ^.

The above equation (26), a normalization process that divides the Vs · cos2θ ^ -Vc · sin2θ ^ inductance reference value L _lm, by suppressing the variation in level due to positive and negative phase difference theta _e, the phase difference A phase difference estimated value θ _E2 that changes within a predetermined range in accordance with θ _e can be calculated. Then, in STEP 9, the angle estimation unit 25 updates and calculates the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 by the follow-up calculation by the observer of the following equation (27).

Where θ ^ (n + 1) is the updated value of the estimated value of the rotor angle, ω ^ (n + 1) is the updated value of the estimated value of the angular velocity of the motor, Δt is the elapsed time since the previous estimated value was calculated, θ ^ (n): Estimated value of previous rotor angle, ω ^ (n): Estimated value of angular velocity of previous motor, K1, K2: Calculation gain, θ _E2 (n): Estimated phase difference, A _ω ( n): Estimated angular acceleration.

In the above equation (27), by correcting the phase difference estimated value θ _E2 by adding an offset value proportional to the d-axis current, the rotor angle when the estimated value of the rotor angle is updated by the above equation (27). The convergence point of the phase difference between the actual value and the estimated value can be shifted to a point closer to zero. Thereby, the estimation accuracy of the rotor angle can be increased. Further, the angular acceleration estimated value A _ω (n) is calculated by the above formula (17) or the above formula (18) as in the first embodiment described above.

Next, in STEP 6, I _d and I _q calculated by the three-phase / dq converter 26 are held for current feedback control. Then, when the counter variable ptr becomes 0 in STEP 7, the process proceeds to STEP 8, and the d-axis voltage Vfb _d and the q-axis voltage Vfb _q are calculated by the above formulas (7) and (8). Further, in STEP 9, the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 are updated by the above equation (27). In this way, by updating the drive voltages Vfb _d and Vfb _q and updating the estimated value θ ^ of the rotor angle at the same timing, deviation of the rotor angle when outputting the drive voltages Vfb _d and Vfb _q is eliminated. ing.

On the other hand, if ptr = 0 is not satisfied in STEP 7, the process branches to STEP 10, and the processing of STEP 8 to STEP 9 is not executed. Therefore, the d-axis voltage Vfb _d and the q-axis voltage Vfb _q, estimated angle theta ^ are periodic signals Vh _u, Vh _v, is updated every cycle of the Vh _w, periodic signals Vh _u, Vh _v, the Vh _w During one period, the d-axis voltage Vfb _{d, the} q-axis voltage Vfb _d, and the estimated angle θ ^ are kept constant.

In STEP 10, the dq / 3-phase converter 20 converts the d-axis voltage Vfb _d and the q-axis voltage Vfb _q into three-phase drive voltages Vfb _u , Vfb _v , Vfb _w , and in STEP 11, the periodic signal superimposing unit 21. Periodic signals Vh _u , Vh _v , Vh _w are superimposed on the three-phase drive voltages Vfb _u , Vfb _v , Vfb _w and output to the power drive unit 22. As a result, three-phase voltages Vu, Vv, and Vw are output from the power drive unit 22 to the motor 1. Then, in the subsequent STEP12 to STEP15, the counter variable ptr described above is incremented or cleared, and one control cycle is completed.

In STEP 11, the process of superimposing the periodic signals Vh _u , Vh _v , and Vh _w by the periodic signal superimposing unit 21 corresponds to the first step of the present invention. In STEP 4, the sine reference value Vs, the cosine reference value Vc, and the inductance The process of calculating the reference value L _lm and calculating the phase difference estimated value θ _E2 in STEP 5 corresponds to the process of the second process of the present invention. In STEP 9, the process of calculating the angular acceleration estimated value A _ω (n) of the above equation (27) corresponds to the third step of the present invention, and the estimated value θ ^ of the rotor angle is calculated by the above equation (27). The process of updating (θ ^ (n) → θ ^ (n + 1)) and updating the estimated angular velocity ω ^ of the motor 1 (ω ^ (n) → ω ^ (n + 1)) is the present invention. This corresponds to the fourth step.

In the second embodiment, the sine reference value Vs, the cosine reference value Vc, and the inductance reference value _Llm are calculated by the above equation (20). However, the sine reference value Vs, the cosine reference value Vc, and the inductance reference value L are calculated. The calculation formula of _lm is determined according to the mode of the periodic signal Vh.

In the second embodiment, the angular acceleration estimated value A _ω (n) is calculated using the d-axis command current Id_c (n) by the above formulas (18) and (19). The angular acceleration estimated value A _ω (n) may be calculated using the d-axis current Id instead of the command current Id_c.

In the second embodiment, the estimated value θ ^ of the rotor angle and the estimated value ω ^ of the angular velocity of the motor 1 are updated by the observer of the above equation (27), but the d-axis command current Id_c or the d-axis current is updated. Using the observer that updates the angular velocity estimate ω ^ using the angular acceleration estimate A _ω (n) calculated using, and updates the rotor angle estimate θ ^ using the angular velocity estimate ω ^, The present invention can be applied if the estimated value of the rotor angle is updated.

  In the first embodiment and the second embodiment, the sine reference value, the cosine reference value, and the inductance reference value are calculated using the first-order difference of the detected current. However, the second-order difference of the detected current is calculated. The present invention can also be applied to the case where the sine reference value, cosine reference value, and inductance reference value are calculated using.

In the first embodiment and the second embodiment, the angular velocity estimated value ω ^ is calculated from the angular acceleration estimated value A _ω (n) according to the equations (17) and (27). The effect of the update is particularly great when the engine 6 is rotated by the motor 1 to start the engine 6 or when the rotational speed of the motor 1 is low, such as when the hybrid vehicle is traveling at a low speed. Become. Therefore, the angular velocity estimated value ω ^ may be updated using the angular acceleration estimated value A _ω (n) only when the rotational speed of the motor 1 is equal to or lower than the predetermined rotational speed.

The block diagram of DC brushless motor. The block diagram of the motor control apparatus in 1st Embodiment. Explanatory drawing of the output aspect of the periodic signal in the motor control apparatus shown in FIG. The flowchart of the estimation process of the rotor angle in the motor control apparatus shown in FIG. The timing chart corresponding to the flowchart of FIG. The block diagram of the motor control apparatus in 2nd Embodiment. 7 is a flowchart of rotor angle estimation processing in the motor control device shown in FIG. 6.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... DC brushless motor, 2 ... Rotor, 3 ... U-phase armature, 4 ... V-phase armature, 5 ... W-phase armature, 10 ... Motor control device (of the first embodiment), 20 ... dq / 3-phase conversion unit, 21 ... periodic signal superposition unit (of the first embodiment), 22 ... power drive unit, 23 ... U-phase current sensor, 24 ... W-phase current sensor, 25 ... (first embodiment) Angle estimation unit, 26 ... three-phase / dq conversion unit, 27 ... non-interference calculation unit, 40 ... (second embodiment) motor control device, 50 ... (second embodiment) Angle estimator 51, periodic signal superimposing unit (of the second embodiment)

Claims (10)

A DC brushless motor is handled by converting it into an equivalent circuit having a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor and a d-axis armature on the d-axis orthogonal to the q-axis. Current detecting means for detecting a phase current flowing in the armature of the motor, and d flowing in the d-axis armature based on the phase current detected by the current detecting means and the estimated value of the rotor angle of the motor. Dq current calculation means for calculating a shaft current and a q-axis current flowing in the q-axis armature, and a deviation between a d-axis command current and a d-axis current, which is a command value of a current flowing in the d-axis armature, and The d-axis voltage applied to the d-axis armature at every predetermined control cycle and the q-axis so as to reduce the deviation between the q-axis command current and the q-axis current, which is the command value of the current flowing through the q-axis armature. The q-axis voltage applied to the shaft armature is determined, and the motor The motor control device for performing energization control, a method of estimating a rotor angle of the motor,
A first step of superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage;
A sine corresponding to a double angle of the phase difference between the estimated value and the actual value of the rotor angle of the motor, which is calculated based on changes in the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. A second step of calculating a phase difference estimated value according to a phase difference between an actual value and an estimated value of the rotor angle of the motor, using at least one of a reference value and a cosine reference value;
A third step of calculating an angular acceleration estimated value corresponding to the angular acceleration of the motor based on the d-axis command current or the d-axis current;
Based on the phase difference estimated value, the previous estimated value of the rotor angle of the motor, and the estimated value of the previous angular velocity of the motor so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The estimated value of the rotor angle of the motor is calculated while being sequentially updated, and the angular velocity of the motor is calculated based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor. A rotor angle estimation method for a DC brushless motor, comprising: a fourth step of updating the estimated value of the rotor angle of the motor by an observer that calculates the estimated value while sequentially updating the estimated value. A DC brushless motor is handled by converting it into an equivalent circuit having a q-axis armature on the q-axis that is the magnetic flux direction of the field of the motor and a d-axis armature on the d-axis orthogonal to the q-axis. Current detecting means for detecting a phase current flowing in the armature of the motor, rotor angle estimating means for calculating an estimated value of the rotor angle of the motor, phase current detected by the current detecting means, and rotor of the motor Dq current calculation means for calculating a d-axis current flowing in the d-axis armature and a q-axis current flowing in the q-axis armature based on the estimated value of the angle, and a command for the current flowing in the d-axis armature The deviation between the d-axis command current and the d-axis current, which is a value, and the deviation between the q-axis command current and the q-axis current, which is a command value of the current flowing through the q-axis armature, is reduced every predetermined control cycle. D-axis voltage applied to the d-axis armature and It determines the q-axis voltage applied to the q-axis armature, a control device of a DC brushless motor and a current supply control means for performing energization control of the motor,
High-frequency superimposing means for superimposing a high-frequency voltage on the d-axis voltage and the q-axis voltage;
A sine reference value and a cosine according to the double angle of the phase difference between the estimated value and the actual value of the rotor angle of the motor based on the change of the d-axis current and the q-axis current of the motor when the high-frequency voltage is superimposed. A phase difference estimated value calculating means for calculating a phase difference estimated value corresponding to the phase difference between the actual value and the estimated value of the rotor angle of the motor, using at least one of the reference values;
Using an estimated value of the output torque of the motor based on the d-axis command current or the d-axis current, and an angular acceleration estimated value calculating means for calculating an angular acceleration estimated value according to the angular acceleration of the motor,
The rotor angle estimation means is configured to cancel the phase difference between the actual value and the estimated value of the rotor angle of the motor, and the estimated value of the rotor angle of the previous motor and the estimated value of the previous motor. Based on the estimated value of the angular velocity, the estimated value of the rotor angle of the motor is calculated while being sequentially updated, and based on the estimated value of the phase difference, the estimated angular acceleration, and the estimated value of the angular velocity of the previous motor. A controller for a DC brushless motor, wherein the estimated value of the rotor angle of the motor is updated by an observer that calculates the angular velocity of the motor while sequentially updating it. 3. The control apparatus for a DC brushless motor according to claim 2, wherein the rotor angle estimation means updates the estimated value of the rotor angle of the motor and the angular velocity of the motor by an observer of the following formula (1).

Where θ ^ (n + 1) is the updated value of the estimated value of the rotor angle, ω ^ (n + 1) is the updated value of the estimated value of the angular velocity of the motor, Δt is the elapsed time since the previous estimated value was calculated, θ ^ (n): Estimated value of previous rotor angle, ω ^ (n): Estimated value of angular velocity of previous motor, K1, K2: Calculation gain, θ _E1 (n): Estimated value of phase difference, A _ω ( n): Estimated angular acceleration. 4. The control apparatus for a DC brushless motor according to claim 2, wherein the angular acceleration estimation means calculates the estimated angular acceleration value by the following equation (2).

Where A _ω (n): Estimated angular acceleration, T _t (n): Estimated value of motor output torque, I _e : Inertia of motor load, Id_c (n): d-axis command current, K _t : Motor Torque coefficient of 4. The control apparatus for a DC brushless motor according to claim 2, wherein the angular acceleration estimation means calculates the estimated angular acceleration value by the following equation (3).

Where A _ω (n): estimated angular acceleration value, T _t : estimated value of motor output torque, F _t : estimated value of friction torque of motor load, I _e : inertia of motor load, K _t : motor Torque coefficient, Id_c: d-axis command current, K _f : friction coefficient of motor load, ω ^ (n): estimated value of the previous motor angular velocity. Claim 4 or characterized by comprising temperature detecting means for detecting the ambient temperature of the motor, and a torque coefficient correction means for correcting the torque coefficient K _t according to the detected temperature of the temperature detecting means 5. A control device for a DC brushless motor according to 5. DC according to claim 5, characterized by comprising temperature detecting means for detecting the ambient temperature of the motor, and a friction coefficient correcting means for correcting the friction coefficient K _f in accordance with the detected temperature of the temperature detecting means Brushless motor control device.   The DC brushless motor control device according to any one of claims 2 to 7, wherein the motor is connected to an engine and is started by rotating the engine. A drive voltage applied to the motor armature of the motor using an estimated value of the rotor angle of the motor at every predetermined control cycle so as to reduce the deviation between the phase current flowing through the armature of the DC brushless motor and the target current. In the motor control device that determines the energization of the motor and determines the rotor angle of the motor,
A first step of superimposing a high frequency voltage on the drive voltage;
Of the sine reference value and cosine reference value according to the double angle of the actual value of the rotor angle of the motor, calculated based on the change in the phase current of the motor when the high-frequency voltage is superimposed on the drive voltage A second step of calculating a phase difference estimated value corresponding to a phase difference between an actual value and an estimated value of the rotor angle of the motor, using at least one of the following:
A third step of calculating an angular acceleration estimated value corresponding to the angular acceleration of the motor based on the target current;
Based on the phase difference estimated value, the previous estimated value of the rotor angle of the motor, and the estimated value of the previous angular velocity of the motor so as to eliminate the phase difference between the actual value and estimated value of the rotor angle of the motor. The estimated value of the rotor angle of the motor is calculated while being sequentially updated, and the angular velocity of the motor is calculated based on the estimated phase difference value, the estimated angular acceleration value, and the estimated value of the angular velocity of the previous motor. A rotor angle estimation method for a DC brushless motor, comprising: a fourth step of updating the estimated value of the rotor angle of the motor by an observer that calculates the estimated value while sequentially updating the estimated value. Current detection means for detecting the phase current flowing in the armature of the DC brushless motor, rotor angle estimation means for calculating an estimated value of the rotor angle of the motor, and detected by the current detection means at every predetermined control cycle An energization control means for controlling energization of the motor by determining a drive voltage to be applied to the armature of the motor using the estimated value of the rotor angle so as to reduce the deviation between the phase current and the predetermined target current In a DC brushless motor control device comprising:
High frequency superimposing means for superimposing a high frequency voltage on the drive voltage;
Before SL based on the change in the phase current of the motor when the high frequency voltage is superimposed, using at least one of the sine reference value and the cosine reference value corresponding to the double angle of the actual value of the rotor angle of the motor Phase difference estimated value calculating means for calculating a phase difference estimated value according to a phase difference between an actual value and an estimated value of the rotor angle of the motor;
Using an estimated value of the output torque of the motor based on the target current or phase current, and an angular acceleration estimated value calculating means for calculating an angular acceleration estimated value corresponding to the angular acceleration of the motor,
The rotor angle estimation means is configured to cancel the phase difference between the actual value and the estimated value of the rotor angle of the motor, and the estimated value of the rotor angle of the previous motor and the estimated value of the previous motor. Based on the estimated value of the angular velocity, the estimated value of the rotor angle of the motor is calculated while being sequentially updated, and based on the estimated value of the phase difference, the estimated angular acceleration, and the estimated value of the angular velocity of the previous motor. A controller for a DC brushless motor, wherein the estimated value of the rotor angle of the motor is updated by an observer that calculates the angular velocity of the motor while sequentially updating it.

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