JP7075002B2 - Synchronous motor position sensorless controller - Google Patents

Description

The present invention relates to a position sensorless control device for a synchronous motor that drives and controls the synchronous motor without detecting the magnetic pole position of the synchronous motor using a position sensor.

In order to increase the output of a synchronous motor (hereinafter, simply referred to as a motor if necessary), it is necessary to drive the motor in the high frequency region.

Patent Document 1 proposes a control device by voltage vector control suitable for driving a motor in a high frequency region.

Further, from the viewpoint of preventing disconnection and reducing the cost of parts, position sensorless control that drives and controls the motor without detecting the position of the magnetic pole of the motor by the position sensor is desired.

Japanese Unexamined Patent Publication No. 2017-184395

Therefore, it is considered to adopt the position sensorless control for the motor control in the high frequency region using the voltage vector control device of Patent Document 1.

This is because in the voltage vector control, when the zero voltage vector is set, the input voltage of the motor becomes zero, and there is a possibility that only the induced voltage can be detected.

The state at this time will be described in detail as follows.
V = L * di / dt + R * i + e (1)
e = -Ke * ω * sin (ω * t) (2)
L * di / dt >> R * i (3)

Here, V is the input voltage of the motor, e is the induced voltage, and di / dt is the amount of current change per unit time. The equivalent circuit when the zero voltage vector is set is shown in FIG. 8A, and V becomes zero.
At this time, the equations (1) to (3) are as follows.
di / dt = (Ke * ω / L) * sin (ωt) (4)

When the current change amount di / dt is calculated and plotted by the equation (4), the induced voltage becomes a sinusoidal shape as shown in FIG. 8 (b). If the induced voltage is sinusoidal, the magnetic pole position of the motor can be estimated, for example, based on zero cross timing.

However, in the high frequency region of the motor drive frequency, the amount of current change (induced voltage) becomes a sinusoidal shape as shown in FIG. 9 due to the influence of the motor current detection error, the accuracy of the current detection processing resolution, noise, and the like. It doesn't become. Therefore, there remains the problem that the magnetic pole position of the motor cannot be properly estimated based on the induced voltage in the high frequency region simply by using the zero cross timing.

The present invention has been made focusing on such a problem, and an object of the present invention is to appropriately estimate the magnetic pole position of the synchronous motor in the position sensorless control device of the synchronous motor so that it can be applied even in a high frequency region. There is.

The present invention has taken the following measures in order to solve such a problem.

That is, the position sensorless control device of the synchronous motor according to the present invention has a voltage vector generation unit that generates a voltage vector based on a multi-phase current command, and a plurality of switching elements driven according to the voltage vector. A position sensorless control device for a synchronous motor that outputs a voltage to the synchronous motor by driving the switching element according to the voltage vector. The voltage vector generator sets the voltage vector to a zero voltage vector. The detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector at the current control cycle and the normative signal for the estimated magnetic pole position of the synchronous motor at the previous control cycle. Based on this, it is provided with a sensorless control unit that calculates a magnetic pole position error correction value according to the current deviation difference between the detection signal and the reference signal and corrects the estimated magnetic pole position based on the magnetic pole position error correction value. It is a feature.

In the present invention, the detection is based on the detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector in the current control cycle and the reference signal for the estimated magnetic pole position of the synchronous motor in the previous control cycle. The magnetic pole position error correction value according to the current deviation difference between the signal and the reference signal is calculated, and the estimated magnetic pole position is corrected based on the magnetic pole position error correction value. Therefore, the detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector at the current control cycle is not sinusoidal due to the influence of the detection error of the synchronous motor, the accuracy of the resolution of the current detection process, noise, etc. By correcting the estimated magnetic pole position based on the magnetic pole position error correction value according to the current deviation difference from the sinusoidal reference signal for the estimated magnetic pole position of the synchronous motor during the control cycle of In the region, the magnetic pole position of the synchronous motor can be properly estimated based on the detection signal detected at the time of zero voltage vector output to which no voltage is applied.

Moreover, in the position sensorless control device of the synchronous motor according to the present invention, the sensorless control unit is the product of the detection signal of one of the two phases included in the plurality of phases and the reference signal of the other, and the plurality of phases. The current deviation difference is detected based on the difference between the product of the other detection signal of the two phases and the reference signal of one of the two phases included in the above.

In the present invention, the product of one detection signal of two phases contained in the plurality of phases and the other normative signal, and the product of the other detection signal of the two phases included in the plurality of phases and one normative signal are calculated. With the same amplitude and the same phase difference, the difference is only the DC component, and the processing load is smaller than the case where the current deviation difference is detected by focusing on only one phase included in the plurality of phases.

In the position sensorless control device of the synchronous motor according to the present invention, the sensorless control unit includes the product of one of the two phases included in the plurality of phases and the other reference signal, and the plurality of phases. The difference between the product of the other detection signal of the two phases and the reference signal of one of the two phases is detected by passing the low-pass filter to detect the current deviation difference.

In the present invention, the difference between the product of one detection signal of two phases included in the plurality of phases and the other normative signal and the product of the other detection signal of the two phases included in the plurality of phases and one normative signal. Therefore, high frequency noise can be removed. Therefore, the detection accuracy of the current deviation difference is improved.

In the position sensorless control device of the synchronous motor according to the present invention, the detection signal is obtained based on the amount of current change at predetermined time intervals.

In the present invention, a detection signal can be easily obtained based only on the amount of change in current.

The position sensorless control device for the synchronous motor according to the present invention is driven by a PWM control unit that generates a voltage vector based on a current command on the d-axis and the q-axis and a current detection value on the d-axis and the q-axis, and a drive according to the voltage vector. The PWM control unit is a position sensorless control device for a synchronous motor that has a plurality of switching elements and outputs a voltage to the synchronous motor by driving the switching element according to the voltage vector. It includes control to set the voltage vector to zero voltage vector, and includes a detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector at the current control cycle and the synchronous motor at the previous control cycle. Based on the reference signal for the estimated magnetic pole position, the magnetic pole position error correction value according to the current deviation difference between the detection signal and the reference signal is calculated, and the estimated magnetic pole position is determined based on the magnetic pole position error correction value. The sensorless control unit includes a sensorless control unit for correction, and the sensorless control unit includes a product of the detection signal on the d-axis and the reference signal on the q-axis, and the detection signal on the q-axis and the reference signal on the d-axis. It is characterized in that the current deviation difference is detected based on the difference from the product .

In the present invention, the detection is based on the detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector in the current control cycle and the reference signal for the estimated magnetic pole position of the synchronous motor in the previous control cycle. The magnetic pole position error correction value according to the current deviation difference between the signal and the reference signal is calculated, and the estimated magnetic pole position is corrected based on the magnetic pole position error correction value. Therefore, the detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector at the current control cycle is not sinusoidal due to the influence of the detection error of the synchronous motor, the accuracy of the resolution of the current detection process, noise, etc. The PWM modulation method was used by correcting the estimated magnetic pole position based on the magnetic pole position error correction value according to the current deviation difference from the sinusoidal reference signal for the estimated magnetic pole position of the synchronous motor during the control cycle of. Even in the position sensorless control device, the magnetic pole position of the synchronous motor can be estimated appropriately.
Moreover, in the position sensorless control device of the synchronous motor according to the present invention, the sensorless control unit includes the product of the detection signal on the d-axis and the reference signal on the q-axis, the detection signal on the q-axis and the detection signal. The current deviation difference is detected based on the difference between the product and the reference signal of the d-axis.
In the present invention, the product of the detection signal on the d-axis and the reference signal on the q-axis and the product of the detection signal on the q-axis and the reference signal on the d-axis have the same amplitude and the same phase difference. Therefore, the difference between them is only the DC component, and the processing load is smaller than the case where the current deviation difference is detected by focusing only on the d-axis or the q-axis.

As described above, according to the present invention, the detection signal for the actual magnetic pole position of the synchronous motor in the zero voltage vector in the current control cycle is affected by the detection error of the synchronous motor, the accuracy of the resolution of the current detection process, noise, and the like. Although it is not sinusoidal, by correcting the estimated magnetic pole position based on the magnetic pole position error correction value according to the current deviation difference from the sinusoidal reference signal for the estimated magnetic pole position of the synchronous motor in the previous control cycle. In particular, in the medium-high speed and high-frequency regions, the magnetic pole position of the synchronous motor can be properly estimated based on the detection signal detected at the time of zero voltage vector output to which no voltage is applied.

It is a control block diagram which shows the schematic structure of the voltage type inverter device which concerns on embodiment of this invention. It is a figure which showed the control block diagram of FIG. 1 in detail. It is a circuit diagram which shows the schematic structure of a gate circuit. It is a figure which shows the voltage vector generated by the voltage type inverter device which concerns on embodiment. In the current deviation vector diagram, it is a figure which shows the coordinate system when the current deviation of three phases is converted into two axes. It is a figure which shows each region in the current deviation vector diagram. In the current deviation vector diagram, it is a figure which shows an example of the current deviation vector. It is a figure explaining the detection principle of an induced voltage. It is a figure which shows the induced voltage component extracted from a motor current. It is a figure which shows the control of a sensorless control circuit. It is a figure explaining the definition of a voltage vector. It is a figure which shows the control which calculates the current deviation difference ΔIdc based on the detection signal ΔIu, ΔIw and the normative signal Iun, Iwn. It is a figure which shows the change of the waveform in the control of FIG. It is a figure which shows the control which corrects the estimated magnetic pole position θest (n) based on the current deviation difference ΔIdc. It is a control block diagram which shows the schematic structure of the voltage type inverter device which concerns on the modification of this invention.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

(overall structure)
The voltage type inverter device 1 (position sensorless control device) according to the embodiment of the present invention generates a voltage vector based on a three-phase current command, and a plurality of switching elements 31, 32, 41, according to the voltage vector. It is a device that outputs a voltage to the motor 2 by driving 42, 51, 52. The motor 2 is a three-phase AC motor.

As shown in FIG. 1, the voltage type inverter device 1 includes an output voltage vector control circuit 3, an inverter circuit 4, a current detection circuit 5, a sensorless control circuit 6, and a speed detection circuit 7.

Specifically, as shown in FIG. 2, the voltage type inverter device 1 includes a current command generation unit 11, a two-phase three-phase conversion unit 12, a voltage vector generation unit 13, a gate command generation unit 14, and a gate circuit. It includes 15, a three-phase two-phase conversion unit 16, a torque estimation unit 17, a current detection circuit 5, a sensorless control circuit (sensorless control unit) 6, and a speed detection circuit 7.

The torque command Trq ^* is input to the current command generation unit 11 from the outside. The current command generation unit 11 generates current commands Id ^* and Iq ^* on the d-axis and the q-axis based on the input torque command Trq ^* .

The two-phase three-phase conversion unit 12 converts the d-axis and q-axis current commands Id ^* and Iq ^* output from the current command generation unit 11 into three-phase current commands Iu ^* , Iv ^* , and Iw ^* . Specifically, the two-phase three-phase converter 12 uses the current commands Id ^* , Iq ^* on the d-axis and the q-axis and the estimated magnetic pole position θest output from the sensorless control circuit 6, to use the U-phase, V-phase, and U-phase. Generates W-phase current commands Iu ^* , Iv ^* , and Iw ^* .

The voltage vector generation unit 13 calculates a current deviation vector using the three-phase current commands Iu ^* , Iv ^* , Iw ^* and the three-phase output currents Iu, Iv, Iw of the motor 2, and the current in the current deviation vector diagram. The voltage vector is set according to the region to which the deviation vector belongs. The voltage vector generation unit 13 outputs the set voltage vector as a voltage vector command Vector.

The voltage vector generation unit 13 includes a current deviation calculation unit 20, a current deviation vector calculation unit 21, a current deviation vector region determination unit 22, a voltage vector setting unit 23, and a storage unit 24. The detailed configuration of the voltage vector generation unit 13 will be described later.

The gate command generation unit 14 generates gate commands Up, Un, Vp, Vn, Wp, Wn for driving the gate circuit 15 based on the voltage vector command Vector output from the voltage vector generation unit 13. The gate command generation unit 14 issues commands to the switching elements 31, 32, 41, 42, 51, 52 of the switching arms 30, 40, 50 described later in the gate circuit 15 in each of the U phase, V phase, and W phase. Generate.

Specifically, the gate command generation unit 14 has a gate command Up for the switching element 31 of the upper arm in the U-phase switching arm 30, a gate command Un for the switching element 32 of the lower arm, and a V-phase switching arm 40. The gate command Vp for the switching element 41 of the upper arm, the gate command Vn for the switching element 42 of the lower arm, the gate command Wp for the switching element 51 of the upper arm in the W phase switching arm 50, and the switching element 52 of the lower arm. A gate command Wn is generated and output.

The gate commands Up, Un, Vp, Vn, Wp, and Wn generated by the gate command generation unit 14 include signals for turning each switching element into an ON state or an OFF state.

The gate circuit 15 has a plurality of switching elements 31, 32, 41, 42, 51, 52 constituting a three-phase bridge circuit. Specifically, as shown in FIG. 3, the gate circuit 15 has switching arms 30, 40, 50 connected to each of the U phase, V phase, and W phase of the motor 2, respectively. In the switching arm 30, a pair of switching elements 31 and 32 are connected in series. Similarly, in the switching arm 40, a pair of switching elements 41 and 42 are connected in series. Similarly, in the switching arm 50, a pair of switching elements 51 and 52 are connected in series.

One of the switching elements 31, 41, 51 in the switching arms 30, 40, 50 corresponds to the upper arm of the switching arms 30, 40, 50, respectively. The other switching elements 32, 42, 52 in the switching arms 30, 40, 50 correspond to the lower arms of the switching arms 30, 40, 50, respectively.

In this embodiment, for the switching elements 31, 32, 41, 42, 51, 52, for example, an IGBT is used. The switching elements 31, 32, 41, 42, 51, 52 may be other switching devices such as MOSFETs.

Diodes 31a, 32a, 41a, 42a, 51a, and 52a are provided in parallel to the switching elements 31, 32, 41, 42, 51, and 52, respectively. The diodes 31a, 32a, 41a, 42a, 51a, 52a are provided so as to allow a current flow in the direction opposite to the current flowing through the switching elements 31, 32, 41, 42, 51, 52. The diodes 31a, 32a, 41a, 42a, 51a, 52a are so-called freewheeling diodes.

The switching elements 31, 32, 41, 42, 51, 52 of the gate circuit 15 are in the ON state or are in the ON state according to the gate commands Up, Un, Vp, Vn, Wp, Wn output from the gate command generation unit 14, respectively. It is configured to be in the OFF state. FIG. 4 shows a voltage vector diagram of the output voltage of the gate circuit 15. In FIG. 4, V0 and V7 are zero voltage vectors in which no potential difference occurs in each phase. In FIG. 4, in V1, a voltage is generated in the U phase, in V2, a voltage is generated in the V phase, and in V4, a voltage is generated in the W phase.

In the following description, when the ON / OFF states of the switching elements 31, 32, 41, 42, 51, 52 are shown according to each voltage vector, the switching element of the upper arm is in the ON state and the switching of the lower arm is switched. The OFF state of the element is set to "1", the switching element of the upper arm is set to the OFF state, and the switching element of the lower arm is set to "0". Then, the voltage vector is expressed by a three-digit number having the W phase switching state as the third digit, the V phase switching state as the second digit, and the U phase switching state as the first digit.

For example, if the voltage vector is V1, it is expressed as V1 = [001]. In this case, as for the switching element of each arm, the switching element of the upper arm of the W phase is in the OFF state, the switching element of the lower arm of the W phase is in the ON state, and the switching element of the upper arm of the V phase is in the OFF state. The switching element of the lower arm of the V phase is in the ON state, the switching element of the upper arm of the U phase is in the ON state, and the switching element of the lower arm of the W phase is in the OFF state. The zero voltage vector is for V0 and V7, and V0 = [000] (all the switching elements of the lower arm of each phase are ON) and V7 = [111] (the switching element of the upper arm of each phase is ON). All are ON state).

As shown in FIG. 2, the three-phase two-phase conversion unit 16 converts the output currents Iu and Iw of the motor 2 into the d-axis current Id and the q-axis current Iq. Specifically, the three-phase two-phase conversion unit 16 obtains the d-axis current Id and the q-axis current Iq based on the U-phase current Iu and the W-phase current Iw of the motor 2.

The torque estimation unit 17 estimates the output torque of the motor 2 using the d-axis current Id and the q-axis current Iq output from the three-phase two-phase conversion unit 16.

The current detection circuit 5 detects the output currents Iu and Iw of the motor 2.

The output current detected by the current detection circuit 5 may be a combination other than Iu and Iw. Therefore, the current detection circuit 5 may detect the output current of any two of the U phase, the V phase, and the W phase.

The sensorless control circuit 6 estimates the magnetic pole position of the motor 2 based on the output currents Iu and Iw of the motor 2 output from the current detection circuit 5. The control of the sensorless control circuit 6 will be described later.

The speed detection circuit 7 estimates the magnetic pole speed ωest of the motor 2 based on the estimated magnetic pole position θest of the motor 2 output from the sensorless control circuit 6.

(Voltage vector generator)
Next, the configuration of the voltage vector generation unit 13 will be described in detail with reference to FIGS. 1 to 7.

As described above, the voltage vector generation unit 13 includes a current deviation calculation unit 20, a current deviation vector calculation unit 21, a current deviation vector region determination unit 22, a voltage vector setting unit 23, and a storage unit 24. ..

The current deviation calculation unit 20 uses the three-phase current commands Iu ^* , Iv ^* , Iw ^* and the three-phase output currents Iu, Iv, Iw of the motor 2 to calculate the current deviations Δiu, Δiv, Δiw of each phase. calculate.

Specifically, the current deviation calculation unit 20 obtains the three-phase current deviations Δiu, Δiv, and Δiw by the following equation.
Δiu = Iu ^* -Iu
Δiv = Iv ^* -Iv
Δiw = Iw ^* -Iw

The current deviation vector calculation unit 21 obtains a current deviation vector using the current deviations Δiu, Δiv, and Δiw of each phase calculated by the current deviation calculation unit 20. Specifically, the current deviation vector calculation unit 21 obtains the magnitude of the current deviation vector using the current deviations Δiu, Δiv, and Δiw.

Specifically, the current deviation vector calculation unit 21 converts the three-phase current deviations Δiu, Δiv, and Δiw into the two-axis current deviations Δiα, Δiβ as shown in FIG. By obtaining the sum of the squares, Δi ² corresponding to the magnitude of the current deviation vector is calculated.
Δiα = √ (3/2) × Δiu
Δiβ = √ (1/2) × (Δiv-Δiw)
Δi ² = (Δiα ² + Δiβ ² )

The current deviation vector region determination unit 22 uses the three-phase current deviations Δiu, Δiv, and Δiw calculated by the current deviation vector calculation unit 21, and the current deviation vector is any of the current deviation vectors in the current deviation vector diagram as shown in FIG. Determine if it belongs to the area of. Here, as shown in FIG. 6, in the current deviation vector diagram, when the regions are divided by 60 degrees, it is determined which region of the regions A1 to A6 the current deviation vector belongs to.

Specifically, the current deviation vector region determination unit 22 obtains the difference in the absolute value of the current deviation between the two phases of the three-phase current deviations Δiu, Δiv, and Δiw, and determines whether the difference is 0 or more. After narrowing down the region to which the current deviation vector belongs, the region to which the current deviation vector belongs is specified depending on whether the current deviations Δiu, Δiv, and Δiw of the three phases are 0 or more.

For example, in the case of | Δiu | ― | Δiv | ≧ 0, | Δiv | ― | Δiw | ≧ 0 and | Δiw | ― | Δiu | Belongs. If Δiu ≧ 0, the current deviation vector belongs to the region A1 of FIG. 6, and if Δi <0, the current deviation vector belongs to the region A4 of FIG.

The voltage vector setting unit 23 determines the switching mode when the switching elements 31, 32, 41, 42, 51, 52 of the gate circuit 15 operate, and also determines the determined mode and the current deviation vector in the current deviation vector diagram. Set the voltage vector according to the region to which it belongs.

Specifically, as shown in FIG. 7, the voltage vector setting unit 23 determines the switching mode using the magnitude of the current deviation vector X. In the present embodiment, the voltage vector setting unit 23 determines whether or not the magnitude of the current deviation vector is equal to or less than the first predetermined value P and whether or not the second predetermined value Q or less is larger than the first predetermined value P. do. In the example shown in FIG. 7, the size of the current deviation vector X (thick arrow in FIG. 7) is equal to or larger than the first predetermined value P and smaller than the second predetermined value Q.

The first predetermined value P is set to the magnitude of the current deviation vector that does not affect the output of the motor 2 so much. Further, the second predetermined value Q gives the switching loss and the current distortion factor more than the influence on the output of the motor 2 when the influence on the output of the motor 2 and the influence on the switching loss and the current distortion factor are taken into consideration. The magnitude of the current deviation vector is set so that the influence is larger.

When the magnitude of the current deviation vector is equal to or less than the first predetermined value P, the voltage vector setting unit 23 sets the switching mode to the reflux mode. Specifically, when the magnitude of the current deviation vector is equal to or less than the first predetermined value P, the voltage vector setting unit 23 sets the output voltage vector to either zero voltage vector V0 or V7. Therefore, when the V0 and V7 vectors are selected and output to the gate, the motor current returns.
(Sensorless control circuit 6)

In the position sensorless control of the voltage type inverter 1 of the present embodiment, a method of estimating the magnetic pole position of the motor 2 will be described using the flow shown in FIG.

In the present embodiment, as shown in FIG. 11A, the U phase (corresponding to the U phase in FIG. 4) is used as the origin for estimating the magnetic pole position of the motor 2. With the U-phase vector as a reference, the angle formed by the U-phase vector and the N pole direction of the permanent magnet of the motor 2 is defined as the actual angle θ (actual angle of the motor 2).

Further, as shown in FIG. 11B, the magnetic pole position is estimated so that the estimated magnetic pole position θest becomes the actual angle θ based on the magnetic pole position error θh described later.

When the flow shown in FIG. 10 starts (starts), first, it is determined whether or not the voltage vector set by the voltage vector setting unit 23 is a zero voltage vector (V0, V7). (Step 1)

When the zero voltage vector (V0, V7) is set, the sensorless control circuit 6 detects the motor U phase detection signal ΔIu and the W phase detection signal ΔIw. The detection signal ΔIu is the amount of current change di / dt when the U-phase zero voltage vector is set.

In the present embodiment, by fixing the differential time dt, the detection signal ΔIu is treated as the amount of current change during a preset period. The detection signal ΔIw is also detected in the same manner as the detection signal ΔIu. (Step 2)

After detecting the detection signals ΔIu and ΔIw, the sensorless control circuit 6 calculates the normative signal Iun of the motor U phase and the normative signal Iwn of the W phase. As shown in FIG. 12, the normative signal Iun is calculated based on the estimated magnetic pole position θest (estimated electric angle) of the motor 2 with reference to a trigonometric function table or the like. In this embodiment, a Sin function value is used as a normative signal.

In addition, the normative signal Iun is 2π / 3 with respect to the estimated magnetic pole position θest because the W-phase motor current is 2π / 3 ahead of the U-phase motor current (there is a phase difference of 2π / 3). Calculated in consideration of the phase difference of / 3. (Step 3)

After calculating the normative signals Iun and Iwn, the sensorless control circuit 6 calculates the current deviation difference ΔIdc from the detection signals ΔIu and ΔIw and the normative signals Iun and Iwn.

As shown in FIG. 13, the current deviation difference ΔIdc is calculated by subtracting ΔIu * Iwn (the product of the detection signal ΔIu and the reference signal Iwn) and ΔIw * Iun (the product of the detection signal Iun and the detection signal ΔIw). ..

The phase of the detection signal ΔIu is the actual phase of the induced voltage of the U phase, and the phase of the normative signal Iun is the phase based on the estimated magnetic pole position θest. Similarly, the phase of the detection signal ΔIw is the actual phase of the induced voltage of the W phase, and the phase of the normative signal Iwn is the phase based on the estimated magnetic pole position θest.

Therefore, in the current deviation difference ΔIdc, if the amplitude and phase of ΔIu * Iwn and the amplitude and phase of ΔIw * Iun match, the amplitude difference and phase difference between ΔIu * Iwn and ΔIw * Iun become zero.

On the other hand, if the amplitude and phase of ΔIu * Iwn and the amplitude and phase of ΔIw * Iun do not match, the values correspond to the phase difference between ΔIu * Iwn and ΔIw * Iun (step 4).

Next, the method of calculating the current deviation difference ΔIdc will be specifically described.

The U-phase norm signal Iun and the W-phase norm signal Iwn are as follows.

Here, as described above, the normative signal Iwn needs to consider the phase difference of 2π / 3 with respect to the estimated magnetic pole position θest.
Iun (t) = sin (ω * t) (5)
Iwn (t) = sin (ω * t-2π / 3) (6)

The U-phase detection signal ΔIu and the U-phase detection signal ΔIw are as follows. Here, Ke is the counter electromotive force constant, L is the inductance, ω is the motor speed, and φ is the phase difference between the actual phase of the motor and the estimated magnetic pole.
ΔIu (t) = Ke / L * ω * sin (ω * t + φ) (7)
ΔIw (t) = Ke / L * ω * sin (ω * t－2π / 3 ＋ φ) (8)

ΔIu * Iwn and ΔIw * Iun are as follows from the equations (5) to (8).
ΔIw (t) * Iun (t) = Ke / L * ω * sin (ω * t－2π / 3 ＋ φ) * sin (ω * t)
= -Ke / 2 L * ω * (cos (2ω * t + φ-2π / 3) -cos (φ-2π / 3)) (9)
ΔIu (t) * Iwn (t) = Ke / L * ω * sin (ω * t ＋ φ) * sin (ω * t－2π / 3)
= -Ke / 2 L * ω * (cos (2ω * t + φ-2π / 3) -cos (φ + 2π / 3)) (10)
Therefore, the current deviation difference ΔIdc is as follows from the equations (9) and (10).

ΔIdc (t) = ΔIw (t) * Iun (t) －ΔIu (t) * Iwn (t)
= Ke / 2 L * ω * (cos (φ + 2π / 3) -cos (φ-2π / 3))
= -Ke / L * sin2π / 3 * ω * sinφ (11)

Here, if sinφ = φ is approximated, it becomes as follows.
ΔIdc (t) ＝ －Ke / L * sin2π / 3 * ω * φ (12)

When the current deviation difference ΔIdc calculated by the equation (12) is removed by an LPF (low-pass filter) and the motor speed ω = the estimated motor speed ω_est, the result is as follows.
ΔIdc (t) = －√3 / 2 * Ke / L * ω_est (t) * φ (13)

In particular, when the inertia of the motor is large, the fluctuation of the motor speed ω is small, so ΔIdc (t) can be regarded as a proportional function of φ.

After calculating the current deviation difference ΔIdc in step 4 of FIG. 10, the sensorless control circuit 6 calculates the magnetic pole position error correction value θh. As shown in FIG. 14, a feedback circuit is configured in which the current deviation difference ΔIdc is brought closer to the target value Io ^* . Specifically, in order to bring the current deviation difference ΔIdc close to zero, control is performed with the target value Io ^* as zero, and the magnetic pole position error correction value θh, which is the operation amount used for estimating the magnetic pole position, is calculated. The control applied here refers to PLL control, P control, PI control, and the like. (Step 5)

After calculating the magnetic pole position error correction value θh, the sensorless control circuit 6 corrects the previous estimated magnetic pole position θest [n] based on the magnetic pole position error correction value θh. Specifically, as shown in FIG. 14, the previous estimated magnetic pole position θest [n] is subtracted by the magnetic pole position error correction value θh to obtain the current estimated magnetic pole position θest [n + 1], and the estimated magnetic pole position is set. Update. (Step 6)

After calculating the current estimated magnetic pole position θest [n + 1], the sensorless control circuit 6 inputs the current estimated magnetic pole position θest [n + 1] to the two-phase three-phase conversion unit 12 as the estimated magnetic pole position θest. .. Further, the process returns to step 1 and the estimation of the magnetic pole position is continued (step 7).

On the other hand, when a voltage vector (V1 to V6) other than the zero voltage vector is set in step 1 of FIG. 10, the magnetic pole position of the motor 2 cannot be estimated based on the current change amount di / dt. Therefore, the magnetic pole position of the motor 2 is estimated by using the latest estimated magnetic pole position θest1 estimated at the time of setting the zero voltage vector and the latest motor estimated speed ω_est.

First, the sensorless control circuit 6 calculates the advance angle Δθest of the magnetic pole position of the motor 2 in the control cycle of the voltage type inverter 1. (Step 8)

After calculating the lead angle Δθest of the magnetic pole position, the sensorless control circuit 6 adds the advance angle Δθest of the magnetic pole position to the latest estimated magnetic pole position θest1 estimated at the time of setting the zero voltage vector, and the current magnetic pole position of the motor 2. Calculate θest [n + 1]. (Step 9)

After calculating the current estimated magnetic pole position θest [n + 1], the sensorless control circuit 6 inputs the current estimated magnetic pole position θest [n + 1] to the two-phase three-phase conversion unit 12 as the estimated magnetic pole position θest. .. Further, the process returns to step 1 and the estimation of the magnetic pole position is continued (step 10).

Next, the specific calculation method of the current deviation difference ΔIdc is as follows.
Δθest = ω_est * Ts (14)
θest [n + 1] = θest [n] + Δθest (15)

Here, Δθest is the advance angle of the magnetic pole position of the motor 2, and Ts is the control cycle of the voltage type inverter 1.

The voltage-type inverter device 1 of the present embodiment includes a voltage vector generation unit 13 that generates a voltage vector based on a three-phase current command, and a plurality of switching elements 31, 32, 41, 42 driven according to the voltage vector. , 51, 52, and is a voltage type inverter device 1 of the motor 2 that outputs a voltage to the motor 2 by driving a switching element according to the voltage vector, and the voltage vector generation unit 13 is a voltage vector. Is controlled to set to the zero voltage vector, and the detection signals ΔIu and ΔIw for the actual magnetic pole position of the motor 2 in the zero voltage vector at the current control cycle and the motor 2 at the previous control cycle Based on the normative signals Iun and Iwn for the estimated magnetic pole position, the magnetic pole position error correction value θh according to the current deviation difference ΔIdc between the detected signal and the normative signal is calculated, and the magnetic pole is based on the magnetic pole position error correction value θh. A sensorless control unit 6 for correcting an estimated position is provided.

Therefore, in the voltage type inverter device 1 of the present embodiment, the detection signal for the actual magnetic pole position of the motor 2 in the zero voltage vector at the current control cycle is the detection error of the motor 2 and the accuracy of the resolution of the current detection process. Although it is not sinusoidal due to the influence of noise, etc., the estimated magnetic pole position of the estimated magnetic pole position of the motor 2 in the previous control cycle is based on the magnetic pole position error correction value θh according to the current deviation difference ΔIdc from the sinusoidal reference signal. By correcting the position, the magnetic pole position of the motor 2 can be appropriately estimated based on the detection signal detected at the time of zero voltage vector output to which no voltage is applied, especially in the medium and high speed and high frequency regions.

In the voltage type inverter device 1 of the present embodiment, the sensorless control unit 6 has the product of the U-phase detection signal ΔIu and the W-phase reference signal Iwn contained in the three phases, and the W-phase detection signal ΔIw and the U-phase. The current deviation difference ΔIdc is detected based on the difference from the product of the normative signal Iun.

As a result, in the voltage type inverter device 1, the product of the U-phase detection signal ΔIu contained in the three phases and the W-phase reference signal Iwn, and the product of the W-phase detection signal ΔIw and the U-phase reference signal Iun. Is the same amplitude and the same phase difference, and the difference is only the DC component, and the processing load is smaller than the case where the current deviation difference ΔIdc is detected by focusing on only one phase included in the three phases. ..

In the voltage type inverter device 1 of the present embodiment, the sensorless control unit 6 has the product of the U-phase detection signal ΔIu and the W-phase reference signal Iwn included in the three phases, and the W-phase detection signal ΔIw and the U-phase. The current deviation difference ΔIdc is detected through a low-pass filter for the difference between the product and the product of the normative signal Iun.

As a result, in the voltage type inverter device 1, the product of the U-phase detection signal ΔIu contained in the three phases and the W-phase reference signal Iwn, and the product of the W-phase detection signal ΔIw and the U-phase reference signal Iun. From the difference with, the high frequency noise can be removed. Therefore, the detection accuracy of the current deviation difference ΔIdc is improved.

In the voltage type inverter device 1 of the present embodiment, the detection signals ΔIu and ΔIw are obtained based on the amount of current change at predetermined time intervals.

As a result, in the voltage type inverter device 1, a detection signal can be easily obtained based only on the amount of change in current.

As described above, in the present invention, even the voltage type inverter device 1 having no position sensor can drive and control the synchronous motor with high accuracy.

In the present invention, the position sensorless control is applied to the voltage type inverter, but it is not necessary to adopt the PWM modulation method of triangular wave comparison based on the dq axis vector control, which is a general current control method.

In the extended induced voltage method, which is a position sensorless control in the middle and high speed range, which is common in the dq-axis vector control system, the magnetic poles are estimated by an observer using a motor model. However, in the present invention, the magnetic pole position can be estimated with a small amount of calculation without being affected by model parameter fluctuations.

Although one embodiment of the present invention has been described above, the specific configuration of each part is not limited to the above-described embodiment, and various modifications can be made without departing from the spirit of the present invention.

Although the case where the voltage type inverter device 1 of the above embodiment is not a dq-axis vector control system (PWM control) has been described, the present invention is applicable to the dq-axis vector control system (PWM control).

That is, as shown in FIG. 15, the inverter control device (position sensorless control device) 101 of the present invention has a plurality of phases by the PWM modulation method based on the d-axis and q-axis current commands and the d-axis and q-axis current detection values. A PWM control unit 103 that generates a voltage vector for the above, an inverter circuit 115 having a plurality of switching elements driven according to the voltage vector generated by the PWM control unit 103, a current detection circuit 5, and a three-phase two-phase conversion unit. A position sensorless control device for a motor 2 having a sensorless control circuit 106, a sensorless control circuit 106, and a speed detection circuit 7, and outputting a voltage to the motor 2 by driving a switching element through a PWM control unit 103. The control unit 103 has a first state (corresponding to zero voltage vector V0) in which all the switching elements of the lower arm of each phase are in the ON state, and a second state in which all the switching elements of the upper arm of each phase are in the ON state. The sensorless control unit 106 includes a control (corresponding to the zero voltage vector V7), and the sensorless control unit 106 includes a detection signal for the actual magnetic pole position of the motor 2 in the first state or the second state in the current control cycle. Based on the reference signal for the estimated magnetic pole position of the motor 2 in the previous control cycle, the magnetic pole position error correction value according to the current deviation difference between the detection signal and the reference signal is calculated and used as the magnetic pole position error correction value. Correct the estimated magnetic pole position based on.

The sensorless control in the sensorless control unit 106 is the same as the sensorless control in the sensorless control unit 6 of the above embodiment.

As a result, as in the above embodiment, the detection signal about the actual magnetic pole position of the motor 2 in the first state or the second state in the current control cycle and the estimated magnetic pole of the motor 2 in the previous control cycle. Based on the reference signal for the position, the magnetic pole position error correction value according to the current deviation difference between the detection signal and the reference signal is calculated, and the estimated magnetic pole position is corrected based on the magnetic pole position error correction value. Therefore, the detection signal for the actual magnetic pole position of the motor 2 in the first state or the second state in the current control cycle is sinusoidal due to the influence of the detection error of the motor 2, the accuracy of the resolution of the current detection process, noise, and the like. However, PWM modulation is performed by correcting the estimated magnetic pole position based on the magnetic pole position error correction value according to the current deviation difference from the sinusoidal reference signal for the estimated magnetic pole position of the motor 2 in the previous control cycle. Even in the position sensorless control device using the method, the magnetic pole position of the motor 2 can be estimated appropriately.

In the above embodiment, the voltage type inverter devices 1 and 101 having no position sensor have been described, but the present invention can be applied to the voltage type inverter device having a position sensor. By applying the present invention, it can be applied to abnormality detection (failure, disconnection, connector disconnection, etc.) by duplication of the magnetic pole position detection system and correction of the sensor detection value.

1,101 Voltage type inverter device 2 Motor 6, 106 Sensorless control circuit 13 Voltage vector generator 31, 32, 41, 42, 51, 52 Switching element 103 PWM control unit

Claims (4)

By having a voltage vector generation unit that generates a voltage vector based on a multi-phase current command and a plurality of switching elements driven according to the voltage vector, and driving the switching element according to the voltage vector. , A position sensorless control device for a synchronous motor that outputs a voltage to the synchronous motor.
The voltage vector generation unit includes a control for setting the voltage vector to a zero voltage vector.
The detection signal and the above-mentioned detection signal based on the detection signal about the actual magnetic pole position of the synchronous motor in the zero voltage vector in the current control cycle and the reference signal about the estimated magnetic pole position of the synchronous motor in the previous control cycle. A sensorless control unit that calculates a magnetic pole position error correction value according to the current deviation difference from the reference signal and corrects the estimated magnetic pole position based on the magnetic pole position error correction value is provided .
The sensorless control unit includes a product of one of the two phases of the detection signal included in the plurality of phases and the other of the normative signals, and one of the two phases of the detection signal contained in the plurality of phases. A position sensorless control device for a synchronous motor, characterized in that the current deviation difference is detected based on the difference between the product and the reference signal . The sensorless control unit includes a product of one of the two phases of the detection signal included in the plurality of phases and the other of the reference signals, and one of the two phases of the detection signal included in the plurality of phases. The position sensorless control device for a synchronous motor according to claim 1 , wherein the difference between the product and the reference signal is detected by a low-pass filter to detect the current deviation difference. The position sensorless control device for a synchronous motor according to claim 1 or 2 , wherein the detection signal is obtained based on a current change amount at predetermined time intervals. It has a PWM control unit that generates a voltage vector based on a d-axis and q-axis current command and a d-axis and q-axis current detection value, and a plurality of switching elements driven according to the voltage vector, and has the voltage. A position sensorless control device for a synchronous motor that outputs a voltage to the synchronous motor by driving the switching element according to a vector.
The PWM control unit includes a control for setting the voltage vector to a zero voltage vector.
The detection signal and the above-mentioned detection signal based on the detection signal about the actual magnetic pole position of the synchronous motor in the zero voltage vector in the current control cycle and the reference signal about the estimated magnetic pole position of the synchronous motor in the previous control cycle. A sensorless control unit that calculates a magnetic pole position error correction value according to the current deviation difference from the reference signal and corrects the estimated magnetic pole position based on the magnetic pole position error correction value is provided .
The sensorless control unit is based on the difference between the product of the detection signal on the d-axis and the reference signal on the q-axis and the product of the detection signal on the q-axis and the reference signal on the d-axis. A position sensorless control device for a synchronous motor, characterized in that the current deviation difference is detected .

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