JP6129972B2 - AC motor control device, AC motor drive system, fluid pressure control system, positioning system - Google Patents

Description

  The present invention relates to an AC motor control device, an AC motor drive system, a fluid pressure control system, and a positioning system using the same.

  Three-phase AC motors are widely used in various fields such as industry, home appliances, and automobiles. In particular, the application of small and highly efficient permanent magnet synchronous motors (hereinafter abbreviated as PMSM) is increasing.

  Conventionally, in order to control the rotational speed and torque of an AC motor with high accuracy, a rotational speed sensor (or rotational position sensor in the case of PMSM) that detects the rotational state of the motor, and a phase current that flows through each phase of the motor And a phase current sensor for detecting. However, when these sensors are used, problems such as an increase in failure rate, variation in detection results, installation accuracy, and installation space occur. Therefore, it is desirable to make the sensorless so that the rotation state and phase current of the motor can be detected without using a sensor.

  A method for realizing the sensorless phase current is described in, for example, Patent Documents 1 and 2 below.

  In the phase current detection method of Patent Document 1, the current flowing in the DC side of the inverter that controls the driving of the electric motor is detected, and the current change before and after switching is obtained based on the detection result. Seeking.

  In the phase current detection method of Patent Document 2, two continuous peaks or valleys of a triangular wave carrier signal are obtained in two-phase modulation type pulse width modulation (PWM) in which two-phase switching elements are switched among three phases within one cycle. In this part, basic voltage vector components whose phases are different from each other by 180 degrees are alternately output. Then, the current flowing through the shunt resistor provided on the DC bus of the inverter is detected at this timing, converted from analog to digital, and taken into the microcomputer. Thereby, the timing of current detection is fixed, and an inexpensive microcomputer can be used.

Japanese Laid-Open Patent Publication No. 8-19263 Japanese Unexamined Patent Publication No. 2005-12934

  During driving of the electric motor, a current corresponding to a phase current of any one of the U, V, and W phases flows through the shunt resistor according to the switch state of each phase. Therefore, by detecting the current of the shunt resistor at the timing synchronized with the switching of the switch state of each phase, it is possible to realize the sensorless phase current as described above.

  However, in the case of general PWM, switching of the switch state occurs every half cycle of the carrier signal, and the generation timing is not constant. Therefore, in the phase current detection method as in Patent Document 1, it is necessary to perform current detection and analog-digital conversion of the detected value twice in a half cycle of the carrier signal. It is also necessary to change the current detection timing in accordance with the switching of the switch state. Since a microcomputer capable of realizing such a function is expensive, there is a problem that the cost of the apparatus is increased. Further, since the detected current includes an error such as a current ripple generated when the switch state is switched, there is a problem that the accuracy of torque control is deteriorated.

  On the other hand, in the phase current detection method as disclosed in Patent Document 2, since the current flowing through the shunt resistor can be detected at the peak or valley timing of the triangular wave carrier signal, the above-described problem relating to cost increase does not occur. However, in the previous PWM period (first period), the voltage command vector is corrected and output so as to be larger than the original voltage command vector, and in the subsequent PWM period (second period), the first By outputting a voltage command vector that is 180 degrees out of phase with the voltage command vector in the period, the voltage command vector is matched with the original voltage command vector. Therefore, the current ripple in the first period increases, and as a result, the current detection value includes a large detection error. Therefore, the problem that the accuracy of torque control deteriorates cannot be overcome.

  An object of the present invention is to provide an AC motor control device capable of detecting a phase current with high accuracy using an inexpensive microcomputer in view of the problems in the conventional phase current detection method as described above.

The control apparatus for an AC motor according to one aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and in accordance with the switching state of a switching element provided with three phases on each of the positive electrode side and the negative electrode side in the inverter. It controls the drive of the AC motor. The inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements. The voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all the switching elements on the positive electrode side are OFF and all the switching elements on the negative electrode side are ON are included. The AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After outputting the zero vector, the inverter is controlled to output a voltage vector obtained by inverting the first voltage vector as a third voltage vector , and the first voltage vector, the second voltage vector, the zero vector, and the third vector are output. The inverter is controlled so as to sequentially output the voltage vectors of the first and the third voltage vectors as the first voltage vector and the third voltage vector. By controlling the inverter so as to output different types of voltage vectors to the controls the driving of the AC motor.
The control apparatus for an AC motor according to another aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and the inverter is switched to a switching state in which three phases are provided on each of the positive electrode side and the negative electrode side. In response to the drive of the AC motor, a voltage command generation unit that generates a three-phase voltage command, a voltage correction unit that corrects the voltage command, and a corrected voltage command corrected by the voltage correction unit And a PWM generator that generates a three-phase PWM waveform based on the comparison result, and outputs the PWM waveform generated by the PWM generator to the inverter. The driving of the AC motor is controlled by controlling the switching state of the switching element . The inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements. The voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all the switching elements on the positive electrode side are OFF and all the switching elements on the negative electrode side are ON are included. The AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After outputting the zero vector, the inverter is controlled so that a voltage vector obtained by inverting the first voltage vector is output as a third voltage vector. In the first period of two consecutive triangular wave carriers, the voltage correction unit sets the intermediate voltage command to 0 in the three-phase voltage command, and further outputs the maximum voltage command and the minimum voltage command in the three-phase voltage command. The voltage command is corrected by shifting, and in the next cycle following the first cycle, the minimum voltage command is set to 0, and the voltage command is corrected by shifting the maximum voltage command and the intermediate voltage command. .
The control apparatus for an AC motor according to another aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and the inverter is switched to a switching state in which three phases are provided on each of the positive electrode side and the negative electrode side. In response to the drive of the AC motor, a voltage command generation unit that generates a three-phase voltage command, a voltage correction unit that corrects the voltage command, and a corrected voltage command corrected by the voltage correction unit And a PWM generator that generates a three-phase PWM waveform based on the comparison result, and outputs the PWM waveform generated by the PWM generator to the inverter. The driving of the AC motor is controlled by controlling the switching state of the switching element. The inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements. The voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all the switching elements on the positive electrode side are OFF and all the switching elements on the negative electrode side are ON are included. The AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After the zero vector is output, the inverter is controlled so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector, and when the triangular wave carrier takes an intermediate value between the minimum value and the maximum value, Acquires the detection result of the direct current flowing in the direct current bus .
A control apparatus for an AC motor according to still another aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and the switching state of switching elements provided on the positive electrode side and the negative electrode side in the inverter in three phases respectively. The drive of the AC motor is controlled according to the above. The inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements. The voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all the switching elements on the positive electrode side are OFF and all the switching elements on the negative electrode side are ON are included. The AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After outputting the zero vector, the inverter is controlled to output a voltage vector obtained by inverting the first voltage vector as the third voltage vector, and in the low speed region where the AC motor is driven at low speed, The inverter is controlled so that only two-phase switching elements are switched and the 120-degree conduction drive is performed. In the speed range, to perform a three-phase conduction drive by switching the switching state of the switching elements of three phases, it controls the inverter. The AC motor control device also includes a first current controller that outputs a first voltage command used in three-phase energization driving, and a second current commonly used in 120-degree energization driving and three-phase energization driving. A first current controller that outputs a voltage command, and the first current controller performs a feedforward calculation based on a current command input from the outside or a detection result of a current flowing in the AC motor, by a first method. Outputs voltage command .
AC motor drive system according to the present onset Ming, a control device ac motor, is connected to the controller for an AC motor, an inverter switching element is provided in the positive and negative sides by respective three-phase AC motor The control device includes a three-phase AC motor that is driven and controlled in accordance with the switching state of the switching element, and the AC motor control device, the inverter, and the AC motor are integrated.
Fluid pressure control system according to the present onset Ming, a control device ac motor, is connected to the controller for an AC motor, an inverter switching element is provided in the positive and negative sides by respective three-phase AC motor The control device includes a three-phase AC motor that is driven and controlled according to the switching state of the switching element, and a pump that operates according to the driving of the AC motor and controls the fluid pressure.
Positioning system according to the present onset Ming, a control device ac motor, is connected to the controller for an AC motor, an inverter for the positive and negative sides in each respective three-phase switching elements are provided, the AC motor control device Thus, a three-phase AC motor that is driven and controlled according to the switching state of the switching element, and a positioning device that operates according to the driving of the AC motor and adjusts the position of the object.

  ADVANTAGE OF THE INVENTION According to this invention, the control apparatus of the alternating current motor which can detect a phase current with high precision using an inexpensive microcomputer is realizable.

It is a figure which shows the structure of the alternating current motor drive system containing the control apparatus of the alternating current motor which concerns on the 1st Embodiment of this invention. It is a figure which shows the example of the PWM waveform and switch state when not correcting a voltage command. It is a figure which shows the mode of the electric current which each flows into a switching element in each switch state. It is a figure which shows the relationship between the motor current of each phase which flows into an alternating current motor, the shunt current which flows into a direct current bus, and a switch state. It is a figure which shows the example of a PWM waveform at the time of correcting a voltage command based on the conventional two-phase modulation system, and a switch state. FIG. 6 is a diagram illustrating an example of a PWM waveform and a switch state when a voltage command is corrected by a method different from that in FIG. 5. It is a figure which shows the example of the PWM waveform at the time of correcting the voltage command by this invention, and a switch state. 6 is a table showing a relationship between a voltage vector in a period A and a detected phase current. 6 is a table showing a relationship between a voltage vector in a period B and a detected phase current. It is a figure which shows the mode of the phase current detected as a shunt electric current in the 1st voltage vector and the 3rd voltage vector. It is the figure which represented each voltage vector output from an inverter on the (alpha) (beta) coordinate. It is a figure which shows the detail of correction | amendment of the voltage command by a voltage correction part. It is a figure which shows the structure of the control apparatus which concerns on the 2nd Embodiment of this invention. It is a figure which shows the structure of the alternating current motor drive system which concerns on the 3rd Embodiment of this invention. It is a figure which shows the structure of the hydraulic control system which concerns on the 4th Embodiment of this invention. It is a figure which shows the modification of a structure of the hydraulic control system which concerns on the 4th Embodiment of this invention. It is a figure which shows the structure of the positioning system which concerns on the 5th Embodiment of this invention.

(First embodiment)
FIG. 1 is a diagram showing a configuration of an AC motor drive system including an AC motor control device 2 according to a first embodiment of the present invention. This AC motor drive system includes a q-axis current command generator 1, a control device 2, an inverter 3, and an AC motor 4. The AC motor 4 is a three-phase AC motor, for example, a permanent magnet type synchronous motor (PMSM) is used.

  The q-axis current command generator 1 is a circuit that generates a q-axis current command Iq * corresponding to the output torque of the AC motor 4. The q-axis current command generator 1 is positioned above the control device 2 and normally has a mechanism for generating the q-axis current command Iq * so that the rotational speed ωr of the AC motor 4 becomes a predetermined speed. ing. The q-axis current command Iq * output from the q-axis current command generator 1 is input to the subtracter 6b in the control device 2.

  The control device 2 controls the driving of the AC motor 4 according to the switching state of each switching element provided in the inverter 3, and the q-axis current command Iq * input from the q-axis current command generator 1 is controlled. The AC motor 4 operates so as to generate a corresponding torque. The control device 2 includes a voltage command generation unit 20, a voltage correction unit 10, a PWM generation unit 11, a current estimation unit 21, a position / speed estimation unit 15, and a speed conversion unit 16.

  The voltage command generator 20 is a portion that generates three-phase, that is, U-phase, V-phase, and W-phase voltage commands based on the q-axis current command Iq * input from the q-axis current command generator 1. The voltage command generator 20 includes a d-axis current command generator 5, a subtractor 6 a, a subtractor 6 b, a d-axis current controller 7, a q-axis current controller 8, and a dq inverse converter 9.

  The d-axis current command generator 5 generates a d-axis current command Id * corresponding to the excitation current of the AC motor 4. If the AC motor 4 is a non-salient permanent magnet motor, the d-axis current command Id * is normally zero. The d-axis current command Id * output from the d-axis current command generator 5 is input to the subtractor 6a.

  The subtractor 6 a is a subtracter for obtaining a d-axis current deviation between the d-axis current command Id * input from the d-axis current command generator 5 and the d-axis current Id estimated by the current estimation unit 21. . This d-axis current deviation is input to the d-axis current controller 7.

  The subtractor 6b is a subtracter for obtaining a q-axis current deviation between the q-axis current command Iq * input from the q-axis current command generator 1 and the q-axis current Iq estimated by the current estimation unit 21. . This q-axis current deviation is input to the q-axis current controller 8.

  The d-axis current controller 7 calculates the d-axis voltage command Vd * on the dq coordinate axis so that the d-axis current deviation input from the subtractor 6a becomes zero. The d-axis voltage command Vd * calculated by the d-axis current controller 7 is input to the dq inverse converter 9.

  The q-axis current controller 8 calculates the q-axis voltage command Vq * on the dq coordinate axis so that the q-axis current deviation input from the subtractor 6b becomes zero. The q-axis voltage command Vq * calculated by the q-axis current controller 8 is input to the dq inverse converter 9.

  The dq reverse converter 9 receives the d-axis voltage command Vd * and the q-axis voltage Vq * of the dq coordinate axis (magnetic flux axis-magnetic flux axis orthogonal axis) system input from the d-axis current controller 7 and the q-axis current controller 8. This is a circuit that converts the voltage command into a three-phase AC coordinate. The dq inverse converter 9 converts the d-axis voltage command Vd * and the q-axis voltage command Vq * into the voltage command Vu * of the three-phase AC coordinate system based on the phase angle θdc input from the position / velocity estimation unit 15. Convert to Vv * and Vw *. Each voltage command after conversion, that is, the U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * is output to the voltage correction unit 10.

  The voltage command generator 20 can generate three-phase voltage commands Vu *, Vv * and Vw * by the operation of each component as described above.

  The voltage correction unit 10 corrects the three-phase voltage commands Vu *, Vv *, and Vw * output from the voltage command generation unit 20, respectively, and corrects the corrected voltage commands Vu **, Vv **, and Output to the PWM generator 11 as Vw **. The voltage command correction method performed by the voltage correction unit 10 will be described in detail later.

  The PWM generation unit 11 compares the corrected three-phase voltage commands Vu **, Vv **, and Vw ** output from the voltage correction unit 10 with a triangular wave carrier of a predetermined period, and based on the comparison result, Generate phase PWM waveform. From this PWM generator 11, as three-phase PWM waveforms, PWM signals UP, VP, WP corresponding to the U-phase, V-phase, and W-phase, and PWM signals UN, VN, WN is generated. These PWM signals generated by the PWM generator 11 are output from the control device 2 to the inverter 3.

  The current estimation unit 21 acquires a detection result of the DC current flowing in the DC bus 34 in the inverter 3 based on the DC current signal IDC input from the inverter 3, and based on the detection result, the phase current of the AC motor 4 and This is a part for obtaining the current on the dq coordinate. The current estimation unit 21 includes a sample and hold circuit 12, a current reproducer 13, and a dq converter 14.

  The sample hold circuit 12 acquires the detection result of the direct current flowing through the direct current bus 34 of the inverter 3 by sampling the direct current signal IDC input from the inverter 3 at a predetermined sampling timing. The sampling timing by the sample and hold circuit 12 is determined based on the triangular wave carrier used in the PWM generator 11. This point will be described in detail later.

  The current reproducer 13 reproduces the phase currents Iu, Iv, and Iw flowing in the U phase, V phase, and W phase of the AC motor 4 based on the detection result of the DC current acquired by the sample hold circuit 12, respectively. The reproduced values Iuc, Ivc, and Iwc of the phase current are obtained. The phase current reproduction values Iuc, Ivc, and Iwc obtained by the current reproducer 13 are output to the dq converter 14.

  The dq converter 14 is a circuit that receives the phase current reproduction values Iuc, Ivc, and Iwc from the current reproduction unit 13 and converts them into current values Id and Iq on the dq coordinates. The dq converter 14 converts the reproduction values Iuc, Ivc, Iwc of the three-phase phase currents into the d-axis current Id and the q-axis current Iq based on the phase angle θdc input from the position / velocity estimation unit 15. To do. The converted values of the d-axis current Id and the q-axis current Iq are output to the subtracters 6a and 6b, respectively, and used for the calculation of the d-axis current deviation and the q-axis current deviation as described above.

  The current estimation unit 21 obtains the reproduction values Iuc, Ivc, and Iwc of the three-phase phase currents and the values of the d-axis current Id and the q-axis current Iq by the operation of each component as described above. it can.

  The position / speed estimation unit 15 estimates and calculates the rotor position of the AC motor 4 (when the AC motor 4 is a synchronous motor) or the secondary magnetic flux phase (when the AC motor 4 is an induction motor), and calculates the phase of the calculation result. Output as angle θdc. Further, based on the value of the phase angle θdc, the electric angular velocity ω1 of the AC motor 4 is estimated and calculated, and the calculation result is output. In addition, since these estimation calculation methods are well-known, description is abbreviate | omitted.

  The speed conversion unit 16 converts the electrical angular velocity ω1 output from the position / speed estimation unit 15 into a rotational speed ωr based on the number P of poles of the AC motor 4. The rotational speed ωr obtained by the speed converter 16 is input to the q-axis current command generator 1 and used to generate the q-axis current command Iq *.

  The inverter 3 is connected to the AC motor 4, generates an AC voltage based on the PWM signal input from the PWM generator 11 of the control device 2, and supplies the AC voltage to the AC motor 4. The inverter 3 includes a DC power supply 31, an inverter main circuit 32, an output predriver 33, a DC bus 34, and a shunt current detector 35.

  DC power supply 31 is connected to inverter main circuit 32 via DC bus 34 and generates DC voltage VDC.

  The inverter main circuit 32 includes three-phase switching elements Sup, Svp, and Swp provided on the positive electrode side, and three-phase switching elements Sun, Svn, and Swn provided on the negative electrode side, respectively. The switching elements Sup, Svp and Swp are connected to the positive electrode side of the DC power supply 31 via the DC bus 34, and the switching elements Sun, Svn and Swn are connected to the negative electrode side of the DC power supply 31 via the DC bus 34. Each is connected. Between the positive-side switching elements Sup, Svp, Swp and the negative-side switching elements Sun, Svn, Swn, AC lines connected to the U-phase, V-phase, and W-phase of the AC motor 4 are connected.

  The output pre-driver 33 is based on the aforementioned PWM signals UP, VP, WP, UN, VN and WN output from the PWM generator 11 of the control device 2, and the switching elements Sup, Svp, Swp, The switching states of Sun, Svn and Swn are controlled respectively. By controlling the switching state of each switching element by the output pre-driver 33, the DC voltage VDC supplied from the DC power supply 31 is converted into a three-phase AC voltage Vu, Vv, Vw and output to the AC motor 4. The

  The shunt current detector 35 is a circuit for overcurrent protection that detects a direct current flowing through the direct current bus 34, and is provided on the direct current bus 34. The detection result of the direct current by the shunt current detector 35 is output to the current estimation unit 21 of the control device 2 as the above-described direct current signal IDC.

  Note that the configuration of the AC motor drive system described above is based on a sensorless vector control system, but the present invention can also be applied to vector control with a sensor.

  The AC motor drive system according to the present embodiment has the configuration as described above. The main characteristic part of the present invention in this configuration is the voltage correction unit 10 for correcting the voltage command and the sampling timing of the DC current signal IDC by the sample hold circuit 12. These will be described in detail below.

  First, the state of pulse width modulation (PWM) when voltage command correction according to the present invention is not performed will be described with reference to the examples of FIGS. FIG. 2 is a diagram illustrating an example of a PWM waveform and a switch state when the voltage command is not corrected. In the control device 2, it is assumed that, for example, voltage commands Vu *, Vv * and Vw * as shown in FIG. When the voltage command is not corrected by the voltage correction unit 10, the PWM generation unit 11 compares these voltage commands with a triangular wave carrier, respectively, to thereby generate PWM signals UP, VP, WP as shown in FIG. PWM signals UN, VN, and WN are generated.

  In the inverter 3, the switching state of the switching elements Sup, Svp, Swp, Sun, Svn, Swn of the inverter main circuit 32 is changed by the output pre-driver 33 according to the PWM signals UP, VP, WP, UN, VN, WN. Each is controlled. As a result, the switch state of each switching element changes as shown in FIG. In FIG. 2, the switching states of the U phase, the V phase, and the W phase are respectively expressed as voltage vectors in association with 1 or 0 in parentheses. For example, in the voltage vector V (1, 0, 0), the switching element Sup on the positive electrode side is on (the switching element Sun on the negative electrode side is off) in the U phase, and the switching element Svn on the negative electrode side in the V phase and W phase. , Swn is ON (switching elements Svp and Swp on the positive electrode side are OFF).

  The inverter 3 can output eight types of voltage vectors to the AC motor 4 according to combinations of switch states of the switching elements Sup, Svp, Swp, Sun, Svn, and Swn. That is, by turning on all the switching elements Sup, Svp, Swp on the positive side and turning off all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (1, 1, 1) is output. be able to. On the contrary, by turning off all the switching elements Sup, Svp, Swp on the positive side and turning on all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (0, 0 , 0) can be output. These voltage vectors are all zero vectors in which no phase current flows. Further, in at least one of the U phase, the V phase, and the W phase, the positive side switching element is turned on, the negative side switching element is turned off, and in the other phases, the positive side switching element is turned off. The negative side switching element is turned on. Thus, in addition to the zero vector, each of the six types of voltage vectors V (1, 0, 0), V (0, 1, 0), V (0, 0, 1), V (1, 1, 0), V (1, 0, 1) and V (0, 1, 1) can be output.

  FIG. 3 is a diagram showing the states of currents flowing through the switching elements Sup, Svp, Swp, Sun, Svn, and Swn of the inverter main circuit 32 in the respective switch states indicated by voltage vectors in FIG. FIG. 3A shows the state of current in the voltage vector V (1,1,1), and FIG. 3B shows the state of current in the voltage vector V (1,1,0). 3 (c) shows the state of current in the voltage vector V (1, 0, 0), and FIG. 3 (d) shows the state of current in the voltage vector V (0, 0, 0). Show. In these drawings, the portion through which the current flows is indicated by a thick line.

As shown in FIGS. 3 (b) and 3 (c), the voltage vector V (1,1,0) or V (1,0,0) has a DC bus of the inverter 3 via the shunt current detector 35. A direct current shunt current I _DC corresponding to the W-phase phase current Iw or the U-phase phase current Iu flows through 34. Therefore, by sampling as described above the direct current signal IDC obtained by detecting the shunt current I _DC at switch state corresponding to these voltage vectors, two phase currents Iw, can be detected Iu It becomes.

4, each phase of the motor current flowing to the AC motor 4 (phase currents) Iu, and Iv and Iw, a diagram showing the relationship between the shunt current I _DC and the switch state flows through the DC bus 34. For example, for each switch state shown in FIG. 4 (c), together with the flow phase current as shown in FIG. 4 (a), the shunt current I _DC shown in FIG. 4 (b) flows to the DC bus 34. Accordingly, as shown in FIG. 4 (b), by sampling the shunt current I _DC for each switch state, it is possible to detect the phase current Iw, Iu.

When the phase current detection method as described above is employed, it is necessary to perform current sampling and analog-digital conversion twice within a half cycle of the carrier signal. In addition, it is necessary to arbitrarily set the current sampling timing in accordance with the change in the switch state. However, there is a problem that microcomputers that can realize these functions are generally expensive. Further, immediately after the switch state is changed, because it contains a lot of current ripple to shunt current I _DC in accordance with the switching of the switch, there is also this problem current detection error is sampled increases. In particular, such a current detection error tends to include an offset component, which greatly affects torque accuracy.

  One method for solving these problems is to correct the voltage commands Vu *, Vv * and Vw * to fix the current sampling timing. The specific method will be described below with reference to FIGS. FIG. 5 is a diagram showing an example of a PWM waveform and a switch state when a voltage command is corrected based on the conventional two-phase modulation method, and FIG. 6 is a diagram different from FIG. It is a figure which shows the example of the PWM waveform at the time of correct | amending and a switch state.

In the example of FIG. 5, the voltage command Vu *, Vv *, and Vw * before correction shown in FIG. 2 is raised without changing their interval, and the highest voltage command (here, Vu *). Is corrected so as to coincide with the apex of the triangular wave carrier to obtain corrected voltage commands Vu **, Vv ** and Vw **. In this case, when a current is sampled at the timing of the apexes of the triangular wave carrier, it can detect the phase currents Iu of the U phase as a shunt current I _DC, can not be detected for the other phase currents.

  Therefore, as shown in FIG. 6, each voltage command Vu *, Vw **, Vw ** is corrected so that the magnitude of the corrected voltage commands Vu ** and Vw ** alternately coincide with the apex of the triangular wave carrier. Correct Vv * and Vw *. The corrected voltage commands Vu **, Vv **, and Vw ** obtained by this correction are the intervals between the maximum value and the intermediate value (in the example of FIG. 6, when the voltage command Vu ** is the maximum, the voltage command The interval between Vv ** and the interval between voltage command Vu ** when voltage command Vw ** is maximum is set to be equal to or greater than a predetermined minimum voltage. Further, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged for two consecutive cycles of the triangular wave carrier, the interval between the average values is the original voltage command Vu * before correction. , Vv * and Vw * are set to be equal to each other.

When the voltage commands Vu *, Vv *, and Vw * are corrected as shown in FIG. 6 and current sampling is performed at the timing of the apex of the triangular wave carrier, the U-phase phase current Iu and the W-phase phase current Iw are shunted alternately. it can be detected as a current _{I DC.} 5 and 6, the example in which the voltage commands Vu *, Vv *, and Vw * are corrected in accordance with the apex portion of the triangular wave carrier has been described. Conversely, the voltage command Vu * is adjusted in accordance with the valley portion of the triangular wave carrier. , Vv * and Vw * can be corrected.

However, when the voltage command correction method and the phase current detection method as shown in FIG. 6 are adopted, the corrected voltage commands Vu **, Vv **, and Vw ** are biased toward the positive peak direction of the triangular wave carrier. Corrected. In this way, there is a tendency that the current ripples contained in the shunt current I _DC increases. Furthermore, since current sampling is performed at the timing of the apex of the triangular wave carrier, a current containing many harmonic ripples is sampled. As a result, there is a problem that current detection error becomes large and torque accuracy is deteriorated.

Therefore, in the present invention, the voltage correction unit 10 corrects the voltage commands Vu *, Vv *, and Vw * by the method described below. Then, the voltage command Vu ** after correction, the DC current signal IDC in accordance with the shunt current _{I DC} flowing through the DC bus 34 of inverter 3 by the PWM signal based on the Vv ** and Vw **, sample and hold circuit 12 The sampling is performed at the timing as described below.

  FIG. 7 is a diagram showing an example of a PWM waveform and a switch state when the voltage command is corrected according to the present invention.

  In FIG. 7, in the first period of the triangular wave carrier (period [A] in FIG. 7), the voltage commands Vu *, Vv *, and Vw * before correction shown in FIG. Thus, the corrected voltage commands Vu **, Vv ** and Vw ** are obtained and PWM control is performed. Here, the original voltage command before correction has a relationship of Vu *> Vv *> Vw *. In the following description, the phase with the maximum voltage command value before correction is the maximum phase (here, U phase), the minimum phase is the minimum phase (here, W phase), and the medium phase is the intermediate phase (here, V phase).

  The correction amount of each voltage command in the period A is such that the sum of the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier, is 0 by subtraction. Is set to That is, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged for the period A, the intervals between the average values are the original voltage commands Vu *, Vv *, and Vw before correction. * It is set to be equal to the distance between each other.

  By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in period A, zero vector V (1, 1, 1), voltage vector V (1, 1, 0), voltage vector V (1, 0, 0), zero vector V (0, 0, 0), The voltage vector V (0, 0, 1) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.

When the voltage vector other than zero vectors V (1, 1, 1) or V (0,0,0) is output flows to the DC bus 34 phase current corresponding to the voltage vector as a shunt current _{I DC,} It is detected by the shunt current detector 35. FIG. 8 is a table showing the relationship between the voltage vector in the period A and the detected phase current. In the table of FIG. 8, each voltage vector other than the zero vector output in order in the period A is represented as a first voltage vector, a second voltage vector, and a third voltage vector, respectively.

From the table of FIG. 8, when the first voltage vector V (1, 1, 0) is output and when the third voltage vector V (0, 0, 1) is output, the W-phase current Iw is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 1, 0) and the third voltage vector V (0, 0, 1), the signs representing the switch states of the respective phases are inverted, so that the phase current Iw flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I _DC, which are not subject to sampling by the sample and hold circuit 12.

10, in the first voltage vector V (1,1,0) and a third voltage vector V (0,0,1), is a diagram illustrating a state of a phase current is detected as a shunt current _{I DC} . FIG. 10A shows the state of the switch state and the phase current in the first voltage vector V (1,1,0), and FIG. 10B shows the third voltage vector V (0,0). , 1) shows the state of the switch and the phase current.

As shown in FIG. 10A, the phase current Iw flows from the AC motor 4 to the inverter 3 in the first voltage vector V (1, 1, 0). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the phase current -Iw sign is negative is detected as a shunt current I _DC. On the other hand, as shown in FIG. 10B, the phase current Iw flows from the inverter 3 to the AC motor 4 in the third voltage vector V (0, 0, 1). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the code is positive matrix phase current Iw is detected as a shunt current I _DC.

  Here, when a voltage vector other than the zero vector is output from the inverter 3, the current waveform changes greatly, so that the phase current of each phase includes a pulsation component other than the fundamental wave component necessary for control. . However, like the first voltage vector V (1, 1, 0) and the third voltage vector V (0, 0, 1), voltage vectors having opposite signs are included in one period of the triangular wave carrier. If the average value is obtained by detecting the phase currents that are applied and flowing through each of them, the pulsation component can be canceled, so that highly accurate detection becomes possible.

  In other words, the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG. The phase currents -Iw and Iw can be detected in accordance with the voltage vector. Thereby, it is possible to detect the phase current Iw excluding the pulsation component.

Further, as shown in FIG. 7, in the period A, the corrected voltage command Vv ** for the intermediate phase is always corrected to zero. Thus, the zero cross timing of the triangular wave carrier, i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I _DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current -Iw corresponding to (1, 1, 0) and the phase current Iw corresponding to the third voltage vector V (0, 0, 1) can be detected. Accordingly, in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period A and the downstream section of the triangular wave carrier in the second half of period A. .

  In FIG. 7, in the next period of the triangular wave carrier (period [B] in FIG. 7), the voltage commands Vu *, Vv *, and Vw * before correction shown in FIG. It is deformed as shown in the figure by a method different from the period A. Thus, corrected voltage commands Vu **, Vv ** and Vw ** are obtained and PWM control is performed.

  As for the correction amount of each voltage command in the period B, similarly to the period A described above, the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier. The total is set to be 0 by subtraction. That is, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged over the period B, the intervals between the average values are the original voltage commands Vu *, Vv *, and Vw before correction. It is set to be equal to the interval between *.

  By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in the period B, the zero vector V (1, 1, 1), the voltage vector V (1, 0, 1), the voltage vector V (1, 0, 0), the zero vector V (0, 0, 0), The voltage vector V (0, 1, 0) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.

When the voltage vector other than zero vectors V (1, 1, 1) or V (0,0,0) is output flows to the DC bus 34 phase current corresponding to the voltage vector as a shunt current _{I DC,} It is detected by the shunt current detector 35. FIG. 9 is a list showing the relationship between the voltage vector in the period B and the detected phase current. In the table of FIG. 9, similarly to FIG. 8, the voltage vectors other than the zero vector output in order in the period B are represented as first, second, and third voltage vectors, respectively.

From the table of FIG. 9, when the first voltage vector V (1, 0, 1) is output and when the third voltage vector V (0, 1, 0) is output, the V-phase current Iv is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 0, 1) and the third voltage vector V (0, 1, 0), the signs representing the switch states of the respective phases are inverted, so that the phase current Iv flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I _DC, which are not subject to sampling by the sample and hold circuit 12.

  As described above, the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG. The phase currents -Iv and Iv can be detected according to the first and third voltage vectors. Thereby, it is possible to detect the phase current Iv excluding the pulsation component.

Further, as shown in FIG. 7, in the period B, the corrected voltage command Vw ** for the minimum phase is always corrected to zero. Thus, the zero cross timing of the triangular wave carrier, i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I _DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current −Iv corresponding to (1, 0, 1) and the phase current Iv corresponding to the third voltage vector V (0, 1, 0) can be detected. Accordingly, in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period B and the downstream section of the triangular wave carrier in the second half of period B. .

As described above, the period A, in B, by detecting the shunt current I _DC respectively fixed timing the triangular wave carrier is zero (the middle value), it is possible to detect the phase currents of different two-phase . Therefore, the sample hold circuit 12 may sample the DC current signal IDC at a fixed timing every half cycle of the triangular wave carrier. Therefore, the sample hold circuit 12 can be realized by using an inexpensive microcomputer.

The above periods A, Repeat B, the sample and hold circuit 12 to sample the DC current signal IDC in accordance with the shunt current I _DC, it is possible to detect the phase current of two phases. Based on the detection result of the phase current obtained in this way, the current reproduction unit 13 can obtain the reproduction values Iuc, Ivc, and Iwc of the phase current of each phase.

  FIG. 11 is a diagram showing the voltage vectors output from the inverter 3 on the αβ coordinates. When voltage vectors having different signs are expressed on αβ coordinates, they have a relationship as shown in FIG.

  As described above, the control device 2 controls the inverter 3 so that the voltage correction unit 10 and the PWM generation unit 11 output any one of the voltage vectors excluding the zero vector as the first voltage vector. Then, after outputting the first voltage vector, the inverter 3 is controlled so as to output any voltage vector different from the first voltage vector, excluding the zero vector, as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and then the inverter 3 is output so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector. To control. Such control is performed in the period A corresponding to the first period of the triangular wave carrier, and in the period B corresponding to the next period, the first voltage vector and the third voltage vector are different types of voltages from the previous period. The inverter 3 is controlled so as to output each vector. By controlling the driving of the AC motor 4 by controlling the inverter 3, the sampling of the DC current signal IDC in the sample and hold circuit 12 is performed at a fixed timing every half cycle of the triangular wave carrier, and the result is used to reproduce the current. The reproduction value Iuc, Ivc, Iwc of the three-phase phase current can be obtained in the unit 13.

  In the above description, the content of each voltage vector output other than the zero vector changes according to the magnitude relationship of the original voltage command before correction. Accordingly, the voltage vectors output as the first, second, and third voltage vectors in the periods A and B are not limited to the above example.

  Next, details of the operation of the voltage correction unit 10 will be described. FIG. 12 is a diagram illustrating details of correction of the voltage command by the voltage correction unit 10.

  As described above with reference to FIG. 7, in the control device 2 of the present embodiment, the two shunt current detectors 35 of the inverter 3 perform two periods in two consecutive periods of the triangular wave carrier, that is, the period A and the period B described above. The voltage correction unit 10 corrects the voltage command for each phase so that the phase current can be detected. In FIG. 12, two continuous cycles of the triangular wave carrier are divided into half cycles, and the respective periods are represented as Tc0, Tc1, Tc2, and Tc3.

  In the periods Tc0 and Tc1 corresponding to the first period (period A) of the two consecutive periods of the triangular wave carrier, the voltage correction unit 10 outputs the intermediate phase voltage command without changing the interval between the voltage commands of each phase. Always 0. On the other hand, in the periods Tc2 and Tc3 corresponding to the next cycle (period B), the voltage command of the minimum phase is always set to 0 without changing the interval between the voltage commands of each phase. In FIG. 12, the voltage commands for the respective phases thus determined are shown as the maximum phase voltage command Vmax, the intermediate phase voltage command Vmid and the minimum phase voltage command Vmin before shifting.

  The voltage correction unit 10 calculates the voltage commands Vmax0, Vmid0, and Vmin0 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc0 by the following equations (1) to (3). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax of the maximum phase before the shift and the voltage command Vmid of the intermediate phase. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the difference between the voltage command Vmin for the minimum phase before the shift and the voltage command Vmid for the intermediate phase and a predetermined required shift amount Vshift. By shifting the voltage command for the maximum phase and the minimum phase based on these shift amounts, the voltage command Vmax0 for the maximum phase and the voltage command Vmin0 for the minimum phase after the shift can be obtained. In FIG. 12, the shift amounts of the maximum phase and the minimum phase represented by the equations (1) and (3) are indicated by arrows in the drawing. Equation (2) indicates that the shift amount with respect to the intermediate phase is zero.

Maximum phase: Vmax0 = 2 (Vmax−Vmid) (1)
Intermediate phase: Vmid0 = Vmid = 0 (2)
Minimum phase: Vmin0 = 2 (Vmin−Vmid) −Vshift (3)

Vshift in the equation (3) represents a necessary shift amount set in advance. This necessary shift amount Vshift can be set based on the output time of the voltage vector necessary for current detection. For example, required before the basis ringing and size of shunt current I _DC, and the like set value of the dead time of the switching elements, to detect the shunt current I _DC flowing from the change of the voltage vector to the DC bus 34 of the inverter 3 Time is determined, and the required shift amount Vshift can be determined according to the time. Alternatively, by observing the waveform of the shunt current I _DC, it may determine the required shift amount Vshift based on the observation result. From equation (3), it can be seen that the corrected minimum phase voltage command Vmin0 is smaller than the value obtained by subtracting the required shift amount Vshift from 0, as shown in FIG.

  The voltage correction unit 10 calculates the voltage commands Vmax1, Vmid1, and Vmin1 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc1 by the following equations (4) to (6). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the necessary shift amount Vshift. The maximum phase voltage command Vmax1 and the minimum phase voltage command Vmin1 can be obtained by shifting the maximum phase and minimum phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the minimum phase represented by the equations (4) and (6) are indicated by arrows in the drawing. Equation (5) indicates that the shift amount with respect to the intermediate phase is zero.

Maximum phase: Vmax1 = 0 (4)
Intermediate phase: Vmid1 = Vmid = 0 (5)
Minimum phase: Vmin1 = Vshift (6)

  From equation (6), as shown in FIG. 12, it can be seen that the corrected minimum phase voltage command Vmin1 is equal to the required shift amount Vshift.

  The voltage correction unit 10 calculates the voltage commands Vmax2, Vmid2, and Vmin2 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc2 by the following equations (7) to (9). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax for the maximum phase before the shift and the voltage command Vmin for the minimum phase. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the necessary shift amount Vshift. Then, the maximum phase voltage command Vmax2 and the intermediate phase voltage command Vmid2 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the intermediate phase represented by the equations (7) and (8) are indicated by arrows in the drawing. Equation (9) indicates that the shift amount with respect to the minimum phase is zero.

Maximum phase: Vmax2 = 2 (Vmax-Vmin) (7)
Intermediate phase: Vmid2 = −Vshift (8)
Minimum phase: Vmin2 = Vmin = 0 (9)

  From equation (8), it can be seen that the corrected intermediate phase voltage command Vmid2 is equal to the value obtained by subtracting the required shift amount Vshift from 0, as shown in FIG.

  The voltage correction unit 10 calculates the voltage commands Vmax3, Vmid3, and Vmin3 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc3 by the following equations (10) to (12). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the difference between the voltage command Vmid for the intermediate phase before the shift and the voltage command Vmin for the minimum phase and the required shift amount Vshift. The maximum phase voltage command Vmax3 and the intermediate phase voltage command Vmid3 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the intermediate phase represented by the expressions (10) and (11) are indicated by arrows in the drawing. Expression (12) indicates that the shift amount with respect to the minimum phase is zero.

Maximum phase: Vmax3 = 0 (10)
Intermediate phase: Vmid3 = 2 (Vmid−Vmin) + Vshift (11)
Minimum phase: Vmin3 = Vmin = 0 (12)

  From equation (10), it can be seen that the corrected intermediate phase voltage command Vmid3 is larger than the required shift amount Vshift, as shown in FIG.

  In the control device 2 of the present embodiment, the voltage correction unit 10 performs the correction as described above, whereby the sample and hold circuit 12 detects the current of the minimum phase and the intermediate phase having different signs in each period of Tc0 to Tc3. It can be done once. Based on the result of this current detection, the current reproducer 13 can obtain the reproduced values Iuc, Ivc, and Iwc of the three-phase currents. As is well known, in PWM control, as long as the relationship of the line voltage of each phase is maintained, changing the phase voltage of each phase does not affect the fundamental wave component to AC motor 4. In each period of the above Tc0 to Tc3, the voltage correction unit 10 corrects the voltage command of each phase so that the average value of the line voltage in these periods matches the original line voltage before correction. Yes. Therefore, there is no influence on the fundamental wave of the AC motor 4 due to the correction of the voltage command.

  According to the 1st Embodiment of this invention demonstrated above, there exist the following effects.

(1) The control device 2 of the AC motor 4 is connected to an inverter 3 connected to the three-phase AC motor 4, and in the inverter 3, switching elements Sup, Svp, provided with three phases respectively on the positive electrode side and the negative electrode side, The drive of the AC motor 4 is controlled according to the switching states of Swp, Sun, Svn and Swn. The inverter 3 can output eight voltage vectors to the AC motor 4 in accordance with the switching state of each switching element. This voltage vector includes the switching state in which all the switching elements Sup, Svp, Swp on the positive side are on and all the switching elements Sun, Svn, Swn on the negative side are off, and all the switching elements Sup on the positive side. , Svp, Swp are off, and two types of zero vectors V (0,0,0) and V (1, corresponding to the switching states where all the switching elements Sun, Svn, Swn on the negative side are on, respectively. 1, 1). The control device 2 of the AC motor 4 controls the inverter 3 so as to output any voltage vector except the zero vector as the first voltage vector, and after outputting the first voltage vector, the zero vector is removed. The inverter 3 is controlled so as to output any voltage vector different from the first voltage vector as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and after outputting the zero vector, a voltage vector obtained by inverting the first voltage vector is used as the third voltage vector. The inverter 3 is controlled to output. Thereby, the drive of the AC motor 4 is controlled. Since it did in this way, the control apparatus of the alternating current motor which can detect a phase current with high precision using an inexpensive microcomputer is realizable.

(2) The control device 2 controls the inverter 3 so as to repeatedly output the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector in order. At this time, the inverter 3 is controlled so that different types of voltage vectors from the previous time are output as the first voltage vector and the third voltage vector, respectively. Since it did in this way, the phase current of two phases which are different from each other can be detected, and the phase current of each phase can be obtained based on this.

(3) The control device 2 uses the current estimation unit 13 based on the detection result of the DC shunt current I _DC flowing in the DC bus 34 of the inverter 3 during the output period of the first voltage vector and the third voltage vector. Then, the phase current of the AC motor 4 is obtained. Since it did in this way, in the electric current estimation part 13, the phase current of each phase can be calculated | required and the reproduction values Iuc, Ivc, and Iwc can be calculated | required.

(4) The control device 2 includes a voltage command generator 20 that generates three-phase voltage commands Vu *, Vv *, and Vw *, a voltage corrector 10 that corrects these voltage commands, and a PWM generator 11. Prepare. The PWM generator 11 compares the corrected voltage commands Vu **, Vv **, and Vw ** corrected by the voltage correction unit 10 with a triangular wave carrier having a predetermined period, and based on the comparison result, the three-phase PWM waveform is generated. By outputting the PWM waveform generated by the PWM generator 11 to the inverter 3 in this way, the control device 2 controls the switching state of the switching elements Sup, Svp, Swp, Sun, Svn and Swn, and the AC motor 4 Control the drive. Since it did in this way, according to the voltage command after correction | amendment, the drive of the alternating current motor 4 can be controlled reliably.

(5) The voltage correction unit 10 sets the intermediate voltage command Vmid to 0 in the three-phase voltage command and the maximum voltage command Vmax in the three-phase voltage command in the first cycle out of the two consecutive cycles of the triangular wave carrier. And the voltage command of each phase is corrected by shifting the minimum voltage command Vmin. In the next cycle following the first cycle, the minimum voltage command Vmin is set to 0, and the maximum voltage command Vmax and the intermediate voltage command Vmid are shifted to correct the voltage command of each phase. Since it did in this way, while said 1st, 2nd and 3rd voltage vector is output sequentially, these voltage vectors can be changed for every period of a triangular wave carrier.

(6) In the first half period Tc0 of the first cycle, the voltage correction unit 10 determines the maximum value based on the difference between the maximum voltage command Vmax and the intermediate voltage command Vmid according to the above-described equations (1) and (3). The shift amount with respect to the minimum voltage command Vmin is determined based on the difference between the minimum voltage command Vmin and the intermediate voltage command Vmid and the predetermined required shift amount Vshift. Further, in the second half period Tc1 of the first cycle, the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (4) and (6), and the necessary shift amount. A shift amount with respect to the minimum voltage command Vmin is determined based on Vshift. On the other hand, in the first half period of the next cycle, the shift amount with respect to the maximum voltage command Vmax is determined based on the difference between the maximum voltage command Vmax and the minimum voltage command Vmin according to the above-described equations (7) and (8). At the same time, the shift amount for the intermediate voltage command Vmid is determined based on the necessary shift amount Vshift. In the second half of the next cycle, the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (10) and (11), and the intermediate voltage command Based on the difference between Vmid and the minimum voltage command Vmin and the required shift amount Vshift, the shift amount for the intermediate voltage command Vmid is determined. Since it did in this way, correction | amendment of the voltage command of each phase as mentioned above can be performed appropriately and reliably.

(7) above needs shift Vshift can be set according to the time to detect a DC shunt current I _DC flowing from the switching state is changed to the DC bus 34 of the inverter 3 of each switching element. In this way, it is possible to accurately detect the shunt current I _DC corresponding to the phase current.

(8) The control device 2, the sample hold circuit 12, obtains the detection result of the DC shunt current I _DC to when the triangular wave carrier takes an intermediate value between the minimum and maximum values flowing through the DC bus 34 of the inverter 3. Since this is done, it is possible to obtain the shunt current I _DC corresponding to the phase currents of two different phases for each fixed timing.

(Second Embodiment)
Next, a second embodiment of the present invention will be described. FIG. 13 is a diagram showing a configuration of a control device 2B according to the second embodiment of the present invention.

  In the first embodiment described above, as shown in FIG. 12, an example is shown in which the present invention is applied to sensorless drive control in which the AC motor 4 is driven with a sinusoidal current. Such sensorless drive control has been developed for various uses, but performance degradation at low speeds is a problem. As a solution, for example, a technique based on 120-degree energization driving as disclosed in Japanese Patent Application Laid-Open No. 2009-189176 is well known. In this method, 120-degree energization driving is performed by switching the switching state of only the two-phase switching elements among the three-phase switching elements, so that it can be realized with an inexpensive microcomputer. On the other hand, since it is based on 120-degree conduction drive, there is a problem that torque pulsation is large and there are many harmonic components included in the current, so that loss is large.

  Therefore, in this embodiment, by switching the switching state of only the two-phase switching elements among the three-phase switching elements in the low-speed range where the AC motor is driven at a low speed, The inverter 3 is controlled to perform 120-degree energization driving. Further, in other medium and high speed ranges, the inverter 3 is controlled to perform the three-phase energization drive by switching the switching state of each of the three-phase switching elements by the sensorless drive control as described in the first embodiment. To do. Thereby, it is set as the structure which compensates both faults. Since the sine wave sensorless driving method of the present invention is a method suitable for an inexpensive microcomputer as described above, it is very suitable for a combination with 120-degree energization driving that can be realized originally by an inexpensive microcomputer.

  As shown in FIG. 13, in the control device 2B according to the present embodiment, 120-degree energization drive capable of sensorless drive control in the low speed region is realized in the configuration of the control device 2 according to the first embodiment shown in FIG. The structure for this is added, and these can be switched now. By using this control device 2B instead of the control device 2 of FIG. 1, an AC motor drive system according to the second embodiment of the present invention can be realized.

  13, the same components as those in FIG. 1 are denoted by the same reference numerals. In addition to the configuration of FIG. 1, the control device 2B shown in FIG. 13 includes a 120-degree energization unit 17, a 120-degree PWM unit 18 that performs PWM for 120-degree energization drive, a 120-degree energization drive, and a three-phase It consists of change-over switches 19a to 19e for switching energization drive. Further, in place of the d-axis current controller 7 and the position / speed estimation unit 15 of FIG. 1, a d-axis current controller 7B and a position / speed estimation unit 15B are provided.

  The 120-degree energization unit 17 outputs a phase value θdcB for performing 120-degree energization driving, and a speed estimation value ω1cB obtained by calculation from the result of performing 120-degree energization driving. The phase value θdcB is a discrete value every 60 degrees. The phase value θdcB and the phase value θdcS which is a continuous value output from the position / velocity estimation unit 15B are switched by the changeover switch 19c and output to the dq inverse converter 9. Each of the changeover switches 19a to 19e is switched to the “0” side during 120-degree energization driving, and is switched to the “1” side during three-phase energization driving.

  The changeover switch 19e switches between the estimated speed value ω1cB output from the 120-degree energization unit 17 and the estimated speed value ω1cS output from the position / speed estimation unit 15B, and either one is used as the electrical angular velocity ω1c of the AC motor 4. Output to the speed converter 16.

  The changeover switch 19a switches between a PWM waveform for three-phase energization driving output from the PWM generator 11 and a PWM waveform for 120-degree energization driving output from the 120-degree PWM unit 18, and either one is switched to the inverter 3 Output to.

Changeover switch 19d is switched as a feedback current to the q-axis current controller 8, a q-axis current Iq which are coordinate transformation by dq converter 14, the sample-and-hold circuit 12 and a detection result of shunt current I _DC sampled . At the time of 120-degree conduction drive, the current control may be one, or may be directly fed back to the detection value of the shunt current I _DC.

  The changeover switch 19b switches between 0 and a value output from the d-axis current controller 7B, and outputs one of them to the dq inverse converter 9 as a d-axis voltage command Vd *. There is no concept of “d-axis” during 120-degree energization drive. Therefore, the changeover switch 19b switches so that the d-axis voltage command Vd * becomes zero. The q-axis voltage command Vq * output from the q-axis current controller 8 is commonly used during 120-degree energization driving and three-phase energization driving.

  The d-axis current controller 7B calculates the d-axis voltage command Vd * used in the three-phase energization drive directly from the q-axis current command Iq * without performing feedback control of the d-axis current Id. Here, when switching between 120-degree energization drive and three-phase energization drive, processing of values in the d-axis current controller 7B becomes a problem. For example, when the d-axis voltage command Vd * is obtained by PI control (proportional integral control), there are problems relating to the initial value setting of the integrator and problems relating to the operation during 120-degree conduction drive. Therefore, in the present embodiment, the d-axis current controller 7B determines the d-axis voltage command Vd * by feedforward calculation based on the q-axis current command Iq *, and does not perform feedback control. As a result, switching between the 120-degree energization drive and the three-phase energization drive can be performed smoothly. Instead of the q-axis current command Iq *, the d-axis voltage command Vd * may be determined by a feedforward calculation based on the q-axis current Iq estimated by the current estimation unit 21.

  The d-axis current controller 7B includes a first-order lag filter 71, a multiplier 72, a q-axis inductance setter 73, and a sign inverter 74. Since the arithmetic processing realized by these configurations is a feedforward calculation for the d-axis voltage of the AC motor 4, three-phase energization driving can be performed stably. Note that the time constant Tr of the first-order lag filter 71 may be set according to the response set value of the q-axis current controller 8, for example.

  The position / velocity estimation unit 15B includes an axis error estimator 151, a zero setter 152, a PI controller 153, a subtractor 6c, and a phase calculator 154. Such a configuration of the position / speed estimation unit 15B is well known as a medium / high speed sensorless position / speed estimator.

  The axis error estimator 151 estimates and calculates the deviation between the actual rotor position of the AC motor 4 that is a permanent magnet motor and the rotor position calculated by the control device 2B, and outputs the result as an axis error Δθdc. .

  The subtractor 6 c calculates a difference between “0” output from the zero setting unit 152 and the axis error Δθdc from the axis error estimator 151, and outputs the calculation result to the PI controller 153.

  The PI controller 153 performs control for setting the value of the axis error Δθdc to 0 by calculating the speed estimation value ω1cS so that the difference from the subtractor 6c becomes 0. This PI controller 153 constitutes a PLL (Phase Locked Loop). The estimated speed value ω1cS obtained by the PI controller 153 corresponds to the electrical angular velocity of the AC motor 4.

  The phase calculator 154 integrates the estimated speed value ω1cS from the PI controller 153 to obtain the phase value θdcS. The phase value θdcS is output from the position / velocity estimation unit 15B to the changeover switch 19c.

  According to the second embodiment of the present invention described above, in addition to the functions and effects (1) to (8) according to the first embodiment, the following functions (9) and (10) are further provided. There is an effect.

(9) In the low speed region where the AC motor 4 is driven at a low speed, the control device 2B switches the switching state of only the two-phase switching elements among the switching elements of the inverter 3 so as to perform 120-degree energization driving. 3 is controlled. Further, in the speed range other than the low speed range, the inverter 3 is controlled so that the switching state of the three-phase switching element is switched to perform the three-phase energization drive. Since it did in this way, a high-performance sensorless drive is realizable in all the speed ranges from a low speed range to a high speed range.

(10) The control device 2B includes a d-axis current controller 7B that outputs a d-axis voltage command Vd * used in three-phase current drive, and a q-axis voltage command commonly used in 120-degree current drive and three-phase current drive. A q-axis current controller 8 for outputting Vq *. The d-axis current controller 7B outputs a d-axis voltage command Vd * by a feedforward calculation based on a q-axis current command Iq * input from the outside or a q-axis current Iq that is a detection result of a current flowing in the AC motor 4. To do. Since it did in this way, switching with 120 degree energization drive and three phase energization drive can be performed smoothly.

  In the present embodiment, the d-axis voltage command Vd * is obtained by feedback calculation using the d-axis current controller 7 as described in the first embodiment without using the d-axis current controller 7B. It can also be output. However, the configuration as shown in FIG. 13 is more desirable from the viewpoint of realizing processing by a low-function microcomputer, which is a feature of the present invention.

(Third embodiment)
Next, a third embodiment of the present invention will be described. FIG. 14 is a diagram showing a configuration of an AC motor drive system 40 according to the third embodiment of the present invention.

  The AC motor drive system 40 shown in FIG. 14 is configured by integrating the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 described in the first embodiment. The q-axis current command generator 1, the control device 2, and the inverter 3 are packaged inside the AC motor 4 as a drive control board 41 mounted on one circuit board.

  The wiring to the integrated AC motor drive system 40 can be, for example, only the power line and the communication line shown in FIG. The power supply line is connected to the inverter 3 and is used to supply a DC power supply or the like to the switching element. The communication line is connected to the q-axis current command generator 1 and the control device 2 to transmit a rotational speed command input to the q-axis current command generator 1 and to transmit an operation state output from the control device 2. Used.

  According to the third embodiment of the present invention described above, since the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 are all integrated as the AC motor drive system 40, a small size is achieved. Can be realized. In addition, since the wiring between the respective components can be eliminated, the handling property can be improved.

  In the AC motor drive system 40, the configuration as described in the above-described second embodiment may be applied by using the control device 2B instead of the control device 2.

(Fourth embodiment)
Next, a fourth embodiment of the present invention will be described. FIG. 15 is a diagram showing a configuration of a hydraulic control system according to the fourth embodiment of the present invention.

  The hydraulic control system shown in FIG. 15 is used, for example, for hydraulic control of a transmission inside a vehicle, hydraulic control of a brake, and the like. This hydraulic control system includes a hydraulic control unit 25 and a hydraulic circuit 50.

  The hydraulic control unit 25 includes the q-axis current command generator 1, the control device 2, the inverter 3, the AC motor 4, and the oil pump 26 attached to the AC motor 4 described in the first embodiment. The oil pump 26 operates according to the driving of the AC motor 4 and controls the hydraulic pressure of the hydraulic circuit 50.

  The hydraulic circuit 50 includes a tank 51 that stores hydraulic oil, a relief valve 52 that keeps the hydraulic pressure below a set value, a solenoid valve 53 that switches a discharge destination of the hydraulic oil, and a cylinder 54 that operates as a hydraulic actuator.

  The oil pump 26 operates according to the drive of the AC motor 4 to generate hydraulic pressure, and drives the cylinder 54 that is a hydraulic actuator. At this time, when the hydraulic oil discharge destination is switched by the solenoid valve 53 in the hydraulic circuit 50, the load of the oil pump 26 changes, and a load disturbance to the hydraulic control unit 25 occurs. As a result, the AC motor 4 may stop due to a load that is several times greater than the steady-state pressure. Thus, even if the AC motor 4 is stopped, the hydraulic drive system according to the present embodiment performs the drive control of the AC motor 4 by the method described in the first embodiment, so that the rotor in the stopped state is obtained. The position can be estimated. Therefore, no problem occurs.

  In the conventional hydraulic control system, it has been essential to release the hydraulic pressure that is a great load on the AC motor 4 by the relief valve 52. However, according to the present embodiment, the relief valve 52 may be eliminated as described below.

  FIG. 16 is a diagram showing a modification of the configuration of the hydraulic control system according to the fourth embodiment of the present invention. This hydraulic control system is not provided with the relief valve 52 which is present in FIG. That is, in the hydraulic control system according to the present embodiment, the control performance in the entire speed range for the AC motor 4 can be improved, so that the hydraulic pressure can be controlled without a relief valve which is a mechanical protection device for avoiding an excessive load. It becomes.

  According to the fourth embodiment of the present invention described above, the present invention can be applied in hydraulic control for various uses.

  In the fourth embodiment, the application example to the hydraulic control for controlling the pressure of the hydraulic oil has been described. However, the fourth embodiment may be applied to the pressure control of another fluid such as water. That is, the present invention can be applied to various fluid pressure control systems that control fluid pressure by operating a pump by driving an AC motor.

  In the hydraulic control unit 25, the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.

(Fifth embodiment)
Next, a fifth embodiment of the present invention will be described. FIG. 17 is a diagram showing a configuration of a positioning system according to the fifth embodiment of the present invention.

  The positioning system shown in FIG. 17 includes the control device 2, the inverter 3, the AC motor 4, the q-axis current command generator 1C, and the positioning device 70 described in the first embodiment.

  The q-axis current command generator 1C includes a position controller 101, a speed controller 102, a subtracter 6d, and a subtractor 6e.

  The subtractor 6 d calculates the difference between the position command θ * input from the outside and the phase angle θdc obtained by the control device 2, and outputs the calculation result to the position controller 101. The position controller 101 obtains the speed command ωr * based on the calculation result from the subtractor 6d. The subtractor 6 e calculates the difference between the speed command ωr * from the position controller 101 and the rotational speed ωr of the AC motor 4 obtained by the control device 2, and outputs the calculation result to the speed controller 102. The speed controller 102 obtains a q-axis current command Iq * based on the calculation result from the subtractor 6 e and outputs it to the control device 2.

  The positioning device 70 is connected to the rotating shaft of the AC motor 4 as a load of the AC motor 4. This positioning device 70 is a device using, for example, a ball screw and can adjust the position of the object by operating in accordance with the driving of the AC motor 4. Note that no position sensor is attached to the positioning device 70, and the position controller 101 performs drive control of the AC motor 4 using the phase angle θdc output from the control device 2 as it is. Thereby, the operation of the positioning device 70 can be controlled, and the position of the object can be adjusted according to the position command θ *. Therefore, the position of the object can be controlled without using a position sensor.

  According to the fifth embodiment of the present invention described above, the present invention can be applied in position control of various objects.

  In the positioning system, the configuration as described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.

  Although various embodiments of the present invention have been specifically described above, the present invention is not limited to these embodiments, and it goes without saying that various modifications can be made without departing from the scope of the present invention. Yes. The above-described embodiments and modifications are merely examples, and the present invention is not limited to these contents as long as the features of the invention are not impaired.

The disclosure of the following priority application is hereby incorporated by reference.
Japanese patent application 2013 No. 172202 (filed on August 22, 2013)

  1: q-axis current command generator, 2: control device, 3: inverter, 4: AC motor, 5: d-axis current command generator, 6a to 6e: subtractor, 7: d-axis current controller, 8: q Axis current controller, 9: dq inverse converter, 10: voltage correction unit, 11: PWM generation unit, 12: sample hold circuit, 13: current reproduction unit, 14: dq converter, 15: position / speed estimation unit, 16: Speed conversion unit, 17: 120-degree energization unit, 18: 120-degree PWM unit, 19a to 19e: changeover switch, 20: voltage command generation unit, 21: current estimation unit, 31: DC power supply, 32: inverter main circuit 33: Output pre-driver, 34: DC bus, 35: Shunt current detector

Claims (10)

Control of an AC motor that is connected to an inverter connected to a three-phase AC motor and controls the driving of the AC motor according to the switching state of switching elements provided on the positive side and the negative side in the inverter, respectively. A device,
The inverter can output eight voltage vectors to the AC motor according to the switching state of the switching element,
The voltage vector includes a switching state in which all the switching elements on the positive electrode side are on and all the switching elements on the negative electrode side are off, and all the switching elements on the positive electrode side are off, Two types of zero vectors respectively corresponding to switching states in which all the switching elements are on,
The control device for the AC motor is:
Controlling the inverter to output any voltage vector except the zero vector as a first voltage vector;
After outputting the first voltage vector, the inverter is controlled to output any voltage vector different from the first voltage vector excluding the zero vector as a second voltage vector;
Controlling the inverter to output the zero vector after outputting the second voltage vector;
Controlling the inverter to output a voltage vector obtained by inverting the first voltage vector as a third voltage vector after outputting the zero vector ;
Controlling the inverter to repeatedly output the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector sequentially;
The control apparatus for an AC motor that controls the drive of the AC motor by controlling the inverter to output different types of voltage vectors as the first voltage vector and the third voltage vector, respectively. . In the control apparatus for an AC motor according to claim 1 ,
An AC electric motor comprising a current estimating unit that obtains a phase current of the AC electric motor based on a detection result of a direct current flowing through the DC bus of the inverter during an output period of the first voltage vector and the third voltage vector Control device. Control of an AC motor that is connected to an inverter connected to a three-phase AC motor and controls the driving of the AC motor according to the switching state of switching elements provided on the positive side and the negative side in the inverter, respectively. A device,
A voltage command generator for generating a three-phase voltage command;
A voltage correction unit for correcting the voltage command;
Comparing the corrected voltage command corrected by the voltage correction unit with a triangular wave carrier of a predetermined period, and a PWM generation unit that generates a three-phase PWM waveform based on the comparison result,
By outputting the PWM waveform generated by the PWM generator to the inverter, the switching state of the switching element is controlled, and the drive of the AC motor is controlled,
The inverter can output eight voltage vectors to the AC motor according to the switching state of the switching element,
The voltage vector includes a switching state in which all the switching elements on the positive electrode side are on and all the switching elements on the negative electrode side are off, and all the switching elements on the positive electrode side are off, Two types of zero vectors respectively corresponding to switching states in which all the switching elements are on,
The control device for the AC motor is:
Controlling the inverter to output any voltage vector except the zero vector as a first voltage vector;
After outputting the first voltage vector, the inverter is controlled to output any voltage vector different from the first voltage vector excluding the zero vector as a second voltage vector;
Controlling the inverter to output the zero vector after outputting the second voltage vector;
Controlling the inverter to output a voltage vector obtained by inverting the first voltage vector as a third voltage vector after outputting the zero vector;
The voltage correction unit is
In the first of two consecutive cycles of the triangular wave carrier, the intermediate voltage command is set to 0 by the three-phase voltage command, and the maximum voltage command and the minimum voltage command are shifted by the three-phase voltage command. By correcting the voltage command,
In the next period following the first period, the minimum voltage command is set to 0, and the maximum voltage command and the intermediate voltage command are shifted to correct the voltage command. In the control apparatus for an AC motor according to claim 3 ,
The voltage correction unit is
In the first half of the first cycle, a shift amount for the maximum voltage command is determined based on a difference between the maximum voltage command and the intermediate voltage command, and the minimum voltage command and the intermediate voltage command Determining a shift amount for the minimum voltage command based on the difference and a predetermined required shift amount;
In the latter half of the first cycle, the shift amount for the maximum voltage command is determined so that the corrected voltage command is 0, and the shift amount for the minimum voltage command is determined based on the required shift amount. And
In the first half of the next cycle, a shift amount for the maximum voltage command is determined based on a difference between the maximum voltage command and the minimum voltage command, and the intermediate voltage command is determined based on the required shift amount. Determine the shift amount for
In the second half of the next cycle, the shift amount for the maximum voltage command is determined so that the corrected voltage command becomes 0, and the difference between the intermediate voltage command and the minimum voltage command and the necessary shift A control apparatus for an AC motor that determines a shift amount with respect to the intermediate voltage command based on the amount. In the control device for an AC motor according to claim 4 ,
The required shift amount is a control device for an AC motor that is set according to a time from when a switching state of the switching element is changed until a DC current flowing through a DC bus of the inverter is detected. Control of an AC motor that is connected to an inverter connected to a three-phase AC motor and controls the driving of the AC motor according to the switching state of switching elements provided on the positive side and the negative side in the inverter, respectively. A device,
A voltage command generator for generating a three-phase voltage command;
A voltage correction unit for correcting the voltage command;
Comparing the corrected voltage command corrected by the voltage correction unit with a triangular wave carrier of a predetermined period, and a PWM generation unit that generates a three-phase PWM waveform based on the comparison result,
By outputting the PWM waveform generated by the PWM generator to the inverter, the switching state of the switching element is controlled, and the drive of the AC motor is controlled,
The inverter can output eight voltage vectors to the AC motor according to the switching state of the switching element,
The voltage vector includes a switching state in which all the switching elements on the positive electrode side are on and all the switching elements on the negative electrode side are off, and all the switching elements on the positive electrode side are off, Two types of zero vectors respectively corresponding to switching states in which all the switching elements are on,
The control device for the AC motor is:
Controlling the inverter to output any voltage vector except the zero vector as a first voltage vector;
After outputting the first voltage vector, the inverter is controlled to output any voltage vector different from the first voltage vector excluding the zero vector as a second voltage vector;
Controlling the inverter to output the zero vector after outputting the second voltage vector;
Controlling the inverter to output a voltage vector obtained by inverting the first voltage vector as a third voltage vector after outputting the zero vector;
A control apparatus for an AC motor that acquires a detection result of a DC current flowing in a DC bus of the inverter when the triangular wave carrier takes an intermediate value between a minimum value and a maximum value. Control of an AC motor that is connected to an inverter connected to a three-phase AC motor and controls the driving of the AC motor according to the switching state of switching elements provided on the positive side and the negative side in the inverter, respectively. A device,
The inverter can output eight voltage vectors to the AC motor according to the switching state of the switching element,
The voltage vector includes a switching state in which all the switching elements on the positive electrode side are on and all the switching elements on the negative electrode side are off, and all the switching elements on the positive electrode side are off, Two types of zero vectors respectively corresponding to switching states in which all the switching elements are on,
The control device for the AC motor is:
Controlling the inverter to output any voltage vector except the zero vector as a first voltage vector;
After outputting the first voltage vector, the inverter is controlled to output any voltage vector different from the first voltage vector excluding the zero vector as a second voltage vector;
Controlling the inverter to output the zero vector after outputting the second voltage vector;
Controlling the inverter to output a voltage vector obtained by inverting the first voltage vector as a third voltage vector after outputting the zero vector;
In the low speed range where the AC motor is driven at a low speed, the inverter is controlled such that only the two-phase switching elements among the switching elements are switched to perform a 120-degree conduction drive.
In the speed range other than the low speed range, the inverter is controlled so as to switch the switching state of the three-phase switching element and perform the three-phase energization drive,
A first current controller that outputs a first voltage command used in the three-phase energization drive;
A second current controller that outputs a second voltage command commonly used in the 120-degree energization drive and the three-phase energization drive,
The first current controller is a control device for an AC motor that outputs the first voltage command by a feedforward calculation based on a current command input from the outside or a detection result of a current flowing in the AC motor. A control device for an AC motor according to any one of claims 1 to 7 ,
An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
A three-phase AC motor that is driven and controlled in accordance with the switching state of the switching element by the AC motor control device,
The AC motor control device, the inverter, and the AC motor are integrated with each other. A control device for an AC motor according to any one of claims 1 to 7 ,
An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
A three-phase AC motor that is driven and controlled according to the switching state of the switching element by the AC motor control device;
A fluid pressure control system comprising: a pump that operates according to driving of the AC motor and controls fluid pressure. A control device for an AC motor according to any one of claims 1 to 7 ,
An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
A three-phase AC motor that is driven and controlled according to the switching state of the switching element by the AC motor control device;
A positioning system comprising a positioning device that operates according to driving of the AC motor and adjusts the position of an object.

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