JP2006320127A - Controller for inverter - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To compensate for an output voltage error Vtd caused by a short-circuit preventing period of an output voltage of an inverter 1 without discriminating the polarity of the output current of the inverter 1. <P>SOLUTION: A controller 100A includes a PWM output device 14 for outputting a PWM signal to the inverter 1 which converts a DC power into a three-phase electric power, and outputs it to a three-phase electric motor 2; a current controller 10 for calculating a command value V*dq of the output voltage of the inverter 1 in a rotational coordinate system, on the basis of the detected current value iαβ and the command current value i* of the inverter 1; a dead time compensation device 13a for correcting the command value V*dq of the output voltage in the rotational coordinate system to determine a compensation voltage value V*dq+ΔVdq, on the basis of a phase value θi of a command current value Idq; and coordinate conversion devices 11, 12 for performing coordinate transform of the compensation voltage value V*dq+ΔVdq from the rotational coordinate system to a three-phase fixed coordinate system to give it to the inverter 1. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to an inverter control device.

  2. Description of the Related Art Conventionally, some three-phase electric motor control devices include an inverter 1 that converts DC power into three-phase AC power and supplies the three-phase electric motor 2 as shown in FIG. 1).

  The inverter 1 has six transistors (specifically, IGBT, MOS-FET, etc.) U−, V−, W−, U +, V +, and W + that are bridge-connected. U−, V−, W−, U +, V +, and W + are sequentially switched based on the PWM signal (see FIG. 9A) to convert the DC power output from the DC power source Ba into three-phase AC power. Configured to convert.

  Here, when the six transistors U−, V−, W−, U +, V +, and W + are switched, if the switching is delayed and the upper arm transistor 1a and the lower arm transistor 1b are turned on at the same time, DC A short circuit occurs between the two terminals of the power supply Ba, causing overcurrent breakdown in both transistors 1a and 1b.

  Therefore, in order to prevent a short circuit between both terminals of the DC power supply Ba, a short-circuit prevention period td (see FIGS. 9B and 9C) in which both transistors 1a and 1b are simultaneously turned off in the transition period. The output is stopped at the output voltage of the inverter 1 for a certain period.

  However, when the output of the output voltage of the inverter 1 is stopped, an output voltage error (see FIG. 9D) occurs between the actual output voltage of the inverter 1 and the theoretical value of the output voltage of the inverter 1.

  Along with this, the actual output voltage of the inverter 1 becomes a voltage waveform (see symbol c in FIG. 10) offset from an ideal sine wave of the inverter 1 output voltage (see symbol b in FIG. 10). The phase electric motor cannot be properly controlled.

  Therefore, in the above-described Non-Patent Document 1, an inverter control device that corrects each output phase voltage of the inverter 1 is proposed in order to compensate for an output voltage error of the inverter 1. The control system of the inverter control device can be divided into two systems, a current system and a voltage system. In particular, as a control apparatus that can realize the control apparatus with software (ie, a computer program), It has become mainstream that the method is adopted.

  Here, as an example of a current-type inverter control device, the configuration of the control device 100 described in Non-Patent Documents 2 to 4 will be described with reference to FIG.

  The control device 100 includes a current control unit 10, a dq / αβ conversion unit 11, an αβ-uvw conversion unit 12, a dead time compensation unit 13, a PWM output unit 14, a vertical arm short circuit prevention period generation unit 15, and a uvw / αβ conversion. Means 16 and position estimation means 17 are provided.

  First, the current control means 10 uses the command current value i *, the detected current value iαβ, the estimated rotational position θ ^ re, and the estimated rotational speed ω ^ rm to generate signals such as PI control, current vector control, and current foodback control. Processing is performed to calculate a command voltage value V * dq in the rotating coordinate system.

  Here, the command current value i * is a command value (that is, target value) of the current value output from the inverter 1, and the detected current value iαβ is a detected value of the current value output from the inverter 1. The estimated rotational position θ ^ re is the estimated position of the rotor of the three-phase electric motor 2, the estimated rotational speed ω ^ rm is the estimated speed of the rotor of the three-phase electric motor 2, and the command voltage value V * dq. Is a command value of the voltage value output from the inverter 1.

  The detected current value iαβ, the estimated rotational position θ ^ re, and the estimated rotational speed ω ^ rm are calculated as described later.

  Next, the dq / αβ conversion means 11 converts the command voltage value V * dq from the rotating coordinate system (dq) to the two-phase fixed coordinate system (α-β), and the command voltage in the two-phase fixed coordinate system. The value V * αβ is calculated.

  Next, the αβ-uvw conversion means 12 converts the command voltage value V * αβ from the two-phase fixed coordinate system (α-β) to the three-phase fixed coordinate system (uvv-w), and the three-phase fixed coordinates. A command voltage value V * uvw in the system is calculated.

  Next, the dead time compensation means 13 detects the output current phase of the inverter 1 based on the detected current value Iuvw of the current detection sensor 22, and uses this detected output current phase to command the command voltage value V * uvw. In contrast, the “output error due to the short-circuit prevention period (dead time) td of the output voltage of the inverter 1” is compensated.

  Here, since the average error voltage (see symbol d in FIG. 10) due to the short-circuit prevention period has the same phase and opposite polarity as the output current of the inverter 1 (see symbol a in FIG. 10), the dead time compensation means 13 performs current detection. The polarity of the detected current value Iuvw of the sensor 3 is determined. Based on the determined polarity, a correction value ΔVuvw having a polarity opposite to the error average voltage is calculated, and the correction value ΔVuvw is added to the command voltage value V * uvw to obtain a compensation voltage command value (V * uvw + ΔVuvw). Is calculated.

  The detected current value Iuvw is an output current value of the inverter 1 detected by the current detection sensor 3.

  Next, the PWM output means 14 compares the compensation voltage command value (V * uvw + ΔVuvw) with a carrier wave (for example, a reference triangular wave), and calculates a PWM signal for each phase as a comparison result. Then, the upper and lower arm short circuit prevention period generation means 15 sets a short circuit prevention period for each PWM signal and outputs the set PWM signal to the inverter 1.

  Then, since the inverter 1 converts the DC power output from the DC power supply Ba into three-phase AC power, the three-phase electric motor 2 rotates based on the three-phase AC power output from the inverter 1.

  On the other hand, when the output current Iuvw output from the inverter 1 is detected by the current detection sensor 3, the detected current value Iuvw is coordinate-converted from the three-phase fixed coordinate system to the two-phase fixed coordinate system by the uvw / αβ conversion means 16. The detected current value iαβ in the two-phase fixed coordinate system is calculated.

  Then, the position estimation means 17 estimates the estimated rotational position θ ^ re and the estimated rotational speed ω ^ rm using the detected current value iαβ and the mathematical model of the three-phase electric motor 2 (that is, the circuit equation of the electric motor). To the current control means 10.

  Further, since the output current of the inverter 1 includes many ripples due to electromagnetic noise or the like, when the polarity of the output current of the inverter 1 is determined, the ripple may cause chattering due to the ripple.

Therefore, Patent Document 1 proposes a method in which a value that is offset from “0” is adopted as a threshold for polarity determination, and the polarity is determined by comparing the threshold with the output current of the inverter 1.
Power electronics circuit Chen, Tomita, Michiki, Okuma, 1999 IEEJ National Conference Proceedings. No. 1026. "Extended induced voltage observer for sensorless control of salient pole type brushless DC motor" Chen, Tomita, Michiki, Okuma 1999 Proceedings of National Conference on Industrial Applications of the Institute of Electrical Engineers of Japan, page 45, 1999. No. 229. "Realization of sensorless position / speed estimation of salient pole type brushless DC motor by disturbance observer". Chen, Tomita, Senju, Michiki, Okuma Electrical Engineering D, 118: 828-835, 1998-7 / 8 “Sensorless position / speed sensorless control of cylindrical brushless DC motor by disturbance observer and speed adaptive identification” Japanese Patent No. 2756049

  By the way, in the control devices described in Non-Patent Documents 2 to 4, the polarity of the current value Iuvw is determined by the current detection sensor 3, and the compensation voltage command value (V * uvw + ΔVuvw) is determined based on the determined polarity. By calculating, the output error due to the short-circuit prevention period of the output voltage of the inverter 1 is compensated.

  However, when the value of the output current of the inverter 1 is small and the peak value of the ripple is larger than the output current value, the polarity may not be accurately determined.

  Further, as in the control device described in Patent Document 1 described above, even if the polarity is determined by using a value offset from “0” as a threshold value, the value of the output current of the inverter 1 itself is small, and the ripple wave If the high value is large, the polarity may not be accurately determined.

  An object of the present invention is to provide an inverter control apparatus that compensates for an output voltage error of an inverter without determining the polarity of the output current of the inverter.

In order to achieve the above object, in the invention described in claim 1,
PWM output means (14) for outputting a PWM signal to the inverter (1) that converts DC power into three-phase power and outputs it to the three-phase electric motor (2);
Calculation means (10) for calculating a command value (V * dq) of the output voltage of the inverter based on the output current (iαβ) of the inverter and a command value (i *) of the output current of the inverter;
The command value of the output voltage is corrected and the corrected command value is output as the command value of the output voltage to the PWM output means, and the output voltage error due to the short-circuit prevention period of the output voltage of the inverter is calculated. In an inverter control device comprising compensation means (13a) for compensation,
The calculation means calculates a command value (V * dq) of the output voltage of the inverter in the rotating coordinate system,
The compensation means corrects the command value of the output voltage of the inverter based on the phase angle (θi) of the command value of the output current in the rotating coordinate system,
Conversion means (11, 12) for converting the command value (V * dq + ΔVdq) of the output voltage corrected by the compensation means from the rotating coordinate system to the three-phase fixed coordinate system and applying the converted value to the PWM output means is provided. An inverter control device characterized by being a thing.

  Therefore, the command value (V * dq + ΔVdq) of the corrected output voltage is coordinate-converted from the rotating coordinate system to the three-phase fixed coordinate system and applied to the PWM output means. Therefore, based on the PWM signal output from the PWM output means. When the inverter outputs three-phase power, it is possible to compensate for the output voltage error due to the short-circuit prevention period of the output voltage of the inverter.

  Here, as described above, since the command value (V * dq) of the output voltage of the inverter is corrected based on the phase angle (θi) of the command value of the output current, the polarity of the output current of the inverter is discriminated. The output voltage error due to the short-circuit prevention period of the output voltage of the inverter can be compensated.

  According to the first aspect of the present invention, the polarity of the output current of the inverter is not discriminated, and the detection accuracy of the sensor that detects the output current of the inverter can be made lower than that of the conventional one. In general, if the detection accuracy of the sensor is lowered, the physique of the sensor can be reduced.

  On the other hand, moving bodies such as vehicles, airplanes, and trains are limited in mounting space and are required to be compact in size. For this reason, as in the invention described in claim 3, the inverter control device described in claim 1 or 2 can reduce the size of the sensor as described above. It is effective.

  Specifically, the compensation means corrects the command value (V * dq) of the output voltage of the inverter as in the second aspect of the invention.

That is, in the inverter control device according to claim 1, the compensation means includes:
Setting a norm (ΔVdq) of a compensation voltage corresponding to an average value obtained by coordinate-converting the theoretical value of the output voltage error caused by the short-circuit prevention period from the three-phase fixed coordinate system to the rotating coordinate system and averaging the time,
A command value (V * dq) of the output voltage of the inverter is corrected according to a norm of the compensation voltage and a phase angle (θi) of the command value of the output current.

According to a fourth aspect of the present invention, the inverter control device according to any one of the first to third aspects and a three-phase electric motor are provided.
The three-phase electric motor has a structure cooled by the fluid.

  Here, in the conventional techniques described in Non-Patent Documents 2 to 4 described above, if the output current of the inverter becomes small, the polarity of the current cannot be accurately determined as described above, so that the output voltage error is compensated. The three-phase electric motor could not be controlled.

  On the other hand, according to the first aspect of the present invention, it is possible to compensate for the output voltage error due to the short-circuit prevention period of the output voltage of the inverter without determining the polarity of the output current of the inverter. Therefore, even when the three-phase electric motor rotates at a low rotation speed range and a low load, the output voltage error can be compensated.

  Therefore, since it is possible to control a three-phase electric motor even at a lower rotational speed range than before, abnormalities such as fluid shortage when rotating at a low rotational speed and low load (that is, even when the inverter output current is small). Can be detected, and secondary destruction such as seizure of the bearing can be prevented.

  In addition, the code | symbol in the bracket | parenthesis of each said means shows the correspondence with the specific means as described in embodiment mentioned later.

(First embodiment)
FIG. 1 shows a schematic configuration of a first embodiment of an inverter control device 100A according to the present invention. Note that the same parts as those of the above-described conventional technology are denoted by the same reference numerals, and description thereof is omitted.

  The control device 100A of this embodiment includes a current control unit 10, dq / αβ conversion units 11 and 18, αβ-uvw conversion unit 12, dead time compensation unit 13a, PWM output unit 14, and upper and lower arm short circuit prevention period generation unit. 15, uvw / αβ conversion means 16, and position estimation means 17.

  First, the current control means 10 calculates a command voltage value V * dq in the rotating coordinate system using the command current value i *, the detected current value iαβ, the estimated rotation position θ ^ re, and the estimated rotation speed ω ^ rm.

  Next, the dead time compensator 13a, based on the phase value θi of the command current value Idq in the rotating coordinate system and the norm ΔVdq of the compensating voltage value, compensates the compensation voltage value ΔVd in the rotating coordinate system (dq coordinate system), ΔVq is obtained.

  Specifically, as shown in FIG. 2, the component ΔVd on the d-axis of the compensation voltage value ΔVdq is ΔVdq · cos θi, and the component ΔVd on the q-axis of the compensation voltage value ΔVdq is ΔVdq · sin θi. .

  Here, the norm ΔVdq of the compensation voltage value is calculated by the dead time compensation means 13a as follows.

  That is, the theoretical value Vtd (see the figure) for the output voltage error caused by the short-circuit prevention period td is obtained from the base carrier frequency fc and the base output voltage Ve of the DC power supply Ba, and the coordinates are converted from the three-phase fixed coordinate system to the rotating coordinate system. Then, an average value obtained by time averaging is calculated in advance, and the average value is set as a base compensation voltage norm.

  Here, since the theoretical value Vtd is the carrier frequency fc × the short-circuit prevention period time td × the output voltage Ve of the DC power supply Ba (that is, the voltage value output from the DC power supply Ba to the inverter 1), the carrier frequency fc, the DC power supply When the output voltage Ve of Ba is different from the base carrier frequency fc and the base output voltage Ve of the DC power supply Ba, the base compensation voltage norm is multiplied by a proportional coefficient to obtain a compensation voltage norm ΔVdq.

  The compensation voltage value ΔVdq calculated as described above is a voltage compensation value having a polarity opposite to that of the output voltage error caused by the short-circuit prevention period td.

  Thereafter, the dead time compensation means 13a calculates the compensation voltage command value V * dq + ΔVdq by adding the compensation voltage value ΔVdq to the command voltage value V * dq in the rotating coordinate system.

  Here, since the output voltage error due to the short-circuit prevention period has the same phase and opposite polarity as the output current of the inverter 1, “the compensation voltage command value calculated using the phase angle θi of the command current value i * of the inverter 1. As V * dq + ΔVdq ”, the output voltage error (Vtd) due to the short-circuit prevention period can be compensated.

  Next, the dq / αβ conversion means 11 converts the compensation voltage command value V * dq + ΔVdq from the rotational coordinate system (dq) to the two-phase fixed coordinate system (α-β), and compensates in the two-phase fixed coordinate system. The voltage command value V * αβ + ΔVαβ is calculated.

  Next, the αβ-uvw conversion means 12 converts the compensation voltage command value V * αβ + ΔVαβ from the two-phase fixed coordinate system (α-β) to the three-phase fixed coordinate system (uvv-w), thereby fixing the three-phase fixed value. A compensation voltage command value V * uvw + ΔVuvw in the coordinate system is calculated.

  Next, the PWM output unit 14 compares the compensation voltage command value V * uvw + ΔVuvw with the carrier wave and calculates a PWM signal for each phase. Then, the upper and lower arm short circuit prevention period generation means 15 sets a short circuit prevention period for each PWM signal and outputs the set PWM signal to the inverter 1.

  Then, since the inverter 1 converts the DC power output from the DC power supply Ba into three-phase AC power, the three-phase electric motor 2 rotates based on the three-phase AC power output from the inverter 1.

  In the present embodiment, as the three-phase electric motor 2, for example, a sensorless motor that does not use a position detection sensor that detects the absolute position of the rotor, and a permanent magnet type synchronous motor is used. Yes. The three-phase electric motor 2 is used to drive a compressor for an in-vehicle air conditioner.

  On the other hand, when the current detection sensor 3 detects the output current Iuvw output from the inverter 1, the current detection sensor I3 converts the detected current value Iuvw from the three-phase fixed coordinate system to the two-phase fixed coordinate system by the uvw / αβ conversion means 16. A detected current value iαβ is calculated.

  Further, the dq / αβ conversion means 18 calculates the command voltage value V * αβ by converting the command voltage value V * dq from the rotating coordinate system to the two-phase fixed coordinate system.

  Then, the position estimating means 17 estimates the estimated rotational position θ ^ re and the estimated rotational speed ω ^ rm on the basis of the detected current value iαβ and the command voltage value V * αβ in the two-phase fixed coordinate system. 10 is given.

When the processing of each of the means 10, 11... 18 is repeated, the three-phase AC power is output from the inverter 1 to the three-phase electric motor 2, and the three-phase electric motor 2 rotates to be used for the on-vehicle air conditioner. The compressor 100 is driven. As the control device 100A, a microcomputer, a digital signal processor or the like is used, and various means 10 to 18 such as a current control means are configured by a computer program.

  Next, the effect of this embodiment is demonstrated.

  That is, the control device 100A of the present embodiment includes a PWM output means 14 that outputs a PWM signal to the inverter 1 that converts DC power into three-phase power and outputs the three-phase power to the three-phase electric motor 2, and detection of the inverter 1 Based on the current value iαβ and the command current value i *, the current control means 10 for calculating the command value V * dq of the output voltage of the inverter 1 in the rotation coordinate system, and the rotation coordinate based on the phase value θi of the command current value Idq. Dead time compensation means 13a for correcting the command output voltage V * dq in the system (dq coordinate system) to obtain the compensation voltage value V * dq + ΔVdq, and the compensation voltage value V * dq + ΔVdq from the rotating coordinate system to the three-phase fixed coordinate system Coordinate conversion means 11 and 12 which convert the coordinates into the inverter 1 and provide them to the inverter 1.

  Therefore, since the compensation voltage value V * dq + ΔVdq is coordinate-transformed from the rotating coordinate system to the three-phase fixed coordinate system and applied to the inverter 1, the output voltage error Vtd caused by the short-circuit prevention period of the output voltage of the inverter 1 is compensated Can do.

  Here, as described above, the command value (V * dq) of the output voltage of the inverter is corrected based on the phase angle (θi) of the command value (i *) of the output current. The output voltage error Vtd caused by the short-circuit prevention period of the output voltage of the inverter 1 can be compensated without discriminating the polarity.

  Further, in the control device 100 of the prior art described in Patent Document 1, the command output voltage is corrected in the three-phase fixed coordinate system to obtain the compensation voltage value. For this reason, the compensation voltage command value input to the PWM output unit 14 is a rectangular wave signal waveform having a large amplitude change, as indicated by a symbol y in FIG.

  On the other hand, in the control device of the present embodiment, the command output voltage is corrected in the rotating coordinate system to obtain a compensation voltage value, and the compensation voltage is coordinate-converted from the rotating coordinate system to the three-phase fixed coordinate system to the PWM output unit 14. Output.

  Here, the compensation voltage command value input to the PWM output means 14 is a sinusoidal signal waveform having a smooth amplitude change, as indicated by a symbol X in FIG. When the PWM output means 14 calculates the PWM signal using the compensation voltage command value of the signal waveform having a smooth amplitude change as described above, the change in the pulse width of the PWM signal becomes smoother than that in the prior art.

  When such a PWM signal is used to control the inverter 1 to generate three-phase power and apply it to the three-phase electric motor 2, even if the three-phase electric motor 2 is an electric motor with a small time constant, Pulsation hardly occurs.

  Further, according to the present embodiment, the polarity of the output current of the inverter is not discriminated, and the detection accuracy of the sensor that detects the output current of the inverter can be made lower than before. In general, if the detection accuracy of the sensor is lowered, the physique of the sensor can be reduced.

  On the other hand, moving bodies such as vehicles, airplanes, and trains are limited in mounting space and are required to be compact in size. On the other hand, as described above, in the present embodiment, it is possible to use a sensor with low sensor detection accuracy, so that the size of the sensor itself can be reduced. Therefore, it is effective in reducing the size of the physique when the inverter control device 100A is mounted on the vehicle.

  Further, in the conventional control devices described in Non-Patent Documents 2 to 4, when the output current of the inverter 1 becomes small, the polarity of the output current cannot be determined, and the output voltage error (Vtd) can be compensated. The three-phase electric motor 2 could not be controlled.

  On the other hand, the three-phase electric motor 2 is incorporated into the on-vehicle air conditioner electric compressor as described above, and compresses the compressor to apply pressure to the refrigerant. The three-phase electric motor 2 is cooled by a refrigerant (fluid) flowing therethrough.

  Here, according to the present embodiment, since the output voltage error (Vtd) of the inverter 1 is compensated without determining the polarity of the output current of the inverter 1, even if the output current of the inverter 1 is small, The three-phase electric motor 2 can be controlled by compensating for the output voltage error (Vtd).

  Here, even if the output current of the inverter 1 is small, stable control can be performed, so that load abnormality such as lack of refrigerant (fluid) can be detected accurately, and damage to sliding parts such as rotor bearings and compression mechanisms can occur in advance. Can be prevented.

(Second Embodiment)
In the control device 100A of the first embodiment described above, the position estimation unit 17 has been described with respect to the example in which the estimated rotational position θ ^ re and the estimated rotational speed ω ^ rm are estimated in the two-phase fixed coordinate system. Instead, in the second embodiment, the control device 100A may be configured as shown in FIG.

  That is, in this embodiment, the αβ / dq conversion means 18 is connected to the subsequent stage of the uvw / αβ conversion means 16. When the uvw / αβ conversion means 16 performs coordinate conversion of the detected current value Iuvw from the three-phase fixed coordinate system to the two-phase fixed coordinate system to calculate the detected current value iαβ, the αβ / dq conversion means 18 calculates the detected current value iαβ. The detected current value Idq is calculated by performing coordinate conversion from the two-phase fixed coordinate system to the rotating coordinate system.

  Then, the position estimation means 17 estimates the estimated rotational position θ ^ re and the estimated rotational speed ω ^ rm based on the detected current value Idq and the command voltage value V * dq calculated by the current control means 10. Other configurations are the same as those in the first embodiment.

(Third embodiment)
In the control device 100A of the second embodiment described above, the uvw / αβ conversion unit 16 performs coordinate conversion of the detected current value Iuvw from the three-phase fixed coordinate system to the two-phase fixed coordinate system, and then the αβ / dq conversion unit 18 detects the detected current value Iuvw. Although the example in which the current value iαβ is coordinate-transformed from the two-phase fixed coordinate system to the rotating coordinate system has been described, instead of this, in the third embodiment, as shown in FIG. 5, the uvw / αβ conversion means 16 and the αβ Instead of the / dq conversion means 18, uvw / dq conversion means 19a may be adopted. In this case, the uvw / dq conversion means 19a directly converts the detected current value Iuvw from the three-phase fixed coordinate system to the rotating coordinate system to calculate the detected current value Idq.

  Further, the compensation voltage value V * dq + ΔVdq calculated by the dead time compensation means 13a is directly converted by the dq / uvw conversion means 19b from the rotating coordinate system to the three-phase fixed coordinate system to obtain the compensation voltage value V * uvw + ΔVuvw. calculate. Other configurations are the same as those in the first embodiment.

(Fourth embodiment)
In the fourth embodiment, the control device 100A may be configured to correct the command output voltage as if in the two-phase fixed coordinate system. The configuration of the control device 100A in this case is shown in FIG.

  First, the dead time compensation means 13a stores in advance a plurality of candidate values of the compensation voltage value ΔVαβ in the memory, and the dead time compensation means 13a is based on the estimated rotational position θ ^ re among the plurality of candidate values. A compensation voltage value ΔVαβ is selected.

  On the other hand, the dq / αβ converter 11 converts the command voltage value V * dq from the rotating coordinate system to the fixed coordinate system to calculate the compensation voltage value V * αβ. Accordingly, the compensation voltage command value V * αβ + ΔVαβ is calculated by adding the compensation voltage value ΔVαβ and the compensation voltage value V * αβ by the adding means 18b.

  Then, the compensation voltage command value V * αβ + ΔVαβ is converted by the αβ-uvw conversion means 12 from the two-phase fixed coordinate system (α-β) to the three-phase fixed coordinate system (uvv), and the three-phase fixed coordinates are converted. A compensation voltage command value V * uvw + ΔVuvw in the system is calculated. Other configurations are the same as those in the first embodiment.

(Other embodiments)
In implementing the present invention, not only the case where a three-phase modulation method for switching all three phases as a PWM signal is adopted, but also a two-phase modulation method for setting a pause period Tw (see FIG. 7) for only one of the three phases. In addition, an overmodulation method in which a pause period Tw is set for another phase may be employed.

  Here, when the phase and phase interval of the pause period Tw change depending on the rotation speed, load, etc., the compensation voltage norm ΔVdq is changed according to the partial integration value of the pause period.

  FIG. 7 shows the compensation voltage command value output from the inverter 1 in the two-phase modulation method. The one-phase compensation voltage command value changes stepwise, but the interphase voltage has a smooth sinusoidal signal waveform.

  In the above-described embodiment, an example in which a permanent magnet type synchronous motor is used as the three phase electric motor 2 has been described. However, the present invention is not limited thereto, and various three phase electric motors other than the permanent magnet type synchronous motor may be used. .

  In the above-described embodiment, as the three-phase electric motor 2, an example using a sensorless motor that does not use a position detection sensor for detecting the absolute position of the rotor has been described. You may use the motor with a sensor to which is added.

  In carrying out the present invention, the three-phase electric motor 2 may be rotated regardless of whether the torque is positive or negative.

  Hereinafter, the correspondence relationship between the above-described embodiment and the configuration within the scope of the claims will be described. The current control means 10 corresponds to a calculation means, the dead time compensation means 13a corresponds to a compensation means, and the dq / αβ conversion means 11 , Αβ-uvw conversion means 12 corresponds to the conversion means, and the electric compressor corresponds to the fluid machine.

1 is a block diagram showing a first embodiment of an inverter control device according to the present invention. FIG. It is a figure for demonstrating the electric current phase in FIG. It is a figure which shows the voltage waveform of the compensation voltage command value in FIG. It is a block diagram which shows 2nd Embodiment of the control apparatus of the inverter which concerns on this invention. It is a block diagram which shows 3rd Embodiment of the control apparatus of the inverter which concerns on this invention. It is a block diagram which shows 4th Embodiment of the control apparatus of the inverter which concerns on this invention. It is a figure which shows the output waveform of the inverter which concerns on the modification which concerns on this invention. It is a figure which shows the structure of the conventional inverter. It is a figure for demonstrating the problem of the output voltage of the conventional inverter. It is a figure for demonstrating the problem of the output voltage of the conventional inverter. It is a figure which shows the structure of the control apparatus of the conventional inverter.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 ... Inverter, 100a ... Control apparatus, 2 ... Three-phase electric motor,
14 ... PWM output means, 10 ... current control means,
11 ... dq / αβ conversion means, 12 ... αβ-uvw conversion means, θi ... phase value,

Claims (4)

PWM output means (14) for outputting a PWM signal to the inverter (1) that converts DC power into three-phase power and outputs it to the three-phase electric motor (2);
Calculation means (10) for calculating a command value (V * dq) of the output voltage of the inverter based on the output current (iαβ) of the inverter and a command value (i *) of the output current of the inverter;
The command value of the output voltage is corrected and the corrected command value is output as the command value of the output voltage to the PWM output means, and the output voltage error due to the short-circuit prevention period of the output voltage of the inverter is calculated. In an inverter control device comprising compensation means (13a) for compensation,
The calculation means calculates a command value (V * dq) of the output voltage of the inverter in the rotating coordinate system,
The compensation means corrects the command value of the output voltage of the inverter based on the phase angle (θi) of the command value of the output current in the rotating coordinate system,
Conversion means (11, 12) for converting the command value (V * dq + ΔVdq) of the output voltage corrected by the compensation means from the rotating coordinate system to the three-phase fixed coordinate system and applying the converted value to the PWM output means is provided. An inverter control device characterized by being a thing. The compensation means includes
Setting a norm (ΔVdq) of a compensation voltage corresponding to an average value obtained by coordinate-converting the theoretical value of the output voltage error caused by the short-circuit prevention period from the three-phase fixed coordinate system to the rotating coordinate system and averaging the time,
The inverter output voltage command value (V * dq) is corrected according to a norm of the compensation voltage and a phase angle (θi) of the output current command value. Control device. A control device for an inverter mounted on a mobile unit, comprising the control device for an inverter according to claim 1. The inverter control device according to claim 1 and a three-phase electric motor are provided,
The three-phase electric motor has a structure in which a fluid flows through and is cooled.

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