JP2011211818A - Power conversion equipment, method of converting power, and motor drive system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide an asynchronous PWM control method for preventing an increase in a low-order harmonic and a beat phenomenon from occurring even at a low carrier frequency, and hence to achieve a compact and efficient power converter system without deteriorating control performance.SOLUTION: A fundamental wave component of a PWM pulse is compared with a voltage command value, and the voltage command value is guaranteed based on the comparison result, thus suppressing a beat. More specifically, a motor drive having a small torque ripple owing to the beat can be realized by the power converter system including a function of extracting the fundamental wave component of the PWM pulse, a function of comparing the extracted fundamental wave component with the voltage command value, and a function of correcting the voltage command value of the PWM according to the comparison result.

Description

  The present invention relates to a power conversion device, a power conversion method, and a motor drive system.

  The power converter turns on and off the semiconductor switching element based on the voltage command value to obtain an output with a desired frequency. The so-called PWM control is known as the control for determining the on / off timing of the semiconductor switching element. In the PWM control, typically, a voltage command value is compared with a triangular wave carrier to create a PWM pulse. At present, many of them are digitized, calculate a pulse width corresponding to a command value for every fixed period (corresponding to the period of a carrier wave), and perform PWM control by turning on / off the bits. For example, it is known to “Semiconductor power conversion circuit”, Institute of Electrical Engineers, 6.3.2 Asynchronous PWM inverter) (Non-patent Document 1).

  It has been known that problems such as the occurrence of a beat phenomenon occur and torque pulsation occurs when the motor is driven in this state. If torque pulsation occurs, it may cause shaft vibration. In particular, shaft vibration is a problem in systems that drive compressors and the like.

  For torque pulsation, if the carrier frequency is set sufficiently higher than the frequency of the command value, these problems will be reduced, but other problems will occur such as increased switching loss and degraded efficiency. In particular, in a large-capacity power converter, there is a limit to semiconductor elements to be used, and the carrier frequency cannot be physically increased.

Japanese Patent Laid-Open No. 08-251930

“Semiconductor Power Conversion Circuit” The Institute of Electrical Engineers of Japan, 6.3.2 Asynchronous PWM Inverter)

  In creating a PWM pulse, the voltage command value is compared with the carrier. When the voltage command value is close to the frequency of the carrier signal, the command value is within the cycle of the carrier signal. In view of the fact that it is caused by a large change, a technique is known in which an average value of voltage command values is estimated and the output pulse width is controlled accordingly in order to suppress the occurrence of a beat phenomenon. Such a technique is described in, for example, Japanese Patent Application Laid-Open No. 08-251930 (Patent Document 1).

  However, in this prior art, since estimation is used, deviation from actual control is likely to occur, and it has not always been possible to sufficiently suppress an increase in low-order harmonics or a beat phenomenon.

  So-called synchronous PWM control is known in which the carrier frequency and the number of pulses (ratio of the carrier frequency and the command value frequency) are switched depending on the command value frequency. Although the carrier frequency and the number of pulses can be selected so as to suppress the beat phenomenon, the control circuit for switching the carrier frequency and the number of pulses becomes complicated, and pulsation (switching shock) accompanying the switching of the number of pulses occurs. However, it is actually difficult to apply in a system in which the frequency of the command value changes.

  An object of the present invention is to provide a power conversion device, a power conversion method, and a motor drive system capable of suppressing an increase in low-order harmonics or a beat phenomenon even with a low carrier frequency with a relatively simple configuration. is there.

  In the present invention, in order to achieve the above object, voltage command information of a predetermined frequency is obtained based on the knowledge that a component that causes an increase in a low-order harmonic or a beat phenomenon is superimposed when a power command value is PWM-converted. Is generated based on carrier wave information to generate PMW information for turning on and off the semiconductor switching element, and the component of the predetermined frequency is extracted from the PMW information and compared with the voltage command information. A frequency component different from the predetermined one that is superimposed when voltage command information is converted into the PMW information is detected, and PWM information in which the detected component is suppressed is generated.

  According to the present invention, an increase in low-order harmonics or a beat phenomenon can be suppressed even with a low carrier frequency with a relatively simple configuration.

The circuit diagram which shows the 1st-4th embodiment of this invention. Explanatory drawing of PWM conversion. The control block diagram which shows a part of 1st Embodiment of this invention. The control block diagram which shows a part of 1st Embodiment of this invention. The control block diagram which shows a part of 1st Embodiment of this invention. Explanatory drawing of a beat component. The control block diagram which shows a part of 2nd Embodiment of this invention. The control block diagram which shows a part of 2nd Embodiment of this invention. The control block diagram which shows a part of 2nd Embodiment of this invention. The control block diagram which shows a part of 3rd Embodiment of this invention. The control block diagram which shows a part of 3rd Embodiment of this invention. The control block diagram which shows a part of 4th Embodiment of this invention. The control block diagram which shows a part of 4th Embodiment of this invention. The control block diagram which shows a part of 4th Embodiment of this invention. The control block diagram which shows a part of 4th Embodiment of this invention. The effect of 1st Embodiment of this invention is shown.

  Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings.

  A first embodiment will be described with reference to FIG. First, the configuration of the power conversion system 10 of the first embodiment will be described. The power conversion system of the first embodiment includes an AC / DC converter and a DC / AC inverter. In this embodiment, as shown in FIG. 1, the AC / DC converter is configured as a diode rectifier 2, and the DC / AC inverter is configured as an IGBT inverter 1. Of course, the AC / DC converter may be formed of an IGBT element in addition to the diode rectifier 2. The IGBT inverter 1 is connected to the motor 4, and the IGBT inverter 1 is connected to the three-phase AC 3 via the diode rectifier 2 in order to establish the DC voltage of the DC capacitor 12 of the IGBT inverter 1.

  In the following description, it is assumed that the capacity of the DC capacitor 12 is sufficiently large and the DC capacitor voltage can be regarded as substantially constant. However, the same effect can be obtained by adding a control block for correcting the voltage command value of the IGBT inverter with the DC voltage.

  First, the operation of the power conversion system of the first embodiment will be described. The DC capacitor 12 of the IGBT inverter 1 is charged by the three-phase AC 3 and the diode rectifier 2 as described above, and a DC voltage is established. Each gate of each IGBT 101 i of the IGBT inverter 1 is controlled by an output signal of the PWM controller 150, outputs an AC voltage by PWM modulating the DC voltage, and drives the motor 4.

  When the induction motor is controlled at a variable speed by vector control, the slip frequency ωs is added to the speed command value ωr * to obtain the inverter frequency command value ω1 *, and based on this, the D-axis voltage command value (Vd *) , Q-axis voltage command value (Vq *) and phase θ are calculated, and the AC voltage supplied to the induction motor is vector-controlled to vary the speed. The command value to be PWM modulated is a periodic function such as a sine wave.

  The D-axis voltage command value (Vd *) and the Q-axis voltage command value (Vq *) are subjected to 2-phase / 3-phase coordinate conversion by the phase θ corresponding to the frequency command value ω1 * to have a frequency corresponding to the frequency command value. Three-phase AC voltage command values Vu *, Vv *, and Vw * are generated, and each of them is compared with a carrier wave, and variable speed control is performed by turning on and off the switching elements. In normal vector control, a component parallel to the magnetic flux of the motor is defined as the D axis, and a component perpendicular to the magnetic flux is defined as the Q axis.

  When a three-phase voltage command 750 in FIG. 3 to be described later in detail is output as a command value indicated by reference numeral 760 without being corrected, the switching element based on the three-phase AC voltage command values Vu *, Vv *, and Vw * is output. The control operation will be described.

  Although FIG. 2 shows only one phase of Vu *, Vv *, and Vw *, the other phases operate in the same manner. When the AC voltage command value (V *) (meaning one phase represented by Vu *, Vv *, Vw *) becomes larger than the carrier wave Cr31, a positive on signal is output. The negative-side IGBT operates complementarily to the positive-side IGBT. For example, taking the U phase as an example, the upper IGBT 101iP described above is turned on by the on signal described below, and the positive side voltage of the DC capacitor 12 is applied to the U phase terminal of the motor 4 as shown in FIG. When the AC voltage command (V *) 760 becomes smaller than the carrier wave Cr31, an off signal is output to the IGBT 101iP. Negative side IGBT 101 iN becomes conductive, and the negative side voltage of DC capacitor 12 is applied to the U-phase terminal of motor 4.

  Next, the configuration of the PWM controller 150 is shown in FIG. The PWM controller 150 includes a modulation wave calculation block 901 and a PWM calculation block 902. The modulation wave calculation block 901 corrects the voltage command value for PWM control, and the PWM calculation block 902 performs triangular wave comparison PWM control based on the voltage command correction value.

  The modulation wave calculation block 901 includes a voltage command generation block 801, a compensation waveform calculation block 802, a compensation block 803, and the like.

  In FIG. 2, the voltage command generation block 801 includes a D-axis voltage command value of the D-axis voltage command block 601, a Q-axis voltage command value of the Q-axis voltage command block 602, a 2-phase / 3-phase conversion block 603, and the like. . The D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602 are calculated values of a current control block not shown in the figure.

  In the voltage command generation block 801, a sine wave 791 and a cosine wave 792 are added to the D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602, respectively, so-called DQ / After αβ conversion, the two-phase / three-phase conversion block 603 converts it to three-phase alternating current, and outputs a three-phase voltage command 750.

  Next, the compensation waveform calculation block 802 that is the point of the present invention will be described. In FIG. 2, the compensation waveform calculation block 802 includes a triangular wave comparison PWM block 701, a moving average block 702, and the like.

  In the compensation waveform calculation block 802, the fundamental wave component waveform of the PWM pulse is extracted for each phase using DFT. First, based on the three-phase voltage command 750 that is the output of the two-phase / three-phase conversion block 603, the triangular wave comparison PWM block 701 creates a PWM pulse for each phase. Next, the PWM pulse, the sine wave 791 of the fundamental frequency, and the cosine wave 792 are integrated for each phase, and the moving average of the period of the fundamental frequency is calculated in the moving average block 702, respectively. Next, for each phase, the sine wave 791 and the cosine wave 792 are added to each moving average calculation value, and then the calculation value 713 is calculated by adding the integration value 711 and the integration value 712. As shown in FIG. 6, when a PWM pulse is converted, it should be the same as the power conversion command 760, but the beat signal is transformed into a superposition in the process of PWM conversion. The calculated value 713 indicates the fundamental wave component waveform of PWM. The calculation value 713 is a Fourier transform of the PWM conversion of the D-axis voltage command value of the D-axis voltage command block 601 and the PWM conversion of the Q-axis voltage command value of the Q-axis voltage command block 602. In this embodiment, so-called DFT is used, but other transformations may be used as long as the fundamental wave component can be extracted. That is, the difference between the calculated value 713 and the three-phase voltage command 750 corresponds to the PWM fundamental wave component output error, and causes a beat. In some cases, it may be necessary to remove other harmonics other than the fundamental wave component waveform. In that case, the same applies to the following embodiments, but the extracted component of the DFT can be changed to remove the component.

  The compensation block 803 modifies the three-phase voltage command 750 so as to correct the error. Specifically, the difference between the three-phase voltage command 750 and the calculation value 713 is calculated to calculate a difference calculation value 755, and the difference calculation value 755 is further added to the three-phase voltage command 750 to correct the voltage command. The value 760 is calculated. By converting the voltage command correction value 760 into a PWM pulse by the PWM calculation block 902, a PWM pulse with few beats can be created. By controlling the IGBT 101 using the PWM pulse, the motor can be smoothly turned.

  The beat component with respect to the fundamental component of the voltage command correction value 760 is added with a beat component having a reverse polarity so as to suppress the beat component of the PWM pulse. It is larger than the beat for the fundamental wave component of the PWM pulse.

  Since the triangular wave comparison PWM does not have a linear relationship between the voltage command value and the PWM pulse, as shown in FIG. 4, in order to correct the compensation amount, a difference calculation value 755 between the three-phase voltage command 750 and the calculation value 713 is further added. If the compensation calculation block 758 is added and the compensation calculation is performed, the beat suppression effect can be further increased. Specifically, in the correction calculation block, it is preferable to integrate and output a real number of 1 or less with respect to the input.

  FIG. 16 shows the effect of this embodiment when the correction calculation block 758 outputs a value obtained by multiplying the input value by 0.8. The IGBT inverter 1 is a neutral point clamp three-level inverter. In each waveform, 997 indicates the fundamental wave component amplitude of the inverter output when this embodiment is not applied, and 998 indicates the output of the compensation calculation block 758, that is, the fundamental wave component amplitude of the compensation amount of the voltage command value. Reference numeral 999 denotes the fundamental wave component amplitude of the inverter output when PWM is performed using the voltage command correction value 760. The fundamental wave component amplitude of the inverter output is preferably close to a constant value.

  Compared to beat 997 of the fundamental wave component amplitude of the inverter when this embodiment is not applied, the beat of the fundamental wave component amplitude of the inverter output when PWM is performed using the voltage command correction value 760 has less beats, and the effect of this embodiment is achieved. I know that there is.

  In FIG. 3 and FIG. 4, the voltage command value of the three-phase voltage command 750 is corrected by the sum / difference calculation of the voltage command. However, as shown in FIG. However, since zero division cannot be calculated, a zero division compensation block 718 needs to be added.

  The modulation wave calculation block 901 is preferably constituted by a CPU, the PWM calculation block 902 is constituted by an FPGA, and the PWM calculation block 902 is preferably calculated at a calculation cycle earlier than that of the modulation wave calculation block 901.

  Further, the triangular wave comparison PWM block 701, the compensation waveform computation block 802, and the compensation block 803 may be configured with an FPGA having an early computation cycle.

  In the first embodiment, the three-phase voltage command 750 is corrected. In the second embodiment, the DQ-axis voltage command value is corrected and then converted into a three-phase AC voltage correction value. . The configuration of the PWM controller 150 of the second embodiment is shown in FIG.

  In this embodiment, the PWM controller 150 includes a modulation wave calculation block 901 and a PWM calculation block 902. The modulation wave calculation block 901 includes a voltage command generation block 801, a compensation waveform calculation block 802, a compensation block 803, and the like.

  The voltage command generation block 801 includes a D-axis voltage command block 601, a Q-axis voltage command block 602, a 2-phase / 3-phase conversion block 603, and the like. The D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602 are calculation values such as current control not shown in the figure. The D-axis voltage command value in the D-axis voltage command block 601 and the Q-axis voltage command value in the Q-axis voltage command block 602 are each converted to a three-phase AC value in the 2-phase / 3-phase conversion block 603 after so-called DQ / αβ conversion. The three-phase voltage command 750 is calculated by conversion.

  Next, the compensation waveform calculation block 802 that is a point will be described. In the second embodiment, the compensation waveform calculation is performed using different sine waves 791U to W and cosine waves 792U to W for each phase. That is, the modulation wave calculation block 901 of the power conversion system 1 of the present embodiment includes three compensation waveform calculation blocks 802U to 802W that are the same as the number of phases. The compensation calculation blocks 802U to 802W for each phase differ only in the phases of the sine waves 791U to W and cosine waves 792U to W to be integrated, and the phases of the sine waves and cosine waves differ by 120 degrees.

  As described above, the compensation waveform calculation block 802 in FIG. 7 extracts the fundamental wave component waveform of the PWM pulse using a sine wave and a cosine wave that are different for each phase in the DFT.

  First, based on the three-phase voltage command 750 that is the output of the two-phase / three-phase conversion block 603, the triangular wave comparison PWM block 701 creates a PWM pulse for each phase. Next, for each phase, the PWM pulse, the sine waves 791U to W of the fundamental frequency, and the cosine waves 792U to W are integrated, and the moving average of the period of the fundamental frequency is calculated by the moving average block 702, respectively. . The U phase uses a sine wave generation block 791U and a cosine wave generation block 792U, the V phase uses a sine wave generation block 791V and a cosine wave generation block 792V, and the W phase uses a sine wave generation block 791W and a cosine wave generation block 792W. The fundamental wave amplitude components 711U to W and 712U to W are extracted.

  Next, for each phase, the calculated value of the moving average, that is, the fundamental wave amplitude components 711U to W and 712U to W, the integrated value, the D axis voltage command value of the D axis voltage command block 601 and the Q axis voltage command block 602 are used. The calculated values 756U to W and 757U to W are calculated by calculating the difference from the Q-axis voltage command value. The calculated values 756U to W and 757U to W correspond to beat component compensation amounts. In the second embodiment, the compensation amount of the beat component can be extracted by comparing the D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602, which are DC amounts. It is.

  The D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602 are added to the calculated values 756U to W and 757U to W as shown in FIG. The corrected D-axis voltage command values 766U to W and the corrected Q-axis voltage command values 767U to W are calculated.

  As shown in FIG. 7, a sine wave 791U to W and a cosine wave 792U to W are integrated for each phase to the corrected D-axis voltage command value 766U to W and the corrected Q-axis voltage command value 767U to W to obtain a three-phase Voltage command correction values 760U to 760W that are alternating current are calculated.

  By converting the voltage command correction value 760 into a PWM pulse by the PWM controller 150, a PWM pulse with few beats can be created. By controlling the IGBT 101 using the PWM pulse, the motor can be smoothly turned.

  The beat component for the fundamental component of voltage command correction value 760 is smaller than the beat for the fundamental component of the PWM pulse.

  Since the triangular wave comparison PWM does not have a linear relationship between the voltage command value and the PWM pulse, if the compensation calculation is further performed to the difference calculation value 755 between the three-phase voltage command 750 and the calculation value 713 in order to correct the compensation amount, the beat is further increased. The suppression effect can be increased. In FIG. 8, the voltage command correction value 760 is calculated by adding the correction value to the difference calculation value 755. The correction value is preferably 1 or less.

  7 and 8, the correction amount is calculated from the difference between the fundamental wave amplitude component, the D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602. As shown in FIG. 9, the correction amount may be calculated using the ratio of the fundamental wave amplitude component to the D-axis voltage command value of the D-axis voltage command block 601 and the Q-axis voltage command value of the Q-axis voltage command block 602. .

  The third embodiment is characterized in that the voltage command value is a voltage command value based on the predictive PWM.

  The configuration of the third embodiment is different from the configuration of the first embodiment only in the block configuration of the voltage command generation block 801.

  In the first embodiment, the original three-phase voltage command 750 is a sine wave, whereas in the third embodiment, the three-phase voltage command 750 is a prediction PWM voltage command Y. The prediction PWM block 995 calculates the prediction PWM voltage command value so that the sine wave voltage command value and the prediction PWM voltage command value are the same within the control cycle. Since the PWM based on the voltage command of the prediction PWM has a large beat suppression effect, the beat suppression effect can be further increased by combining the compensation waveform calculation block 802 and the compensation block 803 of FIGS. 10 to 12 of the third embodiment. it can.

The fourth embodiment is characterized in that the voltage command value is a voltage command value based on the predictive PWM.
The configuration of the fourth embodiment is different from the configuration of the first embodiment only in the block configuration of the voltage command generation block 801.

  In the first embodiment, the original three-phase voltage command 750 is a sine wave, whereas in the third embodiment, the three-phase voltage command 750 is a prediction PWM voltage command. The prediction PWM block 995 calculates the prediction PWM voltage command value so that the sine wave voltage command value and the prediction PWM voltage command value are the same within the control cycle. Since the PWM based on the voltage command of the prediction PWM has a large beat suppression effect, the beat suppression effect can be further increased by combining the compensation waveform calculation block 802 and the compensation block 803 of FIGS. 10 to 12 of the third embodiment. it can.

DESCRIPTION OF SYMBOLS 1 IGBT inverter 2 Diode rectifier 3 Three-phase alternating current 10 Power conversion system 12 DC capacitor 150 PWM controller 601 D-axis voltage command block 602 Q-axis voltage command block 603 2-phase / 3-phase conversion block 701 Triangular wave comparison PWM block 702 Moving average block 711 Integrated value 713, 756U to W, 757U to W Calculated value 750 Three-phase voltage command 750Y Three-phase prediction PWM voltage command 766U to W Modified D-axis voltage command value 767U to W Modified Q-axis voltage command value 718 Zero percent compensation block 755 Difference calculation value 760 Voltage command correction value 791, 791U to W Sine wave 792, 792U to W Cosine wave 801 Voltage command generation block 802 Compensation waveform calculation block 803 Compensation block 901 Modulation wave calculation block 902 PWM calculation block 995 Prediction PWM block

Claims (16)

  In a power converter having a PMW information generating means for generating PMW information for turning on / off a semiconductor switching element by converting voltage command information of a predetermined frequency based on carrier wave information, the power conversion device has the predetermined frequency of the PMW information. Component detection means for detecting a frequency component different from the predetermined superimposed when the voltage command information is converted into the PMW information by extracting a component and comparing with the voltage command information, and the detected component A power conversion device comprising correction PWM information generation means for generating suppressed PWM information.   2. The suppressed component according to claim 1, wherein the suppressed component is extracted as a beat component superimposed on a fundamental wave, and has voltage command correcting means for correcting the voltage command value based on the beat component. A power converter.   The power converter according to claim 2, wherein the beat component is extracted using a DFT, and the voltage command value is corrected based on the beat component.   3. The fundamental wave component of the PWM information is extracted by DFT, a beat component is extracted from a comparison between the voltage command value and the extracted fundamental wave component, and the voltage command value is calculated based on the beat component. The power converter characterized by correcting.   5. The power converter according to claim 4, wherein a positive real number less than or equal to 1 is added to the beat component, and then added to a voltage command value for PWM control.   5. The PWM control voltage command value is corrected by dividing the extracted PWM fundamental wave component by a voltage command value and dividing the voltage command value by a calculated value. Power converter.   2. A fundamental wave component amplitude of the PWM information is extracted by DFT using a sine wave and a cosine wave having different phases for each phase, and the voltage command value on the DQ axis and the fundamental wave component of the extracted PWM information are extracted. A power converter that extracts a beat component from a comparison with an amplitude and corrects the voltage command value based on the beat component.   8. The beat command is extracted from the difference between the voltage command value on the DQ axis and the fundamental wave component amplitude of the extracted PWM information, and the voltage command value for PWM control is corrected based on the beat component. The power converter characterized by the above-mentioned.   7. The voltage command according to claim 6, wherein a beat component is extracted from a difference between the voltage command value on the DQ axis and the fundamental component amplitude of the extracted PWM information, and added to the voltage command value on the DQ axis. The power converter characterized by correcting a value.   2. The fundamental wave component amplitude of the PWM information is extracted by DFT with a sine wave and a cosine wave having different phases for each phase, and the voltage command value on the DQ axis and the fundamental wave of the extracted PWM information are extracted. A beat component is extracted from a difference from the component amplitude, and after adding a positive real number of 1 or less to the beat component, the voltage command value is corrected by adding to the voltage command value on the DQ axis. A power converter.   2. The fundamental wave component amplitude of the PWM information is extracted by DFT with a sine wave and a cosine wave having different phases for each phase, and the extracted fundamental wave component amplitude is divided by the voltage command value on the DQ axis. Then, the voltage command value on the DQ axis is divided by the divided operation value to correct the voltage command value.   2. The power conversion device according to claim 1, wherein a beat component of the AC voltage output from the power conversion system is smaller than a beat component of the voltage command value.   PMW information generating means for generating PMW information by comparing voltage command information of a predetermined frequency with carrier wave information, and a power conversion system for turning on / off a semiconductor switching element based on the PMW information. Component detection means for detecting a frequency component different from the predetermined superimposed when the voltage command information is converted into the PMW information by extracting a component and comparing with the voltage command information, and the detected component An electric motor drive system comprising correction PWM generation means for generating PWM information to be suppressed, and driving an AC electric motor with electric power supplied via the switching element.   14. The electric motor drive system according to claim 13, wherein the AC electric motor drives a compressor.   15. The motor drive system according to claim 14, wherein the AC motor drives a rolling mill.   By converting voltage command information of a predetermined frequency based on carrier wave information, PMW information for turning on and off the semiconductor switching element is generated, and the component of the predetermined frequency is extracted from the PMW information and the voltage command information and A power conversion method for detecting, by comparing, the frequency component different from the predetermined superimposed when the voltage command information is converted into the PMW information, and generating PWM information in which the detected component is suppressed.

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