JP2009124799A - Motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To quickly generate a phase voltage command equivalent to a phase voltage command with a third-harmonic wave superimposed thereon through simple computation and thereby enhance the voltage utilization ratio of an inverter. <P>SOLUTION: At a basic command value generation unit 20a, phase voltage command values Vu<SP>cmd</SP>, Vv<SP>cmd</SP>, Vw<SP>cmd</SP>are generated as basic command values of sine wave. These phase voltage command values are obtained by converting a d-axis voltage command value Vd<SP>cmd</SP>and a q-axis voltage command value Vq<SP>cmd</SP>, generated by proportional-integral control based on the difference between a d-axis current command value id<SP>cmd</SP>and a d-axis current detection value id<SP>feed</SP>and the difference between a q-axis current command value iq<SP>cmd</SP>and a q-axis current detection value iq<SP>feed</SP>, from two-phase to three-phase. At a voltage correction unit 20b, the phase voltage command values Vu<SP>cmd</SP>, Vv<SP>cmd</SP>, Vw<SP>cmd</SP>are amplified in the direction of amplitude by proportional gain and a crest value is limited by a limiter value to generate a waveform approximate to a waveform with a third-harmonic wave superimposed thereon. The generated waveform is used as new phase voltage command values Vu<SP>cmd2</SP>, Vv<SP>cmd2</SP>, Vw<SP>cmd2</SP>for PWM control and the voltage utilization ratio of the inverter is thereby enhanced. <P>COPYRIGHT: (C)2009,JPO&INPIT

Description

  The present invention relates to a motor control device that controls a three-phase permanent magnet synchronous motor by pulse width modulation of an inverter.

  Generally, PWM (Pulse Width Modulation) control is widely used as a control method for an inverter that drives an AC motor. In a three-phase permanent magnet synchronous motor, each of U, V, and W phases is used. A three-phase PWM control is performed by generating a sine wave phase voltage command and comparing the phase voltage command with a triangular wave carrier.

  However, in this triangular wave carrier comparison PWM control, in order to perform the modulation normally, the voltage command of each phase needs to be within the amplitude of the carrier, and the amplitude of the fundamental wave of the output voltage of the inverter Is limited to √3 / 2 or less of the DC voltage. For this reason, there is a problem that the voltage utilization rate of the DC voltage is low, which is an obstacle to improving the efficiency of the motor and driving at a high speed.

To cope with this, as described in Non-Patent Document 1 and Patent Document 1, the peak value of the voltage command is obtained by superimposing the third harmonic of the frequency three times the fundamental wave on the voltage command value of each phase. A technique for increasing the maximum value of the fundamental component of the output voltage to improve the voltage utilization factor of the inverter is known.
“Theory and Design of AC Servo Systems” by Hidehiko Sugimoto; Comprehensive Electronic Publishing July 10, 2005 7th Edition, P44-46 JP 2005-45846 A

  However, in the conventional technique for superimposing the third harmonic, not only the voltage command of the fundamental wave but also a sine wave having a frequency three times that of the fundamental wave needs to be generated. There is a problem that the control system becomes complicated because it is necessary to synchronize with a sine wave of three times the frequency. In addition, a complicated calculation is required to increase the processing time, and the system load may increase and the controllability may decrease.

  The present invention has been made in view of the above circumstances, and generates a phase voltage command equivalent to a phase voltage command in which the third-order harmonics are superimposed at high speed by a simple calculation, and improves the voltage utilization rate of the inverter for control. An object of the present invention is to provide a motor control device capable of improving the performance.

  In order to achieve the above object, a motor control device according to the present invention is a motor control device that controls a three-phase permanent magnet synchronous motor by pulse width modulation of an inverter, and controls the motor via the inverter. A basic command value generator for generating a phase voltage command value as a sine wave basic command value, amplifying the basic command value with a predetermined proportional gain, and limiting a peak value with a predetermined limiter value, And a voltage correction unit for correcting the basic command value as a new three-phase voltage command value.

  According to the present invention, a phase voltage command equivalent to a phase voltage command in which the third harmonic is superimposed can be generated at high speed by a simple calculation, and the controllability is improved by improving the voltage utilization rate of the inverter. be able to.

  Embodiments of the present invention will be described below with reference to the drawings. 1 to 5 relate to a first embodiment of the present invention, FIG. 1 is an overall configuration diagram of a motor control device, FIG. 2 is a block diagram of a motor control system, FIG. 3 is a block diagram of a voltage correction unit, and FIG. Is a flowchart showing voltage command correction processing, and FIG. 5 is a waveform diagram showing a waveform of a voltage command value.

  As shown in FIG. 1, the motor control device 1 of the present embodiment includes an inverter 3 that drives a three-phase permanent magnet synchronous motor (PM motor) 2 and a controller that controls the inverter 3 via a drive circuit unit 10. 20 is mainly configured. The inverter 3 is a three-phase bridge type inverter using switching elements Q1 to Q6 such as a power MOSFET, and includes an arm configured by connecting switching elements Q1 and Q2 in series and switching elements Q3 and Q4 connected in series. An arm configured and an arm configured by connecting switching elements Q5 and Q6 in series are connected in parallel. In addition, recirculation diodes D1 to D6 are connected in parallel to the switching elements Q1 to Q6 in the reverse direction, respectively.

  The switching elements Q1, Q3, Q5 on the upper side (high side) of each arm of the inverter 3 are connected to the positive side of the power storage device 4 disposed on the DC power supply side such as a smoothing capacitor or a storage battery. Switching elements Q <b> 2, Q <b> 4, Q <b> 6 on the lower side (low side) of the arm are connected to the negative side of the power storage device 4. Further, the intermediate connection point between the high side and the low side of each arm is connected to each of the U, V, and W phase terminals of the armature winding of the PM motor 2.

  The controller 20 is composed mainly of a microcomputer and includes various timers / counters, logic circuits, arithmetic units, etc., and performs PWM control of the inverter 3 in accordance with a torque command generated in the control system for the PM motor 2. The three-phase voltage command is generated. That is, the controller 20 detects the voltage sensor 5 that detects the voltage of the power storage device 4, the current sensors 6, 7, and 8 that detect the U, V, and W phase AC currents of the PM motor 2, and the rotor position of the PM motor 2. Based on the signal from the position sensor 9 to be detected, three-phase voltages of U, V, and W phases are controlled by vector control of the d axis (direction of magnetic flux generated by the magnet) and q axis (direction orthogonal to the d axis). Generate directives. Further, the controller 20 compares the voltage command value of each phase with the triangular wave carrier signal, generates a PWM control signal, and outputs it to the drive circuit unit 10.

  The drive circuit unit 10 generates a gate signal for driving the switching elements Q1 to Q6 of the inverter 3 in accordance with the PWM control signal from the controller 20, and drives the switching elements Q1 to Q6 at a predetermined frequency. Thereby, the amplitude and phase of the current supplied to the PM motor 2 are controlled, and the PM motor 2 rotates at a predetermined rotational speed.

  The PM motor 2 is controlled by the controller 20 in consideration of, for example, a high-speed motor such as a traveling motor mounted on an electric vehicle. Characteristically, a third-order phase voltage command based on a sine wave is used. A voltage command equivalent to the voltage command in which the harmonics are superimposed is generated at high speed by a simple calculation, and the voltage utilization rate in the inverter 3 is improved.

  Therefore, as shown in FIG. 2, the controller 20 generates a basic command value for generating a basic command value for generating a three-phase voltage command value for controlling the PM motor 2 via the inverter 3 as a sine wave basic command value. The main components are a value generation unit 20a and a voltage correction unit 20b that corrects the three-phase voltage command value generated by the basic command value generation unit to obtain a new phase voltage command value for PWM control. The basic command value generation unit 20a includes a current command value generation unit 21, a three-phase to two-phase coordinate conversion unit 22, subtractors 23 and 24, PI (proportional integration) control units 25 and 26, and a two-phase to three-phase coordinate conversion unit. 27 etc.

The basic command value generation unit 20a generates a d-axis current command value id ^cmd and a q-axis current command value iq ^cmd from a torque command value or the like input from the outside in the current command value generation unit 21, and in 3-phase to two-phase coordinate conversion unit 22, U detected by the current sensor 6, 7, 8, V, each of the current i _u ^feed the W-phase, i _v ^feed, the i _w ^feed, was detected by the position sensor 9 electrically Based on the angle θe, the current detection value id ^feed on the d axis is converted into the current detection value iq ^feed on the q axis.

Then, the subtracters 23 and 24 calculate the deviation between the d-axis current command value id ^cmd and the d-axis current detection value id ^feed, and the deviation between the q-axis current command value iq ^cmd and the q-axis current detection value iq ^feed , The PI control units 25 and 26 perform proportional-integral control, respectively, to generate a d-axis voltage command value Vd ^cmd and a q-axis voltage command value Vq ^cmd .

The d-axis voltage command value Vd ^cmd from the PI control unit 25 and the q-axis voltage command value Vq ^cmd from the PI control unit 26 are input to the two-phase to three-phase coordinate conversion unit 27, and two-phase based on the electrical angle θe By -3 phase conversion, phase voltage command values Vu ^cmd , Vv ^cmd , and Vw ^cmd for U, V, and W phases are generated as basic command values of sine waves.

Specifically, the conversion between the two phases and the three phases is executed using the following equations (1) and (2).

The phase voltage command values Vu ^cmd , Vv ^cmd , and Vw ^cmd of the U, V, and W phases after the two-phase to three-phase conversion are basic sine wave command values having a phase difference of 2 / 3π, and the peak value Vac Can be expressed by the following equations (3) to (5).
Vu ^cmd = Vac × sin θe (3)
Vv ^cmd = Vac × sin (θe−2 / 3 × π) (4)
Vw ^cmd = Vac × sin (θe + 2/3 × π) (5)

Next, the voltage correction unit 20b applies d to the sinusoidal phase voltage command values Vu ^cmd , Vv ^cmd , and Vw ^cmd for each of the U, V, and W phases generated by the two-phase to three-phase coordinate conversion unit 27. Correction using the axis voltage command value Vd ^cmd and the q axis voltage command value Vq ^cmd is performed, and new phase voltage command values Vu ^cmd2 , Vv ^cmd2 , and Vw ^{cmd2 in} which the third harmonic is approximately superimposed are generated.

In general, a voltage command value obtained by superimposing a third harmonic on a basic sine wave is a third order integer multiple having a peak value with a gain G (G <1) in addition to the voltage command value expressed by the equations (3) to (5). Are generated in synchronism so that the phase 0 points coincide with each other. Voltage command values Vu ^cmd3 , Vv ^cmd3 , and Vw ^{cmd3 on} which the third harmonic is superimposed are as shown in the following equations (6) to (8). In this case, the gain G is optimally set to G = 1/6, but can be slightly adjusted according to the motor and circuit characteristics.
Vu ^cmd3 = Vac (sin θe + G × sin 3θe) (6)
Vv ^cmd3 = Vac (sin (θe−2 / 3 × π) + G × sin3θe) (7)
Vw ^cmd3 = Vac (sin (θe + 2/3 × π) + G × sin3θe) (8)

  The waveform of the voltage command on which the third harmonic is superimposed becomes a waveform as shown by a solid line in FIG. 5. The voltage utilization rate of the inverter 3 is improved to increase the output voltage, and the PM motor 2 is driven to a high speed. It becomes possible. However, in order to generate such a waveform, a complicated calculation with an alternating current value similar to the equations (1) and (2) is required, the calculation amount becomes large, the load on the computer increases, and the control is increased. There is a risk of deteriorating the performance.

Therefore, in the present embodiment, by approximating the superposition of the third harmonic by a simple calculation, the same effect as the voltage command in which the third harmonic is superimposed is exhibited without increasing the calculation load. . Specifically, sinusoidal phase voltage command values Vu ^cmd , Vv ^cmd , Vw ^cmd from the two-phase / three-phase conversion unit 27 are amplified in the amplitude direction by the proportional gain K in the voltage correction unit 20 b and amplified. A substantially trapezoidal waveform is generated by limiting the upper and lower peak values of the waveform with the limiter value Vlim, and the waveform on which the third harmonic is superimposed is approximated.

Therefore, the voltage correction unit 20b corrects the phase voltage command values Vu ^cmd , Vv ^cmd , and Vw ^cmd generated by the two-phase / three-phase coordinate conversion unit 27 with the proportional gain K, as shown in FIG. Correction unit 30, 31, 32, limiter value calculation unit 33 for calculating limiter value Vlim from d-axis voltage command value Vd ^cmd and q-axis voltage command value Vq ^cmd , voltage command value of each phase corrected by proportional gain K Limiter circuit portions 34, 35, and 36 that are limited by the value Vlim are provided.

In the present embodiment, the proportional gain K and the limiter value Vlim are applied to the same value in each of the U, V, and W phases, and the gain correction units 30, 31, and 32 use the proportional gain K as system-fixed data. Assume that you have it. Hereinafter, the proportional gain K and the limiter value Vlim, will be described with reference to the AC voltage command values Vu ^cmd of U-phase typically.

First, regarding the proportional gain K, the term relating to the triangular function on the right side of the equation (6) can be decomposed as the following equation (9) using the triple angle formula.
sin θe + G × sin 3θe = sin θe + G × ( ³ sin θe−4 sin ³ θe)
= Sin θe + 3G × sin θe−4G × sin ³ θe (9)

Here, the term of 4G × sin ³ θe in the equation (9) can be ignored if the electrical angle θe is small, and when the electrical angle θe is small, the equation (6) in which the third harmonic is superimposed is as follows: (10).
Vu ^cmd3 = Vac (sin θe + G × sin 3θe)
≒ Vac (sinθe + 3G × sinθe)
= Vac × (1 + 3G) × sin θe (10)

Therefore, from the equation (10), the phase voltage command value Vu ^cmd3 on which the third harmonic is superimposed is obtained by ^changing the basic sine wave phase voltage command value Vu ^cmd to the following equation (11) in a region where the electrical angle θe is small. It can be approximated by a value amplified by the proportional gain K shown. In the present embodiment, it is assumed that G = 1/6 and the proportional gain K is K = 3/2.
K = 1 + 3G (11)

  In the approximation using the proportional gain K, the error increases as the electrical angle θe increases, so the amplitude of the approximated waveform is limited using the limiter value Vlim. This limiter value Vlim can be obtained as follows.

First, the relationship between the peak value Vac of the voltage command of each phase obtained from the equations (1) and (2) and the voltages Vd and Vq on the dq axis can be calculated by the following equation (12). .
Vac = (2/3) ^1/2 × (Vd ² + Vq ² ) ^1/2 (12)

Here, using the electrical angle θe near the peak of the voltage waveform so that the peak value of the voltage command corrected with the proportional gain K is substantially equal to the peak value of the voltage command value superimposed with the third harmonic, The term sinθe + G × sin3θe on the right side of the equation (6) is calculated. For example, when θe = 90 ° at which the voltage command value of the sine wave is maximum, the term sinθe + G × sin3θe on the right side of equation (6) is calculated as shown in equation (13) below.
sin90 ° + G × sin270 ° = 1−G (13)

Therefore, the limit value Vlim is set by multiplying the peak value Vac by using the value (1-G) of the equation (13) as a coefficient, and the upper and lower limits of the voltage command value corrected by the proportional gain K are limited. In the present embodiment, G = 1/6, and the limiter value Vlim is set as in the following equation (14) with respect to the upper and lower limits of positive and negative. Note that Vd and Vq in the equation (14) are calculated using the d-axis voltage command value Vd ^cmd and the q-axis voltage command value Vq ^cmd , respectively.
± Vlim = ± (1-G) × Vac
= ± 5/6 × (2/3) ^1/2 × (Vd ² + Vq ² ) ^1/2 (14)

  In this case, the limiter value Vlim is not necessarily limited to the calculation result at θe = 90 °, and takes the vicinity of the peak of the voltage command waveform on which the third harmonic is superimposed in consideration of motor characteristics, circuit characteristics, and the like. It may be set from a calculation result at an electrical angle θe (for example, θe = 60 ° to 90 °). By setting a limiter value according to actual motor characteristics and circuit characteristics, the power utilization rate can be effectively increased. Can be improved.

Next, the phase voltage command value correction process using the proportional gain K and the limiter value Vlim obtained in this way will be described with reference to the flowchart of FIG. Although an example in which the U-phase voltage command value Vu ^cmd is corrected will be described here, the same correction is performed for the V-phase and the W-phase.

This phase voltage command correction process is executed every predetermined calculation time. First, in the first step S1, the sine wave phase voltage command value Vu ^cmd is corrected by the proportional gain K, and the command corrected by this proportional gain K is corrected. It is checked whether or not the value Vu ^cmd × K is larger than the upper limit value Vlim shown by the above-described equation (14).

As a result, when Vu ^cmd × K> Vlim, the process proceeds from step S1 to step S2, the command value Vu ^cmd × K corrected by the proportional gain K is limited by the upper limit value Vlim, and a new phase voltage command value Vu ^Output as ^cmd2 (Vu ^cmd2 = Vlim).

On the other hand, when Vu ^cmd × K ≦ Vlim in step S1, the process proceeds from step S1 to step S3 to check whether or not the command value Vu ^cmd × K corrected with the proportional gain K is smaller than the lower limit limit value −Vlim. .

When Vu ^cmd × K <−Vlim, the process proceeds from step S3 to step S4, the command value Vu ^cmd × K corrected by the proportional gain K is limited by the lower limiter value −Vlim, and a new phase voltage command value is set. ^Output as Vu ^cmd2 (Vu ^cmd2 = −Vlim).

If Vu ^cmd × K ≧ −Vlim in step S3, the process proceeds from step S3 to step S5, and the command value Vu ^cmd × K corrected with the proportional gain K is output as a new phase voltage command value Vu ^cmd2. (Vu ^cmd2 = Vu ^cmd × K).

FIG. 5 shows a result of generating a new voltage command value Vu ^cmd2 by the above processing. In FIG. 5, the waveform indicated by the one-dot chain line is the voltage command value Vu ^cmd of the sine wave before correction, and the two-dot chain line. The waveform shown is the third harmonic, the waveform shown by the solid line is the voltage command value Vu ^cmd3 generated by superimposing the third harmonic by the conventional method, and the waveform shown by the broken line is after correction using the proportional gain K and the limiter value Vlim. ^Is a new voltage command value Vu ^cmd2 . This result is the corrected voltage command value Vu ^cmd2, it can be seen that can well approximate the conventional voltage command values Vu ^cmd3 generated by superimposing a third harmonic wave method.

  In the above calculation example, the example of performing the voltage correction calculation using only the voltage command value has been described. However, a circuit for detecting a three-phase voltage is provided, and the voltage correction calculation is performed using the detected three-phase voltage. May be. The limiter value in this case is the average value of the detected peak values of the three-phase voltage, or the peak value obtained by converting to the voltage on the dq axis using the two-phase to three-phase conversion as in the case of current detection. Therefore, it is possible to perform stable control even for a motor having a distorted voltage characteristic.

As described above, in the present embodiment, the voltage command values Vu ^cmd , Vv ^cmd , and Vw ^cmd generated by performing the two-phase to three-phase conversion on the d-axis voltage command value Vd ^cmd and the q-axis voltage command value Vq ^cmd are obtained. For limiting the crest value calculated from the proportional gain K for making the amplitude substantially the same as the voltage waveform obtained by superimposing the third harmonic on the basic sine wave, the d-axis voltage command value Vd ^cmd, and the q-axis voltage command value Vq ^cmd And the inverter 3 is controlled as new phase voltage command values Vu ^cmd2 , Vv ^cmd2 , Vw ^cmd2 . As a result, it is possible to approximate a waveform in which the third harmonic is superimposed by a simple calculation, improve the voltage utilization rate of the inverter 3 that drives the PM motor 2, improve the controllability, and rotate the PM motor 2 at a high speed. Can be driven.

  Next, a second embodiment of the present invention will be described. 6 and 7 relate to the second embodiment of the present invention, FIG. 6 is a block diagram of the voltage correction unit, and FIG. 7 is a flowchart showing variable control processing of the proportional gain and limiter value.

  In the first embodiment, the proportional gain K and the limiter value Vlim are fixed values (K = 3/2, Vlim = 5/6 × Vac), respectively, but in the second embodiment, the proportional gain K and the limiter are set. By changing the value Vlim, it is possible to smoothly shift to the optimum control according to the rotation range of the PM motor 2.

  For this reason, in the second mode, the voltage correction unit 20b of the first mode is changed as shown in FIG. In other words, the voltage correction unit 20b ′ of the second form has a proportional gain K ′ with the variable value of the proportional gain K of the first form as a variable value compared to the voltage correction unit 20b of the first form. In addition to the variable gain calculation unit 50 to be output to 32, the limiter value Vlim ′ having the limiter value Vlim as a variable value is replaced with the limiter circuit units 34 and 35 in place of the limiter value calculation unit 33 for calculating the limiter value Vlim as a fixed value. , 36 is provided with a variable limiter value calculation unit 51.

The variable gain calculation unit 50 and the variable limiter value calculation unit 51 compare the crest value Vac ′ of the phase voltage command value obtained from the d-axis voltage command value Vd ^cmd and the q-axis voltage command value Vq ^cmd with a threshold value, and compare them. Depending on the results, the proportional gain K ′ and the limiter value Vlim ′ are varied from K ′ = 1 to 3/2 and ± Vlim ′ = ± Vac to ± 5/6 × Vac, respectively. Here, the peak value Vac ′ is a value represented by (Vd ² + Vq ² ) ^1/2 ), omitting the coefficient (2/3) ^1/2 in the above-described equation (12).

  In this embodiment, as the threshold value, three set values of the first threshold value Vdq1, the second threshold value Vdq2, and the third threshold value Vdq3 (Vdq1 <Vdq2 <Vdq3) are used, and the peak value Vac ′ and each threshold value are used. Depending on the magnitude relationship with Vdq1, Vdq2, Vdq3, proportional gain K 'and limiter value Vlim' are variably set, phase voltage command for basic sine wave, phase voltage command approximating superposition of third harmonic, field weakening control Smooth transition to different control states such as phase voltage commands.

  Hereinafter, the variable control processing of the proportional gain and the limiter value will be described with reference to the flowchart shown in FIG.

  First, in step S11, the peak value Vac ′ of the phase voltage command value is compared with the first threshold value Vdq1, and when Vac ′ ≦ Vdq1, that is, when the peak value Vac ′ is equal to or less than the first threshold value Vdq1, step In S12, the proportional gain K ′ is set to 1 and the limiter value Vlim ′ is set as an initial set value. The initial set value in this case is a design value for the purpose of safety protection of the inverter 3.

  In this case, the control is provided with a safety protection limiter for driving by the basic sine wave by K '= 1, but when the limiter process is not performed, the normal sine wave driving is performed. As a result, a smooth transition of the control system from the sine wave drive control to the control by the correction command value can be realized, and also when the shift from the control by the correction command value to the sine wave drive, the transition is smooth. It becomes possible.

  On the other hand, when Vac ′> Vdq1 in step S11, the process proceeds from step S11 to step S13, and a value obtained by adjusting the calculation value F with a predetermined adjustment gain A1 is set as a proportional gain K ′ (K ′ = A1 × F). Further, a value obtained by dividing the fixed limiter value Vlim by the calculated value F and adjusting with a predetermined adjustment gain A2 is set as the limiter value Vlim ′ (± Vlim ′ = ± A2 × 1 / F × Vlin).

The calculated value F is calculated as a value representing the amount of change in the peak value Vac ′ between the first threshold value Vdq1 and the second threshold value Vdq2, for example, as shown in the following equation (15). As long as the condition of Vac ′ ≦ Vdq2 is satisfied in the next step S14, the proportional gain K ′ and the limiter value Vlim ′ are within the range of 1 ≦ K ′ <3/2, 5/6 × Vac ≦ Vlim ′ <Vac. Slowly variable.
F = (Vac′−Vdq1) / (Vdq2−Vdq1) (15)

  Next, when Vac ′> Vdq2, the process proceeds from step S14 to step S15, and the proportional gain K ′ and limiter value Vlim ′ are set to their maximum values (K ′ = 3/2, Vlim ′ = 5/6 × Vac). The process proceeds to step S16. In step S16, the peak value Vac ′ is compared with the third threshold value Vdq3. If Vac ′ ≦ Vdq3, the process exits and holds the current proportional gain K ′ and the limiter value Vlim ′, and Vac ′> Vdq3. In this case, the process proceeds to field weakening control in step S17.

  The third threshold value Vdq3 is set to a value equal to or less than the maximum voltage value that can be output by the inverter 3 or equal to the maximum voltage value, and a smooth transition to field weakening control is possible while ensuring a voltage margin. In this case, the third threshold value Vdq3 may be the same value as the second threshold value Vdq2. However, the final voltage command is limited to a range where the inverter 3 can be used safely.

  In the above description, the example in which the peak value Vac ′ of the phase voltage command value is compared with the three threshold values Vdq1, Vdq2, and Vdq3 has been described. However, a circuit for detecting a three-phase voltage is provided, and the detected three-phase voltage is expressed on the dq axis. Comparison with a plurality of threshold values may be performed using the peak value obtained by coordinate conversion to the upper voltage value, and stable control can be performed even for a motor having distorted voltage characteristics.

  Each threshold value may be set according to the rotational speed of the PM motor 2, the DC voltage of the inverter 3, and the like. According to control purposes such as stability improvement, reliability improvement, performance improvement, etc. It is possible to set.

  For example, by setting a plurality of threshold values according to the motor rotation speed, the PM motor 2 is efficiently driven by sinusoidal drive in the low rotation speed region, and the third harmonic is superimposed in the high rotation speed region. As with the voltage command, the voltage utilization rate in the inverter 3 can be increased and the controllability can be improved.

  Moreover, when setting a some threshold value according to the direct current voltage of the inverter 3, the influence of the fluctuation | variation of direct current voltage can be considered and controllability can be improved. At that time, it is possible to further improve the stability of the control system by setting each threshold value according to the current command value or the torque command value.

  As described above, in the second embodiment, smooth control switching can be performed by varying the proportional gain and the limiter value. In addition, when the peak value of the phase voltage command value or the average value of the detected peak values of the three-phase voltage exceeds the first threshold value Vdq1, the proportional gain and the limiter value are varied, and the peak value exceeds the second threshold value Vdq2. In this case, the proportional gain and the limiter value are fixed, and when the third threshold value Vdq3 is exceeded, the control state is smoothly switched and the control stability after switching can be secured by shifting to the field weakening control. .

1 is a diagram illustrating an overall configuration of a motor control device according to a first embodiment of the present invention. Same as above, block diagram of motor control system Same as above, voltage correction block diagram Same as above, flowchart showing voltage command correction processing Same as above, waveform diagram showing waveform of voltage command value The block diagram of a voltage correction part concerning 2nd Embodiment of this invention. Same as above, flowchart showing variable control processing of proportional gain and limiter value

Explanation of symbols

1 Motor controller 2 3-phase permanent magnet synchronous motor 3 inverter 20 controller 20a basic command value generating section 20b voltage correcting unit K proportional gain Vlim limiter value Vac peak value Vu ^cmd, Vv ^cmd, Vw ^cmd voltage command value (basic command value)
Vu ^cmd2 , Vv ^cmd2 , Vw ^cmd2 voltage command value (corrected voltage command value)
θe Electrical angle Vdq1, Vdq2, Vdq3 Threshold

Claims (8)

A motor control device for controlling a three-phase permanent magnet synchronous motor by pulse width modulation of an inverter,
A basic command value generation unit for generating a three-phase voltage command value for controlling the motor via the inverter as a basic command value of a sine wave;
A voltage correction unit that amplifies the basic command value with a predetermined proportional gain, limits a peak value with a predetermined limiter value, and corrects the basic command value as a new three-phase voltage command value for pulse width modulation; A motor control device comprising the motor control device.   The proportional gain is set in consideration of the electrical angle when the third harmonic is superimposed on the basic command value, and the limiter value is set based on the voltage on the dq axis for generating the basic command value. The motor control device according to claim 1, wherein the motor control device is set.   3. The motor control device according to claim 1, wherein the proportional gain and the limiter value are varied based on a comparison result between a parameter based on a three-phase voltage or a voltage on the dq axis and a threshold value.   4. The motor control device according to claim 3, wherein the threshold value is set to a plurality of threshold values according to the rotation speed of the motor. 4. The motor control device according to claim 3, wherein the threshold value is set to a plurality of threshold values according to the DC voltage of the inverter. The motor control device according to claim 3, wherein the threshold value is set to a plurality of threshold values according to a current command value or a torque command value of the inverter.   4. The motor control device according to claim 3, wherein the proportional gain is varied within an amplification range of 1 to 1.5 times the basic command value.   4. The motor control device according to claim 3, wherein the limiter value is varied within a limit range of 1 to 5/6 times the peak value of the basic command value.

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