JP6287715B2 - Rotating machine control device - Google Patents

Description

  The present invention relates to a control device for a rotating machine that is applied to a rotating machine that is electrically connected to a power conversion circuit having a switching element, and controls a control amount of the rotating machine to a target value by operating the switching element.

  As this type of control device, as can be seen in Patent Document 1 below, in order to control the control amount (torque) of the rotating machine to its target value, the phase of the output voltage vector of the power conversion circuit and the output voltage vector It is known to manipulate the amplitude (norm) of. Specifically, this control device performs control amount control based on the amplitude of the output voltage vector and control amount control based on the phase of the output voltage vector.

JP 2012-23943 A

  By the way, since the maximum value that the amplitude of the output voltage vector can take is determined according to the input voltage of the power conversion circuit that applies the voltage to the rotating machine, there is an upper limit value for the amplitude of the output voltage vector. Here, when the control of the control amount based on the amplitude causes so-called voltage saturation in which the amplitude reaches the upper limit value, the control amount of the rotating machine can vary greatly. In this case, the control accuracy of the control amount is lowered and the control becomes unstable.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a control device for a rotating machine that can suitably suppress a decrease in controllability of a control amount when voltage saturation occurs. .

  In order to achieve the above object, the present invention is applied to a rotating machine (10) electrically connected to a power conversion circuit (20) having switching elements (SUp to SWn), and the control amount of the rotating machine is set as the control amount. As the operation amount for feedback control to the target value, the phase of the output voltage vector of the power conversion circuit in the rotating coordinate system of the rotating machine, or a component according to the phase among the orthogonal biaxial components of the output voltage vector A phase setting means (30e; 30k) for setting a certain phase operation amount, and an operation amount for feedback control of the current flowing through the rotating machine to a command current corresponding to the target value, or the amplitude of the output voltage vector, or Amplitude setting means (32g; 32i) for setting an amplitude manipulated variable that is a component in accordance with the amplitude among the orthogonal two-axis components of the output voltage vector, and the control variable An operation means (30j) for operating the switching element based on the phase operation amount set by the phase setting means and the amplitude operation amount set by the amplitude setting means to control to the target value, Before the modulation factor determined by the fundamental wave component of the voltage output from the power conversion circuit to the rotating machine and the input voltage of the power conversion circuit reaches its upper limit value, the amplitude setting means controls the command current with respect to the command current. Amplitude response lowering means (34c; 36b) for reducing the responsiveness of the current flowing through the rotating machine in advance according to the modulation rate is provided.

  In a state where the response of the current flowing through the rotating machine to the command current by the amplitude setting means is set high, the control amount greatly fluctuates when so-called voltage saturation occurs in which the modulation rate increases and reaches its upper limit value. This is due to the fact that the increase rate (slope) of the amplitude of the output voltage vector immediately before the modulation rate reaches the upper limit value increases due to the high responsiveness of the amplitude setting means. Therefore, in the above invention, before the modulation factor reaches the upper limit value, the responsiveness of the current flowing through the rotating machine to the command current by the amplitude setting means is reduced in advance according to the modulation factor. Thereby, it can be avoided that the control amount fluctuates greatly when the modulation rate reaches the upper limit value. Here, as the amplitude response lowering means, specifically, one that lowers the responsiveness of the current flowing through the rotating machine to the command current by the amplitude setting means in the second range than in the first range can be adopted. . As a result, when the modulation rate approaches the upper limit value, the rate of increase in the amplitude of the output voltage vector decreases, and it is possible to suitably avoid the control amount from fluctuating greatly when the modulation rate reaches the upper limit value.

  Further, when the responsiveness by the amplitude setting means is reduced in the second range, the responsiveness of the control amount by the phase setting means is not lowered in the second range than in the first range. For this reason, the responsiveness of the controlled variable can be maintained as much as possible. As described above, the controllability of the control amount can be maintained high in the first range where the modulation rate has a margin with respect to the upper limit value, and the controllability of the control amount is preferably lowered when the modulation rate reaches the upper limit value. Can be suppressed.

1 is an overall configuration diagram of a motor control system according to a first embodiment. The block diagram of motor control. The characteristic view which shows the relationship between a voltage phase and a torque. The figure for demonstrating (lambda) axis | shaft. The figure which shows the calculation method of the angle which d axis | shaft and (lambda) axis | shaft make. The figure for demonstrating (lambda) axis | shaft. The figure which shows the calculation method of (lambda) axis current. The block diagram of a gain setting part and a responsiveness setting part. The figure which shows the relationship between a rotational speed, a voltage phase, and a proportional gain. The figure which shows the relationship between a rotational speed, a voltage phase, and a torque. The figure which shows the relationship between a modulation factor and responsiveness. The figure which shows the effect which made the response of amplitude control variable. The figure which shows the effect which made the response of amplitude control variable. The block diagram of the motor control concerning 2nd Embodiment. The figure which shows the effect which made the response of amplitude control variable. The figure which shows the effect which made the response of amplitude control variable. The figure which shows the relationship between the modulation factor concerning 3rd Embodiment, and responsiveness. The figure which shows the effect which made the response of amplitude control variable. The figure which shows the relationship between the dq coordinate system concerning 4th Embodiment, and pl coordinate system. The block diagram of motor control. The figure which shows the relationship between the modulation rate concerning other embodiment, and responsiveness. The figure which shows the effect which made the response of amplitude control variable.

(First embodiment)
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a vehicle (for example, an electric vehicle or a hybrid vehicle) including a rotating machine as an in-vehicle main machine will be described with reference to the drawings.

  As shown in FIG. 1, the motor control system includes a motor generator 10, an inverter 20 as a “power conversion circuit”, and a control device 30 that controls the motor generator 10. In the present embodiment, the motor generator 10 is an in-vehicle main machine and is connected to drive wheels (not shown). In the present embodiment, an IPMSM that is a salient pole machine is used as the motor generator 10.

  The motor generator 10 is connected to a battery 22 as a DC power source via an inverter 20. The output voltage of the battery 22 is, for example, 100 V or more. A smoothing capacitor 24 that smoothes the input voltage of the inverter 20 is provided between the battery 22 and the inverter 20.

  The inverter 20 includes a series connection body of switching elements SUp, SVp, SWp on the upper arm side and switching elements SUn, SVp, SWn on the lower arm side. Specifically, the inverter 20 includes three sets of serially connected bodies of switching elements on the upper arm side and switching elements on the lower arm side. The connection points of the upper arm side switching elements SUp, SVp, SWp and the lower arm side switching elements SUn, SVn, SWn are connected to the U, V, W phases of the motor generator 10. Incidentally, in the present embodiment, voltage-controlled semiconductor switching elements are used as the switching elements SUp to SWn, and more specifically, IGBTs are used. Free wheel diodes DUp to DWn are connected in antiparallel to the switching elements SUp to SWn.

  The motor control system further includes a V-phase current sensor 42V that detects a current flowing in the V-phase of the motor generator 10, a W-phase current sensor 42W that detects a current flowing in the W-phase, and an input voltage (output from the battery 22) of the inverter 20. And a rotation angle sensor 46 (for example, a resolver) for detecting the rotation angle (electrical angle θ) of the motor generator 10.

  The control device 30 is mainly composed of a microcomputer, and operates the inverter 20 so as to feedback-control a control amount (torque in this embodiment) related to the torque of the motor generator 10 to a target value (hereinafter, target torque Trq *). . Specifically, the control device 30 generates the operation signals gUp to gWn based on the detection values of the various sensors so as to turn on and off the switching elements SUp to SWn constituting the inverter 20, and the generated operation signals gUp to gWn. Are output to the drive circuits DrUp to DrWn (gate drive circuit). Here, the operation signals gUp, gVp, gWn on the upper arm side and the corresponding operation signals gUn, gVn, gWn on the lower arm side are complementary to each other. That is, the switching elements SUp, SVp, SWp on the upper arm side and the corresponding switching elements SUn, SVn, SWn on the lower arm side are alternately turned on. The target torque Trq * is, for example, a control device provided outside the control device 30, and is output from a control device higher than the control device 30.

  Next, torque control of the motor generator 10 executed by the control device 30 will be described using FIG. This control includes phase control and amplitude control.

  First, phase control will be described. The two-phase converter 30a is based on the V-phase current IV detected by the V-phase current sensor 42V, the W-phase current IW detected by the W-phase current sensor 42W, and the electrical angle θ detected by the rotation angle sensor 46. The U-phase current IU, V-phase current IV, and W-phase current IW in the three-phase fixed coordinate system are converted into the d-axis current Idr and the q-axis current Iqr in the two-phase rotational coordinate system (dq coordinate system). The U-phase current IU may be calculated from the V-phase current IV and the W-phase current IW based on Kirchhoff's law.

  Torque estimator 30b calculates estimated torque Te of motor generator 10 based on the d and q axis currents Idr and Iqr output from two-phase converter 30a. Here, the estimated torque Te may be calculated using a map in which the d-axis current Idr and the q-axis current Iqr are associated with the estimated torque Te, or may be calculated using a model formula. In the present embodiment, the torque estimator 30b corresponds to “torque estimation means”.

  The filter 30c removes a high frequency component from the estimated torque Te calculated by the torque estimator 30b. In the present embodiment, the filter 30c is configured as a low-pass filter. The torque deviation calculation unit 30d calculates the torque deviation ΔT by subtracting the estimated torque Te from which the high frequency component has been removed from the target torque Trq *.

  The phase setting unit 30e (corresponding to the “phase setting unit”) uses a phase operation as an operation amount for performing feedback control of the estimated torque Te to the target torque Trq * based on the torque deviation ΔT calculated by the torque deviation calculation unit 30d. The voltage phase φ as a quantity is calculated. Specifically, the voltage phase φ is calculated by proportional integral control using the torque deviation ΔT as an input. More specifically, the output value “Kpφ × ΔT” of the proportional controller having the torque deviation ΔT as an input, and the output value “Kiφ × ∫ (ΔT × dt)” of the integral controller having the torque deviation ΔT as an input, The voltage phase φ is calculated as an added value of. Here, “Kpφ” indicates a proportional gain in phase control, and “Kiφ” indicates an integral gain in phase control. In the present embodiment, the voltage phase φ is defined with the positive direction of the d-axis as a reference, and the counterclockwise direction from this reference (the direction rotating from the positive direction of the d-axis to the positive direction of the q-axis) is defined as the positive direction. Has been. Therefore, when the estimated torque Te is insufficient with respect to the target torque Trq *, the voltage phase φ is increased (advanced) as shown in FIG. 3, and the estimated torque Te is excessive with respect to the target torque Trq *. In this case, the voltage phase φ is decreased (retarded). FIG. 3 also shows that the torque also depends on the rotational speed (electrical angular speed) of the motor generator 10.

  Next, amplitude control will be described. As shown in FIG. 2, the command voltage setting unit 30f calculates the standardized voltage amplitude “Vn / ω” using the target torque Trq * as an input. Here, the normalized voltage amplitude “Vn / ω” is a value obtained by dividing the amplitude command value (hereinafter, voltage amplitude Vn) of the output voltage vector of the inverter 20 in the two-phase rotating coordinate system by the electrical angular velocity ω. . The voltage amplitude Vn is defined as the square root of the sum of the square value of the d-axis component Vd and the square value of the q-axis component Vq of the output voltage vector. In the present embodiment, the normalized voltage amplitude is calculated using a map in which the target torque Trq * and the normalized voltage amplitude are related.

  The speed calculation unit 30g calculates the electrical angular speed ω of the motor generator 10 based on the electrical angle θ detected by the rotation angle sensor 46. The speed multiplication unit 30h calculates the voltage amplitude Vn by multiplying the normalized voltage amplitude “Vn / ω” by the electrical angular velocity ω. Voltage amplitude Vn is an operation amount for performing feedforward control of torque of motor generator 10 to target torque Trq *.

  The correction unit 30i corrects the voltage amplitude Vn by adding the amplitude correction amount ΔV calculated by the correction amount calculation unit 32 to the voltage amplitude Vn output from the speed multiplication unit 30h. The correction amount calculation unit 32 will be described in detail later.

  The operation signal generation unit 30j (corresponding to “operation means”) includes the voltage amplitude “Vn + ΔV” output from the correction unit 30i, the voltage phase φ output from the phase setting unit 30e, and the input voltage detected by the voltage sensor 44. Based on VINV, operation signals gUp to gWn are generated and output to the drive circuits DrUp to DrWn. In the present embodiment, the operation signal is generated as follows.

  The operation signal generator 30j first calculates a sine wave three-phase command voltage whose phases are shifted from each other by 120 degrees in electrical angle. Then, the operation signals gUp to gWn are generated by the triangular wave comparison PWM control based on the magnitude comparison between the calculated three-phase command voltage and a carrier signal (for example, a triangular wave signal).

  Incidentally, in the present embodiment, in the case of an overmodulation region in which the modulation factor Mr exceeds the first specified value M1 (eg, 1.15), the above-described torque control (overmodulation control) is performed. In the present embodiment, the modulation factor Mr is a value obtained by normalizing the voltage amplitude Vn with the input voltage VINV. More specifically, the modulation factor Mr is a value obtained by dividing a value obtained by dividing the voltage amplitude Vn by “½” of the input voltage VINV by “√ (1.5)”. In the overmodulation region, the amplitude or effective value of the fundamental wave component included in the inverter 20 output voltage (specifically, the line voltage) is included in the output voltage of the inverter 20 when the output voltage of the inverter 20 is a sine wave. It becomes larger than the amplitude or effective value of the fundamental wave component. In particular, when the modulation factor Mr becomes a second specified value M2 (for example, 1.27, which corresponds to the “upper limit value”) higher than the first specified value M1, rectangular wave control (one-pulse control) is performed. Is called. When rectangular wave control is performed, the pulse pattern includes a period for turning on the upper arm side switching element and a period for turning on the lower arm side switching element in one cycle of the electrical angle θ. Pulse pattern (one pulse waveform) is selected. The rectangular wave control is also performed when the modulation factor Mr becomes larger than the second specified value M2. On the other hand, when the modulation factor Mr is equal to or less than the first specified value M1, sine wave PWM control is performed in which the output voltage of the inverter 20 is a sine wave of the electrical angular velocity ω.

  Subsequently, a design method of the correction amount calculation unit 32 will be described with reference to FIGS.

  The voltage equation of the permanent magnet synchronous machine is expressed by the following equation (eq1).

In the above equation (eq1), “p” indicates a differential operator, “R” indicates an armature winding resistance, “Ld” and “Lq” indicate d and q axis inductances, and “ψ” is permanent. The effective value of the armature flux linkage of the magnet is shown. In the above equation (eq1), assuming a steady state in which the rotation speed of the motor generator 10 is constant and imposing a condition that the transient phenomenon is ignored, “p = 0” is obtained. Further, a condition that the rotational speed of the motor generator 10 is sufficiently high and the relationship of “R << ω · Ld” and “R << ω · Lq” is satisfied is given to the above equation (eq1). From the above, the above equation (eq1) is expressed as the following equation (eq2).

The relationship between the d and q axis voltages Vd and Vq, the voltage phase φ, and the voltage amplitude Vn is expressed by the following equation (eq3).

Here, the voltage equation when the voltage phase φ is changed by a minute amount Δφ is expressed by the following equation (eq4) using the above equations (eq2) and (eq3).

When the above equation (eq2) is subtracted from the above equation (eq4), the following equation (eq5) is derived.

In the above equation (eq5), “Idφ−Id” on the right side is the d-axis current change amount ΔIdφ, and “Iqφ−Iq” is the q-axis current change amount ΔIqφ. When the above equation (eq5) is solved for each current change amount ΔIdφ, ΔIqφ, the following equation (eq6) is derived.

FIG. 4 shows a voltage vector Vnvt and a current vector Invt in the dq coordinate system. Here, the current vector is defined as the square root of the sum of the square value of the d-axis current and the square value of the q-axis current. In FIG. 4, the change amount of the current vector Invt when the voltage phase φ is changed by a minute amount Δφ is indicated by “ΔIφ”. Further, a change amount of the current vector Invt when the voltage amplitude is changed by a minute amount ΔVn is indicated by “ΔIvn”. FIG. 5 shows an enlarged view of the change in the current vector Invt. From the above equation (eq6), when the voltage phase φ is slightly changed, the change direction α of the current vector Invt with respect to the d axis is represented by the following equation (eq7).

For example, the change direction α can be calculated between “−π to + π” by the arctangent calculation of the above equation (eq7). Particularly in the present embodiment, when the denominator in the parenthesis is 0 and the numerator is a positive value on the right side of the above equation (eq7), the change direction α is calculated as “π / 2”. On the other hand, in the right side of the above equation (eq7), when the denominator in the parenthesis is 0 and the numerator is a negative value, the change direction α is calculated as “−π / 2”. Here, in FIG. 6, a coordinate axis extending in a direction orthogonal to the changing direction of the current vector Invt is shown as a λ axis. Of the change ΔIvn of the current vector when the voltage amplitude Vn changes by a minute amount ΔVn, the λ-axis component (that is, the component obtained by mapping the change ΔIvn to the λ-axis) is affected by the change of the voltage phase φ. There is no current. In the present embodiment, this current is used as the λ-axis current Iλ for calculating the amplitude correction amount ΔV. By using the λ-axis current Iλ, interference between amplitude control and phase control can be suppressed. Here, the angle λ formed by the d-axis and the λ-axis, which is a parameter necessary for setting the λ-axis, is expressed by the following equation (eq8).

Based on the above, returning to FIG. 2, the correction amount calculation unit 32 will be described.

  The λ-axis setting unit 32a is based on the d and q-axis inductances Ld and Lq and the voltage phase φ output from the phase setting unit 30e, and the angle formed between the d-axis and the λ-axis based on the above equation (eq8). λ is calculated. The λ-axis setting unit 32a corresponds to a non-interacting axis setting unit that sets a non-interacting coordinate axis (λ axis) in which a change in the current vector Invt is made non-interfering in the dq coordinate system. The λ axis set in the λ axis setting unit 32a changes each time the driving state of the motor generator 10 changes.

  The command current setting unit 32b sets the d and q axis command currents Id * and Iq * based on the target torque Trq *. The command currents Id * and Iq * are set to values that can realize the target torque Trq *. In the present embodiment, currents for realizing minimum current / maximum torque control are set as d and q-axis command currents Id * and Iq *. The minimum current / maximum torque control is described, for example, on page 23 of “Design and control of embedded magnet synchronous motor: Yoji Takeda et al., 3 others, Ohmsha, April 20, 2006, first edition”. Has been.

  Based on the command currents Id * and Iq * output from the command current setting unit 32b and the angle λ output from the λ axis setting unit 32a, the λ-axis command current calculation unit 32c is based on the following equation (eq9). Then, the λ-axis command current Iλ * is calculated (see FIG. 7).

In FIG. 7, the current command current vector is indicated by “In *”, and the current current vector is indicated by “Invt”.

  The λ-axis actual current calculation unit 32d (corresponding to “non-interacting current calculation unit”) includes the d and q-axis currents Idr and Iqr output from the two-phase conversion unit 30a and the angle output from the λ-axis setting unit 32a. Based on λ, the λ-axis current Iλr is calculated based on the following equation (eq10) (see FIG. 7).

Since the λ axis changes as the driving state of the motor generator 10 changes, the λ axis current Iλr and the λ axis command current Iλ * also change as the driving state of the motor generator 10 changes.

  The filter 32e removes a high frequency component from the λ-axis current Iλr calculated by the λ-axis actual current calculation unit 32d. In the present embodiment, the filter 32e is configured as a low-pass filter.

  The current deviation calculator 32f calculates the current deviation ΔIλ by subtracting the λ-axis current Iλr from which the high-frequency component has been removed from the λ-axis command current Iλ *. Based on the current deviation ΔIλ, the amplitude correction amount calculation unit 32g (corresponding to the “amplitude setting means”) performs an operation amount (in other words, an estimated torque Te) for feedback control of the λ-axis current Iλr to the λ-axis command current Iλ *. Is calculated as an amplitude manipulated variable, and an amplitude correction amount ΔV as an amplitude manipulated variable is calculated. Specifically, the amplitude correction amount ΔV is calculated by proportional-integral control using the current deviation ΔIλ as an input. More specifically, as shown in the following equation (eq11), the amplitude is obtained by adding the output value of the proportional controller having the current deviation ΔIλ as an input and the output value of the integral controller having the current deviation ΔIλ as an input. A correction amount ΔV is calculated.

In the above equation (eq11), “Kpλ” represents a proportional gain in amplitude control, and “Kiλ” represents an integral gain in amplitude control.

  As described above, according to the present embodiment, the proportional gain Kpλ and the integral gain Kiλ in the amplitude correction amount calculation unit 32g and the phase setting unit 30e can be increased. For this reason, the responsiveness of feedback control by amplitude control can be improved to a level equivalent to the responsiveness of feedback control by phase control. Thereby, both high torque controllability and high current controllability can be maintained even when a disturbance occurs or a transient state occurs. Further, according to the present embodiment, even when the feedforward control in the amplitude control is inappropriate, such as when the accuracy of the map in the command voltage setting unit 30f is low, the high torque controllability and the high current controllability are achieved. Both can be maintained.

  Next, a feedback gain variable method in amplitude control and phase control will be described with reference to FIG.

  In the present embodiment, as shown in FIG. 8, the correction amount calculation unit 32 corresponds to an amplitude gain setting unit 34a (corresponding to “amplitude gain setting unit”) and a phase gain setting unit 34b (corresponding to “phase gain setting unit”). ) And a responsiveness setting unit 34c (corresponding to "amplitude response lowering means"). The proportional gain Kpλ and the integral gain Kiλ used by the amplitude correction amount calculation unit 32g are variably set by the amplitude gain setting unit 34a. Further, the proportional gain Kpφ and the integral gain Kiφ used in the phase setting unit 30e are variably set by the phase gain setting unit 34b. Such a setting is made to maintain high feedback control responsiveness in each of amplitude control and phase control even when the driving state of the motor generator 10 changes. Hereinafter, a gain setting method will be described.

  First, the gain used in the amplitude correction amount calculation unit 32g will be described.

  The voltage equation when the voltage amplitude Vn changes by a minute amount ΔVn is expressed by the following equation (eq12) using the above equations (eq2) and (eq3).

When the above equation (eq2) is subtracted from the above equation (eq12), the following equation (eq13) is derived.

In the above equation (eq13), “Idv−Id” is the d-axis current change amount ΔIdv, and “Iqv−Iq” is the q-axis current change amount ΔIqv. Solving the above equation (eq13) for each current change amount ΔIdv, ΔIqv, the following equation (eq14) is derived.

A coordinate system obtained by rotating the dq coordinate system about the origin 0 clockwise by λ is defined as a λo coordinate system. Each current change amount ΔIλ, ΔIo in this coordinate system can be expressed by the following equation (eq15) based on the above equation (eq14).

“Λ” in the above equation (eq15) is represented by the above equation (eq8). Therefore, the proportional gain Kpλ changes according to the voltage phase φ and the rotational speed (electrical angular velocity ω) as shown in FIG. When the feedback control is performed with the proportional gain and the integral gain kept constant despite the gain characteristic changing, the response varies depending on the driving state of the motor generator 10. For this reason, a driving state in which the responsiveness is relatively lowered may occur. Therefore, in order to improve the responsiveness of the amplitude control in all the drive regions of the motor generator 10, it is required to variably set the gain used in the feedback control in accordance with the drive state. In the present embodiment, the amplitude gain setting unit 34a sets the proportional gain Kpλ and the integral gain Kiλ based on the voltage phase φ and the electrical angular velocity ω in order to maintain constant feedback control response in amplitude control regardless of the driving state. Variable setting. Specifically, the gains Kpλ and Kiλ are set to be larger as the electrical angular velocity ω is higher or the voltage phase φ is more advanced.

  In order to maintain the responsiveness constant in amplitude control, for example, the λ-axis command current Iλ * is stepped in a state where the command voltage setting unit 30f and the speed multiplication unit 30h are removed from the configuration of FIG. This means that the time constant of the λ-axis current Iλr when it is changed is maintained at the target time. Further, the gains Kpλ and Kiλ may be calculated as follows, for example. Specifically, in the case of a proportional gain as an example, the inverse of the gain shown in FIG. 9 is mapped as a correction gain. Then, a proportional gain is set by multiplying a separately set basic gain by a correction gain.

  Next, the gain used in the phase setting unit 30e will be described.

  The following equation (eq16) is derived by solving the above equation (eq2) for d and q-axis currents Id and Iq assuming a steady state and ignoring the influence of the armature winding resistance R.

On the other hand, torque τ of motor generator 10 is represented by the following equation (eq17), where the number of pole pairs is “Pn”.

From the above equations (eq16), (eq17), and (eq3), a relational expression between the voltage phase φ and the torque τ is derived as the following equation (eq18).

According to the above equation (eq18), as shown in FIG. 10, the torque τ changes according to the voltage amplitude Vn, the voltage phase φ, and the electrical angular velocity ω. Therefore, in order to improve the phase control responsiveness in all the drive regions of the motor generator 10, it is required to variably set the gain used in the feedback control in accordance with the drive state. In the present embodiment, in order to keep the phase control responsiveness constant regardless of the driving state, the phase gain setting unit 34b is based on the voltage phase φ, the electrical angular velocity ω, and the voltage amplitude “Vn + ΔV”, and the proportional gain Kpφ and the integral The gain Kiφ is variably set. Specifically, the gains Kpφ and Kiφ are set larger as the electrical angular velocity ω is higher, the voltage phase φ is retarded, or the voltage amplitude “Vn + ΔV” is smaller.

  In the phase control, maintaining the responsiveness constant means, for example, maintaining the time constant for the estimated torque Te when the target torque Trq * is changed stepwise at the target time. The gains Kpφ and Kiφ may be calculated as follows, for example. Specifically, the slope of torque τ with respect to voltage phase φ under each driving condition of motor generator 10 shown in FIG. 10 is mapped as a correction gain. Then, the gains Kpφ and Kiφ are set by multiplying the separately set basic gain by the correction gain.

  According to the above configuration in which the gain used in the feedback control is variably set, the responsiveness of each of the amplitude control and the phase control can be maintained high regardless of the driving state of the motor generator 10. However, since the responsiveness of the amplitude control can be increased, torque fluctuation (torque shock) is increased when so-called voltage saturation occurs when the modulation factor Mr reaches the second specified value M2. In order to cope with such a problem, in this embodiment, a responsiveness setting unit 34c is provided. Hereinafter, the processing of the responsiveness setting unit 34c will be described with reference to FIGS.

  The responsiveness setting unit 34c calculates the modulation rate Mr based on the voltage amplitude “Vn + ΔV” and the input voltage VINV. Then, the multiplication coefficient Kt is variably set based on the calculated modulation factor Mr. Here, the multiplication coefficient Kt is a coefficient for multiplying the feedback gain of only the amplitude control among the amplitude control and the phase control. In the present embodiment, the multiplication coefficient Kt is calculated using the following equation (eq19).

In the above equation (eq19), “Mb” represents a reference modulation rate. In the present embodiment, the reference modulation factor Mb is set to a value (1.2) that is higher than the first specified value M1 and less than the second specified value M2. In the present embodiment, the first predetermined value Rs1 is set to 1, and the second predetermined value Rs2 is set to a value less than 1 (0.3). In the present embodiment, “0 ≦ Mr <Mb” corresponds to the first range, and “Mr ≧ Mb” corresponds to the second range.

  According to the above equation (eq19), the multiplication coefficient Kt monotonously decreases in “Mb ≦ Mr ≦ M2”. Further, the multiplication coefficient Kt is set to a value higher than 0 in “Mr> M2”. This is a setting for ensuring control stability when the modulation factor Mr is set to an excessively large value.

  The multiplication coefficient Kt set by the responsiveness setting unit 34c is multiplied by the gains Kpλ and Kiλ output from the amplitude gain setting unit 34a. Then, the amplitude correction amount ΔV is calculated by the amplitude correction amount calculation unit 32g using the gains Kpλ and Kiλ multiplied by the multiplication coefficient Kt. Thereby, the responsiveness of the amplitude control can be lowered at “Mr ≧ Mb” rather than “0 ≦ Mr <Mb”. Incidentally, the feedback gains Kpφ and Kiφ in the phase control are not multiplied by the multiplication coefficient Kt. For this reason, according to the above configuration, in “Mb ≦ Mr ≦ M2”, the higher the modulation rate Mr, the lower the responsiveness of the amplitude control with respect to the responsiveness of the phase control.

  Next, the effect of including the responsiveness setting unit 34c will be described with reference to FIGS. First, FIG. 12 shows the transition of torque or the like in the torque step response for each of the present embodiment and related technologies. Here, the related technology is a configuration in which the responsiveness setting unit 34c is removed from the present embodiment. In FIG. 12, in each of the present embodiment and related technologies, the vertical scales are the same and the horizontal scales are also the same.

  In the present embodiment, after the target torque Trq * changes suddenly, the modulation rate Mr exceeds the reference modulation rate Mb (1.2), thereby reducing the responsiveness of the amplitude control. For this reason, as the modulation factor Mr gradually approaches the second specified value M2, the rate of increase of the modulation factor Mr (voltage amplitude “Vn + ΔV”) decreases. Thereby, voltage saturation can be performed in a state where the increasing speed of the voltage amplitude “Vn + ΔV” is low, and torque shock can be reduced. On the other hand, in the related art, since the amplitude gain setting unit 34a is provided, the responsiveness of the amplitude control is made constant regardless of the driving state. For this reason, even when the modulation factor Mr gradually approaches the second specified value M2, the increasing speed of the voltage amplitude “Vn + ΔV” does not decrease. This increases the torque shock when voltage saturation occurs.

  FIG. 13 shows transitions of d, q-axis current and the like in the torque step response for each of the present embodiment and related technologies. In FIG. 13, in each of the present embodiment and related technologies, the vertical scales are the same and the horizontal scales are also the same.

  In the present embodiment, since the modulation factor Mr exceeds the reference modulation factor Mb, the responsiveness of the amplitude control is lowered, and therefore fluctuations in the d and q axis currents (current shock) when voltage saturation occurs can be suitably suppressed. . On the other hand, in the related technology, a large current shock occurs.

  According to the embodiment described above, the following effects can be obtained.

  (1) The multiplication coefficient Kt (<1) in “Mr> Mb” is made lower than the multiplication coefficient Kt in “0 ≦ Mr ≦ Mb”. Therefore, torque controllability and current controllability can be maintained high in a situation where the modulation factor Mr has a margin with respect to the second specified value M2, and torque controllability and current controllability are reduced when voltage saturation occurs. It can suppress suitably. In particular, in the present embodiment, decreasing the multiplication coefficient Kt as the modulation factor Mr becomes higher in “Mr> Mb” maintains the torque controllability and the like, and decreases the torque controllability when the voltage is saturated. It greatly contributes to the coexistence of suppression.

  (2) Amplitude control using the λ-axis current Iλr was performed. In this case, since the responsiveness of amplitude control can be increased, torque shock when voltage saturation occurs tends to increase. For this reason, in this embodiment in which a torque shock is likely to increase, the merit provided with the responsiveness setting unit 34c is great.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In this embodiment, the method for reducing the response of amplitude control is changed.

  FIG. 14 shows a block diagram of torque control according to the present embodiment. In FIG. 14, the same processes as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  In the present embodiment, the control device 30 includes a low-pass filter 36a and a responsiveness setting unit 36b in addition to the amplitude gain setting unit 34a and the phase gain setting unit 34b. The voltage amplitude “Vn + ΔV” output from the correction unit 30 i is input to the low-pass filter 36 a. The voltage amplitude “Vn + ΔV” subjected to the low-pass filter process in the low-pass filter 36a is input to the operation signal generation unit 30j.

  In the present embodiment, the responsiveness setting unit 36b variably sets the multiplication coefficient Kt by the same method as the responsiveness setting unit 34c of the first embodiment. Specifically, the responsiveness setting unit 36b calculates the modulation factor Mr based on the voltage amplitude “Vn + ΔV” subjected to the low-pass filter process and the input voltage VINV. Based on the calculated modulation factor Mr, the multiplication coefficient Kt is set as shown in FIG.

  The multiplication coefficient Kt set by the responsiveness setting unit 36b is multiplied by the cut-off frequency fc of the low-pass filter 36a. For this reason, the cutoff frequency is set to “Rs1 × fc = fc” in “0 ≦ Mr <Mb”. Further, in “Mb ≦ Mr ≦ M2,” the cutoff frequency is set lower as the modulation factor Mr is higher. That is, in “Mb ≦ Mr ≦ M2,” the cutoff frequency monotonously decreases as the modulation rate Mr increases. Further, the cutoff frequency is set to “Rs2 × fc = 0.3fc” in “Mr> M2”. Since the cutoff frequency has a positive correlation with the reciprocal of the time constant of the low-pass filter 36a, the time constant increases as the cutoff frequency decreases. Thereby, the responsiveness of the amplitude control can be lowered at “Mr> Mb” rather than “0 ≦ Mr ≦ Mb”.

  Next, the effect of including the low-pass filter 36a and the response setting unit 36b will be described with reference to FIGS. First, FIG. 15 shows the transition of torque or the like in the torque step response for each of the present embodiment and related technologies. FIG. 15 corresponds to FIG. As illustrated, according to the present embodiment, the torque shock can be reduced as compared with the related art.

  FIG. 16 shows the transition of d, q-axis current and the like in the torque step response for each of the present embodiment and related technology. FIG. 16 corresponds to FIG. As illustrated, according to the present embodiment, the current shock can be reduced as compared with the related art.

  According to the present embodiment described above, it is possible to obtain the same effect as that obtained in the first embodiment.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, as shown in FIG. 17, the setting method of the multiplication coefficient Kt in the responsiveness setting unit 34c is changed. Specifically, the multiplication coefficient Kt is uniformly set to the second predetermined value Rs2 in the range where the modulation factor Mr exceeds the reference modulation factor Mb.

  FIG. 18 shows the transition of torque or the like in the torque step response for each of the present embodiment and related technology. FIG. 18 corresponds to FIG. As illustrated, according to the present embodiment, the torque shock can be reduced as compared with the related art. The reason why the rate of increase of the modulation factor Mr greatly decreases at time t1 after the sudden change in the target torque Trq * is that the multiplication coefficient Kt is greatly changed from 1 to 0.3.

  According to the present embodiment described above, it is possible to obtain an effect according to the effect obtained in the first embodiment.

(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In this embodiment, the design method of the correction amount calculation unit is changed. Specifically, the λ axis is derived not in the polar coordinate system but in the biaxial orthogonal coordinate system.

  When the condition that the transient phenomenon is ignored in the above equation (eq1), the following equation (eq20) is derived.

In the above equation (eq20), the following equation (eq21) is obtained from the relationship between the minute change amounts ΔId and ΔIq of the d and q axis currents Id and Iq when the d and q axis voltages Vd and Vq slightly change by ΔVd and ΔVq. Led.

Here, as shown in FIG. 19, a coordinate system including an l-axis extending from the origin 0 in a direction parallel to the voltage vector Vnvt and a p-axis extending from the origin 0 in a direction orthogonal to the voltage vector Vnvt is defined as a pl coordinate system. To do. Similarly to the λ axis, the pl coordinate system also changes as the driving state of the motor generator 10 changes. Here, the relationship between the voltage vector Vnvt converted from the dq coordinate system to the pl coordinate system and the current vector Invt converted from the dq coordinate system to the λo coordinate system different from the pl coordinate system is derived. The pl coordinate system is a coordinate system obtained by rotating the dq coordinate system about the origin 0 counterclockwise by an angle η (= φ−π / 2). The λo coordinate system is a coordinate system obtained by rotating the dq coordinate system about the origin 0 counterclockwise by an angle λ. For this reason, if the minute change amounts of the p and l axis voltages on the p and l axes corresponding to ΔVd and ΔVq are ΔVp and ΔVl, the following equation (eq22) is derived from the above equation (eq21).

Solving the above equation (eq22) for current leads to the following equation (eq23).

Here, increasing / decreasing ΔVl is equivalent to increasing / decreasing the amplitude of the voltage vector Vnvt. Further, increasing or decreasing ΔVp is equivalent to increasing or decreasing the phase of voltage vector Vnvt. Therefore, similarly to the first embodiment, when controlling Vp to control torque and controlling Vl to control current, only the l-axis voltage Vl is not affected by the change in Vp. To control, on the right side of the above equation (eq23), an angle λ is set such that “Rc−ω · Loλ = 0”, and the l-axis voltage Vl is controlled using the λ-axis current Iλ. That's fine. Alternatively, in order to control only the l-axis voltage Vl without being affected by the change in Vp, an angle λ is set on the right side of the above equation (eq23) such that “Rs−ω · Lλ = 0”. The l-axis voltage Vl may be controlled using the o-axis current Io.

  Here, if the electrical angular velocity ω is sufficiently high, “Rc << ω · Loλ” and “Rs << ω · Lλ” are established. In order to control the l-axis voltage Vl using the λ-axis current Iλ, “−ω · Loλ = 0”, that is, “Loλ = 0” is sufficient. For this reason, the following formula (eq24) is derived.

On the other hand, in order to control the l-axis voltage Vl using the o-axis current Io, “−ω · Lλ = 0”, that is, “Lλ = 0” is sufficient. For this reason, the following formula (eq25) is derived.

The above equation (eq25) is obtained by rotating the above equation (eq24) by “−π / 2”, and the λ axis in the above equation (eq24) coincides with the o axis in the above equation (eq25). For this reason, in order to control the l-axis voltage Vl, the effect is the same when the λ-axis current Iλ is used and when the o-axis current Io is used. The derived angle λ coincides with the angle represented by the above equation (eq8) that can suppress interference between the amplitude control and the phase control.

  FIG. 20 shows a block diagram of torque control according to the present embodiment. In FIG. 20, the same processes as those shown in FIG. 2 are given the same reference numerals for the sake of convenience.

  Based on the torque deviation ΔT, the phase setting unit 30k calculates the p-axis voltage Vp as an operation amount for performing feedback control of the estimated torque Te to the target torque Trq *. Specifically, the p-axis voltage Vp is calculated by proportional-integral control with the torque deviation ΔT as an input. The p-axis voltage Vp corresponds to a phase operation amount that is a p-axis component of the voltage vector Vnvt, and is a voltage component according to the voltage phase φ.

  The command voltage setting unit 301 receives the target torque Trq * as an input and calculates a standardized voltage amplitude “Vl / ω”. In the present embodiment, the normalized voltage amplitude “Vl / ω” is a value obtained by dividing the l-axis voltage Vl by the electrical angular velocity ω. Note that the normalized voltage amplitude may be calculated using a map in which the target torque Trq * and the normalized voltage amplitude are related.

  The speed multiplication unit 30h calculates the l-axis voltage Vl by multiplying the normalized voltage amplitude “Vl / ω” by the electrical angular speed ω. The l-axis voltage Vl is an operation amount for performing feedforward control of the torque of the motor generator 10 to the target torque Trq *.

  The amplitude correction amount calculation unit 32i that constitutes the correction amount calculation unit 32 is based on the current deviation ΔIλ, and uses the correction amount of the l-axis voltage Vl as an operation amount for feedback control of the λ-axis current Iλr to the λ-axis command current Iλ *. ΔVl is calculated. Specifically, the correction amount ΔVl is calculated by proportional-integral control using the current deviation ΔIλ as an input. The correction amount ΔVl corresponds to an amplitude operation amount that is an l-axis component of the voltage vector Vnvt, and is a voltage component according to the voltage amplitude Vn. Here, in the amplitude correction amount calculation unit 32i, the proportional and integral gains used in the feedback control are set by the amplitude and phase gain setting units 34a and 34b as in the first embodiment. At this time, the responsiveness setting unit 34c multiplies the proportional and integral gains by the multiplication coefficient Kt as in the first embodiment.

  The correction unit 30i corrects the l-axis voltage Vl by adding the correction amount ΔVl calculated by the correction amount calculation unit 32 to the l-axis voltage Vl output from the speed multiplication unit 30h.

  The operation signal generation unit 30m generates operation signals gUp to gWn based on the l-axis voltage “Vl + ΔVl” output from the correction unit 30i, the p-axis voltage Vp output from the phase setting unit 30k, and the input voltage VINV. And output to the drive circuits DrUp to DrWn. In the present embodiment, the operation signal is generated as follows.

  The operation signal generation unit 30m sets the l-axis voltage “Vl + ΔVl” output from the correction unit 30i as the amplitude Vn of the voltage vector. Further, the operation signal generation unit 30m sets the voltage phase φ based on the relationship “φ = η + π / 2”. In this embodiment, the angle η is subjected to low-pass filter processing in order to remove a noise component of the calculated value of the angle η formed by the d axis and the p axis. For this reason, the voltage phase φ is set by the following equation (eq26).

In the above equation (eq26), the first term on the right side corresponds to the change in the angle η formed by the d-axis and the p-axis during the period from the previous processing cycle to the current processing cycle. In order to calculate the voltage phase φ in the next processing cycle, the angle η of the second term on the right-hand side of the above equation (eq26) is “π / 2” from the voltage phase φ calculated in the current processing cycle for each processing cycle. ”Is updated to the subtracted value. That is, the previously updated angle η is used as the angle η of the second term on the right side of the above equation (eq26). The operation signal generation unit 30m calculates a three-phase command voltage based on the set voltage amplitude Vn and voltage phase φ, and performs an operation signal by sinusoidal PWM control based on a magnitude comparison between the calculated three-phase command voltage and the carrier signal. Generate g ¥ #.

  Incidentally, in the present embodiment, the λ-axis setting unit 32h constituting the correction amount calculation unit 32 includes the p-axis voltage Vp output from the phase setting unit 30k and the l-axis voltage “Vl + ΔVl” output from the correction unit 30i. Based on the above, the angle λ formed by the d-axis and the λ-axis is calculated. Specifically, first, the voltage phase φ is calculated based on the above equation (eq26) with p and l-axis voltages as inputs. Then, using the calculated voltage phase φ as an input, the angle λ is calculated based on the above equation (eq24).

  The torque control described above is such that the voltage as the operation amount is operated in the pl coordinate system, and has the estimated torque Te and the λ-axis current Iλr as the control amounts. Also according to the present embodiment described above, the same effect as that obtained in the first embodiment can be obtained.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the first embodiment, the feedback gains Kpλ and Kiλ used in the amplitude correction amount calculation unit 32g may be variably set based on either the electrical angular velocity ω or the voltage phase φ. In the second embodiment, the feedback gains Kpλ and Kiλ may be variably set based on the current and torque flowing through the motor generator 10. This will be described below.

  Assuming a steady state and ignoring the influence of the armature winding resistance R, the d and q axis voltages Vd and Vq are expressed by the following equation (eq27).

In this case, the voltage phase φ is expressed by the following equation (eq28).

When the minimum current / maximum torque control is performed, when the target torque Trq * is determined, the currents Id and Iq are uniquely determined from the above equation (eq17). Therefore, when the target torque Trq * is determined, the voltage phase φ is also determined from the above equation (eq28). Therefore, each of the feedback gains Kpλ and Kiλ can be set by using the target torque Trq * instead of the voltage phase φ and using the electrical angular velocity ω. Here, the smaller the target torque Trq *, the larger the feedback gains Kpλ and Kiλ may be set.

  For example, when the SPMSM that is a non-salient pole machine is used as the motor generator 10, “Ld = Lq”. Therefore, according to the above equations (eq17) and (eq28), each feedback gain can be set based on the q-axis current Iq and the electrical angular velocity ω.

  In the first embodiment, the feedback gains Kpφ and Kiφ used in the phase setting unit 30e may be variably set based on any one or two of the electrical angular velocity ω, the voltage phase φ, and the voltage amplitude Vn. In the second embodiment, the feedback gains Kpφ and Kiφ may be variably set based on the current and torque flowing through the motor generator 10. Specifically, the feedback gains Kpφ and Kiφ may be set larger as the current is smaller or the torque is smaller. For the setting method based on current and torque, refer to Japanese Patent Application Laid-Open No. 2012-85485.

  The λ-axis derivation method is not limited to the one exemplified in the first embodiment. For example, it may be described below.

  Based on the above equation (eq6), each current change amount ΔIλ, ΔIo in the λo coordinate system obtained by rotating the dq coordinate system by λ is represented by the following equation (eq29).

In the above equation (eq29), if the term relating the λ-axis current change amount ΔIλ and the minute change amount Δφ is 0, the λ-axis current Iλ is not affected by the voltage phase change. From this, the following formula (eq30) is derived.

Solving the above equation (eq30) for λ leads to the following equation (eq31).

The above equation (eq31) represents that the coordinate axis in the direction in which the current vector change when the current voltage phase φ changes slightly can be set as the λ axis. When setting the λ axis from the above equation (eq31), for example, the sign of each gain Kpλ, Kiλ of the amplitude correction amount calculation unit 32g is inverted from the sign of each gain Kpλ, Kiλ in the first embodiment. It should be.

  In each of the above embodiments, the feedback control in at least one of the phase setting unit and the amplitude correction amount calculation unit may be performed only by integration control, for example. For example, the feedback control is not limited to proportional control or integral control, but may be differential control.

  In each of the above embodiments, the command voltage setting unit 30f and the speed multiplication unit 30h may be removed from the control device 30. Even in this case, the gains Kpλ and Kiλ in the amplitude correction amount calculation unit can be set large by using the λ-axis current Iλr. For this reason, high torque controllability can be maintained.

  In each of the above embodiments, a coordinate axis extending in a direction slightly deviated from a direction orthogonal to the direction of change of the current vector Invt when the voltage phase φ is slightly changed may be set as a non-interacting coordinate axis (λ axis). Even in this case, it is possible to obtain the effects according to the effects of the above embodiments.

  The method for setting the non-interacting coordinate axis (λ axis) is not limited to those exemplified in the above embodiments. For example, the armature winding resistance R may be considered in the calculation of the λ axis under the low rotation conditions of the motor generator 10 where the relationship of “R << ω · Ld” and “R << ω · Lq” is not established. . In this case, the above equation (eq6) is represented as the following equation (eq32), and the above equation (eq7) is represented as the following equation (eq33). Therefore, the angle λ can be calculated by substituting the change direction α of the current vector Invt calculated from the following equations (eq32) and (eq33) into the above equation (eq8).

For example, when the difference between the d-axis inductance Ld and the q-axis inductance Lq is small, “Ld / Lq” is a value close to 1. Therefore, the λ axis can be set based only on the voltage phase φ. When motor generator 10 is SPMSM, “Ld / Lq” is 1, so that the λ axis can be set based only on voltage phase φ.

  The operation signal generation unit is not limited to one that generates an operation signal by triangular wave comparison PWM control, and may be, for example, described below. For example, the control device 30 includes storage means (for example, a memory such as a ROM) in which a line voltage pattern for realizing the voltage amplitude Vn is stored in advance. In such a configuration, the line voltage pattern stored in the storage means is converted into a pulse pattern for the gate of each switching element, and the operation signal is generated by setting the output timing of the converted pulse pattern based on the voltage phase φ. To do. As for the method for converting the line voltage pattern to the pulse pattern for the gate, for example, “High-efficiency motor control by harmonic modulation type pulse-saving drive: Kimihisa Furukawa et al., 7 others, 2010 IEEJ Industrial Application Division Conference 1-134, pp. I-627 to I-632 ”.

  Further, the operation signal generation unit is not limited to one that calculates a sine wave three-phase command voltage whose phases are shifted from each other by 120 degrees in electrical angle based on the voltage amplitude and phase. For example, a method of superposing a third-order high frequency on a sine wave may be used. This is described in, for example, “Theory and Design of AC Servo Systems: Hidehiko Sugimoto, General Electronic Publishing Company”.

  -The torque value used for phase control is not limited to the estimated value. For example, the control system may include torque detection means (for example, a torque measuring device) that detects the torque of the motor generator 10, and the detected torque may be used for phase control.

  -The motor output used for phase control is not limited to that related to torque. For example, an o-axis current that is a direction orthogonal to the λ-axis may be used. In this case, phase control may be performed so that the o-axis current follows the command current. Further, for example, when the rotational speed control of the motor generator 10 exists as a higher-level control system of the torque control, the configuration for controlling the estimated torque with the target torque that is the output of the speed control as an input is omitted. In such a configuration, the phase control may be directly performed based on an error between the actual rotational speed as the control amount and the command rotational speed. At this time, the amplitude control is performed by the λ-axis current Iλr, and the λ-axis command current Iλ * is set using a map based on the estimated torque. Even with such a configuration, it is possible to obtain the same effect as that of the first embodiment.

  The λ-axis current used for amplitude control may be directly calculated from the phase current without being calculated via the d and q-axis currents, for example, as in the above equation (eq10). Specifically, for example, the λ-axis current may be directly calculated using the phase current, the electrical angle θ, and the angle λ. The λ-axis command current Iλ * may be calculated using a map related to the d and q-axis command currents Id * and Iq *, or using a map related to the target torque Trq *. It may be calculated.

  In the first and second embodiments, as the modulation factor Mr is higher with “Mb ≦ Mr <M2”, the multiplication factor Kt is not limited to be continuously set lower, but is set to be lower stepwise. May be.

  The configuration using the low-pass filter of the second embodiment may be applied to the fourth embodiment, or the method for setting the multiplication coefficient Kt of the third embodiment may be applied.

  As shown in the first to third embodiments, the responsiveness is not limited to two steps using the first and second predetermined values Rs1 and Rs2, but is changed to three or more steps. May be. Specifically, for example, by using the first to third predetermined values Rs1 to Rs3 (Rs1> Rs2> Rs3), the responsiveness can be changed in three stages.

  The amplitude response reducing means (gain reducing means) explained in FIGS. 8 and 11 of the first embodiment, and the amplitude response reducing means (filter cutoff frequency reducing means) explained in FIG. 14 of the second embodiment. May be used in combination. Specifically, for example, in the configuration of FIG. 2 of the first embodiment, the processing unit further includes a processing unit that performs a filter process (low-pass filter process) on the voltage amplitude Vn output from the speed multiplication unit 30h. As described in the second embodiment, the frequency may be variably set according to the modulation rate Mr. In addition, the output value of the said process part is input into the correction | amendment part 30i.

In each of the above embodiments, the region in which the responsiveness is changed is defined by the first range and the second range, but is not limited thereto. For example, the multiplication coefficient Kt in the first embodiment may be changed with a continuous function in the total modulation rate Mr as shown in FIG. In this case, it is not necessary to define the region in which the responsiveness is changed between the first range and the second range. Here, in FIG. 21, the responsiveness is expressed by a function represented by the following equation (eq34). When the following equation (eq34) is used, the multiplication coefficient Kt in the modulation factor Mr between 0 and the second specified value M2 tends to increase slightly, but there is no problem because the increase amount is small.
FIG. 22 shows changes in torque and the like between the present embodiment and the related technology when the above equation (eq34) is used. FIG. 22 corresponds to FIG. As shown in the drawing, the same effect as that obtained in the first embodiment can be obtained by the configuration using the above equation (eq34).

  The configuration for reducing the response of the amplitude control when “Mr ≧ Mb” is not based on the assumption of the amplitude control using the λ-axis current. For example, when the target torque Trq * is low, the voltage phase φ is set near “π / 2” as shown in FIG. For this reason, the angle λ output from the λ-axis setting unit is close to zero. Under such circumstances, since the λ-axis current is substantially not used for amplitude control, it is not necessary to assume amplitude control using the λ-axis current. That is, the amplitude control is performed using the d-axis current that is not the non-interference current but the angle λ = 0.

  -"Rotating machine" is not limited to IPMSM but may be SPMSM or a wound field type synchronous machine. Here, for example, when the SPMSM is adopted in the first embodiment, the torque of the rotating machine is determined by the q-axis current, and therefore the control amount related to the torque may be the q-axis current instead of the torque. Further, the “rotating machine” is not limited to being used as an in-vehicle main machine, but may be used as an in-vehicle auxiliary machine such as an electric power steering device or an electric motor constituting an electric compressor for air conditioning. In addition, the “rotating machine” is not limited to a vehicle-mounted type.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 20 ... Inverter, 30 ... Control apparatus, SUp-SWn ... Switching element.

Claims (12)

Applied to a rotating machine (10) electrically connected to a power conversion circuit (20) having switching elements (SUp to SWn);
As an operation amount for feedback control of the control amount of the rotating machine to its target value, the phase of the output voltage vector of the power conversion circuit in the rotating coordinate system of the rotating machine, or the orthogonal biaxial component of the output voltage vector Phase setting means (30e; 30k) for setting a phase operation amount which is a component in accordance with the phase,
As an operation amount for feedback-controlling the current flowing through the rotating machine to a command current according to the target value, the amplitude of the output voltage vector or a component according to the amplitude of the orthogonal two-axis components of the output voltage vector An amplitude setting means (32g; 32i) for setting an amplitude operation amount which is
Operation means (30j) for operating the switching element based on the phase operation amount set by the phase setting means and the amplitude operation amount set by the amplitude setting means to control the control amount to the target value. )When,
Before the modulation factor determined by the fundamental wave component of the voltage output from the power conversion circuit to the rotating machine and the input voltage of the power conversion circuit reaches its upper limit value, An apparatus for controlling a rotating machine, comprising: amplitude response lowering means (34c; 36b) for reducing the response of the current flowing through the rotating machine in advance according to the modulation factor.   The amplitude response lowering means has a range in which the modulation rate is less than a specified value higher than 0 as a first range, a range in which the modulation rate includes the specified value to an upper limit value as a second range, and the phase setting means The responsiveness of the current flowing through the rotating machine to the command current by the amplitude setting means is reduced without lowering the responsiveness of the control amount to the target value by the amplitude setting means in the second range than the first range. The control device for a rotating machine according to claim 1, wherein the controller is lowered in the second range than the range. The amplitude setting means sets the amplitude operation amount by feedback control based on a deviation between a current flowing through the rotating machine and the command current,
Regardless of the value of the first setting parameter including at least one of the rotational speed of the rotating machine, the torque of the rotating machine, the current flowing through the rotating machine, and the phase, the current flowing through the rotating machine with respect to the command current An amplitude gain setting means (34a) for variably setting an amplitude feedback gain used in feedback control by the amplitude setting means so that the responsiveness is constant;
3. The rotation according to claim 2, wherein the amplitude response lowering unit increases the degree of decrease in the amplitude operation amount calculated using the amplitude feedback gain when the modulation rate is higher in the second range than when the modulation rate is low. Machine control device. The amplitude setting means sets the amplitude operation amount by feedback control based on a deviation between a current flowing through the rotating machine and the command current,
Regardless of the value of the first setting parameter including at least one of the rotational speed of the rotating machine, the torque of the rotating machine, the current flowing through the rotating machine, and the phase, the current flowing through the rotating machine with respect to the command current An amplitude gain setting means (34a) for variably setting an amplitude feedback gain used in feedback control by the amplitude setting means so that the responsiveness is constant;
Filter means (36a) for performing low-pass filter processing on the amplitude manipulated variable calculated using the amplitude feedback gain;
The operation means operates the switching element based on the amplitude operation amount subjected to low-pass filter processing by the filter means,
The control apparatus for a rotating machine according to claim 2 or 3, wherein the amplitude response lowering means (36b) increases a time constant in the low-pass filter process when the modulation factor is higher in the second range than when the modulation factor is low. The phase setting means sets the phase operation amount by feedback control based on a deviation between the control amount and the target value,
Regardless of the value of the second setting parameter including at least one of the rotational speed of the rotating machine, the torque of the rotating machine, the current flowing through the rotating machine, the phase, and the output voltage of the power conversion circuit, the target value 5. The rotating machine according to claim 3, further comprising phase gain setting means (34 b) for variably setting a phase feedback gain used in feedback control by the phase setting means so that the responsiveness of the control amount with respect to is constant. Control device.   In the second range, the amplitude response lowering unit increases the control over the target value by the amplitude setting unit with respect to the responsiveness of the control amount to the target value by the phase setting unit as the modulation rate is higher. 6. The control device for a rotating machine according to claim 5, wherein the responsiveness of the quantity is lowered.   6. The control device for a rotating machine according to claim 2, wherein the amplitude response lowering unit lowers the response as the modulation rate is higher in the second range. 7.   The amplitude response lowering means is configured to reduce a current flowing through the rotating machine with respect to the command current by the amplitude setting means so as to reduce the increase speed of the amplitude as the modulation rate approaches the upper limit value in the second range. The control device for a rotating machine according to any one of claims 2 to 7, wherein the responsiveness is lowered in the second range. In the rotating coordinate system, a non-interfering coordinate axis is a coordinate axis in which a change in the current vector flowing through the rotating machine with respect to a change in the phase manipulated variable is made non-interfering, and non-interference is a non-interfering coordinate axis direction component of the current vector. A non-interacting current calculating means (32d) for calculating
The amplitude setting unit sets the amplitude operation amount as an operation amount for feedback-controlling the non-interacting current calculated by the non-interacting current calculating unit to the command current. The control device for a rotating machine according to Item 1.   10. The rotating machine control device according to claim 9, wherein the non-interacting coordinate axis is a coordinate axis in a direction perpendicular to a direction in which the current vector changes when the current phase operation amount is slightly changed.   10. The rotating machine control device according to claim 9, wherein the non-interacting coordinate axis is a coordinate axis in a direction in which a change amount of the current vector becomes zero when the current phase operation amount is slightly changed. Torque estimation means (30b) for estimating the torque of the rotating machine based on the current flowing through the rotating machine,
The control device for a rotating machine according to any one of claims 1 to 11, wherein the control amount is a torque estimated by the torque estimating means.