JP6361569B2 - Control device for rotating electrical machine - Google Patents

Description

  The present invention relates to a rotating electrical machine control apparatus which is applied to a rotating electrical machine electrically connected to a power conversion circuit having a switching element and controls a control amount of the rotating electrical machine to a command value by operation of the switching element.

  As this type of control device, as seen in Patent Document 1 below, in order to control the control amount of the rotating electrical machine to its command value, the phase of the output voltage vector of the power conversion circuit and the amplitude of the output voltage vector ( Norm) is known. Here, the amplitude of the output voltage vector is set using a map in which the rotational speed and amplitude of the rotating electrical machine are associated with each other. For this reason, when the accuracy of the map is low, there is a concern that the controllability of the control amount is lowered.

  Therefore, in the control device described in Patent Document 1 below, the amplitude of the output voltage vector is corrected based on the d-axis current flowing through the rotating electrical machine. As a result, even if the accuracy of the map is low, a reduction in controllability is avoided.

JP 2012-23943 A

  Here, even when the inventor uses the correction method described in Patent Document 1, the present inventors have faced a situation in which the control accuracy of the control amount of the rotating electrical machine decreases.

  The present invention has been made to solve the above-described problems, and an object of the present invention is to provide a control device for a rotating electrical machine that can improve the controllability of the control amount of the rotating electrical machine.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

  The present invention is applied to a rotating electrical machine (10) electrically connected to a power conversion circuit (20) having switching elements (Sup to Swn), and feedback control the control amount of the rotating electrical machine to its command value. The phase manipulated variable is either the phase of the output voltage vector of the power conversion circuit in the rotating coordinate system of the rotating electrical machine or a component in accordance with the phase among the orthogonal two-axis components of the output voltage vector. A phase setting unit (30d; 40b; 30n) that sets the output voltage vector amplitude or the output as an operation amount for feedback control of the current flowing through the rotating electrical machine to a command current corresponding to the command value An amplitude setting unit (32f; 32h) for setting an amplitude manipulated variable that is one of the components in accordance with the amplitude among the orthogonal two-axis components of the voltage vector; An operation unit for operating the switching element based on the phase operation amount set by the phase setting unit and the amplitude operation amount set by the amplitude setting unit to control the control amount to the command value. (30j; 30q), and the phase setting unit sets the phase operation amount so as to suppress a change in the control amount accompanying a change in the amplitude operation amount.

  In the present invention, the control based on the phase operation amount for feedback control of the control amount of the rotating electrical machine to the command value is referred to as phase control, and the current flowing through the rotating electrical machine is feedback controlled to the command current corresponding to the command value. Control based on the amplitude manipulated variable for the purpose is referred to as amplitude control.

  The present inventor has found that the factor that decreases the control accuracy of the control amount is interference from the amplitude control based on the amplitude operation amount to the phase control based on the phase operation amount. Therefore, in the present invention, the phase operation amount is set by the phase setting unit so as to suppress the change in the control amount accompanying the change in the amplitude operation amount. Thereby, interference from amplitude control to phase control can be suppressed. Therefore, the controllability of the control amount can be improved.

  Here, the present invention can be embodied as follows, for example. Specifically, in the rotating coordinate system, a coordinate axis in which a change in the current vector flowing in the rotating electrical machine with respect to a change in the phase operation amount is made non-interfering is a phase non-interfering axis, and the phase non-interfering axis direction component of the current vector is A phase current calculation unit (32d) for calculating a phase non-interference current is further provided, and the amplitude setting unit performs an operation for feedback control of the phase non-interference current calculated by the phase current calculation unit to the command current. The amplitude operation amount is set as the amount.

  In the above invention, in the rotating coordinate system, the coordinate axis in which the change of the current vector flowing to the rotating electrical machine with respect to the change of the phase operation amount is made non-interfering is used as the phase non-interfering axis, The current is calculated. Then, the amplitude operation amount is set by the amplitude setting unit as an operation amount for feedback control of the phase non-interference current to the command current. Thereby, interference from phase control to amplitude control can be suppressed. In addition to the suppression of this interference, in the above invention, the phase operation amount is set by the phase setting unit so as to suppress the change in the control amount accompanying the change in the amplitude operation amount. The configuration in which the phase operation amount is set so as to suppress the interference from the phase control to the amplitude control and also suppress the change in the control amount accompanying the change in the amplitude operation amount is for reducing the loss of the rotating electrical machine. .

  That is, the control amount of the phase control changes with the change of the amplitude operation amount set in the amplitude control. Specifically, for example, the control amount increases as the amplitude operation amount increases. Thereby, the follow-up speed to the command value of the control amount becomes higher than the follow-up speed to the command current of the phase non-interference current. As a result, a weak magnetic flux current flows through the rotating electrical machine, and the loss of the rotating electrical machine increases. Therefore, in the above invention, the phase operation amount is set so as to suppress the interference from the phase control to the amplitude control and to suppress the change in the control amount accompanying the change in the amplitude operation amount. Thereby, interference from amplitude control to phase control can be suppressed, and loss of the rotating electrical machine can be reduced while improving controllability of the control amount.

1 is an overall configuration diagram of a motor control system according to a first embodiment. The block diagram of motor control. The figure for demonstrating the change of the current vector accompanying the change of an output voltage vector. The figure which shows the calculation method of the angle which d axis | shaft and (lambda) axis | shaft make. The figure for demonstrating (lambda) axis | shaft. The figure which shows the calculation method of (lambda) axis current. The block diagram of a gain setting part. The time chart which shows the relationship between a torque change speed and a lambda-axis current change speed. The figure which shows the relationship between the torque change speed in a dq coordinate system, and (lambda) axis current change speed. The figure which shows the effect of 1st Embodiment. The figure which shows the effect of 1st Embodiment. The block diagram of the motor control concerning 2nd Embodiment. The figure which shows the calculation method of the angle which d-axis and s-axis make. The figure for demonstrating the s-axis. The figure which shows the calculation method of s-axis current. The figure which shows the relationship between the dq coordinate system and pl coordinate system concerning 3rd Embodiment. The block diagram of motor control.

(First embodiment)
Hereinafter, a first embodiment in which a control device according to the present invention is applied to a vehicle (for example, an electric vehicle or a hybrid vehicle) including a three-phase rotating electrical machine as an in-vehicle main unit will be described with reference to the drawings.

  As shown in FIG. 1, the motor control system includes a motor generator 10, an inverter 20 as a “power conversion circuit”, and a control device 30 that controls the motor generator 10. In the present embodiment, the motor generator 10 is an in-vehicle main machine and is connected to drive wheels (not shown). In the present embodiment, an IPMSM that is a salient pole machine is used as the motor generator 10.

  The motor generator 10 is connected to a high voltage battery 21 as a DC power source via an inverter 20. The output voltage of the high voltage battery 21 is, for example, 100 V or more. A smoothing capacitor 22 that smoothes the input voltage of the inverter 20 is provided between the high voltage battery 21 and the inverter 20.

  The inverter 20 includes three sets of serially connected bodies of upper arm switches Sup, Svp, Swp and lower arm switches Sun, Svn, Swn. The U phase of the motor generator 10 is connected to a connection point between the U phase upper and lower arm switches Sup and Sun. The V phase of the motor generator 10 is connected to a connection point between the V phase upper and lower arm switches Svp and Svn. The W phase of the motor generator 10 is connected to a connection point between the W phase upper and lower arm switches Swp and Swn. Incidentally, in this embodiment, a voltage-controlled semiconductor switching element is used as each switch Sup, Sun, Svp, Svn, Swp, Swn, and more specifically, an IGBT is used. The free wheel diodes Dup, Dun, Dvp, Dvn, Dwp, Dwn are connected in antiparallel to the switches Sup, Sun, Svp, Svn, Swp, Swn.

  The motor control system further includes a phase current detection unit that detects a current of at least two phases among the phase currents flowing through the motor generator 10. In the present embodiment, the phase current detection unit includes a V-phase current sensor 23V that detects a current flowing in the V-phase of the motor generator 10 and a W-phase current sensor 23W that detects a current flowing in the W-phase. In addition, the motor control system detects the rotation angle (electrical angle θ) of the voltage sensor 24 as a voltage detection unit that detects the power supply voltage of the inverter 20 (DC voltage output from the high voltage battery 21) and the motor generator 10. A rotation angle sensor 25 (for example, a resolver) is provided as an angle detection unit.

  The control device 30 is mainly composed of a microcomputer, and operates the inverter 20 to feedback-control a control amount (torque in the present embodiment) related to the torque of the motor generator 10 to a command value (hereinafter, command torque Trq *). . Specifically, the control device 30 controls each switch Sup, Sun, Svp, Svn, Swp, Swn constituting the inverter 20 based on the detection values of the various sensors to turn on / off the switches Sup, Sun, Svp, Svn, Swp, Swn. , Gvn, gwp, and gwn are generated, and the generated operation signals gup, gun, gvp, gvn, gwp, and gwn are output to each drive circuit Dr (gate drive circuit) corresponding to each switch. Here, the upper arm side operation signals gup, gvp, gwp and the corresponding lower arm side operation signals gun, gvn, gwn are complementary to each other. That is, the upper arm switch and the corresponding lower arm switch are alternately turned on. The command torque Trq * is, for example, a control device provided outside the control device 30, and is output from a control device higher than the control device 30.

  Next, torque control of the motor generator 10 executed by the control device 30 will be described using FIG. This control includes phase control and amplitude control.

  First, phase control will be described. The two-phase conversion unit 30a is based on the V-phase current IV detected by the V-phase current sensor 23V, the W-phase current IW detected by the W-phase current sensor 23W, and the electrical angle θ detected by the rotation angle sensor 25. The U-phase current IU, V-phase current IV, and W-phase current IW in the three-phase fixed coordinate system are converted into the d-axis current Idr and the q-axis current Iqr in the two-phase rotational coordinate system (dq coordinate system).

  Torque estimator 30b calculates estimated torque Te of motor generator 10 based on d and q axis currents Idr and Iqr output from two-phase converter 30a. Here, the estimated torque Te may be calculated using a map in which the d-axis current Idr and the q-axis current Iqr are associated with the estimated torque Te, or may be calculated using a model formula.

  The torque deviation calculation unit 30c calculates the torque deviation ΔT by subtracting the estimated torque Te from the command torque Trq *. Note that the estimated torque Te input to the torque deviation calculating unit 30c may be subjected to a low-pass filter process for removing high frequency components.

  The phase setting unit 30d calculates the voltage phase φ as an operation amount for feedback control of the estimated torque Te to the command torque Trq * based on the torque deviation ΔT calculated by the torque deviation calculation unit 30c. In the present embodiment, the voltage phase φ is calculated by proportional-integral control with the torque deviation ΔT as an input. More specifically, the voltage phase φ is calculated as an addition value of the output value of the proportional controller having the torque deviation ΔT as an input and the output value of the integral controller having the torque deviation ΔT as an input. In the present embodiment, the voltage phase φ is defined with the positive direction of the d-axis as a reference, and the counterclockwise direction from this reference (the direction rotating from the positive direction of the d-axis to the positive direction of the q-axis) is defined as the positive direction. Has been. Therefore, when the estimated torque Te is insufficient with respect to the command torque Trq *, the voltage phase φ is increased (advanced), and when the estimated torque Te is excessive with respect to the command torque Trq *, the voltage The phase φ is decreased (retarded).

  The phase correction unit 30e calculates the corrected phase φc by subtracting the phase correction amount Δph calculated by the gain multiplication unit 34 from the voltage phase φ set by the phase setting unit 30d. The gain multiplier 34 will be described in detail later.

  Next, amplitude control will be described. The command voltage setting unit 30f receives the command torque Trq * and calculates a normalized voltage amplitude “Vn / ω”. Here, the normalized voltage amplitude “Vn / ω” is a value obtained by dividing the amplitude command value (hereinafter, voltage amplitude Vn) of the output voltage vector of the inverter 20 in the two-phase rotating coordinate system by the electrical angular velocity ω. . The voltage amplitude Vn is defined as the square root of the sum of the square value of the d-axis component Vd and the square value of the q-axis component Vq of the output voltage vector. In the present embodiment, the normalized voltage amplitude is calculated using a map in which the command torque Trq * and the normalized voltage amplitude are related.

  The speed calculation unit 30g calculates the electrical angular speed ω of the motor generator 10 based on the electrical angle θ detected by the rotation angle sensor 25. The speed multiplication unit 30h calculates the voltage amplitude Vn by multiplying the normalized voltage amplitude “Vn / ω” by the electrical angular velocity ω. Voltage amplitude Vn is an operation amount for performing feedforward control of torque of motor generator 10 to command torque Trq *.

  The amplitude correction unit 30i calculates the corrected amplitude Vc by adding the amplitude correction amount ΔC calculated by the correction amount calculation unit 32 to the voltage amplitude Vn output from the speed multiplication unit 30h. The correction amount calculation unit 32 will be described in detail later.

  The operation signal generation unit 30j is based on the corrected amplitude Vc output from the amplitude correction unit 30i, the corrected phase φc output from the phase correction unit 30e, and the input voltage VINV detected by the voltage sensor 44. Each operation signal gup, gun, gvp, gvn, gwp, gwn is generated and output to each drive circuit Dr. In this embodiment, the operation signals gup, gun, gvp, gvn, gwp, and gwn are generated as follows.

  The operation signal generator 30j first calculates a sine wave three-phase command voltage whose phases are shifted from each other by 120 degrees in electrical angle. Then, the operation signals gup, gun, gvp, gvn, gwp, and gwn are generated by triangular wave comparison PWM control based on the magnitude comparison between the calculated three-phase command voltage and a carrier signal (for example, a triangular wave signal).

  Next, the correction amount calculation unit 32 and the gain multiplication unit 34 will be described.

<1. About correction amount calculation unit>
First, a design method of the correction amount calculation unit 32 will be described.

  The voltage equation of the permanent magnet synchronous machine is expressed by the following equation (eq1).

In the above equation (eq1), “p” indicates a differential operator, “R” indicates an armature winding resistance, “Ld” and “Lq” indicate d and q axis inductances, and “ψ” is permanent. The effective value of the armature flux linkage of the magnet is shown. In the above equation (eq1), assuming a steady state in which the rotation speed of the motor generator 10 is constant and imposing a condition that the transient phenomenon is ignored, “p = 0” is obtained. Further, a condition that the rotational speed of the motor generator 10 is sufficiently high and the relationship of “R << ω · Ld” and “R << ω · Lq” is satisfied is given to the above equation (eq1). From the above, the above equation (eq1) is expressed as the following equation (eq2).

The relationship between the d and q axis voltages Vd and Vq, the voltage phase φ, and the voltage amplitude Vn is expressed by the following equation (eq3).

Here, the voltage equation when the voltage phase φ is changed by a minute amount Δφ is expressed by the following equation (eq4) using the above equations (eq2) and (eq3).

When the above equation (eq2) is subtracted from the above equation (eq4), the following equation (eq5) is derived.

In the above equation (eq5), “Idφ−Id” on the right side is the d-axis current change amount ΔIdφ, and “Iqφ−Iq” is the q-axis current change amount ΔIqφ. When the above equation (eq5) is solved for each current change amount ΔIdφ, ΔIqφ, the following equation (eq6) is derived.

FIG. 3 shows the voltage vector Vnvt and the current vector Invt in the dq coordinate system. Here, the current vector is defined as the square root of the sum of the square value of the d-axis current and the square value of the q-axis current. In FIG. 3, the change amount of the current vector Invt when the voltage phase φ is changed by a minute amount Δφ is indicated by “ΔIφ”. Further, the change amount of the current vector Invt when the voltage amplitude Vn is changed by a minute amount ΔVn is indicated by “ΔIvn”. FIG. 4 shows an enlarged view of the change in the current vector Invt. From the above equation (eq6), when the voltage phase φ is slightly changed, the change direction α of the current vector Invt with respect to the d axis is represented by the following equation (eq7).

For example, the change direction α can be calculated between “−π to + π” by the arctangent calculation of the above equation (eq7). Particularly in the present embodiment, when the denominator in the parenthesis is 0 and the numerator is a positive value on the right side of the above equation (eq7), the change direction α is calculated as “π / 2”. On the other hand, in the right side of the above equation (eq7), when the denominator in the parenthesis is 0 and the numerator is a negative value, the change direction α is calculated as “−π / 2”. Here, FIG. 5 shows a coordinate axis extending in a direction orthogonal to the changing direction of the current vector Invt as a λ axis (corresponding to a “phase non-interference axis”). That is, the λ axis is a coordinate axis in a direction in which the change amount of the current vector Invt becomes 0 when the voltage phase φ is slightly changed. Of the change ΔIvn of the current vector when the voltage amplitude Vn changes by a minute amount ΔVn, the λ-axis component obtained by mapping the change ΔIvn to the λ-axis is a current that is not affected by the change in the voltage phase φ. In the present embodiment, this current is used as the λ-axis current Iλ (corresponding to “phase non-interference current”) to calculate the amplitude correction amount ΔC. By using the λ-axis current Iλ, interference from phase control to amplitude control can be suppressed. Here, the angle λ formed by the d-axis and the λ-axis, which is a parameter necessary for setting the λ-axis, is expressed by the following equation (eq8).

Based on the above, returning to FIG. 2, the correction amount calculation unit 32 will be described.

  The λ-axis setting unit 32a forms the d-axis and the λ-axis based on the above equation (eq8) based on the d and q-axis inductances Ld and Lq and the voltage phase φ output from the phase setting unit 30d. The angle λ is calculated. The λ axis set in the λ axis setting unit 32a changes each time the driving state of the motor generator 10 changes.

  The command current setting unit 32b sets d and q-axis command currents Id * and Iq * for realizing the command torque Trq * based on the command torque Trq *. In the present embodiment, currents for realizing minimum current / maximum torque control are set as d and q-axis command currents Id * and Iq *.

  The λ-axis command current calculation unit 32c calculates the following equation (eq9) based on the command currents Id * and Iq * output from the command current setting unit 32b and the angle λ output from the λ-axis setting unit 32a. First, the λ-axis command current Iλ * is calculated (see FIG. 6).

Here, in FIG. 6, the current command current vector is indicated by “In *”, and the current current vector is indicated by “Invt”.

  The λ-axis actual current calculation unit 32d (corresponding to the “phase current calculation unit”) includes the d and q-axis currents Idr and Iqr output from the two-phase conversion unit 30a, and the angle λ output from the λ-axis setting unit 32a. Based on the above, the λ-axis current Iλr is calculated based on the following equation (eq10) (see FIG. 6).

Since the λ axis changes as the driving state of the motor generator 10 changes, the λ axis current Iλr and the λ axis command current Iλ * also change as the driving state of the motor generator 10 changes.

  The λ-axis current deviation calculation unit 32e calculates the λ-axis current deviation ΔIλ by subtracting the λ-axis current Iλr from the λ-axis command current Iλ *. Note that the λ-axis current Iλr input to the λ-axis current deviation calculation unit 32e may be subjected to a low-pass filter process for removing high frequency components.

  Based on the λ-axis current deviation ΔIλ, the amplitude correction amount calculation unit 32f calculates an amplitude correction amount ΔC as an amplitude operation amount as an operation amount for feedback-controlling the λ-axis current Iλr to the λ-axis command current Iλ *. . Specifically, the amplitude correction amount ΔC is calculated by proportional integral control using the λ-axis current deviation ΔIλ as an input. More specifically, the amplitude correction amount ΔC is calculated as an addition value of the output value of the proportional controller having the λ-axis current deviation ΔIλ as an input and the output value of the integral controller having the current deviation ΔIλ as an input. The amplitude correction amount ΔC calculated by the amplitude correction amount calculation unit 32f is input to the amplitude correction unit 30i.

  In the present embodiment, the control device 30 further includes an amplitude gain setting unit 36a and a phase gain setting unit 36b as shown in FIG. The amplitude gain setting unit 36a variably sets the proportional gain Kpv and the integral gain Kiv used for feedback control in the amplitude correction amount calculation unit 32f. Further, the phase gain setting unit 36b variably sets the proportional gain Kpφ and the integral gain Kiφ used for feedback control in the phase setting unit 30d. This setting is made to maintain high feedback control responsiveness in amplitude control and phase control even when the driving state of the motor generator 10 changes.

  The amplitude gain setting unit 36a variably sets the proportional gain Kpv and the integral gain Kiv based on the voltage phase φ and the electrical angular velocity ω so as to maintain constant feedback control response in amplitude control regardless of the driving state. Specifically, the gains Kpv and Kiv are set to be larger as the electrical angular velocity ω is higher or the voltage phase φ is more advanced. In order to maintain the responsiveness constant in amplitude control, for example, the λ-axis command current Iλ * is stepped in a state where the command voltage setting unit 30f and the speed multiplication unit 30h are removed from the configuration of FIG. This means that the time constant of the λ-axis current Iλr when it is changed is maintained at the target time.

  The phase gain setting unit 36b variably sets the proportional gain Kpφ and the integral gain Kiφ based on the voltage phase φ, the electrical angular velocity ω, and the voltage amplitude Vn so as to keep the phase control responsiveness constant regardless of the driving state. . Specifically, the gains Kpφ and Kiφ are set larger as the electrical angular velocity ω is higher, the voltage phase φ is retarded, or the voltage amplitude Vn is smaller. In the phase control, maintaining the responsiveness constant means, for example, maintaining the time constant of the estimated torque Te at the target time when the command torque Trq * is changed stepwise.

<2. About Gain Multiplier 34>
Next, the gain multiplication unit 34 will be described.

  The gain multiplication unit 34 calculates the phase correction amount Δph by multiplying the amplitude correction amount ΔC calculated by the amplitude correction amount calculation unit 32f by the correction gain k. In the following, the reason for providing the gain multiplication unit 34 in the control device 30 will be described, and then the design method of the correction gain k will be described.

  By using the λ axis in which the change of the current vector Invt with respect to the change of the voltage phase φ is made non-interfering, the interference from the phase control to the amplitude control can be suppressed. However, there is also interference from amplitude control to phase control. Due to this interference, as shown in FIG. 8, the speed at which the λ-axis current Iλr follows the λ-axis command current Iλ * is lower than the speed at which the estimated torque Te of the motor generator 10 follows the command torque Trq *. Become. Here, in FIG. 8, the transition of each waveform is shown so that the command torque Trq * and the λ-axis command current Iλ * coincide.

  When the speed at which the λ-axis current Iλr follows the λ-axis command current Iλ * is lower than the speed at which the estimated torque Te follows the command torque Trq *, a flux weakening current is generated. As a result, the loss of the motor generator 10 increases. FIG. 9A shows an ideal state where a weak flux current is not generated, and FIG. 9B shows a state where a weak flux current is generated. In the figure, d and q axis currents Id and Iq corresponding to the minimum current and maximum torque control (MTPA) are indicated by a one-dot chain line, and an equal torque line and a constant voltage ellipse are also indicated. FIG. 9 shows a quadrant in which the d-axis current Id has a negative value and the q-axis current Iq has a positive value in the dq coordinate system.

  In FIG. 9A, the operating point representing the current λ-axis current Iλr (t) and the estimated torque Te (t) is indicated by A. In addition, an operating point that represents the λ-axis command current Iλ * and the command torque Trq * and that should be shifted to the next processing cycle is indicated by B. In an ideal state, in the next processing cycle, the estimated torque Te (t + Δt) becomes the command torque Trq *, and the λ-axis current Iλr (t + Δt) becomes the λ-axis command current Iλ *.

  On the other hand, as shown in FIG. 9B, when the follow-up speed of the λ-axis current Iλr is lower than the follow-up speed of the estimated torque Te, the estimated torque Te (t + Δt) becomes the command torque in the next processing cycle. Although it becomes Trq *, the λ-axis current Iλr (t + Δt) is smaller than the λ-axis command current Iλ *. For this reason, the operating point shifts to C instead of B. As a result, the voltage becomes insufficient, a weak magnetic flux current flows, and the loss of motor generator 10 increases.

  Here, if the follow-up speed of the estimated torque Te and the follow-up speed of the λ-axis current Iλr are equal, no flux-weakening current is generated. Therefore, in the present embodiment, the gain multiplication unit 34 and the phase correction unit 30e are provided in order to make the follow-up speed of the estimated torque Te equal to the follow-up speed of the λ-axis current Iλr.

  Next, a design method for the correction gain k will be described.

  The torque T of the permanent magnet synchronous machine is expressed by the following equation (eq11).

In the above equation (eq11), Pn represents the number of pole pairs of the motor generator 10, and Ke represents the counter electromotive force constant (corresponding to the effective value ψ of the armature flux linkage). Further, assuming a steady state and ignoring the influence of the armature winding resistance R, the d and q axis currents Id and Iq are expressed by the following equation (eq12).

From the above equations (eq3), (eq11), and (eq12), the relationship between the torque and the voltage phase φ can be expressed as the following equation (eq13).

Using the above equation (eq13), the torque Tφ when the voltage phase φ is changed by a minute amount Δφ is represented by the following equation (eq14).

Here, the following expression (eq15) is established.

From the above equations (eq13) to (eq15), the following equation (eq16) is derived.

When “Δφ ^ 2≈0” in the above equation (eq16), the following equation (eq17) is derived.

On the other hand, using the above equation (eq13), the torque Tφ when the voltage amplitude Vn changes by a minute amount ΔV is represented by the following equation (eq18).

From the above equations (eq13) and (eq18), the following equation (eq19) is derived.

If “ΔV ^ 2≈0” in the above equation (eq19), the following equation (eq20) is derived.

The above equation (eq20) shows the torque change amount when the voltage amplitude Vn changes by a minute amount ΔV. On the other hand, the above equation (eq17) shows the torque change amount when the voltage phase φ changes by a minute amount Δφ. Therefore, in order to correct the voltage phase φ so as to cancel the torque change when the voltage amplitude Vn changes, the correction gain k may be set so as to satisfy the following equation (eq21).

That is, the correction gain k is expressed by the following equation (eq22).

From the above equation (eq22), the correction gain k is determined by the voltage phase φ, the voltage amplitude Vn, the electrical angular velocity ω, and the d and q axis inductances Ld and Lq. Therefore, in the present embodiment, the gain multiplication unit 34 variably sets the correction gain k based on the voltage amplitude Vn, the voltage phase φ, the electrical angular velocity ω, and the d and q axis inductances Ld and Lq. Then, the gain multiplication unit 34 calculates the phase correction amount Δph by multiplying the amplitude correction amount ΔC by the set correction gain k. The d and q axis inductances Ld and Lq may be set to fixed values or may be variably set according to the amount of current flowing through the motor generator 10 or the like.

  Incidentally, for example, the gain multiplying unit 34 receives the voltage amplitude Vn, the voltage phase φ, the electrical angular velocity ω, and the d and q axis inductances Ld and Lq as inputs for each processing cycle based on the above equation (eq22). May be calculated. Further, the gain multiplication unit 34 generates map data that defines the correction gain k associated with the voltage amplitude Vn, the voltage phase φ, and the electrical angular velocity ω, for example, when d and q-axis inductances Ld and Lq are fixed values. Based on this, the correction gain k may be calculated.

  Subsequently, the effects of the present embodiment will be described with reference to FIGS. 10 and 11. Here, in the figure, the related technique refers to the voltage phase set by the phase setting unit 30d in place of the corrected phase φc in the phase information input to the operation signal generation unit 30j in the configuration of FIG. It is a technology that uses φ.

  In FIG. 10, the alternate long and short dash lines indicate the trajectories of the d and q axis currents Id and Iq according to the minimum current and maximum torque control when the command torque Trq * is gradually increased. FIG. 10 shows a quadrant in which the d-axis current Id has a negative value and the q-axis current Iq has a positive value in the dq coordinate system. In the present embodiment, the gain multiplication unit 34 and the phase correction unit 30e suppress the change in the voltage phase accompanying the change in the amplitude correction amount ΔC, and suppress the interference from the amplitude control to the phase control. Thereby, the follow-up speed of the estimated torque Te to the command torque Trq * can be made equal to the follow-up speed of the λ-axis current Iλr to the λ-axis command current Iλ *. As a result, as shown in FIG. 10, it is possible to greatly reduce the flux-weakening current flowing through the motor generator 10 as compared with the related art. Therefore, as shown in FIG. 11, the loss of motor generator 10 can be greatly reduced.

  Thus, according to the present embodiment, it is possible to reduce the loss of the motor generator 10 while improving the torque controllability.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, a method for suppressing interference from amplitude control to phase control is changed. Specifically, in addition to the λ axis, a method using an s axis that is a coordinate axis that is not affected by a change in voltage amplitude is employed. For this reason, in the present embodiment, as shown in FIG. 12, the control device 30 includes an s-axis processing unit 40. In FIG. 12, the same processes as those shown in FIG. 2 are denoted by the same reference numerals for the sake of convenience. In the present embodiment, as shown in FIG. 12, the amplitude control is performed by the same method as in the first embodiment. Further, in the present embodiment, the λ-axis command current Iλ * corresponds to the “first command current”.

  First, a design method of the s-axis processing unit 40 will be described.

  The voltage equation when the voltage amplitude Vn changes by a minute amount ΔV is expressed by the following equation (eq23) using the above equations (eq2) and (eq3).

When the above equation (eq2) is subtracted from the above equation (eq23), the following equation (eq24) is derived.

In the above equation (eq24), “Idv−Id” on the right side is the d-axis current change amount ΔIdv, and “Iqv−Iq” is the q-axis current change amount ΔIqv. When the above equation (eq24) is solved for each current change amount ΔIdv, ΔIqv, the following equation (eq25) is derived.

In FIG. 3, the change amount of the current vector Invt when the voltage amplitude Vn is changed by a minute amount ΔV is indicated by “ΔIvn”. FIG. 13 shows an enlarged view of the change in the current vector Invt. From the above equation (eq25), when the voltage amplitude Vn changes slightly, the change direction m of the current vector Invt with respect to the d axis is represented by the following equation (eq26).

For example, the change direction m can be calculated between “−π to + π” by the arctangent calculation of the above equation (eq26). Here, in FIG. 14, a coordinate axis extending in a direction orthogonal to the changing direction of the current vector Invt is shown as an s-axis (corresponding to an “amplitude non-interference axis”). That is, the s-axis is a coordinate axis in the direction in which the change amount of the current vector Invt becomes 0 when the voltage amplitude Vn changes slightly. Of the current vector change ΔIφ when the voltage phase φ changes by a minute amount Δφ, the s-axis component obtained by mapping the change ΔIφ to the s-axis is a current that is not affected by the change in the voltage amplitude Vn. In the present embodiment, this current is used as the s-axis current Is (corresponding to “amplitude non-interfering current”) for calculation of the voltage phase φ. By using the s-axis current Is, interference from amplitude control to phase control can be suppressed. Here, an angle γ formed by the d-axis and the s-axis, which is a parameter necessary for setting the s-axis, is expressed by the following equation (eq27).

Based on the above, returning to FIG. 12, the s-axis processing unit 40 will be described.

  Based on the above equation (eq27), the γ-axis setting unit 40a is based on the d and q-axis inductances Ld and Lq and the voltage phase φ output from the phase setting unit 40b described later. Is calculated. The γ-axis set in the γ-axis setting unit 40a changes each time as the driving state of the motor generator 10 changes.

  The s-axis command current calculation unit 40c is configured as a second command current based on the command currents Id * and Iq * output from the command current setting unit 32b and the angle γ output from the γ-axis setting unit 40a. The s-axis command current Is * is calculated (see FIG. 15). The s-axis actual current calculation unit 40d (corresponding to the “amplitude current calculation unit”) includes the d and q-axis currents Idr and Iqr output from the two-phase conversion unit 30a, and the angle γ output from the γ-axis setting unit 40a. S-axis current Isr is calculated based on (see FIG. 15).

  The s-axis current deviation calculation unit 40e calculates the s-axis current deviation ΔIs by subtracting the s-axis current Isr from the s-axis command current Is *. Based on the s-axis current deviation ΔIs, the phase setting unit 40b calculates the voltage phase φ as an operation amount for performing feedback control of the s-axis current Isr to the s-axis command current Is *. Specifically, the voltage phase φ is calculated by proportional-integral control using the s-axis current deviation ΔIs as an input. Note that, similarly to the phase gain setting unit 36b described in the first embodiment, the feedback gain used in the phase setting unit 40b may be variably set.

  Based on the corrected amplitude Vc output from the amplitude correction unit 30i, the voltage phase φ output from the phase setting unit 40b, and the input voltage VINV, the operation signal generation unit 30j outputs the operation signals gup, gun, and gvp. , Gvn, gwp, and gwn.

  According to the present embodiment described above, the same effect as that of the first embodiment can be obtained.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In this embodiment, the design method of the correction amount calculation unit is changed. Specifically, the λ axis is derived not in the polar coordinate system but in the biaxial orthogonal coordinate system.

  When the condition that the transient phenomenon is ignored in the above equation (eq1), the following equation (eq28) is derived.

In the above equation (eq28), the following equation (eq29) is obtained from the relationship between the minute change amounts ΔId and ΔIq of the d and q axis currents Id and Iq when the d and q axis voltages Vd and Vq are slightly changed by ΔVd and ΔVq. Led.

Here, as shown in FIG. 16, a coordinate system including an l-axis extending from the origin 0 in a direction parallel to the voltage vector Vnvt and a p-axis extending from the origin 0 in a direction orthogonal to the voltage vector Vnvt is defined as a pl coordinate system. To do. Similarly to the λ axis, the pl coordinate system also changes as the driving state of the motor generator 10 changes. Here, the relationship between the voltage vector Vnvt converted from the dq coordinate system to the pl coordinate system and the current vector Invt converted from the dq coordinate system to the λo coordinate system different from the pl coordinate system is derived. The pl coordinate system is a coordinate system obtained by rotating the dq coordinate system about the origin 0 counterclockwise by an angle η (= φ−π / 2). The λo coordinate system is a coordinate system obtained by rotating the dq coordinate system about the origin 0 counterclockwise by an angle λ. For this reason, if the minute change amounts of the p and l axis voltages on the p and l axes corresponding to ΔVd and ΔVq are ΔVp and ΔVl, the following equation (eq30) is derived from the above equation (eq29).

When the above equation (eq30) is solved for the current, the following equation (eq31) is derived.

Here, increasing / decreasing ΔVl is equivalent to increasing / decreasing the amplitude of the voltage vector Vnvt. Further, increasing or decreasing ΔVp is equivalent to increasing or decreasing the phase of voltage vector Vnvt. For this reason, as in the first embodiment, the p-axis voltage Vp is controlled to control the torque, and the l-axis voltage Vl is controlled to control the current. In order to control only the l-axis voltage Vl without receiving, an angle λ is set on the right side of the above equation (eq32) so that “Rc−ω · Loλ = 0”, and then the λ-axis current Iλ is used. The l-axis voltage Vl may be controlled. Alternatively, in order to control only the l-axis voltage Vl without being affected by the change of the p-axis voltage Vp, an angle λ is set such that “Rs−ω · Lλ = 0” on the right side of the above equation (eq32). Above, the l-axis voltage Vl may be controlled using the o-axis current Io.

  Here, if the electrical angular velocity ω is sufficiently high, “Rc << ω · Loλ” and “Rs << ω · Lλ” are established. In order to control the l-axis voltage Vl using the λ-axis current Iλ, “−ω · Loλ = 0”, that is, “Loλ = 0” is sufficient. For this reason, the following expression (eq32) is derived.

On the other hand, in order to control the l-axis voltage Vl using the o-axis current Io, “−ω · Lλ = 0”, that is, “Lλ = 0” is sufficient. For this reason, the following formula (eq33) is derived.

The above equation (eq33) is obtained by rotating the above equation (eq32) by “−π / 2”, and the λ axis in the above equation (eq32) coincides with the o axis in the above equation (eq33). For this reason, in order to control the l-axis voltage Vl, the effect is the same when the λ-axis current Iλ is used and when the o-axis current Io is used. The derived angle λ coincides with the angle represented by the above equation (eq8) that can suppress interference from phase control to amplitude control.

  FIG. 17 shows a block diagram of torque control according to the present embodiment. In FIG. 17, the same processes as those shown in FIG. 2 are denoted by the same reference numerals for the sake of convenience.

  The command voltage setting unit 30k calculates the standardized voltage amplitude “Vl / ω” using the command torque Trq * as an input. In the present embodiment, the normalized voltage amplitude “Vl / ω” is a value obtained by dividing the l-axis voltage Vl by the electrical angular velocity ω. Note that the normalized voltage amplitude may be calculated using a map in which the command torque Trq * and the normalized voltage amplitude are associated with each other.

  The speed multiplication unit 301 calculates the l-axis voltage Vl by multiplying the normalized voltage amplitude “Vl / ω” by the electrical angular speed ω. The l-axis voltage Vl is an operation amount for performing feedforward control of the torque of the motor generator 10 to the command torque Trq *.

  The amplitude correction amount calculation unit 32h constituting the correction amount calculation unit 32 uses the amplitude of the l-axis voltage Vl as an operation amount for performing feedback control of the λ-axis current Iλr to the λ-axis command current Iλ * based on the current deviation ΔIλ. A correction amount Δl is calculated. Specifically, the amplitude correction amount Δl is calculated by proportional-integral control using the current deviation ΔIλ as an input. The amplitude correction amount Δl corresponds to an amplitude operation amount that is an l-axis component of the voltage vector Vnvt, and is a voltage component according to the voltage amplitude Vn. The amplitude correction unit 30m calculates the corrected l-axis voltage Vlc by adding the amplitude correction amount Δl calculated by the correction amount calculation unit 32 to the l-axis voltage Vl output from the speed multiplication unit 30l.

  The phase setting unit 30n calculates the p-axis voltage Vp as an operation amount for feedback control of the estimated torque Te to the command torque Trq * based on the torque deviation ΔT. Specifically, the p-axis voltage Vp is calculated by proportional-integral control with the torque deviation ΔT as an input.

  The gain multiplier 38 variably sets the correction gain k based on the l-axis voltage Vl, the p-axis voltage Vp, and the electrical angular velocity ω. Here, the reason why the l-axis voltage Vl is used for setting the correction gain k is that, as described above, changing the l-axis voltage Vl is equivalent to changing the amplitude of the voltage vector Vnvt. The reason why the p-axis voltage Vp is used for setting the correction gain k is that changing the p-axis voltage Vp is equivalent to changing the phase of the voltage vector Vnvt. The gain multiplication unit 38 calculates the phase correction amount Δr by multiplying the set correction gain k by the amplitude correction amount Δl.

  The phase correction unit 30p calculates the corrected p-axis voltage Vpc by subtracting the phase correction amount Δr calculated by the gain multiplication unit 38 from the p-axis voltage Vp set by the phase setting unit 30n. The corrected p-axis voltage Vpc corresponds to a phase operation amount that is a p-axis component of the voltage vector Vnvt, and is a voltage component according to the voltage phase φ. The corrected l-axis voltage Vlc and the corrected p-axis voltage Vpc are input to the operation signal generation unit 30q.

  The operation signal generator 30q sets the corrected l-axis voltage Vlc as the voltage amplitude Vn. Further, the operation signal generation unit 30q sets the voltage phase φ based on the relationship “φ = η + π / 2”. In this embodiment, the angle η is subjected to low-pass filter processing in order to remove a noise component of the calculated value of the angle η formed by the d axis and the p axis. For this reason, the voltage phase φ is set by the following equation (eq34).

In the above equation (eq34), the first term on the right side corresponds to the change in the angle η formed by the d-axis and the p-axis in the period from the previous processing cycle to the current processing cycle. In order to calculate the voltage phase φ in the next processing cycle, the angle η of the second term on the right side of the above equation (eq34) is “π / 2” from the voltage phase φ calculated in the current processing cycle for each processing cycle. ”Is updated to the subtracted value. That is, the previously updated angle η is used as the angle η of the second term on the right side of the above equation (eq34). The operation signal generation unit 30q calculates a three-phase command voltage based on the set voltage amplitude Vn and voltage phase φ, and performs sinusoidal PWM control based on a magnitude comparison between the calculated three-phase command voltage and the carrier signal. Operation signals gup, gun, gvp, gvn, gwp, gwn are generated.

  Incidentally, in the present embodiment, the λ-axis setting unit 32i constituting the correction amount calculation unit 32 includes the corrected p-axis voltage Vpc output from the phase correction unit 30p and the corrected l-axis output from the amplitude correction unit 30m. Based on the voltage Vlc, an angle λ formed by the d-axis and the λ-axis is calculated. Specifically, first, the corrected p-axis voltage Vpc and the corrected l-axis voltage Vlc are input, and the voltage phase φ is calculated based on the above equation (eq34). The λ axis setting unit 32i receives the calculated voltage phase φ and calculates the angle λ based on the above equation (eq32).

  In the present embodiment described above, the voltage as the operation amount is operated in the pl coordinate system, and the estimated torque Te and the λ-axis current Iλr are included as the control amounts. According to this embodiment, the same effect as the first embodiment can be obtained.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In the first embodiment, when the salient pole ratio of the motor generator 10 is small, “Ld≈Lq”. In this case, the above equation (eq22) becomes the following equation (eq35).

For this reason, the correction gain k may be variably set based on the voltage amplitude Vn and the voltage phase φ without using the electrical angular velocity ω.

  Further, the correction gain k may be variably set based on the command torque Trq * and the electrical angular velocity ω. This is because, when the minimum current / maximum torque control is performed, the d and q axis currents Id and Iq are uniquely determined with respect to the command torque Trq *, and from the d and q axis currents Id and Iq, This is because the voltage amplitude Vn and the voltage phase φ are also uniquely determined.

In the first embodiment, the feedback gains Kpv and Kiv used in the amplitude correction amount calculation unit 32f may be variably set based on either the electrical angular velocity ω or the voltage phase φ. Further, the feedback gains Kpv and Kiv may be variably set based on the current and torque flowing through the motor generator 10. This will be described below.

  Assuming a steady state and ignoring the influence of the armature winding resistance R, the d and q axis voltages Vd and Vq are expressed by the following equation (eq29).

In this case, the voltage phase φ is expressed by the following equation (eq38).

When the minimum current / maximum torque control is performed, when the command torque Trq * is determined, the d and q axis currents Id and Iq are uniquely determined. Therefore, when the command torque Trq * is determined, the voltage phase φ is also determined from the above equation (eq38). Therefore, the feedback gains Kpv and Kiv can be set by using the command torque Trq * instead of the voltage phase φ and using the electrical angular velocity ω. Here, the feedback gains Kpv and Kiv may be set larger as the command torque Trq * is smaller.

  For example, when the SPMSM that is a non-salient pole machine is used as the motor generator 10, “Ld = Lq”. Therefore, according to the above equations (eq11) and (eq38), each feedback gain can be set based on the q-axis current Iq and the electrical angular velocity ω.

  In the first embodiment, the phase correction amount Δph is calculated by multiplying the amplitude correction amount ΔC by the correction gain k. However, the present invention is not limited to this. For example, the phase correction amount Δph may be calculated by adding / subtracting a predetermined correction amount to / from the amplitude correction amount ΔC. Here, the predetermined correction amount may be set based on the voltage amplitude Vn, the voltage phase φ, and the electrical angular velocity ω, as in the method of setting the correction gain k.

  In the first embodiment, the feedback gains Kpφ and Kiφ used in the phase setting unit 30d may be variably set based on any one or two of the electrical angular velocity ω, the voltage phase φ, and the voltage amplitude Vn. . Further, the feedback gains Kpφ and Kiφ may be variably set based on the current and torque flowing through the motor generator 10. Specifically, the feedback gains Kpφ and Kiφ may be set larger as the current is smaller or the torque is smaller.

  In each of the above embodiments, the feedback control in at least one of the phase setting unit 30d and the amplitude correction amount calculation unit 32f may be performed only by integration control, for example. Further, for example, the feedback control is not limited to proportional control or integral control, but may be differential control.

  In each of the above embodiments, the command voltage setting units 30f and 30k and the speed multiplication units 30h and 30l may be removed from the control device 30. Even in this case, the feedback gains Kpv and Kiv in the amplitude correction amount calculation unit 32f can be set large by using the λ-axis current Iλr. For this reason, high torque controllability can be maintained.

  The method for generating the operation signal is not limited to the one based on the triangular wave comparison PWM control, and for example, the method described below may be used. For example, the control device 30 includes storage means (for example, a memory such as a ROM) in which a line voltage pattern for realizing the voltage amplitude Vn is stored in advance. In such a configuration, the line voltage pattern stored in the storage means is converted into a pulse pattern for the gate of each switching element, and the operation signal is generated by setting the output timing of the converted pulse pattern based on the voltage phase φ. To do.

  Further, the method of generating the operation signal is not limited to the calculation of the sine wave three-phase command voltage whose phases are shifted from each other by 120 degrees in electrical angle based on the voltage amplitude and the voltage phase. For example, a method of superposing a third-order high frequency on a sine wave may be used.

  In the first and third embodiments, the torque value used for phase control is not limited to the estimated value. For example, the control system may include a torque detection unit (for example, a torque measuring device) that detects the torque of the motor generator 10, and the detected torque may be used for phase control.

  In each of the above embodiments, when the command torque Trq * is low, the voltage phase φ is set near “π / 2”. For this reason, the angle λ output from the λ-axis setting unit is close to zero. Under such circumstances, since the λ-axis current is substantially not used for amplitude control, it is not necessary to assume amplitude control using the λ-axis current.

  The rotary electric machine is not limited to IPMSM, but may be SPMSM or a wound field type synchronous machine. Here, for example, when SPMSM is employed in the first embodiment, the torque of the rotating electrical machine is determined by the q-axis current, and therefore the control amount related to the torque may be the q-axis current instead of the torque. Further, the rotating electrical machine is not limited to being used as an in-vehicle main machine, but may be used as an in-vehicle auxiliary machine such as an electric power steering device or an electric motor constituting an air-conditioning electric compressor. In addition, the rotating electrical machine is not limited to a vehicle-mounted type.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 20 ... Inverter, 30 ... Control apparatus, Sup-Swn ... Switch.

Claims (10)

Applied to a rotating electrical machine (10) electrically connected to a power conversion circuit (20) having switching elements (Sup to Swn);
As an operation amount for feedback control of the control amount of the rotating electrical machine to its command value, the phase of the output voltage vector of the power conversion circuit in the rotating coordinate system of the rotating electrical machine, or the orthogonal biaxial component of the output voltage vector A phase setting unit (30d; 40b; 30n) for setting a phase operation amount which is one of the components according to the phase;
As an operation amount for feedback-controlling the current flowing through the rotating electrical machine to a command current corresponding to the command value, the amplitude of the output voltage vector, or a component according to the amplitude of the orthogonal two-axis components of the output voltage vector An amplitude setting unit (32f; 32h) for setting an amplitude operation amount that is any one of
An operation unit for operating the switching element based on the phase operation amount set by the phase setting unit and the amplitude operation amount set by the amplitude setting unit to control the control amount to the command value. (30j; 30q)
The control apparatus for a rotating electrical machine, wherein the phase setting unit sets the phase operation amount so as to suppress a change in the control amount accompanying a change in the amplitude operation amount. In the rotating coordinate system, a coordinate axis in which a change in the current vector flowing through the rotating electrical machine with respect to a change in the phase manipulated variable is made non-interfering is a phase non-interfering axis, and the phase non-interfering axis direction component of the current vector is a phase non-interfering axis direction component. A phase current calculation unit (32d) for calculating the interference current;
2. The rotating electrical machine according to claim 1, wherein the amplitude setting unit sets the amplitude operation amount as an operation amount for feedback-controlling the phase non-interference current calculated by the phase current calculation unit to the command current. Control device. A phase for correcting the phase operation amount set by the phase setting unit (30d; 30n) based on the change amount of the amplitude operation amount in order to suppress the change of the control amount accompanying the change of the amplitude operation amount. A correction unit (30e; 30p);
The control device for a rotating electrical machine according to claim 1, wherein the operation unit operates the switching element based on the phase operation amount corrected by the phase correction unit and the amplitude operation amount.   The control device for a rotating electrical machine according to claim 3, wherein the phase correction unit corrects the phase operation amount by multiplying a change amount of the amplitude operation amount by a correction gain.   The rotating electrical machine according to claim 4, further comprising a gain setting unit (34; 38) that variably sets the correction gain based on at least one of an amplitude of the output voltage vector and a phase of the output voltage vector. Control device.   The correction gain is set to a value obtained by dividing the amount of change in the control amount when the amplitude of the output voltage vector changes slightly by the amount of change in the control amount when the phase of the output voltage vector changes slightly. The rotating electrical machine control device according to claim 5. A torque estimation unit (30b) for estimating the torque of the rotating electrical machine based on the current flowing through the rotating electrical machine;
The control device for a rotating electrical machine according to claim 1, wherein the control amount is a torque estimated by the torque estimating unit. The command current is a first command current,
In the rotating coordinate system, a coordinate axis in which a change in the current vector flowing through the rotating electrical machine with respect to a change in the amplitude manipulated variable is made non-interfering is an amplitude non-interfering axis, and the amplitude non-interfering axis direction component of the current vector An amplitude current calculation unit (40d) for calculating an interference current;
The phase setting unit (40b) uses the amplitude operation amount as an operation amount for feedback-controlling the amplitude non-interference current calculated by the amplitude current calculation unit to a second command current according to the command value. The control device for a rotating electrical machine according to claim 2 to be set.   Based on at least one of the rotational speed of the rotating electrical machine, the torque of the rotating electrical machine, the current flowing through the rotating electrical machine, and the phase of the output voltage vector, the feedback gain used in feedback control by the amplitude setting unit is variable. The control device for a rotating electrical machine according to any one of claims 1 to 8, further comprising an amplitude gain setting unit (36a) for setting.   Feedback by the phase setting unit based on at least one of the rotational speed of the rotating electrical machine, the torque of the rotating electrical machine, the current flowing through the rotating electrical machine, the phase of the output voltage vector, and the output voltage of the power conversion circuit. The control apparatus for a rotating electrical machine according to any one of claims 1 to 9, further comprising a phase gain setting unit (36b) that variably sets a feedback gain used in the control.