JP5444983B2 - Rotating machine control device - Google Patents

Description

  The present invention provides an electrical angle of the rotating machine when controlling a control amount of the rotating machine by operating a power conversion circuit including a switching element that selectively connects a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine. A voltage having a frequency different from the frequency and having an arbitrary phase angle and a frequency signal in which a voltage having a phase angle different from the phase angle by 180 degrees is periodically switched are superimposed on the output voltage of the power conversion circuit. The present invention relates to a control device for a rotating machine that estimates a rotation angle of the rotating machine based on a current signal that is actually propagated through the rotating machine by the superposition.

  As this type of control device, for example, as shown in Patent Document 1 below, when a voltage is alternately applied in the positive and negative directions of the estimated d-axis of a three-phase motor, a current signal that is actually propagated to the motor is used. There has also been proposed a method for estimating the electric angle of an electric motor (paragraph “0038”).

Japanese Patent No. 3454212

  However, when the voltage for electrical angle detection is applied as described above, noise in the audible frequency band is generated by the application of this voltage. For this reason, in the situation where noise is disliked, it tends to be avoided to apply the above technique as sensorless control of an electric motor.

  The present invention has been made in order to solve the above-described problems, and an object of the present invention is to provide a voltage having a frequency different from the electrical angle frequency of the rotating machine and having an arbitrary phase angle, and the phase angle 180. A frequency signal at which voltages having different phase angles are periodically switched is superimposed on the output voltage of the power conversion circuit, and the rotation angle of the rotating machine is determined based on a current signal that is actually propagated through the rotating machine by the superposition. It is an object of the present invention to provide a control device for a rotating machine that can suitably reduce noise when estimating.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

In the configuration 1 , when the control amount of the rotating machine is controlled by operating a power conversion circuit including a switching element that selectively connects a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine having saliency, The power converter circuit outputs a voltage signal having a frequency different from the electrical angle frequency of the machine and having an arbitrary phase angle and a frequency signal periodically switching a voltage having a phase angle different from the phase angle by 180 degrees. In the controller for a rotating machine that estimates the rotation angle of the rotating machine based on a current signal that is superimposed on the voltage and that is actually propagated through the rotating machine, the current that is actually propagated by operating the frequency signal Limiting means for limiting the rate of change of the signal is provided.

  In the above invention, by providing the limiting means, the rate of change of the current flowing through the rotating machine can be reduced, and as a result, noise caused by the superposition of frequency signals can be suitably suppressed.

In Configuration 2 , in Configuration 1 or 2 , the limiting unit periodically switches between the plurality of pulses as the voltage having the arbitrary phase angle and the plurality of pulses as the voltage having the phase angle different by 180 degrees. It is a means.

  In the above invention, the average value of the pair of voltages can be reduced by making each of a voltage having an arbitrary phase angle and a voltage having a phase angle different by 180 degrees into a plurality of pulses. For this reason, the speed of change of the current actually propagated through the rotating machine can be reduced by the pair of voltages.

Configuration 3 is the configuration 1 , wherein the DC power supply is an output terminal of a converter for making the input voltage of the power conversion circuit variable, and the limiting means operates the converter to A control voltage, which is a voltage applied to the rotating machine to control a control amount, and a composite voltage of the voltage having the arbitrary phase angle, and a composite voltage of the control voltage and the voltage having a phase angle different from the 180 degrees. The input voltage is limited so that a value obtained by subtracting the larger one of the values from the input voltage of the power conversion circuit is equal to or less than a specified value.

  Although the power conversion circuit can vary the average value of its output voltage, the instantaneous voltage depends on the input voltage. And the change rate of the electric current which flows through a rotary machine with the change of the output voltage of an inverter becomes large, so that this input voltage is large. In this regard, in the above invention, by limiting the input voltage, it is possible to limit the rate of change of the actually propagated current signal.

In the configuration 4 , in the configuration 3 , the limiting unit is configured such that the control voltage and the combined voltage of the voltage having the arbitrary phase angle are combined with the combined voltage of the control voltage and the voltage having the phase angle different from the 180 degrees. The converter is operated so that the larger one of them and the input voltage of the power conversion circuit are matched.

  According to the above invention, the change rate of the current signal can be reduced as much as possible while avoiding a decrease in controllability of the control amount.

The configuration 5 further includes means for setting a command voltage for controlling the control amount in the configuration 3 or 4 , and means for calculating a norm of the frequency signal superimposed on the set command voltage, The limiting means operates the converter with the calculated norm as an input.

  The norm represents a combined voltage of the control voltage and the voltage having the arbitrary phase angle and a combined voltage of the control voltage and the voltage having the phase angle different from the 180 degrees. For this reason, the process which restrict | limits an input voltage based on the one with a larger synthetic | combination voltage can be performed based on an appropriate input parameter.

Configuration 6 is characterized in that, in Configuration 3 or 4 , the limiting means operates the converter with at least one of an electrical angular velocity of the rotating machine and a torque of the rotating machine as an input.

  The electrical angular velocity and torque are parameters that have a correlation with the voltage applied to the rotating machine. In the above invention, in view of this point, at least one of these parameters is used as an input of a process for limiting the input voltage based on the higher composite voltage.

According to Configuration 7 , in any one of Configurations 1 to 6 , the limiting means is a current that is actually propagated through the rotating machine so that a detected value of the audio signal generated by the superposition of the frequency signal is within an allowable range. It is characterized by limiting the rate of change of the signal.

  In the said invention, it can avoid reliably that an audio | voice signal becomes large excessively by carrying out feedback control of an audio | voice signal.

1 is a system configuration diagram according to a first embodiment. FIG. The block diagram which shows the cross-sectional structure of the motor generator concerning the embodiment. The figure which shows the setting method of the power supply voltage concerning the embodiment. The figure which shows the effect of the same embodiment. The figure for demonstrating the effect of the embodiment. The system block diagram concerning 2nd Embodiment. The system block diagram concerning 3rd Embodiment. The system block diagram concerning 4th Embodiment. The system block diagram concerning 5th Embodiment. The figure which shows the setting method of the power supply voltage concerning the embodiment. The figure which shows the voltage application method concerning 6th Embodiment.

(First embodiment)
A first embodiment in which a rotating machine control device according to the present invention is applied to a rotating machine control device mounted on a hybrid vehicle will be described below with reference to the drawings.

  FIG. 1 shows an overall configuration of a control system according to the present embodiment.

  The illustrated motor generator 10 is an embedded magnet synchronous motor (IPMSM). That is, as shown in FIG. 2, the rotor 10a of the motor generator 10 is configured by embedding a permanent magnet in an iron body.

  As shown in FIG. 1, the phase currents (U-phase actual current iu, V-phase actual current iv, and W-phase actual current iw) that actually flow through motor generator 10 are fixed by αβ converter 20. It is converted into a current in a two-dimensional coordinate system (αβ coordinate system) (actual current iα on the α axis and actual current iβ on the β axis). The real current iα on the α-axis is taken in by the low-pass filter 22 and the real current iβ on the β-axis is taken into the dq conversion unit 26 after the harmonic components are removed by the low-pass filter 24, respectively. The current on the αβ axis that actually flows through the motor generator 10 as the output of these low-pass filters 22 and 24 is converted by the dq converter 26 into current in the rotating two-dimensional coordinate system, that is, current vector components of the d-axis and q-axis. Is done. In this conversion, the electrical angle θ of the output shaft of the motor generator 10 is used.

  On the other hand, the command current setting unit 28 sets the command current idr on the d-axis and the command current iqr on the q-axis based on the command value (requested torque Tr) of the control amount (torque) for the motor generator 10. . Then, the difference between the command current idr on the d axis and the actual current id is calculated by the deviation calculating unit 30, and the difference between the command current iqr on the q axis and the actual current iq is calculated by the deviation calculating unit 32. The current controller 34 calculates the command voltage vdr on the d-axis and the command voltage vqr on the q-axis based on the outputs of the deviation calculation units 30 and 32. This calculation process basically includes an operation amount for feedback control of the actual current id to the command current idr on the d axis, and feedback control of the actual current iq to the command current iqr on the q axis. Therefore, the operation amount calculation process is performed. This feedback control may be proportional integral control, for example. Specifically, at this time, the electric angular velocity ω is input, an operation amount for feedforward control by non-interference control or induced voltage compensation processing is calculated, and the command voltage vdr is calculated as the sum of this and the operation amount for feedback control. , Vqr are calculated.

  The command voltages vdr and vqr are converted into a command voltage vαr on the α axis and a command voltage vβr on the β axis in the αβ conversion unit 36. For this conversion, the electrical angle θ is used.

  In the three-phase conversion unit 42, the output of the adder 38 corresponding to the command voltage vαr on the α axis and the output of the adder 40 corresponding to the command voltage vβr on the β axis are the U-phase command voltage vur. , V-phase command voltage vvr, and W-phase command voltage vwr.

  Based on the command voltages vur, vvr, and vwr, the PWM processing unit 44 generates an operation signal for the inverter 50 for applying the command voltages vur, vvr, and vwr to the motor generator 10. This process is a process for generating a signal for turning on and off a switching element SW of the inverter 50 in accordance with a comparison between each of the command voltages vur, vvr, and vwr and the carrier. Here, the fluctuation range of the carrier is the input voltage (power supply voltage VDC) of the inverter 50. For this reason, this process is performed using the input voltage of the inverter 50 as an input. Here, the inverter 50 includes a plurality of switching elements SW that selectively connect a pair of input terminals and the terminals of the motor generator 10, and the pseudo sine is derived from the input DC voltage by the operation of the switching elements SW. An AC output voltage such as a wave voltage is generated. In the present embodiment, the inverter 50 is connected to a converter 54 that boosts the voltage of the high voltage battery 52. Incidentally, the power supply voltage VDC is detected by the voltage sensor 56.

  Next, processing related to acquisition of the electrical angle θ of the motor generator 10 according to the present embodiment will be described. FIG. 1 shows processing related to acquisition of the electrical angle θ in the region where the rotational speed of the motor generator 10 is low, and does not describe processing related to acquisition of the electrical angle θ in the high rotational speed region. In the high rotation speed region, for example, a method of estimating the electrical angle θ based on the induced voltage may be employed.

  In the present embodiment, when the motor generator 10 is driven, a high-frequency voltage having a cycle shorter than the rotation angle of the electrical angle of the motor generator 10 and oscillating between the d-axis positive direction and the negative direction is applied to the inverter 50. Superimpose on output. In other words, a high frequency voltage having a cycle shorter than the cycle of the current actually flowing through the motor generator 10 is superimposed according to the command currents idr and iqr. Specifically, the high frequency voltage is a signal having the same period as the period of the carrier signal in the PWM processing unit 44. Specifically, for example, the carrier is a triangular wave signal having the same increasing speed and decreasing speed, and a positive voltage on the d-axis and a negative voltage on the d-axis are assigned to the increasing period and the decreasing period. The received signal may be a high frequency voltage.

  The electrical angle θ of the motor generator 10 is calculated based on a high frequency signal (current signal) that is actually propagated through the motor generator 10 by the superposition of the high frequency voltage. This is done in view of the fact that the motor generator 10 has saliency.

  Specifically, the high frequency voltage generation unit 66 converts the high frequency voltage signal (vhd, 0) expressed on the dq axis into αβ components and outputs them as high frequency voltages vhα and vhβ. As a result, the high frequency voltage vhα is added to the command voltage vαr output from the αβ conversion unit 36 in the adder 38, and the high frequency voltage vhβ is added to the command voltage vβr output from the αβ conversion unit 36 in the adder 40. The For this reason, the motor generator 10 is applied with voltages in which the high-frequency voltages vhα and vhβ are superimposed on the command voltages vαr and vβr, respectively.

  On the other hand, the actual current iα on the α axis output from the αβ converter 20 is a harmonic current by the high-pass filter 60, and the actual current iβ on the β axis output from the αβ converter 20 is a harmonic current by the high-pass filter 62, respectively. Ingredients are extracted. The electrical angle θ is estimated and calculated by the position estimator 64 using the outputs of the high-pass filters 60 and 62 and the high-frequency voltages vhα and vhβ as inputs.

  That is, the motor generator 10 has a minimum inductance in the d-axis direction and a maximum inductance in the q-axis direction due to its structure. Accordingly, since current flows more easily in the d-axis direction than in the q-axis direction, when the high-frequency voltages vhα and vhβ are superimposed, the high-frequency current signal actually propagated through the motor generator 10 is deflected in the d-axis direction. To do. Specifically, when the estimated d-axis (estimated d-axis) is advanced with respect to the actual d-axis (real d-axis), the high-frequency voltages vhα and vhβ are superimposed in the estimated d-axis direction. At this time, the direction of the actually propagated current signal is deviated to the actual d-axis side, so that it is deviated from the estimated d-axis. When the estimated d-axis is retarded with respect to the actual d-axis, when the high-frequency voltages vhα and vhβ are superimposed on the estimated d-axis, the direction of the current signal that is actually propagated is the actual d-axis side. To deviate toward the estimated d-axis.

  If the above property is used, the d-axis can be estimated and calculated, and the electrical angle θ can be calculated. Specifically, for example, the phase angle of the high-frequency voltages vhα and vhβ and the phase angle of the actually propagated current signal may be calculated, and the electrical angle θ may be estimated so that these errors are zero. The electrical angular velocity ω is calculated by differentiating the electrical angle θ output from the position estimator 64 by the speed calculation unit 68.

  By the way, the high-frequency voltages vhα and vhβ are normally “several hundred to several kHz”. For this reason, the frequency of noise generated by the high-frequency current signal flowing in the motor generator 10 by superimposing the high-frequency voltages vhα and vhβ is in the audible frequency band. For this reason, there is a possibility that noise may be generated by a current signal generated by superimposing the high frequency voltages vhα and vhβ.

  Therefore, in the present embodiment, the noise is suppressed by limiting the input voltage (power supply voltage VDC) of the inverter 50 as much as possible within a range in which the controllability (here, torque) of the motor generator 10 is not lowered. This is because the rate of change of current due to switching of the switching of the switching element SW increases as the input voltage of the inverter IV increases. Specifically, a power supply voltage changer 70 is provided, and an output voltage command value (power supply voltage command value VDCr) for the converter 54 is variably set according to the outputs of the adders 38 and 40.

  Here, the power supply voltage command value VDCr has a magnitude equal to the larger one of the combined voltages of the control amount control voltage and the high-frequency voltages vhα and vhβ. That is, as shown in FIG. 3B, the power supply voltage VDC is controlled by the high-frequency voltage vectors vh (+ d) and vh (−d) having phase angles in the positive and negative directions of the d-axis and the control amount. Is set to a value obtained by multiplying the larger one of the norm of a pair of vectors consisting of the sum (the combined voltage vector in the two-dimensional coordinate system) with the square voltage of “8/3”. This corresponds to setting the modulation factor at the time of applying the control voltage and the high-frequency voltage to “1”.

  According to such setting, the input voltage of the inverter 50 can be set to the minimum necessary voltage. On the other hand, as shown in FIG. 3A, when the power supply voltage VDC is smaller than the value obtained by multiplying the larger norm of the combined voltage vector by the square root of “8/3”, the actual voltage of the inverter IV is reduced. Since the output voltage cannot be the combined voltage vector, the control accuracy of the control amount is lowered. On the other hand, as shown in FIG. 3C, when the power supply voltage VDC is larger than the value obtained by multiplying the larger norm of the combined voltage vector by the square root of “8/3”, the input voltage of the inverter 50 is excessive. As a result, the change rate of the current flowing through the motor generator 10 increases, and noise may increase.

  FIG. 4 shows the effect of this embodiment in comparison with the conventional example.

  In the present embodiment in which the power supply voltage VDC is “20 V”, the current change rate can be reduced as compared with the conventional example in which the power supply voltage VDC is “100 V”. For this reason, the harmonic component of the Fourier component (current FFT) of the current is significantly suppressed in the present embodiment as compared with the conventional example. Incidentally, in FIG. 4, the current waveform of the conventional example is approximated by a rectangular wave, and the Fourier series expansion in the case of approximating the current waveform of the present embodiment by a triangular wave is used.

  Although FIG. 4 shows an example in which only the high frequency voltage is the output voltage of the inverter 50, the behavior of the high frequency current signal even when the high frequency voltage is superimposed on the control amount control voltage. Would be the same. Incidentally, in FIG. 4, the pulse width of the present embodiment is larger due to the processing of the PWM processing unit 44. That is, since the ratio of the power supply voltage VDC to the amplitude of the high-frequency voltage vector is larger in this embodiment, the pulse width is larger. The norm of the high-frequency voltage vector (vhα, vhβ) is set to a level at which a high-frequency current signal can be detected with high accuracy, and is normally a fixed value.

  FIG. 5 shows the relationship between the input voltage (power supply voltage) of the inverter 50 and the intensity of the audio signal in the audible frequency range. As shown in the figure, the strength of the audio signal decreases as the power supply voltage decreases.

  As described above, in the present embodiment, by limiting the input voltage of the inverter 50 as much as possible, when the electrical angle is estimated using the high-frequency voltages vhα and vhβ, the input voltage is set to an allowable maximum value (in a high rotation speed region). The maximum value that can be taken) can be made smaller. For this reason, compared with the case where the input voltage of the inverter 50 utilized when estimating an electrical angle with another method is applied also at the time of the superimposition process of a high frequency voltage, a noise can be reduced suitably.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) The converter 54 is operated so that the larger one of the combined voltages of the control voltage of the motor generator 10 and the high-frequency voltages vhα and vhβ matches the input voltage of the inverter 50. Thereby, the change rate of the current signal can be reduced as much as possible while avoiding a decrease in controllability of the control amount.

  (2) A command voltage (output of the adders 38 and 40) in the two-dimensional coordinate system for the motor generator 10 is input, and the converter 54 is operated based on the vector norm. Thereby, the process which restrict | limits an input voltage based on the one with the larger said synthetic | combination voltage can be performed based on an appropriate input parameter.

(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 6 shows the overall configuration of the control system according to the present embodiment. In FIG. 6, processes corresponding to the processes shown in FIG. 1 are given the same reference numerals for convenience.

  In the present embodiment, the converter 54 is operated based on the torque of the motor generator 10 and the electrical angular velocity ω. That is, the power supply voltage changer 70 receives the required torque Tr and the electrical angular velocity ω, and generates a power supply voltage command value VDCr according to the input. Here, since the voltage to be applied to the motor generator 10 increases as the torque of the motor generator 10 increases, the torque is a parameter having a correlation with the magnitude of the voltage for controlling the control amount. Further, since the induced voltage increases as the electrical angular velocity ω increases, it is necessary to increase the applied voltage as the electrical angular velocity ω increases. Therefore, the electrical angular velocity ω is a parameter having a correlation with the magnitude of the output voltage of the inverter 50.

  According to the present embodiment described above, the following effects can be obtained in addition to the effect (1) of the first embodiment.

  (3) The converter 54 is operated with the electric angular velocity ω and torque of the motor generator 10 as inputs. Thereby, the process which restrict | limits an input voltage based on the one where the said synthesized voltage is larger can be performed appropriately.

(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 7 shows the overall configuration of the control system according to the present embodiment. In FIG. 7, processes corresponding to the processes shown in FIG. 1 are given the same reference numerals for convenience.

  In the present embodiment, the converter 54 is operated so as to feedback control the noise level resulting from the superposition of the high-frequency voltages vhα and vhβ within an allowable range. That is, the sound level detector 80 outputs the intensity of frequency noise in the audible frequency region (amount quantified by the magnitude of amplitude) to the power supply voltage changer 70 as a detection signal. The power supply voltage changer 70 sets a power supply voltage command value VDCr as an operation amount for feedback control of the noise intensity within an allowable range.

  According to the present embodiment described above, the following effects can be obtained in addition to the effect (1) of the first embodiment.

  (4) The input voltage of the inverter 50 was manipulated so that the detected value of the audio signal generated with the superposition of the high-frequency voltages vhα and vhβ was within the allowable range. Thereby, it is possible to reliably avoid an excessive increase in the audio signal.

(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 8 shows the overall configuration of the control system according to the present embodiment. In FIG. 8, processes corresponding to the processes shown in FIG.

  In the present embodiment, the electrical angle θ is estimated by using the outer product value of the high-frequency voltage vector (vhα, vhβ) and the current signal vector (ihα, ihβ) as an error parameter. That is, the position estimator 64a receives the outer product value calculated by the outer product value calculation unit 82, and operates the electrical angle θ as an operation amount for feedback control of the outer product value to zero. As a result, the electrical angle θ is manipulated so that the phase angles of the high-frequency voltage vectors (vhα, vhβ) and the current signal vectors (ihα, ihβ) coincide.

  According to this embodiment described above, the following effects can be obtained in addition to the effects (1) and (2) of the first embodiment.

  (5) By using the outer product value as an error parameter and calculating the electrical angle θ so that it becomes zero, the electrical angle θ can be estimated without performing an arctangent function calculation or the like when estimating the electrical angle θ. .

(Fifth embodiment)
Hereinafter, a fifth embodiment will be described with reference to the drawings, focusing on differences from the first embodiment.

  FIG. 9 shows the overall configuration of the control system according to the present embodiment. In FIG. 9, processes corresponding to the processes shown in FIG. 1 are given the same reference numerals for convenience.

  In the present embodiment, a two-phase modulation unit 84 is provided, and two-phase modulation processing of command voltages vur, vvr, and vwr is performed. That is, the largest one of the command voltages vur, vvr, and vwr matches the potential of the positive input terminal of the inverter 50 while maintaining the same voltage difference (line voltage) between the two phases of the command voltages vur, vvr, and vwr. Or the smallest one of the command voltages vur, vvr, vwr is made to coincide with the potential of the negative input terminal of the inverter 50. The PWM processing unit 44 generates an operation signal for each switching element SW of the inverter 50 based on the magnitude of the two-phase modulated command voltage and the carrier.

  According to the command voltage that has been subjected to the two-phase modulation process, the same current that can be passed to the motor generator 10 by the command voltage before the two-phase modulation can be supplied. In addition, when the two-phase modulation process is performed, the maximum value of the line voltage of the motor generator 10 can be increased. For this reason, in the present embodiment, as shown in FIG. 10B, the setting of the power supply voltage VDC is performed by high-frequency voltage vectors vh (+ d), vh (−) with the positive and negative directions of the d-axis as phase angles. d) is set to a value obtained by multiplying the larger of the norms of a pair of vectors formed by the sum of the control amount control voltage vector va (synthesized voltage vector) by the square root of “2”.

  Thus, according to the present embodiment, by performing the two-phase modulation process, it is possible to further reduce the input voltage of the inverter IV while maintaining the controllability of the control amount. For this reason, the noise resulting from the superposition of the high-frequency voltages vhα and vhβ can be further reduced.

(Sixth embodiment)
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 11 shows a method for applying the high-frequency voltages vhα and vhβ according to the present embodiment. As shown in the figure, in the present embodiment, a process of applying a plurality of pulse voltages in the positive direction of the d-axis and a process of applying a plurality of pulse voltages in the negative direction of the d-axis are periodically repeated. Thus, a process of applying a high-frequency voltage that oscillates between the positive and negative directions of the d-axis is performed.

  Thus, as compared with the case where the number of pulse voltages is 1 (FIG. 11B), the period in which the voltage is applied in the d-axis positive direction and the period in which the voltage is applied in the d-axis negative direction The average voltage in each can be reduced. For this reason, since the change speed of the average value of the current in the microscopic time scale of about twice the pulse width can be reduced, the noise can be reduced also by this.

  In particular, in the present embodiment, the pulse width and the pulse interval are set to be equal to each other, thereby suppressing fluctuations in the change speed of the average value of the current in the microscopic time scale.

  Here, also in this embodiment, when the carrier of the PWM processing unit 44 is a triangular wave signal having the same gradually increasing speed and gradually decreasing speed, the d-axis positive voltage and d A signal to which the negative voltage of the axis is assigned may be a high frequency voltage. That is, for example, the d-axis positive direction voltage is a voltage (a plurality of pulse voltages in the d-axis positive direction) that periodically switches between the d-axis positive direction voltage and the zero voltage during the gradual increase period. The voltage may be a voltage (a plurality of pulse voltages in the d-axis negative direction) at which the d-axis negative direction voltage and the zero voltage are periodically switched during the gradual decrease period.

  At this time, one method for equalizing the pulse width and the pulse interval is to divide the ON command signal of the switching element SW, which is a signal generated by the PWM processing unit 44 shown in FIG. This is a method of processing so as to sandwich an off command signal.

  The pulse interval may be set longer than the pulse width. Thereby, the change speed of the average value of the current in the microscopic time scale can be further reduced.

  According to the embodiment described above, the following effects can be obtained.

  (6) The high frequency voltage was superimposed by periodically switching between a plurality of pulses as a voltage in the d-axis positive direction and a plurality of pulses as a voltage in the d-axis negative direction. Thereby, the average value of these pair of voltages can be reduced. For this reason, the change rate of the current actually propagated through motor generator 10 can be reduced by the pair of voltages.

  (7) By making the pulse width and pulse interval constant, it is possible to suppress fluctuations in the current change rate.

(Other embodiments)
Each of the above embodiments may be modified as follows.
<About the rotation angle estimator>
The rotation angle estimator is not limited to the one that receives the αβ component of the high-frequency voltage and the αβ component of the detected value of the high-frequency current (including the one that uses the outer product value as an error parameter). For example, it may be one that receives the dq component of the high-frequency voltage and the dq component of the detected value of the high-frequency current (including a product whose outer product value is the error parameter).

  Further, the present invention is not limited to estimating the electrical angle based on the angular difference between the vibration direction of the high-frequency current that actually propagates through the rotating machine when the high-frequency voltage is superimposed and the superposition direction of the high-frequency voltage. For example, as described in JP 2008-125260 A, the electrical angle may be estimated based on the amplitude of the high-frequency current.

Furthermore, the operation target for feedback-controlling the detected error (error parameter such as outer product value) to zero is not limited to the electrical angle θ but may be the electrical angular velocity ω. In this case, the rotation angle estimator may be configured to include, for example, an estimator that estimates the electrical angular velocity ω based on the error parameter, and a unit that calculates the electrical angle θ by integrating the estimated electrical angular velocity ω. Good.
<Input voltage of inverter IV>
The input voltage of the inverter IV is not limited to one that matches the larger one of the combined vector norms of each of the pair of high-frequency voltage vectors and the voltage vector for controlling the control amount. For example, it may be larger by a specified value than the larger one. Here, it is desirable that the specified value be as small as possible (for example, several V). In this case, it is also possible to set the prescribed value as the feedback operation amount of the sound level with zero as the lower limit. Here, the feedback control can be performed in the manner of the third embodiment.

In the sixth embodiment, the converter 54 may not be provided.
<About frequency signals>
The high frequency voltage is not limited to one whose phase angle periodically changes in the positive and negative directions of the d-axis. For example, as described in Japanese Patent Application Laid-Open No. 2009-148017, the vibration direction may be variably set according to the operating state of the motor generator 10. Furthermore, it is not limited to the variable setting, but may be fixed to a specific phase angle other than the positive direction and negative direction of the d-axis. Even in this case, for example, the rotation angle can be estimated by calculating the rotation angle so as to reduce the difference between the norm of the actually propagated high-frequency current vector and its target value.

Further, as the one that periodically switches between a plurality of pulses as a voltage having an arbitrary phase angle and a plurality of pulses as a voltage having a phase angle that is 180 degrees different from this, it is exemplified in the sixth embodiment. It is not limited to what you did. For example, the plurality of pulses may be two pulses, three pulses, or five or more pulses. Further, the pulse width and pulse interval of each pulse are not equal to each other, and the pulse interval is not necessarily longer than the pulse width.
<Others>
The final control amount of the motor generator 10 is not limited to torque, and may be, for example, a rotational speed. Further, the present invention is not limited to performing current vector control, and for example, torque feedback control may be performed.

  -You may change the said 4th, 5th embodiment by the change of 3rd Embodiment with respect to the said 1st Embodiment. In the sixth embodiment, the sound level may be feedback-controlled as in the third embodiment. In this case, the operation amount for feedback control may be the pulse width of the pulse signal. In such cases, when the pulse width is less than or equal to the specified value, the total amount of change in current due to the change in pulse width can be set by variably setting the number of pulses according to the pulse width, such as increasing the number of pulses. It is desirable to suppress fluctuations in

  -Due to the structure, the electric motor having saliency is not limited to the motor generator 10 described above. For example, a synchronous reluctance motor (SynRM) may be used.

  In each of the above embodiments, the control device according to the present invention is applied to a hybrid vehicle. However, the present invention is not limited to this, and may be applied to, for example, an electric vehicle. Furthermore, the control device of the present invention may be applied to an electric motor as power transmission means such as power steering in a vehicle using an internal combustion engine as a power source.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator (one Embodiment of the rotating machine which has saliency), 50 ... Inverter, 54 ... Converter, 70 ... Power supply voltage changer.

Claims (7)

When controlling a control amount of the rotating machine by operating a power conversion circuit including a switching element that selectively connects a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine having saliency, the electrical angle of the rotating machine A voltage having a frequency different from the frequency and having an arbitrary phase angle and a frequency signal in which a voltage having a phase angle different from the phase angle by 180 degrees is periodically switched are superimposed on the output voltage of the power conversion circuit. In the control device for a rotating machine that estimates the rotation angle of the rotating machine based on the current signal that is actually propagated through the rotating machine with the superposition,
With limiting means for limiting the rate of change of the current signal that actually propagates through the rotating machine with the superposition ,
The limiting means sets a phase angle that is 180 degrees different from the plurality of pulses as the voltage having the arbitrary phase angle in order to limit the change speed of the current signal that actually propagates through the rotating machine with the superposition. A control device for a rotating machine, characterized in that it is means for periodically switching between a plurality of pulses as a voltage it has. When controlling a control amount of the rotating machine by operating a power conversion circuit including a switching element that selectively connects a positive electrode and a negative electrode of a DC power source to a terminal of the rotating machine having saliency, the electrical angle of the rotating machine A voltage having a frequency different from the frequency and having an arbitrary phase angle and a frequency signal in which a voltage having a phase angle different from the phase angle by 180 degrees is periodically switched are superimposed on the output voltage of the power conversion circuit. In the control device for a rotating machine that estimates the rotation angle of the rotating machine based on the current signal that is actually propagated through the rotating machine with the superposition,
With limiting means for limiting the rate of change of the current signal that actually propagates through the rotating machine with the superposition,
The DC power supply is an output terminal of a converter for making the input voltage of the power conversion circuit variable,
The limiting means is a control voltage that is a voltage applied to the rotating machine in order to control a control amount of the rotating machine in order to limit a change rate of a current signal that actually propagates through the rotating machine with the superposition. The larger value of the combined voltage of the voltage and the voltage having the arbitrary phase angle and the combined voltage of the control voltage and the voltage having the phase angle different by 180 degrees is subtracted from the input voltage of the power conversion circuit. controller times turning point characterized in that values to limit the input voltage by operating the converter to be equal to or less than the specified value. The limiting means sets a phase angle that is 180 degrees different from the plurality of pulses as the voltage having the arbitrary phase angle in order to limit the change speed of the current signal that actually propagates through the rotating machine with the superposition. 3. The control device for a rotating machine according to claim 2, wherein the control device is means for periodically switching between a plurality of pulses as a voltage having the voltage. The limiting unit is configured to convert a power of the control voltage and a composite voltage of the voltage having an arbitrary phase angle and a composite voltage of the control voltage and the voltage having a phase angle different from 180 degrees by the power conversion. 4. The control device for a rotating machine according to claim 2 , wherein the input voltage is limited by operating the converter so that the input voltages of the circuits are matched. Means for setting a command voltage for controlling the control amount;
Means for calculating a norm of the frequency signal superimposed on the set command voltage;
5. The control of a rotating machine according to claim 2, wherein the limiting unit performs the limitation of the input voltage by operating the converter with the calculated norm as an input. apparatus. It said limiting means limits the input voltage, one of the claims 2 to 4, characterized in that by operating the converter at least one of the torque of the electric angular speed and the rotating machine of the rotating machine as an input The control apparatus of the rotary machine of Claim 1 .   The limiting means limits the change rate of the current signal that is actually propagated through the rotating machine in accordance with the superposition so that the detected value of the audio signal generated along with the superposition of the frequency signal is within an allowable range. The control device for a rotating machine according to any one of claims 1 to 6.