JP5573714B2 - Rotating machine control device - Google Patents

Description

  The present invention provides a switching element for selectively connecting a terminal of a rotating machine having saliency to each of a positive electrode and a negative electrode of a DC voltage source and a DC / AC converter circuit including a diode connected in reverse parallel to the switching element. In controlling the control amount of the rotating machine, superimposing means for superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the DC-AC converter circuit, and the superimposed high-frequency voltage signal And a estimator that estimates a rotation angle of the rotating machine based on a detected value of a high-frequency current signal flowing through the rotating machine.

  As this type of control device, for example, as shown in Patent Document 1 below, when a high-frequency voltage signal that vibrates in the positive and negative directions of the estimated d-axis of a three-phase motor is applied, the high-frequency that actually propagates to the motor Some have also been proposed that estimate the electrical angle of an electric motor based on a current signal.

Japanese Patent No. 3312472

  By the way, since the frequency of the high-frequency voltage signal is usually within an audible frequency band, noise perceived by a person when estimating the electrical angle may occur. In order to reduce this noise, it is effective to reduce the high-frequency voltage signal. However, in this case, the inventors have found that the estimation accuracy of the electrical angle is lowered.

  The present invention has been made in the process of solving the above-mentioned problems, and its object is to detect by superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the DC-AC converter circuit. Another object of the present invention is to provide a new control device for a rotating machine capable of estimating the rotation angle of the rotating machine based on the detected value of the high-frequency current signal.

  Hereinafter, means for solving the above-described problems and the operation and effect thereof will be described.

According to a first aspect of the present invention, there is provided a switching element for selectively connecting a terminal of a rotating machine having saliency to each of a positive electrode and a negative electrode of a DC voltage source, and a DC / AC conversion circuit including a diode connected in reverse parallel to the switching element. When controlling the control amount of the rotating machine by operation, superimposing means for superimposing a high frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the DC / AC converter circuit, and the superimposed high frequency In a control device for a rotating machine comprising: estimation means for estimating a rotation angle of the rotating machine based on a detected value of a high-frequency current signal flowing through the rotating machine according to a voltage signal; and the switching element connected to the positive electrode and the negative electrode One of the switching elements connected to the other and the other from the state where the other is turned on and off, respectively When switching to a state in which each is turned off and on, a dead time period in which both are turned off is provided, and the rotating machine includes a plurality of sets of stators connected to each other at a neutral point, and the estimation The means uses the detected value of the high-frequency current signal for each of at least two of the plurality of sets of the stators for estimation of the rotation angle.

  When the DC / AC converter circuit is used, the voltage applied to the terminal of the rotating machine during the dead time period depends on the polarity of the current flowing through the terminal. The voltage applied to the terminal of the rotating machine during this time can be an error with respect to the high frequency voltage signal intended to be superimposed by the superimposing means. The ratio of the error voltage to the high frequency voltage signal increases as the high frequency voltage signal is reduced. For this reason, the smaller the high-frequency voltage signal, the greater the error that the high-frequency voltage signal actually superimposed has on the intended one.

  In the above invention, by using the detection value of the high-frequency current signal for each of at least two sets of the set of stators, one having a small error due to the dead time period can be selectively used, or a plurality of By simultaneously using a set of high-frequency current signals, the influence of errors can be reduced.

According to a second invention, in the first invention, the dead time compensation compensates for an error caused by the dead time by shifting the start point and the end point of the ON operation command period of the operation signal of the DC / AC converter circuit by the same time. And a stator phase separation means for differentiating the phase of the current flowing through the stator with respect to at least a pair of the plurality of sets, and the estimation means is configured to estimate the rotation angle of the stator. Of the plurality of sets, the degree of use of the detected value of the high-frequency current relating to the absolute value of the current flowing through the stator as the constituent element being equal to or greater than the predetermined value It comprises a selective use means for making the degree of use larger than the degree of use.

  In the above invention, by having a dead time compensation function, it is possible to avoid the occurrence of an error in the high-frequency voltage signal due to the dead time when there is nothing that zero-crosses the current (phase current) flowing through the terminal. . And even if there is something that crosses zero, by increasing the utilization of the high-frequency current signal related to other than the set of stators, the high-frequency current signal affected by the error of the high-frequency voltage signal becomes the estimated value of the rotation angle. Therefore, the influence of the error of the high-frequency voltage signal due to the dead time can be suitably reduced.

According to a third invention, in the second invention, the selection and use means prohibits the use of the detected value of the high-frequency current signal relating to an input parameter for estimating the rotation angle that is less than or equal to the predetermined value. Means are provided.

  In the above invention, since the high-frequency current signal affected by the error of the high-frequency voltage signal is not used for the estimation of the rotation angle by including the prohibiting means, in the situation where the error of the high-frequency voltage signal due to the dead time has not occurred. The rotation angle can be estimated.

According to a fourth invention, in the second or third invention, the stator phase separation means is configured by making the directions of vectors determined by the stators different from each other for the at least one pair. To do.

According to a fifth invention, in the second or third invention, the plurality of sets include those in which directions of vectors determined by the stators coincide with each other, and the stator phase separation means It is characterized by comprising vector phase spreading means for making the phases of flowing current vectors different from each other.

In a sixth aspect based on any one of the second to fifth aspects, the predetermined value is set to be equal to or greater than a ripple current value accompanying switching of a switching state of a switching element constituting the DC / AC converter circuit. It is characterized by.

  When the current flowing through the rotating machine becomes zero, in the microscopic time scale, the current is actually inverted from negative to positive due to the ripple current accompanying switching of the switching state of the switching element. This phenomenon causes an error of the high-frequency voltage signal due to the dead time period. In the above invention, in view of this point, the above setting makes it possible to reliably identify those in which an error occurs in the high-frequency voltage signal due to the dead time period.

According to a seventh invention, in any one of the second to fifth inventions, the predetermined value is set to be equal to or larger than an amplitude value of the high-frequency current signal.

  When the current flowing through the rotating machine becomes zero, in the microscopic time scale, the current is actually inverted from negative to positive due to the ripple current accompanying switching of the switching state of the switching element. This phenomenon causes an error of the high-frequency voltage signal due to the dead time period. Here, the high-frequency current signal is formed by a ripple current. In the above invention, in view of this point, the above setting makes it possible to reliably identify those in which an error occurs in the high-frequency voltage signal due to the dead time period.

According to an eighth invention, in any one of the first to seventh inventions, there is provided disconnection detecting means for detecting presence or absence of disconnection of an electrical path between each of the plurality of sets of stators and the DC-AC converter circuit. In addition, when the disconnection detecting unit detects a disconnection, the estimation unit uses the detected value of the high-frequency current signal for the one of the plurality of sets in which the disconnection is not detected for the estimation of the rotation angle. It is characterized by.

  In the said invention, even if it is a case where a disconnection arises, a rotation angle can be estimated.

1 is a system configuration diagram according to a first embodiment. FIG. The figure which shows arrangement | positioning of the stator concerning the embodiment. The block diagram regarding the estimation process of the rotation angle concerning the embodiment. The time chart which shows the PWM process concerning the embodiment. The time chart which shows the dead time compensation process concerning the embodiment. The time chart explaining the error which arises in the high frequency voltage signal concerning the embodiment. The time chart explaining the error which arises in the high frequency voltage signal concerning the embodiment. The vector figure which shows the error which arises in the high frequency voltage signal concerning the embodiment. The time chart which shows the estimation accuracy of the rotation angle at the time of making a high frequency voltage small. The flowchart which shows the procedure of the selection process of the high frequency current signal concerning the said embodiment. The block diagram regarding the estimation process of the rotation angle concerning 2nd Embodiment. The system block diagram concerning 3rd Embodiment. The figure which shows the arrangement | positioning of the stator and setting of command electric current concerning the embodiment. The block diagram regarding the estimation process of the rotation angle concerning the embodiment. The block diagram regarding the estimation process of the rotation angle concerning 4th Embodiment. The block diagram regarding the estimation process of the rotation angle concerning 5th Embodiment. The flowchart which shows the procedure of the selection process of the high frequency current signal concerning 6th Embodiment.

<First Embodiment>
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a rotating machine as an in-vehicle main machine will be described with reference to the drawings.

  FIG. 1 shows a system configuration diagram according to the present embodiment.

  The motor generator 10 is a permanent magnet synchronous motor. The motor generator 10 is a rotating machine (saliency pole machine) having saliency. Specifically, the motor generator 10 is an embedded magnet synchronous motor (IPMSM). Motor generator 10 includes two sets of three-phase stators connected at neutral points (U phase, V phase and W phase, and X phase, Y phase and Z phase).

  The set of U, V, and W phases is connected to the high voltage battery 12 via the inverter IV1. Here, inverter IV1 includes three sets of series connection bodies of switching elements S * p, S * n (* = u, v, w), and the connection point of each series connection body is the U of motor generator 10. , V and W phases. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements S * p and S * n. In addition, diodes D * p and D * n are connected in antiparallel to these.

  The set of X, Y, and W phases is connected to the high voltage battery 12 via the inverter IV2. Here, inverter IV2 is provided with three sets of series connection bodies of switching elements S * p, S * n (* = x, y, z), and the connection point of these series connection bodies is the X of motor generator 10. , Y and Z phases are connected to each other. In the present embodiment, insulated gate bipolar transistors (IGBTs) are used as the switching elements S * p and S * n. In addition, diodes D * p and D * n are connected in antiparallel to these.

  FIG. 2A shows a geometrical arrangement relationship between the U, V, and W phases and the X, Y, and Z phases. As shown in the figure, in this embodiment, an X phase, a Y phase, and a Z phase are respectively present between the U phase and the V phase, between the V phase and the W phase, and between the W phase and the U phase. Allocated. That is, these U, V, W, X, Y, and Z phases are arranged so that vectors corresponding to each other are not parallel. For this reason, when the current flowing through the motor generator 10 is controlled to the command currents idr and iqr, the phases of the phase currents are different from each other as shown in FIG.

  The control device 14 shown in FIG. 1 controls the control amount of the motor generator 10 by operating the inverters IV1 and IV2. The control device 14 detects a current sensor that detects a current i * (* = u, v, w, x, y, z) flowing through each phase of the motor generator 10 and an input voltage (power supply voltage VDC) of the inverter IV. An operation signal for operating the inverter IV is generated and output based on the detected value of the voltage sensor. Here, signals for operating the switching elements S * p and S * n of the inverters IV1 and IV2 are the operation signals g * p and g * n.

  FIG. 3 shows processing performed by the control device 14. In the following, first, “control amount control” will be described, and then “rotation angle estimation processing” will be described.

`` Control amount control ''
The command current setting unit 21 sets a command current idr on the d axis and a command current iqr on the q axis, which are current command values of the rotating two-phase coordinate system, based on the required torque Tr. On the other hand, currents iu, iv, and iw flowing through the U, V, and W phases of motor generator 10 are, in dq conversion unit 20a, real current id and q axis on the d axis, which are real currents in the rotating two-phase coordinate system. It is converted into the above actual current iq. The actual currents id and iq output from the dq converter 20a are filtered by the low-pass filters 22a and 24a, respectively. The deviation calculator 26a calculates the difference between the d-axis command current idr and the actual current id (output of the low-pass filter 22a), and the deviation calculator 28a calculates the q-axis command current iqr and the actual current iq (low-pass filter 24a). Output). The current controller 30a feedback-controls the command voltage vdr on the d-axis as an operation amount for feedback-controlling the actual current id on the d-axis to the command current idr, and the actual current iq on the q-axis to the command current iqr. A command voltage vqr on the q-axis is calculated as an operation amount for this. Here, the calculation is performed by adding the output of the proportional element and the output of the integral element.

  The three-phase conversion unit 32a converts the command voltages vdr and vqr in the rotating two-phase coordinate system into three-phase command voltages vur, vvr and vwr, and normalizes them with the power supply voltage VDC, whereby the duty signal Du. , Dv, Dw are calculated. The dead time compensation unit 34a calculates dead time correction amounts Δvu, Δvv, Δvw for feedforward correction of the duty signals Du, Dv, Dw based on the corresponding phase currents iu, iv, iw. In each of the correction units 36a, 38a, and 40a, the duty signals Du, Dv, and Dw are corrected based on the dead time correction amounts Δvu, Δvv, and Δvw, respectively. The operation signal generation unit 42a generates an operation signal g * # (* = u, v, w; # = p, n) by PWM processing based on the magnitude comparison between the duty signals Du, Dv, Dw and the carrier.

  FIG. 4 shows details of processing by the operation signal generation unit 42a. In the present embodiment, based on the magnitude comparison between the triangular wave carrier CS in which the gradual increase speed and the gradual decrease speed are the same and the gradual increase period and the gradual decrease period are the same, and the duty signals Du, Dv, Dw of each phase, PWM signals gu, gv, and gw are generated. Based on the PWM signal g * (* = u, v, w), an upper arm operation signal g * p and a lower arm operation signal g * n are generated. At this time, by performing dead time generation processing, the operation signal g * # (* = u, v, w; # = p, n) is delayed in the rising timing by the dead time DT with respect to the PWM signal g *. Will be. Note that the update cycle of the duty signals Du, Dv, Dw (command voltages vur, vvr, vwr) is made to coincide with the update cycle of the carrier CS. More specifically, in the present embodiment, the duty signals Du, Dv, Dw are updated at the timing when the carrier CS reaches a peak.

  FIG. 5 shows details of the processing of the dead time compensation unit 34a.

  As shown in FIG. 5A, when the phase current i * (* = u, v, w) is positive, current flows through the lower arm diode D * n during the dead time period. The ON period of the operation signal g * p is shorter than the ON period of the PWM signal g * by the dead time DT, and the rising edge is delayed by the dead time DT. For this reason, the dead time compensation unit 34 corrects the duty signal D * to be increased by the dead time correction amount Δv *, so that both the rising edge and the falling edge of the PWM signal g * are “½ of the dead time DT. Correct by one by one. Thereby, the ON period of the operation signal g * p can be matched with the ON period of the PWM signal g * before correction, and the delay amount of the rising edge can be halved.

  As shown in FIG. 5B, when the phase current i * (* = u, v, w) is negative, current flows through the upper arm diode D * p during the dead time period. The on period of the operation signal g * p is longer by the dead time DT than the on period of the PWM signal g *. Therefore, the dead time compensation unit 34 corrects the duty signal D * to be decreased by the dead time correction amount Δv *, so that both the rising edge and the falling edge of the PWM signal g * are “½ of the dead time DT. Correct by one by one. Thereby, the ON period of the operation signal g * p can be matched with the ON period of the PWM signal g * before correction. However, at this time, the rising edge of the operation signal g * p is delayed by “½” of the dead time DT with respect to the rising edge of the PWM signal g * before correction.

  As shown in FIG. 5C, when the phase current i * (* = u, v, w) reverses from negative to positive during the period from the rising edge to the falling edge of the PWM signal g *, the dead corresponding to the rising edge During the time period, current flows through the upper arm diode D * p, and during the dead time period corresponding to the falling edge, current flows through the lower arm diode D * n. For this reason, the ON period of the operation signal g * p coincides with the ON period of the PWM signal g *. Therefore, in this case, the dead time correction amount Δv * is set to zero.

  As shown in FIG. 3, the currents flowing through the X, Y, and Z phases are also feedback controlled to the command currents idr and iqr. Regarding the processing related to this, in the above description, the letter a assigned to the code is b.

"Rotation angle estimation process"
The high frequency voltage signal setting unit 52 shown in FIG. 3 sets the high frequency voltage command signal Vhr = (vdhr, vqhr). Here, in this embodiment, vqhr = 0 and vdhr is a signal that reverses its polarity every half cycle of the PWM processing. The superimposing unit 44a corrects the d-axis command voltage vdr output from the current controller 30a with the d-axis component vdhr of the high-frequency voltage command signal and outputs the corrected signal to the three-phase conversion unit 32a.

  On the other hand, the high pass filter 50a extracts harmonic components (high frequency current signals idh, iqh) from the actual currents id, iq output from the dq converter 20a. Here, the high frequency component is a component having a higher frequency than the fundamental wave component. In particular, here, the same frequency component as the high-frequency voltage command signal Vhr is extracted. As the high-pass filter 50a, for example, a means for outputting a difference between values before and after a half cycle of the PWM signal for the actual currents id and iq may be used.

  The outer product value calculation unit 54a calculates the outer product value of the high-frequency voltage command signal Vhr and the high-frequency current signals idh and iqh. This outer product value has a correlation with the angle between the vectors of the high-frequency voltage signal and the high-frequency current signals idh and iqh, and is a parameter (angle correlation amount) having a correlation with the rotation angle of the motor generator 10. In particular, in this embodiment, the error correlation amount has a correlation with the error of the rotation angle θ. The outer product value as the error correlation amount is input to the speed calculation unit 58. The speed calculation unit 58 calculates the electrical angular speed ω as the sum of a proportional element and an integral element with the outer product value as an input. Then, the angle calculation unit 60 calculates the rotation angle θ as a time integral value of the electrical angular velocity ω. As a result, the rotation angle θ becomes an operation amount for feedback control of the outer product value to its target value of zero.

  The target value of the outer product value is zero because the motor generator 10 is IPMSM, and therefore the d-axis inductance Ld is smaller than the q-axis inductance Lq. That is, in this case, if the high-frequency voltage in the d-axis direction is superimposed on the voltage for controlling the control amount as the output voltage of the inverter IV, the high-frequency current signal is also in the d-axis direction, and the outer product value becomes zero. When the outer product value is not zero, the rotation angle θ is manipulated so that the outer product value becomes zero, and the rotation angle θ matches the correct angle.

  Note that, as shown in FIG. 3, the process of estimating the rotation angle θ based on the current flowing through the X, Y, and Z phases is also executed. In this process, in the above description, the letter a assigned to the code is b.

  By the way, if the ratio of the change amount of the ON time and OFF time of the operation signal g * # due to the superposition is reduced by decreasing the high frequency voltage signal, it is actually superimposed. The error due to the dead time DT of the high-frequency voltage signal becomes large, and as a result, the estimation accuracy of the rotation angle θ is lowered. Such an error is caused by providing the dead time compensation units 34a and 34b, so that the phase current i * (* = u, v, w, x, y, z) is generated in the period from the rising edge to the falling edge of the PWM signal g *. This can be avoided except in the case of inversion from negative to positive. This is because, as shown in FIG. 4, the ON period of the operation signal g * # is defined by the PWM signal g * by the compensation by the dead time compensators 34a and 34b, and the phase is “DT / 2. This is because the line voltage matches that specified by the PWM signal g * before correction. In other words, this case is equivalent to the case where the phase of the carrier CS is delayed by “DT / 2” and PWM processing is performed, and no error occurs in the line voltage.

  However, if the phase current i * (* = u, v, w, x, y, z) is inverted from negative to positive during the period from the rising edge to the falling edge of the PWM signal g *, the operation signal for that phase Since the phase of g * n is not delayed, it is equivalent to advancing only that phase by “DT / 2” compared to the other phases. For this reason, in this case, the line voltage deviates from that defined by the PWM signal g * before correction, and an error occurs in the high-frequency voltage signal. FIG. 6 shows the PWM signal g * after processing by the dead time compensation unit 34. In the example shown in the figure, the U phase has a zero-cross period, and in this case, only the U phase is in the same state as when the voltage is advanced by “DT / 2”. In other words, when the PWM process is performed by retarding the phase of the carrier CS by “DT / 2”, it is equivalent to the voltage of only the U phase being advanced by “DT / 2”. As a result, as indicated by the one-dot chain line in the upper part of the figure, if the high-frequency voltage signal (vdh) is sequentially superimposed in the positive and negative directions of the U-axis every PWM half cycle, As indicated by the two-dot chain line, the amplitude of the actually superimposed high-frequency voltage signal increases. On the other hand, if the high-frequency voltage signal (vdh) is sequentially superimposed in the negative and positive directions of the U-axis every PWM half cycle, the amplitude of the actually superimposed high-frequency voltage signal is reduced, which is the worst. As shown in FIG. 7, the polarity of the high frequency voltage signal is inverted. Note that what is indicated by a one-dot chain line in FIGS. 6 and 7 is precisely the d-axis component vdhr of the high-frequency voltage command signal normalized by the power supply voltage VDC.

  For this reason, as shown in FIG. 8, when the phase current zero-crosses, the high frequency voltage signal Vh actually superimposed on the high frequency voltage command signal Vhr has an error. FIG. 9 shows the time change of the angular correlation amount (outer product value) calculated according to the high-frequency current signal. As shown in the figure, the amount of angular correlation largely fluctuates near the zero cross of the phase current. In the figure, the fluctuation of the angular correlation amount outside the vicinity of the zero cross of the phase current shown in the figure corresponds to the zero cross period of the other phase not shown.

  In order to avoid such a problem, in the present embodiment, the high-frequency current signal included in the current (iu, iv, iw) flowing through the U, V, and W phases and the current (ix, iy, Among the high-frequency current signals included in iz), a process of switching the current used for the process of estimating the rotation angle θ is performed. That is, the output of the outer product value calculation units 54 a and 54 b shown in FIG. 3 is output to the speed calculation unit 58 via the selector 56, and the switching process can be performed by switching the selector 56. It has become.

  FIG. 10 shows the procedure of the high-frequency current signal selection process according to this embodiment. This process is repeated and executed by the control device 14 at a predetermined cycle, for example.

  In this series of processes, first, in step S10, it is determined whether or not the absolute value of the current (iu, iv, iw) flowing through the U, V, and W phases is equal to or greater than a predetermined value it. This process is for determining whether or not to use the high-frequency current signal included in the currents (iu, iv, iw) flowing through the U, V, and W phases for the process of estimating the rotation angle θ. Here, the predetermined value ith may be set to be equal to or greater than the magnitude of the ripple current accompanying switching of the switching state of the inverter IV1. This is a condition for preventing zero crossing of the phase current in the U, V, and W phases. Alternatively, the predetermined value it may be set to be equal to or greater than the amplitude value of the high-frequency current signals idh and iqh.

  If an affirmative determination is made in step S10, the cross product value calculated based on the high-frequency current signal included in the currents (iu, iv, iw) flowing through the U, V, and W phases is adopted in step S12. On the other hand, when a negative determination is made in step S10, the outer product value calculated based on the high-frequency current signal included in the current (ix, iy, iz) flowing through the X, Y, and Z phases is adopted. Here, when a negative determination is made in step S10, it is considered that the minimum absolute value of the current (ix, iy, iz) flowing through the X, Y, and Z phases is equal to or greater than the predetermined value it. This is because the directions of the U, V, and W phases are different from the directions of the X, Y, and Z phases, so the zero cross period of any of the U, V, and W phases and the zero cross of any of the X, Y, and Z phases. This is because the period is different. On the contrary, the currents (ix, iy, iz) that flow through the X, Y, and Z phases when the negative determination is made in step S10 for setting the directions of the U, V, and W phases and the X, Y, and Z directions. Is set so that the minimum value of the absolute value is equal to or greater than a predetermined value ith.

  In addition, when the process of said step S12, S14 is completed, this series of processes are once complete | finished.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) The dead time compensation units 34a and 34b are provided. As a result, when there is no zero crossing, it is possible to avoid an error in the high frequency voltage signal due to the dead time.

(2) The direction of the U, V, and W phases is made different from the direction of the X, Y, and Z phases, and the rotation angle θ is estimated based on the current flowing through the set of phases that do not reach the zero cross period. Thereby, when an error actually occurs in the high-frequency voltage signal due to the dead time, the estimation of the rotation angle θ using the corresponding high-frequency current signals idh and iqh can be avoided.
<Second Embodiment>
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 11 is a block diagram relating to processing of the control device 14 according to the present embodiment. In FIG. 11, the same reference numerals are assigned for convenience to the processing corresponding to the processing shown in FIG.

In the present embodiment, the input parameter of the speed calculation unit 58 is a weighted average value of the outputs of the outer product value calculation units 54a and 54b. That is, the sum of the output of the outer product value calculation unit 54 a multiplied by the weighting factor α and the output of the outer product value calculation unit 54 b multiplied by the weighting factor β is output to the speed calculation unit 58. Here, it is desirable that the weighting coefficients α and β are equal if they are not zero cross periods of the U, V, and W phases and the X, Y, and Z phases. On the other hand, in the zero cross period of any of the U, V, and W phases, the weighting coefficient α is made smaller than the weighting coefficient β (preferably zero), and in any of the zero crossing periods of the X, Y, and Z phases, The weighting factor β is made smaller than the weighting factor α (preferably zero).
<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 12 shows a system configuration according to the present embodiment. In FIG. 12, the same reference numerals are attached for convenience to those corresponding to the members shown in FIG.

  In the present embodiment, each of the U, V, and W phases is matched with each of the X, Y, and Z phases.

  FIG. 13A shows the relationship between the phase of the current vector flowing in the U, V, and W phases and the phase of the current vector flowing in the X, Y, and Z phases. As shown in the drawing, in this embodiment, command values for currents flowing in the U, V, and W phases (command currents idra, iqra) and command values for currents flowing in the X, Y, and Z phases (command currents idrb, iqrb). ) Are different from each other. A combined vector of these current vectors is set to realize minimum current / maximum torque control. As a result, while the minimum current / maximum torque control can be realized for the current flowing through the motor generator 10, the phase of the current flowing through each of the U, V, and W phases, as shown in FIG. , Y and Z phases can be made different from each other.

  FIG. 14 is a block diagram relating to processing of the control device 14 according to the present embodiment. In the process shown in FIG. 14, processes corresponding to the process shown in FIG. 3 are given the same reference numerals for convenience.

  As shown in the figure, the command current setting units 21a and 21b respectively have command values for currents flowing in the U, V, and W phases (command currents idra and iqra) and command values for currents flowing in the X, Y, and Z phases ( Command current idrb, iqrb).

  According to this embodiment described above, in addition to the effect (1) of the first embodiment, the following effect can be obtained.

(3) A combined vector of the command currents idra and iqra related to the U, V, and W phases and the command currents idrb and iqrb related to the X, Y, and Z phases is configured to realize the minimum current / maximum torque control. As a result, the minimum current maximum torque control can be realized while the phase of the current flowing in each of the U, V, and W phases is different from the phase of the current flowing in each of the X, Y, and Z phases. .
<Fourth Embodiment>
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the third embodiment.

  FIG. 15 is a block diagram relating to processing of the control device 14 according to the present embodiment. In the process shown in FIG. 15, processes corresponding to the processes shown in FIG. 3 are given the same reference numerals for convenience.

  As shown in the figure, in the present embodiment, the actual currents id and iq output from the dq converter 20a (specifically, the outputs of the low-pass filters 22a and 24a) and the actual currents id and iq output from the dq converter 20b ( Specifically, the outputs of the low-pass filters 22b and 24b) are combined by the combining units 61 and 62. In the current controller 30 a, the command voltages vdr and vqr are fed back to the command currents idr and iqr set by the command current setting unit 21 in order to feedback the synthesized d-axis actual current and q-axis actual current. Set. In the three-phase conversion unit 32a, when the duty signals Du, Dv, Dw are set based on this, the rotation angle θ is increased and corrected by the offset amount Δ, while in the three-phase conversion unit 32b, the duty signals Du, Dv are used. , Dw, the rotation angle θ is corrected by decreasing the offset amount Δ.

Thus, the current flowing through the motor generator 10 is reduced to the minimum current maximum torque while the phase of the current flowing through each of the U, V, and W phases is different from the phase of the current flowing through each of the X, Y, and Z phases. The current can be controlled for control.
<Fifth Embodiment>
Hereinafter, the fifth embodiment will be described with reference to the drawings with a focus on differences from the fourth embodiment.

  FIG. 16 is a block diagram relating to processing of the control device 14 according to the present embodiment. In the process shown in FIG. 16, the processes corresponding to the process shown in FIG.

In the present embodiment, the superimposing unit 44 superimposes the d-axis component vdhr of the high-frequency voltage command signal on the command voltage vdr, and the superimposing unit 45 superimposes the q-axis component vqhr of the high-frequency voltage command signal on the command voltage vqr. Alternately. Then, in the multiplication value calculation units 70a and 70b, the vector norm of the high-frequency current signals idh and iqh accompanying the process of superimposing the d-axis component vdhr and the high-frequency current signals idh and iqh accompanying the process of superimposing the q-axis component vqhr A multiplication value with the vector norm is calculated. On the other hand, the target value calculation unit 72 calculates the target value of the multiplication value based on the combined current output from the combining units 61 and 62. Then, the rotation angle θ is manipulated to feedback control the multiplication value to the target value. Here, selection of which of the multiplication value calculation units 70a and 70b is used is the same as the method described in the first embodiment. In calculating the target value, the electrical angular velocity ω may be further taken into consideration.
<Sixth Embodiment>
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 17 shows a procedure for selecting a high-frequency current signal according to the present embodiment. This process is repeated and executed by the control device 14 at a predetermined cycle, for example.

  In this series of processes, first, in step S20, it is determined whether or not a disconnection has occurred in any of the U, V, and W phases. Here, the disconnection detection method may be a conventional method. When an affirmative determination is made in step S20, the cross product value calculated based on the current (ix, iy, iz) flowing through the X, Y, and Z phases is used as the fail-safe process in step S22 to estimate the rotation angle θ. Use.

  On the other hand, if a negative determination is made in step S10, it is determined in step S24 whether or not a break has occurred in any of the X, Y, and Z phases. If an affirmative determination is made in step S24, the cross product value calculated based on the currents (iu, iv, iw) flowing through the U, V, W phases is estimated as the fail-safe process in step S26. Use.

In addition, when the process of said step S22, S26 is completed, or when negative determination is made in step S24, this series of processes are once complete | finished. If an affirmative determination is made in steps S20 and S24, it is desirable to suppress errors due to dead time by increasing the vector norm of the high-frequency voltage command signal.
<Other embodiments>
Each of the above embodiments may be modified as follows.

“Selective use”
The parameter used when determining the utilization degree is not limited to the phase current. For example, the rotation angle θ (electrical angle) may be used. That is, since the zero-cross period of each phase has periodicity depending on the rotation angle θ, the zero-cross period can be specified by paying attention to this.

  In the third to fifth embodiments (FIGS. 14 to 16), the final error parameter may be calculated using the weighting coefficients α and β in the manner of the second embodiment.

  Further, for example, the rotation angle θ may be once calculated using the outputs of the high-pass filters 50a and 50b, and the final rotation angle may be calculated by weighted average processing. Instead of this, it is also possible to select which one of the calculated values of the pair of rotation angles is used in the manner of the first embodiment.

"Vector phase spreading means"
In the third and fourth embodiments, instead of fixing which one of the current vector related to the current flowing in the U, V, W phase and the current vector related to the current flowing in the X, Y, Z phase is the advance side, You may switch by division.

  The combined current vector is not limited to the current for minimum current / maximum torque control. For example, it may be a current for performing maximum efficiency control.

"Stator phase separation means"
The direction of the voltage vector determined by the stator is not limited to the setting in which there is no polarity inversion of the current in the phase by switching the switching state of a specific phase, but is determined by the stator. While appropriately changing the direction of the voltage vector, the phases of the current vectors flowing through the respective sets of stators may be different from each other as in the third and fourth embodiments.

"Dead time compensation function"
The dead time compensation means for controlling the error of the command voltage due to the dead time voltage as a direct control amount to zero is not limited to the one that feed-forward corrects the command voltage (Duty signal) based on the polarity of the phase current. . For example, the detected value of the output voltage of each phase of the inverter may be feedback controlled to a command value. Even in this case, if correction is performed such that the start point and end point of the on-operation command period are shifted by the same time, it is possible to preferably avoid occurrence of an error in the high-frequency voltage signal outside the zero-cross period.

  Further, even if no dead time compensation means for controlling the error of the command voltage due to the dead time voltage as a direct control amount to zero is provided, for example, as in the current feedback control shown in FIG. You may employ | adopt what has a dead time compensation function in the function itself which feedback-controls the control amount of the generator 10. FIG. That is, according to the current feedback control, an error that occurs in the average line voltage within one cycle Tc of the PWM processing due to an error in the voltage of each phase due to the dead time becomes a control amount error. Will be corrected. Here, when the triangular wave PWM process is performed, the command voltage is corrected by shifting the start point and the end point of the ON operation command period of the operation signal by the same time. For this reason, when there is no zero crossing, the line voltage deviation is completely compensated. On the other hand, when there is a zero-crossing phase, no error occurs in the voltage of this phase, and the phase shift of this phase voltage does not cause an error in the average line voltage within one cycle Tc of the PWM processing. For this reason, the situation similar to the case of said each embodiment arises. However, there is a difference in that no error occurs in the high-frequency voltage signal due to the dead time in a period where there is no zero crossing, after feedback correction by feedback control is completed.

  Even if the dead time compensation function is not provided, if both of the outputs of the plurality of high-pass filters (high-pass filters 50a and 50b) are used simultaneously, as in the second embodiment (FIG. 11), for example. It is also possible to reduce the influence of errors caused by the dead time period compared to the case where only one is used. Even when the dead time compensation function is not provided, it is also effective to selectively use the one having less influence of errors due to the dead time period.

"About rotating machines"
The present invention is not limited to including two sets of three stators connected to each other at neutral points, and may include three or more sets. Further, the number of stators connected to each other at a neutral point is not limited to three, and may be two or four or more, for example.

“About Carrier CS”
The timing when the carrier CS becomes a peak may be the update timing of the command voltages vur, vvr, vwr.

  The carrier CS is not limited to a triangular wave, and may be any one as long as the gradual increase period and the gradual decrease period are symmetrical by setting the gradual increase speed and the gradual decrease speed to be equal to each other and the gradual increase period and the gradual decrease period being equal to each other. . In this case, the dead time compensation function makes it easy to set the correction to shift the start point and the end point of the ON operation command period of the operation signal by the same time.

"others"
The final control amount of the motor generator 10 is not limited to torque, and may be, for example, a rotational speed. Further, the present invention is not limited to performing current vector control, and for example, torque feedback control may be performed. At this time, if the command voltage is set as the operation amount for controlling the control amount, and the operation signal is set based on the magnitude comparison between the carrier having symmetry and the command voltage, the dead time is set by feedback control of the control amount. A time compensation function can be provided.

  The structural rotating machine having saliency is not limited to the motor generator 10 described above. For example, a synchronous reluctance motor (SynRM) may be used.

  -The rotating machine is not limited to the in-vehicle main machine. For example, an electric motor mounted on a vehicle-mounted power steering may be used.

  DESCRIPTION OF SYMBOLS 10 ... Motor generator, 14 ... Control apparatus, 50 ... High frequency voltage signal setting part, 54a, 54b ... Outer product calculating part.

Claims (7)

By operating a DC / AC converter circuit comprising a switching element that selectively connects a terminal of a rotating machine having saliency to each of a positive electrode and a negative electrode of a DC voltage source, and a diode connected in reverse parallel to the switching element, the rotating machine When controlling the control amount, superimposing means for superimposing a high-frequency voltage signal having a frequency higher than the electrical angular frequency of the rotating machine on the output voltage of the DC-AC converter circuit, and depending on the superimposed high-frequency voltage signal, In a control device for a rotating machine, comprising: estimation means for estimating a rotation angle of the rotating machine based on a detection value of a high-frequency current signal flowing through the rotating machine,
When switching from the state where either one of the switching element connected to the positive electrode and the switching element connected to the negative electrode and the other are turned on and off to the state where one and the other are turned off and on, respectively, There is a dead time period to turn off,
The rotating machine includes a plurality of sets of a plurality of stators connected to each other at neutral points,
The estimation means uses the detected value of the high-frequency current signal for each of at least two sets of the plurality of sets of the stators to estimate the rotation angle ,
A dead time compensation function that compensates for an error caused by the dead time by shifting the start point and the end point of the on operation command period of the operation signal of the DC / AC converter circuit by the same time, and
For at least a pair of the plurality of sets, a stator phase separation means for making the phase of the current flowing through the stator different, and
In the estimation of the rotation angle, the estimation means uses the detected value of the high-frequency current with respect to the plurality of sets of stators in which the absolute value of the current flowing through the stator as a constituent element is a predetermined value or more A control device for a rotating machine, comprising: a selection and use means for making the value larger than the degree of use of the detected value of the high-frequency current relating to a value equal to or less than the predetermined value . The selection available means as input parameters when estimating the rotational angle, according to claim 1, characterized in that it comprises inhibiting means for inhibiting the use of the detection value of the high-frequency current signal regarding those equal to or less than the predetermined value Rotating machine control device. Said stator phase separating means, the at least the pair of the set, the control device for a rotary machine according to claim 1 or 2 wherein, characterized in that it is constituted by causing differences the direction of the vector determined by the mutual stator. The plurality of sets include those in which the directions of vectors determined by the stator coincide with each other;
It said stator phase separating means is a control device for a rotary machine according to claim 1, wherein further comprising a vector phase diffusion means for mutually different phases of the current vector flowing through each set of said matched. 5. The rotating machine according to claim 1 , wherein the predetermined value is set to be equal to or greater than a ripple current value associated with switching of a switching state of a switching element constituting the DC / AC conversion circuit. Control device. Wherein the predetermined value, the control device for a rotary machine according to any one of claims 1 to 4, characterized in that it is set above the amplitude value of the high-frequency current signal. Further comprising a disconnection detecting means for detecting the presence or absence of disconnection of the electrical path between each of the plurality of sets of stators and the DC-AC conversion circuit;
The estimation means uses, when the disconnection detection means detects a disconnection, the detected value of the high-frequency current signal related to one of the plurality of sets in which the disconnection is not detected, for estimation of the rotation angle. The control device for a rotating machine according to any one of claims 1 to 6 .

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