JP2007151344A - Magnetic pole position estimating method, motor speed estimating method, and motor controller - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To reduce an error of an estimated magnetic pole position based on a harmonic component included in a motor current, while suppressing a computing load as compared with a conventional one in a synchronous motor. <P>SOLUTION: The harmonic component i<SB>γ_h</SB>of a γ-axis current value i<SB>γ</SB>is extracted as the harmonic component of the motor current, then a compensation amount i<SB>comp</SB>to the estimated magnetic pole position θ<SB>re</SB>is directly computed from the extracted harmonic component i<SB>γ_h</SB>. In other words, in view of the harmonic component of the motor current becoming an error factor when the estimated magnetic pole position θ<SB>re</SB>is obtained, the compensation amount i<SB>comp</SB>corresponding to the error of the estimated magnetic pole position caused by its harmonic component is directly obtained from the harmonic component i<SB>γ_h</SB>of the γ-axis current value i<SB>γ</SB>by which the computing load to obtain the compensation amount i<SB>comp</SB>is reduced as compared with the conventional one. <P>COPYRIGHT: (C)2007,JPO&INPIT

Description

  The present invention relates to a magnetic pole position estimation method for estimating the magnetic pole position of a synchronous motor, a motor speed estimation method for estimating the motor speed of the synchronous motor, and synchronous motor control for controlling the synchronous motor based on the estimated magnetic pole position and the estimated motor speed. Relates to the device.

  For example, when operating a synchronous motor (SPMSM, IPMSM) using a permanent magnet as a rotor, the energization phase to the stator winding is controlled in synchronization with the position of the rotor (magnetic pole position). A position detector for detecting the magnetic pole position of the child is required. As this position detector, a Hall element, an encoder, a resolver, or the like can be used. However, when such a position detector is used, it is difficult to reduce the size of the motor, and it is necessary to perform connection wiring with a control device that controls the motor.

  For this reason, in recent years, it has been studied to obtain the rotor magnetic pole position necessary for motor control by estimation without using a position detector for detecting the magnetic pole position of the rotor. For example, as described in Non-Patent Document 1, AC voltage and AC current on two-phase AC coordinates represented by αβ coordinates are configured based on an extended induced voltage model that is a motor model on the two-phase AC coordinates. It has been proposed to directly estimate the rotor magnetic pole position by inputting it into an extended induced voltage observer.

  However, the actual voltage and current include harmonic components generated due to various causes. For example, harmonic components are also generated by dead time of the inverter, unbalanced winding of the synchronous motor, and induced voltage caused by the structure of the synchronous motor. On the other hand, many motor models, including the extended induced voltage model, are configured on the assumption of an ideal sinusoidal magnetic flux distribution. Therefore, harmonic components included in voltage and current cause errors in the estimated magnetic pole position. It becomes a cause.

For this reason, in Non-Patent Document 2, current higher-order harmonics are removed using repetitive control and Fourier transform. As a result, the motor current can be brought close to an ideal sine wave, and errors included in the estimated magnetic pole position can be reduced.
“Extended Inductive Voltage Observer for Sensorless Control of Salient-Pole Brushless DC Motor” 1999 Annual Conference of the Institute of Electrical Engineers of Japan 1026 "Improvement of IPMSM sensorless control system by harmonic current suppression using Fourier transform and repetitive control" IEEJ Trans. IA, Vol.123, No.10, 2003

  Here, the technique described in Non-Patent Document 2 will be specifically described. First, the harmonic component signal extracted by the Fourier transform process is divided and averaged and stored in the repetitive compensator of the repetitive control unit. The repetitive compensator adds the input to the output delayed by the control period of the input repetitive control. The harmonic components stored in the repetitive compensator are phase-adjusted by the time advance compensator of the repetitive control unit in order to stabilize the control system.

  Thus, when the compensation signal for the harmonic component is generated and stored in the compensation signal generator, the harmonic component of the current control system is reduced by the compensation signal. Further, extraction by the Fourier transform process and learning by repetitive control are continuously performed on the reduced harmonic component. Such a series of processes is repeated until the harmonic component falls below the set value.

  As described above, in Non-Patent Document 2, in order to learn a compensation signal for harmonic components, it is necessary to repeatedly extract and store harmonic components by Fourier transform. For this reason, there exists a problem that the calculation load becomes large.

  The present invention has been made in view of the above points, and can reduce an estimated magnetic pole position error and a motor speed error based on a harmonic component while suppressing a calculation load as compared with the conventional one. It is an object of the present invention to provide a position estimation method, a motor speed estimation method, and a synchronous motor control device that controls a synchronous motor based on the estimated magnetic pole position and estimated motor speed.

In order to achieve the above object, a magnetic pole position estimation method according to claim 1 estimates a magnetic pole position of a synchronous motor,
An estimation step for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current, and a model formula of a synchronous motor;
An extraction step for extracting the harmonic component of the motor current;
A correction amount for the estimated magnetic pole position is calculated based on the harmonic component of the motor current, and a correction step for correcting the estimated magnetic pole position obtained in the estimation step by the calculated correction amount is provided.

  Thus, in the magnetic pole position estimation method according to the first aspect, the harmonic component of the motor current is extracted, and the correction amount for the estimated magnetic pole position is directly calculated from the extracted motor current harmonic component. In other words, in view of the fact that the harmonic component of the motor current becomes an error factor when obtaining the estimated magnetic pole position, the correction amount corresponding to the estimated magnetic pole position error due to the harmonic component is obtained directly from the harmonic component. is there. Therefore, it is possible to suppress the calculation load for obtaining the correction amount as compared with the conventional case.

  As described in claim 2, in the estimation step, first, when the induced voltage is calculated based on the model equation of the synchronous motor and then the estimated magnetic pole position is obtained based on the induced voltage, the correction step includes the estimation step. As a result, the estimated magnetic pole position may be corrected by correcting the induced voltage calculated in step.

  Here, when the estimated motor speed is obtained based on the change of the estimated magnetic pole position, if the error based on the harmonic component of the estimated magnetic pole position is reduced as described in the above-mentioned claim 1, the result is as follows. In addition, the estimated motor speed error is also reduced. However, particularly when it is necessary to remove an error with respect to the estimated motor speed, the method described in claim 3 can be employed.

That is, the motor speed estimation method for estimating the motor speed of the synchronous motor according to claim 3 is:
An estimation step for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current, and a model formula of a synchronous motor;
An estimated motor speed calculating step for obtaining an estimated motor speed based on a change in the estimated magnetic pole position;
An extraction step for extracting the harmonic component of the motor current;
A correction amount is calculated based on a harmonic component of the motor current, and a motor speed correction step for correcting the estimated motor speed based on the calculated correction amount is provided.

  Thereby, the estimated motor speed in which the error based on the harmonic component is reduced can be obtained.

  A control apparatus for a synchronous motor according to claims 4 and 5 controls the synchronous motor based on the configuration for obtaining the magnetic pole position by the magnetic pole position estimation method according to claims 1 and 2 and the obtained estimated magnetic pole position. The structure for this is provided. Accordingly, the function and effect are similar to those of the first aspect, and the description thereof is omitted.

In the above-described synchronous motor control apparatus, as described in claim 6,
The control means
Generating means for generating a motor applied voltage for each phase of the synchronous motor based on the estimated magnetic pole position;
Three-phase modulation means for generating a three-phase pulse width modulation signal by applying pulse width modulation to each phase applied voltage generated by the generation means in accordance with a three-phase modulation method;
Two-phase modulation means for generating a two-phase pulse width modulation signal by subjecting each phase applied voltage generated by the generation means to pulse width modulation according to a two-phase modulation method;
It is preferable to include switching means for switching a switching control signal supplied to the inverter to either a three-phase pulse width modulation signal or a two-phase pulse width modulation signal based on the motor speed of the synchronous motor.

  As is well known, in the three-phase modulation system, each phase applied voltage is compared with a carrier wave such as a triangular wave, and a pulse width modulation (PWM) signal of each phase is generated according to the comparison result. As a result, the line voltage of the synchronous motor approximates a sine wave. On the other hand, in the two-phase modulation method, for example, the voltage applied to each phase is set to the peak value of the carrier wave for a period of about 60 ° in electrical angle, thereby stopping switching of one of the three phases. The PWM signal is generated so that switching is performed only in the remaining two phases. In this two-phase modulation method, the line voltage of the synchronous motor can be increased as compared with the three-phase modulation method, and the voltage effective rate can be improved. Furthermore, since the number of times of switching can be reduced, switching loss can be reduced. The two-phase modulation method is disclosed in Japanese Patent Laid-Open Nos. 59-216476 and 7-46855.

  However, when a PWM signal for driving a synchronous motor is generated by the two-phase modulation method, the motor current includes more harmonic components than when a PWM signal is generated by the three-phase modulation method. Will be.

  When the motor speed is high, the induced voltage correlated with the magnetic pole position of the synchronous motor is larger than that when the motor speed is low, thereby increasing the motor current. It is possible to determine the correct estimated magnetic pole position. However, when the motor speed is low and the harmonic component of the motor current increases, the so-called S / N ratio is lowered, and the estimation accuracy of the magnetic pole position decreases.

  Therefore, in the synchronous motor control device according to claim 6, the switching control signal supplied to the inverter is switched between the three-phase pulse width modulation signal and the two-phase pulse width modulation signal according to the motor speed. It is.

  Specifically, as described in claim 7, when the motor speed is higher than the first predetermined speed, the switching unit switches the switching control signal supplied to the inverter to the two-phase pulse width modulation signal. When the speed is smaller than the second predetermined speed, the mode is switched to the three-phase pulse width modulation signal. In this way, when the motor speed decreases, the synchronous motor can be driven by the PWM signal based on the three-phase modulation method, which is a modulation method in which harmonic components are unlikely to occur in the motor current. On the other hand, when the motor speed increases and the influence of the harmonic component of the motor current decreases, the synchronous motor is driven by a PWM signal by a two-phase modulation method that can drive the synchronous motor with high efficiency. Can do.

  The second predetermined speed is set to be equal to or lower than the first predetermined speed, and is preferably set to a value smaller than the first predetermined speed. Thereby, it can prevent that a modulation system switches frequently.

Further, as described in claim 8,
The control means
A fundamental wave generating means for generating a fundamental wave of a motor applied voltage of each phase of the synchronous motor based on the estimated magnetic pole position;
Third harmonic generation means for generating the third harmonic of the fundamental wave of each phase applied voltage generated by the fundamental wave generation means;
According to the motor speed of the synchronous motor, there may be provided switching means for switching the motor applied voltage of each phase to one of a synthesized wave obtained by combining the fundamental wave and the third harmonic and a fundamental wave. good.

  In the above-described sixth aspect, the modulation method for generating the PWM signal is switched according to the motor speed. However, as described in the eighth aspect, the motor application for each phase is performed according to the motor speed. The same effect can be obtained by synthesizing the third harmonic with the fundamental wave of the voltage or canceling the synthesis. That is, the effective voltage ratio can be improved by synthesizing the third harmonic into the fundamental wave and distorting the applied voltage of each phase so as to stay longer in the vicinity of the peak value from the sine wave. However, in this case, the harmonic component in the motor current increases. Therefore, in the case of claim 8, the synthesis of the third harmonic with respect to the fundamental wave and the suspension of the synthesis are switched according to the motor speed.

  Specifically, as described in claim 9, when the motor speed is higher than the first predetermined speed, the switching unit synthesizes the motor applied voltage of each phase with the fundamental wave and the third harmonic. In addition to switching to the composite wave, it may be switched to the fundamental wave when the speed is lower than the second predetermined speed. The second predetermined speed is set to be equal to or lower than the first predetermined speed.

  Preferably, the switching unit calculates the motor speed based on a change in the estimated magnetic pole position. This eliminates the need to provide a sensor or the like in order to detect the motor speed. In addition to the motor speed calculated from the change in the estimated magnetic pole position, the switching means may use, for example, a motor speed command value as a motor speed for switching control, or a motor detected by a sensor or the like. Speed may be used.

  A control apparatus for a synchronous motor according to claim 11 is configured to obtain an estimated motor speed by the motor speed estimation method according to claim 3, and is configured to control the synchronous motor based on the obtained estimated motor speed. Is provided. Accordingly, the function and effect are similar to those of the third aspect, and the description thereof is omitted.

  The synchronous motor control device according to claim 12 supplies the inverter to the inverter according to the motor speed without assuming that the estimated magnetic pole position error is corrected by the correction amount based on the harmonic component of the motor current. The configuration for switching the switching control signal to either a three-phase pulse width modulation signal or a two-phase pulse width modulation signal is described independently. Also in this case, since the harmonic component of the motor current can be reduced when the motor speed is reduced, the error in the estimated magnetic pole position can be reduced.

  According to a fourteenth aspect of the present invention, there is provided a synchronous motor control device according to a motor speed, based on a motor speed without assuming that an estimated magnetic pole position error is corrected by a correction amount based on a harmonic component of a motor current. In this case, the third harmonic is synthesized with the fundamental wave of the motor applied voltage and the configuration for canceling the synthesis is independently described. In this case, the same effect as that of claim 12 can be obtained. it can.

  In addition, since the effect regarding the control apparatus of the synchronous motor of Claim 13, 15 and 16 is the same as that regarding the above-mentioned Claim 7, 9 and 10, description is abbreviate | omitted.

  Hereinafter, a control apparatus for a synchronous motor according to an embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a block diagram showing the overall configuration of the control device for the synchronous motor 10 according to the present embodiment. Note that the synchronous motor 10 whose driving state is controlled by the control device according to the present embodiment includes a three-phase (U-phase, V-phase, W-phase) stator winding and a rotor composed of permanent magnets.

In Figure 1, the speed control section 2 on the basis of the deviation between the first speed command value calculated by the deviation calculation unit 1 omega ^* and the estimated motor speed omega _re, gamma-axis current command value i ^* _gamma and δ The shaft current command value i ^* _δ is calculated and output. The second deviation calculation units 3a and 3b are configured to output the γ-axis current command value i ^* _γ and the δ-axis current command value i ^* _δ, and a γ-axis current value i _γ and δ-axis current from an αβ / dq conversion unit 13 described later. A value i _δ is inputted, and deviations (γ-axis current deviation Δi _γ , δ-axis current deviation Δi _δ ) of these input values are calculated and output. Then, the current control unit 4 performs the γ-axis voltage command value v ^* _γ and the δ-axis voltage command value v so that the respective deviations approach zero based on the γ-axis current deviation Δi _γ and the δ-axis current deviation Δi _δ. ^* Calculate and output _δ . The γ-axis voltage command value v ^* _γ and the δ-axis voltage command value v ^* _δ output from the current control unit 4 are output to the dq / αβ coordinate conversion unit 5.

  Here, as is well known, the dq coordinate is, for example, a rotational coordinate defined by a q axis perpendicular to the d axis, with the direction from the south pole to the north pole of the rotor being the d axis. The γδ coordinates are rotational coordinates defined by a δ axis perpendicular to the γ axis, where the estimated direction (estimated magnetic pole position) from the S pole to the N pole of the rotor is the γ axis. Therefore, when the estimated magnetic pole position does not include an error, the dq coordinate and the γδ coordinate coincide with each other. The αβ coordinates are stationary coordinates defined by an α axis and a β axis that are orthogonal to each other with the rotation axis of the rotor as the origin.

The coordinate conversion from the γδ rotation coordinate to the αβ static coordinate in the dq / αβ coordinate conversion unit 5 is performed based on the corrected estimated magnetic pole position θ _{re_comp} of the rotor output from the addition unit 16 described later. This is because if the magnetic pole position of the rotor can be estimated, the relative phase relationship between the γδ rotation coordinates and the αβ stationary coordinates can be specified, and coordinate conversion can be performed. By this coordinate conversion, the dq / αβ coordinate conversion unit 5 outputs the α-axis voltage command value v ^* _α and the β-axis voltage command value v ^* _β .

The α-axis voltage command value v ^* _α and the β-axis voltage command value v ^* _β from the dq / αβ coordinate conversion unit 5 are given to the αβ / uvw conversion unit 6 which is a two-phase / three-phase conversion unit. The αβ / uvw conversion unit 6 calculates each stator winding (U phase, V phase, W) of the three-phase synchronous motor 10 from the α axis voltage command value v ^* _α and the β axis voltage command value v ^* _β in αβ static coordinates. U-phase voltage command value to be output to phase) ^v _{* u,} ^V-phase voltage command value ^v _{* v,} to produce a W-phase voltage command value ^{v *} _w. The α-axis voltage command value v ^* _α and the β-axis voltage command value v ^* _β are also given to the magnetic pole position estimation unit 14 for calculating the estimated magnetic pole position.

The PWM signal generation unit 7 performs switching corresponding to each stator winding of the inverter 8 based on the U-phase voltage command value v ^* _u , the V-phase voltage command value v ^* _v , and the W-phase voltage command value v ^* _w. A PWM signal for driving the element is generated.

Here, in this embodiment, as shown in FIG. 2, the PWM signal generator 7 outputs a two-phase PWM signal according to a two-phase modulation method, and a three-phase PWM according to a three-phase modulation method. And a three-phase modulation unit 22 for outputting a signal. The PWM signal generator 7 further _supplies the U-phase, V-phase, and W-phase voltage command values v ^* _u , v ^* _v , and v ^* _w to the two-phase modulator 21 and the three-phase according to the motor speed ωre. It has a changeover switch unit 20 for switching to which of the modulation unit 22 an input. As the motor speed, the estimated motor speed omega _re outputted from the speed estimation unit 17 described later is used.

  Here, in the two-phase modulation method in the two-phase modulation unit 21, for example, one phase corresponding to the voltage command value for a period of about 60 ° at an electrical angle including a position where the voltage command value of each phase reaches a peak. And the PWM signal is generated so that the switching is performed only in the remaining two phases. Thereby, the line voltage of the synchronous motor 10 can be increased to improve the voltage effective rate, and the switching loss can be reduced. On the other hand, in the three-phase modulation method in the three-phase modulation unit 22, the voltage command value of each phase is formed, for example, in a sine wave shape, compared with a carrier wave such as a triangular wave, and each phase of the three phases according to the comparison result. The PWM signal is generated so that the switching is performed at. As a result, the line voltage of the synchronous motor 10 approximates a sine wave.

The inverter 8 has switching elements corresponding to the respective stator windings of the synchronous motor 10, and these switching elements are driven on / off according to the PWM signal output from the PWM signal generator 7. As a result, voltages corresponding to the U-phase voltage command value v ^* _u , the V-phase voltage command value v ^* _v, and the W-phase voltage command value v ^* _w can be applied to each stator winding. Reference numeral 9 denotes a DC power source. When each switching element of the inverter 8 is turned on / off, a current flows from the DC power source 9 to the synchronous motor 10 and the synchronous motor 10 is rotationally driven.

The current detection unit 11 includes a sensor that detects a motor current flowing through each stator winding, and detects current values i _u , i _v , and i _w that are passed through the stator windings using the sensor. . The current detection value output from the current detection unit 11 is input to the uvw / αβ conversion unit 12 and converted into an α-axis current value i _α and a β-axis current value i _β in αβ coordinates. The current detection unit 11 does not need to detect the current values of the stator windings of all three phases, but detects the current values of the two phases, and the remaining one-phase current value is the current of the two phases. It may be estimated from the value.

The α-axis current value i _α and β-axis current value i _β output from the uvw / αβ conversion unit 12 are input to the αβ / dq conversion unit 13 and the magnetic pole position estimation unit 14. The αβ / dq converter 13 converts the α-axis current value i _α and the β-axis current value i _β into the γ-axis current value i _γ based on the corrected estimated magnetic pole position θ _{re_comp} of the rotor output from the adder 16 described later. And the δ-axis current value i _δ and output to the second deviation calculation units 3a and 3b. The γ-axis current value i _γ is also input to a correction amount generation unit 15 that generates a correction amount for reducing an error that would be included in the estimated magnetic pole position, based on the harmonic component of the motor current. .

Here, the magnetic pole position estimation unit 14 and the correction amount generation unit 15 will be described in detail. First, the magnetic pole position estimation unit 14 receives the α-axis and β-axis voltage command values v ^* _α and v ^* _β input from the dq / αβ coordinate conversion unit 5 and the α-axis input from the uvw / αβ conversion unit 12 and Based on the β axis current values i _α and i _β , the magnetic pole position of the rotor is estimated using a predetermined model equation. As this magnetic pole position estimator 14, for example, “IPMSM position / velocity sensorless control using extended induced voltage disturbance observer” Ichikawa, Chen, Tomita, Michiki, Okuma, 2000 The described disturbance observer can be applied. Here, the disturbance observer will be briefly described.

  First, as the model of the synchronous motor, an extended induced voltage model shown in the following formula 1 can be used.

上記の数式１において、右辺第２項が拡張誘起電圧として定義される。この拡張誘起電圧モデルは、永久磁石による誘起電圧に加え、リラクタンストルクを生じるインダクタンスの差を新たに磁束成分として扱うものである。このような拡張誘起電圧モデルを用いると、磁極位置の推定演算を容易にするメリットがある。そして、上記数式１に示されるモデルに対して、拡張誘起電圧を外乱とみなした外乱オブザーバを構成することにより、拡張誘起電圧を求めることができる。この場合、拡張誘起電圧ｅ_α、ｅ_βは、以下の数式２によって表される。

In the above mathematical formula 1, the second term on the right side is defined as the extended induced voltage. In this extended induced voltage model, a difference in inductance that generates reluctance torque is newly treated as a magnetic flux component in addition to an induced voltage caused by a permanent magnet. Use of such an extended induced voltage model has an advantage of facilitating the calculation of the magnetic pole position. Then, an extended induced voltage can be obtained by configuring a disturbance observer in which the extended induced voltage is regarded as a disturbance with respect to the model expressed by Equation 1 above. In this case, the expansion induced voltages e _α and e _β are expressed by the following formula 2.

上記の数式２によって得られた拡張誘起電圧ｅ_α、ｅ_βに対して、以下の数式３に示すように逆正接演算を行うことにより、回転子の推定磁極位置θ_ｒｅを求めることができる。

The estimated magnetic pole position θ _re of the rotor can be obtained by performing an arctangent calculation on the extended induced voltages e _α and e _β obtained by the above mathematical formula 2 as shown in the following mathematical formula 3.

上述した例では、詳細なモータモデルを基に誘起電圧を演算するため、その誘起電圧から演算される推定磁極位置θ_ｒｅについて、速度応答性や負荷変動応答性を良好に保つことができる。ただし、この場合、モータモデルには微分項が含まれるので、演算量が増加する。この点を考慮し、微分項を省略した近似モデルにより誘起電圧を演算するようにしても良い。これにより、推定磁極位置θ_ｒｅを求めるための演算量を低減することができる。微分項を省略する場合、電流の波高値Ｉ_ａは一定であり、その時間微分はゼロであるとみなす。この仮定により、例えばα軸電流ｉ_αの微分項は以下の数式４のように近似できる。

In the above-described example, since the induced voltage is calculated based on the detailed motor model, the speed responsiveness and the load fluctuation responsiveness can be kept good with respect to the estimated magnetic pole position θ _re calculated from the induced voltage. In this case, however, the amount of calculation increases because the motor model includes a differential term. In consideration of this point, the induced voltage may be calculated by an approximate model in which the differential term is omitted. Thereby, the calculation amount for _obtaining the estimated magnetic pole position θ _re can be reduced. When omitting the differential term, the peak value I _a current is constant, the time derivative is regarded as being zero. Based on this assumption, for example, the differential term of the α-axis current i _α can be approximated as shown in Equation 4 below.

同様にして、β軸電流ｉ_βの微分項の近似も行うことにより、上記した数式１は、以下の数式５のように簡略化できる。

Similarly, the mathematical expression 1 described above can be simplified as the following mathematical expression 5 by approximating the differential term of the β-axis current i _β .

上述したようにして、磁極位置推定部１４において、拡張誘起電圧ｅ_α、ｅ_β及び回転子の推定磁極位置θ_ｒｅが算出される。この算出された推定磁極位置θ_ｒｅは、加算部１６に与えられる。

As described above, the magnetic pole position estimation unit 14 calculates the expansion induced voltages e _α and e _β and the estimated magnetic pole position θ _{re of the} rotor. The calculated estimated magnetic pole position θ _re is given to the adding unit 16.

On the other hand, the correction amount generation unit 15 generates a correction amount for reducing the estimated magnetic pole position error based on the harmonic component included in the motor current, based on the γ-axis current value i _γ . Specifically, gamma extracts a harmonic component i _{R_h} of the motor current from the axis current value i _gamma by the high-pass filter, as shown in Equation 6 below, a predetermined gain on the extracted harmonic component i _{R_h} The correction amount i _comp is calculated by multiplication.

なお、モータ電流の高調波成分は、δ軸電流値ｉ_δに対してハイパスフィルタ処理を施すことにより求めても良い。さらに、モータ電流の高調波成分は、高調波成分を含まないγ軸電流指令値ｉ^＊ _γと高調波成分を含むγ軸電流値ｉ_γとの差から求めることもできるし、δ軸電流指令値ｉ^＊ _δとδ軸電流値ｉ_δとの差から求めることもできる。

The harmonic component of the motor current may be obtained by performing a high-pass filter process on the δ-axis current value i _δ . Further, the harmonic component of the motor current can be obtained from the difference between the γ-axis current command value i ^* _γ not including the harmonic component and the γ-axis current value i _γ including the harmonic component, or the δ-axis current command. It can also be obtained from the difference between the value i ^* _δ and the δ-axis current value _iδ .

Here, an example of the relationship between the estimated magnetic pole position error and the harmonic component of the γ-axis current value i _γ obtained as the harmonic component of the motor current is shown in the graph of FIG. As shown in FIG. 3, the harmonic component of the γ-axis current value i _γ is synchronized with the estimated magnetic pole position error. This is because, for example, a harmonic component is generated by the dead time of the inverter 8, the unbalance of the winding of the synchronous motor 10, and the induced voltage caused by the structure of the synchronous motor 10. The estimated magnetic pole position θ _re is included in the current value i _γ , and is calculated using the motor current (α-axis current value i _α , β-axis current value i _β ) including the harmonic component. Because.

As shown in FIG. 3, since the estimated magnetic pole position error and the harmonic component _{ir_h} of the γ-axis current value i _γ are synchronized, the harmonic component _{ir_h} is multiplied by a predetermined gain as shown in Equation 6. Thus, the correction amount i _comp for reducing the estimated magnetic pole position error can be calculated. In this case, the gain K is given a code that can reduce the error included in the estimated magnetic pole position θ _re by adding the correction amount i _comp to the estimated magnetic pole position θ _re in the adding unit 16. . For this reason, the corrected estimated magnetic pole position θ _{re_comp} with reduced error is output from the adder 16. The corrected estimated magnetic pole position θ _{re_comp} is given to the dq / αβ coordinate conversion unit 5, the αβ / dq conversion unit 13, and the speed estimation unit 17.

The speed estimator 17 calculates a motor speed ω _re that is the rotational speed of the rotor of the synchronous motor 10 based on the change in the corrected estimated magnetic pole position θ _{re_comp} output from the adder 17, and a first deviation calculator Output to 1. For example, the speed estimation unit 17 can obtain the motor speed ω _re by _performing a differential operation on the corrected estimated magnetic pole position θ _{re_comp} as shown in Expression 7.

上述したように、本実施形態では、モータ電流の高調波成分ｉ_γ＿ｈを抽出し、抽出したモータ電流高調波成分ｉ_γ＿ｈから、直接的に、推定磁極位置θ_ｒｅに対する補正量ｉ_ｃｏｍｐを算出する。つまり、モータ電流の高調波成分ｉ_γ＿ｈが、推定磁極位置θ_ｒｅを求める際の誤差要因となることに鑑み、その高調波成分ｉ_γ＿ｈによる推定磁極位置誤差に対応する補正量ｉ_ｃｏｍｐを、高調波成分ｉ_γ＿ｈから直接的に求めるのである。従って、補正量ｉ_ｃｏｍｐを求めるための演算負荷を従来に比較して減少させることができる。

As described above, in the present embodiment, the harmonic component i _{γ_h} of the motor current is extracted, and the correction amount i _comp for the estimated magnetic pole position θ _re is directly _calculated from the extracted motor current harmonic component i _{γ_h.} . That is, considering that the harmonic component i _{γ_h} of the motor current becomes an error factor when _obtaining the estimated magnetic pole position θ _re , the correction amount i _comp corresponding to the estimated magnetic pole position error due to the harmonic component i _{γ_h is calculated} by It is obtained directly from the wave component i _{γ_h} . Therefore, the calculation load for _obtaining the correction amount i _comp can be reduced as compared with the conventional case.

Furthermore, in the present embodiment, as described above, PWM signal generating unit 7 has a two-phase modulation unit 21 and the three-phase modulation unit 22, according to the motor speed omega _re, in order to generate a PWM signal , Which modulation method is used is switched. Specifically, as shown in FIG. 4, when the motor speed omega _re is greater than a first predetermined speed N1 (e.g., 800 rpm) generates a PWM signal according to the two-phase modulation scheme, the second predetermined speed When it is smaller than N2 (for example, 500 rpm), the modulation method to be adopted is switched so as to generate the PWM signal according to the three-phase modulation method. The reason for switching the modulation scheme in accordance with the motor speed omega _re is as follows.

  As described above, the two-phase modulation method has advantages over the three-phase modulation method, such that the voltage effective rate can be improved and the switching loss can be reduced. However, when the PWM signal is generated by the two-phase modulation method, the motor current includes more harmonic components than when the PWM signal is generated by the three-phase modulation method.

When the motor speed _ωre is high, an induced voltage correlated with the magnetic pole position of the synchronous motor 10 becomes larger than that when the motor speed _ωre is low, thereby increasing the motor current, so that the harmonic component slightly increases. However, it is possible to obtain a correct estimated magnetic pole position. However, when the motor speed is low and the harmonic component of the motor current increases, the so-called S / N ratio is lowered, and the estimation accuracy of the magnetic pole position decreases.

Therefore, in the present embodiment, as described above, according to the motor speed omega _re, a switching control signal supplied to the inverter 8, so that to switch to one of the 3-phase PWM signal and the 2-phase PWM signal is there. An example of the effect of switching the modulation method will be described with reference to the graph of FIG.

  FIG. 5 shows changes in the estimated magnetic pole position and modulation rate when the modulation method is switched from the two-phase modulation method to the three-phase modulation method with the synchronous motor 10 rotating at a motor speed of 500 rpm. . As shown in FIG. 5, by switching from the two-phase modulation method to the three-phase modulation method, harmonic noise superimposed on the waveform indicating the estimated magnetic pole position is reduced. In addition, the modulation rate has changed to an appropriate level.

Furthermore, the effect by this embodiment is demonstrated based on the graph of FIG. 6 (a), (b) and FIG. 7 (a), (b). FIGS. 6A and 7A show the case where the error of the estimated magnetic pole position is not reduced by the correction amount based on the harmonic component of the motor current, and the above-described modulation method is not switched. This shows the FFT analysis result of the harmonic component included in the estimated magnetic pole position θ _re and the region where the synchronous motor 10 can be driven stably. FIG. 6 (b) and 7 (b), in the case of controlling the driving of the synchronous motor 10 by the control apparatus according to the present embodiment, FFT analysis result of the harmonic components included in the estimated magnetic pole position theta _re and synchronization An area in which the motor 10 can be stably driven is shown.

FIG. 6 (a), the As is apparent from comparison (b), when using a control apparatus according to the present embodiment, harmonic components (3f contained in the estimated magnetic pole position _{θ re, 6f, 9f, 12f} , ... ) Has greatly decreased. Further, from FIGS. 7A and 7B, it can be seen that the stable drive region of the synchronous motor 10 can be expanded to the low speed and high load regions by the control device according to the present embodiment.

In addition, when the synchronous motor 10 is driven in a high load state, the motor applied voltage increases and the spatial inductance increases, so that the harmonic component increases. For this reason, errors such as the estimated magnetic pole position _θre also increase, and stable driving becomes difficult as compared with a low load state.

As described above, in this embodiment, depending on the motor speed omega _re, a switching control signal supplied to the inverter 8 is switched to either the 3-phase PWM signal and the 2-phase PWM signal. For this reason, when the motor speed _ωre decreases, the synchronous motor 10 can be stably driven by the three-phase PWM signal based on the three-phase modulation method, which is a modulation method in which a harmonic component hardly occurs in the motor current. On the other hand, when the motor speed _ωre increases and the influence of the harmonic component of the motor current decreases, the synchronous motor is driven by the two-phase PWM signal based on the two-phase modulation method that can drive the synchronous motor 10 with high efficiency. 10 can be driven.

  In the present embodiment, as shown in FIG. 4, between the first predetermined speed N1 that is the switching speed to the two-phase modulation system and the second predetermined speed N2 that is the switching speed to the three-phase modulation system. Is provided with hysteresis. For this reason, it can prevent that a modulation system switches frequently.

  The synchronous motor control device according to the preferred embodiment of the present invention has been described above. However, the present invention is not limited to the above-described embodiment, and various modifications may be made without departing from the spirit of the present invention. Is possible.

(Modification 1)
For example, in the above-described embodiment, and that the harmonic component i _{Ganma_h} of the motor current, and calculates a correction amount _icomp for reducing the error of the estimated magnetic pole position theta _re, obtaining a correction estimated magnetic pole position theta _{re_comp,} motor the switching of the modulation scheme according to the speed omega _re were those carried out simultaneously. As described above, by simultaneously implementing the two countermeasures against the generation of the harmonic component, it is possible to effectively reduce the error of the estimated magnetic pole position θ _re based on the harmonic component. However, the above-mentioned two measures do not necessarily have to be performed at the same time, and even if each of them is used alone, it can contribute to a reduction in the error of the estimated magnetic pole position θ _re based on the harmonic component.

(Modification 2)
In the embodiment described above, the estimated magnetic pole position θ _re calculated by the magnetic pole position estimating unit 14 is added to the estimated magnetic pole position θ _re by adding the correction amount i _comp generated by the correction amount generating unit 15 in the adding unit 16. A correction estimated magnetic pole position θ _{re_comp} that is a correction value of the position θ _re was calculated. However, the method of correcting the estimated magnetic pole position _θre is not limited to such an example.

For example, as shown in FIG. 8, the correction amount generation unit 15a calculates the correction amounts i _{comp_α} and i _{comp_β} in the α axis and the β axis based on the high frequency components such as the γ axis current value i _γ and estimates the magnetic pole position. To part 14a. As shown in Formula 8, the magnetic pole position estimation unit 14a calculates corrected extended induced voltages e _{α_comp} and e _{β_comp obtained} by correcting the extended induced voltages e _α and e _β with the input correction amounts i _{comp_α} and i _{comp_β} .

そして、この補正拡張誘起電圧ｅ_{α＿ｃｏｍｐ}、ｅ_{β＿ｃｏｍｐ}を用いて、逆正接演算を行うことによって補正推定磁極位置θ_{ｅｓ＿ｃｏｍｐ}を求める。このようにしても、上述した実施形態と実質的に同じ補正推定磁極位置θ_{ｅｓ＿ｃｏｍｐ}を求めることができる。

Then, the corrected estimated magnetic pole position θ _{es_comp} is obtained by performing an arctangent calculation using the corrected extended induced voltages e _{α_comp} and e _{β_comp} . Even in this case, it is possible to obtain the corrected estimated magnetic pole position θ _{es_comp} that is substantially the same as in the above-described embodiment.

(Modification 3)
In the embodiment and the modification 2 described above, the corrected estimated magnetic pole position θ _{es_comp} is obtained, and the estimated motor speed ω _re is obtained based on the change in the corrected estimated magnetic pole position θ _{es_comp} . In this case, since the estimated motor speed ω _re is obtained from the corrected estimated magnetic pole position θ _{es_comp} in which the error based on the harmonic component of the estimated magnetic pole position θ _es is reduced, the error of the estimated motor speed ω _re is also reduced as a result. Is done. However, particularly when it is necessary to remove an error regarding the estimated motor speed ω _re , the estimated motor speed ω _re may be directly corrected.

Specifically, as shown in FIG. 9, the correction amount calculation unit 15b calculates a correction amount i _comp for reducing an error based on the harmonic component of the estimated motor speed ω _re , and the correction amount i _comp The estimated motor speed ω _re may be added by the adding unit 18. As a result, it is possible to obtain the corrected estimated motor speed _{ωre_comp} in which the error based on the harmonic component is reduced.

(Modification 4)
In the embodiment described above, the PWM signal generator 7 was achieved, switching the modulation scheme in accordance with the motor speed omega _re. As a result, the driving stability of the synchronous motor 10 can be ensured particularly in the low speed region and the high load region where the driving stability of the synchronous motor 10 is a problem.

However, the same effect can be obtained by a method other than switching the modulation method. That is, in the PWM signal generation unit 7, as shown in FIG. 10, the U-phase, V-phase, and W-phase voltage command values v ^* _u , v ^* _v , and v ^* _w input from the αβ / UVW conversion unit 6 are changed. Based on this, a sine wave-like fundamental wave and a third harmonic whose frequency is increased to three times the fundamental wave are generated. Then, in order to generate a PWM signal to be output to the inverter 8 according to the motor speed _ωre , a voltage command value to be compared with the carrier wave (triangular wave) is combined with a combined wave obtained by combining the fundamental wave and the third harmonic. Switch to one of the fundamental waves.

Here, referring to the waveform diagram of FIG. 10, the synthesized wave obtained by synthesizing the fundamental wave and the third harmonic wave has a distorted waveform shape so as to stay long in the vicinity of the peak value. Therefore, a synthesized wave obtained by synthesizing the fundamental wave and the third harmonic is adopted as the voltage command values v ^* _u , v ^* _v , and v ^* _w for each phase, and a PWM signal is generated under comparison with the carrier wave. Thus, as in the case of adopting the two-phase modulation method, it is possible to improve the voltage effective rate and reduce the switching loss.

However, when a composite wave is adopted as the voltage command values v ^* _u , v ^* _v , and v ^* _w for each phase, the harmonic component in the motor current increases as in the case of the two-phase modulation method. Therefore, to employ the composite wave is the time the motor speed omega _re is greater than the first predetermined speed, when the motor speed omega _re falls below the second predetermined speed, the voltage command fundamental wave of each phase The values v ^* _u , v ^* _v , and v ^* _w are adopted.

(Modification 5)
In the embodiment described above, the magnetic pole position estimation unit 14 calculates the estimated magnetic pole position θ _es in the αβ static coordinate system and corrects it by the correction amount generation unit 15 and the addition unit 16.

However, the calculation and correction of the estimated magnetic pole position θ _es may be performed in the dq (γδ) rotating coordinate system. In this case, however, the calculated error is not the actual magnetic pole position but the estimated magnetic pole position. The estimated error is corrected using the correction amount calculated from the harmonic component of the motor current. Will do. The same applies to the case where correction is made to the expansion induced voltage and the motor voltage.
(Modification 6)
In the above-described embodiment, switching between the two-phase PWM signal and the three-phase PWM signal is performed based on the motor speed ω _re calculated from the change in the estimated magnetic pole position θ _es . However, for example, the motor speed command value ω ^* may be used as a motor speed for performing switching control, or a motor speed detected by a sensor or the like may be used.

It is a block diagram which shows the whole structure of the control apparatus of the synchronous motor by embodiment. It is a block diagram which shows the structure of the PWM signal generation part in FIG. 6 is a graph showing an example of a relationship between an estimated magnetic pole position error and a harmonic component of a γ-axis current value i _γ obtained as a harmonic component of a motor current. It is explanatory drawing for demonstrating the switching logic of the modulation system in a PWM signal generation part. It is a graph for demonstrating an example of the effect by switching of a modulation system. (A) shows a harmonic included in the estimated magnetic pole position θ _re when the error of the estimated magnetic pole position is not reduced by the correction amount based on the harmonic component of the motor current and the above-described modulation method is not switched. It is a graph which shows the FFT analysis result of a component, (b) shows the FFT analysis result of the harmonic component contained in the estimation magnetic pole position (theta) _re when the drive of the synchronous motor 10 is controlled by the control apparatus by this embodiment. It is a graph to show. (A) is a region in which the synchronous motor 10 can be driven stably without reducing the error in the estimated magnetic pole position by the correction amount based on the harmonic component of the motor current and without switching the modulation method described above. (B) is a graph which shows the area | region which can drive the synchronous motor 10 stably when the drive of the synchronous motor 10 is controlled by the control apparatus by this embodiment. It is a block diagram which shows the structure of the synchronous motor control apparatus by the modification 2 of embodiment. It is a block diagram which shows the structure of the synchronous motor control apparatus by the modification 3 of embodiment. It is a wave form diagram for demonstrating the synchronous motor control apparatus by the modification 4 of embodiment.

Explanation of symbols

DESCRIPTION OF SYMBOLS 2 ... Speed controller 4 ... Current controller 5 ... dq / αβ coordinate converter 6 ... αβ / uvw converter 7 ... PWM signal generator 8 ... Inverter 10 ... Synchronous motor 11 ... Current detector 12 ... uvw / αβ converter 13 ... αβ / dq conversion unit 14 ... magnetic pole position estimation unit 15 ... correction amount generation unit 16 ... addition unit 17 ... speed estimation unit

Claims (16)

A magnetic pole position estimation method for estimating a magnetic pole position of a synchronous motor,
An estimation step for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current, and a model formula of the synchronous motor;
An extraction step of extracting harmonic components of the motor current;
A correction amount for the estimated magnetic pole position is calculated based on a harmonic component of the motor current, and a correction step for correcting the estimated magnetic pole position obtained in the estimation step with the calculated correction amount is provided. Magnetic pole position estimation method. In the estimation step, first, an induced voltage is calculated based on a model formula of the synchronous motor, and then an estimated magnetic pole position is obtained based on the induced voltage.
2. The magnetic pole position estimation method according to claim 1, wherein in the correction step, the estimated magnetic pole position is corrected as a result by correcting the induced voltage calculated in the estimation step. A motor speed estimation method for estimating a motor speed of a synchronous motor,
An estimation step for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current, and a model formula of the synchronous motor;
An estimated motor speed calculating step for obtaining an estimated motor speed based on a change in the estimated magnetic pole position;
An extraction step of extracting harmonic components of the motor current;
A motor speed estimation step comprising: calculating a correction amount for the motor speed based on the harmonic component of the motor current, and correcting the estimated motor speed based on the calculated correction amount. Method. A control device for a synchronous motor,
Magnetic pole position estimating means for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current and a model formula of the synchronous motor;
Extraction means for extracting harmonic components of the motor current;
A correction unit that calculates a correction amount based on the harmonic component of the motor current and corrects the estimated magnetic pole position obtained by the magnetic pole position estimation unit based on the calculated correction amount;
A synchronous motor control device comprising: control means for generating the motor applied voltage and applying it to the synchronous motor based on the estimated magnetic pole position corrected by the correcting means. The magnetic pole position estimating means calculates an induced voltage based on a model formula of the synchronous motor, and obtains an estimated magnetic pole position based on the induced voltage,
5. The synchronous motor control device according to claim 4, wherein the correction means corrects the induced voltage calculated in the estimating step, and as a result, corrects the estimated magnetic pole position. The control means includes
Generating means for generating a motor applied voltage for each phase of the synchronous motor based on the estimated magnetic pole position;
Three-phase modulation means for generating a three-phase pulse width modulation signal by applying pulse width modulation to each phase applied voltage generated by the generation means according to a three-phase modulation method;
Two-phase modulation means for generating a two-phase pulse width modulation signal by applying pulse width modulation to each phase applied voltage generated by the generation means in accordance with a two-phase modulation method;
Switching means for switching a switching control signal supplied to the inverter to either the three-phase pulse width modulation signal or the two-phase pulse width modulation signal based on the motor speed of the synchronous motor. The control device for a synchronous motor according to claim 4 or 5.   The switching means switches the switching control signal supplied to the inverter to the two-phase pulse width modulation signal when the motor speed is higher than the first predetermined speed, and when the motor speed is lower than the second predetermined speed. 7. The synchronous motor control device according to claim 6, wherein the control is switched to the three-phase pulse width modulation signal. The control means includes
A fundamental wave generating means for generating a fundamental wave of a motor applied voltage of each phase of the synchronous motor based on the estimated magnetic pole position;
Third harmonic generation means for generating a third harmonic of the fundamental wave of each phase applied voltage generated by the fundamental wave generation means;
Based on the motor speed of the synchronous motor, switching means is provided for switching the motor applied voltage of each phase to one of a synthesized wave obtained by synthesizing the fundamental wave and the third harmonic and the fundamental wave. 6. The control device for a synchronous motor according to claim 4 or 5, wherein:   The switching means switches the motor applied voltage of each phase to a combined wave obtained by combining the fundamental wave and the third harmonic when the motor speed is higher than a first predetermined speed, 9. The synchronous motor control device according to claim 8, wherein the control is switched to the fundamental wave when the speed is lower than a predetermined speed.   10. The synchronous motor control device according to claim 6, wherein the switching unit calculates a motor speed based on a change in the estimated magnetic pole position. A control device for a synchronous motor,
Magnetic pole position estimating means for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current and a model formula of the synchronous motor;
Estimated motor speed calculation means for obtaining an estimated motor speed based on a change in the estimated magnetic pole position;
Extraction means for extracting harmonic components of the motor current;
While calculating a correction amount based on the harmonic component of the motor current, motor speed correction means for correcting the estimated motor speed by the calculated correction amount;
A synchronous motor control device comprising: a control unit that generates the motor applied voltage based on the estimated motor speed corrected by the correction unit and applies the generated voltage to the synchronous motor. A control device for a synchronous motor,
Magnetic pole position estimating means for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current and a model formula of the synchronous motor;
Generating means for generating a motor applied voltage for each phase of the synchronous motor based on the estimated magnetic pole position;
Three-phase modulation means for generating a three-phase pulse width modulation signal by applying pulse width modulation to each phase applied voltage generated by the generation means according to a three-phase modulation method;
Two-phase modulation means for generating a two-phase pulse width modulation signal by applying pulse width modulation to each phase applied voltage generated by the generation means in accordance with a two-phase modulation method;
And switching means for switching a switching control signal supplied to the inverter to either the three-phase pulse width modulation signal or the two-phase pulse width modulation signal based on the motor speed of the synchronous motor. Motor control device.   The switching means switches the switching control signal supplied to the inverter to the two-phase pulse width modulation signal when the motor speed is higher than the first predetermined speed, and when the motor speed is lower than the second predetermined speed. The synchronous motor control device according to claim 12, wherein the control is switched to the three-phase pulse width modulation signal. A control device for a synchronous motor,
Magnetic pole position estimating means for obtaining an estimated magnetic pole position based on a motor applied voltage and a motor current and a model formula of the synchronous motor;
A fundamental wave generating means for generating a fundamental wave of a motor applied voltage of each phase of the synchronous motor based on the estimated magnetic pole position;
Harmonic generation means for generating a third harmonic of the fundamental wave of each phase applied voltage generated by the fundamental wave generation means;
Based on the motor speed of the synchronous motor, switching means is provided for switching the motor applied voltage of each phase to one of a synthesized wave obtained by synthesizing the fundamental wave and the third harmonic and the fundamental wave. The control apparatus of the synchronous motor characterized by this.   The switching means switches the motor applied voltage of each phase to a combined wave obtained by combining the fundamental wave and the third harmonic when the motor speed is higher than a first predetermined speed, 15. The synchronous motor control device according to claim 14, wherein the control is switched to the fundamental wave when the speed is lower than a predetermined speed.   16. The synchronous motor control apparatus according to claim 12, wherein the switching unit calculates a motor speed based on a change in the estimated magnetic pole position.

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