JP5543388B2 - Control device for permanent magnet synchronous motor - Google Patents

Control device for permanent magnet synchronous motor Download PDF

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JP5543388B2
JP5543388B2 JP2011018139A JP2011018139A JP5543388B2 JP 5543388 B2 JP5543388 B2 JP 5543388B2 JP 2011018139 A JP2011018139 A JP 2011018139A JP 2011018139 A JP2011018139 A JP 2011018139A JP 5543388 B2 JP5543388 B2 JP 5543388B2
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裕樹 有馬
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Description

本発明の実施形態は、回転子位置の推定機能を有する永久磁石同期電動機の制御装置に関する。   Embodiments described herein relate generally to a permanent magnet synchronous motor control apparatus having a rotor position estimation function.

インバータによる可変速電動機制御装置は各分野に適用されており、今後はトルクや速度制御精度の向上、高効率、低騒音など更なる高性能化と信頼性の向上が期待されている。特に、速度センサや位置センサを用いない位置速度センサレス制御は、信頼性向上や設置環境の制約の改善などの点で有用である。   Variable speed motor control devices using inverters are applied in various fields, and further improvements in performance and reliability such as improved torque and speed control accuracy, high efficiency and low noise are expected in the future. In particular, position / speed sensorless control without using a speed sensor or position sensor is useful in terms of improving reliability and improving constraints on the installation environment.

1980年頃から、位置センサを用いることなく、永久磁石などにより回転子に磁束を有する電動機の回転子磁束方向を検出しようとする技術が開発されている。しかし、電動機の停止時または低速回転時には誘起電圧が0または非常に小さいため、デッドタイム補償誤差による出力電圧誤差や推定式に用いる一次抵抗の設定誤差などの影響を受け易く、誘起電圧に依存した回転子磁束方向の推定手段だけでは十分な制御特性が得られない。   Since about 1980, a technique for detecting the rotor magnetic flux direction of an electric motor having a magnetic flux in the rotor by a permanent magnet or the like without using a position sensor has been developed. However, since the induced voltage is 0 or very small when the motor is stopped or rotated at a low speed, it is easily affected by the output voltage error due to the dead time compensation error and the setting error of the primary resistance used in the estimation formula, and depends on the induced voltage. Sufficient control characteristics cannot be obtained only by means for estimating the rotor magnetic flux direction.

そこで、回転子磁束位置推定技術の1つとして、制御装置内で有する推定された回転子磁束方向に、高周波の交番電流あるいは交番電圧を重畳する手段がある。推定された回転子磁束の方向をD軸とし、トルクに寄与する電流ベクトルの方向をQ軸と定め、それぞれの電流成分を各々制御する電流制御装置を備える。   Therefore, as one of the rotor magnetic flux position estimation techniques, there is means for superimposing a high-frequency alternating current or alternating voltage on the estimated rotor magnetic flux direction in the control device. The estimated direction of the rotor magnetic flux is defined as the D axis, the direction of the current vector contributing to the torque is defined as the Q axis, and a current control device that controls each current component is provided.

回転子磁束方向を推定するには、高周波の交番電流を重畳する場合には、高周波電流が流されない軸方向の電圧値を用いればよい。つまり、D軸に高周波電流を重畳し、高周波電流を重畳しないQ軸の電圧値を用いて回転子磁束方向を推定することができる。またこれと等価な手段として、高周波の交番電圧を重畳する場合には、高周波電圧が重畳されない軸方向の電流値を用いればよい。すなわち、D軸に高周波電圧を重畳し、高周波電圧を重畳しないQ軸の電流値を用いて回転子磁束位置を推定することもできる。   In order to estimate the rotor magnetic flux direction, when a high-frequency alternating current is superimposed, a voltage value in the axial direction in which no high-frequency current is passed may be used. That is, the rotor magnetic flux direction can be estimated using the voltage value of the Q axis that superimposes the high frequency current on the D axis and does not superimpose the high frequency current. Further, as an equivalent means, when a high-frequency alternating voltage is superimposed, an axial current value where the high-frequency voltage is not superimposed may be used. That is, it is possible to estimate the rotor magnetic flux position by superimposing a high frequency voltage on the D axis and using a Q axis current value without superimposing the high frequency voltage.

米国特許第4763058号明細書US Patent No. 4763058 特開昭62−138074号公報JP-A-62-138074 特許第3454212号公報Japanese Patent No. 3454212

Schroedl,M.、「DETECTION OF THE ROTOR POSITION OF A PERMANENT MAGNET MACHINE AT STANDSTILL」、ICEM(International Conference on Electrical Machine) 1988 in Pisa(Italy)、p.195-197Schroedl, M., `` DETECTION OF THE ROTOR POSITION OF A PERMANENT MAGNET MACHINE AT STANDSTILL '', ICEM (International Conference on Electrical Machine) 1988 in Pisa (Italy), p.195-197

高周波電圧を重畳し、電機子電流に流れる高周波電流を用いて回転子磁束方向を推定するためには、D軸電流またはQ軸電流に現れる高周波電流成分を抽出し、それに基づいて実際の回転子磁束方向と推定した回転子磁束方向との誤差Δθの情報を得る必要がある。高周波電流成分を抽出する方法として最も簡単な方法の1つは、重畳している高周波電圧に基づいて、高周波電流値が最大となる時と最小となる時の差分を抽出する方法である。   In order to estimate the rotor magnetic flux direction using the high-frequency current that is superimposed on the high-frequency voltage and that flows through the armature current, the high-frequency current component that appears in the D-axis current or the Q-axis current is extracted, and the actual rotor is based on that. It is necessary to obtain information on the error Δθ between the magnetic flux direction and the estimated rotor magnetic flux direction. One of the simplest methods for extracting a high-frequency current component is a method of extracting a difference between when the high-frequency current value is maximum and when it is minimum based on the superimposed high-frequency voltage.

しかし、この方法によれば高周波電圧周期に2回のみの電流検出値によって高周波成分を抽出することになるため、インバータ回路のスイッチングノイズ等の影響を受け易い。上記2回の何れかの電流検出値に誤差が生じた場合、高周波電流成分から算出される位置誤差情報は、当該高周波電圧周期において検出誤差を含んでいるため、回転子磁束方向の推定結果に影響を及ぼす虞がある。   However, according to this method, the high frequency component is extracted by the current detection value only twice in the high frequency voltage period, and therefore, it is easily affected by the switching noise of the inverter circuit. When an error occurs in one of the two current detection values, the position error information calculated from the high-frequency current component includes a detection error in the high-frequency voltage period, and therefore the estimation result of the rotor magnetic flux direction May have an effect.

特にインダクタンスが大きい場合には、重畳する高周波電圧の周波数を高くすると高周波電流成分が小さくなるので、重畳する高周波電圧の周波数を高く取れない。この場合には、次の高周波電流の最大値または最小値の検出値が得られるまでの時間が長くなる。その結果、回転子位置の推定誤差が大きくなった場合には、トルクを維持できず脱調に至る可能性もある。   In particular, when the inductance is large, if the frequency of the superimposed high-frequency voltage is increased, the high-frequency current component is reduced, so that the frequency of the superimposed high-frequency voltage cannot be increased. In this case, the time until the next maximum or minimum detection value of the high-frequency current is obtained is increased. As a result, when the estimation error of the rotor position becomes large, there is a possibility that the torque cannot be maintained and a step-out occurs.

本発明が解決しようとする課題は、高周波電圧を印加することにより高周波電流を流して磁極位置を推定する構成において、高周波電流の検出誤差の影響を受けにくい永久磁石同期電動機の制御装置を提供することである。   A problem to be solved by the present invention is to provide a controller for a permanent magnet synchronous motor that is less susceptible to the detection error of a high frequency current in a configuration in which a high frequency current is applied by applying a high frequency voltage to estimate a magnetic pole position. That is.

実施形態の永久磁石同期電動機の制御装置は、永久磁石同期電動機の回転子磁束の推定軸をD軸とし、D軸に対しπ/2進んだ位置にある軸をQ軸とし、永久磁石同期電動機に高周波電圧を印加することにより高周波電流を流して回転子の磁極位置を推定し、この推定磁極位置に基づいて永久磁石同期電動機を制御する。電流検出手段は、永久磁石同期電動機に流れる電流を所定の電流検出周期で検出する。高周波電圧印加手段は、印加する高周波電圧の半周期を電流検出周期のN倍(N=1、2、…)となるように選択し、その高周波電圧の位相を(2π)÷(2N)を単位とする離散的な位相として取り扱い、この離散的な位相の正弦波状関数の電圧成分をD軸電圧に重畳する。第1の位置誤差情報抽出手段は、離散的な位相の余弦波状関数の極性反転信号を検出したD軸電流およびQ軸電流のうち少なくともQ軸電流に乗算し、その乗算値を高周波電圧の0から2Mπ(M=1、2、…)までの位相区間にわたり加算して、推定磁極位置の誤差情報を含む位置誤差抽出量を演算する。第2の位置誤差情報抽出手段は、前記乗算値を、前記位相区間に対しMπだけ位相が異なる位相区間にわたり加算して、推定磁極位置の誤差情報を含む位置誤差抽出量を演算する。収束演算手段は、第1の位置誤差情報抽出手段から得られた位置誤差抽出量と、第2の位置誤差情報抽出手段から得られた位置誤差抽出量に基づいて、推定磁極位置の誤差を打ち消すように出力周波数を収束演算する。第1の位置誤差情報抽出手段および第2の位置誤差情報抽出手段は、乗算値を高周波電圧の各位相区間にわたり加算する場合、乗算値を同一のメモリ領域に順次加算する。
In the control device for a permanent magnet synchronous motor of the embodiment, the estimated axis of the rotor magnetic flux of the permanent magnet synchronous motor is the D axis, the axis at a position advanced by π / 2 with respect to the D axis is the Q axis, and the permanent magnet synchronous motor A high frequency voltage is applied to the rotor to cause a high frequency current to flow to estimate the magnetic pole position of the rotor, and the permanent magnet synchronous motor is controlled based on the estimated magnetic pole position. The current detection means detects the current flowing through the permanent magnet synchronous motor at a predetermined current detection cycle. The high frequency voltage application means selects a half cycle of the applied high frequency voltage to be N times the current detection cycle (N = 1, 2,...), And sets the phase of the high frequency voltage to (2π) ÷ (2N). Treated as a discrete phase as a unit, the voltage component of the sinusoidal function of this discrete phase is superimposed on the D-axis voltage. The first position error information extraction means multiplies at least the Q-axis current of the detected D-axis current and Q-axis current of the polarity inversion signal of the cosine wave function having a discrete phase, and multiplies the multiplied value by 0 of the high-frequency voltage. To 2Mπ (M = 1, 2,...) To calculate a position error extraction amount including error information of the estimated magnetic pole position. The second position error information extraction unit adds the multiplication value over a phase interval whose phase is different by Mπ with respect to the phase interval, and calculates a position error extraction amount including error information of the estimated magnetic pole position. The convergence calculation means cancels the error of the estimated magnetic pole position based on the position error extraction amount obtained from the first position error information extraction means and the position error extraction amount obtained from the second position error information extraction means. Thus, the output frequency is converged. The first position error information extraction unit and the second position error information extraction unit sequentially add the multiplication values to the same memory area when adding the multiplication values over each phase interval of the high frequency voltage.

第1の実施形態を示す電動機制御装置のブロック構成図The block block diagram of the motor control apparatus which shows 1st Embodiment. 重畳する正弦波状関数を示す図Diagram showing superimposed sinusoidal function M−T軸座標とD−Q軸座標との関係を示す図The figure which shows the relationship between MT axis coordinates and DQ axis coordinates パルス状の高周波電圧vd(k)を重畳して高周波成分ΔIqhを抽出する過程で用いる信号値および算出値を示す図The figure which shows the signal value and calculation value which are used in the process of extracting the high frequency component ΔIqh by superimposing the pulsed high frequency voltage vd (k) 第2の実施形態を示す図4相当図FIG. 4 equivalent view showing the second embodiment 第3の実施形態を示す図4相当図FIG. 4 equivalent view showing the third embodiment

(第1の実施形態)
図1ないし図4を参照しながら第1の実施形態を説明する。図1は、回転子磁極位置の推定機能を備えた電動機制御装置の構成を示している。電動機制御装置1は、速度センサと位置センサを用いることなく永久磁石同期電動機2を駆動するセンサレス制御装置である。インバータ3は、3相ブリッジ接続された6つのスイッチング素子から構成され、直流電源線4p、4n間の直流電圧を指令電圧vuref、vvref、vwrefに基づいて3相交流電圧に変換し、電動機2に供給する。U相およびW相に配された電流検出器5は、電動機2に流れる電流を検出する電流検出手段である。
(First embodiment)
The first embodiment will be described with reference to FIGS. 1 to 4. FIG. 1 shows a configuration of an electric motor control device having a function of estimating a rotor magnetic pole position. The motor control device 1 is a sensorless control device that drives the permanent magnet synchronous motor 2 without using a speed sensor and a position sensor. The inverter 3 is composed of six switching elements connected in a three-phase bridge. The inverter 3 converts a DC voltage between the DC power supply lines 4p and 4n into a three-phase AC voltage based on the command voltages uref, vvref, and vwref. Supply. The current detectors 5 arranged in the U phase and the W phase are current detection means for detecting the current flowing through the electric motor 2.

電動機制御装置1は、回転子磁束の推定軸(推定した回転子磁束方向)をD軸とし、D軸に対しπ/2進んだ位置にあるトルク電流方向の軸をQ軸とする。そして、電動機2に高周波電圧vdhfを印加して固定子巻線に高周波電流を流すことにより、回転子磁束方向である回転子磁極位置θest(推定磁極位置)を推定し、この推定磁極位置に基づいて電動機2を制御する。   The motor control device 1 uses the estimated axis of the rotor magnetic flux (estimated rotor magnetic flux direction) as the D axis, and sets the axis in the torque current direction at a position advanced by π / 2 with respect to the D axis as the Q axis. Then, the rotor magnetic pole position θest (estimated magnetic pole position) that is the direction of the rotor magnetic flux is estimated by applying a high frequency voltage vdhf to the motor 2 and causing a high frequency current to flow through the stator winding, and based on this estimated magnetic pole position. To control the motor 2.

座標変換部6は、推定した回転子磁極位置θestを用いて、3相静止軸上の検出電流iu、iwを同期DQ軸上の検出電流id、iqに変換する。電流制御部7は、指令電流idref、iqrefと検出電流id、iqとを一致させるように、例えば電流偏差に対するPI演算により指令電圧vdref、vqrefを出力する。   The coordinate conversion unit 6 converts the detected currents iu and iw on the three-phase stationary axis into the detected currents id and iq on the synchronous DQ axis by using the estimated rotor magnetic pole position θest. The current control unit 7 outputs the command voltages vdref and vqref, for example, by PI calculation with respect to the current deviation so that the command currents idref and iqref and the detected currents id and iq match.

高周波電圧印加手段である加算器8は、回転子磁極位置θestを推定するために、D軸の電流制御結果である指令電圧vdrefに高周波電圧vdhfを加算する。重畳する高周波電圧vdhfは正弦波状関数である。正弦波状関数は、正弦波関数と同様に、高周波電圧vdhfの位相が0からπまでの期間で正の値を持ち、πから2πまでの期間で負の値を持つ関数である。正弦波状関数を例示すれば、図2に示すように(a)正弦波電圧(正弦波関数)、(b)相異なる2つの電圧レベルを有する50%デューティのパルス電圧(例えば正と負の向きに同じ電圧振幅を持つ方形波電圧(方形波関数))、(c)三角波電圧(三角波関数)、(d)鋸波電圧(鋸波関数)などである。本実施形態では、後述するように正弦波電圧と方形波電圧(パルス電圧)を例に説明する。   An adder 8 serving as a high-frequency voltage application means adds the high-frequency voltage vdhf to the command voltage vdref that is the current control result of the D axis in order to estimate the rotor magnetic pole position θest. The superposed high-frequency voltage vdhf is a sinusoidal function. Similar to the sine wave function, the sine wave function is a function having a positive value in the period from 0 to π and a negative value in the period from π to 2π, as in the case of the high frequency voltage vdhf. As an example of a sine wave function, as shown in FIG. 2, (a) a sine wave voltage (sine wave function), (b) a 50% duty pulse voltage having two different voltage levels (for example, positive and negative directions) Square wave voltage (square wave function)), (c) triangular wave voltage (triangular wave function), (d) sawtooth voltage (sawtooth wave function), and the like. In this embodiment, as will be described later, a sine wave voltage and a square wave voltage (pulse voltage) will be described as an example.

電流制御の応答は、重畳する高周波電圧vdhfの周波数より十分に遅い応答として調整されている。すなわち、ここで言う高周波とは、電動機2の運転周波数に対して十分に高い周波数であって、電流制御の周波数帯域に対しても十分に高い周波数である。   The response of the current control is adjusted as a response sufficiently slower than the frequency of the superimposed high frequency voltage vdhf. That is, the high frequency mentioned here is a sufficiently high frequency with respect to the operating frequency of the electric motor 2 and a sufficiently high frequency with respect to the frequency band of current control.

座標変換部9は、推定した回転子磁極位置θestを用いて、同期DQ軸上の指令電圧vdref、vqrefを3相静止軸上の指令電圧vuref、vvref、vwrefに変換する。指令電圧vuref、vvref、vwrefは、三角波からなる搬送波と比較されてPWM変調され、そのPWM変調された駆動信号は、インバータ3のスイッチング素子に付与される。   The coordinate conversion unit 9 converts the command voltages vdref and vqref on the synchronous DQ axis into command voltages vuref, vvref and vwref on the three-phase stationary axis using the estimated rotor magnetic pole position θest. The command voltages vuref, vvref, and vwref are compared with a carrier wave formed of a triangular wave and subjected to PWM modulation, and the PWM-modulated drive signal is applied to the switching element of the inverter 3.

位置誤差情報抽出部10(位置誤差情報抽出手段)は、詳しくは後述するように余弦波状関数の極性反転信号をD軸検出電流idおよびQ軸検出電流iqに乗算し、それぞれその乗算値を高周波電圧vdhfのM周期(M=1、2、…)にわたり加算平均して、推定した回転子磁極位置θestの誤差Δθを含む位置誤差抽出量(後述する高周波成分ΔIdh、ΔIqhに相当)を演算する。   As will be described in detail later, the position error information extraction unit 10 (position error information extraction means) multiplies the polarity inversion signal of the cosine wave function by the D axis detection current id and the Q axis detection current iq, and respectively multiplies the multiplied values by high frequency. A position error extraction amount (corresponding to high-frequency components ΔIdh and ΔIqh described later) including the error Δθ of the estimated rotor magnetic pole position θest is calculated by averaging over M periods (M = 1, 2,...) Of the voltage vdhf. .

収束演算部11(収束演算手段)は、位置誤差抽出量に基づいて回転子磁極位置θestの誤差Δθを求め、この誤差Δθを打ち消すように回転子速度ωstat(出力周波数)を収束演算する。積分器12は、回転子速度ωstatを積分して回転子磁極位置θestを得る。なお、上述した座標変換、電流制御、高周波電圧の重畳、PWM変調、位置誤差抽出量の演算、収束演算等の処理は、予め不揮発性メモリに記憶された制御プログラムに従ってマイクロコンピュータにより実行されるようになっている。   The convergence calculation unit 11 (convergence calculation means) obtains an error Δθ of the rotor magnetic pole position θest based on the position error extraction amount, and performs a convergence calculation on the rotor speed ωstat (output frequency) so as to cancel this error Δθ. The integrator 12 integrates the rotor speed ωstat to obtain the rotor magnetic pole position θest. The above-described processing such as coordinate conversion, current control, high-frequency voltage superposition, PWM modulation, position error extraction amount calculation, convergence calculation, and the like is executed by a microcomputer in accordance with a control program stored in advance in a nonvolatile memory. It has become.

以下、位置誤差情報抽出部10および収束演算部11による回転子磁極位置θestの推定演算について詳しく説明する。回転子に磁束を有する回転機の数学的モデルは、回転子の磁束方向をM軸とし、M軸からπ/2進んだ位置をT軸と定めると、一般的に(1)式のように示される。iM、iTはM軸電流、T軸電流、Rは固定子巻線の抵抗、Ld、Lqは固定子巻線のD軸、Q軸インダクタンス、φは回転子の固定子鎖交磁束、ωmeは回転速度、pは微分演算子である。

Figure 0005543388
Hereinafter, the estimation calculation of the rotor magnetic pole position θest by the position error information extraction unit 10 and the convergence calculation unit 11 will be described in detail. In a mathematical model of a rotating machine having a magnetic flux in the rotor, when the magnetic flux direction of the rotor is defined as the M axis and the position advanced by π / 2 from the M axis is defined as the T axis, generally, Indicated. iM and iT are M-axis current, T-axis current, R is the resistance of the stator winding, Ld and Lq are D-axis and Q-axis inductance of the stator winding, φ is the stator flux linkage of the rotor, and ωme is The rotational speed, p is a differential operator.
Figure 0005543388

電動機2の実際の磁束方向であるM軸方向をθとし、M軸とD軸の誤差角Δθ(推定磁極位置の誤差Δθ)を(2)式のように定める。このM−T軸とD−Q軸との関係は、図3に示す通りである。

Figure 0005543388
The M-axis direction, which is the actual magnetic flux direction of the electric motor 2, is defined as θ, and the error angle Δθ between the M-axis and the D-axis (the estimated magnetic pole position error Δθ) is determined as shown in Equation (2). The relationship between the MT axis and the DQ axis is as shown in FIG.
Figure 0005543388

制御上の推定軸であるD軸からπ/2進んだ位置をQ軸と定義したので、(1)式に示されるM−T軸上の電圧方程式は、D−Q軸上における数学的モデルとして(3)式のように表される。

Figure 0005543388
Since the position advanced by π / 2 from the D axis, which is an estimated axis for control, is defined as the Q axis, the voltage equation on the MT axis shown in the equation (1) is a mathematical model on the DQ axis. (3) is expressed as follows.
Figure 0005543388

微分項を左辺に整理すると(4)式のようになる。

Figure 0005543388
When the differential term is arranged on the left side, the following equation (4) is obtained.
Figure 0005543388

ここで、L0、L1は(5)式のように定義される。

Figure 0005543388
Here, L0 and L1 are defined as in equation (5).
Figure 0005543388

重畳する高周波電圧vdhfの周波数は電動機2の運転周波数に対して十分に高いため、巻線抵抗の電圧降下はインダクタンスに係る項に比べ十分に小さくなる。そこで、固定子巻線の抵抗Rに係る項を無視し、さらに停止時や低速域において小さな値となる回転速度ωmeを含む項を無視すると、(4)式は(6)式の近似式で表すことができる。

Figure 0005543388
Since the frequency of the superimposed high-frequency voltage vdhf is sufficiently higher than the operating frequency of the electric motor 2, the voltage drop of the winding resistance is sufficiently smaller than the term related to the inductance. Therefore, ignoring the term related to the resistance R of the stator winding and further ignoring the term including the rotational speed ωme that becomes a small value at the time of stopping or in the low speed region, the equation (4) is an approximation of the equation (6). Can be represented.
Figure 0005543388

高周波電圧信号をD軸方向にのみ重畳すると、vd=vdhf、vq=0となるので、(6)式は(7)式となる。

Figure 0005543388
If the high-frequency voltage signal is superimposed only in the D-axis direction, vd = vdhf and vq = 0, so that equation (6) becomes equation (7).
Figure 0005543388

これをD軸成分、Q軸成分ごとに表すと(8)式、(9)式となる。

Figure 0005543388
When this is expressed for each of the D-axis component and the Q-axis component, the following equations (8) and (9) are obtained.
Figure 0005543388

これより、D軸電流idとQ軸電流iqに推定磁極位置誤差Δθの2倍に相当する位置誤差情報が含まれることが分かる。そこで、D軸に重畳する高周波電圧vdhfを正弦波関数である正弦波電圧とすると(10)式のように表現できる。ただし、ΔVhは、振幅を表し正の値とする。

Figure 0005543388
From this, it can be seen that the D-axis current id and the Q-axis current iq include position error information corresponding to twice the estimated magnetic pole position error Δθ. Therefore, when the high-frequency voltage vdhf superimposed on the D-axis is a sine wave voltage that is a sine wave function, it can be expressed as in equation (10). However, ΔVh represents an amplitude and is a positive value.
Figure 0005543388

このとき、(6)式は(11)式となる。

Figure 0005543388
At this time, equation (6) becomes equation (11).
Figure 0005543388

正弦波状関数である高周波電圧vdhfを重畳した際にどのような電流が流れるかを考察するため、(11)式の両辺を積分すると(12)式となる。id0、iq0は、積分を開始し始めた時刻t=0における初期値である。(12)式の高周波成分のみを書き出すと(13)式となる。

Figure 0005543388
In order to consider what kind of current flows when the high-frequency voltage vdhf, which is a sinusoidal function, is superimposed, equation (12) is obtained by integrating both sides of equation (11). id0 and iq0 are initial values at time t = 0 when the integration is started. If only the high-frequency component of equation (12) is written, equation (13) is obtained.
Figure 0005543388

(13)式によれば、高周波電流は、印加している正弦波高周波電圧vdhfに対してπ/2だけ位相が遅れた高周波電流、言い換えると余弦波の極性反転成分(余弦波状関数の極性反転信号)として観測される。(5)式によればL0−L1cos(2Δθ)≧0なので、推定磁極位置誤差Δθに応じて、D軸に関してはその振幅が変化し、Q軸に関しては振幅のみならず極性も変化することが分かる。   According to the equation (13), the high-frequency current is a high-frequency current whose phase is delayed by π / 2 with respect to the applied sine wave high-frequency voltage vdhf, in other words, the polarity reversal component of the cosine wave (polarity reversal of the cosine wave function) Signal). According to the equation (5), L0−L1cos (2Δθ) ≧ 0, so that the amplitude of the D axis changes according to the estimated magnetic pole position error Δθ, and not only the amplitude but also the polarity of the Q axis changes. I understand.

推定磁極位置誤差Δθに応じて変化する高周波電流の振幅および極性を表す部分(位置誤差抽出量)をD軸、Q軸それぞれ高周波成分ΔIdh、ΔIqhと表記すれば、(13)式は(14)式のようになる。

Figure 0005543388
If the portion (position error extraction amount) representing the amplitude and polarity of the high-frequency current that changes in accordance with the estimated magnetic pole position error Δθ is expressed as high-frequency components ΔIdh and ΔIqh on the D axis and Q axis, equation (13) becomes (14) It becomes like the formula.
Figure 0005543388

(14)式は、高周波成分ΔIdh、ΔIqhが推定磁極位置誤差Δθに応じて一意に決定することを意味している。電動機2の検出電流値から高周波成分ΔIdh、ΔIqhを抽出できれば、(15)式で示す関係式を得ることができる。

Figure 0005543388
Equation (14) means that the high frequency components ΔIdh and ΔIqh are uniquely determined according to the estimated magnetic pole position error Δθ. If the high-frequency components ΔIdh and ΔIqh can be extracted from the detected current value of the electric motor 2, the relational expression represented by the equation (15) can be obtained.
Figure 0005543388

(15)式において、推定磁極位置誤差Δθの2倍角に対して正弦波状の情報となる分子部分に相当する高周波成分ΔIqhのみを用いて位置推定を行ってもよい。さらに、cos2Δθ≒1、sinΔθ≒2Δθの近似を施せば、(15)式は(16)式のように変形できる。

Figure 0005543388
In equation (15), position estimation may be performed using only the high-frequency component ΔIqh corresponding to the numerator portion that is sinusoidal information with respect to the double angle of the estimated magnetic pole position error Δθ. Further, if approximation of cos2Δθ≈1 and sinΔθ≈2Δθ is performed, Expression (15) can be transformed into Expression (16).
Figure 0005543388

Q軸の高周波成分ΔIqhとともにD軸の高周波成分ΔIdhも用いることで、推定磁極位置誤差Δθの情報は(17)式のように表せる。

Figure 0005543388
By using the high-frequency component ΔIdh of the D-axis together with the high-frequency component ΔIqh of the Q-axis, information on the estimated magnetic pole position error Δθ can be expressed as shown in Equation (17).
Figure 0005543388

(17)式によれば、高周波成分ΔIdh、ΔIqhの情報から推定磁極位置誤差Δθの情報を得る際、電動機2について得られたインダクタンス値Ld、Lqを用いることにより、位置推定のための収束演算用のゲインを最適に保つことができることが分かる。すなわち、回転子速度ωstatのPI収束演算において、比例項のゲインを1(最適値)に設定することができ、ゲイン調整を行う必要がないという効果が得られる。なお、(17)式で分子部分に相当するQ軸の高周波成分ΔIqhのみを用いることとすれば、高周波成分ΔIdhを、例えば、ΔIdh=ΔVh/(ωh・Ld)のように定数化することにより、(17)式の除算演算に替えて乗算演算を用いることができる。上述したΔIdh=ΔVh/(ωh・Ld)の関係式は、(14)式のただし書に記載されたΔIdhの式において、Δθ=0とすることで導くことができる。 According to the equation (17), when obtaining information on the estimated magnetic pole position error Δθ from information on the high-frequency components ΔIdh and ΔIqh, the convergence calculation for position estimation is performed by using the inductance values Ld and Lq obtained for the motor 2. It can be seen that the gain can be kept optimal. That is, in the PI convergence calculation of the rotor speed ωstat, the gain of the proportional term can be set to 1 (optimum value), and there is an effect that it is not necessary to perform gain adjustment. If only the Q-axis high-frequency component ΔIqh corresponding to the molecular part is used in the equation (17), the high-frequency component ΔIdh is made constant by, for example, ΔIdh = ΔVh / (ωh · Ld). , (17), a multiplication operation can be used instead of the division operation. The relational expression of ΔIdh = ΔVh / (ωh · Ld) described above can be derived by setting Δθ = 0 in the expression of ΔIdh described in the proviso of Expression (14).

次に、推定磁極位置誤差Δθの式である(17)式等に含まれる高周波成分ΔIdh、ΔIqhを抽出する手段について説明する。D軸に印加する高周波電圧vdhfによって流れる高周波電流は、(14)式に示されるように、正弦波関数(sinωht成分)で与えた高周波電圧信号に対してπ/2だけ位相の遅れた電流となる。言い換えると、余弦波関数の極性反転信号(−cosωht成分)として表されていることが分かる。   Next, means for extracting the high-frequency components ΔIdh and ΔIqh included in equation (17), which is the equation of the estimated magnetic pole position error Δθ, will be described. The high-frequency current flowing by the high-frequency voltage vdhf applied to the D-axis is a current delayed in phase by π / 2 with respect to the high-frequency voltage signal given by the sine wave function (sinωht component), as shown in equation (14). Become. In other words, it can be seen that the cosine wave function is expressed as a polarity inversion signal (-cosωht component).

例えばQ軸の高周波電流の振幅と極性を表す高周波成分ΔIqhの値を抽出するためには、検出電流iqに−cosωhtを乗算して、高周波電圧vdhfの1周期(M=1の場合)であるTh間だけ積分することで(18)式のように得ることができる。   For example, in order to extract the value of the high-frequency component ΔIqh representing the amplitude and polarity of the Q-axis high-frequency current, the detection current iq is multiplied by −cosωht, and is one cycle of the high-frequency voltage vdhf (when M = 1). By integrating only for Th, it can be obtained as in equation (18).

Figure 0005543388
Figure 0005543388

計算式は省略するが、D軸の高周波電流の振幅を表す高周波成分ΔIdhについても、検出電流idに−cosωhtを乗算してTh間だけ積分することにより同様に計算できる。   Although the calculation formula is omitted, the high-frequency component ΔIdh representing the amplitude of the D-axis high-frequency current can be calculated in the same manner by multiplying the detection current id by −cosωht and integrating only for Th.

以上、(10)式に示す正弦波関数である高周波電圧vdhfを重畳する場合について説明したが、以下では方形波関数であるパルス状の高周波電圧vdhfを重畳する場合でも同様にして高周波成分ΔIdh、ΔIqhを抽出できることを説明する。   The case of superimposing the high-frequency voltage vdhf that is a sine wave function shown in the equation (10) has been described above, but in the following, the high-frequency component ΔIdh, The fact that ΔIqh can be extracted will be described.

図2(b)に示すように、重畳する正弦波状関数からなる高周波電圧vdhfを、正弦波状関数が正の期間(0〜π)では+Δvhとし、負の期間(π〜2π)は−Δvh(ただし、Δvh>0)となるパルス電圧とする。すなわち、時刻t=0からt=Th/2の期間は(19)式となり、時刻t=Th/2からt=Thの期間は(20)式で表せる。時刻t=Th以降は再び+Δvhを出力する。このように、Th/2の期間ごとに正負交互の電圧を切り替えて印加する場合を考える。

Figure 0005543388
As shown in FIG. 2B, the high-frequency voltage vdhf composed of the superimposed sinusoidal function is set to + Δvh when the sinusoidal function is positive (0 to π), and −Δvh () during the negative period (π to 2π). However, the pulse voltage is such that Δvh> 0). That is, the period from time t = 0 to t = Th / 2 is expressed by equation (19), and the period from time t = Th / 2 to t = Th can be expressed by equation (20). After time t = Th, + Δvh is output again. In this way, a case is considered where positive and negative alternating voltages are switched and applied every Th / 2 period.
Figure 0005543388

まず、流れる高周波電流がどのように表されるかについて考察する。時刻t=0からt=Th/2の期間における電流は(21)式で記載される。

Figure 0005543388
First, consider how the flowing high-frequency current is represented. The current in the period from time t = 0 to t = Th / 2 is described by equation (21).
Figure 0005543388

(21)式の両辺を積分すると、時刻t=0からt=Th/2までの期間における電流は(22)式のように表せる。id0、iq0は、積分を開始し始めた時刻t=0における初期値である。

Figure 0005543388
When both sides of the equation (21) are integrated, the current in the period from time t = 0 to t = Th / 2 can be expressed as the equation (22). id0 and iq0 are initial values at time t = 0 when the integration is started.
Figure 0005543388

(22)式より、時刻t=Th/2における電流値は(23)式で表せる。

Figure 0005543388
From equation (22), the current value at time t = Th / 2 can be expressed by equation (23).
Figure 0005543388

続いて、時刻t=Th/2からt=Thまでの期間における電流は(24)式で記載される。

Figure 0005543388
Subsequently, the current in the period from time t = Th / 2 to t = Th is described by equation (24).
Figure 0005543388

(24)式の両辺を積分すると、時刻t=Th/2からt=Thまでの期間における電流は(25)式のように表せる。

Figure 0005543388
When both sides of the equation (24) are integrated, the current in the period from the time t = Th / 2 to t = Th can be expressed as the equation (25).
Figure 0005543388

Q軸の高周波電流の振幅と極性を表す高周波成分ΔIqhの値を抽出するために、(22)式と(25)式で表されるQ軸電流iqに−cosωhtを乗算して、高周波電圧vdhfの1周期(M=1の場合)であるTh間だけ積分すると(26)式が得られる。   In order to extract the value of the high-frequency component ΔIqh representing the amplitude and polarity of the Q-axis high-frequency current, the Q-axis current iq represented by the equations (22) and (25) is multiplied by −cosωht to obtain the high-frequency voltage vdhf. (26) is obtained by integrating only during Th, which is one period (when M = 1).

Figure 0005543388
Figure 0005543388

ここで、

Figure 0005543388
であるので、
Figure 0005543388
となる。 here,
Figure 0005543388
So
Figure 0005543388
It becomes.

従って、(26)式の各項について以下のように計算される。

Figure 0005543388
Therefore, the calculation is performed as follows for each term of the equation (26).
Figure 0005543388

その結果、(26)式は(26′)式となる。

Figure 0005543388
As a result, Equation (26) becomes Equation (26 ′).
Figure 0005543388

これにより、高周波電圧vdhfとして方形波関数であるパルス電圧を印加した場合においても、正弦波電圧と同様の演算手法で高周波成分ΔIdh、ΔIqhを得ることができることが分かる。また、パルス電圧の振幅を正弦波の振幅と等しくしたことから、同振幅の正弦波の実効値と比較して4/π倍の高周波電流成分が含まれることが分かる。   Thus, it can be seen that even when a pulse voltage that is a square wave function is applied as the high-frequency voltage vdhf, the high-frequency components ΔIdh and ΔIqh can be obtained by the same calculation method as the sine wave voltage. Further, since the amplitude of the pulse voltage is made equal to the amplitude of the sine wave, it can be seen that a high-frequency current component that is 4 / π times higher than the effective value of the sine wave of the same amplitude is included.

以上説明したように、正弦波状関数の高周波電圧vdhfとして正弦波電圧または方形波電圧を推定の磁束軸であるD軸に印加し、そのときに流れる高周波電流に基づいて回転子磁極位置θestを推定する。この場合、検出したD軸電流id、Q軸電流iqに余弦波状関数の電圧(余弦波電圧)の極性反転信号−cosωhtを乗算し、その乗算値の高周波電圧のM周期間における加算平均値を演算することで、位置誤差抽出量である高周波成分ΔIdh、ΔIqhを得ることができる。そして、これら高周波成分ΔIdh、ΔIqhから推定磁極位置誤差Δθ(推定磁極位置の誤差情報)を得ることができる。この誤差情報に基づいて、推定磁極位置誤差Δθを打ち消すように回転子速度ωstatに対するPI補償などの収束演算を行うことで回転子速度ωstatを得ることができ、最終的に推定磁極位置誤差Δθがゼロとなるように収束させることができる。   As described above, a sine wave voltage or a square wave voltage is applied to the D axis, which is an estimated magnetic flux axis, as the high frequency voltage vdhf of the sine wave function, and the rotor magnetic pole position θest is estimated based on the high frequency current flowing at that time. To do. In this case, the detected D-axis current id and Q-axis current iq are multiplied by the polarity inversion signal −cosωht of the voltage of the cosine wave function (cosine wave voltage), and the addition average value during the M cycles of the high frequency voltage of the multiplied value is obtained. By calculating, it is possible to obtain high frequency components ΔIdh and ΔIqh which are position error extraction amounts. Then, the estimated magnetic pole position error Δθ (the estimated magnetic pole position error information) can be obtained from the high frequency components ΔIdh and ΔIqh. Based on this error information, the rotor speed ωstat can be obtained by performing a convergence operation such as PI compensation on the rotor speed ωstat so as to cancel the estimated magnetic pole position error Δθ. It can be converged to be zero.

以上説明した回転子磁極位置の推定制御を実際に実装する場合には、マイクロコンピュータを備えたディジタル処理装置を用いるため、離散値系かつ離散時間系であるサンプル制御系に変換しておくほうが都合がよい。そこで、上述した制御方法をサンプル制御系に変換する。はじめに、正弦波関数である高周波電圧を重畳する場合を説明する。(10)式で表される高周波電圧vdhfの位相を離散時間系の式として表すと、以下のようになる。

Figure 0005543388
When the estimation control of the rotor magnetic pole position described above is actually implemented, it is more convenient to convert it to a sample control system that is a discrete value system and a discrete time system because a digital processor equipped with a microcomputer is used. Is good. Therefore, the control method described above is converted into a sample control system. First, a case where a high-frequency voltage that is a sine wave function is superimposed will be described. The phase of the high-frequency voltage vdhf expressed by the equation (10) is expressed as follows in the discrete time system.
Figure 0005543388

重畳する高周波電圧vd(k)は(27)式のように表すことができる。

Figure 0005543388
The superposed high-frequency voltage vd (k) can be expressed as in equation (27).
Figure 0005543388

kは1からnまでの整数値であり、nは高周波成分の1周期間のサンプル数(ただし偶数)を表す。ここでは一例としてn=8の場合を考える。具体的には、キャリア周波数が4kHzで、毎キャリア周期の山ごとまたは谷ごとに電流を検出する。重畳する高周波電圧周波数に500Hzを選択すると、キャリア周期の8周期分が重畳する高周波電圧vd(k)の1周期となる。   k is an integer value from 1 to n, and n represents the number of samples (however, even number) during one period of the high frequency component. Here, a case where n = 8 is considered as an example. Specifically, the carrier frequency is 4 kHz, and current is detected for each peak or valley of each carrier cycle. If 500 Hz is selected as the high frequency voltage frequency to be superimposed, one cycle of the high frequency voltage vd (k) to be superimposed is 8 carrier cycles.

検出されるD軸、Q軸の高周波電流id(k)、iq(k)は、(28)式で示すサンプル制御系として表される。

Figure 0005543388
The detected D-axis and Q-axis high-frequency currents id (k) and iq (k) are expressed as a sample control system represented by equation (28).
Figure 0005543388

Q軸の高周波電流iq(k)の振幅と極性を表す高周波成分ΔIqhの値を抽出するためには、(29)式で示すように検出したQ軸電流iq(k)に−cos(k(2π/n))を乗算して高周波電圧vd(k)の1周期(M=1の場合)である1≦k≦nの区間で加算平均を演算すればよい。

Figure 0005543388
In order to extract the value of the high-frequency component ΔIqh representing the amplitude and polarity of the Q-axis high-frequency current iq (k), the detected Q-axis current iq (k) is expressed as −cos (k ( 2π / n)) is multiplied to calculate the addition average in a section of 1 ≦ k ≦ n, which is one cycle of the high-frequency voltage vd (k) (when M = 1).
Figure 0005543388

上述した連続時間系として算出した(18)式と同様に考えることで、Q軸の高周波成分ΔIqhを(30)式として得ることができる。結果は省略するが、D軸の高周波電流id(k)の振幅を表す高周波成分ΔIdhも同様にして得ることができる。   By considering in the same manner as the equation (18) calculated as the continuous time system described above, the high-frequency component ΔIqh of the Q axis can be obtained as the equation (30). Although the result is omitted, the high-frequency component ΔIdh representing the amplitude of the D-axis high-frequency current id (k) can be obtained in the same manner.

Figure 0005543388
Figure 0005543388

続いて、方形波関数であるパルス状の高周波電圧を重畳する場合について説明する。パルス電圧を重畳する場合、サンプル制御系では(31)式のように表せる。

Figure 0005543388
Next, a case where a pulsed high-frequency voltage that is a square wave function is superimposed will be described. When superimposing the pulse voltage, it can be expressed by the equation (31) in the sample control system.
Figure 0005543388

1≦k≦n/2の範囲での電流検出値id(k)、iq(k)は(32)式のように表せる。

Figure 0005543388
The detected current values id (k) and iq (k) in the range of 1 ≦ k ≦ n / 2 can be expressed by the equation (32).
Figure 0005543388

k=n/2+1すなわちt=Th/2の時における電流値は(23)式と同様にして(33)式となる。

Figure 0005543388
The current value when k = n / 2 + 1, that is, t = Th / 2 is expressed by equation (33) in the same manner as equation (23).
Figure 0005543388

n/2+1≦k≦nの範囲での電流検出値id(k)、iq(k)は(34)式のように表せる。

Figure 0005543388
The detected current values id (k) and iq (k) in the range of n / 2 + 1 ≦ k ≦ n can be expressed by the equation (34).
Figure 0005543388

Q軸の高周波成分ΔIqhを抽出するために、検出された電流値を示す(32)式と(34)式で表される電流値id(k)、iq(k)に−cos(k(2π/n))を乗算して高周波電圧vd(k)の1周期である1≦k≦nの区間で加算平均を演算すると、Q軸について(35)式が得られる。結果は省略するが、D軸の高周波成分ΔIdhも同様にして得ることができる。   In order to extract the high-frequency component ΔIqh of the Q axis, the current values id (k) and iq (k) represented by the equations (32) and (34) indicating the detected current value are set to −cos (k (2π / N)) and calculating the addition average in the section of 1 ≦ k ≦ n, which is one cycle of the high-frequency voltage vd (k), the equation (35) is obtained for the Q axis. Although the result is omitted, the D-axis high-frequency component ΔIdh can be obtained in the same manner.

Figure 0005543388
Figure 0005543388

ここで、(35)式中の加算演算に関して連続系の演算結果に対する誤差が小さいと仮定すると、

Figure 0005543388
である。 Here, assuming that the error with respect to the continuous operation result is small with respect to the addition operation in equation (35),
Figure 0005543388
It is.

よって、(35)式は(35′)式となる。

Figure 0005543388
Therefore, equation (35) becomes equation (35 ').
Figure 0005543388

すなわち、サンプル制御系においても連続系の演算と同様にして高周波成分ΔIdh、ΔIqhを抽出することができる。その後の収束演算は、(15)式ないし(17)式に基づいて推定磁極位置誤差Δθを算出し、その推定磁極位置誤差Δθを打ち消すように回転子速度ωstatに対するPI補償などの収束演算を行えばよい。   That is, in the sample control system, the high frequency components ΔIdh and ΔIqh can be extracted in the same manner as in the continuous calculation. In the subsequent convergence calculation, an estimated magnetic pole position error Δθ is calculated based on the equations (15) to (17), and a convergence calculation such as PI compensation is performed on the rotor speed ωstat so as to cancel the estimated magnetic pole position error Δθ. Just do it.

図4は、サンプル制御系においてパルス状の高周波電圧vd(k)を重畳しQ軸の高周波成分ΔIqhを抽出する過程で用いる各種信号値および算出値を示している。先に述べたように、重畳する高周波電圧が正弦波関数である場合と同様に扱うことができる。同図(a)は搬送波を表している。電流検出器5は、この搬送波に同期して電動機2の電流を検出する。ここでは、搬送波の谷のタイミングで電流の検出が行われる例を示している。   FIG. 4 shows various signal values and calculated values used in the process of extracting the high frequency component ΔIqh of the Q axis by superimposing the pulsed high frequency voltage vd (k) in the sample control system. As described above, the high-frequency voltage to be superimposed can be handled in the same manner as when it is a sine wave function. FIG. 4A shows a carrier wave. The current detector 5 detects the current of the electric motor 2 in synchronization with this carrier wave. Here, an example is shown in which current is detected at the timing of a carrier wave valley.

同図(b)に示すように、印加する高周波電圧vd(k)の周期を電流検出周期Tsampの2N倍(N=1、2、…)となるように、すなわち高周波電圧vd(k)の半周期を電流検出周期TsampのN倍となるように選択している。これは、高周波電圧vd(k)の正の半周期と負の半周期とで、加算する乗算値を同数にするためである。これに伴い、同図(c)に示す高周波電圧位相θh(k)は、(2π)÷(2N)を単位とする離散的な位相として取り扱われる。なお、上述したサンプル数nとNとはn=2Nの関係がある。   As shown in FIG. 5B, the cycle of the applied high frequency voltage vd (k) is 2N times (N = 1, 2,...) The current detection cycle Tsamp, that is, the high frequency voltage vd (k) The half cycle is selected to be N times the current detection cycle Tsamp. This is for the same number of multiplication values to be added in the positive half cycle and the negative half cycle of the high-frequency voltage vd (k). Accordingly, the high-frequency voltage phase θh (k) shown in FIG. 5C is handled as a discrete phase with (2π) ÷ (2N) as a unit. Note that the number of samples n and N described above have a relationship of n = 2N.

このサンプル制御系において、座標変換部6は検出された電流値をD軸電流id(k)とQ軸電流iq(k)に変換する。位置誤差情報抽出部10は、同図(d)に示す離散的な位相を持つ余弦波の極性反転信号である−cos(θh(k))=−cos(k(2π/n))を生成し、それをD軸電流id(k)およびQ軸電流iq(k)とそれぞれ乗算する。そして、その乗算結果である同図(f)で示す乗算値(Q軸のみ示す)を高周波電圧vd(k)の1周期(M=1の場合)にわたり加算平均することで、回転子磁極位置θestの誤差Δθを含む高周波成分ΔIdh、ΔIqhを得ている(同図(g)、(h);Q軸のみ示す)。   In this sample control system, the coordinate conversion unit 6 converts the detected current value into a D-axis current id (k) and a Q-axis current iq (k). The position error information extraction unit 10 generates -cos (θh (k)) =-cos (k (2π / n)), which is a polarity inversion signal of a cosine wave having a discrete phase shown in FIG. Then, it is multiplied by the D-axis current id (k) and the Q-axis current iq (k), respectively. Then, the multiplication value (only the Q axis is shown) shown in FIG. 6 (f), which is the multiplication result, is added and averaged over one period (when M = 1) of the high-frequency voltage vd (k), so that the rotor magnetic pole position High-frequency components ΔIdh and ΔIqh including an error Δθ of θest are obtained ((g) and (h) in the figure; only the Q axis is shown).

この演算処理では、加算平均を行う際に、離散的な各位相θh(k)(k=1、2、…)について、電流値id(k)、iq(k)と余弦関数の極性反転信号−cos(θh(k))との乗算値をそれぞれ異なるメモリに記憶せず、サンプル周期ごとに同一のメモリ領域に順次加算している。これにより、揮発性メモリ例えばRAMの使用量を節約でき、ディジタル処理装置のコストダウンが可能となる。   In this arithmetic processing, when performing averaging, the current values id (k), iq (k) and the polarity inversion signal of the cosine function are obtained for each discrete phase θh (k) (k = 1, 2,...). The multiplication value with −cos (θh (k)) is not stored in different memories, but is sequentially added to the same memory area every sampling period. Thereby, the amount of volatile memory such as RAM can be saved, and the cost of the digital processing apparatus can be reduced.

以上述べたように、電動機制御装置1は、離散的な位相θh(k)の正弦波状関数の高周波電圧vd(k)をD軸電圧に重畳し、離散的な位相の余弦波状関数の極性反転信号−cos(θh(k))を検出したD軸電流id(k)およびQ軸電流iq(k)に乗算し、その乗算値を高周波電圧vd(k)のM周期(M=1、2、…)にわたり加算平均することで、回転子磁極位置θestの誤差情報Δθを含む高周波成分ΔIdh、ΔIqhを算出する。そして、この高周波成分ΔIdh、ΔIqhに基づいて推定磁極位置誤差Δθを打ち消すように回転子速度ωstatを収束演算する。   As described above, the motor control apparatus 1 superimposes the high-frequency voltage vd (k) of the sine wave function having the discrete phase θh (k) on the D-axis voltage, and reverses the polarity of the cosine wave function having the discrete phase. The signal -cos (θh (k)) is multiplied by the detected D-axis current id (k) and Q-axis current iq (k), and the multiplied value is M cycles (M = 1, 2) of the high-frequency voltage vd (k). ,...), The high frequency components ΔIdh and ΔIqh including the error information Δθ of the rotor magnetic pole position θest are calculated. Based on the high frequency components ΔIdh and ΔIqh, the rotor speed ωstat is converged so as to cancel the estimated magnetic pole position error Δθ.

この構成によれば、重畳している高周波電圧vd(k)のM周期間の電流検出結果に基づいて高周波成分ΔIdh、ΔIqhを抽出することにより、推定磁極位置誤差Δθの振幅(大きさ)と極性を同時に得ることができる。その結果、推定磁極位置誤差Δθをより確実に且つ安定して収束させることが可能となる。しかも、(17)式に示した演算により推定磁極位置誤差Δθを求めれば、収束演算用のゲインを最適に保つことができ、収束演算のためのゲイン設計が不要となる。   According to this configuration, by extracting the high-frequency components ΔIdh and ΔIqh based on the current detection result for M periods of the superimposed high-frequency voltage vd (k), the amplitude (size) of the estimated magnetic pole position error Δθ and Polarity can be obtained simultaneously. As a result, the estimated magnetic pole position error Δθ can be more reliably and stably converged. Moreover, if the estimated magnetic pole position error Δθ is obtained by the calculation shown in the equation (17), the gain for convergence calculation can be kept optimal, and the gain design for the convergence calculation becomes unnecessary.

高周波電圧vd(k)のM周期間で検出された複数の電流検出結果を用いて高周波成分ΔIdh、ΔIqhを抽出しているので、たとえノイズの影響で電流検出値の一部に検出誤差が含まれていたとしても、検出誤差の影響が平均化されてノイズの影響を低減する効果も得られる。上記説明では高周波電圧vd(k)の1周期(M=1)についての導出過程を示したが、一般に高周波電圧vd(k)のM周期(M=1、2、…)にわたる加算平均を演算してもよい。Mの値を増やすことにより、高周波成分ΔIdh、ΔIqhを抽出するのに用いる電流検出値のサンプル数が増え、インバータ3のスイッチングノイズなどの影響を一層低減することができる。   Since the high-frequency components ΔIdh and ΔIqh are extracted using a plurality of current detection results detected during M periods of the high-frequency voltage vd (k), a detection error is included in part of the current detection value due to the influence of noise. Even if this is the case, the influence of the detection error is averaged, and the effect of reducing the influence of noise can be obtained. In the above description, the derivation process for one cycle (M = 1) of the high-frequency voltage vd (k) is shown. In general, the arithmetic mean of the high-frequency voltage vd (k) over M cycles (M = 1, 2,...) Is calculated. May be. By increasing the value of M, the number of samples of the current detection value used for extracting the high frequency components ΔIdh and ΔIqh increases, and the influence of switching noise and the like of the inverter 3 can be further reduced.

推定磁極位置の誤差情報を含む位置誤差抽出量としてD軸、Q軸の高周波成分ΔIdh、ΔIqhを抽出したが、推定磁極位置誤差Δθに応じて高周波電流の振幅と極性がともに変化するQ軸の高周波成分ΔIqhのみを抽出してもよい。この場合、D軸の高周波成分ΔIdhを一定値として推定磁極位置誤差Δθを算出すれば、(17)式の除算演算を省くことができる。   The D-axis and Q-axis high-frequency components ΔIdh and ΔIqh are extracted as position error extraction amounts including error information of the estimated magnetic pole position. The Q-axis of which both the amplitude and polarity of the high-frequency current change according to the estimated magnetic pole position error Δθ. Only the high frequency component ΔIqh may be extracted. In this case, if the estimated magnetic pole position error Δθ is calculated with the D-axis high-frequency component ΔIdh as a constant value, the division operation of equation (17) can be omitted.

(第2の実施形態)
図5は、第2の実施形態における図4相当図である。第1の実施形態に対し、位置誤差情報抽出部10による加算平均の演算処理方法に変形を加えている。第1の実施形態の演算処理方法を用いると、位置誤差抽出量である高周波成分ΔIdh、ΔIqhは、高周波電圧vd(k)の1周期ごとに更新される。このため、例えば電動機2のインダクタンスが大きく高周波に対するインピーダンスが大きいため、高周波電圧vd(k)の周波数を上げられない場合などには、位置誤差抽出量の更新周期が長くなり、位置推定の応答を下げざるを得ない制約が生じることが懸念される。その結果、速度応答および位置制御系の応答にも制約を生じる虞がある。
(Second Embodiment)
FIG. 5 is a view corresponding to FIG. 4 in the second embodiment. With respect to the first embodiment, a modification is added to the arithmetic processing method of the addition average by the position error information extraction unit 10. When the arithmetic processing method of the first embodiment is used, the high frequency components ΔIdh and ΔIqh, which are position error extraction amounts, are updated every cycle of the high frequency voltage vd (k). For this reason, for example, when the frequency of the high-frequency voltage vd (k) cannot be increased because the inductance of the motor 2 is large and the impedance to the high frequency is large, the position error extraction amount update period becomes long, and the position estimation response is increased. There are concerns that there will be constraints that must be lowered. As a result, the speed response and the response of the position control system may be restricted.

そこで、本実施形態では図5(g)、(h)に示すように、高周波成分ΔIqhを算出する区間として、高周波電圧位相θh(k)の0〜2πの区間で加算平均を算出するとともに、0〜2πの区間からπだけずらした区間でも加算平均を算出する。高周波成分ΔIdhの算出も同様である。これにより、高周波成分ΔIdh、ΔIqhの更新周期を高周波電圧vd(k)の半周期とすることができる。   Therefore, in the present embodiment, as shown in FIGS. 5G and 5H, as an interval for calculating the high frequency component ΔIqh, an addition average is calculated in the interval of 0 to 2π of the high frequency voltage phase θh (k), and The addition average is calculated even in a section shifted by π from a section of 0 to 2π. The same applies to the calculation of the high frequency component ΔIdh. Thereby, the update cycle of the high frequency components ΔIdh and ΔIqh can be set to a half cycle of the high frequency voltage vd (k).

一般に、乗算値を高周波電圧vd(k)のM周期(M=1、2、…)にわたり加算平均する場合でも、高周波電圧vd(k)の0から2Mπまでの位相区間および当該位相区間に対しMπだけ位相が異なる位相区間について、それぞれ乗算値を加算して高周波成分ΔIdh、ΔIqhを演算すれば、更新周期を2MπからMπに短縮できる。この演算処理方法によれば、加算平均値を保持する揮発性メモリの使用量が2倍に増加するものの、位置誤差抽出量の更新周期を2分の1にすることができるので、速度応答および位置制御応答を改善することができる。   In general, even when the multiplication values are averaged over M periods (M = 1, 2,...) Of the high-frequency voltage vd (k), the phase interval from 0 to 2Mπ of the high-frequency voltage vd (k) If the high frequency components ΔIdh and ΔIqh are calculated by adding multiplication values for the phase sections whose phases are different by Mπ, the update cycle can be shortened from 2Mπ to Mπ. According to this arithmetic processing method, although the amount of use of the volatile memory holding the addition average value is doubled, the position error extraction amount update cycle can be halved, so that the speed response and The position control response can be improved.

(第3の実施形態)
図6は、第3の実施形態における図4相当図である。第1、第2の実施形態に対し、位置誤差情報抽出部10による加算平均の演算処理方法が異なる。すなわち、位置誤差情報抽出部10は、図6(g)に示すように、検出した電流値id(k)、iq(k)と余弦関数の極性反転信号−cos(θh(k))との各乗算値を、重畳する高周波電圧vd(k)の1周期(M=1の場合)にわたり揮発性メモリの別々の領域に記憶し、それらデータ群1の加算平均を算出する。
(Third embodiment)
FIG. 6 is a diagram corresponding to FIG. 4 in the third embodiment. Compared to the first and second embodiments, the arithmetic processing method of the addition average by the position error information extraction unit 10 is different. That is, as shown in FIG. 6G, the position error information extraction unit 10 calculates the detected current values id (k), iq (k) and the cosine function polarity inversion signal -cos (θh (k)). Each multiplication value is stored in a separate area of the volatile memory over one period (when M = 1) of the high-frequency voltage vd (k) to be superimposed, and an average of the data group 1 is calculated.

次の電流検出値が得られたときには、同図(h)に示すように、高周波電圧vd(k)の1周期前の乗算値(図中破線で示す)に替えて、新たに得られた電流値id(k)、iq(k)と余弦関数の極性反転信号−cos(θh(k))との乗算値(図中ハッチングで示す)を当該1周期前の乗算値が記憶されていたメモリ領域に記憶(上書き)し、それらデータ群2の加算平均を算出する。   When the next detected current value was obtained, it was newly obtained instead of the multiplication value (indicated by the broken line in the figure) one cycle before the high frequency voltage vd (k) as shown in FIG. The multiplication value (indicated by hatching in the figure) of the current values id (k), iq (k) and the polarity inversion signal −cos (θh (k)) of the cosine function was stored in the previous cycle. It is stored (overwritten) in the memory area, and the average of the data group 2 is calculated.

一般に、乗算値を高周波電圧vd(k)のM周期(M=1、2、…)にわたり加算平均する場合でも、高周波電圧vd(k)の0から2Mπまでの位相区間に対し2MN個の乗算値を記憶するとともに加算して高周波成分ΔIdh、ΔIqhを演算し、以後、新たに乗算値を算出するごとに2Mπだけ前の乗算値に替えて当該新たに算出した乗算値を記憶するとともに加算して高周波成分ΔIdh、ΔIqhを演算すればよい。   In general, even when the multiplication values are averaged over M periods (M = 1, 2,...) Of the high-frequency voltage vd (k), 2MN multiplications are performed for the phase interval from 0 to 2Mπ of the high-frequency voltage vd (k). The value is memorized and added to calculate the high frequency components ΔIdh and ΔIqh. Thereafter, every time a new multiplication value is calculated, the newly calculated multiplication value is stored and added instead of the previous multiplication value by 2Mπ. Thus, the high frequency components ΔIdh and ΔIqh may be calculated.

この演算処理方法によれば、メモリの使用量は、加算平均を行うM周期分の乗算値データを保持するための領域が必要となるが、位置誤差抽出量の更新周期を電流検出周期Tsampと等しくできるので、速度制御応答および位置制御応答を高くとることができる。   According to this calculation processing method, the memory usage requires an area for holding the multiplication value data for M cycles for which the averaging is performed, but the update cycle of the position error extraction amount is defined as the current detection cycle Tsamp. Since they can be made equal, the speed control response and the position control response can be made high.

(その他の実施形態)
以上説明した実施形態に加えて以下のような構成を採用してもよい。
電動機2に高周波電流を重畳し、高周波電流を重畳しない軸の電圧値を用いて位置誤差抽出量を演算する方法においても同様の抽出方法を用いることができる。
(Other embodiments)
In addition to the embodiment described above, the following configuration may be adopted.
A similar extraction method can also be used in the method of calculating the position error extraction amount using the voltage value of the axis on which the high frequency current is not superimposed on the motor 2 and the high frequency current is not superimposed.

印加する高周波電圧の半周期を電流検出周期のN倍(N=1、2、…)となるように選択したが、N=1の場合には高周波電圧の1周期に2つの電流検出値のみを用いることになる。従って、Nは2以上の整数値に選択することが好ましい。   The half cycle of the high frequency voltage to be applied is selected to be N times the current detection cycle (N = 1, 2,...). When N = 1, only two current detection values are included in one cycle of the high frequency voltage. Will be used. Therefore, it is preferable to select N as an integer value of 2 or more.

以上説明した少なくとも一つの実施形態によれば、重畳している高周波電圧の少なくとも1周期間の電流検出結果に基づいて高周波成分を抽出することにより、推定磁極位置誤差の大きさと極性を同時に得ることができる。また、ノイズの影響で電流検出値の一部に検出誤差が含まれていたとしても、高周波電圧の少なくとも1周期間で行われた電流検出回数分の複数の電流検出結果を用いて高周波成分を抽出しているので、検出誤差の影響が平均化されてノイズの影響の低減効果が得られる。   According to at least one embodiment described above, the magnitude and polarity of the estimated magnetic pole position error can be obtained simultaneously by extracting a high frequency component based on a current detection result for at least one cycle of the superimposed high frequency voltage. Can do. Further, even if a detection error is included in a part of the current detection value due to the influence of noise, a high frequency component is obtained by using a plurality of current detection results corresponding to the number of times of current detection performed during at least one cycle of the high frequency voltage. Since the extraction is performed, the influence of the detection error is averaged, and the effect of reducing the influence of noise is obtained.

本発明のいくつかの実施形態を説明したが、これらの実施形態は、例として提示したものであり、発明の範囲を限定することは意図していない。これら実施形態は、その他の様々な形態で実施されることが可能であり、発明の要旨を逸脱しない範囲で、種々の省略、置き換え、変更を行うことができる。これら実施形態やその変形は、発明の範囲や要旨に含まれると同様に、特許請求の範囲に記載された発明とその均等の範囲に含まれるものである。   Although several embodiments of the present invention have been described, these embodiments are presented by way of example and are not intended to limit the scope of the invention. These embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the spirit of the invention. These embodiments and their modifications are included in the scope and gist of the invention, and are also included in the invention described in the claims and the equivalents thereof.

図面中、1は電動機制御装置(永久磁石同期電動機の制御装置)、2は永久磁石同期電動機、5は電流検出器(電流検出手段)、8は加算器(高周波電圧印加手段)、10は位置誤差情報抽出部(位置誤差情報抽出手段)、11は収束演算部(収束演算手段)である。   In the drawings, 1 is a motor control device (control device for a permanent magnet synchronous motor), 2 is a permanent magnet synchronous motor, 5 is a current detector (current detection means), 8 is an adder (high frequency voltage application means), and 10 is a position. An error information extraction unit (position error information extraction unit) 11 is a convergence calculation unit (convergence calculation unit).

Claims (4)

永久磁石同期電動機の回転子磁束の推定軸をD軸とし、D軸に対しπ/2進んだ位置にある軸をQ軸とし、永久磁石同期電動機に高周波電圧を印加することにより高周波電流を流して回転子の磁極位置を推定し、この推定磁極位置に基づいて永久磁石同期電動機を制御する永久磁石同期電動機の制御装置において、
永久磁石同期電動機に流れる電流を所定の電流検出周期で検出する電流検出手段と、
印加する高周波電圧の半周期を前記電流検出周期のN倍(N=1、2、…)となるように選択し、その高周波電圧の位相を(2π)÷(2N)を単位とする離散的な位相として取り扱い、この離散的な位相の正弦波状関数の電圧成分をD軸電圧に重畳する高周波電圧印加手段と、
前記離散的な位相の余弦波状関数の極性反転信号を前記検出したD軸電流およびQ軸電流のうち少なくともQ軸電流に乗算し、その乗算値を前記高周波電圧の0から2Mπ(M=1、2、…)までの位相区間にわたり加算して、前記推定磁極位置の誤差情報を含む位置誤差抽出量を演算する第1の位置誤差情報抽出手段と、
前記乗算値を、前記位相区間に対しMπだけ位相が異なる位相区間にわたり加算して、前記推定磁極位置の誤差情報を含む位置誤差抽出量を演算する第2の位置誤差情報抽出手段と、
前記第1の位置誤差情報抽出手段から得られた前記位置誤差抽出量と、前記第2の位置誤差情報抽出手段から得られた前記位置誤差抽出量に基づいて、前記推定磁極位置の誤差を打ち消すように出力周波数を収束演算する収束演算手段とを備え
前記第1の位置誤差情報抽出手段および前記第2の位置誤差情報抽出手段は、前記乗算値を前記高周波電圧の前記各位相区間にわたり加算する場合、前記乗算値を同一のメモリ領域に順次加算することを特徴とする永久磁石同期電動機の制御装置。
The estimated axis of the rotor magnetic flux of the permanent magnet synchronous motor is D axis, the axis that is π / 2 advanced from the D axis is Q axis, and a high frequency current is applied by applying a high frequency voltage to the permanent magnet synchronous motor. In the control device of the permanent magnet synchronous motor that estimates the magnetic pole position of the rotor and controls the permanent magnet synchronous motor based on the estimated magnetic pole position,
Current detection means for detecting the current flowing in the permanent magnet synchronous motor at a predetermined current detection period;
The half cycle of the high frequency voltage to be applied is selected to be N times the current detection cycle (N = 1, 2,...), And the phase of the high frequency voltage is discrete in units of (2π) ÷ (2N). High-frequency voltage applying means for treating the voltage component of the discrete phase sinusoidal function on the D-axis voltage,
The polarity inversion signal of the discrete phase cosine wave function is multiplied by at least the Q-axis current of the detected D-axis current and Q-axis current, and the multiplication value is 0 to 2Mπ (M = 1, 2,..., And first position error information extraction means for calculating a position error extraction amount including error information of the estimated magnetic pole position;
Second position error information extraction means for calculating the position error extraction amount including error information of the estimated magnetic pole position by adding the multiplication value over a phase period having a phase different by Mπ with respect to the phase interval;
Based on the position error extraction amount obtained from the first position error information extraction means and the position error extraction amount obtained from the second position error information extraction means, the error of the estimated magnetic pole position is canceled out. And a convergence calculation means for calculating a convergence of the output frequency ,
The first position error information extraction unit and the second position error information extraction unit sequentially add the multiplication values to the same memory area when adding the multiplication values over the phase intervals of the high-frequency voltage. A control device for a permanent magnet synchronous motor.
前記第1の位置誤差情報抽出手段および前記第2の位置誤差情報抽出手段は、それぞれ前記離散的な位相の余弦波状関数の極性反転信号を前記検出したD軸電流およびQ軸電流に乗算し、前記D軸電流の乗算値を前記高周波電圧の前記各位相区間にわたり加算して位置誤差抽出量ΔIdhを演算し、前記Q軸電流の乗算値を前記高周波電圧の前記各位相区間にわたり加算して位置誤差抽出量ΔIqhを演算し、
前記収束演算手段は、
Lq/(Ld−Lq)×(ΔIqh/ΔIdh)
ただし、LdはD軸インダクタンス、LqはQ軸インダクタンス
により前記推定磁極位置の誤差Δθを算出することを特徴とする請求項1記載の永久磁石同期電動機の制御装置。
The first position error information extraction means and the second position error information extraction means multiply the detected D-axis current and Q-axis current by the polarity inversion signals of the discrete phase cosine wave functions, respectively. A position error extraction amount ΔIdh is calculated by adding the multiplication value of the D-axis current over each phase interval of the high-frequency voltage, and the position is determined by adding the multiplication value of the Q-axis current over each phase interval of the high-frequency voltage. Calculate the error extraction amount ΔIqh,
The convergence calculation means includes:
Lq / (Ld−Lq) × (ΔIqh / ΔIdh)
2. The control apparatus for a permanent magnet synchronous motor according to claim 1, wherein Ld is D-axis inductance and Lq is Q-axis inductance to calculate the error [Delta] [theta] of the estimated magnetic pole position.
前記第1の位置誤差情報抽出手段および前記第2の位置誤差情報抽出手段は、それぞれ前記離散的な位相の余弦波状関数の極性反転信号を前記検出したQ軸電流に乗算し、前記Q軸電流の乗算値を前記高周波電圧の前記各位相区間にわたり加算して位置誤差抽出量ΔIqhを演算し、
前記収束演算手段は、
Lq/(Ld−Lq)×(ΔIqh/ΔIdh)
ただし、LdはD軸インダクタンス、LqはQ軸インダクタンス、ΔIdh=ΔVh/(ωh・Ld)、ΔVhは前記高周波電圧の振幅、ωhは前記高周波電圧の角周波数
により前記推定磁極位置の誤差Δθを算出することを特徴とする請求項1記載の永久磁石同期電動機の制御装置。
The first position error information extraction unit and the second position error information extraction unit multiply the detected Q-axis current by a polarity inversion signal of the cosine wave function having the discrete phase, respectively, and the Q-axis current To calculate the position error extraction amount ΔIqh by adding the multiplication value of the high frequency voltage over each phase interval of the high frequency voltage,
The convergence calculation means includes:
Lq / (Ld−Lq) × (ΔIqh / ΔIdh)
However, Ld is a D-axis inductance, Lq is a Q-axis inductance, ΔIdh = ΔVh / (ωh · Ld), ΔVh is an amplitude of the high-frequency voltage, and ωh is an error Δθ of the estimated magnetic pole position based on an angular frequency of the high-frequency voltage. The control device for a permanent magnet synchronous motor according to claim 1.
前記正弦波状関数は、正弦波関数、方形波関数、三角波関数または鋸波関数であることを特徴とする請求項1から3の何れか一項に記載の永久磁石同期電動機の制御装置。 The control apparatus for a permanent magnet synchronous motor according to any one of claims 1 to 3, wherein the sine wave function is a sine wave function, a square wave function, a triangular wave function, or a sawtooth function .
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