JP6150212B2 - Digital rotor phase speed estimation device for AC motor - Google Patents

Description

The present invention relates to an AC motor in which a rotor exhibits salient pole characteristics with respect to application of a high-frequency voltage having a frequency higher than the drive fundamental frequency (for example, a permanent magnet synchronous motor having a permanent magnet in the rotor, a wound synchronous motor, and a synchronous reluctance motor). , Rotor phase (synonymous with position), speed and position speed sensor used in drive control device for hybrid field type synchronous motor, induction motor, etc. with permanent magnet and field winding on rotor The present invention relates to a digital rotor phase speed estimation apparatus for estimation without using, that is, sensorless. In particular, the shape of the applied high-frequency voltage is rectangular, and the frequency of the applied high-frequency voltage is about the same as the switching frequency of the power converter (the carrier frequency in the case of PWM) (that is, the same or a fraction of that). The present invention relates to a digital rotor phase speed estimation device that has a critically high frequency.

High-performance control of the AC motor can be achieved by a so-called vector control method. The vector control method requires information on the rotor phase or speed, which is a differential value thereof, and a position speed sensor such as an encoder has been used for some time. However, the use of this type of position / velocity sensor is not preferable in terms of reliability, axial volume, sensor cable routing, cost, and the like, so-called sensorless vector control method that does not require a position / velocity sensor. Research and development has been conducted for many years.

As a powerful sensorless vector control method, a high-frequency voltage with a frequency higher than the drive fundamental frequency is forcibly applied to the motor, and the corresponding high-frequency current (response high-frequency current) is processed to estimate the rotor phase (so-called high-frequency voltage) Various application methods have been developed and reported so far. In the present specification, the constituent frequency components of the stator voltage and the stator current are divided into a driving fundamental wave component and a high-frequency component and divided into two. The driving fundamental wave component is a component directly related to the rotation speed of the motor (that is, a frequency component comparable to the rotor speed (electrical speed)), and the high frequency component is much higher than the driving fundamental wave component. It is a component of a very high frequency (frequency value is known). In particular, in the present invention, the high-frequency component is a component having a marginally high frequency that is about the same as the switching frequency of the power converter (the carrier frequency in the case of PWM) (that is, the same or a fraction thereof). means.

The rotor phase to be estimated may be determined at an arbitrary position of the rotor, but generally, either the negative salient pole phase or the positive salient pole phase of the rotor is selected as the rotor phase. As is well known to those skilled in the art, there is only an electrical phase deviation of ± π / 2 (rad) between the negative salient pole phase and the positive salient pole phase. The other phases are naturally found. Considering the above, hereinafter, the negative salient pole phase of the rotor will be referred to as the rotor phase unless otherwise specified. A biaxial orthogonal coordinate system composed of a d axis synchronized with the rotor phase without phase deviation and a q axis orthogonal to the d axis is called a dq synchronous coordinate system (see FIG. 1).

Although several technical classifications of the high-frequency voltage application method are conceivable, the first classification method is a classification method based on the shape of the applied high-frequency voltage. The high-frequency voltage application method is broadly classified into those in which the shape of the applied high-frequency voltage is a sine shape and a rectangular shape. The shape of the applied high-frequency voltage targeted by the present invention is rectangular.

The high frequency voltage application method of applying a rectangular high frequency voltage is further subdivided from the viewpoint of a coordinate system for applying the high frequency voltage. Specifically, a method of applying a high-frequency voltage on an αβ fixed coordinate system and a high-frequency voltage on a γδ quasi-synchronous coordinate system (a kind of rotating coordinate system) aiming at convergence with zero phase difference to a dq synchronous coordinate system Is subdivided into a method of applying (see FIG. 1). The high-frequency voltage application method targeted by the present invention is to apply a rectangular high-frequency voltage on the γδ quasi-synchronous coordinate system. Furthermore, a rectangular high-frequency voltage is applied only on the γ-axis. That is, no high-frequency voltage is applied on the δ axis.

In this specification, (1) a high-frequency voltage application method for applying a rectangular high-frequency voltage, (2) a rectangular high-frequency voltage is applied only on the γ-axis of the γδ quasi-synchronous coordinate system, and (3) The frequency of the applied high-frequency voltage satisfies the three conditions that the switching frequency of the power converter (carrier frequency in the case of PWM) is high (that is, the same or a fraction thereof). The high-frequency voltage application method is simply referred to as “rectangular limit high-frequency voltage application method”. As prior inventions of the rectangular limit high-frequency voltage application method to which the present invention belongs, there are Patent Document 1 and Non-Patent Documents 1 to 3 listed in the Prior Art Document column.

In the rectangular limit high-frequency voltage application method to which the present invention belongs, regarding the extraction of the rotor phase information and the generation of the rotor phase estimation value, there are different techniques depending on the coordinate system that performs processing of the stator current including high-frequency components. There is a characteristic that it is necessary. Based on this characteristic, the rectangular limit high-frequency voltage application method can be further classified into technologies based on a coordinate system that performs processing of a stator current including a high-frequency component. The coordinate system for processing the stator current is classified into an αβ fixed coordinate system and other coordinate systems. The technical classification of the prior invention based on the stator current processing coordinate system is organized as shown in Table 1 below.

The present invention performs processing of a stator current including a high frequency component on a γδ quasi-synchronous coordinate system. As in the present invention, there is Patent Document 1 as a prior invention in which a stator current including a high frequency component is processed on a γδ quasi-synchronous coordinate system, that is, as a prior invention belonging to the same technical field as the present invention. Hereinafter, the technique of Patent Document 1 will be outlined with reference to the drawings.

FIG. 5 relates to a drive control system for a permanent magnet synchronous motor, and is a typical configuration example of a digital drive control apparatus including a digital rotor phase speed estimation apparatus (phase speed estimator 10) based on a rectangular limit high-frequency voltage application method. It is. 1 is an AC motor (synchronous motor), 2 is a power converter (inverter), 3 is a discrete time current detector, 4a and 4b are 3 phase 2 phase converters, 2 phase 3 phase converters and 5a, respectively. 5b is a vector rotator, 6 is a current controller, 7 is a command converter, 8 is a speed controller, 9 is a low pass filter, 10 is a phase speed estimator, 11 is a coefficient unit, Reference numeral 12 denotes a cosine sine signal generator. In FIG. 5, various devices from 2 to 12 except for one electric motor constitute a drive control device. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

The three-phase stator current detected by the discrete-time current detector 3 is converted into a two-phase current on the αβ fixed coordinate system by the three-phase two-phase converter 4a, and then the rotor phase ( is converted into a two-phase current in a γδ quasi-synchronous coordinate system aimed at phase synchronization with zero phase deviation. A driving current (driving fundamental frequency component of the stator current) is extracted from the converted current through the low-pass filter 9 and is sent to the current controller 6. In the current controller 6, the driving two-phase current on the γδ quasi-synchronous coordinate system is the current command value of each phase (the initial symbol * of the current command value variable means the command value. In addition, the command value of the related signal is expressed by adding a prefix * to the related signal.) A two-phase voltage command value for driving on the γδ quasi-synchronous coordinate system is generated.
Here, the rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value) received from the phase velocity estimator 10 is superimposed on the γ-axis element of the driving two-phase voltage command value, and superimposed and synthesized. A voltage command value (the δ-axis voltage command value remains the drive voltage command value and remains unchanged) is sent to the vector rotator 5b. In 5b, the voltage command value for superposition and synthesis on the γδ quasi-synchronous coordinate system is converted into a two-phase voltage command value in the αβ fixed coordinate system, and sent to the two-phase three-phase converter 4b. In 4b, the two-phase voltage command value is converted into a three-phase voltage command value and output as a final command value to the power converter 2. The power converter 2 generates electric power according to the command value, applies it to the synchronous motor 1 and drives it.

The phase speed estimator 10 receives a stator current on the γδ quasi-synchronous coordinate system, which is an output signal of the vector rotator 5a, as an input, and receives a rotor phase estimated value, a rotor (electrical) speed estimated value, and a rectangular shape. The single-phase high-frequency voltage command value is output. The rotor phase estimation value is converted into a cosine / sine signal by the cosine sine signal generator 12, and then passed to the vector rotators 5a and 5b that determine the γδ quasi-synchronous coordinate system. This means that the rotor phase estimation value is the phase of the γδ quasi-synchronous coordinate system (equivalent to the phase of the γ axis).

As is well known to those skilled in the art, the two-phase current command value on the γδ quasi-synchronous coordinate system is obtained by converting the torque command value through the command converter 7. In the speed controller 8, the rotor speed estimated value (rotor electrical speed estimated value), which is one of the output signals from the phase speed estimator 10, is a constant value, and the coefficient unit 11 is used to calculate the reciprocal of the pole pair number Np. And then converted to a machine speed estimate and sent. In the example of FIG. 5, an example in which the speed control system is configured is shown, and thus the torque command value is obtained as the output of the speed controller 8. As is well known to those skilled in the art, the speed controller 8 is unnecessary when the control purpose is torque control and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

はγ軸からみたｄ軸の位相でもある。図１参照）。FIG. 10 shows an example of the internal structure of the phase velocity estimator 10 in FIG. As shown in the upper part of the figure, the phase velocity estimator 10 basically comprises a magnetic pole position estimating means and a pulsating voltage applying means. The pulsating voltage applying means generates a rectangular single-phase high-frequency voltage command value vγ * (denoted as vhd * in the original drawing). The generated single-phase high-frequency voltage command value vγ * (denoted as vhd * in the original drawing) is output to the outside and simultaneously sent to the magnetic pole position estimating means. The lower part of FIG. 10 shows the detailed internal configuration of the magnetic pole position estimation means used in the upper phase velocity estimator 10. The input signal of the magnetic pole position estimating means includes a γ-axis current (that is, a γ-axis element of the stator current, expressed as Idc in the original drawing) and a rectangular single-phase high-frequency voltage command value vγ * (indicated in the original drawing as Vhd *). And its output signal

Is also the phase of the d-axis viewed from the γ-axis. (See FIG. 1).

FIG. 11 shows a signal processed by the magnetic pole position estimating means and its temporal timing. The waveform diagram (b) in the figure shows a state of a rectangular single-phase high-frequency voltage command value in which a cycle twice the PWM carrier wave is one cycle. In addition, in the waveform diagram (f), the γ-axis current (dc-axis current in the original diagram) is detected in discrete time four times (8 times in one cycle) every half cycle of the rectangular single-phase high-frequency voltage command value. The situation is shown. In addition, the waveform diagram (h) shows a current polarity signal Sp indicating the polarity to be taken by the first-order time difference signal of the γ-axis current (indicated by Idc in the original drawing). The waveform diagram (i) shows the most important signal for generating an axis error generated using the current polarity signal Sp.

As clearly shown in the configuration diagram in the lower part of FIG. 10, the most important signal for generating the axis error is generated in the current change amount calculation means using the current polarity signal Sp as an input signal. This current polarity signal Sp is generated by processing a single-phase high-frequency voltage command value vγ * (indicated as Vhd * in the original drawing) in a current polarity calculator (in other words, it is generated in an estimated manner). ) However, in an actual system (drive control device), generation of a single-phase high-frequency voltage command value, delivery of the command value to the power converter, actual generation and application of a high-frequency voltage, generation and detection of a corresponding high-frequency current, Timing errors easily occur in the processes described. Due to this timing error, the current polarity signal Sp, which is an estimation signal, does not necessarily indicate the intended true polarity. In particular, before and after the true polarity is reversed, the current polarity signal Sp, which is an estimation signal, can easily be erroneously determined as positive or negative or negative as positive. In addition to this, as understood from the internal configuration of the magnetic pole position estimation means shown in the lower diagram of FIG. 10, the method of Patent Document 1 requires a complicated calculation to generate an expected phase estimation value. .

Daigo Kaneko, Yoshinao Iwaji, Kiyoshi Sakamoto, Tsunehiro Endo: “AC Motor Control Device and AC Motor System”, JP 2005-20918 (2003-6-27), Correction (2006-6-15) )

Ryozo Masaki, Satoru Kaneko, Yoshimi Sakurai, Mitsuyuki Motobu: “Position sensorless control system of IPM motor based on voltage superposition synchronized with carrier wave”, IEEJ Transactions, Vol. 122, No. 1, pp. 37-43 (2002-2) Y. D. Yoon, S .; K. Sul, S.M. Morimoto, and K.M. Ide: “High-Bandwidth Sensorless Algorithm for AC Machines Based on Square-Wave-Type Voltage Injection”, IEEE Trans Ind. Appl. , Vol. 47, no. 3, pp. 1361-1370 (2011-5 / 6) S. Kim, J. et al. I. Ha, and S.H. K. Sul: “PWM Switching Frequency Signal Injection Sensor Method in IPMSM”, IEEE Trans Ind. Appl. , Vol. 48, no. 5, pp. 1576-1587 (2012-9 / 10)

The present invention has been made under the above-mentioned background, and its purpose is to accurately detect the required current polarity through direct detection in a phase velocity estimation based on the rectangular limit high-frequency voltage application method without calculating the required current polarity. Accordingly, an object of the present invention is to provide a digital rotor phase velocity estimation device (phase velocity estimator) having high robustness and stability and high accuracy. In addition, an object of the present invention is to provide a digital rotor phase speed estimation device (phase speed estimator) that can generate an estimated value with fewer operations.

To achieve the above object, the invention according to claim 1 operates with a power converter and a period Ti for an AC motor in which a rotor exhibits salient pole characteristics when a high frequency voltage having a frequency higher than a driving fundamental frequency is applied. This is a digital rotor phase speed estimation device used in a digital drive controller equipped with a stator current discrete time detector, and a γ-axis that aims to synchronize with the rotor phase with zero phase deviation. On the γ-axis of the γδ quasi-synchronous coordinate system composed of orthogonal δ-axes, a rectangular high-frequency voltage of period Th = Nh * Ti is synchronized with the discrete time detection time of the stator current when Nh is a positive even number. A high-frequency voltage applying means adapted to be applied in the form of a high-frequency component extracted from a stator current or a stator current detected in discrete time using the stator current discrete-time detector at a period Ti, On a synchronous coordinate system Difference processing is performed to generate a time difference signal, and the polarity of the γ-axis element of the time difference signal is used as the time difference signal to generate at least one of the γ-axis product signal cγ and the δ-axis product signal sδ, and the generated axial product A positive correlation signal generating means for generating a positive correlation signal pc having a positive correlation with respect to the phase deviation between the rotor phase and the γ-axis phase using the signal, and the positive correlation signal so that the positive correlation signal pc becomes zero A rotor phase that generates a rotor phase by processing pc and generating at least one of a γ-axis phase and an estimated value of the rotor phase or an estimated value of the rotor speed that is basically in a calculus with the rotor phase. And a speed generation means.

According to a second aspect of the present invention, in the digital rotor phase speed estimation device according to the first aspect, the positive / even number Nh is set to two.

A third aspect of the invention is the digital rotor phase speed estimation device according to the first aspect, wherein the AC motor is a synchronous motor.

The following description of the effects and the like of the present invention is based on a permanent magnet synchronous motor having a permanent magnet in the rotor if the rotor is an AC motor having salient pole characteristics with respect to application of a high-frequency voltage having a frequency higher than the drive fundamental frequency. The present invention is applicable to any AC motor such as a wound synchronous motor, a synchronous reluctance motor, a hybrid field synchronous motor, and an induction motor. Embedded magnet type permanent magnet synchronous motors, synchronous reluctance motors, and the like exhibit salient pole characteristics with respect to driving voltage and current. These electric motors similarly exhibit salient pole characteristics even when a high frequency voltage is applied. On the other hand, a surface magnet type permanent magnet synchronous motor and an induction motor that do not show salient pole characteristics with respect to the driving voltage / current show salient pole characteristics when a high frequency voltage is applied. The hybrid field type synchronous motor has characteristics of a permanent magnet type and a wound type synchronous motor, and can exhibit salient pole characteristics with respect to application of a high frequency voltage. In particular, the self-excited hybrid field synchronous motor has a strong saliency.

Consider a γδ quasi-synchronous coordinate system that rotates at a coordinate system speed ωγ specified by the control designer, as shown in FIG. The coordinate system speed ωγ can be one of the estimated speed values of the rotor. The rotation from the main axis (γ axis) to the sub axis (δ axis) is defined as the positive direction. The 2 × 1 vector signals expressing the physical quantities of the AC motor to be handled below are all defined on this coordinate system unless otherwise specified. In the mathematical expression, the 2 × 1 vector signal is expressed using bold characters.

First, the effect of the invention of claim 1 will be described. Consider applying a high-frequency voltage for phase estimation superimposed on a voltage for driving an electric motor. In this case, the stator voltage v1, current i1, and linkage flux φ1 can be expressed as a composite vector of two components as follows.

The symbols f and h of the signal on the right side of the expression (1) indicate that they are components of a driving fundamental frequency and a high frequency, respectively. In particular, the three signals v1h, i1h, and φ1h, which are the second terms of the three formulas (1), are closely related to the present invention due to the applied high-frequency voltage, the high-frequency current as a response, and the applied high-frequency voltage. The high-frequency magnetic flux is shown respectively. It is assumed that the frequency ωh of the high-frequency voltage superimposed and applied for phase estimation is sufficiently high so that the relationship of the following equation (2) is satisfied (the frequency relative ratio is practically 0.01 or less).

以降では、記号ｓは微分演算子ｄ／ｄｔを、Ｉは２ｘ２単位行列を、Ｊは次式で定義された２ｘ２交代行列を意味するものとする。

Hereinafter, the symbol s is the differential operator d / dt, I is the 2 × 2 unit matrix, and J is the 2 × 2 alternating matrix defined by the following equation.

When the equation (2) is established, the following equations (4) to (6) are established for the high-frequency component of the stator in the AC motor in which the rotor exhibits salient pole characteristics with respect to the application of the high-frequency voltage. To do.

なお、鏡相インダクタンスＬｍは、回転子位相として負突極位相を選定する場合には負となる。Here, Li and Lm are the in-phase inductance and mirror phase inductance of the stator, and have the following relationship with the d-axis and q-axis inductances.

The mirror phase inductance Lm is negative when a negative salient pole phase is selected as the rotor phase.

Based on the above preparation, the high frequency voltage applying means which is the first means of the invention of claim 1 will be described.
When the high frequency voltage vγh is applied only on the γ-axis under the condition where the relationship of the expression (2) is established, the following expression is established.

When the high-frequency voltage vγh applied on the γ-axis is rectangular with an amplitude Vh, this voltage can be expressed using a signum function as follows.

矩形状高周波電圧の周期Ｔｈの上限は、電動機の電気的時定数の１／２０程度である。周期Ｔｈの上限より、（１０）式における正偶数Ｎｈの最大値は、おのずと定まることになる。According to the invention of claim 1, the period Th of the rectangular high-frequency voltage in the formula (9) is selected so as to satisfy the following formula with respect to the period Ti that is the stator current discrete time detection period.

The upper limit of the period Th of the rectangular high frequency voltage is about 1/20 of the electrical time constant of the electric motor. From the upper limit of the period Th, the maximum value of positive and even Nh in the equation (10) is naturally determined.

Therefore, the high-frequency current generated by the high-frequency voltage applying means which is the first means of the invention of claim 1 is expressed by the following expression from the expressions (8) and (9).

ここに、電流の脚符ｉはｔ＝ｉ＊Ｔｉでの離散時間検出時刻を意味し、電圧の脚符ｉ−１は時間ｔ＝（ｉ−１）＊Ｔｉ〜ｉ＊Ｔｉの間に印加された電圧を意味する。Next, the positive correlation generating means as the second means of the invention of claim 1 will be described. According to the first aspect of the present invention, the application of the rectangular high-frequency voltage is performed in synchronization with the discrete time detection time (sampling time) of the stator current. That is, the waveform switching time of the rectangular high-frequency voltage is made to coincide with one of the discrete time detection times. In this case, the high-frequency current between the discrete time detection times changes linearly. Therefore, in this case, the differential value of the high-frequency current is accurately replaced with the time difference between the discrete time detection values (sample values). That is, the following equation is obtained from the equations (10) and (11).

Here, the current mark i means the discrete time detection time at t = i * Ti, and the voltage mark i-1 is applied between time t = (i−1) * Ti to i * Ti. Means the applied voltage.

FIG. 2 shows the case of Nh = 2 as an example of the waveform switching time of the rectangular high frequency voltage and the discrete time detection time (sampling time) of the current. (A) of the figure corresponds to the discrete time detection time (sampling time) of the period Ti of the stator current, (b) corresponds to the rectangular high frequency voltage of the period Th = 2Ti, and (c) corresponds to the applied high frequency voltage. Mean high-frequency current (the state of the γ-axis high-frequency current, the same applies to the δ-axis high-frequency current) and the same discrete time detection value (denoted by a circle). In addition, (d) and (e) are two examples of the PWM carrier wave for the power converter used when applying the high frequency voltage. In a drive control apparatus using a power converter using a PWM carrier wave, in order to detect the stator current in a discrete time with high accuracy, the detection time needs to be the peak or valley of the PWM carrier wave or both. As the two examples show, the current discrete time detection time is also synchronized with the PWM carrier. FIG. 3 shows the case of Nh = 4 as an example of the waveform switching time of the rectangular high-frequency voltage and the current discrete time detection time (sampling time). The meanings of the waveform diagrams (a) to (e) are the same as those in FIG.

According to the first aspect of the present invention, the stator current (including the driving fundamental frequency component and the high frequency component) detected for a discrete time with the period Ti or the high frequency component extracted from the stator current (that is, the high frequency current) is expressed as γδ. A time difference process is performed on the quasi-synchronous coordinate system to generate a time difference signal, and the γ-axis product signal cγ and the δ-axis product signal sδ are generated using the polarity of the γ-axis element of the time difference signal as the time difference signal. When the signal subjected to time difference processing is a stator current, and when the signal subjected to time difference processing is a high frequency component (high frequency current) of the stator current, the axial product signal in each case is expressed by the following equation (13a): (13b) Expression is used.

Since the discrete time detection cycle is sufficiently small compared to the cycle of the fundamental frequency component for driving and further sufficiently small compared to the electrical time constant of the motor (see the explanation immediately after the equation (10)), The value of the driving fundamental frequency component of the stator current during the discrete time detection period can be regarded as substantially constant. Therefore, the γ-axis product signal cγ and the δ-axis product signal sδ generated according to the equation (13a) are substantially the same as the γ-axis product signal cγ and the δ-axis product signal sδ generated according to the equation (13b). . This fact can be shown as follows using mathematical formulas.

The characteristics of the γ-axis product signal cγ and the δ-axis product signal sδ generated by the expression (13a) or (13b) are analyzed from the expression (12) as follows:

The expansion of Equation (15) from Equation 2 to Equation 3 is as follows: “The polarity of the actually applied rectangular high-frequency voltage and the γ-axis element of the high-frequency current that is the actual response depends on the phase deviation θγ. It is derived based on the new knowledge that it is always the same regardless. This new knowledge can be proved mathematically based on the equation (12). Briefly, this new knowledge can be confirmed from the waveform diagrams (b) and (c) of FIGS. In the actual system (drive control device), the generation of single-phase high-frequency voltage command value, the delivery of command value to the power converter, the actual generation and application of high-frequency voltage, the generation and detection of the corresponding high-frequency current Timing errors are easily generated in the various processes. Even when there is this timing error, the γ-axis product signal cγ and the δ-axis product signal sδ are obtained according to the present invention specified in the expression (13), as clearly shown in the analytical expression of the expression (15). The characteristic of the third formula of the formula (15) is maintained.

The positive correlation signal generating means according to the first aspect of the present invention has a positive correlation with respect to the phase deviation θγ between the rotor (d axis) and the γ axis using at least one of the γ axis product signal cγ and the δ axis product signal sδ. A positive correlation signal pc is generated. The generation of the positive correlation signal pc according to the present invention is generally expressed by the following equation using the function fp ().

（１７）式の正相関信号ｐｃの正相関特性は、（１５）式より、次のように解析される。

One of the generation of a representative positive correlation signal according to the present invention is expressed by the following equation.

The positive correlation characteristic of the positive correlation signal pc in equation (17) is analyzed as follows from equation (15).

図４の正相関特性から理解されるように、正相関領域での正相関信号は、実効的な位相偏差θγであり、位相偏差相当値と呼ぶべきものである（図１参照）。当然のことながら、正相関信号ｐｃの単位は、位相偏差θγの単位と同一のｒａｄである。FIG. 4 shows positive correlation characteristics with respect to various salient pole ratios rs using the equation (18). From the figure, when the salient pole ratio rs = 0.5 is large, a positive correlation is ensured in the region where the phase deviation θγ is within 1.1 [rad], and the phase is also obtained when the salient pole ratio rs = 0.1. It can be seen that a positive correlation is secured in a region where the deviation θγ is within about 0.8 [rad]. As understood from the figure, in the region where the phase deviation θγ is within 0.5 [rad], the linear relationship is approximately established as in the following equation.

As understood from the positive correlation characteristic of FIG. 4, the positive correlation signal in the positive correlation region is an effective phase deviation θγ and should be called a phase deviation equivalent value (see FIG. 1). As a matter of course, the unit of the positive correlation signal pc is the same rad as the unit of the phase deviation θγ.

Next, the rotor phase speed generating means as the third means of the invention of claim 1 will be described. In the first aspect of the present invention, the positive correlation signal pc is signal-processed so that the positive correlation signal pc becomes zero, and an estimated value of the phase of the γδ quasi-synchronous coordinate system is generated. As understood from the example of FIG. 4, driving the positive correlation signal pc in the positive correlation region to zero drives the phase deviation θγ to zero, and thus causes the γδ quasi-synchronous coordinate system to converge to the dq synchronous coordinate system. This means that (see FIG. 1). The phase equivalent to the phase and the same differential value of the γδ quasi-synchronous coordinate system (γ-axis) corresponds to an estimated value of at least one of the rotor phase and the speed in the AC motor. From the phase of the shaft, a rotor phase estimation value and a rotor speed estimation value can be obtained.

As is apparent from the above description, according to the invention of claim 1, the phase of the γδ quasi-synchronous coordinate system (γ-axis phase) and the estimated value of the rotor phase or the rotor phase basically have a calculus relationship. An effect is obtained that an estimated value of at least one of the estimated values of the rotor speed can be obtained. In the actual system (drive control device), the generation of single-phase high-frequency voltage command value, the delivery of command value to the power converter, the actual generation and application of high-frequency voltage, the generation and detection of the corresponding high-frequency current Timing errors are easily generated in the various processes. Even when there is this timing error, according to the first aspect of the invention, it is possible to obtain an effect that a desired estimated value can be stably obtained. This effect is an effect that cannot be obtained by the method of Patent Document 1 belonging to the same technical field as the present invention. Furthermore, as indicated by equations (13), (16), and (17), there is an effect that these values can be obtained with a very small amount of calculation. In the first aspect of the invention, unlike the conventional Non-Patent Documents 1 to 3, since the high-frequency voltage application and the high-frequency current processing are performed on the same γδ quasi-synchronous coordinate system (see Table 1), consistency is achieved. As a result, the phase and the like of the desired γδ quasi-synchronous coordinate system can be generated with accuracy.

Then, the effect by the invention of Claim 2 is demonstrated. As shown in FIGS. 2 and 3, when the current discrete time detection period Ti is used as a reference, the minimum value of the period Th of the rectangular high-frequency voltage is Th = 2Ti. The invention of claim 2 selects this minimum value. In other words, the highest high frequency is selected. As a result, under the same current discrete time detection period, the effect of being able to obtain the highest speed response regarding estimation can be obtained. As a result, the effect that the effect of Claim 1 can be heightened is acquired.

Next, the effect of the invention of claim 3 will be described. The AC motors targeted by the present invention can be broadly classified into synchronous motors and induction motors. In so-called sensorless driving that does not use a position / speed sensor, a more difficult motor is a synchronous motor. Therefore, according to the invention of claim 3 in which the AC motor is a synchronous motor, the effect of the present invention shown in claim 1 and the effect that the usefulness can be further enhanced can be obtained.

  "Figure showing an example of the relationship between the three coordinate systems and the rotor phase"   “Figure showing one example of the relationship between the discrete-time current detection cycle and the high-frequency voltage cycle”   “Figure showing one example of the relationship between the discrete-time current detection cycle and the high-frequency voltage cycle”   “Figure showing positive correlation characteristics of one positive correlation signal example”   "Block diagram showing drive control system for one embodiment"   “Block diagram showing basic configuration of phase velocity estimator in one embodiment”   “Block diagram showing basic configuration of phase velocity estimator in one embodiment”   “Block diagram showing basic configuration of phase velocity estimator in one embodiment”   “Block diagram showing basic configuration of phase velocity estimator in one embodiment”   "Block diagram showing a basic configuration example of a conventional phase velocity generator"   "Figure showing an example of the relationship between various signals from a conventional phase velocity generator"

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 5 shows an example of a digital drive control device provided with the digital rotor phase speed estimation device of the present invention for a permanent magnet synchronous motor which is a typical AC motor. The general configuration of this drive control device is the same as that of the prior art. The main point of the present invention is a digital rotor phase velocity estimation apparatus. Considering the similarity of the general configuration and the main point of the invention, description of the drive control system and the drive control device is omitted. For these, refer to the explanation already given in the “Background Art” section.

正整数Ｎｓの代表値は、１または２である。特に、正整数をＮｓ＝１と選定する場合には、すなわち、制御周期Ｔｓを固定子電流検出周期Ｔｉと同一とする場合には、デジタル式駆動制御装置と整合性の良い形で（換言するならば、システム的にシンプルな形で）、デジタル式回転子位相速度推定装置を構成することができる。The digital drive control device performs feedback control of the stator current in discrete time (see the description of FIG. 5 described above). The control cycle Ts at this time generally has the following relationship with respect to the stator current detection cycle Ti.

A representative value of the positive integer Ns is 1 or 2. In particular, when a positive integer is selected as Ns = 1, that is, when the control cycle Ts is the same as the stator current detection cycle Ti, in other words, in a form that is compatible with the digital drive control device (in other words, If so, a digital rotor phase speed estimation device can be constructed.

In the present specification, so far, the discrete time of the discrete time signal having the period Ti for phase estimation has been shown using the signal mark i. On the other hand, the discrete time of the discrete time signal of the period Ts for stator current control is shown using the signal mark k. The discrete time of the discrete time signal when selecting a positive integer Ns = 1, that is, Ts = Ti, which enables a simple system configuration, indicates the same time for either the signal leg i or the signal leg k. In this specification, when a positive integer is set to Ns = 1 (most typical case), only the signal mark k is used in order to maintain the simplicity of expression.

The core of the present invention is the phase speed estimator 10 which is substantially synonymous with the digital rotor phase speed estimation device. In any of the speed control and the torque control, the phase speed estimator 10 does not require any change. Further, even when the driven motor is another synchronous motor or induction motor, the phase speed estimator 10 does not require any change. In the following, an embodiment of the phase speed estimator 10 will be described without losing generality regarding control modes such as speed control and torque control, and without losing generality with respect to the AC motor to be driven.

FIG. 6 shows an embodiment of the phase velocity estimator 10 using the present invention. The basic devices constituting the phase velocity estimator 10 are three devices: a rectangular high-frequency voltage command device 10-1, a correlation signal generator 10-2, and a phase synchronizer 10-3. The figure shows an example in which Ts = Ti, that is, Ns = 1. This figure shows the operation in one control cycle from the acquisition of the stator current at time t = k * Ti = k * Ts to time t = (k + 1) * Ti = (k + 1) * Ts. is there. The “reciprocal of z” in the figure means a “delay device” (in terms of operation, “delay operator”) that generates a time delay of one control period Ts with respect to a discrete time signal. In the present specification, the “reciprocal of z” is used in the same way in other places.

The rectangular high-frequency voltage command device 10-1 generates a rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value) on the γδ quasi-synchronous coordinate system with the rotor phase estimation value as a base axis (γ-axis) phase. doing. As shown in FIG. 6, the rectangular high frequency voltage command device 10-1 corresponds to the high frequency voltage application means in the phase velocity estimator, and applies the high frequency voltage application together with the devices 5b, 4b, 2 in the drive control device. Has been carried out. The high-frequency voltage command value generated by the rectangular high-frequency voltage command device is as shown in the equations (9) and (10), and the cycle Th is the minimum positive even number Nh = 2 according to the invention of claim 2. The one corresponding to is good. The generated high-frequency voltage command value is actually applied to the electric motor by the power converter over one control cycle from time t = (k + 1) * Ti = (k + 1) * Ts. The high-frequency voltage command value is superimposed on the driving voltage command value and applied to the synchronous motor via the power converter, so that a high-frequency current flows. The high frequency current is included in this as the high frequency component of the stator current.

FIG. 6 shows a positive correlation signal generating means configured according to the invention of claim 1 as a correlation signal generator 10-2. The correlation signal generator shown in the figure is constructed in accordance with the equations (13a) and (16) exactly. The generation process of the γ-axis product signal cγ and the δ-axis product signal sδ based on this configuration can also be expressed as follows using mathematical expressions.

位相同期器は、相関信号生成器からの正相関信号を入力信号として受け、γδ準同期座標系の位相（本例では、γδ準同期座標系の位相は回転子位相推定値と等価）θα＾と回転子（電気）速度推定値ω２ｎ＾を出力している。回転子（電気）速度推定値ω２ｎ＾は、γδ準同期座標系の速度ωγと実効的に同一である。座標系速度そのもののを回転子速度推定値として利用することもあれば、必要に応じローパスフィルタ処理して回転子速度推定値として利用することもある。図６では、この点を考慮して、速度推定値生成用のローパスフィルタを破線ブロックで示している。一般化積分形ＰＬＬ法に基づく位相同期器に関しては、当業者には周知であるので、これ以上の説明は省略する。The phase synchronizer in FIG. 6 implements a rotor phase speed generation means that is a third means. The phase synchronizer is configured based on the digital generalized integral PLL method shown in Equation (22).

The phase synchronizer receives the positive correlation signal from the correlation signal generator as an input signal, and the phase of the γδ quasi-synchronous coordinate system (in this example, the phase of the γδ quasi-synchronous coordinate system is equivalent to the rotor phase estimation value) θα ^ And the estimated rotor (electrical) speed value ω2n ^. The rotor (electrical) speed estimate ω2n ^ is effectively the same as the speed ωγ of the γδ quasi-synchronous coordinate system. The coordinate system speed itself may be used as a rotor speed estimation value, or may be used as a rotor speed estimation value by low-pass filtering if necessary. In FIG. 6, in consideration of this point, the low-pass filter for generating the speed estimation value is indicated by a broken line block. Since the phase synchronizer based on the generalized integral PLL method is well known to those skilled in the art, further explanation is omitted.

る。また、Ｌｍｔは、リミッタ関数を意味する。リミッタの上下限は、（１９ｂ）式のＫθを参考に±Ｋθ程度に選定しておけばよい。FIG. 7 shows a second configuration example of the phase velocity estimator configured according to the present invention. The difference from the embodiment of FIG. 6 is the correlation signal generator 10-2. The rectangular high-frequency voltage command device and the phase synchronizer are the same as those in the embodiment of FIG. This configuration example of the correlation signal generator is basically based on the equation (17), and is specifically expressed as follows.

The Lmt means a limiter function. The upper and lower limits of the limiter may be selected to be about ± Kθ with reference to Kθ in the equation (19b).

すなわち、δ軸積信号ｓδのリミッタ処理信号を正相関信号ｐｃとしている。なお、本正相関信号ｐｃは、位相偏差θγと次の正相関をもつ。

（２４）式の正相関信号の生成におけるリミッタの上下限は、（２５ｂ）式のＫθを参考に±Ｋθ程度に選定しておけばよい。FIG. 8 is a third configuration example of the phase velocity estimator configured according to the present invention. The difference from the embodiment of FIGS. 6 and 7 resides in the correlation signal generator 10-2. The rectangular high-frequency voltage command device and the phase synchronizer are the same as those in the embodiments of FIGS. This configuration example of the correlation signal generator is based on the following equation (24).

That is, the limiter processing signal of the δ-axis product signal sδ is the positive correlation signal pc. The positive correlation signal pc has the following positive correlation with the phase deviation θγ.

The upper and lower limits of the limiter in generating the positive correlation signal of equation (24) may be selected to be about ± Kθ with reference to Kθ of equation (25b).

すなわち、δ軸積信号ｓδに時間差分信号のγ軸要素絶対値を乗じた上で、リミッタ処理した信号を正相関信号ｐｃとしている。結果的には、（２６）式の第２式が示しているように、時間差分信号のγ軸要素、δ軸要素の積を正相関信号としている。なお、本正相関信号ｐｃは、位相偏差θγと次の正相関をもつ。

（２６）式の正相関信号の生成におけるリミッタの上下限は、（２７ｂ）式のＫθを参考に±Ｋθ程度に選定しておけばよい。FIG. 9 shows a third configuration example of the phase velocity estimator configured according to the present invention. The difference from the embodiment of FIGS. 6 to 8 resides in the correlation signal generator 10-2. The rectangular high-frequency voltage command device and the phase synchronizer are the same as those in the embodiments of FIGS. This configuration example of the correlation signal generator is based on the following equation (26).

That is, the signal obtained by multiplying the δ-axis product signal sδ by the absolute value of the γ-axis element of the time difference signal and then the limiter process is used as the positive correlation signal pc. As a result, the product of the γ-axis element and the δ-axis element of the time difference signal is used as a positive correlation signal as indicated by the second expression of the expression (26). The positive correlation signal pc has the following positive correlation with the phase deviation θγ.

The upper and lower limits of the limiter in the generation of the positive correlation signal of equation (26) may be selected to be approximately ± Kθ with reference to Kθ of equation (27b).

The digital rotor phase speed estimation apparatus (phase speed estimator) according to the present invention has been described in detail with reference to four specific examples. It should be pointed out that the method of generating the positive correlation signal, that is, the method of configuring the correlation signal generator 10-2 is not limited to the above four examples. It should be pointed out that the configuration of the correlation signal generator does not necessarily require a limiter process and can be removed in the above four examples. It should be pointed out that the phase synchronizer 10-3 also has various configurations other than the introduction example.

6 to 9, the input signal of the correlation signal is a stator current (including a driving fundamental wave component and a high frequency component), but the input signal of the correlation signal is a high frequency component of the stator current, That is, it should be pointed out that a high-frequency current or an equivalent value may be used. In other words, it should be pointed out that the correlation signal generator may be configured according to equations (13b) and (16). In this case, the high-frequency current is extracted from the stator current detected in discrete time. It should be pointed out that there are various extraction methods such as a method of digitally filtering the stator current and a method of subtracting the current command value of the driving fundamental wave component from the stator current.

The embodiment using FIGS. 6 to 9 is an embodiment in the case where Ts = Ti and Ns = 1, in which the control cycle Ts is the same as the stator current detection cycle Ti in the equation (20). It should be pointed out that even when Ns = 2, a similar embodiment, that is, a similar phase velocity estimator configuration is possible. The person skilled in the art knows how to change this (Non-Patent Document 1 and Non-Patent Document 3 show a signal processing method in the case of Ns = 2), and further explanation is omitted.

For the convenience of specific description of the digital rotor phase speed estimation device, the synchronous motor is used as the driving motor and the drive control device related thereto is taken up. However, the digital rotor phase speed estimation device according to the present invention is a synchronous motor. It is pointed out repeatedly that it is not limited to. The digital rotor phase speed estimation device in the drive control device using another AC motor as the drive motor can be used as it is in the specific embodiment described in detail. Regarding the general arrangement of a drive phase control device and a drive control device in which the drive motor is a synchronous reluctance motor or an induction motor, refer to, for example, Patent Literature (Shinji Shinnaka: “Vector control method of AC motor” And the same device ", Japanese Patent No. 4120775 (1902-3-18)) and the like. Therefore, this description is omitted.

The present invention is suitable for applications that require phase estimation performance with high responsiveness in a speed range including zero speed, among applications in which an AC motor is sensorlessly driven.

1 AC motor (synchronous motor)
DESCRIPTION OF SYMBOLS 2 Power converter 3 Discrete time current detector 4a 3 phase 2 phase converter 4b 2 phase 3 phase converter 5a Vector rotator 5b Vector rotator 6 Current controller 7 Command converter 8 Speed controller 9 Low pass filter 10 Phase speed Estimator 10-1 Rectangular high frequency voltage command device 10-2 Correlation signal generator 10-3 Phase synchronizer 11 Coefficient unit 12 Cosine sine signal generator

Claims (3)

Digital drive with power converter and stator current discrete time detector operating at period Ti for AC motors whose rotor exhibits salient pole characteristics for application of high frequency voltage higher than the drive fundamental frequency A digital rotor phase speed estimation device used in a control device,
On the γ-axis of the γδ quasi-synchronous coordinate system composed of the γ-axis aiming for synchronization with the rotor phase with zero phase deviation and the δ-axis orthogonal thereto, the cycle Th = Nh * Ti A high frequency voltage applying means adapted to apply the rectangular high frequency voltage in synchronization with the discrete time detection time of the stator current;
A time difference signal is obtained by performing time difference processing on the γδ quasi-synchronous coordinate system on a stator current or a high frequency component extracted from the stator current detected at a period Ti using the stator current discrete time detector. And generating at least one of the γ-axis product signal cγ and the δ-axis product signal sδ using the polarity of the γ-axis element of the time difference signal as the time difference signal, and using the generated axis product signal, the rotor phase And a positive correlation signal generating means for generating a positive correlation signal pc having a positive correlation with respect to the phase deviation of the γ-axis phase;
The positive correlation signal pc is subjected to signal processing so that the positive correlation signal pc becomes zero, and the γ-axis phase and the estimated value of the rotor phase or the estimation of the rotor speed basically having a calculus relationship with the rotor phase. A rotor phase velocity generation means for generating an estimated value of at least one of the values;
A digital rotor phase speed estimation apparatus comprising: 2. The digital rotor phase speed estimation apparatus according to claim 1, wherein the positive and even Nh is set to 2. 2. The digital rotor phase speed estimation apparatus according to claim 1, wherein the AC motor is a synchronous motor.

Images