JP4924115B2 - Permanent magnet synchronous motor drive control device - Google Patents

Description

The present invention relates to a drive control device for a permanent magnet synchronous motor, and in particular, to secure the phase of a vector rotator or a vector rotator built-in converter (hereinafter abbreviated as a vector rotator) necessary for drive control. The present invention relates to a sensorless drive control device using a phase determiner. The phase determiner implements vector rotator phase determination means and is also called a phase estimator. The phase determiner which is the subject of the present invention is to estimate the phase for a vector rotator and the like through control of the power factor phase of the terminal power of the permanent magnet synchronous motor.

In order to achieve high control performance by using a permanent magnet synchronous motor, a stator current that contributes to torque generation is set to a predetermined phase difference with respect to the phase of the rotor N pole (hereinafter abbreviated as rotor phase). It is necessary to have. For this reason, it is necessary to know the rotor phase evaluated on the fixed αβ coordinate system in which the α axis is selected as the center of the u-phase winding. The simplest method for detecting the rotor phase is to mount a position sensor such as an encoder or resolver on the rotor. However, mounting the position sensor is problematic in terms of reliability, cost, etc. The rotor phase is estimated by using the stator voltage and current signals.

The estimation of the rotor phase can be performed on a fixed αβ coordinate system or on a quasi-synchronized γδ coordinate system that aims to synchronize with the rotor phase without any phase difference. Note that a coordinate system synchronized with the rotor phase without a phase difference is recently called a synchronized dq coordinate system. The quasi-synchronous γδ coordinate system is substantially equivalent to the synchronous dq coordinate system when synchronization is completed. Considering this point, hereinafter, these including the quasi-synchronous γδ coordinate system and the strict synchronous dq coordinate system will be referred to as a synchronous dq coordinate system in a broad sense. On the other hand, a γδ coordinate system that does not aim at “synchronization without phase difference” with respect to the rotor phase is called a control γδ coordinate system.

FIG. 9 is a block diagram schematically showing a typical example of a drive control device using a phase determiner for a permanent magnet synchronous motor. 1 is a permanent magnet synchronous motor, 2 is a phase determiner, 3 is a power converter, 4 is a current detector, 5a and 5b are 3 phase 2 phase converters, 2 phase 3 phase converters, 6a , 6b are vector rotators, 7 is a cosine sine signal generator, 8 is a current controller, 9 is a command converter, 10 is a speed controller, and 11 is a machine speed estimator. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

In particular, the phase determiner 2 receives an estimated value of the rotor phase evaluated on the fixed αβ coordinate system by obtaining an equivalent value (measured value, command value, etc.) of the stator voltage and current as an input, and the rotor. The estimated electric speed is output. The cosine sine signal generator 7 converts the rotor phase estimation value into a cosine / sine signal and transmits it to the vector rotators 6a and 6b. The electric speed estimated value is divided by the number of pole pairs in the machine speed estimator 11 and converted into a machine speed estimated value. 5, 5a, 5b, 6a, 6b, 7, and 8 capture the stator current contributing to torque generation as a vector signal on the synchronous dq coordinate system, and use the d-axis and q-axis components. A feedback-type current control means is configured to perform control so as to follow each axis current command value. The phase determiner 2 constitutes a vector rotator phase determining means.

The three-phase stator current detected by the current detector 4 is converted into a two-phase current on the fixed αβ coordinate system by the three-phase two-phase converter 5a, and then the two-phase current in the synchronous dq coordinate system by the vector rotator 6a. It is converted into a phase current and sent to the current controller 8. The current controller 8 generates a two-phase voltage command value on the synchronous dq coordinate system so that the two-phase current on the synchronous dq coordinate system follows the current command value of each phase, and sends it to the vector rotator 6b. In 6b, the two-phase voltage command value on the synchronous dq coordinate system is converted into the two-phase voltage command value on the fixed αβ coordinate system and sent to the two-phase three-phase converter 5b. In 5 b, the two-phase signal is converted into a three-phase voltage command value and output as a command value to the power converter 3. The power converter 3 generates electric power according to the command value, applies it to the synchronous motor 1, and drives it. The two-phase current command value on the synchronous dq coordinate system at this time is obtained by converting the torque command value through the command converter 9. In FIG. 9, dq is added to the foot mark to clearly indicate that the stator voltage and current signals present on the left side of the vector rotator are signals on the synchronous dq coordinate system. The vector rotators 6a and 6b may be configured integrally with the three-phase two-phase converter and the two-phase three-phase converters 5a and 5b, respectively. In the present invention, these integrated components are referred to as a vector rotator built-in converter.

In this example of FIG. 9, since the example which comprised the speed control system is shown, the torque command value is obtained as an output of the speed controller 10 which inputs the speed command value and the speed estimated value. As is well known to those skilled in the art, the speed controller 10 and the machine speed estimator 11 are unnecessary when the control purpose is the generated torque and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

Among the above-described components, devices particularly related to the present invention are the phase determiner 2 and the command converter 9. The phase determiner estimates the rotor phase using the equivalent values of the stator voltage and current, and determines the phase to be used for the vector rotator or the like. In addition, the estimated electric speed is determined. The estimation of the rotor phase is performed by first estimating the rotor magnetic flux or the induced voltage and then extracting the rotor phase information contained therein. For estimating the rotor magnetic flux or the induced voltage, a state observer, a disturbance observer, or the like that uses the equivalent values of the stator voltage and current are widely used.

The electric motor is not only a torque generator but also a mechanical energy converter of electric energy, and the energy conversion efficiency of the electric motor is one of the main performances of the electric motor. The issue of improving energy conversion efficiency is, from the viewpoint of motor control, how to implement current control that minimizes losses such as copper loss while achieving the required torque generation, or maximize power factor It can be replaced with a technical problem of how to carry out current control to achieve the above. The command converter 9 plays the role of solving this problem. The command converter generates d-axis and q-axis current commands on the synchronous dq coordinate system that enables high-efficiency operation from the torque command. Examples of the prior invention related to the command converter include the following.

(1) Osawa, Nomura: “Improvement of device utilization efficiency of reverse salient-pole PM motor”, Proc. 348-349
(2) Shinnaka Shinji: “Vector Control Method of Synchronous Motor”, JP-A-11-041998
(3) Shinnaka Shinji: “Analysis by vector signal of salient pole type synchronous motor for current control with emphasis on efficiency”, IEEJ Transactions D, 119, 5, pp. 648-658 (1999-5)

For generating d-axis and q-axis current commands on a synchronous dq coordinate system that achieves high efficiency from a torque command, generally, a torque generation equation describing the relationship between the d-axis current, q-axis current, and generated torque is used. And a method for solving nonlinear simultaneous equations by a orbital expression that mathematically describes a trajectory that achieves high efficiency. As is well known to those skilled in the art, this solution is generally very difficult and attempts have been made to improve the command converter to accomplish this more or less efficiently. However, as long as high efficiency is pursued in the drive control device of the conventional permanent magnet synchronous motor, this command converter is indispensable and cannot be eliminated.

Problems to be solved by the invention

The present invention has been made under the above background, and its object is to provide a drive control device that enables high-efficiency operation without requiring a command converter in sensorless drive control of a permanent magnet synchronous motor. Another object of the present invention is to provide a sensorless drive control device capable of high efficiency operation with a lighter calculation load.

Means for solving the problem

In order to achieve the above object, the invention of claim 1 is a fixed circuit in which a stator current contributing to torque generation is adjusted to follow a current command value using a vector rotator or a converter with a built-in vector rotator. A permanent magnet synchronous motor having current control means for feedback or feedforward control via a child voltage and vector rotator phase determination means for determining a phase to be used for a vector rotator or a converter with a built-in vector rotator A drive control device, wherein the vector rotator phase determining means detects a power factor phase equivalent value of the terminal power of the permanent magnet synchronous motor, and a power factor phase that generates a power factor phase command value Controls the phase used for the vector rotator or the converter with built-in vector rotator so that the command means and the power factor phase equivalent value match the power factor phase command. Power factor phase control means for forming, characterized in that it comprises a.

According to the second aspect of the present invention, the stator current that contributes to the torque generation is fed back via the stator voltage adjusted to follow the current command value by using the vector rotator or the converter with built-in vector rotator. A drive control device for a permanent magnet synchronous motor having current control means for feedforward control, and vector rotator phase determination means for determining a phase used for a vector rotator or a vector rotator built-in converter, The vector rotator phase determining means detects the power factor phase equivalent value of the terminal power of the permanent magnet synchronous motor, and the vector rotator or vector rotator so that the power factor phase equivalent value becomes zero. Power factor phase control means for controlling and generating the phase used for the built-in converter using only fixed parameters.

A third aspect of the present invention is the permanent magnet synchronous motor drive control apparatus according to the first or second aspect of the present invention, which is on the control γδ coordinate system specified by the vector rotator or the vector rotator built-in converter. The power factor phase detecting means is configured to detect the power factor phase equivalent value using the voltage equivalent value.

A fourth aspect of the present invention is the drive control apparatus for the permanent magnet synchronous motor according to the first aspect, wherein the power factor phase command value is obtained by utilizing at least the stator current equivalent value and the rotor speed equivalent value. The power factor phase command means is configured to generate the power factor.

The invention according to claim 5 is the drive control device for the permanent magnet synchronous motor according to claim 1 or 2, wherein the force is controlled so that an integration processing step is involved in a final step of the control generation of the phase. The rate phase control means is constructed.

Next, the operation of the present invention will be described using mathematical expressions describing the electromagnetic relationship of the synchronous motor. As shown in FIG. 1, a control γδ coordinate system that rotates at a speed ω is considered. Further, it is assumed that the rotor N pole of the permanent magnet synchronous motor has a phase θ _r at a certain moment with respect to the γ axis of the main shaft. At this time, it is known that the electromagnetic relationship of the permanent magnet synchronous motor is described by the following equations (1) to (8) using signals evaluated on the control γδ coordinate system.

ω_２ｎは回転子の電気速度であり、Ｒ_ｌは固定子巻線の抵抗である。Ｌ_ｉ，Ｌ_ｍは固定子の同相インダクタンス、鏡相インダクタンスであり、ｄ、ｑインダクタンスとは次の関係を有する。Here, the 2 × 1 vectors ν _l , i _l , and φ _l mean the stator voltage, current, and magnetic flux, respectively. 2 × 1 vectors φ _i and φ _m indicate components constituting the stator magnetic flux φ _l , φ _i is a magnetic flux induced and generated by the stator current i _l , and φ _m is a rotation caused by the rotor permanent magnet. It is a child magnetic flux, and e is an induced voltage. I is a 2 × 2 unit matrix, and J is a 2 × 2 alternating matrix defined by the following equation.

ω _2n is the electrical speed of the rotor and R _l is the resistance of the stator winding. L _i and L _m are the common-phase inductance and mirror-phase inductance of the stator, and have the following relationship with d and q inductances.

また、Ｄ（ｓ，ω）は次式で定義されたＤ因子であり、

ｓは微分演算子ｄ／ｄｔである。

D (s, ω) is a D factor defined by the following equation:

s is a differential operator d / dt.

The electro-magnetic relationship of the equations (1) to (8) is a relationship on the synchronous dq coordinate system when θ _r = 0 is always set. Similarly, in the case where θ _r = θ _α is always set, the relationship is in the fixed αβ coordinate system. However, the signals in the equations (1) to (8) at this time need to be signals evaluated on the related coordinate system.

上式の固定子電流の脚符ｄｑは、電流のｄ軸、ｑ軸成分を意味している。また、最小銅損の状態でトルク発生を行なうには、固定子電流は、同期ｄｑ座標系上で評価した場合、次の（１２）式の最小銅損軌道上に存在しなければならない。

図２に、ある同期電動機のパラメータを利用して、力率１軌道と最小銅損軌道の１例を、参考までに描画した。In order to efficiently drive and control the synchronous motor, the stator current must exist on a specific track. For example, in order to generate torque with a power factor of 1, the stator current evaluated on the synchronous dq coordinate system must be on the trajectory described by the following equation.

The stator current dq in the above equation means the d-axis and q-axis components of the current. Further, in order to generate torque in a state of minimum copper loss, the stator current must exist on the minimum copper loss trajectory of the following equation (12) when evaluated on the synchronous dq coordinate system.

In FIG. 2, an example of a power factor 1 trajectory and a minimum copper loss trajectory is drawn for reference using parameters of a certain synchronous motor.

The power factor phase referred to in the present invention means the phase of the stator voltage as viewed from the stator current. FIG. 3 shows the power factor phase θ _iv defined by the present invention. Also, in the same figure, the stator current is assumed to exist on the δ axis of the control γδ coordinate system in consideration of the convenience of the following explanation. In other words, the phase of the stator current is the δ-axis phase.

In general, when a stator current is present on a specific track, the synchronous motor has a characteristic that the power factor phase of the terminal power of the synchronous motor takes a specific value. For reference, using a certain motor parameter, the stator current (displayed as δ-axis current) is accurately placed on the minimum copper loss track in the power running state in Fig. 4 (a) to achieve the minimum copper loss state. An example of the power factor phase with respect to the stator current in this case was drawn for the cases of electrical speed ω _2n = 3, 30, 150, 300, 600 (rad / s).
If this characteristic regarding the stator current trajectory and the power factor phase is utilized, the stator current can be present on the trajectory that achieves high-efficiency driving through the control of the power factor phase.

According to the first aspect of the present invention, the vector rotator phase determining means includes a power factor phase detecting means for detecting a power factor phase equivalent value of the terminal power of the permanent magnet synchronous motor, and a power factor for generating a power factor phase command value. Phase control means and power factor phase control means for controlling and generating the phase used for the vector rotator or the like so that the power factor phase equivalent value matches the power factor phase instruction. Therefore, according to the first aspect of the present invention, the phase used for the vector rotator or the like necessary for high-efficiency driving can be obtained simply by generating a power factor phase command value for achieving high-efficiency driving in the power factor phase command means. If it can be determined, an effect is obtained. In other words, by solving the nonlinear simultaneous equations by the torque generation equation describing the relationship between the d-axis current, the q-axis current and the generated torque, and the orbit equation mathematically describing the orbit that achieves high efficiency. In other words, it is possible to easily determine the current phase for high-efficiency driving.

Next, the operation of claim 2 of the present invention will be described. When the stator current is always present on the power factor 1 orbit, the power factor phase command corresponding to this is always zero. Therefore, when the stator current is always present on the power factor 1 orbit, the power factor phase command means is not necessarily required (see FIG. 6 described later). Therefore, according to the invention of claim 2, the efficiency drive is limited to the power factor 1 drive, but the same effect as that of the invention of claim 1 can be obtained. In other words, by solving the nonlinear simultaneous equations by the torque generation equation describing the relationship between the d-axis current, the q-axis current and the generated torque, and the orbit equation mathematically describing the orbit that achieves high efficiency. In other words, an effect is obtained that the current phase for driving the power factor of 1 can be easily determined.

The power factor phase control means adjusts the phase so that the power factor phase equivalent value detected by the power factor phase detection means matches the power factor phase command value (when the power factor is 1, the power factor phase is zero). It performs the function of generating control. As a configuration method of the power factor phase control means for performing this function, a method using a variable parameter and a method using a fixed parameter can be considered. In general, when a fixed parameter can be used, the vector rotator phase determining means can determine a more stable phase by using a fixed parameter. This general theory is also applied to the vector rotator phase determination means based on the invention of claim 2. According to the invention of claim 2, since the power factor phase control means, which is one of the main components of the vector rotator phase determination means, is configured using only fixed parameters, more stable power factor phase control is possible. The effect that it becomes possible is also obtained.

（１４）式の関係は、固定αβ座標系上、制御γδ座標系上のいずれの座標系上の固定子電圧、電流を用いても有効であり、また、同一の算定結果を得ることができる。ここで、図３のように、固定電流の位相をδ軸位相とする制御γδ座標系を考える。本制御γδ座標系上では、固定子電流のγ軸成分成分は常時ゼロであるので、（１４）式は、次の（１５）式のように簡略化される。

Then, the effect | action of Claim 3 of this invention is demonstrated. The power factor phase can generally be detected according to the following relationship using a stator voltage and a stator current.

The relationship of the equation (14) is effective even when the stator voltage and current on any coordinate system on the fixed αβ coordinate system and the control γδ coordinate system are used, and the same calculation result can be obtained. . Here, as shown in FIG. 3, a control γδ coordinate system in which the phase of the fixed current is the δ-axis phase is considered. On the present control γδ coordinate system, the γ-axis component of the stator current is always zero, so equation (14) is simplified as the following equation (15).

The identity of equation (15) is also confirmed from FIG. According to the invention of claim 3, the power factor phase detecting means is configured to detect the power factor phase equivalent value using the voltage equivalent value on the control γδ coordinate system specified by the vector rotator or the like. Become. Therefore, the relational expression (15) on the control γδ coordinate system can be utilized. As a result, according to the third aspect of the invention, an effect is obtained that the power factor phase equivalent value can be easily detected. The voltage equivalent value referred to in the present invention means a detected value of the stator voltage, the approximate value, the command value, and the like. If the approximate value of the stator voltage and the command value are good approximations of the detected value, these may be treated substantially as detected values, as is well known to those skilled in the art. Similarly, the power factor phase equivalent value referred to in the present invention means a power factor phase detection value, an approximate value, a power factor phase proportional value, and the like detected based on the equation (14).

Then, the effect | action of Claim 4 of this invention is demonstrated. The example of the power factor phase shown in FIG. 4A is an example of the power factor phase when the stator current (δ-axis current) exists on the minimum copper loss orbit. This figure means that if the stator current and the rotor speed are specified, the power factor phase corresponding to the characteristics of the synchronous motor is inevitably specified. In other words, the figure means that in order to place the stator current on a specific trajectory, it is generally necessary to specify the power factor phase in consideration of the fixed current and the rotor speed. . The few exceptions are when the stator current is present on a power factor orbit. The power factor phase corresponding to the power factor 1 is zero as described in connection with the operation of the invention of claim 2.

According to the invention of claim 4, the power factor phase command means is configured to generate the power factor phase command value by using at least the stator current equivalent value and the rotor speed equivalent value. Therefore, according to the power factor phase command means of the present invention, according to the stator current and the rotor speed, the power factor phase command value for arranging the stator current on the orbit that minimizes the loss is rationally set. The effect that it can be generated is obtained. The current equivalent value referred to in the present invention means a current detection value, an approximate value, a current command value, and the like. Further, the rotor speed equivalent value means a rotor speed estimated value, a rotor speed command value, and the like. As will be readily understood by those skilled in the art, the current detection value is replaced with the command value when appropriate current control is performed. Further, the true value of the rotor speed is replaced with the same estimated value when properly estimated, and is replaced with the same command value when properly controlled.

Then, the effect | action of Claim 5 of this invention is demonstrated. The derivative of phase is speed. Therefore, if an integration process is included in the final process of phase control generation, the integration process signal becomes a differential value of the phase determined to be controlled, that is, a speed. Since the phase at this time is a phase used for a vector rotator or the like, the speed that is the differential value is the speed of the control γδ coordinate system specified by the vector rotator or the like. In the drive control of the synchronous motor, the rotor speed and the speed of the control γδ coordinate system are the same. Therefore, the integrated signal is a rotor speed estimate. According to the fifth aspect of the present invention, the power factor phase control means is configured so as to accompany the integration process in the final process of the phase control generation, so that the rotor speed estimation value can be easily generated from the non-integration process signal. The effect | action that it becomes will be obtained.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 5 shows a basic structure of one embodiment in which the drive control device of the present invention is applied to a permanent magnet synchronous motor. The difference between this apparatus and the conventional drive control apparatus shown in FIG. The second difference is that there is no command converter, the output of the speed controller 10 directly becomes the δ-axis current command value, and the γ-axis current command value is always set to zero. . In FIG. 5, the stator voltage and current signals present on the left side of the vector rotator are signals on the control γδ coordinate system. In order to clarify this fact, γ and δ are added to the foot marks of these signals. On the other hand, in FIG. 9, the stator voltage and current signals present on the left side of the vector rotator are signals on the synchronous dq coordinate system. Special attention should be paid to this difference. Regarding other components, there is no difference between the conventional drive control device and this device.

The core of the present invention that does not require a command converter is the phase determiner 2 that implements a vector rotator phase determining means. FIG. 6 is an example embodiment of a phase determiner constructed in accordance with the present invention. According to the invention of claim 1, the phase determiner 2 realizes a power factor phase detector 2a that realizes a power factor phase detecting means, a power factor phase commander 2b that realizes a power factor phase command means, and a power factor phase control means. Power factor phase controller 2c. In particular, the phase determiner 2 shown as the embodiment in FIG. 6 has a configuration according to the invention of claim 2 as will be apparent from the following description when the power factor commander 2b is removed. I am doing so.

当然のことながら、固定子電圧指令値に代わって、固定子電圧検出値、同近似値を利用してもよい。固定子電圧指令値を利用した（１６）式に従い得た力率位相相当値は、力率位相の近似値となっている。In principle, the power factor phase detector 2a is configured according to the equation (15) based on the invention of claim 3. However, as the voltage equivalent value, a stator voltage command value is used instead of the stator voltage detection value. That is, the power factor phase detector 2 in this embodiment is configured according to the following equation.

Of course, instead of the stator voltage command value, the detected stator voltage value and the approximate value may be used. The power factor phase equivalent value obtained according to equation (16) using the stator voltage command value is an approximate value of the power factor phase.

ている。請求項４の発明に従った、固定子電流相当値、回転子速度相当値を利用した力率位相指令器の構成は、例えば、次のものを考えることができる。

を、Ｌｍｔ（・）はリミッタ処理を意味する。なお、トルク制御には（１７）式第１式あるいは第２式を、速度制御には（１７）式第３式を使用するようにすればよい。図６の実施形態例では、（１７）式第３式に従い、力率位相指令器２ｂを構成している。本例では、電流相当値として電流検出値を、回転子速度相当値として回転子速度指令値を利用した例ともなっている。

ing. The configuration of the power factor phase commander using the stator current equivalent value and the rotor speed equivalent value according to the invention of claim 4 can be considered as follows.

, Lmt (·) means limiter processing. It should be noted that the first or second equation (17) may be used for torque control, and the third equation (17) may be used for speed control. In the embodiment of FIG. 6, the power factor phase commander 2 b is configured according to the third equation (17). In this example, the current detection value is used as the current equivalent value, and the rotor speed command value is used as the rotor speed equivalent value.

（ｒａｄ）は、力率位相として大きめな値である。The design parameters and limiter design guidelines when the power factor phase commander 2b is configured based on the equation (17) are as follows.

(Rad) is a large value as the power factor phase.

描画したものである。

図４（ａ）、（ｂ）の比較より、本例の力率位相指令値は、最小銅損軌道に対応した力率位相

言う性質を考慮するならば、本例からも、（１８）式の設計パラメータを（１７）式に適用して力率位相指令を生成することにより、最小銅損制御に準じた最適な電流制御を遂行できることが確認される。FIG. 4B shows the power factor phase command value based on the third equation of the equation (17) when the design parameter is the following equation (19) according to the design guideline of the equation (18). Use the same motor parameters as

It is drawn.

From the comparison between FIGS. 4A and 4B, the power factor phase command value of this example is the power factor phase corresponding to the minimum copper loss trajectory.

In consideration of the property, the optimum current control according to the minimum copper loss control can be obtained from this example by applying the design parameter of the equation (18) to the equation (17) to generate the power factor phase command. Is confirmed to be able to carry out.

(18) as best seen in formula, the design parameter K _2, it is also possible to adopt a zero. In this case, as is apparent from the equation (17), this means that zero is commanded as the power factor phase command value. As is clear from the configuration of the phase determiner 2 shown in FIG. 6, the zero power factor phase command value is completely equivalent to the removal of the power factor phase commander 2b. The phase determiner 2 of FIG. 6 has a configuration based on the invention of claim 2 when the power factor phase commander 2b is removed. As a matter of course, in this configuration in which the power factor phase command is always equal to zero, drive control with a power factor of 1 is always performed.

The power factor phase controller 2c is configured according to the invention of claim 5. Specifically, as clearly shown in FIG. 6, it is composed of a phase integrator 2c-1, a frequency controller 2c-2, a fundamental frequency gain 2c-3, and the like. These can also be expressed using mathematical formulas as follows.

According to the invention of claim 5, prior to determining the phase of the control γδ coordinate system, first, the velocity ω of the coordinate system is generated. After the coordinate system speed ω is generated, an integration process is performed on the coordinate system speed ω to determine the phase θ used for the vector rotator or the like. As is clear from FIG. 6 or the equations (20) to (24), the integration process is performed in the final step of the phase control generation. FIG. 6 also shows the manner in which the coordinate speed is processed by the low-pass filter 2c-4 to generate an estimated electric speed value of the rotor. As will be readily understood by those skilled in the art, a vector rotator or the like From the viewpoint of generation of the phase used for the integration process, the integration process is the final process.

In this embodiment, the coordinate system speed ω is synthesized as a simple sum of the fundamental frequency ω ₁ and the correction frequency Δω, as clearly shown in the equation (21). The fundamental frequency is generated in a form proportional to the δ-axis voltage equivalent value. In this embodiment, a δ-axis voltage command is used as the δ-axis voltage equivalent value. K _l in the equation (21) is a constant fundamental frequency gain to be designed by the designer. A temporary guide for designing the basic frequency gain is shown at the right end of the same equation.

（２４）式に示しているようにｍ次有理多項式として構成されるが、有理多項式のパラメータは固定であり、これら固定のパラメータは次の（２５）式の（ｍ＋１）次Ｈ（・）がフルビッツ多項式となるように設計されている。

周波数制御器が上記条件を満足するように設計される場合には、力率位相偏差はゼロに収

請求項１及び請求項２の発明における力率位相制御手段の目的が達成される。The correction frequency Δω added to the fundamental frequency is the phase deviation between the power factor phase equivalent value and the command.

As shown in the equation (24), it is configured as an m-order rational polynomial, but the parameters of the rational polynomial are fixed, and these fixed parameters are the (m + 1) -order H (•) of the following equation (25). It is designed to be a Hurwitz polynomial.

If the frequency controller is designed to satisfy the above conditions, the power factor phase deviation will be zero.

The object of the power factor phase control means in the inventions of claims 1 and 2 is achieved.

本目的のローパスフィルタとしては、通常は、次の固定パラメータをもつ１次でよい。

The purpose of the low-pass filter 2c-4 is to easily obtain the estimated electric speed value of the rotor from the coordinate system speed as described in connection with the operation of the invention of claim 5. That is,

The low-pass filter for this purpose is usually a first order having the following fixed parameters.

In the frequency controller 2c-2, the fundamental frequency gain 2c-3, and the low-pass filter 2c-4 constituting the power factor phase controller 2c, parameters whose design is left to the designer are used. As already explained, it should be pointed out that all these parameters are fixed. This meets the requirements of the invention of claim 2.

FIG. 5 shows an embodiment of the speed control mode, but it is also possible to perform a quasi-torque control mode according to the torque control mode. In the quasi-torque control mode, the δ-axis current command may be directly given from the outside. In this case, the power factor phase commander may be configured based on the second equation (17). In order to be based on Equation (2), the electrical speed estimation value of the rotor is necessary. As already described, this is the low-pass filter 2c-4 in the power factor phase controller 2c. Has been generated by. The estimated value of the rotor electrical speed by the low-pass filter may be used for the power factor phase commander.

In the embodiment shown in FIG. 5, the current control is performed in a feedback manner. The present invention can also be applied to the case where the current control is performed in a feedforward manner, and the same operation and effect as in the case of the feedback current control can be obtained.

Although the phase determiner according to the present invention can be implemented in an analog manner, it is preferably constructed digitally in view of the remarkable progress of recent digital technology. Although the digital configuration includes a hardware configuration and a software configuration, as will be apparent to those skilled in the art, any of the present invention can be configured.

Since an experiment was conducted to confirm the characteristics, performance, operation, and effect of the drive control apparatus according to the present invention, an example of the result is presented. The experimental system was configured according to FIG. The state of the test motor and the load device is shown in FIG. The test motor is a 400 (W) salient pole permanent magnet synchronous motor (the left end in FIG. 7). The phase determiner used in the experiment was the same as in FIG. The main design parameters were determined as follows in accordance with the proposed design guidelines.

２次のフルビッツ多項式がｓ＝−７５で安定二重根を持つように２個のパラメータを設計した。回転子速度推定用のローパスフィルタ２ｃ−４は１次とし、その帯域は２０（ｒａｄ／ｓ）とした。電流制御ループは、制御周期１２５（μｓ）を考慮の上、帯域２０００（ｒａｄ／ｓ）が得られるように設計し、速度制御ループは、供試電動機の約５３倍にも及ぶ負荷装置の巨大な慣性モーメントを考慮し、線形速度応答が確保される概ね上限帯域である帯域２（ｒａｄ／ｓ）が得られるように設計した。図５に示した駆動制御システムにおいて、３相２相変換器から２相３相変換器に至るすべての機器は、単一のＤＳＰで離散時間的に実現した。The upper and lower limits of the limiter in the power factor phase commander were also set to ± 0.4 (rad) in accordance with the proposed design guidelines. In addition, the frequency controller 2c-2 is the following primary:

Two parameters were designed so that the second-order Hurwitz polynomial had a stable double root with s = −75. The low-pass filter 2c-4 for estimating the rotor speed is primary, and its band is 20 (rad / s). The current control loop is designed to obtain a band 2000 (rad / s) in consideration of the control cycle 125 (μs), and the speed control loop is a huge load device that is about 53 times as large as the test motor. In consideration of a large moment of inertia, a band 2 (rad / s), which is a substantially upper limit band in which a linear velocity response is ensured, was obtained. In the drive control system shown in FIG. 5, all devices from the three-phase two-phase converter to the two-phase three-phase converter are realized in a discrete time by a single DSP.

子位相の符号反転値−θ_ｒ＝θ−θ_α、ｕ相電流を示している（図１参照）。γ軸から評価した回転子位相θ_ｒはγ軸とｄ軸の位相差であるが、図８では、安定性の高いｄ軸を基準にとり（すなわち符号反転して）、これを表示している。An example of the experimental result when the rated power running load is applied is shown in FIG. In FIGS. 4A, 4B, and 2C, the rated speed of 180 (rad / s) is taken into consideration, and the responses at the machine speeds of 180, 18, and 9 (rad / s) are used. (A) of FIG. 6 shows from the top the γδ coordinate system phase (γ-axis phase) θ evaluated from the α-axis and the circuit evaluated from the α-axis.

The sign inversion value of the child phase −θ _r = θ−θ _α , and the u-phase current are shown (see FIG. 1). The rotor phase θ _r evaluated from the γ-axis is the phase difference between the γ-axis and the d-axis, but in FIG. 8, this is displayed with reference to the highly stable d-axis (ie, the sign is inverted). .

が指示されていることが分かる。力率位相相当値θ_ｉνは同指令値に従い制御され、位相偏

同図（ａ）の合理的な応答は、本発明による所期の特性、性能、作用、効果を裏付けるものである。From FIG. 8A, the power factor phase command value is a value as designed as follows.

It is understood that is indicated. The power factor phase equivalent value θ _iv is controlled according to the command value, and the phase deviation

The reasonable response shown in FIG. 5A supports the expected characteristics, performance, operation and effect of the present invention.

Even for deceleration from the rated speed, the expected characteristics, performance, action and effect were confirmed as shown in FIG. 18 (b) of 18 (rad / s). However, from around 9 (rad / s) corresponding to 1/20 of the rated speed (see (c) in the same figure), the disturbance of the u-phase current waveform and the disturbance of the γδ coordinate system phase that seems to be induced by this are visible. I came out to the extent possible. Note that the disturbance of the u-phase current waveform is caused by the influence of the inverter dead time, as can be understood from the occurrence of the zero-crossing point of the v-phase w-phase current.

The present invention has been described in detail with reference to various embodiments using various drawings. In addition, an actual machine experiment was conducted to verify and confirm the expected characteristics, performance, action, and effect.

The invention's effect

As is clear from the above description, the present invention has the following effects. According to the first aspect of the present invention, the phase to be used for the vector rotator or the like necessary for the high-efficiency drive is determined simply by generating the power factor phase command value for achieving the high-efficiency drive in the power factor phase command means. When it was possible, the effect was obtained. In other words, by solving the nonlinear simultaneous equations by the torque generation equation describing the relationship between the d-axis current, the q-axis current and the generated torque, and the orbit equation mathematically describing the orbit that achieves high efficiency. Thus, it was possible to easily determine the current phase for high-efficiency driving. As a result of this operation, according to the first aspect of the present invention, an effect is obtained that a high-efficiency operation of the permanent magnet synchronous motor becomes possible without using a command converter. As a result, it is possible to obtain an effect that high-efficiency operation is possible with a light calculation load.

Next, the effect of the present invention of claim 2 will be described. According to the invention of claim 2, the efficiency drive is limited to the power factor 1 drive, but the same effect as that of the invention of claim 1 can be obtained. As a result of this operation, according to the second aspect of the present invention, an effect is obtained that power factor 1 driving is possible without using a command converter. In particular, the invention of claim 2 differs from the invention of claim 1 in that the power rotator phase detection means is not required in the configuration of the vector rotator phase determination means, and an efficient power factor 1 operation is possible with a lighter calculation load. The effect of becoming is obtained. According to the invention of claim 2, since the power factor phase control means, which is one of the main components of the vector rotator phase determination means, is configured using only fixed parameters, a more stable power factor phase. The effect that control becomes possible was also obtained. As a result of this action, there is also an effect that the efficient operation can be performed more stably.

Next, the effect of the present invention of claim 3 will be described. According to the invention of claim 3, the effect that the power factor phase equivalent value can be easily detected is obtained. As a result of this action, an effect is obtained that the present invention having the effects of claims 1 and 2 can be realized more easily.

Next, the effect of the present invention of claim 4 will be described. According to the third aspect of the present invention, it is possible to rationally generate a power factor phase command value for arranging the stator current on the orbit that minimizes the loss according to the stator current and the rotor speed. The effect was obtained. As a result of this action, according to the invention of claim 3, the effect that “the effect of enabling high-efficiency operation” according to claim 1 can be obtained reasonably is obtained.

Next, the effect of the invention of claim 5 will be described. According to the fifth aspect of the present invention, there has been obtained an effect that the rotor speed estimation value can be generated by a simple process using a low-pass filter. As is well known to those skilled in the art, when speed control is performed, generation of a speed estimate is indispensable. Therefore, according to the fifth aspect of the invention, it is possible to obtain an effect that the speed control can be easily performed only by adding a simple process using a low-pass filter. As a result of this effect, the effect that the effects and usefulness of claims 1 and 2 can be further enhanced can be obtained.

The above effects were verified through actual machine experiments.

The figure which shows the example of 1 relationship of 3 coordinate systems and a rotor phase Diagram showing examples of minimum copper loss trajectory and power factor 1 trajectory Diagram showing examples of power factor phase between stator current and stator voltage on the δ-axis Diagram showing examples of power factor phase and power factor phase command for stator current and rotor speed 1 is a block diagram of a drive control system including a drive control device according to an embodiment. The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment Figure showing an overview of the actual experimental system Figure showing examples of response results of actual machine experiments A block diagram of a drive control system including a conventional representative drive control device

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Permanent magnet synchronous motor 2 Phase determiner 2a Power factor phase detector 2b Power factor phase commander 2c Power factor phase controller 2c-1 Phase integrator 2c-2 Frequency controller 2c-3 Fundamental frequency gain 2c-4 Low pass filter DESCRIPTION OF SYMBOLS 3 Power converter 4 Current detector 5a 3 phase 2 phase converter 5b 2 phase 3 phase converter 6a Vector rotator 6b Vector rotator 7 Cosine sine signal generator 8 Current controller 9 Command converter 10 Speed controller 11 Machine Speed estimator

Claims (5)

Current that controls the stator current that contributes to torque generation in a feedback or feed-forward manner via a stator voltage that is adjusted to follow the current command value using a vector rotator or a converter with a built-in vector rotator. A permanent magnet synchronous motor drive control device comprising control means and vector rotator phase determining means for determining a phase to be used for a vector rotator or a vector rotator built-in converter,
The vector rotator phase determining means includes a power factor phase detecting means for detecting a power factor phase equivalent value of the terminal power of the permanent magnet synchronous motor, a power factor phase command means for generating a power factor phase command value, and a power factor phase. And a power factor phase control means for controlling and generating a phase to be used for the vector rotator or the vector rotator built-in converter so that the equivalent value matches the power factor phase command value. Drive control device. Current that controls the stator current that contributes to torque generation in a feedback or feed-forward manner via a stator voltage that is adjusted to follow the current command value using a vector rotator or a converter with a built-in vector rotator. A permanent magnet synchronous motor drive control device comprising control means and vector rotator phase determining means for determining a phase to be used for a vector rotator or a vector rotator built-in converter,
The vector rotator phase determining means is a power factor phase detecting means for detecting the power factor phase equivalent value of the terminal power of the permanent magnet synchronous motor, and the vector rotator or the vector rotation so that the power factor phase equivalent value is zero. A drive control apparatus for a permanent magnet synchronous motor, comprising: a power factor phase control means for controlling and generating a phase to be used for a built-in converter using only fixed parameters. The power factor phase detecting means is configured to detect the power factor phase equivalent value using the voltage equivalent value on the control γδ coordinate system designated by the vector rotator or the vector rotator built-in converter. The drive control apparatus for a permanent magnet synchronous motor according to claim 1 or 2. 2. The permanent magnet synchronous motor according to claim 1, wherein the power factor phase command means is configured to generate a power factor phase command value using at least a stator current equivalent value and a rotor speed equivalent value. Drive control device. The drive control apparatus for a permanent magnet synchronous motor according to claim 1 or 2 , wherein the power factor phase control means is configured so as to be accompanied by an integration process step in the final step of the phase control generation.

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