JP6311105B2 - AC motor drive control device - Google Patents

Description

The present invention is a drive control device for an AC motor (permanent magnet synchronous motor, synchronous reluctance motor, induction motor, etc.), and in particular, a vector rotator or a vector rotator built-in converter (hereinafter referred to as a vector rotator) required for drive control. The present invention relates to a sensorless drive control apparatus using a phase determiner for securing the phase of a vector rotator or the like. The phase determiner implements vector rotator phase determination means and is also called a phase estimator.

Consider a case where the stator current treated as a 2 × 1 vector on the “γδ rotating coordinate system” composed of two orthogonal axes of the γ axis and the δ axis exists on the δ axis or the γ axis (see FIG. 2). . The “γδ rotating coordinate system” in which the phase of the stator current and the phase of the coordinate system coincide with each other as in this case is particularly referred to as “γδ current coordinate system”. Further, consider a case where a stator voltage treated as a 2 × 1 vector exists on the δ axis or the γ axis on the γδ rotating coordinate system composed of two orthogonal axes of the γ axis and the δ axis (see FIG. 4). . The “γδ rotating coordinate system” in which the phase of the stator voltage and the phase of the coordinate system coincide with each other as in this case is particularly referred to as “γδ voltage coordinate system”.

In an AC motor, the generated torque is determined by the stator current. In particular, in AC motors such as permanent magnet synchronous motors, synchronous reluctance motors, and induction motors, there are an infinite number of currents that generate the same torque, and particularly important control is minimum copper loss control. In sensorless vector control of a permanent magnet synchronous motor which is a typical AC motor, a method of directly estimating the stator current phase is known so that the stator current exists on the minimum / quasi-minimum loss locus. This representative method is called "power factor phase vector control method" (see Patent Documents (1), (2), (3), Non-Patent Documents (1), (2)). The power factor phase vector control method directly determines and controls the phase of the γδ rotating coordinate system through the control of the “power factor phase”, which is the phase between the stator current and the stator voltage. is there.

The power factor phase vector control method is as follows: “The stator current is captured as a 2 × 1 vector, and the γ-axis current command value on the γδ rotation coordinate system specified by the vector rotator or the converter with built-in vector rotator, δ-axis current command "Feedback current control means for feedback controlling the stator current so as to follow the value", and "capturing the stator voltage as a 2 × 1 vector on the γδ rotational coordinate system, between the stator current and the stator voltage A vector rotator phase determining means for determining a rotator phase used for a vector rotator or a vector rotator built-in converter so that an equivalent value of a power factor phase that is a phase follows a specified value. Have As a simple configuration method of the power factor phase vector control method having this feature, there are a method of configuring on a γδ current coordinate system (Patent Document (1), Non-Patent Document (1)), and a γδ voltage coordinate system. (Patent Documents (2) and (3), Non-Patent Document (2)).

The conventional power factor phase vector control method introduced above does not take into account the existence of a voltage limit of the power converter (inverter). For this reason, the desired efficient drive control is brought about in the speed region below the rated speed, but it cannot be used in the speed region exceeding the rated speed. On the other hand, the “improved power factor phase vector control method” that can be used in a speed range exceeding the rated speed is widely demanded by the industry. The functions and performance required for the improved power factor phase vector control method are:
“(1) In the speed range below the rated speed where there is no effective voltage limit, the performance is equivalent to the conventional power factor phase vector control method”, “(2) The voltage limit exists effectively. In the speed range above the rated speed, it exhibits performance with the same effect as the flux-weakening control so that it can be used under voltage limitation, making it possible to drive the best or quasi-best efficiency under voltage limitation. "That's it.

(1) Shinnaka Shinji: “Driving control device for permanent magnet synchronous motor”, JP-A-2008-199868 (2007-2-10) (2) Shinnaka Shinji: “Driving control device for permanent magnet synchronous motor”, JP 2011-15601 (2011-1-20) (3) Onishi Norio, Kenji Yamanaka, Eiji Goda: “Power Conversion Control Device, Power Conversion Control Method, and Program for Power Conversion Control”, Japanese Patent No. 40222630 (2006-6) -27)

(1) Shinnaka Shinji: “Vector control technology of permanent magnet synchronous motor, second volume (the essence of sensorless drive control)”, Denpa Shimbun pp. 247-273 (2008-12) (2) Shinnaka Shinji: “Power factor phase vector control method for simplified and efficient driving of sensorless PMSM (simplification and conversion to voltage coordinate system)”, IEEJ Transactions D, Vol. 130, no. 2, pp. 215-227 (2010-2)

The present invention has been made under the above-mentioned background, and its object is to provide the above-mentioned (1), (2) as a sensorless drive control device for an AC motor (permanent magnet synchronous motor, synchronous reluctance motor, induction motor, etc.). It is an object of the present invention to newly provide an apparatus based on an improved trajectory-directed vector control method having the function and performance of a term.

In order to achieve the above object, the invention of claim 1 captures the stator current as a 2 × 1 vector, and the γ-axis current command value on the γδ rotating coordinate system specified by the vector rotator or the converter with built-in vector rotator. , Feedback current control means for feedback controlling the stator current so as to follow the δ-axis current command value, and the stator voltage is captured as a 2 × 1 vector on the γδ rotation coordinate system, and the stator current and the stator voltage A vector rotator phase determining means for determining a rotator phase to be used for a vector rotator or a vector rotator built-in converter so that an equivalent value of a power factor phase that is a phase between the vector rotator follows a specified value; A sensorless drive control apparatus for an AC motor, wherein the vector rotator phase determination means includes a voltage limit detection means for detecting whether or not a voltage equivalent value has reached a voltage limit value, and a limit The power factor phase equivalent command value is exponentially within a limited range in the direction corresponding to the weak magnetic flux according to the detection of the arrival of the value, and in the direction corresponding to the anti-weakening magnetic flux according to the detection of the limit value not reached. Index command generating means for automatically adjusting and generating, power factor phase detecting means for detecting the power factor phase equivalent value of the terminal power of the motor, and a rotator so that the detected power factor phase equivalent value follows the power factor phase equivalent command value Phase control means for controlling / determining the phase.

The invention of claim 2 captures the stator current as a 2 × 1 vector and follows the γ-axis current command value and δ-axis current command value on the γδ rotation coordinate system specified by the vector rotator or the converter with built-in vector rotator. As described above, the feedback current control means for feedback control of the stator current and the stator voltage are captured as a 2 × 1 vector on the γδ rotation coordinate system so that the stator voltage exists on the δ axis or the γ axis. A vector rotator phase determining means for determining a phase to be used for the vector rotator or the vector rotator built-in converter, and a sensorless drive control device for an AC motor, wherein the feedback current control means has a voltage equivalent value. Voltage limit detection means for detecting whether or not the voltage limit value has been reached, in the direction corresponding to the weak magnetic flux in response to detection of reaching the limit value, and the limit value not yet reached Based on the generated power factor phase equivalent command value, exponential command generation means for automatically adjusting and generating the power factor phase equivalent command value within a limited range in the direction corresponding to the anti-weakening magnetic flux according to the detection And a current command generating means comprising three means, a power factor phase type current command synthesizing means for synthesizing at least one of the γ-axis current command value and the δ-axis current command value.

The “voltage equivalent value” in the present invention is a general term for signals having a correlation with the true voltage value applied to the AC motor. As a typical voltage equivalent value, there are a measured voltage value and a voltage command value. Further, the “power factor phase equivalent value” in the present invention is a general term for signals having a correlation with the power factor phase. Typical power factor phase equivalent values include the power factor phase itself and the tangent value of the power factor phase. The definition of the power factor phase equivalent command value is based on the definition of the power factor phase.

The effect of the invention of claim 1 will be described. In order to ensure the simplicity of explanation, here, a permanent magnet synchronous motor is taken up as a drive object as a typical AC motor, and the effect of the invention will be described. First, the coordinate system will be described. Consider FIG. FIG. 1 shows an αβ fixed coordinate system, a dq synchronous coordinate system, and a γδ rotating coordinate system. The αβ fixed coordinate system is a coordinate system corresponding to the stator, and generally the α axis is taken at the center of the stator u-phase winding. The d axis of the dq synchronous coordinate system is a coordinate system synchronized with the rotor magnetic flux. That is, in the dq synchronous coordinate system, the phase of the d axis is the same as the phase of the rotor magnetic flux. The speed of the coordinate system is the same as the rotor speed ω2n. The γδ rotating coordinate system is a kind of rotating coordinate system, and its speed is ωγ. The phase of the γ axis viewed from the α axis is θαγ, and the phase of the d axis viewed from the γ axis is θγ.

When the drive control apparatus is configured using the invention of claim 1, the most convenient coordinate system is a γδ current coordinate system which is a kind of γδ rotating coordinate system. FIG. 2 schematically shows an example of the γδ current coordinate system. In this example, the phase of the stator current i1 as a 2 × 1 vector matches the phase of the δ axis. In other words, the γ-axis current is always zero. A γδ current coordinate system is easily realized by constantly controlling the γ-axis current to zero. The phase of the stator voltage v1 as a 2 × 1 vector viewed from the stator current as a 2 × 1 vector is θiv.

FIG. 3 is a block diagram schematically showing a typical example of a drive control device using a phase determiner on a γδ current coordinate system for an AC motor (permanent magnet synchronous motor). 1 is an AC motor (permanent magnet synchronous motor), 2 is a phase determiner, 3 is a power converter, 4 is a current detector, 5a and 5b are 3-phase 2-phase converter and 2-phase 3-phase conversion, respectively. 6a and 6b are vector rotators, 7 is a cosine sine signal generator, 8 is a current controller, 10 is a speed controller, and 11 is a mechanical speed estimator. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

In particular, the phase determiner 2 includes an equivalent value (command value, estimated value, etc.) of the rotor speed and a voltage limit value cv in addition to the equivalent value (actual value, command value, etc.) of the stator voltage and current. It is obtained as an input and outputs the phase θαγ ^ of the current coordinate system and the estimated value of the rotor electrical speed ω2n (the symbol ^ means the estimated value of the corresponding signal). The voltage limit value cv input to the phase determiner is generally time-varying since the bus voltage (also referred to as link voltage) of the power converter 3 varies. The cosine sine signal generator 7 converts the coordinate system phase estimation value into a cosine / sine signal and transmits it to the vector rotators 6a and 6b. The electrical speed estimated value is divided by the pole pair number Np in the mechanical speed estimator 11 and converted into a mechanical speed estimated value. 5, 5a, 5b, 6a, 6b, 7, and 8 capture the stator current as a vector signal on the γδ current coordinate system, and each component of the γ-axis and δ-axis each current command value The feedback current control means is configured to control so as to follow. The phase determiner 2 constitutes a vector rotator phase determining means.

The three-phase stator current detected by the current detector 4 is converted into a two-phase current on the αβ fixed coordinate system by the three-phase two-phase converter 5a, and then on the γδ current coordinate system by the vector rotator 6a. It is converted into a two-phase current and sent to the current controller 8. The current controller 8 generates a two-phase voltage command value on the γδ current coordinate system so that the two-phase current on the γδ current coordinate system follows the current command value of each phase, and sends it to the vector rotator 6b. In 6b, the two-phase voltage command value on the γδ current coordinate system is converted into the two-phase voltage command value on the αβ fixed coordinate system and sent to the two-phase three-phase converter 5b. In 5 b, the two-phase voltage command value is converted into a three-phase voltage command value and output as a command value to the power converter 3. The power converter 3 generates power corresponding to the command value, applies it to the AC motor (permanent magnet synchronous motor) 1 and drives it. The power converter 3 outputs a voltage limit value cv depending on the bus voltage to the phase determiner 2 when power is generated.

The γ-axis current command value constituting the two-phase current command value on the γδ current coordinate system is always set to zero. Under the situation where the current control is exhibiting the desired performance, this zero setting realizes the γδ current coordinate system. The δ-axis current command value is obtained as an output signal of the speed controller. In FIG. 3, γδ is added to the foot mark to clearly indicate that the stator voltage and current signals present on the left side of the vector rotator are signals on the γδ current coordinate system. Similarly, αβ is added to the foot mark to clearly indicate that the stator voltage and current signals present on the right side of the vector rotator are signals on the αβ fixed coordinate system. The vector rotators 6a and 6b may be configured integrally with the three-phase two-phase converter and the two-phase three-phase converters 5a and 5b, respectively. In the present invention, these integrated components are referred to as a vector rotator built-in converter.

3 shows an example in which a speed control system is configured, and thus a δ-axis current command value iδ * is obtained as an output of the speed controller 10 that receives the speed command value and the speed estimation value ( In the present invention, a head value * is added to clearly indicate the command value of the corresponding signal). The phase determiner obtains a speed command value as a speed equivalent value. As is well known to those skilled in the art, the speed controller 10 and the machine speed estimator 11 are unnecessary when the control purpose is current control and the speed control system is not configured. In this case, the δ-axis current command value is directly applied from the outside. In this case, as the speed equivalent value used for the phase determiner, a speed estimated value generated by the phase determiner itself is used.

上式においては、電圧ｖと電流ｉの脚符γ、δは、γ軸要素、δ軸要素を意味している。Ｒ１は巻線抵抗を、Φは回転子磁束強度を、Ｌｉ、Ｌｍは同相、鏡相インダクタンスを示している（詳しい説明は、公知の非特許文献（１）などを参照）。これより、γ軸とｄ軸との位相θγと力率位相θｉｖに関して、次の（２）式の関係が成立していることが確認される。

The vector rotator phase determination means according to the first aspect of the present invention is realized as the phase determiner 2 in the configuration example of FIG. The core of the invention of claim 1 resides in the phase determiner 2. This effect will be described. On the γδ current coordinate system shown in FIG. 2, the relationship between the voltage and current of the stator of the permanent magnet synchronous motor is expressed by the following equation (1).

In the above formula, the symbols γ and δ of the voltage v and the current i mean the γ-axis element and the δ-axis element. R1 represents winding resistance, Φ represents rotor magnetic flux intensity, and Li and Lm represent in-phase and mirror phase inductances (for details, see known non-patent document (1)). From this, it is confirmed that the relationship of the following equation (2) is established regarding the phase θγ and the power factor phase θiv between the γ-axis and the d-axis.

一方、実効的に電圧制限がある場合での高速回転には、「弱め磁束制御」が必須である。最大の弱め磁束制御は、γ軸とｄ軸との位相θγがθγ＝−π／２（正回転、力行を想定）となる。本条件を、（２）式に用いると次の（４）式の関係を得る。

The most basic motor drive under conditions where there is no effective voltage limit is to match the γ and δ axes. This driving means that the phase θγ = 0. For the sake of simplicity, assuming the normal rotation / power running, and using this condition in the equation (2), the relationship of the following equation (3) is obtained.

On the other hand, “weakening magnetic flux control” is essential for high-speed rotation when voltage is effectively limited. In the maximum flux-weakening control, the phase θγ between the γ-axis and the d-axis is θγ = −π / 2 (assuming normal rotation and power running). When this condition is used in equation (2), the following relationship of equation (4) is obtained.

Equations (3) and (4) mean the following four points. {Circle around (1)} If the power factor phase can be controlled according to the voltage limit situation, the drive control of the motor can be performed in a wide range of speeds that require the maximum weakening magnetic flux. (2) The power factor phase is controlled in the direction corresponding to the weak magnetic flux in response to detection of reaching the limit value, and in the direction corresponding to the anti-weak magnetic flux in response to detection of the limit value not reached. There is a need to. (3) In the case of forward rotation and power running, the power factor phase is controlled in the direction in which the stator current is advanced in phase with respect to the stator voltage in response to detection of reaching the limit value, and the limit value is not set. The stator current needs to be in a phase lag direction with respect to the stator voltage in response to detection of arrival. The power factor phase and its tangent value take a wide range from positive to negative. (4) The power factor phase and its tangent value take values in the range from positive to negative. However, the range is limited to the limited range indicated by the equations (3) and (4) at the maximum.

According to the first aspect of the present invention, the vector rotator phase determining means (that is, the phase determiner 2) detects whether or not the voltage equivalent value has reached the voltage limit value, and the limit value reached. The power factor phase equivalent command value is automatically adjusted exponentially within a limited range in the direction corresponding to the weak magnetic flux according to the detection and in the direction corresponding to the anti-weak magnetic flux according to the detection of the limit value not reached. Control / determine the rotor phase so that the power factor phase equivalent value follows the power factor phase equivalent command value, the exponent command generation means to generate, the power factor phase detection means to detect the power factor phase equivalent value of the terminal power of the motor Phase control means. As a result, one end of the “limited range” is set to a value that provides efficient driving when there is no effective voltage limit, and the other end is set to a value corresponding to the expected maximum speed. "Effective driving performance is exhibited in the speed range below the rated speed where there is no effective voltage limit", "(2) In the speed range above the rated speed where there is an effective voltage limit, it can be used under the voltage limit. In this case, an effect equivalent to that of the flux-weakening control is exhibited. In this case, the best or quasi-best efficiency drive under the voltage limit is made possible. In addition, since this adjustment is performed exponentially (in other words, stepwise), it is possible to obtain an effect that the automatic adjustment can be easily stabilized.

Next, the effect of the invention of claim 2 will be described. In order to ensure the simplicity of explanation, a permanent magnet synchronous motor is taken up as a driving object as a typical AC motor here, and the effect of the invention will be described. The invention of claim 2 is particularly characterized in that signal processing including feedback current control is performed on the γδ voltage coordinate system. FIG. 4 schematically shows an example of the γδ voltage coordinate system. In this example, the phase of the stator voltage v1 as a 2 × 1 vector matches the phase of the δ axis. In other words, the γ-axis voltage is always zero. The phase of the stator voltage v1 as a 2 × 1 vector viewed from the stator current i1 as a 2 × 1 vector is θiv. The relative relationship between the stator voltage and the stator current in FIG. 4 is the same as the relative relationship between the stator voltage and the stator current in FIG.

FIG. 5 is a block diagram schematically showing a typical example of a drive control device using a phase determiner 2 on a γδ voltage coordinate system for an AC motor (permanent magnet synchronous motor). Major changes in FIG. 5 with respect to FIG. 3 are two points: a phase determiner 2 and a γ-axis current command generator 9. Since other devices are the same, description of these devices is omitted. The phase determiner of FIG. 5 implements a vector rotator phase determining means. Its role is to generate a phase θαγ ^ for creating a γδ voltage coordinate system. This phase determiner is basically the same as that presented in Patent Document (2) and Non-Patent Document (2). For this reason, further explanation is omitted. In addition to the δ-axis current equivalent value (actual value, command value, etc.) and the rotor speed equivalent value (command value, estimated value, etc.), the γ-axis current command generator 9 generates a stator voltage equivalent value (actual value, command value). Voltage limit value cv as an input, and a γ-axis current command value iγ * is output. The voltage limit value cv obtained from the power converter 3 is generally time-varying because the bus voltage of the power converter 3 varies.

フィードバック電流制御を介し（５）式の関係を達成すべく、たとえば、力率位相相当指令値としてθｉｖ＊の正接値を利用してγ軸電流指令値を以下のように生成する。

力率位相の調整が、非電圧制限下の効率駆動、電圧制限下の弱め磁束と同等の作用を発揮する点は、電流座標系、電圧座標系のいずれにおいても同一である（図２、図４参照）。すなわち、この作用的特性に関しては、座標系の選定如何を問わない。As understood from FIG. 1, on the γδ voltage coordinate system, the following relationship is established between the γ-axis current i _γ and the δ-axis current i _δ .

In order to achieve the relationship of the equation (5) through feedback current control, for example, a γ-axis current command value is generated as follows using a tangent value of θiv * as a command factor corresponding to the power factor phase.

The fact that the adjustment of the power factor phase exhibits the same effect as the efficiency driving under the non-voltage limit and the weak magnetic flux under the voltage limit is the same in both the current coordinate system and the voltage coordinate system (FIG. 2, FIG. 4). That is, regarding this operational characteristic, it does not matter whether the coordinate system is selected.

Therefore, if the γ-axis current command value is generated according to the equation (6), the same effect as that of the invention of claim 1 can be obtained. As a matter of course, the power factor phase equivalent command value (tan θiv * in this example) in the equation (6) at this time is a limited range as described in relation to the equations (3) to (4). So it will be adjusted exponentially.

According to the invention of claim 2, the feedback current control means includes the current command generation means. The current command generation means at this time includes a voltage limit detection means for detecting whether or not the voltage equivalent value has reached the voltage limit value, a direction corresponding to the weak magnetic flux in response to detection of the limit value arrival, and a limit Exponential command generation means for automatically adjusting and generating a power factor phase equivalent command value within a limited range in the direction corresponding to the anti-weakening magnetic flux in response to detection of unreachable value, and the generated power factor phase equivalent Based on the command value, there are provided three means: a power factor phase type current command synthesis means for synthesizing at least one of the γ-axis current command value and the δ-axis current command value. Consequently, as in the first aspect of the invention, one end of the “limited range” of the power factor phase equivalent command value is set to a value that provides efficient driving when there is no effective voltage limit, and the other end is expected. Simply setting the value corresponding to the maximum speed, “(1) Demonstrate efficient drive performance in the speed range below the rated speed where there is no effective voltage limit”, “(2) Effective voltage limit. In the speed range above the existing rated speed, it exhibits the performance with the same effect as the flux-weakening control so that it can be used under the voltage limit, making it possible to drive the best or quasi-best efficiency under the voltage limit. Effect. In addition, since this adjustment is performed exponentially (in other words, stepwise), it is possible to obtain an effect that the automatic adjustment can be easily stabilized.

  The figure which shows the phase relationship example of 3 coordinate system and a rotor   The figure which shows the example of a relationship of (gamma) delta current coordinate system and dq synchronous coordinate system.   1 is a block diagram of a drive control system including a drive control device according to an embodiment.   The figure which shows the example of a relationship of (gamma) delta voltage coordinate system and dq synchronous coordinate system.   1 is a block diagram of a drive control system including a drive control device according to an embodiment.   The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment   1 is a block diagram showing a basic configuration of a γ-axis current command generator in an embodiment.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

An embodiment of the phase determiner 2 that is the core of the drive control system of FIG. 3 will be described. FIG. 6 shows an embodiment of a phase determiner constructed in accordance with the first aspect of the present invention. According to the invention of claim 1, the phase determiner 2 includes a voltage limit detector 2a that realizes a voltage limit detector, a power factor phase commander 2b that realizes an exponent command generator, and a power factor that realizes a power factor phase detector It comprises a phase detector 2c and a phase controller 2d that implements phase control means. The phase determiner 2 places importance on practicality and draws the signal using discrete time representation, assuming that the signal is processed in discrete time. Hereinafter, when the control cycle is Ts, the signal at time t = kTs is simply expressed using (k).

The voltage limit detector 2a detects whether or not the voltage equivalent value has reached the voltage limit value. In the example of FIG. 6, the voltage command value on the γδ current coordinate system and the voltage limit value cv are compared, and 0 is detected when the voltage limit value has not been reached, and 1 is detected when the voltage limit value has not been reached, as in the following equation. The signal u is output.

One implementation example of the power factor phase commander 2b is described by the following equation (8).

いる（８ｄ）式は、指数変化の信号ｘを用いて２個の両端値を加重平均している。総合的には、力率位相指令器２ｂは、制限値到達の検出に応じて弱め磁束に対応する方向に、また、制限値未到達の検出に応じて反弱め磁束に対応する方向に、力率位相相当指令値を限定的範囲内で指数的に自動調整し生成している。Equation (8a) means that the signal x changes exponentially. Α1 in the equation is a design parameter for designating the degree of exponent change, and the selection is left to the designer. The selection range is as specified at the right end of the same formula. The expressions (8b) and (8c) are basically obtained from the expressions (3) and (4) and indicate the two end values of the power factor phase equivalent command. In the equation (8b), iδ ^~ means the rated current of the electric motor. ω2n and ω2m are the electric speed and mechanical speed of the rotor, and the symbol ^ and the symbol * mean the estimated value and command value of the corresponding signal. Np is the number of pole pairs, and Ld and Lq are d-axis and q-axis inductances. Lmt means a limiter function. K2 and K3 are design parameters for which selection is entrusted to the designer, and the selection range is specified in the equation (8b).

In the equation (8d), the two end values are weighted and averaged using the exponent change signal x. Overall, the power factor phase commander 2b moves in the direction corresponding to the weak magnetic flux in response to detection of reaching the limit value, and in the direction corresponding to the anti-weakening magnetic flux in response to detection of the limit value not reached. The rate phase equivalent command value is automatically adjusted and generated exponentially within a limited range.

本例では、電圧実測値に代わって、電圧指令値の相対比の極性反転値を力率位相相当値として利用している。もちろん、電圧検出器のコストが問題ならない場合には、電圧実測値を利用してよい。The power factor phase detector 2c detects the power factor phase equivalent value according to the following equation.

In this example, instead of the actually measured voltage value, the polarity reversal value of the relative ratio of the voltage command value is used as the power factor phase equivalent value. Of course, when the cost of the voltage detector is not a problem, the measured voltage value may be used.

（１０ｅ）式のＦｌ（ｚ−１）は、ローパスフィルタである。（１０）式は、ＰＬＬの一種であり、力率位相相当値が力率位相相当指令値に従うように回転器位相を決定・制御する役割を遂行している。本ＰＬＬは、特許文献（１）、非特許文献（１）にも利用されて、これらにおいて詳しく説明されているので、これ以上の説明は省略する。The phase controller 2d is configured as follows.

Fl (z-1) in the equation (10e) is a low-pass filter. Equation (10) is a kind of PLL and performs the role of determining and controlling the rotator phase so that the power factor phase equivalent value follows the power factor phase equivalent command value. Since this PLL is also used in Patent Document (1) and Non-Patent Document (1) and has been described in detail there, further description is omitted.

（１１ａ）式は、信号ｘは指数的に変化することを意味する。同式のα１は、指数変化の度合いを指定するための設計パラメータであり、その選定は設計者に委ねられている。選定範囲は、同式右端に明示している通りである。（１１ｂ）式は、基本的に、（４）式から得られて

値である。（１１ｃ）式は、指数変化の信号ｘを用いて２個の両端値（他の端値はＫ２）を加重平均している。（１１ｄ）式は、力率位相相当指令の最終生成を担っている。総合的には、力率位相指令器２ｂは、制限値到達の検出に応じて弱め磁束に対応する方向に、また、制限値未到達の検出に応じて反弱め磁束に対応する方向に、力率位相相当指令値を限定的範囲内で指数的に自動調整し生成している。With respect to the phase determiner 2 of FIG. 6, a second embodiment of the power factor phase commander 2b is shown. Considering that the value of the equation (8b) is approximately K2 · iδ at the time of high speed rotation, the following equation (11) can be obtained as an example of the power factor phase commander 2b.

Equation (11a) means that the signal x changes exponentially. Α1 in the equation is a design parameter for designating the degree of exponent change, and the selection is left to the designer. The selection range is as specified at the right end of the same formula. The expression (11b) is basically obtained from the expression (4).

Value. In the equation (11c), two end values (the other end value is K2) are weighted and averaged using the exponent change signal x. Equation (11d) is responsible for the final generation of the power factor phase equivalent command. Overall, the power factor phase commander 2b moves in the direction corresponding to the weak magnetic flux in response to detection of reaching the limit value, and in the direction corresponding to the anti-weakening magnetic flux in response to detection of the limit value not reached. The rate phase equivalent command value is automatically adjusted and generated exponentially within a limited range.

The core of the drive control system of FIG. 5 is in the γ-axis current command generator (current command generation means) 9. Details of the embodiment of the γ-axis current command generator 9 will be described. FIG. 7 shows an embodiment of a γ-axis current command generator (current command generation means) 9 configured according to the invention of claim 2. According to the invention of claim 2, the γ-axis current command generator 9, which is a current command generation means, includes a voltage limit detector 9a that realizes a voltage limit detection means, a power factor phase command device 9b that realizes an exponent command generation means, It comprises a multiplier 9c that implements a rate phase current command combining means. The γ-axis current command generator 9 emphasizes practicality, and signals are processed in discrete time.

The voltage limit detector 9a detects whether or not the voltage equivalent value has reached the voltage limit value. In the example of FIG. 7, the voltage command value on the γδ voltage coordinate system is compared with the voltage limit value cv, and as shown in the equation (7), 0 is reached when the voltage limit value is not reached, and 1 is reached when the voltage limit value is reached. Is output as a detection signal u.
Since the expression (7) has already been described in detail, further description is omitted. One example of realization of the power factor phase commander 9b is of the formula (8). Since the formula (8) has already been described in detail, further description will be omitted. As already described in detail with respect to the equation (8), the power factor phase commander 9b responds to detection of reaching the limit value in the direction corresponding to the weak magnetic flux, and in response to detection of the limit value not reached. The power factor phase equivalent command value is automatically adjusted exponentially within a limited range in the direction corresponding to the anti-weakening magnetic flux.

The multiplier 9c realizing the power factor phase type current command synthesizing means multiplies the tangent value of θiv * as the power factor phase equivalent command value by the measured value of the δ-axis current according to the middle side of the equation (6) to obtain γ The shaft current command value iγ * is synthesized. The synthesized γ-axis current command value iγ * is output from the γ-axis current command generator 9.

Regarding the γ-axis current command generator (current command generating means) 9 in FIG. 9, a second embodiment of the power factor phase commander 9b will be shown. Considering that the value of the equation (8b) is approximately K2 · iδ at the time of high speed rotation, it can be understood that the equation (11) may be used as an example of the power factor phase commander 9b. As already described in detail, the power factor phase commander 9b based on the equation (11) is in a direction corresponding to the weak magnetic flux in response to detection of reaching the limit value, and is anti-weak in response to detection of not reaching the limit value. It has the function of automatically adjusting and generating a power factor phase equivalent command value within a limited range in the direction corresponding to the magnetic flux.

In the third and fourth embodiments, the γ-axis current command value iγ is obtained by multiplying the tangent value of θiv * as the power factor phase equivalent command value by the measured δ-axis current value according to the middle side of the equation (6). * Is synthesized. Alternatively, as shown on the right side of equation (6), the tangent value may be multiplied by the δ-axis current command value to generate the γ-axis current command value iγ *.

As described above, five typical examples of the present invention have been given for permanent magnet synchronous motors as representative AC motors. It should be pointed out that the embodiments of the present invention are not limited to these exemplifications, and there are various embodiments according to the present invention.

Points to be noted when the present invention is applied to an AC motor other than a permanent magnet synchronous motor will be described. Depending on the motor (more precisely, depending on the characteristics of the motor), only the power factor phase commanders 2b and 9b need to be changed in the internal configuration of the device (the role, action, The effect is common to AC motors). For other devices, there is basically no need to change the internal configuration. For this reason, supplementary explanation will be given regarding the equations (8) and (11) that explain the detailed internal configuration of the power factor phase commander.

Also regarding the equation (8) showing the first internal configuration example of the power factor phase commander, the places that need to be changed are limited. The formula (8a), which is exponentially the basis of automatic adjustment, can be applied without modification to all AC motors. Similarly, equation (8d) that generates a final power factor phase equivalent command value by weighted averaging of two power factor phase equivalent command values (two end values) is also unmodified for all AC motors. Applicable. When the final power factor phase equivalent command value is generated, only the equations (8b) and (8c) showing the two end values need to be individually changed according to the individual characteristics of the AC motor.

The expression (11) showing the second internal configuration example of the power factor phase commander will be described. The formula (11a), which is exponentially the basis of automatic adjustment, can be applied to all AC motors without modification. When the final power factor phase equivalent command value is generated, only the equations (11b) to (11d) showing the two end values and the weighted average using the characteristics of the end values are used. It is necessary to change individually according to the characteristics of. Note that the weighted average method for the gain presented in the equation (11c) can also be used for other AC motors.

The present invention is suitable for applications that require efficient sensorless driving in a wide speed range from the middle speed range to the rated speed, and further require a reduction in calculation load.

1 AC motor (permanent magnet synchronous motor)
2 phase determiner 2a voltage limit detector 2b power factor phase commander 2c power factor phase detector 2d phase controller 3 power converter 4 current detector 5a 3 phase 2 phase converter 5b 2 phase 3 phase converter 6a vector rotation 6b Vector rotator 7 Cosine sine signal generator 8 Current controller 9 γ-axis current command generator 9a Voltage limit detector 9b Power factor phase commander 9c Multiplier 10 Speed controller 11 Machine speed estimator

Claims (2)

The stator current is captured as a 2 × 1 vector, and the stator current is adjusted so that the vector rotator or the converter with built-in vector rotator follows the γ-axis current command value and δ-axis current command value specified on the γδ rotation coordinate system. The feedback current control means for feedback control and the stator voltage is captured as a 2 × 1 vector on the γδ rotation coordinate system, and the power factor phase equivalent value that is the phase between the stator current and the stator voltage is specified. And a vector rotator phase determining means for determining a rotator phase used in a vector rotator or a vector rotator built-in converter so as to follow the value of
The vector rotator phase determining means comprises:
Voltage limit detection means for detecting whether or not the voltage equivalent value has reached the voltage limit value and outputting a binary detection signal ;
The detection signal of one of the two values corresponding to the detection of reaching the limit value is used in the direction corresponding to the weak magnetic flux, and the other one of the two values corresponding to the detection of not reaching the limit value Exponent command generation means for automatically adjusting and generating a power factor phase equivalent command value within a limited range in a direction corresponding to the anti-weakening magnetic flux using the detection signal ;
Power factor phase detection means for detecting the power factor phase equivalent value of the terminal power of the motor;
Phase control means for controlling and determining the rotator phase so that the detected power factor phase equivalent value follows the power factor phase equivalent command value;
A sensorless drive control device for an AC electric motor. The stator current is captured as a 2 × 1 vector, and the stator current is adjusted so that the vector rotator or the converter with built-in vector rotator follows the γ-axis current command value and δ-axis current command value specified on the γδ rotation coordinate system. Feedback current control means for feedback control, and the stator voltage is captured as a 2 × 1 vector on the γδ rotation coordinate system, and the vector rotator or vector rotation is performed so that the stator voltage exists on the δ axis or the γ axis. A vector rotator phase determining means for determining a phase to be used for the converter built-in converter, and a sensorless drive control device for an AC motor,
The feedback current control means comprises:
Voltage limit detection means for detecting whether or not the voltage equivalent value has reached the voltage limit value and outputting a binary detection signal ;
The detection signal of one of the two values corresponding to the detection of reaching the limit value is used in the direction corresponding to the weak magnetic flux, and the other one of the two values corresponding to the detection of not reaching the limit value Exponent command generation means for automatically adjusting and generating a power factor phase equivalent command value within a limited range in a direction corresponding to the anti-weakening magnetic flux using the detection signal ;
Current command generating means comprising three means: power factor phase type current command synthesizing means for synthesizing at least one of the γ-axis current command value and the δ-axis current command value based on the generated power factor phase equivalent command value. A sensorless drive control device for an AC motor, comprising:

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