JP2006230174A - Vector control method and device for synchronous reluctance motor - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To newly provide a vector control method and a vector control device that dispense with phase detector for solving the problems caused by the phase detector, such as an encoder fitted to a rotor, for a drive control method and to provide a drive control device of a synchronous reluctance motor. <P>SOLUTION: In a phase determiner 10, a secondary state observer 10a is driven by utilizing a stator current for driving motors and the signal of a stator voltage. Additionally, the secondary state observer utilizes the rotational speed of a rotating dq coordinate system and a rotor speed estimated value, estimates the phase of salient magnetic flux on the rotary dq-coordinate system, and outputs the phase toward a phase synchronizer 10b. The salient magnetic flux is a component, in phase with the positive or negative salient phase of a rotor in stator interlink magnetic flux. In the phase synchronizer, a salient magnetic flux phase estimated value, as seen from a fixed αβ-coordinate system, a rotor speed estimated value used as the differential value, and the rotating dq coordinate system rotational speed are generated and outputted. The salient magnetic flux phase estimated value on the fixed αβ-coordinate system is utilized for vector control rather than the phase signal by the phase detector, thus solving the problems. <P>COPYRIGHT: (C)2006,JPO&NCIPI

Description

The present invention relates to a vector control method and apparatus for a synchronous reluctance motor that does not have a magnetic flux generation source such as a permanent magnet in a rotor itself among the synchronous motors. In particular, to ensure the information of the rotor phase (phase is synonymous with position) necessary for the vector rotator for vector control, the rotor phase detector (synonymous with rotor position detector) attached to the rotor Instead, a phase determiner (synonymous with a phase estimator) is used, and the phase determiner is configured with the stator current and stator voltage for driving the motor or their estimated values as input signals. The present invention relates to a method and apparatus.

In order to make a synchronous reluctance motor exhibit high control performance, control of the stator current is indispensable, and a vector control method is conventionally known as a control method for this purpose. In the vector control method, a stator current that contributes to torque generation is expressed on a rotating dq coordinate system configured by orthogonal d-axis and q-axis with the phase of the positive salient pole or negative salient pole of the rotor as an ideal d-axis phase. A current control step of capturing and controlling as a vector signal.

The phase of the positive salient pole or negative salient pole of the rotor that the rotation dq coordinate system aims to synchronize is hereinafter referred to as the rotor phase. The rotation dq coordinate system is ideally a coordinate system in which the d-axis phase coincides with the rotor phase with a spatial phase difference of zero, but in reality, a small amount of phase error occurs between the two. To do. In the following description relating to the present invention, the “quasi-synchronized” state having a small amount of spatial phase error is treated as “synchronized” with emphasis on practicality.

In order to maintain the phase of the rotating dq coordinate system in a synchronized state with the rotor phase, it is generally necessary to know the rotor phase. The phase information can be obtained by mounting a phase detector on the rotor, but this mounting is not preferable in terms of reliability and cost. Based on this recognition, research and development of a sensorless vector control method aiming at drive control using an estimated phase has recently begun to be developed. The phase determiner plays a role of phase estimation in the sensorless vector control method. The configuration of the phase determiner and the input signal to it are highly dependent on the estimation principle. These are broadly divided into two types, one using a stator voltage and current for driving an electric motor or an estimated value thereof, and one using a high-frequency voltage and current forcibly applied from the outside for phase estimation. The The present invention belongs to the former, and in particular, a model-based phase estimation method for processing a stator voltage, current, or an estimated value thereof in accordance with a mathematical model (circuit equation) of an electric motor to perform rotor phase estimation. And the same device. As a prior invention related to a model-based rotor phase estimation method in a synchronous reluctance motor, there is the following.

(1) Shinnaka Shinji: “Vector Control Method for Synchronous Reluctance Motor”, Japanese Patent Laid-Open No. 2001-268998 (filed on 2000-3-17)
(2) Katsumi Kamisato, Yoshikatsu Tomori, Takeshi Shimabukuro, Toshinobu Senju: “Rotor position sensorless vector control method for reluctance motors”, Proceedings of the 1994 National Conference on Industrial Applications of the Institute of Electrical Engineers of Japan, pp. 59-64 (1994-8)
(3) Toshinori Senju, Seiji Aragaki, Katsumi Kamisato: “Sensorless Vector Control of Reluctance Motors Using Speed Disturbance Torque Observer”, 1999 Annual Conference of the Institute of Electrical Engineers of Japan, pp. 5-8 (1999-8)
(4) A. vagati, M.M. Pastorelli, and G.C. Franceschini: “High-Performance Control of Synchronous Reluctance Motors”, IEEE Trans. Industry Applications, 33, 4, pp. 983-991 (1997-7 / 8)
(5) E.E. Capecchi, P.M. Guglielmi, M .; Pastorelle, and A.M. Vvagata: “Position Sensorless Control of Transverse-Laminated Synchronous Reluctance Motors”, Conference Record of the World 2000 IEEE Industritory Partnership 2000.
(6) Chondaluho: “Synchronous reluctance motor rotation speed control device and method”, Japanese Patent Application Laid-Open No. 2003-33096 (filed in 2002-7-10)
(7) Wonjung-hee, et al .: “Magnetic flux measuring device for synchronous reluctance motor and its sensorless control system”, Japanese Patent Application Laid-Open No. 2004-120993 (2003-6-26 application)
(8) Takeshi Hanamoto, Teruo Tsuji, Yoshiaki Tanaka: “Sensorless Control of Synchronous Reluctance Motors Using Extended Magnetic Flux Observer”, Proc. 981-984 (2000-8)
(9) Wonjung Hee et al .: “Motor rotational speed control device”, Japanese Patent Application Laid-Open No. 2003-18875 (2002-1-25 application)
(10) Chen Zhen, Ikuo Hamada, Shinji Miki, Shigeru Okuma: “Disturbance Observer for Sensorless Control of Synchronous Reluctance Motor”, Proc. 1107-1110 (2000-8)

Conventional phase estimation methods using voltage or current for driving an electric motor or an estimated value thereof can be roughly classified into three types from the viewpoint of a physical quantity to be estimated having rotor phase information. The first method is a method of first estimating the stator flux linkage and estimating the rotor phase from the stator flux linkage estimated value. Documents (1) to (7) belong to this estimation method. The invention of the document (1) is based on the phase estimation based on the mirror phase characteristic of the magnetic flux from the estimated value of the stator flux linkage, and the stator flux linkage estimation is caused by the stator flux linkage. Only a simple process such as a simple integration process for the induced voltage is shown. In literatures (2)-(7), an equivalent high-pass filtered stator flux linkage estimate is obtained through first-order low-pass filtering of the induced voltage evaluated on the fixed coordinate system, while rotating coordinates The stator linkage flux estimated value evaluated on the system is converted into a fixed coordinate system, and then first-order low-pass filtering is performed to obtain a low-pass filtered stator linkage flux estimate, and these two stator linkage fluxes are obtained. The estimated value of the stator flux linkage is obtained by frequency hybrid coupling of the estimated values. The source of this estimation method for stator interlinkage magnetic flux is as shown in Document (2). The method of estimating the rotor phase after obtaining the final estimated value of the stator flux linkage is the literature {(2), (3)} and the literature {(4), (5), (6), (7). } Is different. The documents {(4), (5)} and the documents {(6), (7)} are substantially the same in the subsequent rotor phase estimation using the stator flux linkage estimated value.

In the rotor phase estimation method according to the literatures (1) to (7), it is critically important to accurately estimate the stator flux linkage that is the base of the rotor phase estimation. However, it has been confirmed by numerical experiments and the like that the stator flux linkage estimating method proposed or adopted in these documents cannot be operated in a low speed region.

The second method simultaneously estimates the salient pole magnetic flux and the expansion induced voltage corresponding to the salient pole magnetic flux (note that there are various definitions of the expansion induced voltage and differ depending on the inventor). The final phase is estimated using these. In the literature (8), using the stator voltage, current, etc. evaluated on the fixed coordinate system, the eighth-order state observer (α, β components for estimating the stator current, salient pole magnetic flux, and expansion induced voltage) 4th order, total 8th order). Of course, eight observer gains are required to construct an eighth-order state observer. In general, in order to maintain stability estimates, these become time-varying gains according to speed changes. The method of literature (9) corresponds to the method of literature (8) in which all eight observer gains are selected as zero.

The literature (8) method requires the configuration of a higher-order state observer called eighth order. In addition, the observer gain needs to be time-varying for stable estimation of the state variable. In other words, there is a problem that the amount of calculation for performing the state observer is very large, and further, it is difficult to design the observer gain. In the method (9), the observer gain is set to zero, and the possibility that the estimation by the observer is correctly performed is very low. Even if it is assumed that correct estimation is performed, the operating range must be extremely limited due to the principle of the state observer.

Document (10) uses a stator voltage, current, etc. evaluated on a fixed coordinate system for a synchronous reluctance motor to form a secondary disturbance observer, and this extended induced voltage (the extended induced voltage There are various definitions, which differ depending on the inventor). The extended induced voltage directly includes the rotor phase information, and the phase can be immediately estimated from the extended induced voltage estimated value. The phase estimation method using the disturbance observer is the most compact among the phase estimation methods of the prior invention, and provides a good phase estimation at a constant speed in the high speed region. However, this disturbance observer is constructed under the assumption of a constant speed, and good performance is not necessarily obtained for variable speed operation. In addition, since the expansion induced voltage itself disappears at a low speed, there is a problem that it cannot be used unless the speed is higher than a certain level (for example, 1/4 or higher than the rated speed).

Problems to be solved by the invention

The present invention has been made under the above background, and its purpose is to reduce the reliability of an electric motor system and to reduce the axial volume that has conventionally occurred due to the mounting of a rotor phase detector on the rotor. An object of the present invention is to provide a novel sensorless vector control method and apparatus for a synchronous reluctance motor having the following features in order to overcome various problems such as increase, wiring problem, and various costs.
1) Sensorless vector control is possible in almost the entire speed range excluding zero speed, that is, in a wide speed range with the rotation of the rotor.
2) The amount of calculation for estimating the rotor phase is small. This is a phase estimation pursuing the minimum amount of computation.
3) Easy selection of design parameters for rotor phase estimation.
4) Stable rotor phase estimation is possible, which can contribute greatly to stable system operation.
5) No speed detector is required in addition to the phase detector.
6) Rotor phase estimation is resistant to noise.
7) Simple combination with other phase estimation methods (high frequency signal application method etc.) using signals on the rotating dq coordinate system is possible.
8) Simple combined use with other phase estimation methods (high frequency signal application method etc.) using signals on fixed αβ coordinate system.

Means for solving the problem

In order to achieve the above object, according to the first aspect of the present invention, the stator current that contributes to torque generation is expressed by orthogonal d-axis q with the phase of the positive salient pole or negative salient pole of the rotor as the ideal d-axis phase. A method for controlling a vector of a synchronous reluctance motor, comprising: a current control step for capturing and controlling as a vector signal on a rotation dq coordinate system composed of axes; and a phase determination step for determining a phase of the rotation dq coordinate system. The determination process is performed by estimating the stator current for driving the motor, with the salient pole magnetic flux being the same phase component as the rotor positive salient pole phase or rotor negative salient pole phase in the constituent components of the stator flux linkage, A secondary state observer having at least a stator voltage or an estimated value thereof as an input signal is provided, and the phase of the rotating dq coordinate system is determined using at least the salient pole magnetic flux estimated value obtained through the secondary state observer. Characterized in that it was.

A second aspect of the present invention is the vector control method for a synchronous reluctance motor according to the first aspect, wherein the secondary state observer is realized by using an estimated value of a rotor speed. .

A third aspect of the present invention is the vector control method for a synchronous reluctance motor according to the first or second aspect, wherein the differential value or approximate differential value of the phase of the rotating dq coordinate system is used for the secondary state observer. The estimated value of the rotor speed is used.

The invention according to claim 4 is the vector control method for the synchronous reluctance motor according to claim 1, wherein the secondary state observer is realized by using a constant observer gain for the same direction rotation of the rotor. It is characterized by that.

A fifth aspect of the present invention is the vector control method of the synchronous reluctance motor according to the first and fourth aspects, wherein the secondary state observer uses an observer gain having a structure having a multiplicative exchange characteristic with respect to a substitution matrix. It is characterized by being realized.

A sixth aspect of the present invention is the vector control method for a synchronous reluctance motor according to the first aspect, wherein the stator current and the stator voltage for driving the motor, which are input signals of the secondary state observer, and the estimated values thereof are obtained. , A signal defined on the rotation dq coordinate system.

A seventh aspect of the present invention is the vector control method for a synchronous reluctance motor according to the first aspect, wherein the stator current and the stator voltage for driving the motor, which are input signals of the secondary state observer, and the estimated values thereof are obtained. And a signal defined on a fixed αβ coordinate system which is orthogonal to each other and is composed of a fixed α axis and β axis.

According to the eighth aspect of the present invention, the rotating dq coordinates configured by the orthogonal d-axis and q-axis having the stator current contributing to torque generation as the ideal d-axis phase of the positive or negative salient pole of the rotor. A synchronous reluctance motor vector control device having current control means for capturing and controlling as a vector signal on the system, and phase determining means for determining the phase of the rotating dq coordinate system, the phase determining means comprising: Stator current and stator voltage for motor drive, or their estimated values, with salient magnetic flux being the same phase component as the rotor positive salient pole phase or rotor negative salient pole phase in the magnetic flux components And a phase state of the rotation dq coordinate system is determined using at least a salient pole magnetic flux estimated value obtained through the secondary state observer.

Next, the operation of the present invention will be described. First, let us consider a fixed αβ coordinate system in which the direction of the center of the stator U-phase winding of the synchronous reluctance motor is the basic axis α-axis and the sub-axis orthogonal thereto is the β-axis. The positive salient pole phase or negative salient pole phase of the rotor of the synchronous reluctance motor is used as the rotor phase. Rotor phase during rotation, it is assumed that no electrically theta _alpha relative to alpha axis is instantaneously. Next, consider a rotating dq coordinate system composed of a base axis d-axis and a secondary axis q-axis. This rotation dq coordinate system is a coordinate system directed to synchronization with the rotor, and rotates in synchronization with the rotor while including some phase error. As a third coordinate system and the most general coordinate system, a general γδ coordinate system composed of a base γ axis and a secondary axis δ axis that rotates at an arbitrary speed ω specified by the designer is considered. Further, the rotor phase viewed from the base γ-axis of the general γδ coordinate system is defined as θ _γ . As a special case, the general γδ coordinate system is a coordinate system having high generality including a fixed αβ coordinate system and a rotating dq coordinate system that rotates in synchronization with the rotor. In the three coordinate systems, the direction from the base axis to the sub-axis is the positive direction. FIG. 1 shows three coordinate systems.

On the most general γδ coordinate system, the electromagnetic characteristics of the synchronous reluctance motor can be modeled by the following equations (1) to (7).

ω_２ｎは回転子の電気（角）速度であり、Ｒ_１は固定子巻線の抵抗である。Ｌ_ｉ，Ｌ_ｍは固定子の同相インダクタンス、鏡相インダクタンスであり、いわゆるｄｑインダクタンスとは次の関係を有する。

また、ｓは微分演算子である。Here, 2 × 1 vectors ν ₁ , i ₁ , and φ ₁ mean stator voltage, current, and flux linkage, respectively. I is a 2 × 2 unit matrix, and J is a 2 × 2 alternating matrix defined by the following equation.

ω _2n is the electrical (angular) speed of the rotor, and R ₁ is the resistance of the stator winding. L _i and L _m are the common-phase inductance and mirror-phase inductance of the stator, and have the following relationship with the so-called dq inductance.

S is a differential operator.

A unit vector having a rotor phase is defined as in equation (7), and this vector is hereinafter referred to as a phase vector.

または、

According to the magnetic flux theorem shown in the following document (11) which is the most recent research result, the stator interlinkage magnetic flux φ ₁ is (8) to (10) even when defined on the general coordinate system. ) Equation or (11) to (13).
(11) Shinnaka Shinji: “A Study on Dynamic Mathematical Model of Synchronous Reluctance Motor, Simplified Unified Analysis of Model Characteristics”, IEEJ Transactions D, 124, 11, pp. 11-27. 1149-1154 (2004-11)

Or

It should be noted that in the stator flux linkage model in equations (8) to (10) and (11) to (13), a phase vector having rotor phase information appears explicitly. The magnetic flux defined by the expression (9) or (12) having this phase vector explicitly is the “rotor positive salient pole phase or rotor in the constituent components of the stator interlinkage magnetic flux” according to the present invention. It is a salient pole magnetic flux that is in phase with the negative salient pole phase. According to the above document (11), when the rotor phase is selected as the positive salient pole phase, the formulas (8) to (10) and (11) to (13) are expressed as follows. It is analytically clarified that the selected equations (11) to (13) and (8) to (10) are the same. This fact means that the use of the equations (11) to (13) is the same as the use of the equations (8) to (10) in which the rotor phase is changed. Under this recognition, in the following description of the present invention, the equations (8) to (10) are used as the stator flux linkage model without losing generality.

In order to appropriately generate torque, it is necessary to capture and control the stator current that contributes to torque generation as a vector signal on the rotating dq coordinate system that aims to synchronize with the salient pole magnetic flux. To configure the rotating dq coordinate system, it is necessary to determine the phase of the coordinate system. The phase of the rotating dq coordinate system aiming at synchronization with the salient pole magnetic flux needs to be selected as the phase θ _α of the salient pole magnetic flux as viewed from the fixed αβ coordinate system. In other words, the configuration of the rotating dq coordinate system requires the phase θ _{α of the} salient pole magnetic flux. Conventionally, in order to know the phase of the salient pole magnetic flux, a rotor phase detector such as an encoder has been mounted on the motor rotor.

When the rotor phase detector cannot be used, the salient pole magnetic flux phase may be estimated using available stator current and stator voltage signals. The salient pole magnetic phase is not necessarily estimated on the fixed αβ coordinate system. In general, it may be estimated on a general γδ coordinate system determined by the designer. Since the general γδ coordinate system is a coordinate system determined by the designer, the phase θ _α −θ _γ of the general γδ coordinate system with respect to the fixed αβ coordinate system is naturally known. Therefore, if the salient pole magnetic phase θ _γ seen from the general γδ coordinate system is known, the salient pole magnetic flux phase θ _α seen from the fixed αβ coordinate system is inevitably known.

ここに、Ｇは２次状態オブザーバの構成に不可欠なオブザーバゲインである。オブザーバゲインＧは２ｘ２行列である。From the above viewpoint, the present invention is such that the phase determining means of the vector controller of the synchronous reluctance motor has a secondary state observer as means for estimating the salient pole magnetic flux phase. One of the secondary state observers according to the inventions of claims 1 and 8 is known except for the salient magnetic flux to be estimated, and the salient magnetic flux φ _m

Here, G is an observer gain indispensable for the configuration of the secondary state observer. The observer gain G is a 2 × 2 matrix.

It is clear that the state observer of the equation (14) only estimates salient magnetic flux (2 × 1 vector) having the phase θ _γ directly as the 2 × 1 phase vector u (θ _γ ), and this is the secondary minimum dimension. It is.

Then, according to the second state observer according to the invention of claim 1 and claim 8, when _L m _{i d} constituting the salient pole magnetic flux is _L m _i d = const namely constant, the salient pole magnetic flux Explain what can be estimated. Note that the condition where L _{m id} is constant is achieved through d-axis current control, as illustrated in an exemplary embodiment described later. The synchronous reluctance motor expressed by the formulas (1) and (8) to (10) can be re-expressed as the following state equations (15) and (16) with salient pole magnetic flux as a state variable. .

る。

For the salient pole magnetic flux φ _m of the synchronous reluctance motor described by the equations (15) and (16), the equation (14)

The

収束することを意味している。（１７）式の誤差方程式が明示しているように、オブザーバゲインＧは回転子速度ω_２ｎと積の形で、推定値の真値への収束効果を発揮するので、本発明の２次状態オブザーバによれば、ゼロ速度を除く全速度領域で、すなわち回転子の回転を伴う全速度領域で突極磁束推

は、本発明の実施形態例の記述に際し、詳しく説明する。As can be easily understood by those skilled in the art, the equation (17) can be expressed by the secondary state observer according to the present invention.

It means to converge. As the error equation of the equation (17) clearly indicates, the observer gain G exhibits a convergence effect to the true value of the estimated value in the form of the product of the rotor speed ω _2n, and therefore the secondary state of the present invention. According to the observer, the salient-pole flux estimation is performed in the entire speed range excluding the zero speed, that is, in the entire speed range with the rotation of the rotor.

Will be described in detail in the description of the embodiment of the present invention.

規化あるいは成分比の逆正接をとればよい。The secondary state observer of the equation (14) according to the present invention directly estimates the salient pole magnet φ _m proportional to the phase vector u (θ _γ ). Proportional phase vector estimate from salient pole flux estimate, is

The normalization or the arc tangent of the component ratio may be taken.

As is clear from the above description, according to the invention of claim 1 or claim 8, it can be said that an estimated value of the rotor phase necessary for vector control can be obtained without using a rotor phase detector. The effect is obtained. In addition, it is possible to obtain an effect that these estimated values can be obtained in the entire speed region excluding the zero speed, that is, in the entire speed region accompanying the rotation of the rotor. Since the state observer according to the present invention is the most compact secondary minimum dimension for estimating only a required salient pole magnetic flux, the amount of calculation required to drive the state observer is minimized for the state observer. Can also be obtained. Due to the minimum dimensionality of the state observer, the observer gain is a single 2 × 2 matrix. As a result, the design of the observer gain that guarantees the stable convergence of the salient pole magnetic flux estimated value becomes much easier, and the work required for this design can be significantly simplified compared to the conventional eighth-order state observer or the like. Can also be obtained.

Next, the operation of the second aspect of the present invention will be described. In order to realize the secondary state observer according to the first aspect of the invention, information on the rotor speed is necessary as can be understood from the example of the equation (14). The invention of claim 2 realizes the secondary state observer in the invention of claim 1 by using the estimated value of the rotor speed as the rotor speed information. In other words, the secondary state observer can be realized without detecting the rotor speed. As apparent from the above, according to the invention of claim 2, there is obtained an effect that a secondary state observer can be realized without using a speed detector for detecting the rotor speed. In other words, according to the invention of claim 2, the action described in claim 1 can be obtained without using a speed detector for detecting the rotor speed. A specific method for estimating the rotor speed will be described in detail when describing the embodiment of the present invention.

Next, the operation of the third aspect of the present invention will be described. According to the invention of claim 3, the differential value or approximate differential value of the phase of the rotating dq coordinate system can be used as the rotor speed estimation value used for the secondary state observer in the inventions of claim 1 and claim 3. The rotation dq coordinate system aims to synchronize with the rotor phase, and the differential value or approximate differential value of the phase of the coordinate system is most suitable as the approximate value of the rotor speed, which is the differential value of the rotor phase. is there. In addition, since the phase of the rotation dq coordinate system is obtained in advance, the approximate speed value can be obtained with a few additional operations. As apparent from the above description, according to the invention of claim 3, there is obtained an effect that a reasonable speed estimated value matched with the estimated phase can be obtained by a few additional operations. In other words, according to the third aspect of the present invention, the operation described in the first and second aspects can be obtained with a minimum amount of computation.

ここに、１／ｓは積分処理を意味する。Next, the operation of the fourth aspect of the present invention will be described. According to the invention of claim 4, the observer gain indispensable for realizing the secondary state observer of claim 1 is configured to be constant with respect to the rotation of the rotor in the same direction. Therefore, according to the present invention, when the rotor is rotating in the same direction, the relationship of the following equation (18) is established.

Here, 1 / s means integration processing.

磁束推定値を生成するための中間信号である。As understood by those skilled in the art from the equations (14) and (18), according to the invention of claim 4, the secondary state observer for estimating the salient pole magnetic flux has the following equation (19): Will be available.

It is an intermediate signal for generating a magnetic flux estimate.

The secondary state observer of the equation (19) is based on the secondary state observer of the equation (14), and as a result, has the same error equation as the secondary state observer of the equation (14). Inherits the actions related to the invention. The secondary state observer of the equation (14) has a differential operator s which means a differential process on the right side of the equation (14). Realization of the next state observer requires signal differentiation. On the other hand, the secondary state observer of the equation (19) does not have any differential operator s meaning differential processing on the right side. That is, no signal differentiation process is required to realize the secondary state observer described by equation (19). As is well known to those skilled in the art, the actual signal contains noise to some extent, and the differentiation process has a characteristic that reacts very sensitively to the noise, so the differentiation process for the actual signal is tried. It is desirable to avoid it.

As is apparent from the above description, according to the invention of claim 4, since the secondary state observer can be realized without performing any signal differentiation processing, a secondary state observer robust to noise can be realized. The action to say is obtained. In other words, according to the invention of claim 4, the action described in claim 1 can be obtained in a state that is robust to noise.

ここに、Ｉは２ｘ２単位行列であり、Ｊは既に（５）式において定義されている２ｘ２交代行列である。また、ｇ_１，ｇ_２は共にスカラである。Next, the operation of the fifth aspect of the invention will be described. The invention according to claim 5 is the vector control method according to claim 1 or claim 4, wherein the structure of the observer gain is a structure having a multiplicative exchange characteristic for the substitution matrix. Specifically, the invention of claim 5 realizes the observer gain G in the form of the following equation (20).

Here, I is a 2 × 2 unit matrix, and J is a 2 × 2 alternating matrix already defined in equation (5). Further, g ₁ and g ₂ are both scalars.

When the observer gain has the structure of the equation (20), the observer gain exhibits multiplicative exchange characteristics for the substitution matrix. That is, the following equation (21) is established.

ここに、Ｄ（ｓ，ω）は次の（２３）式で定義されたＤ因子と呼ばれる２ｘ２行列である。

当然のことながら、（２２）式の２次状態オブザーバは、（１９）式の２次状態オブザーバ、ひいては（１４）式の２次状態オブザーバに基づくものであり、この結果、請求項１、請求項４の発明に関する作用を継承している。As understood by those skilled in the art from the equations (19) and (21), according to the invention of claim 5, the following (22) is simplified as a secondary state observer for estimating the salient pole magnetic flux. Will be available.

Here, D (s, ω) is a 2 × 2 matrix called a D factor defined by the following equation (23).

As a matter of course, the secondary state observer of the equation (22) is based on the secondary state observer of the equation (19), and hence the secondary state observer of the equation (14). The operation relating to the invention of item 4 is inherited.

As apparent from the above description, according to the invention of claim 5, the secondary state observer can be realized in the simplest form. As a result, an effect is obtained that a secondary state observer that minimizes the required calculation amount can be realized. In other words, according to the invention of claim 5, the action described in claim 1 and claim 4 can be obtained with a minimum amount of calculation.

Next, the operation of the sixth aspect of the invention will be described. The invention defined in claim 6 is a signal defined on a rotating dq coordinate system that aims to synchronize the rotor phase and the signals such as the stator current and voltage that are the input signals of the secondary state observer in the invention of claim 1. It is what. As described in the description of the operation of the first aspect, the salient pole magnetic flux phase cannot be correctly estimated when the rotor is in the non-rotating zero speed state. This characteristic is based on the basic characteristic of the synchronous reluctance motor, and estimation using the signals of the stator current and the stator voltage contributing to torque generation is impossible not only by the present invention but also by any method. . In order to estimate the rotor phase even in the zero speed state, it is necessary to use an estimation method using a signal such as a high-frequency current forcibly applied from the outside that does not contribute to torque generation. Some of these estimation methods are realized on a rotating dq coordinate system and some are realized on a fixed αβ coordinate system. The combined use of the salient pole magnetic flux phase estimation method in the rotor rotation state according to the present invention and the known salient pole magnetic flux phase estimation method in the non-rotation state of the rotor is most consistent when performed on the same coordinate system. And easy.

As is clear from the above description, according to the invention of claim 6, first, it is said that the salient pole magnetic flux phase can be estimated based on the invention of claim 1 using the signal on the rotating dq coordinate system. The effect is obtained. Next, there is an effect that the combined use of the other estimation method for estimating the salient pole magnetic phase in the zero speed state using the signal on the rotating dq coordinate system and the estimation method according to the invention of claim 1 is facilitated. can get.

Next, the function of claim 7 will be described. According to a seventh aspect of the present invention, the stator current and the stator voltage signal for driving the secondary state observer according to the first aspect of the present invention are fixed αβ that is orthogonal to each other and is composed of a fixed α axis and β axis. The signal is a stator current and a stator voltage defined on the coordinate system. The action of the seventh aspect is as follows, as is apparent from the description of the action of the sixth aspect.

According to the seventh aspect of the invention, first, there is obtained an effect that the salient pole magnetic flux phase can be estimated based on the first aspect of the invention using the signals on the fixed αβ coordinate system. Next, there is an effect that the combined use of the other estimation method for estimating the salient magnetic flux phase in the zero speed state using the signal on the fixed αβ coordinate system and the estimation method according to the invention of claim 1 is facilitated. can get.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 2 shows the basic structure of an embodiment of a vector control apparatus in which the vector control method of the present invention is applied to a synchronous reluctance motor. 1 is a synchronous reluctance motor, 2 is a power converter, 3 is a current detector, 4a and 4b are 3 phase 2 phase converters, 2 phase 3 phase converters, 5a and 5b are both vector rotators. , 6 is a cosine sine signal generator, 7 is a current controller, 8 is a command converter, 9 is a speed controller, and 10 is a phase determiner. In FIG. 2, various devices from 3 to 10 constitute a vector control device. In this figure, a 2 × 1 vector signal closely related to the present invention is represented by a single thick signal line in order to ensure simplicity. The following block diagram expression follows this.

The five types of devices 3, 4 a, 4 b, 5 a, 5 b, 6, and 7 use a stator signal that contributes to torque generation as a vector signal on a rotating dq coordinate system composed of two orthogonal axes of the d axis and q axis. And means for executing a current control process for controlling each component of the d-axis and the q-axis to follow each axis current command. The phase determiner 10 is means for executing a phase determination step of the rotating dq coordinate system. The rotor phase estimation value, which is the output signal of the phase determiner, is converted into a cosine / sine signal by the cosine sine signal generator 6 and then passed to the vector rotators 6a and 6b that determine the rotation dq coordinate system.

The three-phase stator current detected by the current detector 3 is converted into a two-phase current on the fixed αβ coordinate system by the three-phase two-phase converter 4a, and then the vector rotator 5a is set to 2 in the rotation dq coordinate system. It is converted into a phase current and sent to the current controller 7. The current controller 7 generates a two-phase voltage command on the rotation dq coordinate system so that the two-phase current on the rotation dq coordinate system follows the current command of each phase, and sends it to the vector rotator 5b. In 5b, the two-phase voltage command on the rotation dq coordinate system is converted into the two-phase voltage command on the fixed αβ coordinate system and sent to the two-phase three-phase converter 4b. In 4 b, the two-phase signal is converted into a three-phase voltage command and output as a command to the power converter 2. The power converter 2 generates electric power according to the command, applies it to the synchronous reluctance motor 1, and drives it. The two-phase current command on the rotating dq coordinate system at this time is the d-axis current command kept constant, and the q-axis current command is obtained from the command converter 8. The command converter generates a q-axis current command proportional to the torque command.

The speed controller 9 converts, which is one rotor speed estimate value of the output signal from the phase determiner 10 (electric speed estimated value), is divided by the number of pole pairs N _p in the machine (square) velocity estimate Has been sent. In this example of FIG. 2, an example in which a speed control system is configured is shown, and thus a torque command is obtained as an output of the speed controller 9. As is well known to those skilled in the art, the speed controller 9 is unnecessary when the control purpose is the generated torque and the speed control system is not configured. In this case, the torque command is directly applied from the outside.

The core of the present invention is the phase determiner 10. In any of the speed control and the torque control, the phase determiner 10 does not require any change. Hereinafter, an exemplary embodiment of the phase determiner 10 will be described without losing generality regarding control modes such as speed control and torque control. This phase determiner receives as input signals a stator voltage command and a stator current, which are estimated values of the stator voltage evaluated on the orthogonal two-axis rotational dq coordinate system, and receives the final phase estimated value of the rotor. And a speed estimation value (electrical speed estimation value) are output.

FIG. 3 shows the internal structure of one embodiment of the phase determiner 10 in FIG. The phase determiner 10 includes a secondary state observer 10a and a phase synchronizer 10b.

出力している。これには、設計者が任意に選定できる一般γδ座標系を回転ｄｑ座標系に選定するようにすればよい。２次状態オブザーバは、回転ｄｑ座標系上の突極磁束の位相

と固定子電圧指令に加えて、回転ｄｑ座標系の回転速度ωと回転子速度（電気的速度）の

In the present embodiment, the measured value of the stator current and the command value of the stator voltage defined on the rotating dq coordinate system are used as the input signal of the secondary state observer 10a. In response, secondary

Output. For this purpose, a general γδ coordinate system that can be arbitrarily selected by the designer may be selected as the rotation dq coordinate system. The secondary state observer is the phase of the salient pole flux on the rotating dq coordinate system.

In addition to the stator voltage command, the rotational speed ω of the rotating dq coordinate system and the rotor speed (electrical speed)

FIG. 4 shows the structure of one embodiment of the secondary state observer 10a for salient pole magnetic flux estimation. The secondary state observer first estimates the salient pole magnetic flux and outputs an estimated value of the salient pole magnetic flux phase based on the estimated salient pole magnetic flux, and sets the rotor speed (electrical speed) to a true speed value. Except for the point that the speed estimated value is used instead, it is realized following the expression (22). In the use of the secondary state observer, as explained in connection with the operation of the present invention, the observer gain 10a-1 is constant for the same direction rotation of the rotor, and is multiplied by the substitution matrix. The structure must have the following exchange characteristics. An example of a specific observer gain for this purpose is as shown in the following equation (24).

の正負符号情報のみを取り出す機能を有する。

徹矢印で表現している。

Has a function of extracting only the positive / negative sign information.

It is expressed with the arrow.

示）を乗じて突極磁束の推定値をマイナスフィードバックしている。本フィードバックに

あり、突極磁束の推定値を真値に収束させることが可能である。In the present embodiment example using the observer gain of the equation (24), the term corresponding to the second term on the right side of the first equation of the equation (22) is arranged as the following equation (26).

The estimated value of the salient pole magnetic flux is negatively fed back. For this feedback

Yes, the estimated value of the salient pole magnetic flux can be converged to a true value.

(22) The D factor on the left side of the first equation is realized as an inverse matrix thereof in the secondary state observer, as clearly shown as block 10a-3 in FIG. FIG. 5 shows an implementation example of the inverse matrix 10a-3 of the D factor. The inverse matrix of the D factor is realized by one integrator and one multiplier for vector signals.

少な突極磁束位相θ_γの推定値は、例えば次の（２７）式に従い、生成し出力すればよい。

図４の初期位相決定器１０ａ−４は（２７）式で示したような処理を行い、突極磁束への

A small estimated value of the salient magnetic flux phase θ _γ may be generated and output in accordance with the following equation (27), for example.

The initial phase determiner 10a-4 in FIG. 4 performs the processing shown in the equation (27) to

同期器は、次の文献（１２）で提示公開された一般化積分形ＰＬＬ法を忠実に実現したものである。一般化積分形ＰＬＬ法は同文献に詳しく説明されているので、本明細書における本位相同期器のこれ以上の説明は省略する。
（１２）新中新二：「永久磁石同期モータの最小次元Ｄ因子状態オブザーバとこれを用いたセンサレスベクトル制御法の提案」、電気学会論文誌Ｄ、１２３、１２、ｐｐ．１４４６−１４６０（２００３−１２）FIG. 6 shows the internal structure of one embodiment of the phase synchronizer 10b. The phase synchronizer 10b includes a ballast 10b-1 and an integrator 10b-2. On the rotating dq coordinate system

The synchronizer faithfully realizes the generalized integral PLL method presented and disclosed in the following document (12). Since the generalized integral PLL method is described in detail in this document, further description of the phase synchronizer in this specification will be omitted.
(12) Shinnaka Shinji: “Proposal of Minimum Dimension D-factor State Observer of Permanent Magnet Synchronous Motor and Sensorless Vector Control Method Using It”, IEEJ Transactions D, 123, 12, pp. 11-27. 1446-1460 (2003-12)

値に変換された後、回転ｄｑ座標系の位相を決定づけるベクトル回転器５ａ、５ｂの回転信号と

すなわち、図３、図６を用いて示した位相同期器１０ｂは、回転ｄｑ座標系の位相の微分値を回転子速度の推定値とし、これを２次状態オブザーバに利用すべく、２次状態オブザーバに向け出力している。As clearly shown in FIG. 2, in sensorless vector control, is it on a fixed αβ coordinate system?

After being converted into values, the rotation signals of the vector rotators 5a and 5b that determine the phase of the rotation dq coordinate system

That is, the phase synchronizer 10b shown in FIGS. 3 and 6 uses the differential value of the phase of the rotating dq coordinate system as an estimated value of the rotor speed, and uses the secondary state observer as a secondary state observer. Outputs to the observer.

に、回転ｄｑ座標系の位相の近似微分値を、２次状態オブザーバに利用したことになる。Instead of the above embodiment, a signal obtained by low-pass filtering the rotation speed ω of the rotation dq coordinate system.

In addition, the approximate differential value of the phase of the rotating dq coordinate system is used for the secondary state observer.

In the embodiment described with reference to FIGS. 2 to 6, the measured value of the stator current, the fixed value as a signal of the stator current and the stator voltage defined on the rotating dq coordinate system to the secondary state observer are fixed. The same command value was used as the estimated value of the child voltage. Instead, the same command value as the estimated value of the stator current defined on the rotating dq coordinate system, the actual value of the stator voltage, and other signals related to the stator current and the stator voltage and the estimated value are used. Point out that there is no problem.

Next, as another embodiment according to the present invention, a stator current, a stator voltage, or an estimated value thereof for driving a secondary state observer is composed of orthogonal α axis and β axis that are orthogonal to each other. An example using what is defined on the fixed αβ coordinate system. FIG. 7 schematically shows the structure of an example embodiment of this type according to the invention. The decisive difference between the exemplary embodiment of FIG. 2 and the exemplary embodiment of FIG. 7 is that, in the exemplary embodiment of FIG. 7, the input signal to the phase determiner 10 is fixed on the fixed αβ coordinate system. The signal is related to the current and voltage of the child. The other devices in FIG. 7 are the same as those in FIG.

器９へ送られている。As clearly shown in FIG. 7, the phase determiner 10 is input with the measured value of the stator current defined on the fixed αβ coordinate system and the same command value as the estimated value of the stator voltage, and the vector rotation Salient magnetic flux phase estimate from the fixed αβ coordinate system that is ultimately used in the vessel (ie, the rotational dq coordinate system

Sent to vessel 9.

FIG. 8 shows the internal structure of one embodiment of the phase determiner 10 in FIG. The phase determiner 10 includes a secondary state observer 10a and a speed estimator 10c.

きる一般γδ座標系を固定αβ座標系に選定するようにすればよい。この場合は、当然、一般γδ座標系の回転速度はゼロ、すなわちω＝０となる。２次状態オブザーバは、固定

座標系上で定義された固定子電流等の信号に加えて、回転子速度（電気的速度）の推定値

In the present embodiment, the input value to the secondary state observer 10a uses the measured value of the stator current defined on the fixed αβ coordinate system and the same command value as the estimated value of the stator voltage. . Accordingly, the secondary state observer uses the salient pole magnetic flux evaluated on the fixed αβ coordinate system with good consistency.

The general γδ coordinate system that can be used may be selected as the fixed αβ coordinate system. In this case, naturally, the rotation speed of the general γδ coordinate system is zero, that is, ω = 0. Secondary state observer is fixed

In addition to signals such as stator current defined on the coordinate system, an estimate of the rotor speed (electrical speed)

（電気的速度）に関して真値に代わって推定値を利用している点を除けば、（１９）式、（２２）式よるものに忠実に従って実現されている。ただし、一般γδ座標系の回転速度ωは、これを固定αβ座標系に帰着させるべく、ゼロω＝０としている。FIG. 9 shows the structure of one embodiment of the secondary state observer 10a. The secondary state observer first estimates the salient pole magnetic flux, and based on the estimated salient pole magnetic flux,

Except for the point that the estimated value is used instead of the true value with respect to (electrical speed), it is realized in accordance with the equations (19) and (22). However, the rotational speed ω of the general γδ coordinate system is set to zero ω = 0 in order to reduce this to the fixed αβ coordinate system.

To use this secondary state observer, the observer gain must be constant for the same direction of rotation of the rotor, as explained in connection with the operation of the present invention. A specific observer gain for this purpose may be configured as in the above-described equation (24). That is, the observer gain may be the same as that of the embodiment described with reference to FIG. In FIG. 9, the observer

磁束の推定値を真値に収束できる特性も同様に確保される。Since the same observer gain as in the embodiment of FIG. 4 is used, the block 10a-2 that contributes to the feedback of the estimated magnetic flux is also the same as in the embodiment of FIG. As a result, times

The characteristic that the estimated value of the magnetic flux can be converged to the true value is similarly secured.

図９の初期位相決定器１０ａ−４は（２８）式で示したような処理を行い初期位相推定値

The initial phase determiner 10a-4 in FIG. 9 performs the processing shown by the equation (28) and the initial phase estimated value.

は、前述の文献（１２）で提示公開された積分フィードバック形速度推定法を忠実に実現したものである。積分フィードバック形速度推定法は同文献に詳しく説明されているので、本明細書における本速度推定器のこれ以上の説明は省略する。FIG. 10 shows the internal structure of one embodiment of the speed estimator 10c, which is another main component of the phase determiner 10 shown in FIG. The speed estimator 10c includes a ballast 10

Is a faithful implementation of the integral feedback velocity estimation method presented and disclosed in the above-mentioned document (12). Since the integral feedback speed estimation method is described in detail in this document, further description of the speed estimator in this specification will be omitted.

信号発生器６へ送られ余弦正弦値に変換された後、回転ｄｑ座標系の位相を決定づけるベクトル

いて示した速度推定器は、回転ｄｑ座標系の位相の微分値を回転子速度の推定値とし、これを２次状態オブザーバに利用すべく、２次状態オブザーバに向け出力している。

A vector that determines the phase of the rotating dq coordinate system after being sent to the signal generator 6 and converted to a cosine sine value

The speed estimator shown in FIG. 5 uses the differential value of the phase of the rotating dq coordinate system as an estimated value of the rotor speed, and outputs it to the secondary state observer so as to use it as the secondary state observer.

転ｄｑ座標系の位相の近似微分値を、２次状態オブザーバに利用したことになる。Instead of the above embodiment example, the rotation speed of the rotation dq coordinate system is subjected to low-pass filter processing.

This means that the approximate differential value of the phase of the rotating dq coordinate system is used for the secondary state observer.

In the embodiment described with reference to FIGS. 7 to 10, an actual measurement value of the stator current as a signal such as a stator current defined on the fixed αβ coordinate system for driving the secondary state observer, The same command value as an estimated value of the stator voltage was used. Instead, the same command value as the estimated value of the stator current defined on the fixed αβ coordinate system, the actual value of the stator voltage, and other signals related to the stator current and the stator voltage and estimated values are used. Point out that there is no problem.

本仕様では、ｄ軸インダクタンスがｑ軸インダクタンスより大きくなている。これは、正突極位相を回転子位相に選定したことを意味する。反対に、ｄ軸ｑ軸のインダクタスを交換し、ｄ軸インダクタンスをｑ軸インダクタンスより小さく選定する場合には、負突極位相を回転子位相に選定したことになる。Next, an example of the result of a numerical experiment performed in accordance with the embodiment shown in FIGS. Table 1 shows an overview of the specifications of the test motor.

In this specification, the d-axis inductance is larger than the q-axis inductance. This means that the positive salient pole phase is selected as the rotor phase. On the contrary, when the d-axis and q-axis inductances are exchanged and the d-axis inductance is selected to be smaller than the q-axis inductance, the negative salient pole phase is selected as the rotor phase.

The observer gain 10a-1 of the secondary state observer 10a shown in the equations (19) and (22) and FIG. 9 is selected as the value of the following equation (29),

Number 29

速度推定器１０ｃを構成する安定器１０ｃ−１を次の（３０）式に選定し、

The ballast 10c-1 constituting the speed estimator 10c is selected by the following equation (30),

Based on this, the current control system and speed control system bands are designed to be 2000 and 80 (rad / s), respectively, the d-axis current command is fixed at 25.8 (A), and the acceleration / deceleration speed including zero-speed passing is achieved. FIG. 11 shows an example of a response when a command (± 125 (rad / s) trapezoidal wave) is given. FIG. 11 shows a speed command, a speed response, and a speed estimated value. From the figure, even for an acceleration / deceleration command including zero-speed passage, the speed command, speed response, and speed estimation value are in good agreement so that the difference is not visible, and the sensorless vector control system has a very good performance. It is confirmed that it has been achieved. This performance is impossible or very difficult to achieve with the prior art presented in Documents (2) to (10).

The phase determiner according to the present invention has been described specifically and in detail using a plurality of exemplary embodiments with reference to various drawings. The phase determiner of the present invention can be implemented in an analog fashion, but is preferably constructed digitally in view of the recent significant advances in digital technology. Although the digital configuration includes a hardware configuration and a software configuration, as is obvious to those skilled in the art, any of the present invention can be configured.

The invention's effect

As is clear from the above description, the present invention has the following effects. According to the invention of claim 1 or claim 8, an effect is obtained that an estimated value of the rotor phase necessary for vector control can be obtained without using a rotor phase detector. In addition, the estimated value can be obtained in the entire speed region excluding the zero speed, that is, in the entire speed region with the rotation of the rotor. Since the state observer according to the present invention is the most compact second-order minimum dimension for estimating only a required salient magnetic flux, the amount of calculation required for driving the state observer is minimized for the state observer. Obtained. The observer gain is a single 2 × 2 matrix due to the minimum dimensionality of the state observer. As a result, the design of the observer gain that guarantees the stable convergence of the salient pole magnetic flux estimated value becomes much easier, and the work required for this design can be significantly simplified compared to the conventional eighth-order state observer or the like. Was also obtained. As a result of these actions, according to the invention of claim 1 or claim 8, the following effects can be obtained. First, there is an effect that the synchronous reluctance motor can be vector controlled without using a phase detector. In addition, the sensorless vector control can be performed in the entire speed region excluding the zero speed, that is, in the entire speed region accompanying the rotation of the rotor. Since the present invention uses the most compact second-order state observer of the minimum dimension, sensorless vector control having the above-described characteristics can be realized with a much smaller amount of computation than the conventional eighth-order state observer. The effect to say is acquired. In addition to this, it is much easier to design observer gains because of the minimum dimensionality of the state observer.As a result, it is effective to construct a highly stable sensorless vector control that has a highly stable state observer. can get. Furthermore, in the vector control of a synchronous reluctance motor, a reduction in the reliability of the motor system, an increase in axial volume, a wiring problem, etc., which have conventionally occurred due to the mounting of a rotor phase detector on the rotor, The effect of being able to overcome problems such as increased costs is obtained.

Next, the effect of the invention of claim 2 will be described. According to the second aspect of the present invention, there is obtained an effect that a secondary state observer can be realized without using a speed detector for detecting the rotor speed. That is, according to the second aspect of the present invention, there has been obtained an effect that the function described in the first aspect can be realized without using a speed detector for detecting the rotor speed. As a result of these actions, according to the invention of claim 2, the effect that the sensorless vector control having the effect of claim 1 can be realized without using the speed detector is obtained.

The effect of the invention of claim 3 will be described. According to the third aspect of the present invention, there is obtained an effect that a reasonable speed estimated value matched with the estimated phase can be obtained with a small amount of calculation. That is, according to the invention of claim 3, an action is obtained that the action described in claims 1 and 2 can be reasonably obtained with a minimum amount of calculation. As a result of this action, according to the invention of claim 3, there is obtained an effect that sensorless vector control can be realized in which the effect of the invention of claim 1 and claim 2 is reasonably enhanced with a minimum speed estimation calculation amount. Be able to.

The effect of the invention of claim 4 will be described. According to the fourth aspect of the present invention, since the secondary state observer can be realized without performing any signal differentiation processing, an effect is obtained that a secondary state observer robust to noise can be realized. In other words, according to the invention of claim 4, the action described in claim 1 can be obtained in a state robust to noise. As a result of this action, according to the invention of claim 4, in addition to the effect of the invention of claim 1, the effect is obtained that sensorless vector control having characteristics resistant to noise can be realized.

The effect of the invention of claim 5 will be described. According to the invention of claim 5, an effect is obtained that a secondary state observer with a reduced calculation load can be realized. That is, according to the invention of claim 5, an action is obtained that the action described in claims 1 and 4 can be obtained with a minimum amount of calculation. As a result of this action, according to the invention of claim 5, in addition to the effects of the inventions of claims 1 and 4, an effect is obtained that sensorless vector control having a minimum amount of computation can be realized.

Next, the effect of the invention of claim 6 will be described. According to the invention of claim 6, first, an effect is obtained that the salient pole magnetic flux phase can be estimated based on the invention of claim 1 by using a signal on the rotating dq coordinate system. Next, there is an effect that it is easy to use in combination the other estimation method for estimating the salient pole magnetic phase in the zero speed state using the signal on the rotating dq coordinate system and the estimation method according to the invention of claim 1. It was. As a result of this action, according to the invention of claim 6, first, an effect is obtained that sensorless vector control based on the invention of claim 1 can be realized by using a signal on the rotation dq coordinate system. Next, it is possible to obtain an effect that the sensorless vector control based on the first aspect of the invention can be easily used in combination with another phase estimation method using a signal on the rotation dq coordinate system. As a result, the effect that the effect and usefulness of the sensorless vector control according to the first aspect of the invention can be enhanced comprehensively can be obtained.

The effect of the invention of claim 7 will be described. According to the seventh aspect of the invention, first, an effect is obtained that the salient pole magnetic flux phase can be estimated based on the first aspect of the invention using the signal on the fixed αβ coordinate system. Next, there is an effect that it is easy to use the other estimation method for estimating the salient pole magnetic phase in the zero speed state using the signal on the fixed αβ coordinate system and the estimation method according to the invention of claim 1. It was. As a result of this action, according to the invention of claim 7, first, an effect is obtained that sensorless vector control based on the invention of claim 1 can be realized by using signals on the fixed αβ coordinate system. Next, it is possible to obtain an effect that the sensorless vector control based on the first aspect of the invention can be easily used in combination with another phase estimation method using signals on the fixed αβ coordinate system. As a result, the effect that the effect and usefulness of the sensorless vector control according to the first aspect of the invention can be enhanced comprehensively can be obtained.

Among the above effects, effects related to control performance were verified and confirmed through numerical experiments.

Vector diagram showing the relationship between the three coordinate systems and the rotor phase (electrical angle) The block diagram which shows the basic composition of the vector control apparatus in one example of embodiment The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment Block diagram showing one configuration example of secondary state observer for salient pole magnetic flux estimation Block diagram showing one implementation example of the inverse matrix of the D factor Block diagram showing one configuration example of a phase synchronizer The block diagram which shows the basic composition of the vector control apparatus in one example of embodiment The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment Block diagram showing one configuration example of secondary state observer for salient pole magnetic flux estimation Block diagram showing one configuration example of a speed estimator The figure which shows the example of a response by the vector control apparatus in 1 example of embodiment

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Synchronous reluctance motor 2 Power converter 3 Current detector 4a Three-phase two-phase converter 4b Two-phase three-phase converter 5a Vector rotator 5b Vector rotator 6 Cosine sine signal generator 7 Current controller 8 Command converter 9 Speed Controller 10 Phase determiner 10a Secondary state observer 10a-1 Observer gain 10a-4 Initial phase determiner 10b Phase synchronizer 10b-1 Ballast 10b-2 Integrator 10c Speed estimator 10c-1 Ballast 10c-2 Integration vessel

Claims (8)

The stator current that contributes to torque generation is captured as a vector signal on a rotating dq coordinate system composed of orthogonal d-axis and q-axis with the phase of the positive or negative salient pole of the rotor as the ideal d-axis phase. A vector control method for a synchronous reluctance motor having a current control step for controlling and a phase determination step for determining a phase of a rotating dq coordinate system,
The stator for driving an electric motor in which the phase determination step is to estimate a salient pole magnetic flux that is a component in phase with a rotor positive salient pole phase or a rotor negative salient pole phase in the constituent components of the stator linkage flux. A secondary state observer having at least an electric current and a stator voltage or an estimated value thereof as input signals is provided, and the phase of the rotating dq coordinate system is determined using at least a salient pole magnetic flux estimated value obtained through the secondary state observer. A vector control method for a synchronous reluctance motor, characterized in that it is configured as described above. 2. The vector control method for a synchronous reluctance motor according to claim 1, wherein the secondary state observer is realized by using an estimated value of a rotor speed. 3. The synchronous reluctance motor according to claim 1, wherein a differential value or approximate differential value of the phase of the rotating dq coordinate system is used as the rotor speed estimation value used for the secondary state observer. Vector control method. 2. The vector control method for a synchronous reluctance motor according to claim 1, wherein the secondary state observer is realized by using a constant observer gain for the same direction rotation of the rotor. 5. The vector control method for a synchronous reluctance motor according to claim 1, wherein the secondary state observer is realized by using an observer gain having a structure having a multiplicative exchange characteristic with respect to a substitution matrix. . The stator signal and stator voltage for driving the motor, which are input signals of the secondary state observer, and their estimated values are defined as signals defined on the rotating dq coordinate system. Vector control method for synchronous reluctance motor. The stator current and stator voltage for driving the motor, which are input signals of the secondary state observer, or their estimated values are defined on a fixed αβ coordinate system that is orthogonal to each other and is composed of fixed α and β axes. The vector control method for a synchronous reluctance motor according to claim 1, wherein the signal is a generated signal. The stator current that contributes to torque generation is captured as a vector signal on a rotating dq coordinate system composed of orthogonal d-axis and q-axis with the phase of the positive or negative salient pole of the rotor as the ideal d-axis phase. A vector control device for a synchronous reluctance motor having current control means for controlling and phase determining means for determining a phase of a rotating dq coordinate system,
The stator for driving an electric motor in which the phase determining means is for estimating a salient pole magnetic flux that is a component in phase with the rotor positive salient pole phase or the rotor negative salient pole phase in the constituent components of the stator flux linkage. A secondary state observer having at least an electric current and a stator voltage or an estimated value thereof as input signals is provided, and the phase of the rotating dq coordinate system is determined using at least a salient pole magnetic flux estimated value obtained through the secondary state observer. A vector control device for a synchronous reluctance motor, characterized in that it is configured as described above.

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