JP6150211B2 - Digital rotor phase speed estimation device for AC motor - Google Patents

Description

The present invention relates to an AC motor in which a rotor exhibits salient pole characteristics with respect to application of a high-frequency voltage having a frequency higher than the drive fundamental frequency (for example, a permanent magnet synchronous motor having a permanent magnet in the rotor, a wound synchronous motor, and a synchronous reluctance motor). , Rotor phase (synonymous with position), speed and position speed sensor used in drive control device for hybrid field type synchronous motor, induction motor, etc. with permanent magnet and field winding on rotor The present invention relates to a digital rotor phase speed estimation apparatus for estimation without using, that is, sensorless. In particular, the shape of the applied high-frequency voltage is rectangular, and the frequency of the applied high-frequency voltage is about the same as the switching frequency of the power converter (the carrier frequency in the case of PWM) (that is, the same or a fraction of that). The present invention relates to a digital rotor phase speed estimation device that has a critically high frequency.

High-performance control of the AC motor can be achieved by a so-called vector control method. The vector control method requires information on the rotor phase or speed, which is a differential value thereof, and a position speed sensor such as an encoder has been used for some time. However, the use of this type of position / velocity sensor is not preferable in terms of reliability, axial volume, sensor cable routing, cost, and the like, so-called sensorless vector control method that does not require a position / velocity sensor. Research and development has been conducted for many years.

As a powerful sensorless vector control method, a high-frequency voltage with a frequency higher than the drive fundamental frequency is forcibly applied to the motor, and the corresponding high-frequency current (response high-frequency current) is processed to estimate the rotor phase (so-called high-frequency voltage) Various application methods have been developed and reported so far. In the present specification, the constituent components of the stator voltage and the stator current are divided into the driving fundamental wave component and the high-frequency component in two. The driving fundamental wave component is a component directly related to the rotation speed of the motor (that is, a frequency component comparable to the rotor speed (electrical speed)), and the high frequency component is much higher than the driving fundamental wave component. It is a component of a very high frequency (frequency value is known). In particular, in the present invention, the high-frequency component is a component having a marginally high frequency that is about the same as the switching frequency of the power converter (the carrier frequency in the case of PWM) (that is, the same or a fraction thereof). means.

The rotor phase to be estimated may be determined at an arbitrary position of the rotor, but generally, either the negative salient pole phase or the positive salient pole phase of the rotor is selected as the rotor phase. As is well known to those skilled in the art, there is only an electrical phase deviation of ± π / 2 (rad) between the negative salient pole phase and the positive salient pole phase. The other phases are naturally found. Considering the above, hereinafter, the negative salient pole phase of the rotor will be referred to as the rotor phase unless otherwise specified. A biaxial orthogonal coordinate system composed of a d axis synchronized with the rotor phase without phase deviation and a q axis orthogonal to the d axis is called a dq synchronous coordinate system (see FIG. 1).

Although several technical classifications of the high-frequency voltage application method are conceivable, the first classification method is a classification method based on the shape of the applied high-frequency voltage. The high-frequency voltage application method is broadly classified into those in which the shape of the applied high-frequency voltage is a sine shape and a rectangular shape. The shape of the applied high-frequency voltage targeted by the present invention is rectangular.

The high frequency voltage application method of applying a rectangular high frequency voltage is further subdivided from the viewpoint of a coordinate system for applying the high frequency voltage. Specifically, a method of applying a high-frequency voltage on an αβ fixed coordinate system and a high-frequency voltage on a γδ quasi-synchronous coordinate system (a kind of rotating coordinate system) aiming at convergence with zero phase difference to a dq synchronous coordinate system Is subdivided into a method of applying (see FIG. 1). The high-frequency voltage application method targeted by the present invention is to apply a rectangular high-frequency voltage on the γδ quasi-synchronous coordinate system. Furthermore, a rectangular high-frequency voltage is applied only on the γ-axis. That is, no high-frequency voltage is applied on the δ axis.

The high-frequency voltage application method for applying a rectangular high-frequency voltage is based on whether a high-frequency component corresponding to the response high-frequency current needs to be separated and extracted from the stator current by a filter or the like when processing the response high-frequency current. 2 minutes. The method of the present invention belongs to the latter and does not require separation and extraction of high-frequency components from the stator current by a filter or the like.

In this specification, (1) a high-frequency voltage application method for applying a rectangular high-frequency voltage, (2) a rectangular high-frequency voltage is applied only on the γ-axis of the γδ quasi-synchronous coordinate system, and (3) The frequency of the applied high-frequency voltage is as high as the switching frequency of the power converter (carrier frequency in the case of PWM) (that is, the same or a fraction of that), and (4) a filter or the like The high-frequency voltage application method that satisfies the four conditions of not requiring separation and extraction of the high-frequency component (response high-frequency current) from the stator current is simply referred to as “rectangular limit high-frequency voltage application method”. As prior inventions of the rectangular limit high-frequency voltage application method to which the present invention belongs, there are Patent Document 1 and Non-Patent Documents 1 to 4 listed in the Prior Art Document column.

In the rectangular limit high-frequency voltage application method to which the present invention belongs, regarding the extraction of the rotor phase information and the generation of the rotor phase estimation value, there are different techniques depending on the coordinate system that performs processing of the stator current including high-frequency components. There is a characteristic that it is necessary. Based on this characteristic, the rectangular limit high-frequency voltage application method can be further classified into technologies based on a coordinate system that performs processing of a stator current including a high-frequency component. The coordinate system for processing the stator current is classified into an αβ fixed coordinate system and other coordinate systems. The technical classification of the prior invention based on the stator current processing coordinate system is organized as shown in Table 1 below. The third column of the table also shows the current detection period required for generating the rotor phase estimation value when the control period for controlling the fundamental driving component of the stator current is Ts. . Similar to the stator current processing coordinate system, the current detection period required for generating the rotor phase estimation value is deeply related to the technology related to the extraction of the rotor phase information and the generation of the rotor phase estimation value.

The present invention performs processing of a stator current including a high frequency component on a γδ quasi-synchronous coordinate system. As in the present invention, there is Patent Document 1 as a prior invention in which a stator current including a high frequency component is processed on a γδ quasi-synchronous coordinate system, that is, as a prior invention belonging to the same technical field as the present invention. Hereinafter, the technique of Patent Document 1 will be outlined with reference to the drawings.

FIG. 5 relates to a drive control system for a permanent magnet synchronous motor, and is a typical configuration example of a digital drive control apparatus including a digital rotor phase speed estimation apparatus (phase speed estimator 10) based on a rectangular limit high-frequency voltage application method. It is. 1 is an AC motor (synchronous motor), 2 is a power converter (inverter), 3 is a current detector, 4a and 4b are 3 phase 2 phase converters, 2 phase 3 phase converters, 5a and 5b, respectively. Are vector rotators, 6 is a current controller, 7 is a command converter, 8 is a speed controller, 9 is a low-pass filter, 10 is a phase speed estimator, 11 is a coefficient unit, and 12 is a coefficient unit. Each cosine sine signal generator is shown. In FIG. 5, various devices from 2 to 12 except for one electric motor constitute a drive control device. In this figure, a 2 × 1 vector signal is represented by a single thick signal line to ensure simplicity. The following block diagram expression follows this.

The three-phase stator current detected by the current detector 3 is converted into a two-phase current on the αβ fixed coordinate system by the three-phase two-phase converter 4a and then the rotor phase (dq synchronization) by the vector rotator 5a. It is converted into a two-phase current in the γδ quasi-synchronous coordinate system aiming for phase synchronization with zero phase deviation. A driving current (driving fundamental frequency component of the stator current) is extracted from the converted current through the low-pass filter 9 and is sent to the current controller 6. The current controller 6 generates a driving two-phase voltage command value on the γδ quasi-synchronous coordinate system so that the driving two-phase current on the γδ quasi-synchronous coordinate system follows the current command value of each phase. Here, the rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value) received from the phase velocity estimator 10 is superimposed on the γ-axis element of the driving two-phase voltage command value, and superimposed and synthesized. A voltage command value (the δ-axis voltage command value remains the drive voltage command value and remains unchanged) is sent to the vector rotator 5b. In 5b, the voltage command value for superposition and synthesis on the γδ quasi-synchronous coordinate system is converted into a two-phase voltage command value in the αβ fixed coordinate system, and sent to the two-phase three-phase converter 4b. In 4b, the two-phase voltage command value is converted into a three-phase voltage command value and output as a final command value to the power converter 2. The power converter 2 generates electric power according to the command value, applies it to the synchronous motor 1 and drives it.

The phase speed estimator 10 receives a stator current as an output signal of the vector rotator 5a as an input, and receives a rotor phase estimation value, a rotor (electrical) speed estimation value, and a rectangular single-phase high-frequency voltage command. The value is output. The rotor phase estimation value is converted into a cosine / sine signal by the cosine sine signal generator 12, and then passed to the vector rotators 5a and 5b that determine the γδ quasi-synchronous coordinate system. This means that the rotor phase estimation value is the phase of the γδ quasi-synchronous coordinate system (equivalent to the phase of the γ axis).

As is well known to those skilled in the art, the two-phase current command value on the γδ quasi-synchronous coordinate system is obtained by converting the torque command value through the command converter 7. In the speed controller 8, the rotor speed estimated value (rotor electrical speed estimated value), which is one of the output signals from the phase speed estimator 10, is a constant value, and the coefficient unit 11 is used to calculate the reciprocal of the pole pair number Np. And then converted to a machine speed estimate and sent. In the example of FIG. 5, an example in which the speed control system is configured is shown, and thus the torque command value is obtained as the output of the speed controller 8. As is well known to those skilled in the art, the speed controller 8 is unnecessary when the control purpose is torque control and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

相偏差θγと同一。位相偏差はγ軸からみたｄ軸の位相でもある。図１参照）。FIG. 9 shows an example of the internal structure of the phase velocity estimator 10 in FIG. As shown in the upper part of the figure, the present phase velocity estimator 10 basically includes a magnetic pole position estimating means and a pulsating voltage applying means. The pulsating voltage applying means generates a rectangular single-phase high-frequency voltage command value vγ * (denoted as vhd * in the original drawing). The lower part of FIG. 9 shows the detailed configuration of the magnetic pole position estimation means used in the upper phase velocity estimator 10. The input signal of the magnetic pole position estimating means includes a γ-axis current (that is, a γ-axis element of the stator current, expressed as Idc in the original drawing) and a rectangular single-phase high-frequency voltage command value vγ * (in the original drawing, vhd *).

Same as phase deviation θγ. The phase deviation is also the d-axis phase viewed from the γ-axis. (See FIG. 1).

FIG. 10 shows a signal processed by the magnetic pole position estimating means and its temporal timing. The waveform diagram (b) in the figure shows a state of a rectangular single-phase high-frequency voltage command value in which a cycle twice the PWM carrier wave is one cycle. Further, in the waveform diagram (f), discrete time detection of γ-axis current (dc-axis current in the original diagram) is performed four times (8 times in one cycle) every half cycle of the rectangular single-phase high-frequency voltage command value. The situation is shown. The waveform diagram (h) shows a current polarity signal Sp (not a voltage polarity signal) indicating the polarity of the γ-axis current (indicated by Idc in the original drawing). The waveform diagram (i) shows the most important signal for generating an axis error generated by using the current polarity signal Sp (not the voltage polarity signal).

As the waveform diagram (f) clearly shows, the method of Patent Document 1 basically requires four discrete-time current detections (current sample values) in a half cycle of a rectangular high-frequency voltage. . This means that when the control period of the stator current is Ts, current detection is required twice during one control period. In other words, it means that the sampling period of the stator current for estimation is 0.5 Ts (see the third column of Table 1). The feature of the rectangular limit high-frequency voltage application method is that a rotor phase estimation value having a high speed response is obtained after the frequency of the applied high-frequency voltage is increased critically. Since the upper limit of the frequency of the applied high-frequency voltage is also limited by the detection frequency of the stator current including the response high-frequency current, in order to obtain a rotor phase estimation value having a high speed response, the current sample value should be as small as possible. Rotor phase estimation needs to be performed. However, the conventional method of Patent Document 1 requires current detection twice in a half period of the current control period, and cannot always obtain a rotor phase estimation value having sufficiently high speed response (this point is The same applies to the conventional non-patent documents 1 to 3, see the third column of Table 1). Further, as understood from the internal configuration of the magnetic pole position estimating means shown in the lower diagram of FIG. 9, the method of Patent Document 1 requires a complicated calculation to generate the expected phase estimated value.

Daigo Kaneko, Yoshinao Iwaji, Kiyoshi Sakamoto, Tsunehiro Endo: “AC Motor Control Device and AC Motor System”, JP 2005-20918 (2003-6-27), Correction (2006-6-15) )

Ryozo Masaki, Satoru Kaneko, Yoshimi Sakurai, Mitsuyuki Motobu: “Position sensorless control system of IPM motor based on voltage superposition synchronized with carrier wave”, IEEJ Transactions, Vol. 122, No. 1, pp. 37-43 (2002-2) Y. D. Yoon, S .; K. Sul, S.M. Morimoto, and K.M. Ide: “High-Bandwidth Sensorless Algorithm for AC Machines Based on Square-Wave-Type Voltage Injection”, IEEE Trans Ind. Appl. , Vol. 47, no. 3, pp. 1361-1370 (2011-5 / 6) S. Kim, J. et al. I. Ha, and S.H. K. Sul: “PWM Switching Frequency Signal Injection Sensor Method in IPMSM”, IEEE Trans Ind. Appl. , Vol. 48, no. 5, pp. 1576-1587 (2012-9 / 10).

The present invention has been made under the above background, and its purpose is to make the most of the features of the rectangular limit high-frequency voltage application method, and to minimize the current sample value (discrete time detection of the estimation current is a current control current). The phase estimation is performed in the same manner as in discrete time detection), so that a higher frequency voltage can be applied, and a phase estimation value having a high speed response can be generated. Is to provide a digital rotor phase velocity estimation device (phase velocity estimator).

To achieve the above object, the invention of claim 1 is directed to a power converter and a stator current detection for an AC motor in which a rotor exhibits salient pole characteristics with respect to application of a high-frequency voltage having a frequency higher than a drive fundamental frequency. Is a digital rotor phase speed estimation device used for a digital drive control device having a stator current digital control function with a period Ts using a detector, aiming for synchronization with a rotor phase with zero phase deviation On the γ-axis of the γδ quasi-synchronous coordinate system composed of the γ-axis and the δ-axis orthogonal thereto, a rectangular high-frequency voltage with a period Th = Nh * Ts is obtained when Nh is a positive and even number, and the stator current is discrete. A high-frequency voltage applying means that is applied in synchronization with the time detection time, and a stator current including a driving fundamental frequency component and a high-frequency component that are detected in discrete time using the stator current detector at a period Ts. Γδ quasi-synchronous locus A time difference is performed for each period Ts on the system, and at least one of the γ-axis element signal cγ and the δ-axis element signal sδ is obtained by multiplying the time difference stator current by the polarity of the rectangular high-frequency voltage corresponding to the time difference period Ts. A positive correlation signal generating means for generating a positive correlation signal pc having a positive correlation with respect to the phase deviation between the rotor phase and the γ-axis phase using the generated axial element signal for each period Ts; The positive correlation signal pc is signal-processed so that pc becomes zero, and at least one of the γ-axis phase and the estimated value of the rotor phase or the estimated value of the rotor speed that is basically in a calculus with the rotor phase. And a rotor phase speed generating means for generating the estimated value for each period Ts.

According to a second aspect of the present invention, in the digital rotor phase speed estimation device according to the first aspect, the positive / even number Nh is set to two.

A third aspect of the invention is the digital rotor phase speed estimation device according to the first aspect, wherein the AC motor is a synchronous motor.

The following description of the effects and the like of the present invention is based on a permanent magnet synchronous motor having a permanent magnet in the rotor if the rotor is an AC motor having salient pole characteristics with respect to application of a high-frequency voltage having a frequency higher than the drive fundamental frequency. The present invention is applicable to any AC motor such as a wound synchronous motor, a synchronous reluctance motor, a hybrid field synchronous motor, and an induction motor. Embedded magnet type permanent magnet synchronous motors, synchronous reluctance motors, and the like exhibit salient pole characteristics with respect to driving voltage and current. These electric motors similarly exhibit salient pole characteristics even when a high frequency voltage is applied. On the other hand, a surface magnet type permanent magnet synchronous motor and an induction motor that do not show salient pole characteristics with respect to the driving voltage / current show salient pole characteristics when a high frequency voltage is applied. The hybrid field type synchronous motor has characteristics of a permanent magnet type and a wound type synchronous motor, and can exhibit salient pole characteristics with respect to application of a high frequency voltage. In particular, the self-excited hybrid field synchronous motor has a strong saliency.

Consider a γδ quasi-synchronous coordinate system that rotates at a coordinate system speed ωγ specified by the control designer, as shown in FIG. The coordinate system speed ωγ can be one of the estimated speed values of the rotor. The rotation from the main axis (γ axis) to the sub axis (δ axis) is defined as the positive direction. The 2 × 1 vector signals expressing the physical quantities of the AC motor to be handled below are all defined on this coordinate system unless otherwise specified.

First, the effect of the invention of claim 1 will be described. Consider applying a high-frequency voltage for phase estimation superimposed on a voltage for driving an electric motor. In this case, the stator voltage v1, current i1, and linkage flux φ1 can be expressed as a composite vector of two components as follows.

The symbols f and h of the signal on the right side of the expression (1) indicate that they are components of a driving fundamental frequency and a high frequency, respectively. In particular, the three signals v1h, i1h, and φ1h, which are the second terms of each of the three formulas (1), are closely related to the present invention due to the applied high-frequency voltage, the high-frequency current as a response, and the applied high-frequency voltage. The high-frequency magnetic flux is shown respectively. It is assumed that the frequency ωh of the high-frequency voltage superimposed and applied for phase estimation is sufficiently high so that the relationship of the following equation (2) is satisfied (the frequency relative ratio is practically 0.01 or less).

以降では、記号ｓは微分演算子ｄ／ｄｔを、Ｉは２ｘ２単位行列を、Ｊは次式で定義された２ｘ２交代行列を意味するものとする。

Hereinafter, the symbol s is the differential operator d / dt, I is the 2 × 2 unit matrix, and J is the 2 × 2 alternating matrix defined by the following equation.

When the equation (2) is established, the following equations (4) to (6) are established for the high-frequency component of the stator in the AC motor in which the rotor exhibits salient pole characteristics with respect to the application of the high-frequency voltage. To do.

なお、鏡相インダクタンスＬｍは、回転子位相として負突極位相を選定する場合には負となる。Here, Li and Lm are the in-phase inductance and mirror phase inductance of the stator, and have the following relationship with the d-axis and q-axis inductances.

The mirror phase inductance Lm is negative when a negative salient pole phase is selected as the rotor phase.

Based on the above preparation, the high frequency voltage applying means which is the first means of the invention of claim 1 will be described.
When the high frequency voltage vγh is applied only on the γ-axis under the condition where the relationship of the expression (2) is established, the following expression is established.

When the high-frequency voltage v _γh applied on the γ-axis has a rectangular shape with an amplitude V _h , this voltage can be expressed using a signum function as follows.

矩形状高周波電圧の周期Ｔｈの上限は、電動機の電気的時定数の１／２０程度である。周期Ｔｈの上限より、（１０）式における正偶数Ｎｈの最大値は、おのずと定まることになる。According to the invention of claim 1, the period Th of the rectangular high-frequency voltage in the formula (9) is selected so as to satisfy the following formula with respect to the period Ts that is the stator current control period.

The upper limit of the period Th of the rectangular high frequency voltage is about 1/20 of the electrical time constant of the electric motor. From the upper limit of the period Th, the maximum value of positive and even Nh in the equation (10) is naturally determined.

Therefore, the high-frequency current generated by the high-frequency voltage applying means as the first means of the invention of claim 1 is expressed by the following equation.

ここに、電流の脚符ｋはｔ＝ｋＴｓでの離散時間検出時刻を意味し、電圧の脚符ｋ−１は時間ｔ＝（ｋ−１）Ｔｓ〜ｋＴｓの間に印加された電圧を意味する。Next, the positive correlation generating means as the second means of the invention of claim 1 will be described. According to the first aspect of the present invention, the application of the rectangular high-frequency voltage is performed in synchronization with the discrete time detection time (sampling time) of the stator current. That is, the waveform switching time of the rectangular high-frequency voltage is made to coincide with one of the discrete time detection times. In this case, the high-frequency current between the discrete time detection times changes linearly. Therefore, in this case, the differential value of the high-frequency current is accurately replaced with the time difference of the sample value. That is, the following equation is obtained from the equations (10) and (11).

Here, the current mark k means the discrete time detection time at t = kTs, and the voltage mark k-1 means the voltage applied during the time t = (k−1) Ts to kTs. To do.

FIG. 2 shows the case of Nh = 2 as an example of the waveform switching time of the rectangular high frequency voltage and the discrete time detection time (sampling time) of the current. (A) of the figure corresponds to the discrete time detection time (sampling time) of the period Ts of the stator current, (b) corresponds to the rectangular high frequency voltage of the period Th = 2Ts, and (c) corresponds to the applied high frequency voltage. Means the high frequency current and the same discrete time detection value (denoted by a circle). In addition, (d) and (e) are two examples of the PWM carrier wave for the power converter used when applying the high frequency voltage. In a drive control apparatus using a power converter using a PWM carrier wave, in order to detect the stator current in a discrete time with high accuracy, the detection time needs to be the peak or valley of the PWM carrier wave or both. As the two examples show, the current discrete time detection time is also synchronized with the PWM carrier. FIG. 3 shows the case of Nh = 4 as an example of the waveform switching time of the rectangular high-frequency voltage and the current discrete time detection time (sampling time). The meanings of the waveform diagrams (a) to (e) are the same as those in FIG.

上式における頭符＊は、高周波電圧の指令値を意味する。本明細書では、同様に、関連信号の指令値は、関連信号に頭符＊を付して表現している。According to the first aspect of the present invention, a stator current including a drive fundamental frequency component and a high frequency component, which are detected for a discrete time with a period Ts, is time-differed for each period Ts on the γδ quasi-synchronous coordinate system, and a time difference stator is obtained. The current is multiplied by the polarity of the rectangular high-frequency voltage corresponding to the time difference period Ts (not the polarity of the high-frequency current) to generate the γ-axis element signal cγ and the δ-axis element signal sδ. This is expressed by the following equation using a mathematical expression.

The prefix * in the above equation means a command value of the high frequency voltage. In the present specification, similarly, the command value of the related signal is expressed by adding a prefix * to the related signal.

Since the discrete time detection cycle is sufficiently small compared to the cycle of the fundamental frequency component for driving and further sufficiently small compared to the electrical time constant of the motor (see the explanation immediately after the equation (10)), The value of the driving fundamental frequency component of the stator current during the discrete time detection period can be regarded as substantially constant. Therefore, the characteristics of the γ-axis element signal cγ and the δ-axis element signal sδ generated according to the equation (13) are analyzed as follows by the equations (12) and (13).

The positive correlation signal generating means according to the first aspect of the present invention has a positive correlation with respect to the phase deviation θγ between the rotor (d-axis) and the γ-axis using at least one of the γ-axis element signal cγ and the δ-axis element signal sδ. A positive correlation signal pc is generated every period Ts. The generation of the positive correlation signal pc according to the present invention is generally expressed by the following equation.

（１６）式の正相関信号ｐ_ｃの正相関特性は、（１４）式より、次のように解析される。

One of the generation of a representative positive correlation signal according to the present invention is expressed by the following equation.

(16) a positive correlation properties of the positive correlation signal p _c of the expressions (14) from the equation, is analyzed as follows.

図４の正相関特性から理解されるように、正相関領域での正相関信号は、実効的な位相偏差θγであり、位相偏差相当値と呼ぶべきものである（図１参照）。当然のことながら、正相関信号の単位は、位相偏差の単位と同一のｒａｄである。FIG. 4 shows positive correlation characteristics with respect to various salient pole ratios rs using the equation (17). From the figure, when the salient pole ratio rs = 0.5 is large, a positive correlation is ensured in the region where the phase deviation θγ is within 1.1 [rad], and the phase is also obtained when the salient pole ratio rs = 0.1. It can be seen that a positive correlation is secured in a region where the deviation θγ is within about 0.8 [rad]. As understood from the figure, the following relationship is established in the region where the phase deviation θγ is within 0.5 [rad].

As understood from the positive correlation characteristic of FIG. 4, the positive correlation signal in the positive correlation region is an effective phase deviation θγ and should be called a phase deviation equivalent value (see FIG. 1). Of course, the unit of the positive correlation signal is the same rad as the unit of the phase deviation.

Next, the rotor phase speed generating means as the third means of the invention of claim 1 will be described. In the first aspect of the present invention, the positive correlation signal pc is signal-processed so that the positive correlation signal pc becomes zero, and an estimated value of the phase of the γδ quasi-synchronous coordinate system is generated. As understood from the example of FIG. 4, driving the positive correlation signal in the positive correlation region to zero drives the phase deviation θγ to zero, and thus causes the γδ quasi-synchronous coordinate system to converge to the dq synchronous coordinate system. (See FIG. 1). The phase equivalent to the phase and the same differential value of the γδ quasi-synchronous coordinate system (γ-axis) corresponds to an estimated value of at least one of the rotor phase and the speed in the AC motor. From the phase of the shaft, a rotor phase estimation value and a rotor speed estimation value can be obtained.

As is clear from the above description, the claim dealing with the “rectangular limit high-frequency voltage application method” having features such as no need to separate and extract a high-frequency component (response high-frequency current) from the stator current by a filter or the like According to the first aspect of the invention, at least one of the phase of the γδ quasi-synchronous coordinate system (γ-axis phase) and the estimated value of the rotor phase or the estimated value of the rotor speed that is basically in a calculus with the rotor phase. The effect that an estimated value can be obtained is obtained. At this time, as understood from the examples of FIGS. 2 and 3, these values are obtained by the minimum current sample value (the discrete time detection of the estimation current is the same as the discrete time detection of the current control current). The effect that it can be also obtained. This effect cannot be obtained by the conventional “rectangular limit high-frequency voltage application method” (see the third column of Table 1). Furthermore, as indicated by the equations (13), (15), and (16), there is an effect that these values can be obtained with a very small amount of calculation. In the invention of claim 1, unlike the conventional non-patent documents 1 to 3, in order to perform high-frequency voltage application and high-frequency current processing on the same γδ quasi-synchronous coordinate system (see the second column of Table 1), As a result, the phase and the like of the intended γδ quasi-synchronous coordinate system can be generated in a consistent form with high accuracy.

Then, the effect by the invention of Claim 2 is demonstrated. As shown in FIGS. 2 and 3, when the current control period Ts is used as a reference, the minimum value of the period Th of the rectangular high-frequency voltage is Th = 2Ts. The invention of claim 2 selects this minimum value. In other words, the highest high frequency is selected. As a result, under the same current control period, there is an effect that it is possible to obtain the highest speed response regarding estimation. As a result, the effect that the effect of Claim 1 can be heightened is acquired.

Next, the effect of the invention of claim 3 will be described. The AC motors targeted by the present invention can be broadly classified into synchronous motors and induction motors. In so-called sensorless driving that does not use a position / speed sensor, a more difficult motor is a synchronous motor. Therefore, according to the invention of claim 3 in which the AC motor is a synchronous motor, the effect of the present invention shown in claim 1 and the effect that the usefulness can be further enhanced can be obtained.

"Figure showing an example of the relationship between the three coordinate systems and the rotor phase" “Figure showing one example of relationship between control cycle, discrete-time current detection cycle, and high-frequency voltage cycle” “Figure showing one example of relationship between control cycle, discrete-time current detection cycle, and high-frequency voltage cycle” “Figure showing positive correlation characteristics of one positive correlation signal example” "Block diagram showing drive control system for one embodiment" “Block diagram showing basic configuration of phase velocity estimator in one embodiment” “Block diagram showing basic configuration of phase velocity estimator in one embodiment” “Block diagram showing basic configuration of phase velocity estimator in one embodiment” "Block diagram showing a basic configuration example of a conventional phase velocity generator" "Figure showing an example of the relationship between various signals from a conventional phase velocity generator"

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 5 shows an example of a digital drive control device provided with the digital rotor phase speed estimation device of the present invention for a permanent magnet synchronous motor which is a typical AC motor. The general configuration of this drive control device is the same as that of the prior art. The main point of the present invention is a digital rotor phase velocity estimation apparatus. Considering the similarity of the general configuration and the main point of the invention, description of the drive control system and the drive control device is omitted. For these, refer to the explanation already given in the “Background Art” section.

The core of the present invention is the phase speed estimator 10 which is substantially synonymous with the digital rotor phase speed estimation device. In any of the speed control and the torque control, the phase speed estimator 10 does not require any change. Further, even when the driven motor is another synchronous motor or induction motor, the phase speed estimator 10 does not require any change. In the following, an embodiment of the phase speed estimator 10 will be described without losing generality regarding control modes such as speed control and torque control, and without losing generality with respect to the AC motor to be driven.

FIG. 6 shows an embodiment of the phase velocity estimator 10 using the present invention. The basic devices constituting the phase velocity estimator 10 are three devices: a rectangular high-frequency voltage command device 10-1, a correlation signal generator 10-2, and a phase synchronizer 10-3. This figure shows the operation in one control cycle from the acquisition of the stator current at time t = kTs to time t = (k + 1) Ts. Note that “reciprocal of z” in the figure means a “delay device” (in terms of operation, “delay operator”) that generates a time delay of one sampling period Ts with respect to a discrete-time signal. In the present specification, the “reciprocal of z” is used in the same way in other places.

The rectangular high-frequency voltage command device 10-1 generates a rectangular single-phase high-frequency voltage command value (γ-axis high-frequency voltage command value) on the γδ quasi-synchronous coordinate system with the rotor phase estimation value as a base axis (γ-axis) phase. doing. As shown in FIG. 6, the rectangular high frequency voltage command device 10-1 corresponds to the high frequency voltage application means in the phase velocity estimator, and applies the high frequency voltage application together with the devices 5b, 4b, 2 in the drive control device. Has been carried out. The high-frequency voltage command value generated by the rectangular high-frequency voltage command device is as shown in the equations (9) and (10), and the period is set to the minimum positive even number Nh = 2 according to the invention of claim 2. The corresponding one is good. The generated high-frequency voltage command value is actually applied to the motor by the power converter over one control period from time t = (k + 1) Ts. The high-frequency voltage command value is superimposed on the driving voltage command value and applied to the synchronous motor via the power converter, so that a high-frequency current flows. The high frequency current is included in this as the high frequency component of the stator current.

FIG. 6 shows a positive correlation signal generating means configured according to the invention of claim 1 as a correlation signal generator 10-2. The correlation signal generator shown in the figure is constructed in accordance with the equations (13) and (15) exactly.

位相同期器は、相関信号生成器からの正相関信号を入力信号として受け、γδ準同期座標系の位相（本例では、γδ準同期座標系の位相は回転子位相推定値と等価）θα＾とγδ準同期座標系の速度ωγとを出力している。座標系速度は回転子（電気）速度推定値ω２ｎ＾と実効的に同一である。座標系速度そのもののを回転子速度推定値として利用することもあれば（図６では、この様子を示している）、必要に応じローパスフィルタ処理して回転子速度推定値として利用することもある。一般化積分形ＰＬＬ法に基づく位相同期器に関しては、当業者には周知であるので、これ以上の説明は省略する。The phase synchronizer in FIG. 6 implements a rotor phase speed generation means that is a third means. The phase synchronizer is configured based on the digital generalized integral PLL method shown in Equation (19).

The phase synchronizer receives the positive correlation signal from the correlation signal generator as an input signal, and the phase of the γδ quasi-synchronous coordinate system (in this example, the phase of the γδ quasi-synchronous coordinate system is equivalent to the rotor phase estimation value) θα ^ And the velocity ωγ of the γδ quasi-synchronous coordinate system are output. The coordinate system speed is effectively the same as the rotor (electrical) speed estimate ω2n ^. The coordinate system speed itself may be used as a rotor speed estimation value (this state is shown in FIG. 6), or may be used as a rotor speed estimation value by low-pass filtering if necessary. . Since the phase synchronizer based on the generalized integral PLL method is well known to those skilled in the art, further explanation is omitted.

すなわち、高周波電圧指令値の極性に基づく乗算処理は省略できる。この省略可能を考慮し、図７の実施例では、高周波電圧指令値の極性に基づく乗算処理を破線で示した。FIG. 7 shows a second configuration example of the phase velocity estimator configured according to the present invention. The difference from the embodiment of FIG. 6 is the correlation signal generator 10-2. The rectangular high-frequency voltage command device and the phase synchronizer are the same as those in the embodiment of FIG. This configuration example of the correlation signal generator is based on the equation (16). The positive correlation signal of equation (16) can also be evaluated as the following equation, considering equation (13).

That is, the multiplication process based on the polarity of the high-frequency voltage command value can be omitted. In consideration of this omission, the multiplication process based on the polarity of the high-frequency voltage command value is indicated by a broken line in the embodiment of FIG.

すなわち、δ軸要素信号ｓδそのものを正相関信号ｐｃとしている。このため、位相速度推定器への入力信号はδ軸電流（固定子電流のδ軸要素）のみとしている。なお、本正相関信号ｐｃは、位相偏差θγと次の正相関をもつ。

FIG. 8 is a third configuration example of the phase velocity estimator configured according to the present invention. The difference from the embodiment of FIGS. 6 and 7 resides in the correlation signal generator 10-2. The rectangular high-frequency voltage command device and the phase synchronizer are the same as those in the embodiments of FIGS. This configuration example of the correlation signal generator is based on the following equation (21).

That is, the δ-axis element signal sδ itself is used as the positive correlation signal pc. For this reason, the input signal to the phase velocity estimator is only the δ-axis current (δ-axis element of the stator current). The positive correlation signal pc has the following positive correlation with the phase deviation θγ.

The digital rotor phase speed estimation apparatus (phase speed estimator) according to the present invention has been described in detail with reference to three specific examples. For the convenience of specific description of the digital rotor phase speed estimation device, the synchronous motor is used as the driving motor and the drive control device related thereto is taken up. However, the digital rotor phase speed estimation device according to the present invention is a synchronous motor. It is pointed out repeatedly that it is not limited to. The digital rotor phase speed estimation device in the drive control device using another AC motor as the drive motor can be used as it is in the specific embodiment described in detail. Regarding the general arrangement of a drive phase control device and a drive control device in which the drive motor is a synchronous reluctance motor or an induction motor, refer to, for example, Patent Literature (Shinji Shinnaka: “Vector control method of AC motor” And the same device ", Japanese Patent No. 4120775 (1902-3-18)) and the like. Therefore, this description is omitted.

The present invention is suitable for applications that require phase estimation performance with high responsiveness in a speed range including zero speed, among applications in which an AC motor is sensorlessly driven.

1 AC motor (synchronous motor)
DESCRIPTION OF SYMBOLS 2 Power converter 3 Current detector 4a 3 phase 2 phase converter 4b 2 phase 3 phase converter 5a Vector rotator 5b Vector rotator 6 Current controller 7 Command converter 8 Speed controller 9 Low pass filter 10 Phase speed estimator 10-1 Rectangular High Frequency Voltage Commander 10-2 Correlation Signal Generator 10-3 Phase Synchronizer 11 Coefficient Unit 12 Cosine Sine Signal Generator

Claims (3)

A stator current digital control function with a period Ts using a power converter and a stator current detector for an AC motor in which the rotor exhibits salient pole characteristics when a high frequency voltage having a frequency higher than the drive fundamental frequency is applied. A digital rotor phase speed estimation device used in a digital drive control device comprising:
On the γ-axis of the γδ quasi-synchronous coordinate system composed of the γ-axis aiming for synchronization with the rotor phase with zero phase deviation and the δ-axis orthogonal thereto, the cycle Th = Nh * Ts A high frequency voltage applying means adapted to apply the rectangular high frequency voltage in synchronization with the discrete time detection time of the stator current;
A stator current including a driving fundamental frequency component and a high-frequency component, which are detected in a discrete time using the stator current detector at a period Ts, is time-differed for each period Ts on the γδ quasi-synchronous coordinate system, and time Multiplying the difference stator current by the polarity of the rectangular high frequency voltage corresponding to the time difference period Ts to generate at least one of the γ-axis element signal cγ and the δ-axis element signal sδ, and using the generated axis element signal, Positive correlation signal generation means for generating a positive correlation signal pc having a positive correlation with respect to the phase deviation between the rotor phase and the γ-axis phase every period Ts;
The positive correlation signal pc is subjected to signal processing so that the positive correlation signal pc becomes zero, and the γ-axis phase and the estimated value of the rotor phase or the estimation of the rotor speed basically having a calculus relationship with the rotor phase. A rotor phase velocity generating means for generating at least one estimated value of each value for each period Ts;
A digital rotor phase speed estimation apparatus comprising: 2. The digital rotor phase speed estimation apparatus according to claim 1, wherein the positive and even Nh is set to 2. 2. The digital rotor phase speed estimation apparatus according to claim 1, wherein the AC motor is a synchronous motor.

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