CN107276475B - Double-motor series open-phase fault-tolerant prediction type direct torque control method - Google Patents

Double-motor series open-phase fault-tolerant prediction type direct torque control method Download PDF

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CN107276475B
CN107276475B CN201710522578.7A CN201710522578A CN107276475B CN 107276475 B CN107276475 B CN 107276475B CN 201710522578 A CN201710522578 A CN 201710522578A CN 107276475 B CN107276475 B CN 107276475B
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phase
motor
flux linkage
motors
module
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CN107276475A (en
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周扬忠
陈光团
钟天云
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Fuzhou University
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Fuzhou University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation

Abstract

The invention relates to a double-motor series open-phase fault-tolerant prediction type direct torque control method which comprises the links of an inverter, a T6 conversion module, a rotating coordinate conversion module, two motor flux linkage calculation modules, a prediction model module, an evaluation function module and the like. And predicting the power switch combination of the next flapping k +1 inverter from the minimum angle of the loss function by means of a stator flux linkage and an electromagnetic torque prediction model according to the current, the rotor position angle, the rotating speed and other information of the current flapping k. The double-motor series open-phase fault-tolerant prediction type direct torque control method provided by the invention realizes mutual decoupling of operation of two motors after open-phase, improves the open-phase fault-tolerant operation capability of a series driving system, realizes accurate control of electromagnetic torque and stator flux linkage of the two motors, greatly reduces the steady-state pulsation of the electromagnetic torque and the stator flux linkage of the two motors, and ensures that the two motors operate more stably in a steady state.

Description

Double-motor series open-phase fault-tolerant prediction type direct torque control method
Technical Field
The invention relates to a double-motor series open-phase fault-tolerant prediction type direct torque control method.
Background
Two motors in a six-phase series three-phase double-permanent magnet synchronous motor driving system powered by a single six-phase inverter are decoupled from each other, and the decoupling characteristic is achieved by depending on the connection between two motor windings, namely, the tail ends of two phase windings with symmetrical electrical space of the six-phase motor are connected in parallel and then connected in series with one phase winding in the three-phase motor, so that the current of the three-phase winding is uniformly distributed into the two phase windings of the six-phase motor connected in parallel. The current component of the six-phase motor which generates a symmetrical space rotating magnetic field does not flow through the three-phase winding; the current of the three-phase motor flows through the six-phase winding, but does not generate a rotating magnetic field in the six-phase motor. Therefore, the decoupling control between the two motors is realized. By adopting a direct torque control strategy, the dynamic control performance of the torques of the two motors can be further improved, and the control reliability between the two motors can be further improved.
However, when one phase winding in the six-phase winding is broken or one bridge arm in the six-phase inverter bridge has a fault, only the remaining 5-phase winding in the six-phase winding can work, and the phase current of the three-phase winding directly flows through the winding which is electrically symmetrical to the phase-lacking winding. Obviously, the three-phase winding current has adverse effect on the rotating magnetic field of the six-phase motor, and how to keep the torque decoupling control between the two motors is a difficult problem to be solved.
Therefore, the invention provides a fault-tolerant prediction type direct torque control method aiming at the condition that a six-phase motor of a dual-motor series driving system lacks one phase.
Disclosure of Invention
The invention aims to provide a double-motor series default phase fault-tolerant prediction type direct torque control method to overcome the defects in the prior art.
In order to achieve the purpose, the technical scheme of the invention is as follows: a double-motor series open-phase fault-tolerant prediction type direct torque control method provides a six-phase permanent magnet synchronous motor and a three-phase permanent magnet synchronous motor, provides a prediction type direct torque control system, and comprises the following steps: the device comprises a T6 transformation module, a first rotating coordinate transformation module, a second rotating coordinate transformation module, a six-phase motor stator flux linkage calculation module, a three-phase motor stator flux linkage calculation module, a prediction model module and an evaluation function module; the method is realized according to the following steps:
step S1: the T6 conversion module obtains six-phase current i of a six-phase power switch tube in the inverter to be sampledsA～isFAnd after being transformed by the T6 transformation module, the i in the αβ coordinate plane is outputsα(k)、isβ(k)In the xy coordinate plane isx(k)、isy(k)And i in the O1O2 coordinate planeo1(k)、io2(k)；
Step S2: will isα(k)、isβ(k)Rotor position angle theta of six-phase permanent magnet synchronous motorr1(k)The coordinate transformation is carried out on the i-shaped coordinate in the d1q1 coordinate plane, and the i-shaped coordinate is outputd1(k)、iq1(k)(ii) a Will isx(k)、isy(k)The corner position theta of the three-phase permanent magnet synchronous motorr2(k)Transmitted to the second rotary coordinate transformation module and subjected to coordinate transformationI in the transformed output d2q2 coordinate planed2(k)、iq2(k)；
Step S3: will id1(k)、iq1(k)Transmitting the data to the six-phase motor stator flux linkage calculation module to output a stator flux linkage psi in a six-phase motor d1q1 coordinate planed1(k)、ψq1(k)(ii) a Will id2(k)、iq2(k)Transmitting the data to the stator flux linkage calculation module of the three-phase motor, and outputting a stator flux linkage psi in a d2q2 coordinate plane of the three-phase motord2(k)、ψq2(k)；
Step S4: phi (hand-to-hand)d1(k)、ψq1(k)、ψd2(k)、ψq2(k)、id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)DC bus voltage UDCTransmitting the predicted values to the prediction model module to output the k +1 th flapping predicted value T of the electromagnetic torque of the two motorse1(k+1)、Te2(k+1)And predicted value | psi of k +1 th flapping of magnetic linkage amplitudes of two motor statorss1(k+1)|、|ψs2(k+1)|；
Step S5: electromagnetic torque of given motor of two motors
Figure BDA0001337224850000021
Motor stator flux linkage of two motors
Figure BDA0001337224850000022
Predicted value T of k +1 th flapping of electromagnetic torques of two motorse1(k+1)、Te2(k+1)Predicted value of k +1 th flapping of stator flux linkage amplitudes of two motors is | psis1(k+1)|、|ψs2(k+1)| transmitting | to the evaluation function module, calculating corresponding Sb～SfThe evaluation function value cost of (1);
step S6: obtaining S when cost value is minimumb～SfTaking a value according to Sb～SfAnd the value is taken to control the residual healthy five-phase power switch tube, so that the electromagnetic torque and the stator flux linkage of the two motors are subjected to closed-loop control with minimum pulsation.
In an embodiment of the present invention, in the step S1, the T6 transformation module transforms as follows:
Figure BDA0001337224850000031
in an embodiment of the present invention, in the step S2, the first rotational coordinate transformation module is according to
The coordinate transformation is performed as follows:
Figure BDA0001337224850000032
the second rotating coordinate transformation module performs coordinate transformation in the following manner:
Figure BDA0001337224850000033
in an embodiment of the invention, in the step S3, the stator flux linkage ψ in the coordinate plane of the six-phase motor d1q1d1(k)ψq1(k)The method comprises the following steps:
Figure BDA0001337224850000034
wherein psif1Is the rotor flux linkage vector, L, of a six-phase permanent magnet synchronous motord1＝Lsσ1+3Lsm1+3Lrs1、Lq1＝Lsσ1+3Lsm1-3Lrs1，Lsm1＝(Ldm1+Lqm1)/2，Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1Is a six-phase motor phase winding main magnetic flux direct-axis inductor, quadrature-axis inductor, Lsσ1Leakage inductance, L, of phase windings of six-phase machinesd1、Lq1Six-phase motor direct and alternating shaft inductors respectively;
stator flux linkage psi in the three-phase motor d2q2 coordinate planed2(k)ψq2(k)The method comprises the following steps:
Figure BDA0001337224850000035
wherein psif2Is the rotor flux linkage vector, L, of a three-phase permanent magnet synchronous motord2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2，Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2，Lsσ2Is leakage inductance of phase windings of three-phase motor, Lsm2＝(Ldm2+Lqm2)/2，Lrs2＝(Ldm2-Lqm2)/2，Ldm2、Lqm2For three-phase motor phase winding main magnetic flux direct axis inductance, quadrature axis inductance, Ld2、Lq2Three-phase motor direct and alternating shaft inductors respectively.
In an embodiment of the present invention, in the step S4, the method further includes the following steps:
step S41: will be id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、i02(k)Is transmitted to a
Figure BDA0001337224850000041
The calculation module calculates according to the following mode:
Figure BDA0001337224850000042
wherein R iss1Is the winding resistance, omegar1(k)Electrical angular velocity, omega, for six-phase motor rotor rotationr2(k)The electrical angular velocity at which the three-phase motor rotor rotates;
step S42: a set of current Sb～SfValue, DC bus voltage UDCAnd a sending voltage vector calculation link acquires u'sα、usβ、u′sx、usy：
Figure BDA0001337224850000043
Step S43: handle id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、
Figure BDA0001337224850000044
u′sα、usβ、u′sx、usy、ψd1(k)、ψq1(k)、ψd2(k)、ψq2(k)Correspondingly transmitting the data to two motor stator flux linkage prediction modules and outputting the k +1 th beat corresponding Sb～SfPredicted value psi of stator flux linkage of two motorsd1(k+1)、ψq1(k+1)、ψd2(k+1)、ψq2(k+1)The calculation formula is as follows:
Figure BDA0001337224850000051
step S44: will phid1(k+1)ψq1(k+1)、ψd2(k+1)ψq2(k+1)Transmitting the signals to a flux linkage amplitude calculation module, and outputting flux linkage amplitude | psi of the two motors in the following ways1(k+1)|、|ψs2(k+1)|：
Figure BDA0001337224850000052
Step S44: phi (hand-to-hand)d1(k+1)、ψq1(k+1)、ψd2(k+1)、ψq2(k+1)Transmitting to an electromagnetic torque prediction module, and outputting predicted values T of electromagnetic torques of two motors in the following mannere1(k+1)、Te2(k+1)：
Figure BDA0001337224850000053
Figure BDA0001337224850000054
In an embodiment of the present invention, in the step S5, the evaluation function value cost:
Figure BDA0001337224850000055
wherein k is1、k2、k3、k4Is a loss function coefficient.
In an embodiment of the present invention, when the control quantity of the two motors is electromagnetic torque, the electromagnetic torque of the given motor is
Figure BDA0001337224850000056
Given by the system; when the control quantity of the two motors is the rotating speed, the electromagnetic torque of the given motor
Figure BDA0001337224850000057
The two motor speed controllers respectively output given torques; when the control quantity of the two motors is the rotor position angle, the electromagnetic torque of the given motor
Figure BDA0001337224850000058
The output torque is given by two motor position controllers.
In one embodiment of the present invention, in the step S6, S is setb～SfIncreasing the value by 1, and starting the calculation from the step S4; when S isb～SfAfter the values are completely taken, finding out the S with the minimum corresponding cost valueb～Sf。
In one embodiment of the present invention, Sb～SfValue is Sb～Sf＝00000～11111。
Compared with the prior art, the invention has the following beneficial effects:
1) because the decoupling control is carried out on the stator flux linkage and the electromagnetic torque of the two motors considering the flux linkage of the open-phase winding by utilizing the output voltage vector of the inverter considering the open-phase winding voltage, the mutual decoupling of the two motors after the open phase is realized, and the open-phase fault-tolerant operation capability of the series driving system is improved.
2) By means of the prediction control strategy of the electromagnetic torque and the stator flux linkage, the voltage vectors applied to the two motors are always optimal, so that the electromagnetic torque and the stator flux linkage of the two motors are accurately controlled, the electromagnetic torque and the stator flux linkage steady-state pulsation of the two motors are greatly reduced, and the two motors run more stably in a steady state.
Drawings
Fig. 1 is a hardware configuration diagram of a driving system of a two-motor series open-phase fault-tolerant predictive direct torque control method according to an embodiment of the present invention.
Fig. 2 is a schematic connection diagram of a six-phase inverter-powered six-phase series three-phase dual permanent magnet synchronous motor driving circuit according to an embodiment of the present invention.
Fig. 3(a) is a schematic plane view of the electromechanical energy conversion coordinate of the six-phase motor according to an embodiment of the present invention.
Fig. 3(b) is a schematic plane diagram of electromechanical energy conversion coordinate of the three-phase motor according to an embodiment of the present invention.
FIG. 4 is a schematic diagram of a series default phase fault tolerant predictive direct torque control architecture according to an embodiment of the present invention.
FIG. 5 is a block diagram of a prediction model module according to an embodiment of the present invention.
Detailed Description
The technical scheme of the invention is specifically explained below with reference to the accompanying drawings.
As shown in fig. 1, the driving system of the two-motor series open-phase fault-tolerant predictive direct torque control method in the present embodiment includes: the system comprises a rectification circuit, a filter capacitor, a six-phase inverter, a 60-degree offset six-phase symmetrical winding permanent magnet synchronous motor, a three-phase permanent magnet synchronous motor, a six-phase winding current acquisition circuit, two motor rotor position angle acquisition circuits, an isolation drive, a central controller, a man-machine interface and the like. A suitable dc power supply may also be used to provide the six-phase inverter dc bus voltage. The power tube in the inverter adopts IGBT or MOFET, and the central controller adopts DSP or singlechip. The winding current acquisition circuit is formed by combining a Hall current sensor and an operational amplifier, and can also be formed by combining a winding series power resistor and a differential operational amplifier. The Hall scheme can effectively realize the electrical isolation of the control loop and the main loop, and the winding series power resistance scheme can reduce the cost of the driving system. The rotor position angle acquisition circuit can be formed by connecting a rotary encoder with a level conversion circuit and can also be formed by connecting a rotary transformer with a decoding circuit, wherein the cost of the former is lower, but the position angle sampling precision is limited by the number of lines of the encoder, and the cost of the latter is higher, but the position angle sampling precision is higher. Weak voltage signals of the winding current acquisition circuit and the rotor position angle acquisition circuit are sent to the A/D conversion module of the central controller. The control signal to be sent is calculated according to the acquired signal and the fault-tolerant predictive direct torque control strategy of the invention, and the switching action of the power switch tube in the inverter is controlled through the isolation drive.
Further, the connection of the six-phase inverter-powered six-phase series three-phase dual permanent magnet synchronous motor driving circuit is schematically shown in fig. 2, wherein a to F are phase windings of the six-phase motor, and U to W are phase windings of the three-phase motor. The tail ends of the two-phase windings which are spatially symmetrical and connected in parallel are connected in series with the three-phase winding, so that the current component for realizing electromechanical energy conversion in the six-phase motor is ensured not to flow through the three-phase winding; any one-phase three-phase current is divided into two parallel six-phase windings, and the synthetic magnetomotive force generated in the six-phase motor is zero.
Further, in the present embodiment, the definition of the coordinate plane for implementing electromechanical energy conversion in the analysis is shown in fig. 3(a) and 3(b), wherein αβ and xy are stationary coordinate systems, and d1q1And d2q2The synchronous rotating coordinate system is a six-phase and three-phase motor rotor synchronous rotating coordinate system. Thetar1The included angle between the d1 axis and the α axis is the electrical angle of the six-phase motor rotor rotation, and the ω r1 is the electrical angular velocity of the six-phase motor rotor rotation, and the ψ s1, ψ f1, u s1 and is1 are the stator flux linkage vector, the rotor flux linkage vector, the stator voltage vector and the stator current vector of the six-phase motor for realizing the electromechanical energy conversion, and the vectors and the input voltage and the current of the driving system are respectively in d1Axis q1The projections on the axes α, β are respectively indicated by the subscript "d1”、“q1"," α "and" β "indicate that delta 1 is the stator flux linkage and rotation of six-phase motorThe angle between the sub-flux linkages. The same definition of the relevant variables in a three-phase motor is shown in fig. 3 (b). Because the driving system has 6 degrees of freedom, except that the electromagnetic torque and the magnetic linkage in the electromechanical energy conversion planes of the two motors totally have 4 degrees of freedom, the driving system also has 2 degrees of freedom on a non-electromechanical energy conversion shaft system o1o2, and the driving system is called as a zero-sequence shaft system. Electromagnetic torque and magnetic flux linkage pulsation in the electromechanical energy conversion plane directly affect the tangential rotation stability of the two motors, and the variable on the shafting o1o2 directly affects copper loss, iron loss and the like of the six-phase motor. The method comprises the following steps that an A-F phase inversion bridge arm upper power tube and a F-phase inversion bridge arm lower power tube are interlocked to be switched on and switched off, and one switching variable is used for representing the switching condition of the power tubes, namely Si is1 to represent that the upper bridge power tube is switched on and the lower bridge power tube is switched off; si ═ 0 indicates that the upper bridge power tube is off and the lower bridge power tube is on (i ═ a to F).
Further, in this embodiment, a mathematical model of the dual motors in the natural coordinate system a to F may be established according to the relationship among the winding voltage, the current, the flux linkage, and the inductance. In order to reveal the coupling relationship between the two motors, the following formula (1) is used to obtain a constant power transformation matrix T6And transforming the mathematical models in the natural coordinate systems A-F to αβ -xy-o1o 2 stationary coordinate systems.
Figure BDA0001337224850000081
The voltage, flux linkage, electromagnetic torque balance equations in the plane of the six-phase motor αβ are as follows:
Figure BDA0001337224850000082
Figure BDA0001337224850000083
Te1＝p1(ψsαiβ-ψsβiα) (4)
wherein R iss1Is the winding resistance, Lsσ1For self-leakage inductance of each phase winding, Lsm1＝(Ldm1+Lqm1)/2，Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1For six-phase motor phase winding main magnetic flux direct axis inductance, quadrature axis inductance, p1Is the number of pole pairs.
The balance equation of voltage, flux linkage and electromagnetic torque in the xy plane of the three-phase motor is as follows:
Figure BDA0001337224850000084
Figure BDA0001337224850000085
Te2＝p2(ψsxiy-ψsyix) (7)
wherein the relevant variables have a meaning similar to a six-phase motor.
The voltage balance equation in the zero-sequence shafting is as follows:
Figure BDA0001337224850000086
using thetar1Angle, the equations (2) - (4) can be transformed to the coordinate system d1q1And (4) obtaining:
Figure BDA0001337224850000091
Figure BDA0001337224850000092
Te1＝p1(ψd1iq1-ψq1id1) (11)
wherein L isd1＝Lsσ1+3Lsm1+3Lrs1、Lq1＝Lsσ1+3Lsm1-3Lrs1。
Using thetar2Angle, the equations (5) - (7) can be transformed to the coordinate system d2q2And (4) obtaining:
Figure BDA0001337224850000093
Figure BDA0001337224850000094
Te2＝p2(ψd2iq2-ψq2id2) (14)
wherein L isd2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2、Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2。
When the phase A winding of the six-phase motor is broken or the phase A inverter bridge has a fault, the phase A in the six phases does not flow current, only the remaining five phases B-F work, only four controllable degrees of freedom exist, and zero-sequence current does not need to be controlled. In order to realize the decoupling control of the two motors, the A-phase inverter bridge is supposed to always output six-phase A and three-phase U series voltage U0Thus, the six inverter bridge output voltages are as follows:
Figure BDA0001337224850000095
although no current flows in the phase a among the six phases, if the sum of the input voltages of the two-motor series system becomes zero after the phase a winding is considered, U can be obtainedNOThen, formula (15) is generated, and then T6 is used to transform formula (15) into αβ -xy-o1o 2 as follows:
Figure BDA0001337224850000101
order:
Figure BDA0001337224850000102
equation (16) is further changed to:
Figure BDA0001337224850000103
the current [ i ] can be input into the double-motor series systemsAisBisCisDisEisF]Transformed into αβ -xy-o1o 2 by means of T6 matrix, and isAWhen 0, we can solve:
Figure BDA0001337224850000104
thus, the zero sequence voltage uo2The deformation is as follows:
Figure BDA0001337224850000105
if the variable in the synchronous rotating coordinate system is used to represent isα、isxZero sequence voltage uo2The deformation is as follows:
Figure BDA0001337224850000106
wherein the content of the first and second substances,
Figure BDA0001337224850000107
Figure BDA0001337224850000111
wherein the content of the first and second substances,
Figure BDA0001337224850000112
and (3) transforming the rotation of the formula (16) into a synchronous rotating coordinate system to obtain:
Figure BDA0001337224850000113
according to the formulas (9), (10), (12) and (13):
Figure BDA0001337224850000114
the equations (23) (24) solve the differential expression of dq axis system flux linkage and are accordingly expressed in discrete form as follows:
Figure BDA0001337224850000115
since solving the inverse of the 4x4 matrix is time consuming, the leakage inductance of the six-phase motor in the actual system is small. For this reason, the contribution of the leakage inductance of the six-phase motor to the flux linkage of the three-phase motor can be ignored in the actual programming.
This pattern (25) is further simplified to:
Figure BDA0001337224850000121
Tsis a digital control period.
Then the (k + 1) th flapping two motor stator flux linkage amplitude is:
Figure BDA0001337224850000122
Figure BDA0001337224850000123
according to the equations (11) and (14), the predicted values of the electromagnetic torques of the two motors in the next period can be obtained as follows:
Figure BDA0001337224850000124
Figure BDA0001337224850000125
the control purpose of the embodiment is to hope that the voltage vector applied to each digital control period of the two motors realizes the minimum error of the electromagnetic torque and the stator flux linkage amplitude of the two motors, and for this reason, the following evaluation function is selected in the predictive control of the embodiment:
Figure BDA0001337224850000126
obviously, the selected switch combination Sb～SfThe value of equation (31) should be minimized.
The embodiment provides a double-motor series open-phase fault-tolerant prediction type direct torque control method, which aims at two aspects: the method has the advantages that firstly, under the condition that one-phase inverter bridge arm of a double-motor series system is in fault or one-phase windings in six-phase motors are in open circuit, high-performance torque decoupling control of the two motors is still kept, and therefore the running reliability of a series driving system is improved; and secondly, the electromagnetic torque and the stator flux linkage of the two motors are accurately controlled, and the running stability of the two motors is further improved. In the process of constructing the control method, in order to realize the decoupling between the electromagnetic torque and the stator flux linkage of the two motors, the contribution of the open-phase winding coupling flux linkage to the stator flux linkage is considered, namely the six-phase motor stator flux linkage comprises the open-phase winding coupling flux linkage. Similarly, in order to accurately control two motor variables by using the output voltage vector of the residual 5-phase inverter bridge, a phase-lacking phase winding is considered in the output voltage vector of the inverter bridge.
Furthermore, although the six-phase winding lacks one phase, the flux linkage of the phase-lacking winding is still considered in the process of constructing the flux linkage of the stators of the two motors; and simultaneously constructing an inverter voltage vector containing the voltage of the phase-lacking winding. By means of an electromagnetic torque and stator flux prediction control strategy, decoupling control of electromagnetic torque and stator flux minimum pulsation of the two motors is achieved by means of an optimal voltage vector, and therefore reliability of a series driving system is improved. Assuming that the current control is at the kth flutter, the key is to find out the switch combinations Sb-Sf of the next flutter k +1 inverter according to a predictive control method.
The structural block diagram of the prediction type direct torque control system is shown in fig. 4 and comprises an inverter, a T6 transformation, a rotation coordinate transformation, two motor flux linkage calculation links, a prediction model, an evaluation function, two motors and the like. The predictive model is shown in fig. 5. And predicting the power switch combination of the next flapping (k +1 th flapping) inverter from the minimum angle of the loss function by means of a stator flux linkage and an electromagnetic torque prediction model according to the current, the rotor position angle, the rotating speed and other information of the current flapping (k th flapping).
The method comprises the following specific steps:
step (1): six-phase current i is output by the inverter obtained by samplingsA～isFSending to T6 transformation module, and outputting αβ coordinate plane isα(k)isβ(k)In the xy coordinate plane isx(k)isy(k)I in the o1o2 coordinate planeo1(k)io2(k)The computational mathematical model is as follows:
Figure BDA0001337224850000131
step (2): handle isα(k)isβ(k)Six-phase permanent magnet synchronous motor (the motor is abbreviated as PMSM6) rotor position angle thetar1(k)Sending the data to a rotating coordinate transformation module to output i in a d1q1 coordinate planed1(k)iq1(k)(ii) a Handle isx(k)isy(k)Three-phase permanent magnet synchronous motor (the motor is abbreviated as PMSM3) rotor position angle thetar2(k)Sending the data to a rotating coordinate transformation module to output i in a d2q2 coordinate planed2(k)iq2(k)The calculation formula is as follows:
Figure BDA0001337224850000141
and (3): handle id1(k)iq1(k)、id2(k)iq2(k)Respectively sent to stator flux linkage calculation modules of a six-phase motor and a three-phase motor, and respectively output stator flux linkage psi in a d1q1 coordinate plane of the six-phase motord1(k)ψq1(k)Stator flux linkage psi in three-phase motor d2q2 coordinate planed2(k)ψq2(k)The calculation formula is as follows:
Figure BDA0001337224850000142
and (4): phi (hand-to-hand)d1(k)ψq1(k)、ψd2(k)ψq2(k)、id1(k)iq1(k)、id2(k)iq2(k)、θr1(k)、θr2(k)DC bus voltage UDCSending the predicted values to a prediction model link to output the k +1 th predicted value T of the electromagnetic torque of the two motorse1(k+1)Te2(k+1)Predicted value psi of k +1 th flapping amplitude of stator flux linkage of two motorss1(k+1)ψs2(k+1)；
Wherein the step (4) further comprises the following steps
Step (4.1): handle id1(k)iq1(k)、id2(k)iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、i02(k)Is sent to
Figure BDA0001337224850000143
The calculation link has the following calculation formula:
Figure BDA0001337224850000144
step (4.2): a group Sb～SfValue, DC bus voltage UDCSending the voltage vector to a voltage vector calculation link to output u'sαusβu′sxusyThe calculation formula is as follows:
Figure BDA0001337224850000145
step (4.3): handle id1(k)iq1(k)、id2(k)iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、
Figure BDA0001337224850000151
u′sαusβu′sxusy、ψd1(k)ψq1(k)、ψd2(k)ψq2(k)Sending the magnetic flux linkage prediction links to two motor stators and outputting corresponding Sb～SfPredicted value psi of stator flux linkage of two motorsd1(k+1)ψq1(k+1)、ψd2(k+1)ψq2(k+1)The calculation formula is as follows:
Figure BDA0001337224850000152
step (4.4): phi (hand-to-hand)d1(k+1)ψq1(k+1)、ψd2(k+1)ψq2(k+1)Sending the magnetic linkage amplitude value to a magnetic linkage amplitude value calculation link, and outputting magnetic linkage amplitude | psi of the two motorss1(k+1)|、|ψs2(k+1)And the calculation formula is as follows:
Figure BDA0001337224850000153
step (4.5): phi (hand-to-hand)d1(k+1)ψq1(k+1)、ψd2(k+1)ψq2(k+1)Sending the predicted values to an electromagnetic torque prediction link to output predicted values T of the electromagnetic torques of the two motorse1(k+1)、Te2(k+1)The calculation formula is as follows:
Figure BDA0001337224850000154
Figure BDA0001337224850000155
and (5): setting the torques of two motors
Figure BDA0001337224850000156
Stator flux linkage setting of two motors
Figure BDA0001337224850000157
Te1(k+1)Te2(k+1)、ψs1(k+1)ψs2(k+1)Sending to an evaluation function module to calculate the parameters corresponding to Sb-SfThe evaluation function value cost of (1) is calculated as follows:
Figure BDA0001337224850000158
wherein k is1、k2、k3、k4Is a loss function coefficient.
Further, the step Sb～SfThe value is increased by 1 and the calculation is started from step (4). When S isb～SfAfter the values are completely taken, finding out the S with the minimum corresponding cost valueb～SfAnd according to the set Sb～SfAnd value taking is carried out, the output voltage of the inverter is added to a series motor driving system, and the electromagnetic torque and the stator flux linkage of the k +1 flapping two motors are controlled in a minimum pulse closed loop mode.
Wherein the electromagnetic torque is given in the step (5)
Figure BDA0001337224850000161
And
Figure BDA0001337224850000162
depending on the particular two motor control variables. If the electromagnetic torque is controlled, the system directly gives the value; if the rotation speed is controlled, two motor speed controllers (such as PI controllers) respectively output given torque
Figure BDA0001337224850000163
And
Figure BDA0001337224850000164
if the rotor position angle is controlled, the output of the two motor position controllers is the given torque
Figure BDA0001337224850000165
And
Figure BDA0001337224850000166
the above are preferred embodiments of the present invention, and all changes made according to the technical scheme of the present invention that produce functional effects do not exceed the scope of the technical scheme of the present invention belong to the protection scope of the present invention.

Claims (9)

1. A double-motor series open-phase fault-tolerant prediction type direct torque control method provides a six-phase permanent magnet synchronous motor and a three-phase permanent magnet synchronous motor, and is characterized in that a prediction type direct torque control system is provided, and the system comprises: the device comprises a T6 transformation module, a first rotating coordinate transformation module, a second rotating coordinate transformation module, a six-phase motor stator flux linkage calculation module, a three-phase motor stator flux linkage calculation module, a prediction model module and an evaluation function module; the method is realized according to the following steps:
step S1: the T6 conversion module obtains six-phase current i of a six-phase power switch tube in the inverter to be sampledsA～isFAnd after being transformed by the T6 transformation module, the i in the αβ coordinate plane is outputsα(k)、isβ(k)In the xy coordinate plane isx(k)、isy(k)And i in the O1O2 coordinate planeo1(k)、io2(k)；
Step S2: will isα(k)、isβ(k)Rotor position angle theta of six-phase permanent magnet synchronous motorr1(k)Transmitting to the first rotating coordinate transformation module, and outputting i in a d1q1 coordinate plane after coordinate transformationd1(k)、iq1(k)(ii) a Will isx(k)、isy(k)The corner position theta of the three-phase permanent magnet synchronous motorr2(k)Transmitting to the second rotary coordinate transformation module, and outputting i in the d2q2 coordinate plane after coordinate transformationd2(k)、iq2(k)；
Step S3: will id1(k)、iq1(k)Transmitting the data to the six-phase motor stator flux linkage calculation module to output a stator flux linkage psi in a six-phase motor d1q1 coordinate planed1(k)、ψq1(k)(ii) a Will id2(k)、iq2(k)Transmitting the data to the stator flux linkage calculation module of the three-phase motor, and outputting a stator flux linkage psi in a d2q2 coordinate plane of the three-phase motord2(k)、ψq2(k)；
Step S4: phi (hand-to-hand)d1(k)、ψq1(k)、ψd2(k)、ψq2(k)、id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)D.c. currentBus voltage UDCTransmitting the predicted values to the prediction model module to output the k +1 th flapping predicted value T of the electromagnetic torque of the two motorse1(k+1)、Te2(k+1)And predicted value | psi of k +1 th flapping of magnetic linkage amplitudes of two motor statorss1(k+1)|、|ψs2(k+1)|；
Step S5: electromagnetic torque of given motor of two motors
Figure FDA0002338110570000011
Motor stator flux linkage of two motors
Figure FDA0002338110570000012
Predicted value T of k +1 th flapping of electromagnetic torques of two motorse1(k+1)、Te2(k+1)Predicted value of k +1 th flapping of stator flux linkage amplitudes of two motors is | psis1(k+1)|、|ψs2(k+1)| transmitting | to the evaluation function module, calculating corresponding Sb～SfThe evaluation function value cost of (1);
step S6: obtaining S when cost value is minimumb～SfTaking a value according to Sb～SfAnd the value is taken to control the residual healthy five-phase power switch tube, so that the electromagnetic torque and the stator flux linkage of the two motors are subjected to closed-loop control with minimum pulsation.
2. The method as claimed in claim 1, wherein in step S1, the T6 transformation module transforms the direct torque according to the following method:
Figure FDA0002338110570000021
3. the method as claimed in claim 1, wherein in step S2, the first rotating coordinate transformation module performs coordinate transformation as follows:
Figure FDA0002338110570000022
the second rotating coordinate transformation module performs coordinate transformation in the following manner:
Figure FDA0002338110570000023
4. the method as claimed in claim 1, wherein in step S3, the stator flux linkage ψ is determined in the d1q1 coordinate plane of the six-phase motord1(k)、ψq1(k)The method comprises the following steps:
Figure FDA0002338110570000024
wherein psif1Is the rotor flux linkage vector, L, of a six-phase permanent magnet synchronous motord1＝Lsσ1+3Lsm1+3Lrs1、Lq1＝Lsσ1+3Lsm1-3Lrs1，Lsm1＝(Ldm1+Lqm1)/2，Lrs1＝(Ldm1-Lqm1)/2，Ldm1、Lqm1Is a six-phase motor phase winding main magnetic flux direct-axis inductor, quadrature-axis inductor, Lsσ1Leakage inductance, L, of phase windings of six-phase machinesd1、Lq1Six-phase motor direct and alternating shaft inductors respectively;
stator flux linkage psi in the three-phase motor d2q2 coordinate planed2(k)、ψq2(k)The method comprises the following steps:
Figure FDA0002338110570000031
wherein psif2Is the rotor flux linkage vector, L, of a three-phase permanent magnet synchronous motord2＝Lsσ1+2Lsσ2+3Lsm2+3Lrs2，Lq2＝Lsσ1+2Lsσ2+3Lsm2-3Lrs2，Lsσ2Is leakage inductance of phase windings of three-phase motor, Lsm2＝(Ldm2+Lqm2)/2，Lrs2＝(Ldm2-Lqm2)/2，Ldm2、Lqm2For three-phase motor phase winding main magnetic flux direct axis inductance, quadrature axis inductance, Ld2、Lq2Three-phase motor direct and alternating shaft inductors respectively.
5. The method of claim 1, wherein in step S4, the method further comprises the following steps:
step S41: will be id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、io2(k)Is transmitted to a
Figure FDA0002338110570000032
The calculation module calculates according to the following mode:
Figure FDA0002338110570000033
wherein R iss1Is the winding resistance, omegar1(k)Electrical angular velocity, omega, for six-phase motor rotor rotationr2(k)The electrical angular velocity at which the three-phase motor rotor rotates;
step S42: a set of current Sb～SfValue, DC bus voltage UDCAnd a sending voltage vector calculation link acquires u'sα、usβ、u′sx、usy：
Figure FDA0002338110570000034
Wherein u issα、usβRespectively, six phase PMSM stator voltages in a stationary reference frame αComponent of axis and β axis, usx、usyRespectively representing the components of the three-phase PMSM stator voltage on the x-axis and the y-axis of a static coordinate system, uso1、uso2Respectively represents the voltage components, u, of the series system in the zero sequence voltage shafting o1 axis and o2 axiso2Represents a zero sequence voltage;
step S43: handle id1(k)、iq1(k)、id2(k)、iq2(k)、θr1(k)、θr2(k)、ωr1(k)、ωr2(k)、
Figure FDA0002338110570000035
u′sα、usβ、u′sx、usy、ψd1(k)、ψq1(k)、ψd2(k)、ψq2(k)Correspondingly transmitting the data to two motor stator flux linkage prediction modules and outputting the k +1 th beat corresponding Sb～SfPredicted value psi of stator flux linkage of two motorsd1(k+1)、ψq1(k+1)、ψd2(k+1)、ψq2(k+1)The calculation formula is as follows:
Figure FDA0002338110570000041
Figure FDA0002338110570000042
wherein L issσ1Representing the self-leakage inductance of the six-phase PMSM phase winding; l isd1、Lq1The main magnetic flux direct and alternating axis inductors of the six-phase PMSM phase winding are respectively; rs2Resistance of each phase winding of the three-phase PMSM; t issRepresents a control cycle time;
step S44: will phid1(k+1)ψq1(k+1)、ψd2(k+1)ψq2(k+1)Transmitting the signals to a flux linkage amplitude calculation module, and outputting flux linkage amplitude | psi of the two motors in the following ways1(k+1)|、|ψs2(k+1)|：
Figure FDA0002338110570000043
Step S45: phi (hand-to-hand)d1(k+1)、ψq1(k+1)、ψd2(k+1)、ψq2(k+1)Transmitting to an electromagnetic torque prediction module, and outputting predicted values T of electromagnetic torques of two motors in the following mannere1(k+1)、Te2(k+1)：
Figure FDA0002338110570000044
Figure FDA0002338110570000045
Wherein p is1、p2Respectively representing the pole pairs of a six-phase PMSM and a three-phase PMSM; psif1、ψf2The six-phase PMSM and the three-phase PMSM rotor flux linkage vectors are respectively; l isd2、Lq2The direct-quadrature axis inductance of the main magnetic flux of the three-phase PMSM phase winding is shown.
6. The method as claimed in claim 5, wherein in step S5, said evaluation function value cost:
Figure FDA0002338110570000051
wherein k is1、k2、k3、k4Is a loss function coefficient.
7. The method as claimed in claim 1 or 6, wherein when the control variables of the two motors are electromagnetic torques, the electromagnetic torques of the given motor are determined
Figure FDA0002338110570000052
Given by the system; when the control quantity of the two motors is the rotating speed,the given motor electromagnetic torque
Figure FDA0002338110570000053
The two motor speed controllers respectively output given torques; when the control quantity of the two motors is the rotor position angle, the electromagnetic torque of the given motor
Figure FDA0002338110570000054
The output torque is given by two motor position controllers.
8. The method as claimed in claim 1, wherein in step S6, S is selectedb～SfIncreasing the value by 1, and starting the calculation from the step S4; when S isb～SfAfter the values are completely taken, finding out the S with the minimum corresponding cost valueb～Sf。
9. The method according to claim 1, 5 or 8, wherein S isb～SfValue is Sb～Sf＝00000～11111。
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