JP4899788B2 - Drive control method for permanent magnet synchronous motor - Google Patents

Description

The present invention relates to a drive control method for a permanent magnet synchronous motor, and in particular, to ensure the phase of a vector rotator or a vector rotator built-in converter (hereinafter abbreviated as a vector rotator) necessary for drive control. The present invention relates to a sensorless drive control method using a phase determiner (synonymous with a phase estimator). The phase determiner targeted by the present invention is to estimate the phase for a vector rotator or the like through estimation of the rotor magnetic flux or estimation of an induced voltage that is a differential value of the rotor magnetic flux. .

In order to achieve high control performance by using a permanent magnet synchronous motor, a stator current that contributes to torque generation is set to a predetermined phase difference with respect to the phase of the rotor N pole (hereinafter abbreviated as rotor phase). It is necessary to have. For this reason, it is necessary to know the rotor phase evaluated on the fixed αβ coordinate system in which the α axis is selected as the center of the U-phase winding. The simplest method for detecting the rotor phase is to mount a position sensor such as an encoder or resolver on the rotor. However, mounting the position sensor is problematic in terms of reliability, cost, etc. The rotor phase is estimated by using the stator voltage and current signals. The estimation of the rotor phase is substantially achieved by estimating the rotor magnetic flux having the rotor phase information or an induced voltage that is a differential value thereof. As is well known to those skilled in the art, the phase of the rotor flux is the same as the rotor phase. Further, the phase of the induced voltage, which is the differential value of the rotor magnetic flux, has only a simple difference from the rotor phase, that is, + π / 2 or −π / 2. From this relationship, the estimation of the rotor magnetic flux or the estimation of the induced voltage may be considered to substantially mean the estimation of the rotor phase.

The estimation of the rotor magnetic flux or the induced voltage can be performed on the fixed αβ coordinate system or on the quasi-synchronized γδ coordinate system aiming to synchronize with the rotor phase without phase difference. Note that a coordinate system synchronized with the rotor phase without a phase difference is recently called a synchronized dq coordinate system. The quasi-synchronous γδ coordinate system is substantially equivalent to the synchronous dq coordinate system when synchronization is completed. Considering this point, hereinafter, these including the quasi-synchronous γδ coordinate system and the strict synchronous dq coordinate system will be referred to as a synchronous dq coordinate system in a broad sense. On the other hand, a γδ coordinate system that does not aim for synchronization without a phase difference with respect to the rotor phase is called a control γδ coordinate system.

FIG. 8 is a block diagram schematically showing a typical example in which a drive control method using a phase determiner is implemented as an apparatus for a permanent magnet type synchronous motor and is mounted on the drive control method. 1 is a permanent magnet synchronous motor, 2 is a phase determiner, 3 is a power converter, 4 is a current detector, 5a and 5b are 3 phase 2 phase converters, 2 phase 3 phase converters, 6a , 6b are vector rotators, 7 is a cosine sine signal generator, 8 is a current controller, 9 is a command converter, 10 is a speed controller, and 11 is a machine speed estimator. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

In particular, the phase determiner 2 receives the stator voltage and current signals (measured values, command values, etc.) as inputs, and estimates the rotor phase estimated on the fixed αβ coordinate system, and the rotor electrical The estimated speed value is output. The cosine sine signal generator 7 converts the rotor phase estimation value into a cosine / sine signal and transmits it to the vector rotators 6a and 6b. The electric speed estimated value is divided by the number of pole pairs in the machine speed estimator 11 and converted into a machine speed estimated value. 5, 5a, 5b, 6a, 6b, 7, and 8 capture the stator current contributing to torque generation as a vector signal on the synchronous dq coordinate system, and use the d-axis and q-axis components. A feedback-type current control process for controlling the shaft current command value so as to follow each axis is configured. Further, the phase determiner 2 constitutes a phase determination step.

The three-phase stator current detected by the current detector 4 is converted into a two-phase current on the fixed αβ coordinate system by the three-phase two-phase converter 5a, and then the two-phase current in the synchronous dq coordinate system by the vector rotator 6a. It is converted into a phase current and sent to the current controller 8. The current controller 8 generates a two-phase voltage command value on the synchronous dq coordinate system so that the two-phase current on the synchronous dq coordinate system follows the current command value of each phase, and sends it to the vector rotator 6b. In 6b, the two-phase voltage command value on the synchronous dq coordinate system is converted into the two-phase voltage command value on the fixed αβ coordinate system and sent to the two-phase three-phase converter 5b. In 5 b, the two-phase signal is converted into a three-phase voltage command value and output as a command value to the power converter 3. The power converter 3 generates electric power according to the command value, applies it to the synchronous motor 1, and drives it. The two-phase current command value on the synchronous dq coordinate system at this time is obtained by converting the torque command value through the command converter 9. In FIG. 8, dq is added to the foot mark to clearly indicate that the stator voltage and current signals existing on the left side of the vector rotator are signals on the synchronous dq coordinate system. The vector rotators 6a and 6b may be configured integrally with the three-phase two-phase converter and the two-phase three-phase converters 5a and 5b, respectively. In the present invention, these integrated components are referred to as a vector rotator built-in converter.

In this example of FIG. 8, since the example which comprised the speed control system is shown, the torque command value is obtained as an output of the speed controller 10 which inputs the speed command value and the speed estimated value. As is well known to those skilled in the art, the speed controller 10 and the machine speed estimator 11 are unnecessary when the control purpose is the generated torque and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

Among the above-described components, devices particularly related to the present invention are the phase determiner 2 and the command converter 9. The phase determiner estimates the rotor magnetic flux or the induced voltage, which is a differential value, using the stator voltage and current signals, and determines the phase to be used for the vector rotator and the like. In addition, the estimated electric speed is determined. In general, the estimation of the rotor magnetic flux or the induced voltage is obtained by driving a state observer, a disturbance observer or the like by a stator voltage or current signal. Therefore, in the phase determiner, the configuration of a state observer, a disturbance observer, and the like is important, and typical prior inventions related to these include the following.

(1) Shinnaka Shinji: “Synchronous motor vector control method and apparatus”, JP-A-2004-096979
(2) Koko, Machiko Tomioka, Kou Nakano, Tokai Kin: “Position sensorless control of brushless DC motor by adaptive observer”, IEEJ Transactions D, 113, 5, pp. 579-586 (1993-5)
(3) Yasuhiro Yamamoto, Yasuhiro Yoshida, Tadashi Ashikaga: “Sensorless control of PM motors using the same-dimensional magnetic flux observer”, IEEJ Transactions D Volume, 114, 8 pp. 743-749 (2004-8)
(4) Shinnaka Shinji and Sano Kimiori: “A new sensorless vector control method for PMSM using a fixed coordinate fourth-order same-dimensional state observer with integral feedback velocity estimation method”, IEEJ Transactions D Volume, 125, 8, pp . 830-831 (2005-8)
(5) Yoshihiko Kanehara: “Position sensorless control of PM motor using adaptive observer on rotating coordinates”, IEEJ Transactions D, 123, 5, pp. 600-609 (2003-5)
(6) Shinnaka Shinji and Idaisuke: “Applicability of Generalized Integral PLL Method to PMSM Sensorless Vector Control Using Fourth Order Dimensional State Observer”, IEEJ Transactions D Volume, 124, 11, pp. 1164-1165 (2004-11)
(7) Koichi Hirano, Hidehiro Hara, Teruo Tsuji, Ryuichi Oguro: "Sensorless speed control of IPM motor", IEEJ Transactions D Volume, 120, 5, pp. 666-672 (2000-5)
(8) Shinnaka Shinji: “Existence of D-factor disturbance observer for synchronous motor sensorless drive, new phase estimation method of zero phase delay”, IEEJ Transactions D Volume, 126, 9, pp. 1220-1226 (2006-9)
(9) Shinnaka Shinji: “Vector Control Method for Synchronous Motor”, JP-A-2006-223085

  According to these prior inventions, if a state observer, a disturbance observer, etc. are configured, and the stator voltage and current signals (actual measurement values, commands, etc.) for driving the motor are input to the configured observers and the like are driven A rotor phase estimate can be obtained. As is well known to those skilled in the art, motor parameters are required in a configuration such as an observer. In particular, when the motor parameters used for the configuration of the observer and the like, as well as the stator voltage and current signals, which are the drive signals, are correct, the phase estimation value by the observer and the like converges to the true phase value. Actually, there are some differences between the motor parameter true value and the parameter value used for the observer, etc., but the parameter value used should be as close to the true value as possible. This is a major premise for use.

The electric motor is not only a torque generator but also a mechanical energy converter of electric energy, and the energy conversion efficiency of the electric motor is one of the main performances of the electric motor. The issue of improving energy conversion efficiency is, from the viewpoint of motor control, how to implement current control that minimizes losses such as copper loss while achieving the required torque generation, or maximize power factor It can be replaced with a technical problem of how to carry out current control to achieve the above. The command converter 9 can play the role of solving this problem. The command converter generates d-axis and q-axis current commands on the synchronous dq coordinate system that enables highly efficient operation from the torque command. Examples of the prior invention related to the command converter include the following.

(10) Osawa, Nomura: “Improvement of device utilization efficiency of reverse salient-pole PM motor”, Proceedings of National Conference of the Institute of Electrical Engineers of Japan, pp.4. 348-349
(11) Shinnaka Shinji: “Vector control method of synchronous motor”, JP-A-11-041998
(12) Shinnaka Shinji: “Analysis by vector signal of salient pole type synchronous motor for current control with emphasis on efficiency”, IEEJ Transactions D Volume, 119, 5, pp. 648-658 (1999-5)

  For generating d-axis and q-axis current commands on a synchronous dq coordinate system that achieves high efficiency from a torque command, generally, a torque generation equation describing the relationship between the d-axis current, q-axis current, and generated torque is used. And a method for solving nonlinear simultaneous equations by a orbital expression that mathematically describes a trajectory that achieves high efficiency. As is well known to those skilled in the art, this solution is generally very difficult and attempts have been made to improve the command converter to accomplish this more or less efficiently. However, as long as high efficiency is pursued in the drive control method for the conventional permanent magnet synchronous motor, this command converter is indispensable and cannot be eliminated.

Problems to be solved by the invention

The present invention has been made under the above background, and its object is to provide a drive control method that enables high-efficiency operation without requiring a command converter in sensorless drive control of a permanent magnet synchronous motor, and thus Another object of the present invention is to provide a sensorless drive control method capable of high-efficiency operation with a lighter calculation load.

Means for solving the problem

In order to achieve the above object, the invention of claim 1 is a current control for controlling a stator current contributing to torque generation in a feedback manner or a feed forward manner using a vector rotator or a converter incorporating a vector rotator. A method for controlling the driving of a permanent magnet synchronous motor having a process and a phase determination process for determining a phase to be used for a vector rotator or a vector rotator built-in converter, the rotation being present on the d axis of a synchronous dq coordinate system The induced voltage existing on the q-axis, which is a simple differential value of the rotor magnetic flux or the rotor magnetic flux itself only by the permanent magnet , is used as the d-axis on the synchronous dq coordinate system using the stator voltage and stator current signals. And a vector rotator using the phase of the estimated value of the rotor magnetic flux or the induced voltage on the orbit. Alternatively, the phase determination step is configured to determine the phase to be used for the vector rotator built-in converter, and the stator current is applied on the control γδ coordinate system specified by the vector rotator or the vector rotator built-in converter. In the evaluation, the current control process is configured such that the axial current of either the γ-axis or the δ-axis becomes zero.

The invention according to claim 2 is the drive control method of the permanent magnet synchronous motor according to claim 1, wherein the motor parameter having a deviation from the true value of the motor parameter in the estimation determination of the rotor magnetic flux or the induced voltage. By using at least one of the corresponding values, the estimated value of the rotor magnetic flux or the estimated value of the induced voltage is on a different orbit from the d-axis and the q-axis on the synchronous dq coordinate system. It is characterized by that.

The invention of claim 3 is the drive control method for a permanent magnet synchronous motor according to claim 1, characterized in that the orbit on the synchronous dq coordinate system is a hyperbolic or parabolic or elliptical orbit. To do.

A fourth aspect of the present invention is the drive control method for a permanent magnet synchronous motor according to the first aspect, wherein the rotor magnetic flux or the induced voltage is driven by a state observer or a stator voltage signal. It is characterized by using a disturbance observer.

Next, the operation of the present invention will be described. The conventional method aims to converge the estimated value of the estimated rotor voltage or the differential value of the induced voltage to the true value as much as possible. More specifically, the phase of the rotor magnetic flux estimated value converges on the d axis of the synchronous dq coordinate system, or the phase of the induced voltage estimated value converges on the q axis of the synchronous dq coordinate system. It was.

On the other hand, the present invention of claim 1 estimates this so that the rotor magnetic flux estimated value or the induced voltage estimated value converges on a specific trajectory of the dq coordinate system that is neither the d-axis nor the q-axis. It will be decided. In other words, the phase of the rotor magnetic flux estimated value or the phase of the induced voltage estimated value is determined in an estimated manner so that it converges on a specific trajectory. Therefore, according to the first aspect of the present invention, if the phase used for the vector rotator or the like is determined using the phase of these estimated values, the phase of the control γδ coordinate system designated by the vector rotator or the like ( That is, the action of converging (the phase of the γ-axis that is the main axis) on a specific trajectory can be obtained.

In addition to this, according to the invention of claim 1, the current control process is configured such that one of the axial currents becomes zero when the fixed current is evaluated on the control γδ coordinate system specified by the vector rotator or the like. To do. As a result, the controlled stator current can be converged on a specific trajectory. Naturally, the specific trajectory where the fixed current converges will have a phase difference of ± π / 2 (rad) or 0 (rad) from the phase of the control γδ coordinate system. This constant phase difference is ± π / 2 (rad) if the γ-axis component of the stator current is controlled to be zero, and 0 (rad) if the δ-axis component is controlled to be zero. It becomes. As will be described in detail in an embodiment described later, it is easy to control the stator current so that the γ component of the control γδ coordinate system becomes zero. In this case, a constant phase difference of ± π / 2 (rad) always exists between the phase of the control γδ coordinate system and the phase of the stator current, but the control γδ coordinate system and the stator The current is present on the same shape of the trajectory.

Therefore, the stator current can achieve high efficiency without using the command converter, which is indispensable in the prior art, as long as the trajectory left to the designer is selected as the trajectory that achieves high efficiency. The effect that it can exist on an orbit is obtained.

Next, the operation of claim 2 of the present invention will be described. As described in detail in the description of the prior art, observers such as a state observer and a disturbance observer are traditionally used for estimating the rotor magnetic flux or the induced voltage. These observers belong to model matching estimators in a broad sense. In the model matching type estimator, if the true value is used as the motor parameter for the estimator configuration, and further, a signal having no error in the stator voltage and current is used as a signal for driving the estimator, “The phase of the estimated rotor magnetic flux or induced voltage converges to a true value, in other words, these phase estimates converge on the d-axis or q-axis on the synchronous dq coordinate system.” It has a convergence characteristic. If this convergence characteristic is paradoxically captured, a value different from its true value is used as the motor parameter for the estimator configuration, or a signal having a stator voltage or current error as a signal for driving the estimator. Means that the rotor flux or induced voltage phase estimate does not converge on the d-axis or the q-axis. According to the second aspect of the present invention, the motor parameter corresponding value having a deviation from the true value is actively used as the motor parameter for the phase determiner configuration, and the phase estimation value of the rotor magnetic flux or the induced voltage is obtained. An action of converging on a specific trajectory determined by the designer is generated. Hereinafter, the principle of occurrence of this action will be described in detail using mathematical expressions. For simplicity, this is done by taking the estimation of the rotor magnetic flux as an example.

As shown in FIG. 1, a control γδ coordinate system that rotates at a speed ω is considered. Further, it is assumed that the rotor N pole of the permanent magnet synchronous motor has a phase θ _γ instantaneously with respect to the main axis γ-axis. At this time, it is known that the electromagnetic relationship of the permanent magnet synchronous motor is described by the following equations (1) to (8) using signals evaluated on the control γδ coordinate system.

ω_２ｎは回転子の電気速度であり、Ｒ_１は固定子巻線の抵抗（真値）である。Ｌ_ｉ，Ｌ_ｍは固定子の同相インダクタンス、鏡相インダクタンス（真値）であり、ｄ、ｑインダクタンス（真値）とは次の関係を有する。

また、Ｄ（ｓ，ω）は次式で定義されたＤ因子であり、

ｓは微分演算子ｄ／ｄｔである。Here, 2 × 1 vectors ν ₁ , i ₁ , and φ ₁ mean stator voltage, current, and magnetic flux, respectively. 2 × 1 vectors φ _i and φ _m indicate components constituting the stator magnetic flux φ ₁ , φ _i is a magnetic flux induced by the stator current i ₁ (hereinafter abbreviated as a stator induced magnetic flux), and φ _m is a rotor magnetic flux caused by the rotor permanent magnet, and e is an induced voltage which is a differential signal of the rotor magnetic flux as clearly shown in the equation (5). I is a 2 × 2 unit matrix, and J is a 2 × 2 alternating matrix defined by the following equation.

ω _2n is the electric speed of the rotor, and R ₁ is the resistance (true value) of the stator winding. L _i and L _m are the in-phase inductance and mirror phase inductance (true value) of the stator, and have the following relationship with d and q inductance (true value).

D (s, ω) is a D factor defined by the following equation:

s is a differential operator d / dt.

また、インダクタンス対応値に関しては、（１０）式と同一の関係が成立するものとする。この場合、（１２）式で定義されたインダクタンス偏差に関しても、（１０）式と同一の関係が成立する。すなわち、

Permanent magnet synchronous motor stator parameter true value, ie stator winding resistance,

In addition, regarding the inductance corresponding value, the same relationship as in the equation (10) is established. In this case, the same relationship as in equation (10) is established with respect to the inductance deviation defined in equation (12). That is,

（１４）〜（１６）式における磁束上の記号＾は、対応した磁束の推定値を意味する。As an estimation model for estimating the rotor magnetic flux of the permanent magnet synchronous motor described by the equations (1) to (8) in a model matching manner, the following equations (14) to (16) using parameter correspondence values and the like are used. think of.

The symbol ^ on the magnetic flux in the equations (14) to (16) means the estimated value of the corresponding magnetic flux.

図２に明示しているように、同期ｄｑ座標系と制御γδ座標系は、パラメータ偏差、電流信号のレベルに依存した位相差Δθ_ｄをもつ。A coordinate system obtained by using a rotor phase estimation value obtained by model matching in accordance with the estimation model of the equations (14) to (16) as a phase of a vector rotator or the like is defined as a control γδ coordinate system. When the stator current evaluated in the control γδ coordinate system is i _γ , i _δ , the current i _γ , i _δ is the stator current evaluated in the synchronous dq coordinate system in the steady state where ω = ω _2n ≠ 0. It has the following relationship with i _d and i _q (see FIG. 2).

As clearly shown in FIG. 2, the synchronous dq coordinate system and the control γδ coordinate system have a phase difference Δθ _d depending on the parameter deviation and the level of the current signal.

上式は、（１２）式を考慮すると、次の（１９）式として整理することができる。

Consider the case where a true value is used for the value of the stator winding resistance in the model matching estimator. In this case, the condition of ΔR ₁ = 0 is given to the equation (17). Further, as described in the explanation of the operation according to the first aspect of the invention, for the sake of simplicity, the γ-axis current is zero-controlled in a feedback or feed-forward manner. In this case, the condition of i _γ = 0 is given to the equation (17). When two conditions are applied to the equation (17), the following equation (18) is obtained (see FIG. 2).

The above equation can be rearranged as the following equation (19) considering equation (12).

Expression (19) expresses the trajectory that the stator current can take by using only the d-axis current and the q-axis current. In other words, it is expressed on the synchronous dq coordinate system. This equation (19) means an elliptical orbit when ΔL _qd ΔL _q > 0, and a hyperbolic orbit when ΔL _qd ΔL _q <0, and ΔL _qd = 0 and ΔL _q ≠ 0. In the case, it means a parabolic trajectory. In other words, it means that the stator current exists on a specific trajectory on the synchronous dq coordinate system defined by the parameter Φ and the parameter deviations ΔL _qd and ΔL _q (see equation (12)). FIG. 3 shows the relationship between the parameter deviation and the trajectory.

When the stator current is evaluated on the control γδ coordinate system, in this example, the γ component of the stator current is zero. In other words, the stator current is only the δ component. Considering that the stator current is on the δ-axis and the rotor flux estimate is on the γ-axis, the fact that the stator current is on a particular trajectory means that the rotor flux estimate is also It means that it exists on a specific orbit having the same shape with a constant phase difference of ± π / 2 (rad) with respect to the stator current. Similar conclusions can be obtained when the δ-axis current is zero-controlled instead of the γ-axis current. In addition, considering that the induced voltage always has a constant phase difference of ± π / 2 (rad) with respect to the rotor magnetic flux, the same conclusion can be obtained for the induced voltage estimated value.

As apparent from the above description, in accordance with the invention of claim 2, the rotor parameter is obtained by intentionally using a corresponding value having a predetermined deviation from the true value as the motor parameter for phase estimation. An effect is obtained that the phase estimation value of the magnetic flux or the induced voltage can be converged on a specific trajectory determined by the designer.

また、所定のトルクを発生しつつ力率１を達成するには、固定子電流は同期ｄｑ座標系上で評価した場合、次の楕円軌道上に存在しなければならない。

また、固定子巻線抵抗による銅損と磁気抵抗による鉄損とからなる総合損失を最小化する軌道は、（２０）式と（２１）式の中間の軌道となる。総合損失を最小化するには、固定子電流を本軌道上に存在するようにしなければならない。Then, the effect | action of Claim 3 of this invention is demonstrated. In order to minimize the copper loss due to the stator winding resistance while generating a predetermined torque, the stator current must be on the next hyperbolic trajectory when evaluated on the synchronous dq coordinate system.

Further, in order to achieve a power factor of 1 while generating a predetermined torque, the stator current must be present on the next elliptical orbit when evaluated on the synchronous dq coordinate system.

Further, the track that minimizes the total loss including the copper loss due to the stator winding resistance and the iron loss due to the magnetic resistance is an intermediate track between the equations (20) and (21). In order to minimize the total loss, the stator current must be on the orbit.

According to the invention of claim 3, the trajectory on the synchronous dq coordinate system to which the estimated value of the rotor magnetic flux or the induced voltage should converge is a hyperbolic trajectory, a parabolic trajectory, or an elliptical trajectory. The stator current whose phase is determined based on the estimated value of the child magnetic flux or the induced voltage is also present on the hyperbolic or parabolic or elliptical orbit. As is apparent from the above description, the stator current effectively achieves the minimum copper loss, the minimum total loss, or the power factor of 1. As apparent from the above, according to the invention of claim 3, the command converter in which current control for effectively achieving the minimum copper loss, the minimum total loss, or the power factor of 1 is indispensable in the prior art. The effect of being able to accomplish without using is obtained.

と選定する場合には、固定子電流は、（１９）式より、総合損失を準最小化する次の放物線上に存在する。

また、モデルマッチング形推定器に利用するｑ軸インダクタンス対応値を常時ゼロに選定する場合には、すわわち、

と選定する場合には、固定子電流は、（１９）式より、力率１を達成する（２１）式の楕円軌道上に存在する。更には、モデルマッチング形推定器に利用するｑ軸インダクタンス対応値を次の（２３）式に示した範囲に選定する場合には、（１９）式より、固定子電流は準最小銅損を達成する軌道上に存在する。

図４に、ｑ軸インダクタンス対応値を（２２）式に選定した場合の１例を示した。同図では、参考のため、ｑ軸インダクタンス対応値を（２４）、（２５）式に選定した場合の軌道も破線で示している。同図では、回転子磁束推定値の位相をγ軸に選定した制御γδ座標系で評価した場合、固定子電流はδ軸上に存在するものとしている。In addition, if a supplementary explanation is given for reference, when the q-axis inductance corresponding value used for the model matching estimator is selected as the true d-axis inductance value, that is,

In this case, the stator current is present on the next parabola that quasi-minimizes the total loss according to equation (19).

When the q-axis inductance corresponding value used for the model matching type estimator is always set to zero, that is,

Is selected, the stator current exists on the elliptical orbit of equation (21) that achieves a power factor of 1 from equation (19). Furthermore, when the q-axis inductance corresponding value used for the model matching type estimator is selected within the range shown in the following equation (23), the stator current achieves a quasi-minimum copper loss from the equation (19). Exist on orbit.

FIG. 4 shows an example when the q-axis inductance corresponding value is selected in the equation (22). In the figure, for reference, the trajectory when the q-axis inductance corresponding value is selected by the equations (24) and (25) is also indicated by broken lines. In the figure, when the phase of the rotor magnetic flux estimated value is evaluated in the control γδ coordinate system selected for the γ axis, the stator current is assumed to exist on the δ axis.

Then, the effect | action of Claim 4 of this invention is demonstrated. There can be various methods as an estimator for estimating the rotor magnetic flux or the induced voltage in a model matching manner using the signals of the stator voltage and the stator current. As the model matching type estimator, the state observer and the disturbance observer have been most elucidated for the stability estimation. As a result, according to the invention of claim 4, in a form in which the conventional analysis result can be directly used, in other words, the rotor magnetic flux estimated value or the induced voltage estimated value on the specific trajectory is more stably. The effect of being able to estimate is obtained.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 5 shows a basic structure of an embodiment of a drive control apparatus in which the drive control method of the present invention is applied to a permanent magnet synchronous motor. The difference of this apparatus from the conventional drive control apparatus shown in FIG. The second difference is that there is no command converter, the output of the speed controller 11 directly becomes the δ-axis current command value, and the γ-axis current command value is always set to zero. . In FIG. 5, the stator voltage and current signals present on the left side of the vector rotator are signals on the control γδ coordinate system. In order to clarify this fact, γ and δ are added to the foot marks of these signals. On the other hand, in FIG. 8, the stator voltage and current signals present on the left side of the vector rotator are signals on the synchronous dq coordinate system. Special attention should be paid to this difference. Regarding other components, there is no difference between the conventional drive control device and this device.

The core of the present invention that does not require a command converter resides in the phase determiner 2. FIG. 6 is a partial reconfiguration of the phase determiner shown by document (1) according to the invention. The phase determiner 2 includes a minimum dimension state observer 2a and a phase synchronizer 2b.

る。文献（１）では、固定子誘導磁束算定ブッロクに利用するｑ軸インダクタンス値としては、ｑ軸インダクタンス真値あるいはこれに準じた値を利用することを前提としている。これに対して、本発明によるブッロク２ａ−１では、真値に対して所定の偏差を有するｑ軸インダクタンス対応値を利用している。すなわち、ｑ軸インダクタンス対応値としては、（２２）式、または（２４）式、または（２５）式のものを利用している。最小次元状態オブザーバ２ａの構成に関しては、他に相違はない。結果的には、本発明による最小次元状態オブザーバ２ａは、従来の最小次元状態オブザーバにおけるｑ軸インダクタンス真値に代わって、意図的に、真値に対して所定の偏差を有するｑ軸インダクタンス対応値を使用するだけで構成できる。FIG. 7 shows the structure of an embodiment of a minimum dimension state observer in which the rotor magnetic flux is estimated as the minimum dimension state observer 2a. The basic structure is the same as document (1). For this reason, the description regarding the principle of this basic structure is abbreviate | omitted. The only difference between the minimum dimension state observer according to the literature (1) and the minimum dimension state observer 2a according to the present invention is as follows.

The Document (1) assumes that the q-axis inductance value used in the stator-induced magnetic flux calculation block is a true q-axis inductance value or a value based thereon. On the other hand, the block 2a-1 according to the present invention uses a q-axis inductance corresponding value having a predetermined deviation from the true value. That is, as the q-axis inductance corresponding value, the value of the formula (22), the formula (24), or the formula (25) is used. There is no other difference regarding the configuration of the minimum dimension state observer 2a. As a result, the minimum dimension state observer 2a according to the present invention intentionally has a q-axis inductance corresponding value having a predetermined deviation from the true value, instead of the q-axis inductance true value in the conventional minimum dimension state observer. Can be configured simply by using

No motor parameters are used in the phase synchronizer 2b associated with the minimum dimension state observer. For this reason, the configuration of the phase synchronizer 2b is completely the same as the configuration according to the document (1) and the configuration according to the present invention. As is clear from the above description, in the phase determiner using the minimum dimensional state observer for estimating the conventional rotor magnetic flux, instead of the q-axis inductance true value, the true value is intentionally determined. The phase determiner according to the present invention can be configured only by using a q-axis inductance corresponding value having a deviation of.

FIG. 5 is a configuration example of a phase determiner that estimates a rotor magnetic flux using a minimum dimensional state observer configured on a control γδ coordinate system and determines a phase of a vector rotator or the like as an estimator. The same thing can be said in the configuration of “the phase determiner, in place of the q-axis inductance true value, the present invention can be obtained by intentionally using a q-axis inductance corresponding value having a predetermined deviation from the true value. The phase determiner can be configured ". The other configurations mentioned here include a configuration on a fixed αβ coordinate system, a configuration using a non-minimum dimensional state observer, a configuration via induced voltage estimation, a configuration using a disturbance observer, and the like. That is, in the phase determiner configured to estimate the rotor magnetic flux or the induced voltage by the model matching type estimator and determine the phase of the vector rotator or the like through these estimated values, this is configured. Regardless of the coordinate system, the phase determiner according to the present invention can be configured only by intentionally using a q-axis inductance corresponding value having a predetermined deviation from the true value instead of the q-axis inductance true value. Can do. When using the phase determiner according to the present invention, as shown in the embodiment of FIG. 5, the command converter is unnecessary, the γ-axis current command value is set to zero, and the δ-axis current command value is set to The speed controller output will be selected. With this configuration, the operation of the present invention described above can be obtained.

Several more specific example embodiments are illustrated. In the phase determiner shown in the literature (1) to (9), which is configured to use the same true value as the q-axis inductance, intentionally, simply formally with respect to the true value. The phase determiner according to the present invention can be configured only by using a q-axis inductance corresponding value having a predetermined deviation. When using the phase determiner according to the present invention, as shown in the embodiment of FIG. 5, the command converter is unnecessary, the γ-axis current command value is set to zero, and the δ-axis current command value is set to The speed controller output will be selected.

As is clear from these specific embodiments using the literatures (1) to (9), the configuration of the phase determiner according to the present invention has no calculation load compared to the configuration of the conventional phase determiner. Does not increase. The calculation load required for the configuration of the phase determiner is completely the same. On the other hand, since the command converter is unnecessary in the case of the present invention, all the calculation loads for the command converter configuration required by the conventional drive control technique are eliminated.

FIG. 5 shows an embodiment of the speed control mode, but it is also possible to perform a quasi-torque control mode according to the torque control mode. In the quasi-torque control mode, the δ-axis current command may be directly given from the outside.

In the specific embodiment example using the documents (1) to (9) including the embodiment example shown in FIG. 5, the current control is performed in a feedback manner. As mentioned in the description of the operation of the invention, the present invention can also be applied to the case where current control is performed in a feed-forward manner, and the same operation and effect as in the case of feedback current control can be obtained.

In the above embodiment example, the true value is used for the value of the stator winding resistance used in the phase determiner. As apparent from the equation (17) used for the description of the operation of the present invention, the stator current and the rotor magnetic flux are also obtained by utilizing the corresponding values with intentional deviations for the stator winding resistance. It should be pointed out that it is possible for the estimated value and the induced voltage estimated value to exist on a specific trajectory. In this case, the possible trajectory of the stator current is expressed by the following equation if the condition of i _γ = 0 is given to the equation (17).

Although the phase determiner according to the present invention can be implemented in an analog manner, it is preferably constructed digitally in view of the remarkable progress of recent digital technology. Although the digital configuration includes a hardware configuration and a software configuration, as will be apparent to those skilled in the art, any of the present invention can be configured. The present invention has been described in detail with reference to various embodiments using various drawings.

The invention's effect

As is clear from the above description, the present invention has the following effects. According to the first aspect of the present invention, the stator current is not dependent on the command converter, which is indispensable in the prior art, as long as the trajectory entrusted to the designer is selected as a trajectory that achieves high efficiency. In both cases, the effect of being able to exist on a track that achieves high efficiency was obtained. As a result of this action, according to the first aspect of the present invention, there is an effect that the permanent magnet synchronous motor can be operated with high efficiency without using the command converter that is indispensable in the prior art. As a result, it is possible to obtain an effect that high-efficiency operation is possible with a lighter calculation load.

Next, the effect of the present invention of claim 2 will be described. According to the invention of claim 2, the phase estimation of the rotor magnetic flux or the induced voltage is actively performed by intentionally using a corresponding value having a predetermined deviation with respect to the true value as the motor parameter for phase estimation. The effect is that the value can be converged on a specific trajectory determined by the designer. As a result of this operation, the present invention that achieves the effect of claim 1 can be realized more easily.

Next, the effect of the present invention of claim 3 will be described. According to the third aspect of the present invention, there has been obtained an effect that current control for effectively achieving the minimum copper loss, the minimum total loss, or the power factor of 1 can be performed. As a result of this action, according to the invention of claim 3, there is obtained an effect that it is possible to perform an efficient operation that effectively achieves the minimum copper loss, the minimum total loss, or the power factor of 1. As a result, according to the invention of claim 3, the effect that the effect of the invention of claim 1 can be improved to a more useful level is obtained.

Next, the effect of the invention of claim 4 will be described. According to the invention of claim 4, the conventional analysis results relating to the state observer and the disturbance observer can be directly utilized, in other words, more stably, the estimated value of the rotor magnetic flux or the induced voltage on the specific trajectory. The effect that the estimated value can be estimated was obtained. As a result of this action, according to the invention of claim 4, the effect that the effect of claim 1 can be obtained stably can be obtained.

The figure which shows the example of 1 relationship of 3 coordinate systems and a rotor phase The figure which shows the example of 1 relationship of the vector component at the time of evaluating a single stator current by a synchronous dq coordinate system and a control (gamma) delta coordinate system. The figure which shows the example of the trajectory of the stator current corresponding to the trajectory of the rotor magnetic flux estimated value or the induced voltage estimated value obtained using the parameter corresponding value The figure which shows the example of a relationship between the parabolic trajectory of the stator current corresponding to the parabolic trajectory of the rotor magnetic flux estimated value or the induced voltage estimated value obtained using the parameter corresponding value and the control γδ coordinate system The block diagram which shows the basic composition of the drive control apparatus in one example of embodiment. The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment The block diagram which shows the basic composition of the minimum dimension state observer in one example of embodiment. Block diagram showing the basic configuration of a typical conventional drive control device

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Permanent magnet synchronous motor 2 Phase determiner 2a Minimum dimension state observer 2a-1 Stator induction magnetic flux calculation block 2b Phase synchronizer 3 Power converter 4 Current detector 5a 3 phase 2 phase converter 5b 2 phase 3 phase converter 6a Vector rotator 6b Vector rotator 7 Cosine sine signal generator 8 Current controller 9 Command converter 10 Speed controller 11 Machine speed estimator

Claims (4)

Used in the current control process that controls the stator current contributing to torque generation in a feedback or feed-forward manner using a vector rotator or a converter with a built-in vector rotator, and a vector rotator or a converter with a built-in vector rotator A permanent magnet synchronous motor drive control method having a phase determination step for determining a phase to be performed,
The rotor flux generated by only the rotor permanent magnet existing on the d-axis of the synchronous dq coordinate system or a simple differential value of the rotor magnetic flux itself and the induced voltage existing on the q-axis is expressed as a stator voltage and stator current signal. Is used to presumptively determine to exist on a different orbit from the d-axis and q-axis on the synchronous dq coordinate system, and using the phase of the estimated rotor magnetic flux or the estimated voltage on the orbit. Configure the phase determination process to determine the phase to be used for the vector rotator or the vector rotator built-in transducer and fix it on the control γδ coordinate system specified by the vector rotator or the vector rotator built-in transducer A drive control method for a permanent magnet synchronous motor, characterized in that the current control step is configured such that one of the γ-axis and δ-axis currents becomes zero when the child current is evaluated. In the estimation determination of the rotor magnetic flux or the induced voltage, the estimated value of the rotor magnetic flux or the induced voltage of the rotor voltage is obtained by using at least one motor parameter corresponding value having a deviation from the true value of the motor parameter. 2. The drive control method for a permanent magnet synchronous motor according to claim 1, wherein the estimated value is present on a different track from the d-axis and the q-axis on the synchronous dq coordinate system. 2. The drive control method for a permanent magnet synchronous motor according to claim 1, wherein the trajectory on the synchronous dq coordinate system is a hyperbolic trajectory, a parabolic trajectory, or an elliptical trajectory. 2. The permanent magnet synchronous motor according to claim 1, wherein the rotor magnetic flux or the induced voltage is estimated using a state observer or a disturbance observer driven by a signal of a stator current and a stator voltage. Drive control method.

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