JP5991577B2 - Drive control method for permanent magnet synchronous motor - Google Patents

Description

  The present invention relates to a sensorless drive control method for a synchronous motor having a permanent magnet in a rotor, that is, a permanent magnet synchronous motor, and in particular, sensorless in a high speed range subject to voltage limitation caused by a bus voltage (link voltage) of a power converter (inverter). The present invention relates to a drive control method. The target synchronous motor may be any number of permanent magnets mounted on the rotor, and may be salient or non-salient.

  In order to drive a permanent magnet synchronous motor at high speed while achieving as much efficiency as possible, the stator current that contributes to torque generation is set to the phase of the rotor N pole (hereinafter abbreviated as rotor phase). It is necessary to have a predetermined phase difference in consideration of voltage limitation. For this reason, it is necessary to know the rotor phase evaluated on the αβ fixed coordinate system in which the α axis is selected as the center of the stator U-phase winding. The simplest method for detecting the rotor phase is to mount a position sensor such as an encoder or resolver on the rotor. However, mounting the position sensor is problematic in terms of reliability, cost, etc. The rotor phase is estimated by using the stator voltage and current signals. The rotor phase is estimated by estimating the rotor magnetic flux, induced voltage, or extended induced voltage (hereinafter abbreviated as rotor magnetic flux) having the rotor phase information, and extracting the rotor phase information from the estimated value. It is done by.

  The estimation of the rotor magnetic flux etc. can be performed on the αβ fixed coordinate system with α axis and β axis determined according to the stator winding, or γδ quasi-synchronous coordinate system aiming to synchronize with the rotor phase without phase difference It can also be done above. A coordinate system synchronized with the rotor phase without a phase difference is called a dq synchronous coordinate system. The γδ quasi-synchronous coordinate system is substantially equivalent to the dq-synchronous coordinate system when synchronization is completed. On the other hand, it is possible to consider a γδ coordinate system that does not aim for synchronization without a phase difference with respect to the rotor phase. In the present invention, this type of γδ coordinate system is called a γδ control coordinate system.

  FIG. 13 is a block diagram schematically showing a typical example when a drive control method using a phase determiner that plays a role of rotor phase estimation is implemented for a permanent magnet synchronous motor and is mounted on the method. It is a thing. 1 is a permanent magnet synchronous motor, 2 is a phase determiner, 3 is a power converter, 4 is a current detector, 5a and 5b are 3 phase 2 phase converters, 2 phase 3 phase converters, 6a , 6b are vector rotators, 7 is a cosine sine signal generator, 8 is a current controller, 9 is a command converter, 10 is a speed controller, and 11 is a machine speed estimator. In this figure, a 2 × 1 vector signal is represented by one thick signal line to ensure simplicity. The following block diagram expression follows this.

  In particular, the phase determiner 2 obtains an estimated value of the rotor phase obtained on the αβ fixed coordinate system by obtaining an equivalent value (measured value, command value, estimated value, etc.) of the stator voltage and current as an input. The estimated value of the rotor electrical speed is output. The cosine sine signal generator 7 converts the rotor phase estimation value into a cosine / sine signal and transmits it to the vector rotators 6a and 6b. The electrical speed estimated value is divided by the pole pair number Np in the mechanical speed estimator 11 and converted into a mechanical speed estimated value. 5, 5a, 5b, 6a, 6b, 7, and 8 capture the stator current contributing to torque generation as a vector signal on the dq synchronous coordinate system, and use the d-axis and q-axis components. A feedback-type current control process for controlling the shaft current command value so as to follow each axis is configured. Further, the phase determiner 2 constitutes a phase determination step.

  The three-phase stator current detected by the current detector 4 is converted into a two-phase current on the αβ fixed coordinate system by the three-phase two-phase converter 5a, and then the vector rotator 6a 2 in the dq synchronous coordinate system. It is converted into a phase current and sent to the current controller 8. The current controller 8 generates a two-phase voltage command value on the dq synchronous coordinate system so that the two-phase current on the dq synchronous coordinate system follows the current command value of each phase, and sends it to the vector rotator 6b. In 6b, the two-phase voltage command value on the dq synchronous coordinate system is converted into the two-phase voltage command value on the αβ fixed coordinate system and sent to the two-phase three-phase converter 5b. In 5 b, the two-phase signal is converted into a three-phase voltage command value and output as a command value to the power converter 3. The power converter 3 generates electric power according to the command value, applies it to the synchronous motor 1, and drives it. The two-phase current command value on the dq synchronous coordinate system at this time is obtained by converting the torque command value through the command converter 9. In FIG. 13, dq is added to the foot mark to clearly indicate that the stator voltage and current signals existing on the left side of the vector rotator are signals on the dq synchronous coordinate system. The vector rotators 6a and 6b may be configured integrally with the three-phase two-phase converter and the two-phase three-phase converters 5a and 5b, respectively. In the present invention, these integrated components are referred to as a vector rotator built-in converter.

  In this example of FIG. 13, since the speed control system is configured, the torque command value τ * is obtained as the output of the speed controller 10 that receives the speed command value and the speed estimated value. As is well known to those skilled in the art, the speed controller 10 and the machine speed estimator 11 are unnecessary when the control purpose is the generated torque and the speed control system is not configured. In this case, the torque command value is directly applied from the outside.

  Among the above-described components, devices particularly related to the present invention are the phase determiner 2 and the command converter 9. The phase determiner estimates the rotor magnetic flux and the like using the equivalent values of the stator voltage and current, and determines the phase to be used for the vector rotator and the like. In addition, the estimated electric speed is determined. Estimated values of rotor flux, etc. are generally obtained by using equivalent values of stator voltage and current as input signals to state observers, disturbance observers, D-factor filters, etc. (hereinafter simply referred to as “estimator”). Driving and getting. That is, the input signal to the phase determining step is generally a value corresponding to the voltage and current of the stator.

  If an estimator is configured, and the corresponding values of the stator voltage and current for driving the motor are input to the configured estimator and driven, the estimated rotor phase value can be obtained. As is well known to those skilled in the art, motor parameters are required in the estimator configuration. In particular, when the motor parameters used in the configuration of the estimator and the stator voltage and current signals as the drive signals are correct, the phase estimation value by the estimator converges to the true phase value. Actually, there is a slight difference between the motor parameter true value and the parameter value used for the estimator, but it is estimated that the parameter value used is as close to the true value as possible. This is a major premise for the use of the vessel.

  The electric motor is a torque generator and at the same time a mechanical energy converter, and the energy conversion efficiency of the electric motor is one of the main required specifications of the motor drive control. From the viewpoint of motor drive control, the issue of improving energy conversion efficiency is how to implement current control that minimizes losses such as copper loss while achieving the required torque generation, or the maximum power factor. It can be replaced with a technical problem of how to carry out current control for achieving the above. The command converter 9 plays the role of solving this problem. The command converter generates d-axis and q-axis current command values on a dq synchronous coordinate system that enables highly efficient driving from a torque command value. The d-axis current command value and the q-axis current command value at this time have a predetermined phase difference with respect to the rotor phase, and if the stator current is controlled according to the command value, the efficiency drive is Achieved.

  The amplitude (ie, magnitude, level) of the voltage that can be applied to the motor is limited by the bus voltage of the power converter. Therefore, torque generation by the electric motor needs to be performed within the voltage limit caused by the bus voltage. However, as the speed increases, the back electromotive force increases, and under the voltage limit, it becomes impossible to apply a voltage corresponding to the desired torque and current. In such a case, high-speed driving is achieved by controlling the current so as to obtain a negative d-axis current (generally referred to as “weakening magnetic flux control”). The command converter 9 also generates d-axis and q-axis current command values on the dq synchronous coordinate system that enable high-speed driving while pursuing efficiency from the torque command value. At this time, the d-axis current command value and the q-axis current command value have a predetermined phase difference in consideration of voltage limitation with respect to the rotor phase, and if the current is controlled according to the command value, High speed driving is achieved while maintaining the achievable efficiency.

  In general, d-axis current, q-axis current, generated torque, d-axis current, q-axis current command value generation from the torque command value on the dq synchronous coordinate system that achieves high-speed driving considering efficiency and voltage limitation, It is required to solve a nonlinear simultaneous equation describing the four-way relationship of voltage limitation. This solution is generally very difficult.

  Against this background, the phase of the stator current can be estimated presumably without solving complicated simultaneous simultaneous equations by limiting the drive to below the rated speed, which does not need to substantially consider the voltage limit. A direct method has been proposed. These typical methods are listed in Patent Documents 1 and 2 and Non-Patent Document 1 in the column of prior art documents.

一方、特許文献２及び非特許文献１では、位相決定器用インダクタンスＬｉ＾として、次の（２）式の範囲の正値を採用することを求めている

The characteristics of these representative methods are that the current control process is configured so that the γ-axis current is zero when the stator current is evaluated on the γδ control coordinate system specified by the vector rotator or the converter with built-in vector rotator. In addition, the phase determining process is characterized by using a positive value different from the true inductance as the phase determiner inductance corresponding to the δ-axis current. When the d-axis inductance is Ld and the q-axis inductance is Lq, Patent Document 1 requires that a positive value in the range of the following equation (1) is adopted as the phase determiner inductance Li ^.

On the other hand, Patent Document 2 and Non-Patent Document 1 require that a positive value in the range of the following equation (2) be adopted as the phase determiner inductance Li ^.

  The methods based on the proposals in Patent Documents 1 and 2 and Non-Patent Document 1 described above are applicable only at a low speed equal to or lower than the rated speed, and exhibit the expected performance. In other words, it cannot be applied to high-speed driving exceeding the rated speed. If it is applied, high-speed driving cannot be performed. In support of this fact, the inventors of Patent Document 1 and Non-Patent Document 1 that have proposed the use of Equation (1) as the inductance for the phase determiner are required to control to zero conventionally in order to achieve high-speed driving. It has been proposed to control the non-zero γ-axis current (see Non-Patent Document 2).

Tomitsu Hitoshi: “Motor Control Device”, Japanese Patent Application Laid-Open No. 2007-259686 (2006-6-28) Shinnaka Shinji: “Driving Control Method for Permanent Magnet Synchronous Motor”, JP2008-92781 (2006-9-30)

Shinnaka Shinji: “Vector control technology of permanent magnet synchronous motor, second volume (the essence of sensorless drive control)”, Denpa Shimbun (2008-12) Tomitsu Toshio and Kamiyama Kenji: “A Method for Estimating the Maximum Torque Control Axis of an Embedded Magnet Synchronous Motor at an Arbitrary Current Vector”, IEEJ Transactions D, vol. 131, no. 9, pp. 1141-1148 (2011-9)

  The present invention has been made under the above-mentioned background, and the object thereof can be achieved without requiring a command converter in the sensorless drive control of a permanent magnet synchronous motor (and thus can be operated with a lighter computational load). It is to provide a sensorless drive control method capable of achieving high speed drive while maintaining high efficiency.

  In order to achieve the above object, the invention of claim 1 is a current control for controlling a stator current contributing to torque generation in a feedback manner or a feed forward manner using a vector rotator or a converter incorporating a vector rotator. And a phase determination step for determining a phase to be used for the vector rotator or the vector rotator built-in converter, and a stator on the γδ control coordinate system specified by the vector rotator or the vector rotator built-in converter. The current control process is configured so that either the γ-axis or δ-axis current is substantially zero when the current is evaluated, and the phase determiner inductance corresponding to the non-zero other-axis current As a drive control method for a permanent magnet synchronous motor that configures the phase determination step using a value different from the true inductance. Characterized in that the said phase decision dexterity inductance utilized was to select a negative value.

  The invention according to claim 2 is the drive control method for the permanent magnet synchronous motor according to claim 1, wherein the negative value of the inductance for the phase determiner used for the configuration of the phase determination step is caused by the power converter bus voltage. It is characterized in that it can be varied in accordance with at least one of the equivalent value of the voltage limit, the equivalent value of the stator voltage, the equivalent value of the stator current, and the equivalent value of the rotor speed.

  The invention according to claim 3 is the drive control method for the permanent magnet synchronous motor according to claim 1, wherein the negative value of the inductance for the phase determiner used for the configuration of the phase determination step is at least partially calculated repeatedly. It is characterized by being used and defined.

  The invention of claim 4 is the drive control method of the permanent magnet synchronous motor according to claim 1 or claim 2 or claim 3, wherein the rotor magnetic flux or the induced voltage or the The phase determining step is configured by using a negative phase determiner inductance as an estimator capable of appropriately estimating the extended induced voltage.

  The invention of claim 5 is the drive control method for a permanent magnet synchronous motor according to claim 1, 2 or 3, wherein αβ determined according to the stator winding as an input signal to the phase determining step The phase determining step is configured by using at least a stator current equivalent value on a fixed coordinate system or a γδ control coordinate system.

ここに、ｉｄ、ｉｑは、固定子電流をｄｑ同期座標系上で評価した場合のｄ軸電流、ｑ軸電流であり、ｉδは、固定子電流をγδ制御座標系上で評価した場合のδ軸電流である。Φは、回転子磁束の強度（一定）である。また、Ｌｍは、次式で定義された鏡相インダクタンスである。

固定子電流は、γ軸電流が実質的にゼロになるように制御されている。換言するならば、２×１ベクトルしての固定子電流ｉ１は実質的にδ軸上に存在する（図１参照）。Hereinafter, the effect of the invention of claim 1 will be described clearly with reference to the drawings and mathematical expressions. According to the first aspect of the present invention, “a current control step of controlling the stator current contributing to torque generation in a feedback or feedforward manner using a vector rotator or a converter with a built-in vector rotator, and a vector rotator or A permanent magnet synchronous motor drive control method having a phase determining step for determining a phase to be used for a vector rotator built-in converter, particularly on a γδ control coordinate system designated by the vector rotator or the vector rotator built-in converter The current control process is configured so that the γ-axis current becomes zero when the stator current is evaluated at, and a value different from the true inductance is intentionally used as the phase determiner inductance Li ^ corresponding to the δ-axis current. “Think about the situation. Under this situation, the γδ control coordinate system is in contrast to the dq synchronous coordinate system.

Here, id and iq are d-axis current and q-axis current when the stator current is evaluated on the dq synchronous coordinate system, and iδ is δ when the stator current is evaluated on the γδ control coordinate system. It is a shaft current. Φ is the strength (constant) of the rotor magnetic flux. Lm is a mirror phase inductance defined by the following equation.

The stator current is controlled so that the γ-axis current becomes substantially zero. In other words, the stator current i1 as a 2 × 1 vector substantially exists on the δ axis (see FIG. 1).

  In high speed driving under practical circumstances where the bus voltage of the power converter is limited, the stator voltage to be applied must be kept below the voltage limit value due to the bus voltage. As is widely known by those skilled in the art, this maintenance can be achieved by controlling the d-axis current such that the d-axis current has a larger negative value (this type of control is generally a weak flux control). Called).

の位相差に関する理論限界値は、次式で与えられることがわかる。

The same effect as the d-axis current control described above can be obtained by bringing the stator current i1 as a 2 × 1 vector closer to the negative d-axis. The stator current i1 as a 2 × 1 vector is brought close to the negative d-axis for the time being, and the phase difference of the γδ control coordinate system with respect to the dq synchronous coordinate system.

It can be seen that the theoretical limit value for the phase difference is given by the following equation.

要基本式である次式を新たに得る。

The main point of the present invention is the determination of the phase determiner inductance Li ^. From this point of view, let us consider deriving and organizing Equation (3) with respect to the phase determiner inductance Li ^. At this time

The following formula which is a basic formula is newly obtained.

（５）式の条件で導出された（８）式の位相決定器用インダクタンスは、ｑ軸とδ軸（ｄ軸とγ軸）との位相差を±π／２とするための限界値を示している（図１参照）。When the expression (5) is used in the expression (6), the following expression is newly obtained as the phase determiner inductance Li ^ for the expression (5).

The inductance for the phase determiner in the equation (8) derived under the condition of the equation (5) indicates a limit value for setting the phase difference between the q axis and the δ axis (d axis and γ axis) to ± π / 2. (See FIG. 1).

（９）式は、「通常の永久磁石同期電動機においては、（８）式左辺の位相決定器用インダクタンスは負値を取る」ことを示している。すなわち、（８）、（９）式は、「電圧制限（特に、強い電圧制限）を考慮する必要のある高速駆動時に対応した位相決定器用インダクタンスは負値をとる」ことを示している。A normal permanent magnet synchronous motor is designed and manufactured so that the relationship of the following equation (9) is established with respect to the rated current Irat.

The expression (9) indicates that “in a normal permanent magnet synchronous motor, the inductance for the phase determiner on the left side of the expression (8) takes a negative value”. That is, Equations (8) and (9) indicate that “the inductance for the phase determiner corresponding to the time of high-speed driving that needs to consider voltage limitation (particularly strong voltage limitation) takes a negative value”.

  The invention of claim 1 requires that the phase determiner inductance be selected to be a negative value. The negative-value phase determiner inductance makes the δ axis of the γδ control coordinate system asymptotic to the negative d-axis, as is apparent from the above description. Considering that the stator current substantially exists on the δ axis, this asymptotic approach of the δ axis means asymptotic approach to the d axis on the negative side of the stator current. This means that flux-weakening control under voltage limitation is automatically achieved. As described above, according to the present invention of claim 1, which requires that the inductance for the phase determiner be selected as a negative value, the flux weakening control indispensable for the high-speed driving under the voltage limit is applied to the vector rotator or the vector rotator. The effect of being able to be realized automatically can be obtained through the determination of the phase used for the built-in converter.

て、位相決定器用インダクタンスを選定する場合には、位相決定器用インダクタスは固定子電流（あるいはこの相当値）に応じて、変更することが必要となる。Next, the effect of the invention of claim 2 will be described. The value indicated by equation (8) is the phase determiner input.

Thus, when selecting the phase determiner inductance, it is necessary to change the phase determiner inductance according to the stator current (or its equivalent value).

ここに、ｃｖは電力変換器のバス電圧に依存して定まる電圧制限値であり、ω２ｎは回転子の電気速度である。達成可能な効率と高速駆動を考慮した固定子電流は、（１０）式で規定された電圧制限楕円上に存在しなければならない。このときのｄ軸電流は、次式を満足する必要がある。

By the way, the following voltage limit ellipse is formulated as a voltage limit at the time of high-speed driving (see Non-Patent Document 1).

Here, cv is a voltage limit value determined depending on the bus voltage of the power converter, and ω2n is the electric speed of the rotor. The stator current considering the achievable efficiency and high speed driving must be on the voltage limiting ellipse defined by equation (10). The d-axis current at this time needs to satisfy the following equation.

（１２）式を（６）式に代入することにより、位相決定器用インダクタンスＬｉ＾に関して次の関係式を得る。

（１３）式は、位相決定器用インダクタンスＬｉ＾の一つの選定ルールを与えるものである。（１３）式は、固定子電流、回転子速度、電圧制限値の３者を利用している。換言するならば、位相決定器用インダクタンスの選定を（１３）式に従う場合には、固定子電流、回転子速度、電圧制限値の３者に同時に応じて、位相決定器用インダクタンスを変更することになる。If the equation (11) is used in FIG. 1, the following equation is obtained from these.

By substituting the equation (12) into the equation (6), the following relational expression is obtained for the phase determiner inductance Li ^.

Equation (13) gives one selection rule for the phase determiner inductance Li ^. Equation (13) uses three elements: stator current, rotor speed, and voltage limit value. In other words, when the inductance of the phase determiner is selected according to the equation (13), the inductance for the phase determiner is changed simultaneously according to the three of the stator current, the rotor speed, and the voltage limit value. .

  The voltage limitation that prevents high-speed driving is originally caused because the amplitude of the stator voltage to be applied exceeds the voltage limit value caused by the bus voltage of the power converter. Therefore, when the amplitude of the stator voltage exceeds the voltage limit value, the phase determiner inductance Li may be gradually reduced. The upper limit value (positive value) of the inductance for the phase determiner is equal to the in-phase inductance Li as shown by the equations (1) and (2) even in the case of driving in a low speed region where there is substantially no voltage limitation. Never exceed. On the other hand, the lower limit (negative value) of the phase determiner inductance is given by the equation (8) newly presented by the present invention. Therefore, in a situation where the amplitude of the stator voltage takes a value close to the voltage limit value, the phase determiner inductance may be selected so as to decrease sequentially from the upper limit value to the lower limit value. In such a selection, the phase determiner inductance is changed according to the voltage limit value and the stator voltage.

  According to the second aspect of the present invention, the inductance for the phase determiner includes the equivalent value of the voltage limit caused by the power converter bus voltage, the equivalent value of the stator voltage, the equivalent value of the stator current, and the equivalent value of the rotor speed. Since it is made variable according to at least one, the above-described exemplified method for selecting the inductance for the phase determiner can be immediately used. As a result, the effect that the inductance for the phase determiner can be selected rationally or simply is obtained. Furthermore, as a result, the effect of enhancing the usefulness of the invention of claim 1 can be obtained.

連続的に変化することになる。ひいては、本発明が新規に提示した（６）式が示すように、位相決定器用インダクタンスもＬｉ＾も、基本的に連続的に変化することになる。位相決定器用インダクタンスが連続的に変化する場合、繰り返し計算による算定が可能となる（この実例は、実施形態例の欄で詳しく説明する）。繰り返し計算によれば、比較的低い計算量で、駆動状況（印加電圧、電圧制限、電流等の状況）に合致した位相決定器用インダクタンスが決定できるようになる。Next, the effect of the invention of claim 3 will be described. The rotational speed of the motor having inertia must be continuously changed. This characteristic holds regardless of the speed. That is, this characteristic is established even during high-speed driving in which voltage limitation needs to be considered. The speed at high speed continuously changes

It will change continuously. As a result, as the equation (6) newly presented by the present invention shows, both the phase determiner inductance and Li ^ basically change continuously. When the phase determiner inductance continuously changes, it is possible to perform calculation by repeated calculation (this example will be described in detail in the section of the embodiment). According to the iterative calculation, it becomes possible to determine the inductance for the phase determiner that matches the driving condition (applied voltage, voltage limit, current, etc.) with a relatively low calculation amount.

  According to the invention of claim 3, the negative value of the inductance for the phase determiner is determined at least partly by repeated calculation. As is apparent from the above description, according to the present invention, it is possible to determine a phase determiner inductance that matches a driving situation (applied voltage, voltage limit, current, etc.) with a small amount of calculation. Is obtained. As a result, the effect of enhancing the usefulness of the invention of claim 1 can be obtained.

  Next, the effect of the invention of claim 4 will be described. Consider a case in which the rotor magnetic flux, the induced voltage, or the extended induced voltage is estimated using an estimator (state observer, disturbance observer, or D-factor filter). When true inductance is used for these estimators, these estimators correctly estimate the rotor magnetic flux and the like. Moreover, the characteristics of these estimators have already been well analyzed and have the characteristic of equation (6). The most important basic formula relating to the inventions of claim 1, claim 2, and claim 3 is the formula (6) newly presented by the present invention.

  The invention of claim 4 is different from a true stator inductance (positive value) in an estimator that can appropriately estimate a rotor magnetic flux, an induced voltage, or an extended induced voltage when a true stator inductance is used. It is requested to configure a phase determination step using a negative value phase-inductor inductance. According to the invention of claim 4, a well-characterized estimator can be used immediately, which is the most important basis for the inventions of claims 1, 2 and 3 (6). As a result, it is possible to guarantee that the expression is established, and as a result, it is possible to obtain the effect that the effects of the inventions of claims 1, 2, and 3 can be obtained immediately in an effective form.

  Next, the effect of the invention of claim 5 will be described. Signals on various coordinate systems can be used as signals (especially stator current equivalent values) used in the phase determination process. That is, it is also possible to configure using a two-phase signal on a two-axis orthogonal coordinate system and a three-phase signal on a three-axis coordinate system. As the two-phase signal, a signal on the αβ fixed coordinate system, on the dq synchronous coordinate system, on the γδ control coordinate system, or on another two-axis orthogonal coordinate system can be considered. Among these, the signal that can perform the phase determination step in the invention of claim 1 or claim 2 or claim 3 most simply and simply uses a two-phase signal on an αβ fixed coordinate system or a γδ control coordinate system. Is the case.

  The invention of claim 5 requires that the phase determination step is configured using at least a stator current equivalent value on the αβ fixed coordinate system or the γδ control coordinate system as a signal for the phase determiner. Therefore, as is apparent from the above description, according to the invention of claim 5, the effect that the phase determining step can be most easily performed from the viewpoint of the used signal can be obtained. 2. The effect that the effect of the invention of claim 3 can be most easily obtained from the viewpoint of the use signal is obtained.

  The figure which shows the phase difference example of dq synchronous coordinate system and (gamma) delta control coordinate system, and one related example of the vector component at the time of evaluating a single stator current on both coordinate systems   The block diagram which shows the basic composition of the drive control apparatus in one example of embodiment.   The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment   The block diagram which shows the basic composition of the estimator with a variable inductance in the example of 1 embodiment.   The block diagram which shows the basic composition of the estimator part in one example of embodiment.   The block diagram which shows the basic composition of the inductance selection part in the example of 1 embodiment   The block diagram which shows the basic composition of the inductance selection part in the example of 1 embodiment   The block diagram which shows the basic composition of the inductance selection part in the example of 1 embodiment   The block diagram which shows the basic composition of the inductance selection part in the example of 1 embodiment   The block diagram which shows the basic composition of the inductance selection part in the example of 1 embodiment   The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment   The block diagram which shows the basic composition of the estimator part in one example of embodiment.   Block diagram showing the basic configuration of a typical conventional drive control device

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  FIG. 2 shows a basic structure of an embodiment of a drive control apparatus in which the drive control method of the present invention is applied to a permanent magnet synchronous motor. The difference of this apparatus from the conventional drive control apparatus shown in FIG. The second difference is that there is no command converter, the output of the speed controller 10 directly becomes the δ-axis current command value, and the γ-axis current command value is always set to zero. . In FIG. 2, the stator voltage and current signals present on the left side of the vector rotator are signals on the γδ control coordinate system. In order to clarify this fact, γ and δ are added to the foot marks of these signals. On the other hand, in FIG. 13, the stator voltage and current signals present on the left side of the vector rotator are signals on the dq synchronous coordinate system. Special attention should be paid to this difference. Regarding other components, there is no difference between the conventional drive control device and this device.

  The core of FIG. 2 that does not require a command converter is the phase determiner 2. FIG. 3 shows the phase determiner shown in Non-Patent Document 1 improved and reconfigured according to the present invention. The external difference in the phase determiner from the conventional one is that the phase determiner obtains the bus voltage vdc from the power converter. The phase determiner 2 internally includes an estimator with a variable inductance (a state observer having a variable inductance function, a disturbance observer, a D-factor filter, etc.) 2a and a phase synchronizer. Since the configuration of the phase synchronizer is basically the same as that described in detail in Non-Patent Document 1, this description is omitted.

  FIG. 4 shows the internal structure of the estimator 2a with variable inductance, which is composed of an inductance selection unit 2a-1 and an estimator unit 2a-2. The input signal to the estimator 2a with variable inductance includes an actual stator current value i1 as a stator current equivalent value, a stator voltage command value v1 * as a stator voltage equivalent value, and a power converter as a voltage limit equivalent value. There are five types: a measured bus voltage vdc, an estimated rotational speed value ω2n ^ as a rotor speed equivalent value, and a speed ωγ in the γδ control coordinate system. The rotor speed estimated value and the γδ coordinate system speed are reused as generated inside the phase determiner (more precisely, the phase synchronizer).

  FIG. 5 shows the structure of an embodiment using the minimum dimensional state observer originally developed to estimate the rotor magnetic flux as the estimator unit 2a-2. The basic structure is the same as Non-Patent Document 1. For this reason, the detailed description regarding this basic structure is abbreviate | omitted. The only difference between the minimum dimension state observer according to Non-Patent Document 1 and the estimator unit 2a-2 according to the present invention is the introduction of the phase determiner inductance Li ^ in the stator reaction magnetic flux calculation block 2a-2a. In Non-Patent Document 1, it is assumed that the true value of the d-axis and q-axis inductances or a value based thereon is used as the inductance value used for the stator reaction magnetic flux calculation block. On the other hand, the block 2a-2a according to the present invention intentionally uses a phase determiner inductance that can take a negative value different from the true value with respect to the inductance of both the d-axis and the q-axis. . There is no other difference regarding the configuration of the minimum dimension state observer 2a. In the present embodiment, since the γ-axis current is substantially zero-controlled, the inductance corresponding to the γ-axis may be arbitrary. As the inductance corresponding to the δ axis (that is, the original phase determiner inductance), the inductance obtained from the inductance selection unit (which can take a negative value) is used.

  In the embodiment using FIG. 5, the minimum dimension state observer originally developed to estimate the rotor magnetic flux is used as the estimator unit 2a-2. Instead, the rotor magnetic flux is estimated. It is also possible to use a same-dimensional state observer and D-factor filter that have been developed. Further, a developed minimum dimension state observer, disturbance observer, and D-factor filter that should originally estimate the induced voltage and the extended induced voltage may be used. The main point in using these is to use a negative phase determiner inductance instead of the true value (positive value) of the q-axis inductance as the inductance value related to the stator reaction magnetic flux. Since the same-dimensional state observer, disturbance observer, and D-factor filter are described in detail in Non-Patent Document 1, further explanation is omitted.

  Subsequently, in addition to the invention of claim 1, an inductance selecting unit 2 a-1 configured based on the invention of claim 2 will be described. First, FIG. 6 shows a method for selecting the inductance for the phase determiner based on the equation (13). If the power converter bus voltage vdc is obtained, the voltage limit value cv is calculated therefrom. For simplicity, as shown in FIG. The voltage limit value is further divided by the estimated speed value to generate a signal cv / ω2n ^. This value and the absolute value of the δ-axis current are used in the third equation of equation (11) to generate the signal β. If the generated signal β is used in the equation (13), the desired phase determiner inductance Li ^ can be obtained. In the present embodiment, the bus voltage of the power converter, the estimated rotor speed, and the stator current (δ-axis current) are used as input signals for obtaining the phase determiner inductance. In other words, the stator voltage equivalent value is not used.

  When the signal β is generated for each control period according to the third expression of the expression (11), it is required to perform a square root calculation with a large calculation load for each control period. In order to avoid an increase in the computation load, a two-dimensional table using the signal cv / ω2n ^ and the δ-axis current as reference signals is created in advance, and the signal β is output using this table. That's fine. The two-dimensional table may be created at an appropriate signal interval, and the output signal β for the reference signal within the interval may be generated by a function approximation method.

制限を満足するか否かに応じて、繰返し的に算定することを考える。Next, an inductance selection unit 2a-1 configured based on the present invention of claim 3 in addition to the inventions of claims 1 and 2 will be described. Expression (6), which is the most important basic expression relating to the present invention, can be rewritten as the following expression (14).

Consider repeatedly calculating according to whether the limit is satisfied or not.

（１５）式では、制御周期をＴｓで表現するとき、時刻ｔ＝ｋＴｓにおける信号を簡単に（ｋ）で表現している（以降では、同様な表現法を採用する）。（１５ａ）式では、印加すべき固定子電圧を示す固定子電圧指令値が電圧制限値以下ならば、ｕ（ｋ）＝０をセットし、固定子電圧指令値が電圧制限値を超える場合には、ｕ（ｋ）＝１をセットする。これを用いて、（１５ｂ）式に従い、信号ｘ’（ｋ）を繰返し計算により算定する。繰返し計算により算定された信号ｘ’（ｋ）を（１５ｃ）式に従い利用して、信号ｘ（ｋ）を決定する。信号ｘ（ｋ）を（１５ｄ）式に従い用い、Ｌｉ＾（ｋ）を定める。One concrete realization can be given by the following equation if iterative calculation is performed every control period.

In the expression (15), when the control cycle is expressed by Ts, the signal at the time t = kTs is simply expressed by (k) (hereinafter, the same expression method is adopted). In the equation (15a), if the stator voltage command value indicating the stator voltage to be applied is equal to or less than the voltage limit value, u (k) = 0 is set, and the stator voltage command value exceeds the voltage limit value. Sets u (k) = 1. Using this, the signal x ′ (k) is calculated by repeated calculation according to the equation (15b). A signal x (k) is determined by using the signal x ′ (k) calculated by iterative calculation according to the equation (15c). Li ^ (k) is determined using the signal x (k) according to the equation (15d).

If the method of selecting equation (15) is followed, the following relationship is guaranteed.

  The contents indicated by the equation (15) are drawn in FIG. As is apparent from the figure, in this embodiment, as input signals for obtaining the phase determiner inductance, the bus voltage of the power converter, the stator voltage command value (stator voltage equivalent value), the stator current (δ axis) Current). That is, the rotor speed equivalent value is not used.

図８に（１７）式の動作の様子を描画した。（１７）式の動作説明は（１５）式と同様であり、（１５）式の動作を理解した当業者は、容易に（１７）式の動作を理解できると思われるので、この説明は省略する。As another concrete realization based on the equation (14), if iterative calculation is performed every control cycle, it can be given by the following equation.

FIG. 8 shows the behavior of the expression (17). The operation of equation (17) is the same as that of equation (15), and those skilled in the art who understand the operation of equation (15) will be able to easily understand the operation of equation (17). To do.

図９に（１８）式の動作の様子を描画した。簡略化した場合にも、（１５）式の位相決定器用インダクタンスの繰返し計算による算定方法の特性である（１６）式と同様な次式の関係が成立する。

If the formula (8) obtained from the formula (6), which is an important basic formula, is used, the calculation method of the phase determiner inductance in the formula (15) can be simplified as follows. is there.

FIG. 9 shows the behavior of the equation (18). Even in the case of simplification, the relationship of the following equation similar to the equation (16) that is the characteristic of the calculation method by the repeated calculation of the phase determiner inductance of the equation (15) is established.

図１０に（２０）式の動作の様子を描画した。Instead of the equation (18), the calculation method by repetitive calculation of the phase determiner inductance can be simplified as follows.

FIG. 10 shows the behavior of the equation (20).

  Consider the case where the phase determiner is configured on an αβ fixed coordinate system. A typical configuration example in this case is shown in FIG. FIG. 11 shows the phase determiner shown in Non-Patent Document 1 improved and reconfigured according to the present invention. The external difference in the phase determiner from the conventional one is that the phase determiner obtains the bus voltage vdc from the power converter. Internally, the phase determiner 2 includes an estimator with variable inductance (a state observer having a variable inductance function, a disturbance observer, a D-factor filter, etc.) 2a and an electric speed estimator. Since the configuration of the electric speed estimator is basically the same as that described in detail in Non-Patent Document 1, this description is omitted.

  The internal structure of the estimator 2a with variable inductance used for the phase determiner 2 configured on the αβ fixed coordinate system is the same as that of the γδ control coordinate system shown in FIG. In other words, this is composed of an inductance selection unit 2a-1 and an estimator unit 2a-2. The input signal to the estimator 2a with variable inductance includes an actual stator current value i1 as a stator current equivalent value, a stator voltage command value v1 * as a stator voltage equivalent value, and a power converter as a voltage limit equivalent value. There are four types of bus voltage measured values vdc and estimated rotational speed values ω2n ^ as rotor speed equivalent values. As the rotor speed estimation value, a value generated inside the phase determiner (more precisely, an electric speed estimator) is reused. It should be noted that the stator current equivalent value and the stator voltage equivalent value in this embodiment are on the αβ fixed coordinate system.

  FIG. 12 shows the structure of an embodiment using a primary disturbance observer originally developed to estimate an induced voltage or an extended induced voltage as the estimator unit 2a-2. The basic structure is the same as Non-Patent Document 1. For this reason, the detailed description regarding this basic structure is abbreviate | omitted. The only difference between the primary disturbance observer according to Non-Patent Document 1 and the estimator section 2a-2 according to the present invention is the introduction of the phase determiner inductance Li ^ in the stator reaction magnetic flux calculation block 2a-2a. In Non-Patent Document 1, it is assumed that the inductance value used for the stator reaction magnetic flux calculation block is a true value of the stator inductance or a value based thereon. On the other hand, the block 2a-2a according to the present invention intentionally uses a negative estimator inductor that is different from the true value.

  In the embodiment using FIG. 12, the primary disturbance observer originally developed to estimate the induced voltage or the extended induced voltage is used as the estimator unit 2 a-2, but instead of this, the induced voltage is used. Or you may use the minimum dimension state observer and D factor filter developed in order to estimate an expansion induced voltage. Further, a developed minimum dimensional state observer, a same dimensional state observer, and a D-factor filter that should originally estimate the rotor magnetic flux may be used. The main point in using these is to use a negative phase-determiner inductance instead of the true value (positive value) of the stator inductance as the inductance value related to the stator reaction magnetic flux. Since the minimum dimension state observer, the same dimension state observer, and the D-factor filter are described in detail in Non-Patent Document 1, further description thereof is omitted.

  The configuration of the inductance selection unit on the αβ fixed coordinate system is the same as that on the γδ control coordinate system (see FIGS. 6 to 10 and the related description). 6 to 10, the absolute value of the δ-axis current is used. However, in the configuration on the αβ fixed coordinate system, this absolute value can be obtained as the norm value of the stator current (vector quantity) on the αβ fixed coordinate system. Can do. Note that the δ-axis current equivalent value on the γδ control coordinate system in which the current controller exists may be used only for the absolute value of the δ-axis current equivalent value used in the inductance selection unit.

  Even when the inductance selection unit is configured on either the γδ control coordinate system or the αβ fixed coordinate system, the absolute value of the δ-axis current equivalent value is used for this appropriate configuration. In the embodiment using FIG. 4, the absolute value of the measured δ-axis current value (norm of the measured stator current value) was used as this signal. As the absolute value of the δ-axis current equivalent value, the same current command value may be prepared as the δ-axis current equivalent value, and this absolute value may be used.

  FIG. 2 shows an embodiment of the speed control mode, but it is also possible to perform a quasi-torque control mode according to the torque control mode. In the quasi-torque control mode, the δ-axis current command value may be directly given from the outside.

  In the embodiment shown in FIG. 2, the current control is performed in a feedback manner. The present invention can also be applied to the case where the current control is performed in a feedforward manner, and the same effect as in the case of the feedback current control can be obtained.

  The phase determiner including the inductance selecting unit according to the present invention can be realized in an analog manner, but it is preferable that the phase determiner be configured digitally in view of the remarkable progress of recent digital technology. Although the digital configuration includes a hardware configuration and a software configuration, as will be apparent to those skilled in the art, any of the present invention can be configured. The present invention has been described in detail with reference to various embodiments using various drawings.

  The present invention can be widely used in applications where sensorless driving is desired in a high speed range exceeding the rated speed.

DESCRIPTION OF SYMBOLS 1 Permanent magnet synchronous motor 2 Phase determiner 2a Estimator with variable inductance 2a-1 Inductance selection part 2a-2 Estimator part 2a-2a Stator reaction magnetic flux calculation block 3 Power converter 4 Current detector 5a 3 phase 2 phase conversion 5b Two-phase three-phase converter 6a Vector rotator 6b Vector rotator 7 Cosine sine signal generator 8 Current controller 9 Command converter 10 Speed controller 11 Machine speed estimator

Claims (5)

Used in the current control process that controls the stator current that contributes to torque generation in a feedback or feed-forward manner using a vector rotator or a converter with a built-in vector rotator, and a vector rotator or a converter with a built-in vector rotator A phase determining step for determining a phase to be performed,
When the stator current is evaluated on the γδ control coordinate system specified by the vector rotator or the vector rotator built-in converter, the γ-axis or δ-axis current will be substantially zero. A positive d-axis inductance (Ld> 0) that constitutes a current control step and constitutes the phase determination step using a value different from a true inductance as a phase determiner inductance corresponding to a non-zero other-axis current. ) And a positive value q-axis inductance (Lq> 0) .
The phase determining step is configured using a state observer, a disturbance observer, a D-factor filter, or a device conforming thereto, and the phase determiner inductance used for the configuration is selected as a negative value. A drive control method for a permanent magnet synchronous motor having a positive d-axis inductance (Ld> 0) and a positive q-axis inductance (Lq> 0) . The negative value of the phase determiner inductance used in the configuration of the phase determining step is equivalent to the voltage limit equivalent value, the stator voltage equivalent value, the stator current equivalent value, the rotor speed, and the rotor speed. The positive d-axis inductance (Ld> 0) and the positive q-axis inductance (Lq> 0) according to claim 1, wherein the positive d-axis inductance (Ld> 0) and the positive q-axis inductance (Lq> 0) are variable according to at least one of the equivalent values of Control method for permanent magnet synchronous motor with motor. 2. The positive d-axis inductance (Ld ) according to claim 1, wherein the negative value of the inductance for the phase determiner used in the configuration of the phase determination step is determined by at least partially using iterative calculation. > 0) and a drive control method for a permanent magnet synchronous motor having a positive q-axis inductance (Lq> 0) . When using true stator inductance, the phase determination process is configured using a negative phase determiner inductance for the estimator that can properly estimate the rotor flux, induced voltage, or extended induced voltage. 4. A drive control of a permanent magnet synchronous motor having a positive d-axis inductance (Ld> 0) and a positive q-axis inductance (Lq> 0) according to claim 1, 2 or 3 Method. The phase determining step is configured by using at least a stator current equivalent value on an αβ fixed coordinate system or a γδ control coordinate system determined according to a stator winding as an input signal to the phase determining step. A drive control method for a permanent magnet synchronous motor having a positive d-axis inductance (Ld> 0) and a positive q-axis inductance (Lq> 0) according to claim 1, claim 2, or claim 3.

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