JP2015126607A - Motor control system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a motor control system combining high control responsiveness with control stability.SOLUTION: A motor control system 100 is provided which includes current detectors 52, 53 for detecting a current to be supplied to a motor 50 and outputting the current as current signals Iv, Iw and a control device 60 that includes: a signal conversion part 71 for converting the current signals Iv, Iw into feed back current signals Id, Iq; a voltage instruction generation part 74 for performing feedback control on the basis of a deviation between current instructions Idcom, Iqcom to the motor 50 and the feed back current signals Id, Iq and generating current instructions Vd, Vqto the motor 50; and a modulation percentage calculation part 78 for calculating a modulation percentage FM on the basis of the current instructions Vd, Vq. The voltage instruction generation part 74 changes a gain used in performing feedback control, with the modulation percentage FM.

Description

  The present invention relates to a motor control system, and more particularly to a structure of a motor control system that controls a motor with an inverter.

  In recent years, a technique for controlling a motor by converting DC power into AC power by an inverter has been increasingly used. In this case, as a control method, according to a desired AC voltage waveform applied to the motor, a DC voltage is converted into a plurality of pulse trains having different widths, and PWM control for controlling the driving of the motor with a pseudo AC voltage waveform; There are two types of control methods of rectangular wave control in which the driving of the motor is controlled by a rectangular wave pulse whose ratio between the high level period with a high voltage and the low level period with a low voltage is 1: 1. In addition, the PWM control includes a sine wave PWM control in which a desired voltage waveform is a sine wave and a motor drive is controlled by a pseudo sine wave AC voltage, and a sine wave (a rectangular pulse and a distorted desired voltage waveform). There is an overmodulation PWM control in which the motor drive is controlled by a pseudo distorted sine wave AC voltage. In overmodulation PWM control, the waveform of the AC voltage driving the motor is between a waveform close to a sine wave (a waveform with little distortion from the sine wave) to a waveform close to a rectangular pulse (a waveform with a large distortion from the sine wave). Can be changed.

  Since PWM control is suitable for motor control in the low rotation range and rectangular wave control is suitable for high rotation range, motor drive such as PWM control in the low rotation range and rectangular wave control in the high rotation range. In general, the control logic is switched according to the state. The control method between PWM control and rectangular wave control is switched between sine wave PWM control and overmodulation PWM control with a waveform close to a rectangular pulse, and then control is performed between overmodulation PWM control and rectangular wave control. By switching, it is common to suppress the generation of steps in the torque and rotation speed of the motor due to switching of the control logic so that the motor can be smoothly controlled in a wide operating range.

  However, in the overmodulation PWM control immediately before switching to the rectangular wave control, the waveform of the pseudo AC voltage that drives the motor has a shape close to a rectangular pulse, and the waveform of the current flowing through the motor is greatly distorted from the sine wave. Therefore, the control operation may become unstable. For this reason, when the operating state of the motor suddenly changes, the control logic jumps from overmodulation PWM control to rectangular wave control, and a step may occur in the motor torque and rotation speed.

  With respect to this point, a method has been proposed in which control stability in overmodulation PWM control is improved by filtering the detection result of the motor current (see, for example, Patent Document 1).

JP 2010-88205 A

  By the way, in the control of the motor, the output range of the motor is expanded by boosting the voltage of the battery. Normally, a boost converter that boosts the voltage by switching on / off the switching element is used for boosting, and the output voltage of the boost converter is subject to fluctuations. Therefore, a smoothing capacitor for supplying the boosted voltage to the inverter as a smooth and stable DC voltage is provided between the boost converter and the inverter.

  In recent years, there has been a demand for downsizing the motor control device, and for this reason, the smoothing capacitor may be made smaller. In this case, since the DC voltage supplied to the inverter is likely to fluctuate, it is necessary to improve control response in order to stably control the motor. On the other hand, when control responsiveness is simply increased, there is a problem that control stability is lowered.

  In this regard, in the control system of the prior art described in Patent Document 1, although the control can be stabilized, the delay in the current detection caused by the filtering with respect to the detection result of the motor current leads to a decrease in the control responsiveness. There was a problem that it was.

  Therefore, an object of the present invention is to provide a motor control system that has both high control response and control stability.

  The motor control system of the present invention includes at least one current detector that detects a current supplied to a motor and outputs the current signal as a current signal, and a signal converter that converts the current signal output from the current detector into a feedback current signal. A voltage command generation unit that generates a voltage command to the motor by performing feedback control based on a deviation between the current command to the motor and the feedback current signal, and a modulation rate that calculates a modulation rate based on the voltage command And a control device including a calculation unit, wherein the voltage command generation unit changes a gain in the feedback control according to the modulation rate.

  In the motor control system of the present invention, it is preferable that the voltage command generation unit changes the gain when the modulation rate exceeds a predetermined threshold.

  In the motor control system of the present invention, the voltage command generator changes the gain so that when the modulation rate exceeds a predetermined threshold, the modulation rate is smaller than when the modulation rate is equal to or lower than a predetermined threshold. It is also suitable.

  In the motor control system of the present invention, it is also preferable that the voltage command generator is configured such that the gain decreases as the modulation rate increases.

  In addition, the motor control system of the present invention further includes an inverter that supplies power to the motor, and the voltage command generator generates the inverter when the inverter is controlled in a rectangular mode or an overmodulation PWM mode. It is also preferable to make the gain smaller than in the case of control in the sine wave PWM mode.

  The present invention has an effect that it is possible to provide a motor control system having both high control response and control stability.

It is a systematic diagram which shows the structure of the motor control system in embodiment of this invention. It is a control block diagram of the motor control system in the embodiment of the present invention. It is explanatory drawing which shows the voltage waveform at the time of the sine wave PWM control of the motor control system in embodiment of this invention. It is explanatory drawing which shows the basic waveform of the voltage and output voltage of each phase of the motor control system in embodiment of this invention. It is explanatory drawing which shows the application range of the rotation speed, torque, and control mode at the time of controlling a motor with the motor control system in embodiment of this invention. It is a flowchart which shows operation | movement of the motor control system in embodiment of this invention. It is a map of the gain correction coefficient with respect to the modulation factor of the motor control system in the embodiment of the present invention. It is a graph which shows the change of the torque and the modulation factor with respect to time of the motor control system in the embodiment of the present invention.

Embodiments of the invention will be described below. As shown in FIG. 1, a motor control system 100 according to an embodiment of the present invention includes a battery 10, a boost converter 20 that boosts the voltage of DC power supplied from the battery 10 and outputs boosted DC power, a switching element 33a to 35a, the inverter 30 supplies the motor 50 converts the boosted DC power to AC power supplied from boost converter 20 are turned on and off at the carrier frequency f _mg of 33B～35b, supplied to the motor 50 And a control device 60 that adjusts the on / off operation of each of the switching elements 33a to 35a and 33b to 35b of the inverter 30 by feedback control of the generated current.

  The boost converter 20 and the inverter 30 are the same as the negative circuit 11 connected to the negative side of the battery 10, the low piezoelectric circuit 12 connected to the positive side of the battery 10, and the positive output of the boost converter 20. And includes a high-voltage path 13 that is a plus-side input terminal of the inverter 30.

  The step-up converter 20 includes an upper arm switching element 23a disposed between the low piezoelectric path 12 and the high piezoelectric path 13, a lower arm switching element 23b disposed between the negative side electric path 11 and the low piezoelectric path 12, and a low piezoelectric path. 12 includes a reactor 21 arranged in series with the filter 12, a filter capacitor 22 arranged between the low piezoelectric path 12 and the minus-side electric path 11, and a low voltage sensor 27 that detects a low voltage VL across the filter capacitor 22. Yes. Further, diodes 24a and 24b are connected in antiparallel to the switching elements 23a and 23b, respectively. Boost converter 20 turns on lower arm switching element 23b, turns off upper arm switching element 23a and stores electric energy from battery 10 in reactor 21, then turns off lower arm switching element 23b and turns upper arm switching element 23a off. The voltage is increased by the electrical energy accumulated in the reactor 21 and the voltage boosted to the high piezoelectric path 13 is output. Therefore, the output voltage from boost converter 20 varies depending on the ON / OFF cycle of switching elements 23a and 23b.

A smoothing capacitor 31 that makes the stepped-up voltage fluctuating from the step-up converter 20 between the negative side electric circuit 11 and the high-voltage circuit 13 on the input side (step-up converter 20 side) of the inverter 30 smooth, and both ends of the smoothing capacitor 31 The high voltage sensor 32 for detecting the high voltage VH is included. Further, between the negative side electric circuit 11 on the opposite side of the step-up converter 20 of the smoothing capacitor 31 and the high-voltage circuit 13 opposed thereto, the upper arm switching elements 33a to 35a, Six switching elements of the lower arm switching elements 33b to 35b are arranged in series, two for each phase. Output lines 55, 56, and 57 of U, V, and W phases are connected between the upper arm switching elements 33a to 35a and the lower arm switching elements 33b to 35b. The U, V, and W output lines 55, 56, and 57 are connected to the input terminals of the U, V, and W phases of the motor 50, respectively. In addition, U.S. V. Diodes 36a to 38a and 36b to 38b are connected in antiparallel to the W-phase upper arm switching elements 33a to 35a and the lower arm switching elements 33b to 35b, respectively. Inverter 30 includes an upper arm switching element 33a to 35a, the AC power boosting DC power supplied from boost converter 20 turns on and off the six switching elements of the lower arm switching element 33b~35b the carrier frequency f _mg This is converted and supplied to the motor 50 for driving the vehicle.

  Two output lines 56 and 57 for supplying V and W phase power from the inverter 30 to the motor 50 detect currents flowing through the output lines 56 and 57 and output them as current signals Iv and Iw. , 54 are attached. In addition, a resolver 52 that detects the rotational speed or rotational angle θ of the rotor is attached to the motor 50. The control device 60 includes a CPU 61, a storage unit 62, and a device / sensor interface 63 for connecting each device and sensor. The CPU 61, the storage unit 62, the device / sensor interface 63, and the like. Is a computer connected by a data bus 64. In the storage unit 62, control data 65, a PWM control block 70 in which PWM control logic is stored, a rectangular wave control block 80 in which rectangular wave control logic is stored, PWM control and rectangular wave control And a control method switching unit 90 for switching the control method. As shown in FIG. 2, the PWM control block 70 includes a signal converter 71, a current command generator 72, current subtractors 73a and 73b, a voltage command generator 74, a voltage amplitude corrector 75, and a two-phase three-phase converter 76. , A PWM control signal generation output unit 77, and a modulation factor calculation unit 78. In FIG. 1, only the signal conversion unit 71, the voltage command generation unit 74, the PWM control signal generation output unit 77, and the modulation factor calculation unit 78 are illustrated, and the other portions are not illustrated. Further, as shown in FIG. 2, the rectangular wave control block 80 includes a power calculation unit 81, a torque calculation unit 82, a torque subtractor 83, a PI calculation unit 84, a rectangular wave signal generator 85, and a rectangular wave control signal generation output unit. 86. The voltage command generator 74 includes a gain correction coefficient map (FIG. 7) with respect to the modulation rate described later.

  The step-up converter 20 of the motor control system 100 and the switching elements 23a, 23b, 33a to 35a, and 33b to 35b of the inverter 30 are connected to the control device 60 via the device / sensor interface 63, and are driven by commands of the control device 60. It is configured as follows. The V and W phase current sensors 53 and 54 and the resolver 52 attached to the motor 50 are respectively connected to the device / sensor interface 63 of the control device 60, and the current detected by each sensor, the rotational angle θ of the rotor, and the like. The signal is input to the control device 60 via the device / sensor interface 63.

Before describing the operation of the motor control system 100 described above, the PWM control (including overmodulation PWM control) by the inverter 30 (three-phase bridge inverter) shown in FIG. Wave control and “modulation rate” in this specification will be described. In the case of the sine wave PWM control, as shown in FIG. 3A, the sine wave signals e _U , e _V , e _W of the U, V, W phases of the angular frequency ω ₀ and the peak value V ₀ are 1 is compared with the triangular carrier wave e _{c of the} frequency ω _c and the peak value V _c , and the switching elements 33a, 34a, 35a of the upper arm and the switching elements 33b of the lower arm shown in FIG. 34b and 35b are alternately turned on and off. As a result, the voltages V _U , V _V , and V _{W with} respect to the DC midpoint of the U, V, and W output lines 55, 56, and 57 (the midpoint between the high-piezoelectric path 13 and the minus-side electric path 11) b) to VH / 2 or -VH / 2 as shown in FIG. 3 (d), and are shown by the one-dot chain line a, two-dot chain line b, and broken line in FIGS. 3 (b) to 3 (d). It becomes a pulse train modulated like a sine wave d, e, f as shown in c. Further, the line voltage V _{UV that} is a potential difference between the U-phase voltage V _U and the V-phase voltage V _V is a sine wave h between VH and −VH as shown in FIG. The modulated pulse train g is obtained. Here, the frequency of the output sine wave signals e _U, e _V, can be controlled by the angular frequency omega ₀ of e _W, the size thereof, a sine wave signal e _U, e _V, e _W peak value V of it is controlled by ₀ and the ratio of the peak value V _c of the triangular carrier _{e c (α = V 0 /} V c).

The pulse train g shown in FIG. 3 (e) has a basic waveform of the angular frequency ω ₀ (the sine wave h shown in FIG. 3 (e), the sine wave ha shown in FIG. 4 (d)), and a higher harmonic of the angular frequency. Expressed as a Bessel function with wave components. In the case of sine wave PWM control, the peak value Va of the sine wave h, which is the fundamental wave of the output voltage (line voltage V _UV ) shown in FIG. 3E, is Va = 0.86 × α × VH (α = V ₀ / V _c ), and its effective value Vam is Va / √2 = 0.86 / √2 × α × VH = 0.61 × α × VH. In the present specification, “modulation rate FM” is defined as the ratio of the effective value Vam of the fundamental wave component of the motor applied voltage (line voltage) to the high voltage VH. That is, the modulation rate FM = Vam / VH = 0.61 × α. Since α ≦ 1 in the case of sinusoidal PWM control, the modulation factor FM changes between 0 and 0.61 (see FIGS. 4A and 4D). The sine wave PWM control controls the amplitude and phase of the motor applied voltage (alternating current) according to the operating state of the motor 50, so that torque variation is small and fine control can be performed.

In the rectangular wave control, as shown in FIG. 4C, a rectangular wave having a ratio of the high level period to the low level period of 1: 1 is applied to the motor 50. The output voltage in this case is also expressed as a Bessel function of the basic waveform of the angular frequency ω ₀ (the sine wave hc shown in FIG. 4 (f)) and the harmonic component having a higher angular frequency, and is shown in FIG. 4 (f). The peak value Vc of the sine wave hc, which is the basic waveform shown, is Vc = 2 × √3 × VH / π = 1.10 × VH, and its effective value Vcm is Vcm = Vc / √2 = 0.78 × VH. It becomes. Therefore, the “modulation factor FM” in the case of rectangular wave control, that is, the ratio of the effective value Vcm of the fundamental wave component of the motor applied voltage (line voltage) to the high voltage VH is 0.78. Therefore, in the rectangular wave control, an effective voltage higher than that in the sine wave PWM control can be applied to the motor 50 even with the same high voltage VH. On the other hand, in the rectangular wave control, as shown in FIG. 4C, since the amplitude of the motor applied voltage is fixed, the torque is controlled by the phase control of the rectangular wave voltage pulse based on the deviation between the actual torque value and the torque command value. Since control is executed, it is not possible to perform fine control such as sine wave PWM control, and torque fluctuation tends to be larger than that of sine wave PWM control.

Overmodulation PWM control is larger U, V, each phase of the sinusoidal signal e _U of W, e _V, e _W peak value V ₀ than the peak value V _c of the triangular carrier e _c shown in FIG. 3 (a) In the range (α = V ₀ / V _c > 1.0), PWM control of the same method as sine wave PWM control is performed. For this reason, as shown in FIG. 4 (b), the voltages V _U , V _U , V DC, V _V and V _W are distorted sine waves, and the line voltage applied to the motor 50 is also a distorted sine wave. As shown in FIG. 4 (e), the output voltage in the case of overmodulation PWM control is also the basic waveform of the angular frequency ω ₀ (the sine wave hb shown in FIG. 4 (f)) and the harmonic component having a higher angular frequency. And is expressed as a Bessel function. The peak value Vb and the effective value Vbm of the sine wave hb, which are the fundamental wave components, are respectively the peak value Va and effective value Vam of the fundamental wave ha in the case of the sine wave PWM control shown in FIG. ) Between the peak value Vc and the effective value Vcm of the fundamental wave hc in the case of the rectangular wave control shown in FIG. Therefore, the “modulation rate FM” in the case of overmodulation PWM control is an intermediate value between the maximum value 0.61 in the case of sine wave PWM control and 0.78 in the case of rectangular wave control. Therefore, in the overmodulation PWM control, an effective voltage intermediate between the sine wave PWM control and the rectangular wave control can be applied to the motor 50 for the same high voltage VH. Further, in overmodulation PWM control, when the modulation factor FM is close to 0.61, the waveform of the line voltage applied to the motor 50 is close to a sine wave, and torque control is small and fine control is possible. When the modulation factor is close to 0.78, the waveform of the line voltage applied to the motor becomes close to a rectangular wave, and the control operation tends to become unstable.

  In the motor 50, since the induced voltage increases as the rotational speed and output torque increase, the required drive voltage (motor required voltage) increases, and the boost converter 20 boost voltage, that is, the high voltage VH is higher than the motor required voltage. Need to be set higher. On the other hand, the boost voltage (high voltage VH) of the boost converter 20 has a limit value (VH maximum voltage). Therefore, as shown in FIG. 5, sine wave PWM control (sine wave PWM mode) is used in the low rotational speed range I to reduce torque fluctuation, and in the high rotational speed range III, the motor 50 is applied even at the same high voltage VH. Rectangular wave control (rectangular mode) capable of increasing the applied effective voltage is applied, and overmodulation PWM control (overmodulation PWM mode) is applied in the intermediate rotation speed region II. The control logic is switched between overmodulation PWM control (overmodulation PWM mode) and rectangular wave control (rectangular mode). As described above, in the overmodulation PWM control (overmodulation PWM mode), when the modulation rate is close to 0.78, the waveform of the line voltage applied to the motor is close to a rectangular wave, and the control operation is performed. Since there is a tendency to become unstable, in the prior art, when the modulation factor is close to 0.78 in overmodulation PWM control (overmodulation PWM mode), the overmodulation PWM control (overmodulation PWM) is caused by the unstable control operation. The control logic jumps from the (mode) to the rectangular wave control (rectangular mode), and the torque and the rotational speed sometimes fluctuate.

Next, the operation of the motor control system 100 of the present embodiment will be described with reference to FIGS. 2 and 6 to 8. As shown in FIG. 2, the signal conversion unit 71 of the PWM control block 70 receives the V-phase and W-phase current signals Iv input from the V-phase and W-phase current sensors 53 and 54 via the device / sensor interface 63. , Iw to calculate the current value of each phase of U, V, W, and based on the calculated current value of each phase and the rotation angle θ of the rotor detected by the resolver 52, the three-phase values of U, V, W The alternating current is converted into a two-phase feedback current signal Id, Iq composed of a d-axis current Iq and a q-axis current Id and output. On the other hand, the torque command Trqcom generated in another block (not shown) of the control device 60 is input to the current command generation unit 72, and the current command generation unit 72 is a d-axis that is a current command from the torque command Trqcom to the motor 50. A current command Idcom and a q-axis current command Iqcom are generated and output. The d-axis current command Idcom and q-axis current command Iqcom output from the current command generator 72 and the d-axis current Iq and q-axis current Id that are feedback current signals output from the signal converter 71 are current subtractors, respectively. 73a and 73b, and deviations ΔId and ΔIq between the command currents Idcom and Iqcom and the feedback currents Iq and Id are calculated. The current deviations ΔId and ΔIq are input to the voltage command generation unit 74, and voltage commands Vd ^# and Vq ^# to the d-axis and q-axis motors 50 are calculated and output by proportional integration calculation. Thereby, the current of each phase of the motor 50 is fed back (feedback control). The voltage commands Vd ^# and Vq ^# output to the d-axis and q-axis motors 50 output from the voltage command generation unit 74 are input to the voltage amplitude correction unit 75. The voltage amplitude correction unit 75 corrects the amplitude of the applied voltage of the motor 50 with respect to the d-axis and q-axis voltage command values Vd ^# and Vq ^# calculated by the voltage command generation unit 74. Is executed and output. The voltage command values Vd ^# ′ and Vq ^# ′ after the correction processing are converted into U, V, and W phase voltages V _U , V _V , and V _W by the two-phase / three-phase converter 76. As described with reference to FIG. 3A to FIG. 3D, the PWM control signal generation output unit 77 has a voltage command signal (sine wave signal) e _U , e _V , e _{W of} each phase and a triangle. based on the comparison of the carrier e _c, U of the inverter 30, V, W of respective phases of the switching elements 33a to 35a, and generates a signal for turning on and off the 33B～35b, outputs. This signal is input to the gates of the switching elements 33a to 35a and 33b to 35b via the device / sensor interface 63 shown in FIG. 1, and the switching elements 33a to 35a and 33b to 35b are connected to the gates of FIG. As shown in FIG. 3D, an on / off operation is performed, and a pulse train g of a voltage modulated into a sine wave h as shown in FIG.

The d-axis and q-axis voltage command values Vd ^# and Vq ^# output from the voltage command generator 74 are input to the modulation factor calculator 78. The modulation factor calculator 78 calculates the modulation factor FM according to the following (Equation 1).
Modulation rate FM = (Vd ^{# 2} + Vq ^{# 2} ) ^1/2 / VH -------- (Formula 1)
The calculated modulation factor FM is input to the control method switching unit 90 and fed back to the voltage command generation unit 74. When the input modulation factor FM is 0.78 or more, the control method switching unit 90 cannot generate an appropriate AC voltage in the PWM control, so the torque command Trqcom is input to the rectangular wave control block 80. In addition, the control logic is switched so that the motor 50 is controlled by the rectangular wave control block 80. When the calculated modulation factor FM is smaller than 0.78 (when FM <0.78), the PWM control is continued.

Based on the fed back modulation factor FM, the voltage command generation unit 74 corrects the gain of the proportional integral calculation (PI gain G) as shown in the flowchart of FIG. As shown in step S <b> 101 of FIG. 6, the voltage command generation unit 74 acquires the modulation rate FM calculated by the modulation rate calculation unit 78. The modulation factor FM at this time is the previous value calculated at the previous control clock. As shown in step S102 of FIG. 6, the voltage command generation unit 74 compares the threshold A with the acquired modulation rate FM, and when the modulation rate FM exceeds the threshold A (modulation rate FM> A). As shown in step S103 of FIG. 6, the PI gain correction coefficient K for the modulation factor FM is calculated or acquired from the map of the gain of the proportional integral calculation (PI gain) for the modulation factor FM shown in FIG. Then, as shown in step S104 of FIG. 6, the voltage command generation unit 74 multiplies the PI gain correction coefficient K by the PI gain G of the normal control, and uses the corrected PI gain G ′ resulting from the multiplication in the proportional integration calculation. In place of the PI gain G, proportional-integral calculation is performed as shown in step S105 of FIG. 6 to calculate the d-axis and q-axis voltage command values Vd ^# and Vq ^# . Further, when the modulation rate FM does not exceed the threshold A in step S102 of FIG. 6 (modulation rate FM ≦ A), the PI gain G in the case of normal control remains as shown in step S106 of FIG. Proceeding to step S105 in FIG. 6, the voltage command values Vd ^# and Vq ^# for the d-axis and the q-axis are calculated using the normal PI gain G.

By the above operation, as shown in FIG. 7, when the modulation rate FM acquired by the voltage command generation unit 74 does not exceed the threshold A (modulation rate FM ≦ A), the PI gain correction coefficient K is 1.0. Thus, the proportional integral calculation is performed with the normal PI gain G. When the modulation factor FM acquired by the voltage command generation unit 74 exceeds the threshold A, the PI gain correction coefficient K is reduced from 1.0 as shown by lines j and k in FIG. The correction gain G ′ obtained by multiplying the correction coefficient K is reduced from the normal PI gain G. The PI gain correction coefficient K may be linearly reduced from 1.0 to K ₁ when the modulation rate FM is between the threshold A and the upper limit of 0.78, as indicated by the line j in FIG. Then, as shown by the line k in FIG. 7, after the modulation factor FM is reduced from 1.0 to K ₁ between the threshold A and the second threshold B, the upper limit of 0.78 is applied from the second threshold B. it may be a K ₁ constant between. The threshold A may be set to 0.61, which is the upper limit of the modulation factor FM in the case of sine wave PWM control, for example. The threshold value B may be an intermediate value between 0.61 and 0.78. That is, when the PI gain correction coefficient K such as the line j and the line k in FIG. 7 is used, the voltage command generation unit 74 is controlled when the motor 50 or the inverter 30 is controlled in the rectangular mode or the overmodulation PWM mode. First, the gain G is made smaller than when the motor 50 or the inverter 30 is controlled in the sine wave PWM mode.

  In the motor control system 100 of the present embodiment described above, the PI gain correction coefficient K is reduced as the modulation rate FM approaches the upper limit of 0.78, so that the control is switched to rectangular wave control where the control tends to become unstable. In the immediately preceding overmodulation PWM control, the feedback amount of the current of the motor 50 can be reduced. As a result, control stability in overmodulation PWM control immediately before switching to rectangular wave control can be ensured. In this embodiment, the PI gain correction coefficient K is increased as the modulation rate FM decreases from the upper limit of 0.78, and the modulation rate FM is set to 1.0 when the threshold value A (for example, 0.61). Therefore, in the sine wave PWM control, the feedback amount of the motor current is increased, and high-speed response can be ensured. When the threshold value A is 0.61, which is the upper limit of the modulation factor FM in the case of sine wave PWM control, in the overmodulation PWM control between the modulation factor FM of 0.61 and the upper limit of 0.78, the modulation factor FM Increases, that is, distortion from the sine wave of the line voltage waveform applied to the motor 50 increases, and as the harmonic component of the current of the motor 50 increases, the PI gain correction coefficient K decreases. Since the feedback amount of the current of the motor 50 is reduced, the feedback amount can be adjusted in accordance with the magnitude of the harmonics, and both control stability and control response can be achieved.

  In the embodiment described above, the voltage command generation unit 74 has been described as performing feedback control using proportional integral calculation. However, for example, feedback control using proportional calculation or feedback control using integral calculation is performed. You may make it perform.

  Next, the case where the vehicle driving motor 50 is controlled by the conventional motor control system and the case where the vehicle driving motor 50 is controlled by the motor control system 100 of the present embodiment will be compared and described. .

When controlling the motor 50 for driving the vehicle by the conventional motor control system, the over-modulation PWM control is performed in the region where the modulation factor FM is high. The time t _{0 in} FIG. 8A and FIG. In addition, as shown by the line u in FIG. 8B, when the torque Trq of the motor 50 changes suddenly due to vehicle slip, grip or the like, the current value of the motor 50 changes suddenly and the overmodulation PWM control becomes unstable. Thus, as indicated by the one-dot chain line p in FIG. 8A, the control method jumps from overmodulation PWM control to rectangular wave control at time t ₀ . As a result, steps may occur in the rotational speed and torque of the motor 50, and the drivability of the vehicle may be reduced.

On the other hand, when the motor control system 100 of the present embodiment controls the motor 50 for driving the vehicle, the amount of feedback of the motor current is reduced in the region where the modulation factor FM is high. At time t ₀ in FIGS. 8A and 8B in which overmodulation PWM control is performed in a high region, as shown by line u in FIG. Even when the torque Trq suddenly changes, overmodulation PWM control can maintain a stable control state. Then, as indicated by the solid line s in FIG. 8A, the control method does not jump from overmodulation PWM control to rectangular wave control at time t ₀ , and the modulation factor FM of overmodulation PWM control is 0.78 until time t _1. It is possible to change the control logic from overmodulation PWM control to rectangular wave control so that the speed is gradually increased to the vicinity and no step occurs in the rotation speed or torque of the motor 50. Thereby, the fall of the drivability of a vehicle can be suppressed. When the motor control system 100 of the present embodiment controls the motor 50 for driving the vehicle, in the overmodulation PWM control in which the modulation factor FM is near 0.61 or the sine wave PWM control, Since a gain close to the PI gain G is used, high speed response is ensured.

  Furthermore, in the motor control system 100 of the present embodiment, even when the capacity of the smoothing capacitor 31 is reduced and the input voltage to the inverter 30 slightly fluctuates, control stability in overmodulation PWM control is ensured. In addition, high-speed response can be ensured in overmodulation PWM control with a modulation factor FM of around 0.61 and sinusoidal PWM control. Thus, the motor control system 100 of this embodiment can perform control having both control stability and high-speed response.

  DESCRIPTION OF SYMBOLS 10 Battery, 11 Negative side electric circuit, 12 Low piezoelectric circuit, 13 High piezoelectric circuit, 20 Boost converter, 21 Reactor, 22 Filter capacitor, 23a, 23b, 33a-35a, 33b-35b Switching element, 24a, 24b, 36a-38a, 36b -38b Diode, 27 Low voltage sensor, 30 Inverter, 31 Smoothing capacitor, 32 High voltage sensor, 50 Motor, 52 Resolver, 53, 54 Current sensor, 55, 56, 57 Output line, 60 Controller, 61 CPU, 62 Memory Unit, 63 device / sensor interface, 65 control data, 64 data bus, 70 PWM control block, 71 signal conversion unit, 72 current command generation unit, 73a, 73b current subtractor, 74 voltage command generation unit, 75 voltage amplitude correction Part, 76 two-phase three-phase Conversion unit, 77 PWM control signal generation output unit, 78 Modulation rate calculation unit, 80 Rectangular wave control block, 81 Power calculation unit, 82 Torque calculation unit, 83 Torque subtractor, 84 calculation unit, 85 Rectangular wave signal generator, 86 Rectangular wave control signal generation output unit, 90 control system switching unit, 100 motor control system.

Based on the fed back modulation factor FM, the voltage command generation unit 74 corrects the gain of the proportional integral calculation (PI gain G) as shown in the flowchart of FIG. As shown in step S <b> 101 of FIG. 6, the voltage command generation unit 74 acquires the modulation rate FM calculated by the modulation rate calculation unit 78. The modulation factor FM at this time is the previous value calculated at the previous control clock. As shown in step S102 of FIG. 6, the voltage command generation unit 74 compares the threshold A with the acquired modulation rate FM, and when the modulation rate FM exceeds the threshold A (modulation rate FM> A). As shown in step S103 of FIG. 6, the PI gain correction coefficient K for the modulation factor FM is calculated or acquired from the map of the gain of the proportional integral calculation (PI gain) for the modulation factor FM shown in FIG. Then, as shown in step S104 of FIG. 6, the voltage command generation unit 74 multiplies the PI gain correction coefficient K by the PI gain G of the normal control, and uses the corrected PI gain G ′ resulting from the multiplication in the proportional integration calculation. In place of the PI gain G, proportional-integral calculation is performed as shown in step S105 of FIG. 6 to calculate the d-axis and q-axis voltage command values Vd ^# and Vq ^# . If the modulation rate FM does not exceed the threshold A in step S102 of FIG. 6 (modulation rate FM ≦ A) , the PI gain correction coefficient K for the modulation rate FM is 1.0 as shown in the map of FIG. from, as shown in step S106 of FIG. 6, the PI gain G in the case of the normal control by correcting factor 1.0, i.e., remains PI gain G in the case of normal control, the process proceeds to step S105 in FIG. 6 The d-axis and q-axis voltage command values Vd ^# and Vq ^# are calculated using the normal PI gain G.

Claims (5)

At least one current detector for detecting a current supplied to the motor and outputting it as a current signal;
A signal converter for converting the current signal output from the current detector into a feedback current signal, and a feedback control based on a deviation between the current command to the motor and the feedback current signal to generate a voltage command to the motor A control device including a voltage command generation unit that performs a modulation rate calculation unit that calculates a modulation rate based on the voltage command;
A motor control system comprising:
The voltage command generation unit changes the gain of the feedback control according to the modulation rate,
A motor control system characterized by The motor control system according to claim 1,
The voltage command generation unit changes the gain when the modulation rate exceeds a predetermined threshold;
A motor control system characterized by The motor control system according to claim 1 or 2,
The voltage command generation unit, when the modulation rate exceeds a predetermined threshold, changing the gain so that the modulation rate is smaller than the case where the modulation rate is equal to or lower than a predetermined threshold;
A motor control system characterized by The motor control system according to claim 2 or 3,
The voltage command generator is configured to reduce the gain as the modulation rate increases;
A motor control system characterized by The motor control system according to claim 2,
An inverter for supplying power to the motor;
The voltage command generator is configured such that when the inverter is controlled in a rectangular mode or an overmodulation PWM mode, the gain is smaller than when the inverter is controlled in a sine wave PWM mode.
A motor control system characterized by

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