JP2011004541A - Controller of ac motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control configuration which is commonly applied over a wide range of torque control without causing complication of arithmetic processing in feed forward control for changing the voltage phase of the a rectangular voltage applied to an AC motor, based on a torque command value.SOLUTION: On a torque characteristic line 500 having the nonlinear characteristics, a phase change amount for canceling a torque compensation amount with respect to the torque command value is obtained by a linear approximation calculation in accordance with an inclination of the tangent at the present operating point. In inversion regions 600, 610, there is fear that the phase change amount is calculated at the opposite polarity of the original one since the inclination of the tangent is different in the other regions. For this reason, repetitive calculation is performed after correcting the inclination of the tangent to a predetermined positive value when the inclination of the tangent obtained by the linear approximation calculation is negative in the arithmetic processing in which the linear approximation calculation is repeated a plurality of times while updating the operation point.

Description

  The present invention relates to an AC motor control device, and more particularly to motor control in which a DC voltage is converted into a rectangular wave AC voltage by an inverter and applied to the AC motor.

  A motor control system that drives and controls an AC motor by converting a DC voltage into an AC voltage by an inverter is generally used. In such a motor control system, generally, in order to drive an AC motor with high efficiency, the motor current is controlled according to sinusoidal pulse width modulation (PWM) control based on vector control.

  However, the sinusoidal PWM control has a problem that it is difficult to obtain a high output in a high speed region because the fundamental wave component of the output voltage of the inverter cannot be sufficiently increased and the voltage utilization rate is limited. In view of this point, it has been proposed to adopt a control method capable of outputting a voltage having a larger fundamental wave component than the sine wave PWM control.

  Japanese Patent Laying-Open No. 2006-320039 (Patent Document 1) describes a control system in which a rectangular wave voltage having an amplitude of a voltage variably controlled by a converter is applied to an AC motor. In particular, in Patent Document 1, basically, the voltage phase of the rectangular wave voltage is changed according to the torque deviation, and when the motor rotation speed changes abruptly, the converter is changed according to the change ratio of the motor rotation speed. Control for changing the output voltage is described.

JP 2006-320039 A

  In Patent Document 1, phase control of a rectangular wave voltage is performed by feeding back the actual torque of the AC motor. However, in such torque feedback control, when the torque command value changes, the torque phase associated with the change is detected, and then the voltage phase changes according to the control calculation for eliminating the torque deviation. Therefore, there is room for improvement in the control response when the torque command value changes.

  On the other hand, the output torque of the AC motor varies depending not only on the voltage phase which is the operation amount of the rectangular wave voltage control but also on the operating state of the motor represented by the rotation speed. Therefore, it is preferable to directly determine the amount of change in the voltage phase by feedforward control reflecting the motor operating state with respect to the change in the torque command value.

  However, in general, since the change in output torque with respect to the voltage phase in the rectangular wave voltage control has a non-linear characteristic, in the feedforward control as described above, the calculation for obtaining the voltage phase change amount corresponding to the necessary torque compensation amount. There are concerns about the complexity and high load of processing.

  Furthermore, in order to perform the feedforward control corresponding to a wide voltage phase change region (that is, a wide torque region), there is a concern that a special control process reflecting the nonlinear characteristic is required. When such special processing becomes complicated, the load of arithmetic processing is further increased.

  The present invention has been made to solve such problems, and an object of the present invention is to feed forward a voltage phase of a rectangular wave voltage applied to an AC motor based on a torque command value. The purpose of the present invention is to provide a control configuration that can be commonly applied over a wide torque control range without complicating arithmetic processing.

  The control apparatus for an AC motor according to the present invention includes an inverter and a rectangular wave voltage control unit. The inverter converts the DC voltage into an AC voltage for rotationally driving the AC motor. The rectangular wave voltage control unit is configured to control the voltage phase of the rectangular wave voltage applied from the inverter to the AC motor. The rectangular wave voltage control unit includes a linear approximation unit, a polarity check unit, a phase change amount calculation unit, an update processing unit, and a repetition processing determination unit. The linear approximation unit is a first operating point on the torque characteristic line (the torque characteristic line) according to a model expression obtained by substituting the current motor state into a differential expression obtained by differentiating a predetermined torque characteristic expression of the AC motor with a voltage phase. Is configured to calculate a first gradient that is a ratio of a torque change to a voltage phase change. The polarity check unit maintains the first slope when the polarity of the first slope calculated by the linear approximation section matches the assumed polarity, while maintaining the first slope when there is a mismatch. Processing is performed to correct the first slope to a predetermined value. The phase change amount calculation unit calculates a torque compensation amount at the first operating point based on the torque command value, and divides the torque compensation amount by the first slope processed by the polarity check unit. The phase change amount is calculated. The update processing unit updates the first operating point according to the first phase change amount calculated by the phase change amount calculation unit, and in response to the update process, the linear approximation unit, the polarity check unit, and the phase change amount A repetitive process for calculating the first phase change amount at the first operating point updated by the arithmetic unit is instructed. The repetitive process determination unit determines whether or not a repetitive process end condition is satisfied. Further, when it is determined that the end condition is satisfied, the update processing unit is configured to match the output torque of the AC motor with the torque command value based on the updated voltage phase of the first operating point at that time. The voltage phase change amount is determined.

  Preferably, the polarity check unit changes the assumed polarity according to the rotation direction of the AC motor.

  More preferably, the iterative process determination unit determines that the end condition is satisfied when the number of updates of the first operating point reaches a predetermined number. Alternatively, the repetitive processing determination unit determines that the end condition is satisfied when a predetermined time has elapsed from the start of the control calculation.

  Further preferably, when the difference between the torque value calculated according to the torque characteristic equation and the torque command value at the first operating point updated by the update processing unit is smaller than a predetermined value, It is determined that the end condition is satisfied. Alternatively, the iterative processing determination unit determines that the end condition is satisfied when the absolute value of the first phase change amount calculated by the phase change amount calculation unit is smaller than a predetermined determination value.

  Preferably, the rectangular wave voltage control unit further includes a feedback control unit and a calculation unit. The feedback control unit is configured to control the voltage phase of the rectangular wave voltage based on torque deviation feedback with respect to the torque command value. The calculation unit sets the voltage phase command value according to the sum of the voltage phase set by the feedback control unit and the amount of change calculated by the update processing unit.

  Preferably, the AC motor is configured to be mounted on an electric vehicle and generate a vehicle driving force of the electric vehicle.

  According to the present invention, feedforward control for changing the voltage phase of a rectangular wave voltage applied to an AC motor based on a torque command value can be commonly applied over a wide torque control range without incurring complication of arithmetic processing. Such a control configuration can be provided.

1 is an overall configuration diagram of an AC motor control system according to an embodiment of the present invention. FIG. It is a conceptual diagram explaining the control system used for the power conversion in the inverter in the motor control system shown in FIG. It is a conceptual diagram which shows the rough relationship between the driving | running state of an alternating current motor, and control mode. It is a conceptual diagram which shows the correspondence of the voltage phase and torque in rectangular wave voltage control. It is a functional block diagram for demonstrating the specific control structure of rectangular wave voltage control. It is a conceptual diagram explaining the calculation operation | movement of the amount of phase changes based on the linear approximation by the feedforward control part shown in FIG. It is a conceptual diagram explaining the repetition process of the linear approximation calculation by the feedforward control part shown in FIG. It is a functional block diagram for demonstrating the detailed structure of the feedforward control part shown in FIG. It is a 1st conceptual diagram for demonstrating operation | movement of the polarity check part shown in FIG. It is a 2nd conceptual diagram for demonstrating operation | movement of the polarity check part shown in FIG. It is a flowchart which shows the control processing procedure of the rectangular wave voltage control by this Embodiment. It is a flowchart which shows the detailed control processing procedure of the step which calculates the feedforward term shown in FIG. It is a flowchart which shows the detailed control processing procedure of the step which calculates the phase variation shown in FIG.

  Embodiments of the present invention will be described below in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

FIG. 1 is an overall configuration diagram of an AC motor control system according to an embodiment of the present invention.
Referring to FIG. 1, motor control system 100 includes a DC voltage generation unit 10 #, a smoothing capacitor C0, an inverter 14, a control device 30, and an AC motor M1.

  AC electric motor M1 is a driving electric motor that generates torque for driving driving wheels of an electric vehicle such as a hybrid vehicle, an electric vehicle, or a fuel cell vehicle. In other words, in the present embodiment, the electric vehicle includes all vehicles equipped with a motor for generating wheel driving force, including an electric vehicle not equipped with an engine. Note that AC motor M1 is generally configured to have both functions of an electric motor and a generator. Further, this AC motor M1 may be configured to have a function of a generator driven by an engine in a hybrid vehicle. Further, AC electric motor M1 may operate as an electric motor for the engine, and may be incorporated in a hybrid vehicle as one that can start the engine, for example.

  DC voltage generation unit 10 # includes a DC power supply B, system relays SR1 and SR2, a smoothing capacitor C1, and a converter 12.

  The DC power source B is composed of a secondary battery such as nickel metal hydride or lithium ion, a fuel cell, an electric double layer capacitor, or a combination thereof. The DC voltage Vb output from the DC power source B is detected by the voltage sensor 10. Voltage sensor 10 outputs detected DC voltage Vb to control device 30.

  System relay SR 1 is connected between the positive terminal of DC power supply B and power line 6, and system relay SR 2 is connected between the negative terminal of DC power supply B and ground line 5. System relays SR1 and SR2 are turned on / off by signal SE from control device 30. Smoothing capacitor C <b> 1 is connected between power line 6 and ground line 5.

  Converter 12 includes a reactor L1, power semiconductor switching elements Q1, Q2, and diodes D1, D2.

  Power semiconductor switching elements Q 1 and Q 2 are connected in series between power line 7 and ground line 5. On / off of power semiconductor switching elements Q 1 and Q 2 is controlled by switching control signals S 1 and S 2 from control device 30.

In the embodiment of the present invention, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, or a power bipolar transistor is used as a power semiconductor switching element (hereinafter simply referred to as “switching element”). Etc. can be used. Anti-parallel diodes D1 and D2 are arranged for switching elements Q1 and Q2.

  Reactor L1 is connected between a connection node of switching elements Q1 and Q2 and power line 6. Further, the smoothing capacitor C 0 is connected between the power line 7 and the ground line 5.

  Inverter 14 includes a U-phase arm 15, a V-phase arm 16, and a W-phase arm 17 provided in parallel between power line 7 and ground line 5. Each phase arm is composed of a switching element connected in series between the power line 7 and the ground line 5. For example, U-phase arm 15 includes switching elements Q3 and Q4, V-phase arm 16 includes switching elements Q5 and Q6, and W-phase arm 17 includes switching elements Q7 and Q8. Further, antiparallel diodes D3 to D8 are connected to switching elements Q3 to Q8, respectively. Switching elements Q3 to Q8 are turned on / off by switching control signals S3 to S8 from control device 30.

  An intermediate point of each phase arm is connected to each phase end of each phase coil of AC electric motor M1. Typically, AC electric motor M1 is a three-phase permanent magnet motor, and is configured such that one end of three coils of U, V, and W phases are commonly connected to a midpoint. Furthermore, the other end of each phase coil is connected to the midpoint of the switching elements of each phase arm 15-17.

  During the boosting operation, the converter 12 boosts the DC voltage Vb supplied from the DC power supply B (this DC voltage corresponding to the input voltage to the inverter 14 is hereinafter also referred to as “system voltage”) VH to the inverter 14. To supply. In the step-down operation, converter 12 steps down the DC voltage (system voltage) supplied from inverter 14 via smoothing capacitor C0 and charges DC power supply B. During the step-up operation and the step-down operation, on / off of switching elements Q1, Q2 is controlled in response to switching control signals S1, S2 from control device 30, respectively. If switching elements Q1 and Q2 are fixed to ON and OFF, respectively, VH = Vb (voltage ratio = 1.0) can be obtained.

  Smoothing capacitor C 0 smoothes the DC voltage from converter 12 and supplies the smoothed DC voltage to inverter 14. The voltage sensor 13 detects the voltage across the smoothing capacitor C 0, that is, the system voltage VH, and outputs the detected voltage to the control device 30.

  When the torque command value of AC motor M1 is positive (Tqcom> 0), inverter 14 responds to switching control signals S3 to S8 from control device 30 when a DC voltage is supplied from smoothing capacitor C0. The AC motor M1 is driven so as to output a positive torque by converting the DC voltage into an appropriate motor applied voltage (AC voltage) by the switching operation of the elements Q3 to Q8. Further, when the torque command value of the AC motor M1 is zero (Tqcom = 0), the inverter 14 converts the DC voltage to an appropriate motor applied voltage (AC voltage) by a switching operation in response to the switching control signals S3 to S8. The AC electric motor M1 is driven so that the torque becomes zero by converting to. Thus, AC electric motor M1 is driven to generate zero or positive torque designated by torque command value Tqcom.

  Further, during regenerative braking of an electric vehicle equipped with motor control system 100, torque command value Tqcom of AC electric motor M1 is set to be negative (Tqcom <0). In this case, the inverter 14 converts the AC voltage generated by the AC motor M1 into a DC voltage by a switching operation in response to the switching control signals S3 to S8, and converts the converted DC voltage (system voltage) to the smoothing capacitor C0. To the converter 12 via The regenerative braking here refers to braking with regenerative power generation when the driver operating the electric vehicle performs a footbrake operation, or regenerative braking by turning off the accelerator pedal while driving, although the footbrake is not operated. This includes decelerating (or stopping acceleration) the vehicle while generating electricity.

  Current sensor 24 detects a motor current flowing through AC electric motor M <b> 1 and outputs the detected motor current to control device 30. Since the sum of instantaneous values of the three-phase currents iu, iv, and iw is zero, the current sensor 24 has a motor current for two phases (for example, a V-phase current iv and a W-phase current iw) as shown in FIG. It is sufficient to arrange it so as to detect.

  The rotation angle sensor (resolver) 25 detects the rotor rotation angle ANG of the AC electric motor M1 and sends the detected rotation angle ANG to the control device 30. The control device 30 can calculate the rotational speed (represented by the rotational speed per unit time (typically, rpm)) and the rotational angular speed ω (rad / s) based on the rotational angle ANG. Note that the rotation angle sensor 25 may be omitted by directly calculating the rotation angle ANG from the motor voltage or current in the control device 30.

The control device 30 is configured by a CPU (Central Processing Unit) (not shown) and an electronic control unit (ECU) with a built-in memory, and based on a map and a program stored in the memory, an operation using a detection value by each sensor. Perform processing. The control device 30 controls the operation of the motor control system 100 by such arithmetic processing so that the AC motor M1 is operated according to the operation command from the host ECU. Note that a part of the control device 30 may be configured to execute predetermined numerical / logical operation processing by hardware such as an electronic circuit.

  Specifically, the control device 30 determines the torque command value Tqcom, the battery voltage Vb detected by the voltage sensor 10, the system voltage VH detected by the voltage sensor 13, the motor currents iv and iw from the current sensor 24, and the rotation angle. Based on the rotation angle ANG from the sensor 25, the operations of the converter 12 and the inverter 14 are controlled so that the AC motor M1 outputs a torque according to the torque command value Tqcom by a method described later. That is, switching control signals S1 to S8 for controlling converter 12 and inverter 14 as described above are generated and output to converter 12 and inverter 14.

  During the boosting operation of converter 12, control device 30 feedback-controls output voltage VH of smoothing capacitor C0, and generates switching control signals S1 and S2 so that output voltage VH becomes a voltage command value.

  In addition, when control device 30 receives signal RGE indicating that the electric vehicle has entered the regenerative braking mode from the host ECU, switching control signal S <b> 3 to convert the AC voltage generated by AC motor M <b> 1 into a DC voltage. S8 is generated and output to the inverter 14. Thereby, the inverter 14 converts the AC voltage generated by the AC motor M <b> 1 into a DC voltage and supplies it to the converter 12.

  Further, when receiving a signal RGE indicating that the electric vehicle has entered the regenerative braking mode from the host ECU, control device 30 generates switching control signals S1 and S2 so as to step down the DC voltage supplied from inverter 14. And output to the converter 12. As a result, the AC voltage generated by AC motor M1 is converted to a DC voltage, stepped down, and supplied to DC power supply B. Furthermore, control device 30 generates signal SE for turning on / off system relays SR1, SR2 and outputs the signal SE to system relays SR1, SR2.

(Control configuration)
Next, power conversion in the inverter 14 controlled by the control device 30 will be described in detail.

  As shown in FIG. 2, in motor control system 100 according to the embodiment of the present invention, three control modes are switched and used for power conversion in inverter 14.

  The sine wave PWM control is used as a general PWM control, and controls on / off of the switching element in each phase arm according to a voltage comparison between a sine wave voltage command value and a carrier wave (typically a triangular wave). . As a result, for a set of a high level period corresponding to the on period of the upper arm element and a low level period corresponding to the on period of the lower arm element, the duty is set so that the fundamental wave component becomes a sine wave within a certain period. The ratio is controlled. As is well known, in the sine wave PWM control, the fundamental wave component (effective value) of the line voltage applied to the AC motor M1 can be increased only to about 0.61 times the inverter input voltage. Hereinafter, in this specification, the ratio of the fundamental wave component (effective value) of the line voltage of the AC motor M1 to the DC link voltage (that is, the system voltage VH) of the inverter 14 is referred to as “modulation rate”.

  On the other hand, in the rectangular wave voltage control, one pulse of the rectangular wave having a ratio of the high level period to the low level period of 1: 1 is applied to the AC motor M1 within the predetermined period. As a result, the modulation rate is increased to 0.78.

  The overmodulation PWM control performs PWM control similar to the sine wave PWM control in a range where the amplitude of the voltage command is larger than the carrier wave amplitude. In particular, the fundamental wave component can be increased by distorting the voltage command from the original sine wave waveform, and the modulation rate can be increased from the maximum modulation rate in the sine wave PWM control mode to a range of 0.78.

  In the AC motor M1, the induced voltage increases as the rotational speed and the output torque increase, so the required drive voltage (motor required voltage) increases. The boosted voltage by the converter 12, that is, the system voltage VH needs to be set higher than the required voltage of the motor. On the other hand, there is a limit value (VH maximum voltage) in the boosted voltage by converter 12, that is, system voltage VH.

  Therefore, a PWM control mode (sine wave PWM control or overmodulation PWM control) for controlling the amplitude and phase of the motor applied voltage (AC) by feedback of the motor current according to the operating state of AC motor M1, and a rectangular wave voltage One of the control modes is selectively applied. In the rectangular wave voltage control, since the amplitude of the motor applied voltage is fixed, the torque control is executed by the phase control of the rectangular wave voltage pulse based on the deviation between the actual torque value and the torque command value.

FIG. 3 shows a correspondence relationship between the operation state of AC electric motor M1 and the above-described control mode.
Referring to FIG. 3, sine wave PWM control is generally used to reduce torque fluctuation in the low speed region A1, overmodulation PWM control in the medium speed region A2, and rectangular wave in the high speed region A3. Voltage control is applied. In particular, the output of AC electric motor M1 is improved by applying overmodulation PWM control and rectangular wave voltage control. As described above, which of the control modes shown in FIG. 2 is used is basically determined within the range of the modulation rate that can be realized.

  Of the above control modes, any known control configuration can be applied to the sine wave PWM control and overmodulation PWM control. For example, the d-axis and q-axis current command values are obtained from the torque command value Tqcom so that the output torque of the AC motor M1 matches the torque command value Tqcom, and the motor current (Id, Iq) for these current command values is obtained. PWM control can be realized by performing feedback control.

  In the rectangular wave voltage control, the number of times of switching in the inverter 14 (the number of times the switching elements Q3 to Q8 are turned on / off) is greatly reduced, so that efficiency can be improved by reducing switching loss. Therefore, depending on the situation, it is possible to determine the control mode so as to positively select the rectangular wave voltage control even in a low output region to which the sine wave PWM control can be applied.

(Rectangular wave voltage control)
The control system for an AC motor according to the present invention is characterized by the rectangular wave voltage control of the AC motor M1. Therefore, hereinafter, the control configuration of the rectangular wave voltage control will be described in detail.

  The rectangular wave voltage control of the AC motor according to the present embodiment is performed according to the output torque characteristic with respect to the voltage phase θv as shown in FIG.

  Referring to FIG. 4, generally, when a positive torque is generated (Tqcom> 0), the voltage phase θv is advanced when torque is insufficient, while the voltage phase θv is delayed when torque is excessive, according to the torque deviation. Thus, the voltage phase θv is controlled. On the other hand, when negative torque occurs (Tqcom <0), the voltage phase θv is controlled according to the torque deviation so that the voltage phase θv is delayed when the torque is insufficient and the voltage phase θv is advanced when the torque is excessive. The

  As will be described later, the torque characteristics change according to the state of AC electric motor M1. In addition, as shown in FIG. 4, in the region where the voltage phase is small, there is a region where the polarity of the torque change with respect to the change in the voltage phase changes.

FIG. 5 is a functional block diagram for explaining a specific control configuration of rectangular wave voltage control.
Referring to FIG. 5, rectangular wave voltage control unit 400 includes power calculation unit 410, torque calculation unit 420, deviation calculation unit 425, feedback control unit 430, feedforward control unit 440, and addition unit 450. The rectangular wave generator 460 and the signal generator 470 are included.

  Note that each functional block in FIG. 5 is realized by a control program executed by a predetermined program executed by the control device 30 and / or an electronic circuit (hardware) in the control device 30. Then, when the rectangular wave voltage control is applied, the rectangular wave voltage control according to FIG. 5 is executed every predetermined control cycle.

  The power calculation unit 410 uses each phase current obtained from the V-phase current iv and the W-phase current iw detected by the current sensor 24 and each phase (U-phase, V-phase, W-phase) voltage Vu, Vv, Vw. Then, power supplied to the motor (motor power) Pmt is calculated according to the following equation (1).

Pmt = iu · Vu + iv · Vv + iw · Vw (1)
The torque calculation unit 420 uses the rotation angular velocity ω calculated from the motor power Pmt obtained by the power calculation unit 410 and the rotation angle ANG of the AC motor M1 detected by the rotation angle sensor 25 according to the following equation (2). A torque estimated value Tq is calculated.

Tq = Pmt / ω (2)
Note that the estimated torque value Tq is not limited to the estimation method by the power calculation unit 410 and the torque calculation unit 420, but a point that can be obtained by an arbitrary method will be described. Alternatively, the estimated torque value Tq may be obtained by arranging a torque sensor instead of the power calculation unit 410 and the torque calculation unit 420.

  Deviation calculation unit 425 calculates torque deviation ΔTq (ΔTq = Tqcom−Tq) according to estimated torque value Tq and torque command value Tqcom.

  Based on the torque deviation ΔTq, the feedback control unit 430 calculates a control deviation by performing a proportional integration (PI) operation with a predetermined gain, and calculates a feedback term θfb of the rectangular wave voltage phase according to the obtained control deviation. Specifically, as shown in FIG. 4, when positive torque is generated (Tqcom> 0), the voltage phase is advanced when torque is insufficient, while when the torque is excessive, the voltage phase is delayed and negative torque is generated (Tqcom <0). ) Sometimes the feedback term θfb is calculated so that the voltage phase is delayed when the torque is insufficient, while the voltage phase is advanced when the torque is excessive.

  According to the feedback control unit 430, the feedback term θfb set by the feedback control based on the torque deviation is obtained. However, since the operation amount is only the voltage phase in the rectangular wave voltage control, the control response is relatively lowered as compared with the PWM control in which the amplitude and phase of the motor applied voltage can be the operation amount. Further, in the power calculation (equation (1)) in the power calculation unit 410, filter processing for removing noise and the like from the detected motor current value is unavoidable. It becomes difficult to ensure sufficient control response.

  The feedforward control unit 440 reflects the rotational speed Nm and the system voltage VH as variables related to the state of the AC motor M1 (hereinafter also referred to as motor variables), and feeds to respond to changes in the torque command value Tqcom. Set the forward term θff. The rotation speed Nm can be calculated from the rotation angle ANG of the AC motor M1 detected by the rotation angle sensor 25. System voltage VH can be detected by a voltage detected by voltage sensor 13 or a voltage command value VH #.

  Adder 450 sets voltage phase θv corresponding to the phase command of the rectangular wave voltage in accordance with the addition of feedback term θfb by feedback controller 430 and feedforward term θff by feedforward controller 440.

  The rectangular wave generator 460 generates each phase voltage command value (rectangular wave pulse) Vu, Vv, Vw according to the voltage phase θv set by the adding unit 450. The signal generator 470 generates switching control signals S3 to S8 according to the phase voltage command values Vu, Vv, and Vw. When inverter 14 performs a switching operation according to switching control signals S3 to S8, a rectangular wave voltage according to voltage phase θv is applied as each phase voltage of the motor.

  If comprised in this way, control responsiveness can be improved by respond | corresponding to the change of torque command value Tqcom by feedforward control. Furthermore, the offset steady-state deviation can be eliminated by the combination with the feedback control.

  However, in the feedforward control, the setting of the feedforward term θff with respect to changes in the torque command value Tqcom and the motor variable becomes a problem. Generally, in AC motor M1, torque change (torque characteristics) with respect to voltage phase is a non-linear characteristic, so that the computation time is within the control cycle, or the computation load in each control cycle is excessively increased. However, it is a problem to obtain the feedforward term θff. Although it is possible to map the nonlinear characteristics in advance, if the number of map points is increased to improve the control accuracy, the map storage data becomes enormous. Concerned about occupying.

  Therefore, in this embodiment, the feedforward control unit 440 is configured as follows, thereby reducing the calculation load and the storage data capacity required for the feedforward control.

  In the description of the feedforward control according to the present embodiment, first, the characteristics of the output torque with respect to the motor variable indicating the motor state and the voltage phase (hereinafter simply referred to as “torque characteristics”) will be described.

  The torque characteristic reflecting the motor state is grasped by a torque calculation formula described below. As is generally known, voltage equations and torque equations on d-axis and q-axis in a permanent magnet type synchronous motor are expressed by the following equations (3) to (5).

  In the expressions (3) and (4), Ra represents the armature winding resistance, Ψ represents the number of armature linkage fluxes of the permanent magnet, and P represents the number of pole pairs of the AC motor M1. Further, ω represents the electrical angular velocity of the AC motor M1. The electrical angular velocity ω can be obtained by ω = 2π · (Nm / 60) · P) using the motor rotation speed Nm (rpm).

  The voltage component depending on the winding resistance contributes in a very low speed region, and other components become dominant as the rotational speed increases. For this reason, when the rectangular wave voltage control is applied in the high speed region (FIG. 2), the winding resistance component in the equations (3) and (4) can be ignored. Therefore, the above equations (4) and (5) are expressed by the following equations (6) and (7) when the rectangular wave voltage control is applied.

  Furthermore, when the rectangular wave voltage control is performed, considering that the fundamental wave component of the motor applied voltage (line voltage) indicated by the d-axis voltage and the q-axis voltage is 0.78 times the system voltage VH, (6) By applying the expressions (7) and (7) to the above expression (3), the torque calculation expression (8) indicating the relationship between the voltage phase θ of the rectangular wave voltage and the output torque T of the AC motor M1 is obtained. Can do.

  As understood from the equation (8), by substituting the motor variables VH and ω (Nm) indicating the motor state into the torque calculation equation, the relationship between the voltage phase θ and the torque T in the current operation state is It is obtained by calculation without referring to the map. In the equation (8), ψ represents a counter electromotive voltage coefficient of the AC motor M1. Since the constants Ka and Kb are fixed in advance as motor constants, the above equation (8) can be transformed into the following equation (9). That is, the equations (8) and (9) are torque calculation equations using the motor variables VH and ω and the voltage phase θ as variables. That is, the torque characteristic line shown in FIG. 4 corresponds to the expression (9) on the voltage phase-torque plane. It is understood that when the motor variables VH and ω change, the torque characteristic line also changes. Equation (9) corresponds to the “torque characteristic equation” in the present invention.

  Further, by differentiating the above equation (9) by the voltage phase θ, the following equation (10) for calculating the ratio Ktl of the torque change to the voltage phase change in the current motor state and voltage phase is derived. .

  FIG. 6 shows a calculation operation of the phase change amount based on the linear approximation by the feedforward control unit 440.

  Referring to FIG. 6, torque characteristic line 500 is derived by substituting the current motor state (motor variables VH, ω) into equation (9). FIG. 6 is an enlarged view of a part of the torque characteristic line 500.

  In addition, the slope Ktl of the tangent TL at each operating point on the torque characteristic line 500 can be obtained according to a model equation in which the same motor variable as in Equation (9) is substituted into Equation (10). The model formula corresponds to the “model formula” of the present invention.

  The operating point Pa (voltage phase θ0) at the start of calculation in the current control cycle is the operating point corresponding to the current voltage phase (that is, voltage phase θv calculated in the previous control cycle) on the torque characteristic line 500. It is. Then, the slope Ktl of the tangent TL at the operating point Pa is obtained by substituting the voltage phase θ0 into the model equation. This gradient Ktl corresponds to the ratio of torque change to voltage phase change at the operating point Pa. That is, by obtaining the tangential slope Ktl, the torque change characteristic with respect to the voltage phase can be linearly approximated.

  When the torque command value Tqcom changes, a torque compensation amount ΔTtl for the current torque command value Tqcom is calculated. For example, the torque compensation amount ΔTtl can be obtained according to the difference between the current torque command value and the previous torque command value. Alternatively, the torque compensation amount ΔTtl may be obtained from the difference between the torque value calculated according to the equation (9) or the calculated value in the torque calculating unit 420 and the current torque command value.

  Then, on the tangent line TL, the phase change amount Δθ for changing the torque by the torque compensation amount ΔTtl from the current operating point Pa can be calculated according to the following equation (11) based on the gradient Ktl. That is, the slope Ktl corresponds to the “first slope”, and Δθ corresponds to the “first phase change amount”.

Δθ = ΔTtl / Ktl (11)
As shown in FIG. 6, the operating point (voltage phase θ1) of AC electric motor M1 when the voltage phase is changed by phase change amount Δθ is not actually operating point Pc on tangent line TL, but a torque characteristic line. The operating point Pd is 500. The torque value at the operating point Pd has an error (Tqcom−Trq) due to linear approximation with respect to the current torque command value Tqcom.

  On the torque characteristic line 500, the operating point Pb originally corresponds to the current torque command value Tqcom. That is, the phase difference of the voltage phase θ0 (initial value) with respect to the voltage phase θq of the operating point Pb corresponds to the phase change amount Δθ that should be originally obtained.

  The feedforward control unit 440 is configured to accurately obtain the amount of phase change by repeating a linear approximation calculation based on the tangential slope a plurality of times. That is, the same linear approximation operation is performed again with the operation point Pd on the torque characteristic line 500 corresponding to the voltage phase θ1 obtained by the linear approximation operation at the operation point Pa (voltage phase θ0) as the new operation point Pa #. Execute. That is, the operating points Pa and Pa # correspond to the “first operating point”.

FIG. 7 shows a linear operation process executed subsequently to FIG.
Referring to FIG. 7, linear operation calculation similar to that for operating point Pa described in FIG. 6 is executed with operating point Pd in FIG. 6 as a new operating point Pa #.

  That is, the slope Ktl of the tangent line TL at the operating point Pa # is newly obtained according to the above model formula in which the motor variable is substituted into the formula (10). Further, torque value Trq at operating point Pa # is obtained according to equation (10), and torque compensation amount ΔTtl = Tqcom−Trq is obtained. As a result, the phase change amount Δθ can be obtained as in FIG. 6, and the operating points Pc # and Pd # corresponding to the operating points Pc and Pd in FIG. 6 can be obtained. The voltage phase θ1 of the operating points Pc # and Pd # is closer to the original voltage phase θq than the voltage phase θ1 in FIG. 6 (voltage phase θ0 in FIG. 7).

  Furthermore, by setting the operating point Pd # in FIG. 7 as a new operating point Pa #, the next linear arithmetic processing can be executed in the same manner as in FIGS. Thus, by repeating the linear approximation calculation so as to update the operating points Pa, Pa # (voltage phase θ0), the updated operating point Pa # converges to the original operating point Pb. That is, the amount of phase change from the current voltage phase (θ0 in FIG. 6) to the voltage phase θq (operating point Pb) is obtained according to the nonlinear characteristic of equation (9) without storing a huge amount of map data. Is possible.

  FIG. 8 is a functional block diagram of the feedforward control unit 440. Each functional block in FIG. 8 is also realized by a control program executed by a predetermined program executed by the control device 30 and / or an electronic circuit (hardware) in the control device 30.

  Referring to FIG. 8, feedforward control unit 440 includes linear approximation unit 442, polarity check unit 444, phase change amount calculation unit 445, torque calculation unit 446, update processing unit 448, and repetition process determination. Part 447.

  The linear approximation unit 442 obtains the tangent slope Ktl at the voltage phase θ0 given from the update processing unit 448 according to the model formula obtained by substituting the current motor variable (VH, Nm) into the formula (10).

  The torque calculation unit 446 calculates the torque value Trq at the voltage phase θ0 given from the update processing unit 448 according to Expression (9) into which the current motor variable (VH, Nm) is substituted.

  The polarity check unit 444 determines whether or not the tangent slope Ktl obtained by the linear approximation unit 442 matches the assumed polarity, and if it matches, the linear approximation unit 442 The calculated Ktl is output to the phase change amount calculation unit 445 as it is.

  On the other hand, when Ktl calculated by the linear approximation unit 442 does not match the assumed polarity, the slope Ktl is corrected to a predetermined value kc (Ktl = kc). Then, the polarity check unit 444 outputs the corrected gradient Ktl to the phase change amount calculation unit 445. Here, the polarity of the predetermined value kc is the same as the assumed coefficient.

  The phase change amount calculation unit 445 obtains the phase change amount Δθ according to the equation (11) based on the tangent slope Ktl processed by the polarity check unit 444 and the torque compensation amount ΔTtl. The torque compensation amount ΔTtl is obtained based on the torque command value Tqcom and / or the torque value Trq by the torque calculator 446. As described above, in the first linear approximation calculation, the torque compensation amount ΔTtl may be obtained according to the change amount of the torque command value Tqcom without being subjected to the calculation by the torque calculation unit 446.

  The update processing unit 448 sets the voltage phase θv at the start of calculation in the current control cycle (that is, the voltage phase θv set in the previous control cycle) to the initial value of the voltage phase θ0 and calculates the phase change amount. Each time the phase change amount Δθ is calculated by the unit 445, θ0 is updated (θ0 ← θ0 + Δθ). As a result, the operating points Pa and Pa # (first operating point) to be subjected to linear approximation are updated.

  The iterative process determination unit 447 determines whether or not to end the iterative process every time the phase change amount calculation unit 445 calculates the phase change amount Δθ. Then, a flag Frp indicating the determination result is generated. The flag Frp is turned on when a predetermined end condition is satisfied, and is turned off until the end condition is satisfied.

  For example, the flag Frp is turned on when the number of updates of the operating points Pa and Pa # by the update processing unit 448 (that is, the number of repetitions of the linear approximation calculation) exceeds a predetermined number. Alternatively, the calculation may be performed without limiting the number of repetitions, and the flag Frp may be turned on when a predetermined time has elapsed since the start of the current control cycle.

  In addition, each time the phase change amount Δθ is calculated, the iterative processing determination unit 447 determines whether or not | Δθ | is smaller than the predetermined value ε, so that the flag Frp is set when | Δθ | <ε is satisfied. It may be configured to turn on. Alternatively, when the torque value Trq corresponding to the updated voltage phase θ0 is calculated by the torque calculation unit 446, the flag Frp is turned on when the deviation between the torque value Trq and the torque command value Tqcom becomes smaller than a predetermined value. May be.

  When the flag Frp is off, the phase change by the linear approximation unit 442, the polarity check unit 444, the torque calculation unit 446, and the phase change amount calculation unit 445 is performed using the voltage phase θ0 updated by the update processing unit 448. Calculation of the amount Δθ (linear approximation calculation) is executed. That is, in response to the update of the operating points Pa and Pa #, an iterative process for calculating the phase change amount Δθ at the updated operating points Pa and Pa # by linear approximation is continuously executed.

  On the other hand, when the flag Frp is turned on, the repetition process is terminated. The updated voltage phase θ0 at that time is set as a target phase by feedforward control. Therefore, phase change amount θff by feedforward control unit 440 is indicated by the difference between voltage phase θv set in the previous control cycle and voltage phase θ0 (final value) of operating point Pa # after the last update. , Θff = θ0 (final value) −θv. That is, θff is equal to the integral value of the phase change amount Δθ obtained in each of the linear approximation calculations that are repeatedly processed in the current control cycle.

Here, the operation of the polarity check unit 444 will be described with reference to FIGS. 9 and 10.
Referring to FIG. 9, torque characteristic line 500 takes an extreme value in a region where the voltage phase (absolute value) is small due to the influence of the term sin 2θ in equation (9). That is, the polarity (positive / negative) of the tangential slope of the torque characteristic line 500 is not the same, and unlike the other regions, there are inversion regions 600 and 610 where the torque decreases as the voltage phase increases.

  Here, the control operation in the case where the torque command value Tqcom has increased to Ty at the operating point Px in the inversion region 600 will be considered. In this case, the feedforward control should act so that the voltage phase changes toward the voltage phase θy of the operating point Py corresponding to the torque value Ty.

  However, since the gradient of the tangent line of the torque characteristic line 500 is negative in the inversion region 600, in the repeated processing of the linear approximation calculation as described in FIGS. There is a high possibility that That is, it is difficult to perform feedforward control that brings the voltage phase closer to θy.

  Further, when the torque is increased across the reversal region 600, for example, the case where the torque command value Tqcom is increased from the operating point Pz to Ty, the convergence to the operating point Pw is performed in the above-described linear approximation calculation repeated processing. Thus, feedforward control may be performed.

  That is, due to the presence of the inversion regions 600 and 610 in the low torque region, it is difficult to perform feedforward control in which the linear approximation calculation repeated processing as described in FIGS. 6 to 8 is commonly applied over the entire voltage phase region. It is understood that

  Depending on the definition of the voltage phase, the polarity of the voltage phase-torque characteristic is reversed each time the AC motor M1 rotates forward and reverse. For example, while showing the characteristics of FIG. 9 during forward rotation, the torque characteristic line 500 during reverse rotation has torque decreasing as the voltage phase increases, as shown in FIG. However, also in FIG. 10, the torque does not monotonously decrease as the voltage phase increases over the entire voltage phase region, but inversion regions 600 # and 610 # in which the torque increases as the voltage phase increases are shown in FIG. The inversion regions 600 and 610 exist in the same manner.

  Here, as understood from the equation (9), when the motor variable (VH, ω) changes, the torque characteristic line 500 also changes, so that the inversion regions 600 and 610 (600 #, 610 #) also change. In addition, since there are errors in the motor constants Ka and Kb in Equation (9), it is difficult to accurately identify the inversion regions 600 and 610 and reflect them in the calculation.

  In the rectangular wave voltage control according to the present embodiment, in order to deal with the inversion regions 600 and 610 (600 # and 610 #) by a simple calculation process, a polarity check unit 444 is provided to execute a linear approximation calculation repetition process. To do.

  As described above, in FIG. 9, in the regions other than the inversion regions 600 and 610, the slope of the tangent line of the torque characteristic line 500 is positive, and thus the assumed polarity is “positive”. Therefore, the polarity check unit 444 forcibly corrects the gradient Ktl to a predetermined positive value kc when the gradient Ktl calculated by the linear approximation unit 442 becomes a negative value in the inversion regions 600 and 610 of FIG. In the example of FIG. 9, the predetermined value kc needs to be a positive value at a minimum. For example, the torque change coefficient with respect to the voltage phase seen in macro in FIG. 9 (a value corresponding to the overall upward slope) may be used.

  On the other hand, in the case of FIG. 10 (during reverse rotation), the slope of the tangent line of torque characteristic line 500 is negative in the regions other than inversion regions 600 # and 610 #, so the assumed polarity is “negative”. is there. Therefore, the polarity check unit 444 forcibly corrects the gradient Ktl to a negative predetermined value kc when the gradient Ktl calculated by the linear approximation unit 442 becomes a positive value in the inversion regions 600 # and 610 # of FIG. . In the example of FIG. 9, the predetermined value kc needs to be a negative value at a minimum. For example, the change coefficient of the torque with respect to the voltage phase seen as a macro in FIG. 9 (a value corresponding to the overall downward slope) may be used.

  In the examples of FIGS. 9 and 10, the assumed polarity of the gradient Ktl needs to change the “assumed polarity” and the predetermined value kc (polarity) depending on the rotation direction (forward / reverse rotation) of the AC motor M1. There is. However, depending on the definition of the voltage phase, only one of FIG. 9 and FIG. 10 is applied. In this case, the voltage phase-torque characteristic is assumed and the “presumed polarity” and the predetermined value kc (polarity) are obtained. Is fixed.

  By adding the processing of the polarity check unit 444 as described above, the problem described with reference to FIG. 9 can be avoided, and even if the inversion regions 600 and 610 (600 # and 610 #) exist, they can be uniform. The phase change amount Δθ can be correctly obtained by the repetition process of the linear approximation calculation described with reference to FIGS. That is, by applying a common arithmetic logic (configuration of the feedforward control unit 440 shown in FIG. 8) over a wide torque control range, large map data is not stored, calculation is complicated, and processing load is not increased. The feedforward control can be realized.

  Next, a control processing procedure for realizing feedforward control by rectangular wave control according to the embodiment of the present invention will be described with reference to FIGS. Each step of the flowcharts of FIGS. 11 to 13 can be realized by software processing by executing a predetermined program stored in advance by the control device 30 or hardware processing by operating an electronic circuit.

  Referring to FIG. 11, control device 30 obtains torque command value Tqcom of AC electric motor M1 in step S100. In step S110, control device 30 calculates feedback term θfb of voltage phase control by feedback control calculation based on torque deviation ΔTq with respect to torque command value Tqcom. That is, the processing in step S110 corresponds to the function of the feedback control unit 430 in FIG.

  In step S200, control device 30 further sets a feedforward term θff for voltage phase control based on the state (motor variable) of AC electric motor M1 and torque command value Tqcom. That is, the process in step S200 corresponds to the function of the feedforward control unit 440 in FIG.

  Further, in step S130, control device 30 calculates voltage phase θv in the current control cycle according to the sum of feedback term θfb and feedforward term θff. That is, the processing in step S130 corresponds to the function of the adding unit 450 in FIG.

  Then, in step S140, control device 30 performs inverter control command according to voltage phase θv calculated in step S130, specifically, switching command for inverter 14, that is, switching control signal S3 for switching elements Q3 to Q8. ~ S8 is generated. The processing in step S140 corresponds to the functions of the rectangular wave generator 460 and the signal generator 470 in FIG.

  FIG. 12 shows details of the feedforward term calculation in step S200 of FIG.

  Referring to FIG. 12, in step S <b> 210, control device 30 reads the current motor state (the value of the motor variable) to calculate the coefficients of equations (9) and (10). Further, in step S220, control device 30 sets current voltage phase θv (the voltage phase calculated in the previous control cycle) as an initial value of voltage phase θ0. That is, the first operating point Pa is determined.

  In step S230, control device 30 calculates torque compensation amount ΔTtl based on voltage phase θ0 and torque command value Tqcom, and calculates phase change amount Δθ corresponding to torque compensation amount ΔTtl.

  FIG. 13 shows details of the phase change amount calculation processing in step S230 of FIG.

  Referring to FIG. 13, in step S231, control device 30 calculates tangent slope Ktl at voltage phase θ0 in accordance with equation (10). That is, the processing in step S231 corresponds to the function of the linear approximation unit 442 in FIG.

  Further, in step S232, the control device 30 determines whether or not the polarity of the tangential slope Ktl calculated in step S231 matches the assumed polarity. When the polarities match (YES in S232), the control device 30 maintains the gradient Ktl calculated in step S231 as it is in step S233. On the other hand, when the polarities do not match (NO in S232), control device 30 corrects inclination Ktl to predetermined value kc in step S234. At this time, the predetermined value kc is different in polarity from the gradient Ktl calculated in step S232.

  In this way, the processing in steps S232 to 234 corresponds to the function of the polarity check unit 444 in FIG. As described above, the “assumed polarity” in step S232 varies depending on the rotation direction (forward / reverse rotation) of AC electric motor M1, depending on the definition of the voltage phase.

  Further, in step S236, control device 30 calculates torque compensation amount ΔTtl for current torque command value Tqcom, and in step S237, phase for canceling torque compensation amount ΔTtl on tangent line TL of torque characteristic line 500 is calculated. A change amount Δθ is obtained. That is, Δθ = ΔTtl / Ktl is calculated. As described above, the processing in steps S236 and 238 corresponds to the function of the phase change amount calculation unit 445 in FIG.

  Referring to FIG. 12 again, in step S240, control device 30 updates voltage phase θ0 in accordance with phase change amount Δθ obtained in step S230 (θ0 ← θ0 + Δθ). Further, in step S250, control device 30 calculates torque value Tq at updated voltage phase θ0 on torque characteristic line 500 according to equation (9). As a result, the operating points Pa and Pa # (first operating point) to be subjected to the next linear approximation calculation are updated.

  In step S260, control device 30 determines whether or not to repeat the linear approximation calculation iterative process according to a predetermined end condition. The process in step S260 corresponds to the function of the iterative process determination unit 447 in FIG. When the end condition is satisfied, that is, when the flag Frp is turned on by the repetitive process determination unit 447, step S260 is determined as YES, while when it is not (when the flag Frp is turned on), step S260 is A NO determination is made.

  At the time of NO determination in step S260, control device 30 performs a linear approximation operation (step S230) for the updated operating points Pa and Pa # using the updated voltage phase θ0 in step S240. As a result, the linear approximation calculation is repeatedly performed with the update of the operating points Pa and Pa # (voltage phase θ0) until the end condition is satisfied.

  On the other hand, when YES is determined in step S260, control device 30 proceeds to step S270 to follow the difference between the final value of voltage phase θ0 (first voltage phase) and the initial voltage phase θv (step S220). The feedforward term θff is calculated. That is, the feedforward term θff is equal to the integral value of the phase change amount Δθ (S230) calculated for each repetition process in the current control cycle.

  As described above, the feedforward control shown in FIG. 8 can also be realized by the control processing procedure shown in FIGS.

  As described above, according to the control apparatus for an AC motor according to the embodiment of the present invention, the phase change amount by the feedforward control that changes the voltage phase of the rectangular wave voltage applied to the AC motor M1 based on the torque command value. Can be obtained in common over a wide torque control range by simple arithmetic processing. Thereby, it is possible to realize feedforward control in which common arithmetic logic is applied over a wide torque control range without enormous storage of map data. As a result, it is possible to avoid an insufficient storage capacity of the control device 30 and an increase in processing load due to complicated computation.

  In the present embodiment, as a preferred configuration example, a configuration is shown in which DC voltage generation unit 10 # of the motor control system includes converter 12 so that the input voltage (system voltage VH) to inverter 14 can be variably controlled. However, as long as the input voltage to inverter 14 can be variably controlled, DC voltage generation unit 10 # is not limited to the configuration exemplified in the present embodiment. Further, it is not always indispensable that the inverter input voltage is variable. The present invention is also applied to a configuration in which the output voltage of the DC power supply B is directly input to the inverter 14 (for example, a configuration in which the converter 12 is omitted). Applicable.

  Further, the torque characteristic formula is not limited to the formula (9). That is, the torque characteristic formula (torque characteristic line 500) can be configured by other torque calculation formulas including the reflected motor variables. That is, if the employed torque characteristic equation has a non-linear characteristic and does not increase / decrease monotonously (that is, has an extreme value) over the entire range of the voltage phase, the rectangular wave voltage according to the present embodiment. It is possible to apply control feedforward control.

  Further, in the present embodiment, the combination of the feedforward control and the feedback control is shown as a preferable example, but the point that the combination with the feedback control is not necessarily described will be described in a confirming manner.

  As for the AC motor serving as a load of the motor control system, in the present embodiment, a permanent magnet motor mounted for driving a vehicle in an electric vehicle (hybrid vehicle, electric vehicle, etc.) is assumed. The present invention can also be applied to a configuration in which an arbitrary AC motor used in is used as a load.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  The present invention is applied to AC motor control to which rectangular wave voltage control is applied.

  5 Ground wire, 6, 7 Power line, 10, 13 Voltage sensor, 10 # DC voltage generator, 12 Converter, 14 Inverter, 15 U-phase arm, 16 V-phase arm, 17 W-phase arm, 24 Current sensor, 25 Rotation angle Sensor, 30 control unit (ECU), 100 motor control system, 400 rectangular wave voltage control unit, 410 power calculation unit, 420 torque calculation unit, 425 deviation calculation unit, 430 feedback control unit, 440 feedforward control unit, 442 linear approximation Unit, 444 polarity check unit, 445 phase change amount calculation unit, 446 torque calculation unit, 447 repetitive processing determination unit, 448 update processing unit, 450 addition unit, 460 rectangular wave generator, 470 signal generation unit, 500 torque characteristic line , 600, 600 #, 610, 610 # Inversion region, ANG low Rotation angle, B DC power supply, C0, C1 smoothing capacitor, D1-D8 antiparallel diode, Frp flag (repetition end condition), iu, iv, iw three-phase current, Ka, Kb motor constant, kc predetermined value (slope) , Ktl slope (tangent), L1 reactor, M1 AC motor, Nm rotation speed, Pa, Pb, Pc, Pd, Pw, Px, Py, Pz operating point, Pmt motor power, Q1-Q8 power semiconductor switching element, S1 ~ S8 Switching control signal, SE signal, SR1, SR2 System relay, T torque, TL tangent, Tq torque estimated value, Tqcom torque command value, Trq torque value (torque calculation formula), Vb DC voltage (battery voltage), VH DC Voltage (system voltage), VH # voltage command value, VH, ω Motor variable, Vu, Vv, Vw Phase voltage command value, ΔTq torque deviation (FB control), ΔTtl torque compensation amount (FF control), Δθ phase change amount, θ0 voltage phase (first voltage phase), θ1, θq, θv, θy voltage phase, θfb feedback Term, θff feedforward term, ω electrical angular velocity.

Claims (8)

An inverter that converts a DC voltage into an AC voltage for rotationally driving an AC motor;
A rectangular wave voltage control unit configured to control a voltage phase of a rectangular wave voltage applied from the inverter to the AC motor;
The rectangular wave voltage control unit
The voltage phase at the first operating point on the torque characteristic line indicated by the torque characteristic equation according to a model equation in which the current motor state is substituted into a differential equation obtained by differentiating the predetermined torque characteristic equation of the AC motor with a voltage phase. A linear approximation unit configured to calculate a first slope that is a ratio of torque change to change in
When the polarity of the first gradient calculated by the linear approximation unit matches the assumed polarity, the first gradient is maintained as it is, while when the polarity does not match, the predetermined polarity of the assumed polarity is maintained. A polarity checker for processing to correct the first slope to a value;
A first phase change amount is calculated by calculating a torque compensation amount at the first operating point based on a torque command value and dividing the torque compensation amount by the first slope processed by the polarity check unit. A phase change amount calculation unit configured to calculate
Updating the first operating point according to the first phase change amount calculated by the phase change amount calculation unit, and in response to the update process, the linear approximation unit, the polarity check unit, and the phase change An update processing unit configured to instruct a repetition process for calculating the first phase change amount at the first operating point updated by a quantity calculation unit;
A repetition process determination unit that determines whether or not the end condition of the repetition process is satisfied,
When it is determined that the end condition is satisfied, the update processing unit further outputs an output torque of the AC motor based on the updated voltage phase of the first operating point at the time. A control apparatus for an AC motor configured to determine a change amount of the voltage phase for matching with a value.   The control apparatus for an AC motor according to claim 1, wherein the polarity check unit changes the assumed polarity according to a rotation direction of the AC motor.   The control apparatus for an AC motor according to claim 1, wherein the repetitive processing determination unit determines that the end condition is satisfied when the number of updates of the first operating point reaches a predetermined number.   The control apparatus for an AC motor according to claim 1, wherein the repetitive process determination unit determines that the end condition is satisfied when a predetermined time elapses from the start of control calculation.   When the difference between the torque value calculated according to the torque characteristic equation and the torque command value at the first operating point updated by the update processing unit is smaller than a predetermined value, the repetition processing determination unit ends the process. The control apparatus for an AC motor according to claim 1 or 2, wherein it is determined that the condition is satisfied.   The repetition processing determination unit determines that the end condition is satisfied when an absolute value of the first phase change amount calculated by the phase change amount calculation unit is smaller than a predetermined determination value. 3. The control apparatus for an AC motor according to 1 or 2. The rectangular wave voltage control unit
A feedback control unit configured to control a voltage phase of the rectangular wave voltage based on feedback of a torque deviation with respect to the torque command value;
The calculation part which further sets the command value of the said voltage phase according to the sum of the said voltage phase set by the said feedback control part and the said variation | change_quantity calculated by the said update process part is further included. The control apparatus of the alternating current motor of any one of these.   The control apparatus for an AC motor according to any one of claims 1 to 7, wherein the AC motor is configured to be mounted on an electric vehicle and generate a vehicle driving force of the electric vehicle.

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