WO2015025622A1 - Ac electric motor control device, ac electric motor drive system, fluid pressure control system, and positioning system - Google Patents

Ac electric motor control device, ac electric motor drive system, fluid pressure control system, and positioning system Download PDF

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Publication number
WO2015025622A1
WO2015025622A1 PCT/JP2014/067664 JP2014067664W WO2015025622A1 WO 2015025622 A1 WO2015025622 A1 WO 2015025622A1 JP 2014067664 W JP2014067664 W JP 2014067664W WO 2015025622 A1 WO2015025622 A1 WO 2015025622A1
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Prior art keywords
motor
voltage
phase
voltage command
current
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PCT/JP2014/067664
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French (fr)
Japanese (ja)
Inventor
岩路 善尚
高畑 良一
誠己 羽野
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日立オートモティブシステムズ株式会社
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Priority to JP2015532756A priority Critical patent/JP6129972B2/en
Publication of WO2015025622A1 publication Critical patent/WO2015025622A1/en

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    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00Arrangements or methods for the control of AC motors characterised by the kind of supply voltage

Definitions

  • the present invention relates to an AC motor control device, and an AC motor drive system, fluid pressure control system, and positioning system using the same.
  • PMSM permanent magnet synchronous motors
  • phase current detection method of Patent Document 2 two continuous peaks or valleys of a triangular wave carrier signal are obtained in two-phase modulation type pulse width modulation (PWM) in which two-phase switching elements are switched among three phases within one cycle.
  • PWM pulse width modulation
  • basic voltage vector components whose phases are different from each other by 180 degrees are alternately output.
  • the current flowing through the shunt resistor provided on the DC bus of the inverter is detected at this timing, converted from analog to digital, and taken into the microcomputer. Thereby, the timing of current detection is fixed, and an inexpensive microcomputer can be used.
  • the current flowing through the shunt resistor can be detected at the timing of the peak or valley of the triangular wave carrier signal.
  • the voltage command vector is corrected and output so as to be larger than the original voltage command vector
  • the subsequent PWM period the first By outputting a voltage command vector that is 180 degrees out of phase with the voltage command vector in the period, the voltage command vector is matched with the original voltage command vector. Therefore, the current ripple in the first period increases, and as a result, the current detection value includes a large detection error. Therefore, the problem that the accuracy of torque control deteriorates cannot be overcome.
  • An object of the present invention is to provide an AC motor control device capable of detecting a phase current with high accuracy using an inexpensive microcomputer in view of the problems in the conventional phase current detection method as described above.
  • the control device for an AC motor according to the first aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and the inverter is switched to a switching state in which three phases are provided on each of the positive electrode side and the negative electrode side. Accordingly, the driving of the AC motor is controlled.
  • the inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements.
  • the voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all switching elements on the positive electrode side are OFF and all switching elements on the negative electrode side are ON are included.
  • the AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After outputting the zero vector, the inverter is controlled so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector, thereby controlling the driving of the AC motor.
  • the AC motor control device of the first aspect repeatedly outputs the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector sequentially.
  • the control apparatus for an AC motor according to the first or second aspect provides a direct current that flows through the direct current bus of the inverter during the output period of the first voltage vector and the third voltage vector. It is preferable to provide a current estimation unit that obtains the phase current of the AC motor based on the detection result.
  • the AC motor control device includes a voltage command generation unit that generates a three-phase voltage command, and a voltage correction unit that corrects the voltage command.
  • a PWM generation unit that compares the corrected voltage command corrected by the voltage correction unit with a triangular wave carrier having a predetermined period and generates a three-phase PWM waveform based on the comparison result, and generates a PWM. It is preferable to control the driving of the AC motor by controlling the switching state of the switching element by outputting the PWM waveform generated by the unit to the inverter.
  • the voltage correction unit is an intermediate voltage with a three-phase voltage command in the first period of two consecutive periods of the triangular wave carrier. The command is set to 0, and the voltage command is corrected by shifting the maximum voltage command and the minimum voltage command with the three-phase voltage command.
  • the minimum voltage command is set to 0. Further, it is preferable to correct the voltage command by shifting the maximum voltage command and the intermediate voltage command.
  • the voltage correction unit has a maximum value based on a difference between the maximum voltage command and the intermediate voltage command in the first half of the first cycle. And determining the shift amount for the minimum voltage command based on the difference between the minimum voltage command and the intermediate voltage command and a predetermined required shift amount, and in the latter half of the first cycle, The shift amount for the maximum voltage command is determined so that the corrected voltage command becomes 0, and the shift amount for the minimum voltage command is determined based on the necessary shift amount.
  • the maximum shift amount is determined.
  • the shift amount for the maximum voltage command is determined based on the difference between the voltage command and the minimum voltage command, and the shift amount for the intermediate voltage command is determined based on the required shift amount.
  • the shift amount with respect to the maximum voltage command is determined so that the corrected voltage command becomes 0, and the intermediate voltage is determined based on the difference between the intermediate voltage command and the minimum voltage command and the required shift amount. It is preferable to determine the shift amount with respect to the command.
  • the required shift amount is from when the switching state of the switching element changes until the DC current flowing through the DC bus of the inverter is detected.
  • the direct current that flows through the DC bus of the inverter when the triangular wave carrier takes an intermediate value between the minimum value and the maximum value. It is preferable to obtain the detection result of the current.
  • the AC motor control device according to any one of the first to eighth aspects is a two-phase switching element among the switching elements in a low speed range where the AC motor is driven at a low speed.
  • the inverter is controlled so as to perform 120-degree energization drive only by switching the switching state, and in the speed range other than the low speed range, the inverter is configured to perform the three-phase energization drive by switching the switching state of the three-phase switching element. It is preferable to control.
  • the AC motor control device includes a first current controller that outputs a first voltage command used in three-phase energization drive, a 120-degree energization drive, A second current controller that outputs a second voltage command commonly used in three-phase energization drive, and the first current controller is configured to receive a current command input from the outside or a current flowing through the AC motor.
  • An AC motor drive system is connected to the AC motor control device according to any one of the first to tenth modes and the AC motor control device, and has three phases on the positive electrode side and the negative electrode side, respectively.
  • Each having an inverter provided with a switching element and a three-phase AC motor that is driven and controlled according to the switching state of the switching element by an AC motor control device, and the AC motor control device, inverter, and AC motor are: Is integrated.
  • a fluid pressure control system is connected to the control apparatus for an AC motor according to any one of the first to tenth aspects and the control apparatus for the AC motor, and each has three phases on the positive electrode side and the negative electrode side.
  • Three-phase AC motors that are driven and controlled according to the switching state of the switching elements, and inverters that are provided with switching elements and AC motor control devices, and fluid pressure that operates according to the AC motor drive And a pump.
  • a positioning system according to a thirteenth aspect of the present invention is connected to the AC motor control apparatus according to any one of the first to tenth aspects and the AC motor control apparatus, and switches three phases each on the positive electrode side and the negative electrode side.
  • a three-phase AC motor that is driven and controlled according to the switching state of the switching element and an AC motor driven by the inverter provided with the element and the AC motor control device, and adjusts the position of the object A positioning device.
  • FIG. 1 shows the structure of the alternating current motor drive system containing the control apparatus of the alternating current motor which concerns on the 1st Embodiment of this invention. It is a figure which shows the example of the PWM waveform and switch state when not correcting a voltage command. It is a figure which shows the mode of the electric current which each flows into a switching element in each switch state. It is a figure which shows the relationship between the motor current of each phase which flows into an alternating current motor, the shunt current which flows into a direct current bus, and a switch state. It is a figure which shows the example of a PWM waveform at the time of correcting a voltage command based on the conventional two-phase modulation system, and a switch state.
  • FIG. 1 shows the structure of the alternating current motor drive system containing the control apparatus of the alternating current motor which concerns on the 1st Embodiment of this invention. It is a figure which shows the example of the PWM waveform and switch state when not correcting a voltage command. It is
  • 6 is a diagram illustrating an example of a PWM waveform and a switch state when a voltage command is corrected by a method different from that in FIG. 5. It is a figure which shows the example of the PWM waveform at the time of correcting the voltage command by this invention, and a switch state. 6 is a table showing a relationship between a voltage vector in a period A and a detected phase current. 6 is a table showing a relationship between a voltage vector in a period B and a detected phase current. It is a figure which shows the mode of the phase current detected as a shunt electric current in the 1st voltage vector and the 3rd voltage vector. It is the figure which represented each voltage vector output from an inverter on the (alpha) (beta) coordinate.
  • FIG. 1 is a diagram showing a configuration of an AC motor drive system including an AC motor control device 2 according to a first embodiment of the present invention.
  • This AC motor drive system includes a q-axis current command generator 1, a control device 2, an inverter 3, and an AC motor 4.
  • the AC motor 4 is a three-phase AC motor, for example, a permanent magnet type synchronous motor (PMSM) is used.
  • PMSM permanent magnet type synchronous motor
  • the q-axis current command generator 1 is a circuit that generates a q-axis current command Iq * corresponding to the output torque of the AC motor 4.
  • the q-axis current command generator 1 is positioned above the control device 2 and normally has a mechanism for generating the q-axis current command Iq * so that the rotational speed ⁇ r of the AC motor 4 becomes a predetermined speed. ing.
  • the q-axis current command Iq * output from the q-axis current command generator 1 is input to the subtracter 6b in the control device 2.
  • the control device 2 controls the driving of the AC motor 4 according to the switching state of each switching element provided in the inverter 3, and the q-axis current command Iq * input from the q-axis current command generator 1 is controlled.
  • the AC motor 4 operates so as to generate a corresponding torque.
  • the control device 2 includes a voltage command generation unit 20, a voltage correction unit 10, a PWM generation unit 11, a current estimation unit 21, a position / speed estimation unit 15, and a speed conversion unit 16.
  • the voltage command generator 20 is a part that generates voltage commands for three phases, that is, U-phase, V-phase, and W-phase, based on the q-axis current command Iq * input from the q-axis current command generator 1.
  • the voltage command generator 20 includes a d-axis current command generator 5, a subtractor 6 a, a subtractor 6 b, a d-axis current controller 7, a q-axis current controller 8, and a dq inverse converter 9.
  • the d-axis current command generator 5 generates a d-axis current command Id * corresponding to the excitation current of the AC motor 4. If the AC motor 4 is a non-salient permanent magnet motor, the d-axis current command Id * is normally zero.
  • the d-axis current command Id * output from the d-axis current command generator 5 is input to the subtractor 6a.
  • the subtractor 6 a is a subtracter for obtaining a d-axis current deviation between the d-axis current command Id * input from the d-axis current command generator 5 and the d-axis current Id estimated by the current estimation unit 21. .
  • This d-axis current deviation is input to the d-axis current controller 7.
  • the subtractor 6b is a subtracter for obtaining a q-axis current deviation between the q-axis current command Iq * input from the q-axis current command generator 1 and the q-axis current Iq estimated by the current estimation unit 21. .
  • This q-axis current deviation is input to the q-axis current controller 8.
  • the d-axis current controller 7 calculates the d-axis voltage command Vd * on the dq coordinate axis so that the d-axis current deviation input from the subtractor 6a becomes zero.
  • the d-axis voltage command Vd * calculated by the d-axis current controller 7 is input to the dq inverse converter 9.
  • the q-axis current controller 8 calculates the q-axis voltage command Vq * on the dq coordinate axis so that the q-axis current deviation input from the subtractor 6b becomes zero.
  • the q-axis voltage command Vq * calculated by the q-axis current controller 8 is input to the dq inverse converter 9.
  • the dq reverse converter 9 receives the d-axis voltage command Vd * and the q-axis voltage Vq * of the dq coordinate axis (magnetic flux axis-magnetic flux axis orthogonal axis) system inputted from the d-axis current controller 7 and the q-axis current controller 8. This is a circuit that converts the voltage command into a three-phase AC coordinate.
  • the dq inverse converter 9 converts the d-axis voltage command Vd * and the q-axis voltage command Vq * into the voltage command Vu * of the three-phase AC coordinate system based on the phase angle ⁇ dc input from the position / velocity estimation unit 15. Convert to Vv * and Vw *.
  • Each voltage command after conversion that is, the U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * is output to the voltage correction unit 10.
  • the voltage command generator 20 can generate three-phase voltage commands Vu *, Vv *, and Vw * by the operation of each component as described above.
  • the voltage correction unit 10 corrects the three-phase voltage commands Vu *, Vv *, and Vw * output from the voltage command generation unit 20, respectively, and corrects the corrected voltage commands Vu **, Vv **, and Output to the PWM generator 11 as Vw **.
  • the voltage command correction method performed by the voltage correction unit 10 will be described in detail later.
  • the PWM generation unit 11 compares the corrected three-phase voltage commands Vu **, Vv **, and Vw ** output from the voltage correction unit 10 with a triangular wave carrier of a predetermined period, and based on the comparison result, Generate phase PWM waveform. From this PWM generator 11, as three-phase PWM waveforms, PWM signals UP, VP, WP corresponding to the U-phase, V-phase, and W-phase, and PWM signals UN, VN, WN is generated. These PWM signals generated by the PWM generator 11 are output from the control device 2 to the inverter 3.
  • the current estimation unit 21 acquires a detection result of the DC current flowing in the DC bus 34 in the inverter 3 based on the DC current signal IDC input from the inverter 3, and based on the detection result, the phase current of the AC motor 4 and This is a part for obtaining the current on the dq coordinate.
  • the current estimation unit 21 includes a sample and hold circuit 12, a current reproducer 13, and a dq converter 14.
  • the sample hold circuit 12 acquires the detection result of the direct current flowing through the direct current bus 34 of the inverter 3 by sampling the direct current signal IDC input from the inverter 3 at a predetermined sampling timing.
  • the sampling timing by the sample and hold circuit 12 is determined based on the triangular wave carrier used in the PWM generator 11. This point will be described in detail later.
  • the current reproducer 13 reproduces the phase currents Iu, Iv, and Iw flowing in the U phase, V phase, and W phase of the AC motor 4 based on the detection result of the DC current acquired by the sample hold circuit 12, respectively.
  • the reproduced values Iuc, Ivc, and Iwc of the phase current are obtained.
  • the phase current reproduction values Iuc, Ivc, and Iwc obtained by the current reproducer 13 are output to the dq converter 14.
  • the dq converter 14 is a circuit that receives the phase current reproduction values Iuc, Ivc, and Iwc from the current reproduction unit 13 and converts them into current values Id and Iq on the dq coordinates.
  • the dq converter 14 converts the reproduction values Iuc, Ivc, Iwc of the three-phase phase currents into the d-axis current Id and the q-axis current Iq based on the phase angle ⁇ dc input from the position / velocity estimation unit 15. To do.
  • the converted values of the d-axis current Id and the q-axis current Iq are output to the subtracters 6a and 6b, respectively, and used for the calculation of the d-axis current deviation and the q-axis current deviation as described above.
  • the current estimation unit 21 obtains the reproduction values Iuc, Ivc, and Iwc of the three-phase phase currents and the values of the d-axis current Id and the q-axis current Iq by the operation of each component as described above. it can.
  • the position / speed estimation unit 15 estimates and calculates the rotor position of the AC motor 4 (when the AC motor 4 is a synchronous motor) or the secondary magnetic flux phase (when the AC motor 4 is an induction motor), and calculates the phase of the calculation result. Output as angle ⁇ dc. Further, based on the value of the phase angle ⁇ dc, the electric angular velocity ⁇ 1 of the AC motor 4 is estimated and calculated, and the calculation result is output. In addition, since these estimation calculation methods are well-known, description is abbreviate
  • the speed converter 16 converts the electrical angular velocity ⁇ 1 output from the position / speed estimator 15 into a rotational speed ⁇ r based on the number P of poles of the AC motor 4.
  • the rotational speed ⁇ r obtained by the speed converter 16 is input to the q-axis current command generator 1 and used to generate the q-axis current command Iq *.
  • the inverter 3 is connected to the AC motor 4, generates an AC voltage based on the PWM signal input from the PWM generator 11 of the control device 2, and supplies the AC voltage to the AC motor 4.
  • the inverter 3 includes a DC power supply 31, an inverter main circuit 32, an output predriver 33, a DC bus 34, and a shunt current detector 35.
  • the DC power supply 31 is connected to the inverter main circuit 32 via the DC bus 34 and generates a DC voltage VDC.
  • the inverter main circuit 32 has three-phase switching elements Sup, Svp, and Swp provided on the positive electrode side, and three-phase switching elements Sun, Svn, and Swn provided on the negative electrode side, respectively.
  • the switching elements Sup, Svp and Swp are connected to the positive electrode side of the DC power supply 31 via the DC bus 34, and the switching elements Sun, Svn and Swn are connected to the negative electrode side of the DC power supply 31 via the DC bus 34. Each is connected.
  • the positive-side switching elements Sup, Svp, Swp and the negative-side switching elements Sun, Svn, Swn AC lines connected to the U-phase, V-phase, and W-phase of the AC motor 4 are connected.
  • the output pre-driver 33 is based on the aforementioned PWM signals UP, VP, WP, UN, VN and WN output from the PWM generator 11 of the control device 2, and the switching elements Sup, Svp, Swp, The switching states of Sun, Svn and Swn are controlled respectively.
  • the DC voltage VDC supplied from the DC power supply 31 is converted into a three-phase AC voltage Vu, Vv, Vw and output to the AC motor 4.
  • the shunt current detector 35 is a circuit for overcurrent protection that detects a direct current flowing through the direct current bus 34, and is provided on the direct current bus 34. The detection result of the direct current by the shunt current detector 35 is output to the current estimation unit 21 of the control device 2 as the above-described direct current signal IDC.
  • the configuration of the AC motor drive system described above is based on a sensorless vector control system, but the present invention can also be applied to vector control with a sensor.
  • the AC motor drive system according to the present embodiment has the configuration as described above.
  • the main characteristic part of the present invention in this configuration is the voltage correction unit 10 for correcting the voltage command and the sampling timing of the DC current signal IDC by the sample hold circuit 12. These will be described in detail below.
  • FIG. 2 is a diagram illustrating an example of a PWM waveform and a switch state when the voltage command is not corrected.
  • the control device 2 it is assumed that, for example, voltage commands Vu *, Vv * and Vw * as shown in FIG.
  • the PWM generation unit 11 compares these voltage commands with a triangular wave carrier, respectively, to thereby generate PWM signals UP, VP, WP as shown in FIG. PWM signals UN, VN, and WN are generated.
  • the switching state of the switching elements Sup, Svp, Swp, Sun, Svn, Swn of the inverter main circuit 32 is changed by the output pre-driver 33 according to the PWM signals UP, VP, WP, UN, VN, WN. Each is controlled. As a result, the switch state of each switching element changes as shown in FIG. In FIG. 2, the switching states of the U phase, the V phase, and the W phase are respectively expressed as voltage vectors in association with 1 or 0 in parentheses.
  • the switching element Sup on the positive electrode side is on (the switching element Sun on the negative electrode side is off) in the U phase, and the switching element Svn on the negative electrode side in the V phase and W phase.
  • Swn is ON (switching elements Svp and Swp on the positive electrode side are OFF).
  • the inverter 3 can output eight kinds of voltage vectors to the AC motor 4 according to combinations of switch states of the switching elements Sup, Svp, Swp, Sun, Svn, and Swn. That is, by turning on all the switching elements Sup, Svp, Swp on the positive side and turning off all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (1, 1, 1) is output. be able to. On the contrary, by turning off all the switching elements Sup, Svp, Swp on the positive side and turning on all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (0, 0 , 0) can be output. These voltage vectors are all zero vectors in which no phase current flows.
  • each of the six types of voltage vectors V (1, 0, 0), V (0, 1, 0), V (0, 0, 1), V (1, 1, 0), V (1, 0, 1) and V (0, 1, 1) can be output.
  • FIG. 3 is a diagram showing the states of currents flowing through the switching elements Sup, Svp, Swp, Sun, Svn, and Swn of the inverter main circuit 32 in the respective switch states indicated by voltage vectors in FIG.
  • FIG. 3A shows the state of current in the voltage vector V (1,1,1)
  • FIG. 3B shows the state of current in the voltage vector V (1,1,0).
  • 3 (c) shows the state of current in the voltage vector V (1, 0, 0)
  • FIG. 3 (d) shows the state of current in the voltage vector V (0, 0, 0). Show.
  • the portion through which the current flows is indicated by a thick line.
  • the voltage vector V (1,1,0) or V (1,0,0) has a DC bus of the inverter 3 via the shunt current detector 35.
  • a direct current shunt current I DC corresponding to the W-phase phase current Iw or the U-phase phase current Iu flows through 34. Therefore, by sampling as described above the direct current signal IDC obtained by detecting the shunt current I DC at switch state corresponding to these voltage vectors, two phase currents Iw, can be detected Iu It becomes.
  • each phase of the motor current flowing to the AC motor 4 (phase currents) Iu, and Iv and Iw, a diagram showing the relationship between the shunt current I DC and the switch state flows through the DC bus 34.
  • the shunt current I DC shown in FIG. 4 (b) flows to the DC bus 34. Accordingly, as shown in FIG. 4 (b), by sampling the shunt current I DC for each switch state, it is possible to detect the phase current Iw, Iu.
  • phase current detection method As described above, it is necessary to perform current sampling and analog-digital conversion twice within a half cycle of the carrier signal. In addition, it is necessary to arbitrarily set the current sampling timing in accordance with the change in the switch state.
  • microcomputers that can realize these functions are generally expensive.
  • current detection error is sampled increases. In particular, such a current detection error tends to include an offset component, which greatly affects torque accuracy.
  • FIG. 5 is a diagram showing an example of a PWM waveform and a switch state when a voltage command is corrected based on the conventional two-phase modulation method
  • FIG. 6 is a diagram different from FIG. It is a figure which shows the example of the PWM waveform at the time of correct
  • the voltage command Vu *, Vv *, and Vw * before correction shown in FIG. 2 is raised without changing their interval, and the highest voltage command (here, Vu *). Is corrected so as to coincide with the apex of the triangular wave carrier to obtain corrected voltage commands Vu **, Vv ** and Vw **.
  • a current is sampled at the timing of the apexes of the triangular wave carrier, it can detect the phase currents Iu of the U phase as a shunt current I DC, can not be detected for the other phase currents.
  • each voltage command Vu *, Vw **, Vw ** is corrected so that the magnitude of the corrected voltage commands Vu ** and Vw ** alternately coincide with the apex of the triangular wave carrier.
  • the corrected voltage commands Vu **, Vv **, and Vw ** obtained by this correction are the intervals between the maximum value and the intermediate value (in the example of FIG. 6, when the voltage command Vu ** is the maximum, the voltage command
  • the interval between Vv ** and the interval between voltage command Vu ** when voltage command Vw ** is maximum is set to be equal to or greater than a predetermined minimum voltage.
  • the interval between the average values is the original voltage command Vu * before correction.
  • Vv * and Vw * are set to be equal to each other.
  • the corrected voltage commands Vu **, Vv **, and Vw ** are biased toward the positive peak direction of the triangular wave carrier. Corrected. In this way, there is a tendency that the current ripples contained in the shunt current I DC increases. Furthermore, since current sampling is performed at the timing of the apex of the triangular wave carrier, a current containing many harmonic ripples is sampled. As a result, there is a problem that current detection error becomes large and torque accuracy is deteriorated.
  • the voltage correction unit 10 corrects the voltage commands Vu *, Vv *, and Vw * by the method described below. Then, the voltage command Vu ** after correction, the DC current signal IDC in accordance with the shunt current I DC flowing through the DC bus 34 of inverter 3 by the PWM signal based on the Vv ** and Vw **, sample and hold circuit 12 The sampling is performed at the timing as described below.
  • FIG. 7 is a diagram showing an example of a PWM waveform and a switch state when the voltage command is corrected according to the present invention.
  • the voltage commands Vu *, Vv *, and Vw * before correction shown in FIG. thus, the corrected voltage commands Vu **, Vv ** and Vw ** are obtained and PWM control is performed.
  • the original voltage command before correction has a relationship of Vu *> Vv *> Vw *.
  • the phase with the maximum voltage command value before correction is the maximum phase (here, U phase)
  • the minimum phase is the minimum phase (here, W phase)
  • the medium phase is the intermediate phase (here, V phase).
  • the correction amount of each voltage command in the period A is such that the sum of the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier, is 0 by subtraction.
  • the PWM generation unit 11 By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in period A, zero vector V (1, 1, 1), voltage vector V (1, 1, 0), voltage vector V (1, 0, 0), zero vector V (0, 0, 0), The voltage vector V (0, 0, 1) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.
  • FIG. 8 is a table showing the relationship between the voltage vector in the period A and the detected phase current.
  • each voltage vector other than the zero vector output in order in the period A is represented as a first voltage vector, a second voltage vector, and a third voltage vector, respectively.
  • the W-phase current Iw is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 1, 0) and the third voltage vector V (0, 0, 1), the signs representing the switch states of the respective phases are inverted, so that the phase current Iw flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I DC, which are not subject to sampling by the sample and hold circuit 12.
  • FIG. 10 in the first voltage vector V (1,1,0) and a third voltage vector V (0,0,1), is a diagram illustrating a state of a phase current is detected as a shunt current I DC .
  • FIG. 10A shows the state of the switch state and the phase current in the first voltage vector V (1,1,0)
  • FIG. 10B shows the third voltage vector V (0,0).
  • , 1) shows the state of the switch and the phase current.
  • the phase current Iw flows from the AC motor 4 to the inverter 3 in the first voltage vector V (1, 1, 0). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the phase current -Iw sign is negative is detected as a shunt current I DC.
  • the phase current Iw flows from the inverter 3 to the AC motor 4 in the third voltage vector V (0, 0, 1). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the code is positive matrix phase current Iw is detected as a shunt current I DC.
  • the current waveform changes greatly, so that the phase current of each phase includes a pulsation component other than the fundamental wave component necessary for control.
  • the phase current of each phase includes a pulsation component other than the fundamental wave component necessary for control.
  • voltage vectors having opposite signs are included in one period of the triangular wave carrier. If the average value is obtained by detecting the phase currents that are applied and flowing through each of them, the pulsation component can be canceled, so that highly accurate detection becomes possible.
  • the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG.
  • the phase currents -Iw and Iw can be detected in accordance with the voltage vector. Thereby, it is possible to detect the phase current Iw excluding the pulsation component.
  • the corrected voltage command Vv ** for the intermediate phase is always corrected to zero.
  • the zero cross timing of the triangular wave carrier i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current ⁇ Iw corresponding to (1, 1, 0) and the phase current Iw corresponding to the third voltage vector V (0, 0, 1) can be detected. Accordingly, in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period A and the downstream section of the triangular wave carrier in the second half of period A. .
  • the correction amount of each voltage command in the period B similarly to the period A described above, the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier.
  • the total is set to be 0 by subtraction. That is, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged over the period B, the intervals between the average values are the original voltage commands Vu *, Vv *, and Vw before correction. It is set to be equal to the interval between *.
  • the PWM generation unit 11 By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in the period B, the zero vector V (1, 1, 1), the voltage vector V (1, 0, 1), the voltage vector V (1, 0, 0), the zero vector V (0, 0, 0), The voltage vector V (0, 1, 0) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.
  • FIG. 9 is a list showing the relationship between the voltage vector in the period B and the detected phase current.
  • the voltage vectors other than the zero vector output in order in the period B are represented as first, second, and third voltage vectors, respectively.
  • the V-phase current Iv is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 0, 1) and the third voltage vector V (0, 1, 0), the signs representing the switch states of the respective phases are inverted, so that the phase current Iv flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I DC, which are not subject to sampling by the sample and hold circuit 12.
  • the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG.
  • the phase currents -Iv and Iv can be detected according to the first and third voltage vectors. Thereby, it is possible to detect the phase current Iv excluding the pulsation component.
  • the corrected voltage command Vw ** for the minimum phase is always corrected to zero.
  • the zero cross timing of the triangular wave carrier i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current ⁇ Iv corresponding to (1, 0, 1) and the phase current Iv corresponding to the third voltage vector V (0, 1, 0) can be detected.
  • in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period B and the downstream section of the triangular wave carrier in the second half of period B. .
  • the sample hold circuit 12 may sample the DC current signal IDC at a fixed timing every half cycle of the triangular wave carrier. Therefore, the sample hold circuit 12 can be realized by using an inexpensive microcomputer.
  • the sample and hold circuit 12 to sample the DC current signal IDC in accordance with the shunt current I DC, it is possible to detect the phase current of two phases. Based on the detection result of the phase current obtained in this way, the current reproduction unit 13 can obtain the reproduction values Iuc, Ivc, and Iwc of the phase current of each phase.
  • FIG. 11 is a diagram illustrating the voltage vectors output from the inverter 3 on the ⁇ coordinates. When voltage vectors having different signs are expressed on ⁇ coordinates, they have a relationship as shown in FIG.
  • control device 2 controls the inverter 3 so that the voltage correction unit 10 and the PWM generation unit 11 output any voltage vector except the zero vector as the first voltage vector. Then, after outputting the first voltage vector, the inverter 3 is controlled so as to output any voltage vector different from the first voltage vector, excluding the zero vector, as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and then the inverter 3 is output so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector. To control.
  • Such control is performed in the period A corresponding to the first period of the triangular wave carrier, and in the period B corresponding to the next period, the first voltage vector and the third voltage vector are different types of voltages from the previous period.
  • the inverter 3 is controlled so as to output each vector.
  • the sampling of the DC current signal IDC in the sample and hold circuit 12 is performed at a fixed timing every half cycle of the triangular wave carrier, and the result is used to reproduce the current.
  • the reproduction value Iuc, Ivc, Iwc of the three-phase phase current can be obtained in the unit 13.
  • each voltage vector output other than the zero vector changes according to the magnitude relationship of the original voltage command before correction. Accordingly, the voltage vectors output as the first, second, and third voltage vectors in the periods A and B are not limited to the above example.
  • FIG. 12 is a diagram illustrating details of correction of the voltage command by the voltage correction unit 10.
  • the two shunt current detectors 35 of the inverter 3 perform two periods in two consecutive periods of the triangular wave carrier, that is, the period A and the period B described above.
  • the voltage correction unit 10 corrects the voltage command for each phase so that the phase current can be detected.
  • two continuous cycles of the triangular wave carrier are divided into half cycles, and the respective periods are represented as Tc0, Tc1, Tc2, and Tc3.
  • the voltage correction unit 10 In the periods Tc0 and Tc1 corresponding to the first period (period A) of the two consecutive periods of the triangular wave carrier, the voltage correction unit 10 outputs the intermediate phase voltage command without changing the interval between the voltage commands of each phase. Always 0. On the other hand, in the periods Tc2 and Tc3 corresponding to the next cycle (period B), the voltage command of the minimum phase is always set to 0 without changing the interval between the voltage commands of each phase. In FIG. 12, the voltage commands for the respective phases thus determined are shown as the maximum phase voltage command Vmax, the intermediate phase voltage command Vmid and the minimum phase voltage command Vmin before shifting.
  • the voltage correction unit 10 calculates the voltage commands Vmax0, Vmid0, and Vmin0 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc0 by the following formulas (1) to (3). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax of the maximum phase before the shift and the voltage command Vmid of the intermediate phase. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the difference between the voltage command Vmin for the minimum phase before the shift and the voltage command Vmid for the intermediate phase and a predetermined required shift amount Vshift.
  • Equation (2) indicates that the shift amount with respect to the intermediate phase is zero.
  • Vshift in the equation (3) represents a necessary shift amount set in advance.
  • This necessary shift amount Vshift can be set based on the output time of the voltage vector necessary for current detection. For example, required before the basis ringing and size of shunt current I DC, and the like set value of the dead time of the switching elements, to detect the shunt current I DC flowing from the change of the voltage vector to the DC bus 34 of the inverter 3 Time is determined, and the required shift amount Vshift can be determined according to the time. Alternatively, by observing the waveform of the shunt current I DC, it may determine the required shift amount Vshift based on the observation result. From equation (3), it can be seen that the corrected minimum phase voltage command Vmin0 is smaller than the value obtained by subtracting the required shift amount Vshift from 0, as shown in FIG.
  • the voltage correction unit 10 calculates the voltage commands Vmax1, Vmid1, and Vmin1 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc1 by the following equations (4) to (6). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the necessary shift amount Vshift. The maximum phase voltage command Vmax1 and the minimum phase voltage command Vmin1 can be obtained by shifting the maximum phase and minimum phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the minimum phase represented by the equations (4) and (6) are indicated by arrows in the drawing. Equation (5) indicates that the shift amount with respect to the intermediate phase is zero.
  • the voltage correction unit 10 calculates the voltage commands Vmax2, Vmid2, and Vmin2 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc2 by the following formulas (7) to (9). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax for the maximum phase before the shift and the voltage command Vmin for the minimum phase. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the necessary shift amount Vshift. Then, the maximum phase voltage command Vmax2 and the intermediate phase voltage command Vmid2 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the intermediate phase represented by the equations (7) and (8) are indicated by arrows in the drawing. Equation (9) indicates that the shift amount with respect to the minimum phase is zero.
  • the voltage correction unit 10 calculates the voltage commands Vmax3, Vmid3, and Vmin3 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc3 by the following equations (10) to (12). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the difference between the voltage command Vmid for the intermediate phase before the shift and the voltage command Vmin for the minimum phase and the required shift amount Vshift.
  • the maximum phase voltage command Vmax3 and the intermediate phase voltage command Vmid3 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts.
  • the shift amounts of the maximum phase and the intermediate phase represented by the expressions (10) and (11) are indicated by arrows in the drawing.
  • Expression (12) indicates that the shift amount with respect to the minimum phase is zero.
  • the voltage correction unit 10 performs the correction as described above, whereby the sample and hold circuit 12 detects the currents of the minimum phase and the intermediate phase having different signs in each period of Tc0 to Tc3. It can be done once. Based on the result of this current detection, the current reproducer 13 can obtain the reproduced values Iuc, Ivc, and Iwc of the three-phase currents.
  • the voltage correction unit 10 corrects the voltage command of each phase so that the average value of the line voltage in these periods matches the original line voltage before correction. Yes. Therefore, there is no influence on the fundamental wave of the AC motor 4 due to the correction of the voltage command.
  • the control device 2 of the AC motor 4 is connected to an inverter 3 connected to the three-phase AC motor 4, and in the inverter 3, switching elements Sup, Svp, provided with three phases respectively on the positive electrode side and the negative electrode side,
  • the drive of the AC motor 4 is controlled according to the switching states of Swp, Sun, Svn and Swn.
  • the inverter 3 can output eight voltage vectors to the AC motor 4 in accordance with the switching state of each switching element. This voltage vector includes the switching state in which all the switching elements Sup, Svp, Swp on the positive side are on and all the switching elements Sun, Svn, Swn on the negative side are off, and all the switching elements Sup on the positive side.
  • the control device 2 of the AC motor 4 controls the inverter 3 so as to output any voltage vector except the zero vector as the first voltage vector, and after outputting the first voltage vector, the zero vector is removed.
  • the inverter 3 is controlled so as to output any voltage vector different from the first voltage vector as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and after outputting the zero vector, a voltage vector obtained by inverting the first voltage vector is used as the third voltage vector.
  • the inverter 3 is controlled to output. Thereby, the drive of the AC motor 4 is controlled. Since it did in this way, the control apparatus of the alternating current motor which can detect a phase current with high precision using an inexpensive microcomputer is realizable.
  • the control device 2 controls the inverter 3 so as to repeatedly output the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector in order. At this time, the inverter 3 is controlled so that different types of voltage vectors from the previous time are output as the first voltage vector and the third voltage vector, respectively. Since it did in this way, the phase current of two phases which are different from each other can be detected, and the phase current of each phase can be obtained based on this.
  • the control device 2 uses the current estimation unit 13 based on the detection result of the DC shunt current I DC flowing in the DC bus 34 of the inverter 3 during the output period of the first voltage vector and the third voltage vector. Then, the phase current of the AC motor 4 is obtained. Since it did in this way, in the electric current estimation part 13, the phase current of each phase can be calculated
  • the control device 2 includes a voltage command generator 20 that generates three-phase voltage commands Vu *, Vv *, and Vw *, a voltage corrector 10 that corrects these voltage commands, and a PWM generator 11. Prepare.
  • the PWM generator 11 compares the corrected voltage commands Vu **, Vv **, and Vw ** corrected by the voltage correction unit 10 with a triangular wave carrier having a predetermined period, and based on the comparison result, the three-phase PWM waveform is generated.
  • the control device 2 controls the switching state of the switching elements Sup, Svp, Swp, Sun, Svn and Swn, and the AC motor 4 Control the drive. Since it did in this way, according to the voltage command after correction
  • the voltage correction unit 10 sets the intermediate voltage command Vmid to 0 in the three-phase voltage command and the maximum voltage command Vmax in the three-phase voltage command in the first cycle out of the two consecutive cycles of the triangular wave carrier. And the voltage command of each phase is corrected by shifting the minimum voltage command Vmin. In the next cycle following the first cycle, the minimum voltage command Vmin is set to 0, and the maximum voltage command Vmax and the intermediate voltage command Vmid are shifted to correct the voltage command of each phase. Since it did in this way, while said 1st, 2nd and 3rd voltage vector is output sequentially, these voltage vectors can be changed for every period of a triangular wave carrier.
  • the voltage correction unit 10 determines the maximum value based on the difference between the maximum voltage command Vmax and the intermediate voltage command Vmid according to the above-described equations (1) and (3).
  • the shift amount with respect to the minimum voltage command Vmin is determined based on the difference between the minimum voltage command Vmin and the intermediate voltage command Vmid and the predetermined required shift amount Vshift.
  • the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (4) and (6), and the necessary shift amount.
  • a shift amount with respect to the minimum voltage command Vmin is determined based on Vshift.
  • the shift amount with respect to the maximum voltage command Vmax is determined based on the difference between the maximum voltage command Vmax and the minimum voltage command Vmin according to the above-described equations (7) and (8).
  • the shift amount for the intermediate voltage command Vmid is determined based on the necessary shift amount Vshift.
  • the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (10) and (11), and the intermediate voltage command Based on the difference between Vmid and the minimum voltage command Vmin and the required shift amount Vshift, the shift amount for the intermediate voltage command Vmid is determined. Since it did in this way, correction
  • the control device 2 obtains the detection result of the DC shunt current I DC to when the triangular wave carrier takes an intermediate value between the minimum and maximum values flowing through the DC bus 34 of the inverter 3. Since this is done, it is possible to obtain the shunt current I DC corresponding to the phase currents of two different phases for each fixed timing.
  • FIG. 13 is a diagram showing a configuration of a control device 2B according to the second embodiment of the present invention.
  • the inverter 3 by switching the switching state of only the two-phase switching elements among the three-phase switching elements in the low-speed range where the AC motor is driven at a low speed,
  • the inverter 3 is controlled to perform 120-degree energization driving.
  • the inverter 3 is controlled to perform the three-phase energization drive by switching the switching state of each of the three-phase switching elements by the sensorless drive control as described in the first embodiment. To do. Thereby, it is set as the structure which compensates both faults. Since the sine wave sensorless driving method of the present invention is a method suitable for an inexpensive microcomputer as described above, it is very suitable for a combination with 120-degree energization driving that can be realized originally by an inexpensive microcomputer.
  • 120-degree energization drive capable of sensorless drive control in the low speed region is realized in the configuration of the control device 2 according to the first embodiment shown in FIG. The structure for this is added, and these can be switched now.
  • an AC motor drive system according to the second embodiment of the present invention can be realized.
  • the control device 2B shown in FIG. 13 includes a 120-degree energization unit 17, a 120-degree PWM unit 18 that performs PWM for 120-degree energization drive, a 120-degree energization drive, and a three-phase It comprises changeover switches 19a to 19e for switching energization drive.
  • a d-axis current controller 7B and a position / speed estimation unit 15B are provided.
  • the 120-degree energization unit 17 outputs a phase value ⁇ dcB for performing 120-degree energization driving and a speed estimation value ⁇ 1cB obtained by calculation from the result of performing 120-degree energization driving.
  • the phase value ⁇ dcB is a discrete value every 60 degrees.
  • the phase value ⁇ dcB and the phase value ⁇ dcS which is a continuous value output from the position / velocity estimation unit 15B are switched by the changeover switch 19c and output to the dq inverse converter 9.
  • Each of the change-over switches 19a to 19e is switched to the “0” side during 120-degree energization driving, and is switched to the “1” side during three-phase energization driving.
  • the changeover switch 19e switches between the estimated speed value ⁇ 1cB output from the 120-degree energization unit 17 and the estimated speed value ⁇ 1cS output from the position / speed estimation unit 15B, and either one is used as the electrical angular velocity ⁇ 1c of the AC motor 4. Output to the speed converter 16.
  • the changeover switch 19a switches between a PWM waveform for three-phase energization driving output from the PWM generator 11 and a PWM waveform for 120-degree energization driving output from the 120-degree PWM unit 18, and either one is switched to the inverter 3 Output to.
  • Changeover switch 19d is switched as a feedback current to the q-axis current controller 8, a q-axis current Iq which are coordinate transformation by dq converter 14, the sample-and-hold circuit 12 and a detection result of shunt current I DC sampled .
  • the current control may be one, or may be directly fed back to the detection value of the shunt current I DC.
  • the changeover switch 19b switches between 0 and the value output from the d-axis current controller 7B, and outputs either one to the dq inverse converter 9 as the d-axis voltage command Vd *.
  • the changeover switch 19b switches so that the d-axis voltage command Vd * becomes zero.
  • the q-axis voltage command Vq * output from the q-axis current controller 8 is commonly used during 120-degree energization driving and three-phase energization driving.
  • the d-axis current controller 7B calculates the d-axis voltage command Vd * used in the three-phase energization drive directly from the q-axis current command Iq * without performing feedback control of the d-axis current Id.
  • Vd * used in the three-phase energization drive directly from the q-axis current command Iq * without performing feedback control of the d-axis current Id.
  • processing of values in the d-axis current controller 7B becomes a problem.
  • the d-axis voltage command Vd * is obtained by PI control (proportional integral control)
  • PI control proportional integral control
  • the d-axis current controller 7B determines the d-axis voltage command Vd * by feedforward calculation based on the q-axis current command Iq *, and does not perform feedback control. As a result, switching between the 120-degree energization drive and the three-phase energization drive can be performed smoothly.
  • the d-axis voltage command Vd * may be determined by a feedforward calculation based on the q-axis current Iq estimated by the current estimation unit 21.
  • the d-axis current controller 7B includes a first-order lag filter 71, a multiplier 72, a q-axis inductance setting unit 73, and a sign inverter 74. Since the arithmetic processing realized by these configurations is a feedforward calculation for the d-axis voltage of the AC motor 4, three-phase energization driving can be performed stably. Note that the time constant Tr of the first-order lag filter 71 may be set according to the response set value of the q-axis current controller 8, for example.
  • the position / velocity estimation unit 15B includes an axis error estimator 151, a zero setter 152, a PI controller 153, a subtractor 6c, and a phase calculator 154.
  • Such a configuration of the position / speed estimation unit 15B is well known as a medium / high speed sensorless position / speed estimator.
  • the axis error estimator 151 estimates and calculates the deviation between the actual rotor position of the AC motor 4 that is a permanent magnet motor and the rotor position calculated by the control device 2B, and outputs the result as an axis error ⁇ dc. .
  • the subtractor 6 c calculates a difference between “0” output from the zero setting unit 152 and the axis error ⁇ dc from the axis error estimator 151, and outputs the calculation result to the PI controller 153.
  • the PI controller 153 performs control for setting the value of the axis error ⁇ dc to 0 by calculating the speed estimated value ⁇ 1cS so that the difference from the subtractor 6c becomes 0.
  • This PI controller 153 constitutes a PLL (Phase Locked Loop).
  • the estimated speed value ⁇ 1cS obtained by the PI controller 153 corresponds to the electrical angular velocity of the AC motor 4.
  • the phase calculator 154 integrates the estimated speed value ⁇ 1cS from the PI controller 153 to obtain the phase value ⁇ dcS.
  • the phase value ⁇ dcS is output from the position / velocity estimation unit 15B to the changeover switch 19c.
  • the control device 2B switches the switching state of only the two-phase switching elements among the switching elements of the inverter 3 so as to perform 120-degree energization driving. 3 is controlled. Further, in the speed range other than the low speed range, the inverter 3 is controlled so that the switching state of the three-phase switching element is switched to perform the three-phase energization drive. Since it did in this way, a high-performance sensorless drive is realizable in all the speed ranges from a low speed range to a high speed range.
  • the control device 2B includes a d-axis current controller 7B that outputs a d-axis voltage command Vd * used in three-phase current drive, and a q-axis voltage command commonly used in 120-degree current drive and three-phase current drive.
  • a q-axis current controller 8 for outputting Vq *.
  • the d-axis current controller 7B outputs a d-axis voltage command Vd * by a feedforward calculation based on a q-axis current command Iq * input from the outside or a q-axis current Iq that is a detection result of a current flowing in the AC motor 4. To do. Since it did in this way, switching with 120 degree energization drive and three phase energization drive can be performed smoothly.
  • the d-axis voltage command Vd * is obtained by feedback calculation using the d-axis current controller 7 as described in the first embodiment without using the d-axis current controller 7B. It can also be output.
  • the configuration as shown in FIG. 13 is more desirable from the viewpoint of realizing processing by a low-function microcomputer, which is a feature of the present invention.
  • FIG. 14 is a diagram showing a configuration of an AC motor drive system 40 according to the third embodiment of the present invention.
  • the AC motor drive system 40 shown in FIG. 14 is configured by integrating the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 described in the first embodiment.
  • the q-axis current command generator 1, the control device 2, and the inverter 3 are packaged inside the AC motor 4 as a drive control board 41 mounted on one circuit board.
  • the wiring to the integrated AC motor drive system 40 can be, for example, only the power line and the communication line shown in FIG.
  • the power supply line is connected to the inverter 3 and is used to supply a DC power supply or the like to the switching element.
  • the communication line is connected to the q-axis current command generator 1 and the control device 2 to transmit a rotational speed command input to the q-axis current command generator 1 and to transmit an operation state output from the control device 2. Used.
  • the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 are all integrated as the AC motor drive system 40, a small size is achieved. Can be realized. In addition, since the wiring between the respective components can be eliminated, the handling property can be improved.
  • the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2.
  • FIG. 15 is a diagram showing a configuration of a hydraulic control system according to the fourth embodiment of the present invention.
  • the hydraulic control system shown in FIG. 15 is used for, for example, hydraulic control of a transmission inside a vehicle, hydraulic control of a brake, and the like.
  • This hydraulic control system includes a hydraulic control unit 25 and a hydraulic circuit 50.
  • the hydraulic control unit 25 includes the q-axis current command generator 1, the control device 2, the inverter 3 and the AC motor 4 described in the first embodiment, and an oil pump 26 attached to the AC motor 4.
  • the oil pump 26 operates according to the driving of the AC motor 4 and controls the hydraulic pressure of the hydraulic circuit 50.
  • the hydraulic circuit 50 includes a tank 51 that stores hydraulic oil, a relief valve 52 that keeps the hydraulic pressure below a set value, a solenoid valve 53 that switches a discharge destination of the hydraulic oil, and a cylinder 54 that operates as a hydraulic actuator.
  • the oil pump 26 operates according to the driving of the AC motor 4 to generate hydraulic pressure, and drives the cylinder 54 that is a hydraulic actuator.
  • the load of the oil pump 26 changes, and a load disturbance to the hydraulic control unit 25 occurs.
  • the AC motor 4 may stop due to a load that is several times greater than the steady-state pressure.
  • the hydraulic drive system according to the present embodiment performs the drive control of the AC motor 4 by the method described in the first embodiment, so that the rotor in the stopped state is obtained. The position can be estimated. Therefore, no problem occurs.
  • the relief valve 52 may be eliminated as described below.
  • FIG. 16 is a diagram showing a modification of the configuration of the hydraulic control system according to the fourth embodiment of the present invention.
  • This hydraulic control system is not provided with the relief valve 52 which is present in FIG. That is, in the hydraulic control system according to the present embodiment, the control performance in the entire speed range for the AC motor 4 can be improved, so that the hydraulic pressure can be controlled without a relief valve which is a mechanical protection device for avoiding an excessive load. It becomes.
  • the present invention can be applied in hydraulic control for various uses.
  • the fourth embodiment the application example to the hydraulic control for controlling the pressure of the hydraulic oil has been described.
  • the fourth embodiment may be applied to the pressure control of other fluids such as water. That is, the present invention can be applied to various fluid pressure control systems that control fluid pressure by operating a pump by driving an AC motor.
  • the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.
  • FIG. 17 is a diagram showing a configuration of a positioning system according to the fifth embodiment of the present invention.
  • the positioning system shown in FIG. 17 includes the control device 2, the inverter 3, the AC motor 4, the q-axis current command generator 1C, and the positioning device 70 described in the first embodiment.
  • the q-axis current command generator 1C includes a position controller 101, a speed controller 102, a subtracter 6d, and a subtractor 6e.
  • the subtractor 6 d calculates the difference between the position command ⁇ * input from the outside and the phase angle ⁇ dc obtained by the control device 2, and outputs the calculation result to the position controller 101.
  • the position controller 101 obtains the speed command ⁇ r * based on the calculation result from the subtractor 6d.
  • the subtractor 6 e calculates the difference between the speed command ⁇ r * from the position controller 101 and the rotational speed ⁇ r of the AC motor 4 obtained by the control device 2, and outputs the calculation result to the speed controller 102.
  • the speed controller 102 obtains a q-axis current command Iq * based on the calculation result from the subtractor 6 e and outputs it to the control device 2.
  • the positioning device 70 is connected to the rotating shaft of the AC motor 4 as a load of the AC motor 4.
  • This positioning device 70 is a device using, for example, a ball screw and can adjust the position of the object by operating in accordance with the driving of the AC motor 4.
  • no position sensor is attached to the positioning device 70, and the position controller 101 performs drive control of the AC motor 4 using the phase angle ⁇ dc output from the control device 2 as it is.
  • the operation of the positioning device 70 can be controlled, and the position of the object can be adjusted according to the position command ⁇ *. Therefore, the position of the object can be controlled without using a position sensor.
  • the present invention can be applied in position control of various objects.
  • the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

An inverter can output eight voltage vectors including two kinds of zero vectors to an AC electric motor in accordance with the switching states of switching elements. An AC electric motor control device controls the driving of the AC electric motor by: controlling the inverter to output any of the voltage vectors excluding the zero vectors as a first voltage vector; after the first voltage vector is output, controlling the inverter to output any of the voltage vectors excluding the zero vectors and being different from the first voltage vector as a second voltage vector; after the second voltage vector is output, controlling the inverter to output the zero vector; and after the zero vector is output, controlling the inverter to output the voltage vector obtained by inverting the first voltage vector as a third voltage vector.

Description

交流電動機の制御装置、交流電動機駆動システム、流体圧制御システム、位置決めシステムAC motor control device, AC motor drive system, fluid pressure control system, positioning system
 本発明は、交流電動機の制御装置と、これを利用した交流電動機駆動システム、流体圧制御システムおよび位置決めシステムとに関する。 The present invention relates to an AC motor control device, and an AC motor drive system, fluid pressure control system, and positioning system using the same.
 産業、家電、自動車等の様々な分野において、三相交流電動機が幅広く用いられている。特に、小型・高効率の永久磁石型同期電動機(Permanent Magnet Synchronous Motor)(以下、PMSMと略す)の適用が増加傾向にある。 Three-phase AC motors are widely used in various fields such as industry, home appliances, and automobiles. In particular, the application of small-sized, high-efficiency permanent magnet synchronous motors (hereinafter abbreviated as PMSM) is increasing.
 従来、交流電動機の回転速度やトルクを高精度に制御するためには、電動機の回転状態を検出する回転速度センサ(あるいは、PMSMの場合は回転位置センサ)と、電動機の各相に流れる相電流を検出する相電流センサとが必要であった。しかし、これらのセンサ類を用いると、故障率の上昇、検出結果のばらつき、取り付け精度、取り付けスペース等の問題が生じる。そのため、センサを用いずに電動機の回転状態や相電流を検出できるようにするセンサレス化が望ましい。 Conventionally, in order to control the rotational speed and torque of an AC motor with high accuracy, a rotational speed sensor (or rotational position sensor in the case of PMSM) that detects the rotational state of the motor, and a phase current that flows through each phase of the motor And a phase current sensor for detecting. However, when these sensors are used, problems such as an increase in failure rate, variation in detection results, installation accuracy, and installation space occur. Therefore, it is desirable to make the sensorless so that the rotation state and phase current of the motor can be detected without using a sensor.
 相電流のセンサレス化を実現するための方法は、たとえば下記の特許文献1、2に記載されている。 A method for realizing sensorless phase current is described in, for example, Patent Documents 1 and 2 below.
 特許文献1の相電流検出方法では、電動機の駆動を制御するインバータの直流側に流れる電流を検出し、その検出結果に基づいてスイッチング前後の電流変化分を求めることで、各相の相電流を求めている。 In the phase current detection method of Patent Document 1, the current flowing in the DC side of the inverter that controls the driving of the electric motor is detected, and the current change before and after switching is obtained based on the detection result. Seeking.
 特許文献2の相電流検出方法では、1周期内で3相のうち2相のスイッチング素子をスイッチングする2相変調方式のパルス幅変調(PWM)において、三角波キャリア信号の連続する2つの山または谷の部分で、互いに180度位相が異なる基本電圧ベクトル成分を交互に出力する。そして、インバータの直流母線上に設けられたシャント抵抗に流れる電流をこのタイミングで検出し、アナログデジタル変換してマイコンに取り込む。これにより、電流検出のタイミングを固定化して、安価なマイコンを利用可能としている。 In the phase current detection method of Patent Document 2, two continuous peaks or valleys of a triangular wave carrier signal are obtained in two-phase modulation type pulse width modulation (PWM) in which two-phase switching elements are switched among three phases within one cycle. In this part, basic voltage vector components whose phases are different from each other by 180 degrees are alternately output. Then, the current flowing through the shunt resistor provided on the DC bus of the inverter is detected at this timing, converted from analog to digital, and taken into the microcomputer. Thereby, the timing of current detection is fixed, and an inexpensive microcomputer can be used.
日本国特開平8-19263号公報Japanese Laid-Open Patent Publication No. 8-19263 日本国特開2005-12934号公報Japanese Unexamined Patent Publication No. 2005-12934
 電動機の駆動中、シャント抵抗には、各相のスイッチ状態に応じて、U,V,W相のうちいずれか一相の相電流に応じた電流が流れる。したがって、各相のスイッチ状態の切り替えに同期したタイミングでシャント抵抗の電流を検出することで、上記のような相電流のセンサレス化を実現することができる。 During driving of the motor, a current corresponding to the phase current of one of the U, V, and W phases flows through the shunt resistor according to the switch state of each phase. Therefore, by detecting the current of the shunt resistor at the timing synchronized with the switching of the switch state of each phase, it is possible to realize the sensorless phase current as described above.
 しかし、一般的なPWMの場合、キャリア信号の半周期ごとにスイッチ状態の切り替えが発生し、その発生タイミングは一定ではない。したがって、特許文献1のような相電流検出方法では、キャリア信号の半周期内に、電流検出および検出値のアナログデジタル変換を2回ずつ行う必要がある。また、スイッチ状態の切り替えに応じて、電流の検出タイミングを変化させる必要もある。こうした機能を実現可能なマイコンは高価であるため、装置のコストアップにつながるという課題がある。また、検出される電流は、スイッチ状態の切り替えに伴って発生する電流リプル等の誤差を含むため、トルク制御の精度が劣化してしまうという課題もある。 However, in the case of general PWM, switching of the switch state occurs every half cycle of the carrier signal, and the generation timing is not constant. Therefore, in the phase current detection method as in Patent Document 1, it is necessary to perform current detection and analog-digital conversion of the detected value twice in a half cycle of the carrier signal. It is also necessary to change the current detection timing in accordance with the switching of the switch state. Since a microcomputer capable of realizing such a function is expensive, there is a problem that the cost of the apparatus is increased. Further, since the detected current includes an error such as a current ripple generated when the switch state is switched, there is a problem that the accuracy of torque control is deteriorated.
 一方、特許文献2のような相電流検出方法では、三角波キャリア信号の山または谷のタイミングでシャント抵抗に流れる電流を検出できるため、上記のようなコストアップに関する課題は生じない。しかし、先のPWM期間(第1の期間)では、本来の電圧指令ベクトルよりも大きくなるように電圧指令ベクトルを補正して出力し、それに続くPWM期間(第2の期間)では、第1の期間における電圧指令ベクトルと180度位相が異なる電圧指令ベクトルを出力することにより、本来の電圧指令ベクトルに合わせている。そのため、第1の期間における電流リプルが増大し、その結果、電流検出値は大きな検出誤差を含んだものとなる。したがって、トルク制御の精度が劣化してしまうという課題を克服することはできない。 On the other hand, in the phase current detection method as in Patent Document 2, the current flowing through the shunt resistor can be detected at the timing of the peak or valley of the triangular wave carrier signal. However, in the previous PWM period (first period), the voltage command vector is corrected and output so as to be larger than the original voltage command vector, and in the subsequent PWM period (second period), the first By outputting a voltage command vector that is 180 degrees out of phase with the voltage command vector in the period, the voltage command vector is matched with the original voltage command vector. Therefore, the current ripple in the first period increases, and as a result, the current detection value includes a large detection error. Therefore, the problem that the accuracy of torque control deteriorates cannot be overcome.
 本発明の目的は、以上説明したような従来の相電流検出方法における課題に鑑みて、安価なマイコンを用いて高精度に相電流を検出可能な交流電動機の制御装置を提供することにある。 An object of the present invention is to provide an AC motor control device capable of detecting a phase current with high accuracy using an inexpensive microcomputer in view of the problems in the conventional phase current detection method as described above.
 本発明の第1の態様による交流電動機の制御装置は、三相の交流電動機と接続されたインバータに接続され、インバータにおいて正極側と負極側にそれぞれ三相ずつ設けられたスイッチング素子のスイッチング状態に応じて交流電動機の駆動を制御するものである。インバータは、スイッチング素子のスイッチング状態に応じて、8通りの電圧ベクトルを交流電動機に出力可能であり、電圧ベクトルは、正極側の全てのスイッチング素子がオンであり、負極側の全てのスイッチング素子がオフであるスイッチング状態と、正極側の全てのスイッチング素子がオフであり、負極側の全てのスイッチング素子がオンであるスイッチング状態とにそれぞれ対応する2種類の零ベクトルを含む。この交流電動機の制御装置は、零ベクトルを除くいずれかの電圧ベクトルを第1の電圧ベクトルとして出力するようにインバータを制御し、第1の電圧ベクトルを出力した後、零ベクトルを除いた第1の電圧ベクトルとは異なるいずれかの電圧ベクトルを、第2の電圧ベクトルとして出力するようにインバータを制御し、第2の電圧ベクトルを出力した後、零ベクトルを出力するようにインバータを制御し、零ベクトルを出力した後、第1の電圧ベクトルを反転した電圧ベクトルを第3の電圧ベクトルとして出力するようにインバータを制御することにより、交流電動機の駆動を制御する。
 本発明の第2の態様によると、第1の態様の交流電動機の制御装置は、第1の電圧ベクトル、第2の電圧ベクトル、零ベクトルおよび第3の電圧ベクトルを繰り返して順次出力するようにインバータを制御し、第1の電圧ベクトルおよび第3の電圧ベクトルとして、前回とは異なる種類の電圧ベクトルをそれぞれ出力するようにインバータを制御することが好ましい。
 本発明の第3の態様によると、第1または第2の態様の交流電動機の制御装置は、第1の電圧ベクトルおよび第3の電圧ベクトルの出力期間中に、インバータの直流母線に流れる直流電流の検出結果に基づいて、交流電動機の相電流を求める電流推定部を備えることが好ましい。
 本発明の第4の態様によると、第1乃至第3のいずれか一態様の交流電動機の制御装置は、三相の電圧指令を発生する電圧指令発生部と、電圧指令を補正する電圧補正部と、電圧補正部により補正された補正後の電圧指令と、所定周期の三角波キャリアとを比較し、その比較結果に基づいて三相のPWM波形を発生するPWM発生部と、を備え、PWM発生部により発生されたPWM波形をインバータに出力することで、スイッチング素子のスイッチング状態を制御して、交流電動機の駆動を制御することが好ましい。
 本発明の第5の態様によると、第4の態様の交流電動機の制御装置において、電圧補正部は、三角波キャリアの連続する2周期のうち最初の周期では、三相の電圧指令で中間の電圧指令を0とし、さらに三相の電圧指令で最大の電圧指令および最小の電圧指令をシフトさせることにより、電圧指令を補正し、最初の周期に続く次の周期では、最小の電圧指令を0とし、さらに最大の電圧指令および中間の電圧指令をシフトさせることにより、電圧指令を補正することが好ましい。
 本発明の第6の態様によると、第5の態様の交流電動機の制御装置において、電圧補正部は、最初の周期の前半期間では、最大の電圧指令と中間の電圧指令の差分に基づいて最大の電圧指令に対するシフト量を決定すると共に、最小の電圧指令と中間の電圧指令の差分および所定の必要シフト量に基づいて最小の電圧指令に対するシフト量を決定し、最初の周期の後半期間では、補正後の電圧指令が0となるように最大の電圧指令に対するシフト量を決定すると共に、必要シフト量に基づいて最小の電圧指令に対するシフト量を決定し、次の周期の前半期間では、最大の電圧指令と最小の電圧指令の差分に基づいて最大の電圧指令に対するシフト量を決定すると共に、必要シフト量に基づいて中間の電圧指令に対するシフト量を決定し、次の周期の後半期間では、補正後の電圧指令が0となるように最大の電圧指令に対するシフト量を決定すると共に、中間の電圧指令と最小の電圧指令の差分および必要シフト量に基づいて中間の電圧指令に対するシフト量を決定することが好ましい。
 本発明の第7の態様によると、第6の態様の交流電動機の制御装置において、必要シフト量は、スイッチング素子のスイッチング状態が変化してからインバータの直流母線に流れる直流電流を検出するまでの時間に応じて設定されることが好ましい。
 本発明の第8の態様によると、第4乃至第7のいずれか一態様の交流電動機の制御装置は、三角波キャリアが最小値と最大値の中間値をとる時点でインバータの直流母線に流れる直流電流の検出結果を取得することが好ましい。
 本発明の第9の態様によると、第1乃至第8のいずれか一態様の交流電動機の制御装置は、交流電動機が低速駆動している低速域では、スイッチング素子のうち、二相のスイッチング素子のみスイッチング状態を切り替えて120度通電駆動を行うように、インバータを制御し、低速域以外の速度域では、三相のスイッチング素子のスイッチング状態を切り替えて三相通電駆動を行うように、インバータを制御することが好ましい。
 本発明の第10の態様によると、第9の態様の交流電動機の制御装置は、三相通電駆動において用いられる第1の電圧指令を出力する第1の電流制御器と、120度通電駆動および三相通電駆動において共通に用いられる第2の電圧指令を出力する第2の電流制御器と、を備え、第1の電流制御器は、外部から入力される電流指令または交流電動機に流れる電流の検出結果に基づくフィードフォワード演算により、第1の電圧指令を出力することが好ましい。
 本発明の第11の態様による交流電動機駆動システムは、第1乃至第10のいずれか一態様の交流電動機の制御装置と、交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、交流電動機の制御装置により、スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、を備え、交流電動機の制御装置、インバータおよび交流電動機は、一体化されている。
 本発明の第12の態様による流体圧制御システムは、第1乃至第10のいずれか一態様の交流電動機の制御装置と、交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、交流電動機の制御装置により、スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、交流電動機の駆動に応じて動作し、流体圧を制御するポンプとを備える。
 本発明の第13の態様による位置決めシステムは、第1乃至第10のいずれか一態様の交流電動機の制御装置と、交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、交流電動機の制御装置により、スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、交流電動機の駆動に応じて動作し、対象物の位置を調整する位置決め装置とを備える。
The control device for an AC motor according to the first aspect of the present invention is connected to an inverter connected to a three-phase AC motor, and the inverter is switched to a switching state in which three phases are provided on each of the positive electrode side and the negative electrode side. Accordingly, the driving of the AC motor is controlled. The inverter can output eight types of voltage vectors to the AC motor according to the switching state of the switching elements. The voltage vector is such that all switching elements on the positive electrode side are on and all switching elements on the negative electrode side are on. Two types of zero vectors corresponding to the switching state in which the switching state is OFF and the switching state in which all switching elements on the positive electrode side are OFF and all switching elements on the negative electrode side are ON are included. The AC motor control apparatus controls the inverter to output any voltage vector excluding the zero vector as the first voltage vector, outputs the first voltage vector, and then removes the first vector from the first vector. Controlling the inverter to output any one of the voltage vectors different from the voltage vector as a second voltage vector, controlling the inverter to output a zero vector after outputting the second voltage vector, After outputting the zero vector, the inverter is controlled so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector, thereby controlling the driving of the AC motor.
According to the second aspect of the present invention, the AC motor control device of the first aspect repeatedly outputs the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector sequentially. It is preferable to control the inverter so as to output different types of voltage vectors from the previous time as the first voltage vector and the third voltage vector.
According to the third aspect of the present invention, the control apparatus for an AC motor according to the first or second aspect provides a direct current that flows through the direct current bus of the inverter during the output period of the first voltage vector and the third voltage vector. It is preferable to provide a current estimation unit that obtains the phase current of the AC motor based on the detection result.
According to the fourth aspect of the present invention, the AC motor control device according to any one of the first to third aspects includes a voltage command generation unit that generates a three-phase voltage command, and a voltage correction unit that corrects the voltage command. A PWM generation unit that compares the corrected voltage command corrected by the voltage correction unit with a triangular wave carrier having a predetermined period and generates a three-phase PWM waveform based on the comparison result, and generates a PWM. It is preferable to control the driving of the AC motor by controlling the switching state of the switching element by outputting the PWM waveform generated by the unit to the inverter.
According to the fifth aspect of the present invention, in the control apparatus for an AC electric motor according to the fourth aspect, the voltage correction unit is an intermediate voltage with a three-phase voltage command in the first period of two consecutive periods of the triangular wave carrier. The command is set to 0, and the voltage command is corrected by shifting the maximum voltage command and the minimum voltage command with the three-phase voltage command. In the next cycle following the first cycle, the minimum voltage command is set to 0. Further, it is preferable to correct the voltage command by shifting the maximum voltage command and the intermediate voltage command.
According to the sixth aspect of the present invention, in the control apparatus for an AC motor according to the fifth aspect, the voltage correction unit has a maximum value based on a difference between the maximum voltage command and the intermediate voltage command in the first half of the first cycle. And determining the shift amount for the minimum voltage command based on the difference between the minimum voltage command and the intermediate voltage command and a predetermined required shift amount, and in the latter half of the first cycle, The shift amount for the maximum voltage command is determined so that the corrected voltage command becomes 0, and the shift amount for the minimum voltage command is determined based on the necessary shift amount. In the first half of the next cycle, the maximum shift amount is determined. The shift amount for the maximum voltage command is determined based on the difference between the voltage command and the minimum voltage command, and the shift amount for the intermediate voltage command is determined based on the required shift amount. In the latter half of the period, the shift amount with respect to the maximum voltage command is determined so that the corrected voltage command becomes 0, and the intermediate voltage is determined based on the difference between the intermediate voltage command and the minimum voltage command and the required shift amount. It is preferable to determine the shift amount with respect to the command.
According to the seventh aspect of the present invention, in the control apparatus for an AC electric motor according to the sixth aspect, the required shift amount is from when the switching state of the switching element changes until the DC current flowing through the DC bus of the inverter is detected. It is preferable to set according to time.
According to the eighth aspect of the present invention, in the control apparatus for an AC motor according to any one of the fourth to seventh aspects, the direct current that flows through the DC bus of the inverter when the triangular wave carrier takes an intermediate value between the minimum value and the maximum value. It is preferable to obtain the detection result of the current.
According to the ninth aspect of the present invention, the AC motor control device according to any one of the first to eighth aspects is a two-phase switching element among the switching elements in a low speed range where the AC motor is driven at a low speed. The inverter is controlled so as to perform 120-degree energization drive only by switching the switching state, and in the speed range other than the low speed range, the inverter is configured to perform the three-phase energization drive by switching the switching state of the three-phase switching element. It is preferable to control.
According to the tenth aspect of the present invention, the AC motor control device according to the ninth aspect includes a first current controller that outputs a first voltage command used in three-phase energization drive, a 120-degree energization drive, A second current controller that outputs a second voltage command commonly used in three-phase energization drive, and the first current controller is configured to receive a current command input from the outside or a current flowing through the AC motor. It is preferable to output the first voltage command by feed forward calculation based on the detection result.
An AC motor drive system according to an eleventh aspect of the present invention is connected to the AC motor control device according to any one of the first to tenth modes and the AC motor control device, and has three phases on the positive electrode side and the negative electrode side, respectively. Each having an inverter provided with a switching element and a three-phase AC motor that is driven and controlled according to the switching state of the switching element by an AC motor control device, and the AC motor control device, inverter, and AC motor are: Is integrated.
A fluid pressure control system according to a twelfth aspect of the present invention is connected to the control apparatus for an AC motor according to any one of the first to tenth aspects and the control apparatus for the AC motor, and each has three phases on the positive electrode side and the negative electrode side. Three-phase AC motors that are driven and controlled according to the switching state of the switching elements, and inverters that are provided with switching elements and AC motor control devices, and fluid pressure that operates according to the AC motor drive And a pump.
A positioning system according to a thirteenth aspect of the present invention is connected to the AC motor control apparatus according to any one of the first to tenth aspects and the AC motor control apparatus, and switches three phases each on the positive electrode side and the negative electrode side. A three-phase AC motor that is driven and controlled according to the switching state of the switching element and an AC motor driven by the inverter provided with the element and the AC motor control device, and adjusts the position of the object A positioning device.
 本発明によれば、安価なマイコンを用いて高精度に相電流を検出可能な交流電動機の制御装置を実現することができる。 According to the present invention, it is possible to realize an AC motor control device capable of detecting a phase current with high accuracy using an inexpensive microcomputer.
本発明の第1の実施形態に係る交流電動機の制御装置を含む交流電動機駆動システムの構成を示す図である。It is a figure which shows the structure of the alternating current motor drive system containing the control apparatus of the alternating current motor which concerns on the 1st Embodiment of this invention. 電圧指令の補正を行わない場合のPWM波形およびスイッチ状態の例を示す図である。It is a figure which shows the example of the PWM waveform and switch state when not correcting a voltage command. 各スイッチ状態においてスイッチング素子にそれぞれ流れる電流の様子を示す図である。It is a figure which shows the mode of the electric current which each flows into a switching element in each switch state. 交流電動機に流れる各相のモータ電流と、直流母線に流れるシャント電流およびスイッチ状態との関係を示す図である。It is a figure which shows the relationship between the motor current of each phase which flows into an alternating current motor, the shunt current which flows into a direct current bus, and a switch state. 従来の二相変調方式をベースとして電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図である。It is a figure which shows the example of a PWM waveform at the time of correcting a voltage command based on the conventional two-phase modulation system, and a switch state. 図5とは別の方法で電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図である。FIG. 6 is a diagram illustrating an example of a PWM waveform and a switch state when a voltage command is corrected by a method different from that in FIG. 5. 本発明による電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図である。It is a figure which shows the example of the PWM waveform at the time of correcting the voltage command by this invention, and a switch state. 期間Aにおける電圧ベクトルと検出される相電流の関係を示した一覧表である。6 is a table showing a relationship between a voltage vector in a period A and a detected phase current. 期間Bにおける電圧ベクトルと検出される相電流の関係を示した一覧表である。6 is a table showing a relationship between a voltage vector in a period B and a detected phase current. 第1の電圧ベクトルおよび第3の電圧ベクトルにおいて、シャント電流として検出される相電流の様子を示す図である。It is a figure which shows the mode of the phase current detected as a shunt electric current in the 1st voltage vector and the 3rd voltage vector. インバータから出力される各電圧ベクトルをαβ座標上に表した図である。It is the figure which represented each voltage vector output from an inverter on the (alpha) (beta) coordinate. 電圧補正部による電圧指令の補正の詳細を示す図である。It is a figure which shows the detail of correction | amendment of the voltage command by a voltage correction part. 本発明の第2の実施形態に係る制御装置の構成を示す図である。It is a figure which shows the structure of the control apparatus which concerns on the 2nd Embodiment of this invention. 本発明の第3の実施形態に係る交流電動機駆動システムの構成を示す図である。It is a figure which shows the structure of the alternating current motor drive system which concerns on the 3rd Embodiment of this invention. 本発明の第4の実施形態に係る油圧制御システムの構成を示す図である。It is a figure which shows the structure of the hydraulic control system which concerns on the 4th Embodiment of this invention. 本発明の第4の実施形態に係る油圧制御システムの構成の変形例を示す図である。It is a figure which shows the modification of a structure of the hydraulic control system which concerns on the 4th Embodiment of this invention. 本発明の第5の実施形態に係る位置決めシステムの構成を示す図である。It is a figure which shows the structure of the positioning system which concerns on the 5th Embodiment of this invention.
(第1の実施形態)
 図1は、本発明の第1の実施形態に係る交流電動機の制御装置2を含む交流電動機駆動システムの構成を示す図である。この交流電動機駆動システムは、q軸電流指令発生器1、制御装置2、インバータ3および交流電動機4により構成されている。交流電動機4は、三相交流電動機であり、たとえば永久磁石型同期電動機(PMSM)が用いられる。
(First embodiment)
FIG. 1 is a diagram showing a configuration of an AC motor drive system including an AC motor control device 2 according to a first embodiment of the present invention. This AC motor drive system includes a q-axis current command generator 1, a control device 2, an inverter 3, and an AC motor 4. The AC motor 4 is a three-phase AC motor, for example, a permanent magnet type synchronous motor (PMSM) is used.
 q軸電流指令発生器1は、交流電動機4の出力トルクに相当するq軸電流指令Iq*を発生する回路である。このq軸電流指令発生器1は、制御装置2の上位に位置しており、通常、交流電動機4の回転速度ωrが所定速度になるように、q軸電流指令Iq*を発生させる仕組みとなっている。q軸電流指令発生器1から出力されたq軸電流指令Iq*は、制御装置2中の減算器6bに入力される。 The q-axis current command generator 1 is a circuit that generates a q-axis current command Iq * corresponding to the output torque of the AC motor 4. The q-axis current command generator 1 is positioned above the control device 2 and normally has a mechanism for generating the q-axis current command Iq * so that the rotational speed ωr of the AC motor 4 becomes a predetermined speed. ing. The q-axis current command Iq * output from the q-axis current command generator 1 is input to the subtracter 6b in the control device 2.
 制御装置2は、インバータ3に設けられた各スイッチング素子のスイッチング状態に応じて交流電動機4の駆動を制御するものであり、q軸電流指令発生器1から入力されたq軸電流指令Iq*に相当するトルクを交流電動機4が発生するように動作する。この制御装置2は、電圧指令発生部20、電圧補正部10、PWM発生部11、電流推定部21、位置・速度推定部15および速度変換部16により構成される。 The control device 2 controls the driving of the AC motor 4 according to the switching state of each switching element provided in the inverter 3, and the q-axis current command Iq * input from the q-axis current command generator 1 is controlled. The AC motor 4 operates so as to generate a corresponding torque. The control device 2 includes a voltage command generation unit 20, a voltage correction unit 10, a PWM generation unit 11, a current estimation unit 21, a position / speed estimation unit 15, and a speed conversion unit 16.
 電圧指令発生部20は、q軸電流指令発生器1から入力されたq軸電流指令Iq*に基づいて、三相すなわちU相、V相およびW相の各電圧指令を発生する部分である。この電圧指令発生部20は、d軸電流指令発生器5、減算器6a、減算器6b、d軸電流制御器7、q軸電流制御器8およびdq逆変換器9により構成される。 The voltage command generator 20 is a part that generates voltage commands for three phases, that is, U-phase, V-phase, and W-phase, based on the q-axis current command Iq * input from the q-axis current command generator 1. The voltage command generator 20 includes a d-axis current command generator 5, a subtractor 6 a, a subtractor 6 b, a d-axis current controller 7, a q-axis current controller 8, and a dq inverse converter 9.
 d軸電流指令発生器5は、交流電動機4の励磁電流に相当するd軸電流指令Id*を発生する。なお、交流電動機4が非突極型の永久磁石モータであれば、通常はd軸電流指令Id*は零となる。d軸電流指令発生器5から出力されたd軸電流指令Id*は、減算器6aに入力される。 The d-axis current command generator 5 generates a d-axis current command Id * corresponding to the excitation current of the AC motor 4. If the AC motor 4 is a non-salient permanent magnet motor, the d-axis current command Id * is normally zero. The d-axis current command Id * output from the d-axis current command generator 5 is input to the subtractor 6a.
 減算器6aは、d軸電流指令発生器5から入力されたd軸電流指令Id*と、電流推定部21により推定されたd軸電流Idとの間のd軸電流偏差を求める減算器である。このd軸電流偏差は、d軸電流制御器7に入力される。 The subtractor 6 a is a subtracter for obtaining a d-axis current deviation between the d-axis current command Id * input from the d-axis current command generator 5 and the d-axis current Id estimated by the current estimation unit 21. . This d-axis current deviation is input to the d-axis current controller 7.
 減算器6bは、q軸電流指令発生器1から入力されたq軸電流指令Iq*と、電流推定部21により推定されたq軸電流Iqとの間のq軸電流偏差を求める減算器である。このq軸電流偏差は、q軸電流制御器8に入力される。 The subtractor 6b is a subtracter for obtaining a q-axis current deviation between the q-axis current command Iq * input from the q-axis current command generator 1 and the q-axis current Iq estimated by the current estimation unit 21. . This q-axis current deviation is input to the q-axis current controller 8.
 d軸電流制御器7は、減算器6aから入力されたd軸電流偏差が零になるように、dq座標軸上のd軸電圧指令Vd*を演算する。d軸電流制御器7により演算されたd軸電圧指令Vd*は、dq逆変換器9に入力される。 The d-axis current controller 7 calculates the d-axis voltage command Vd * on the dq coordinate axis so that the d-axis current deviation input from the subtractor 6a becomes zero. The d-axis voltage command Vd * calculated by the d-axis current controller 7 is input to the dq inverse converter 9.
 q軸電流制御器8は、減算器6bから入力されたq軸電流偏差が零になるように、dq座標軸上のq軸電圧指令Vq*を演算する。q軸電流制御器8により演算されたq軸電圧指令Vq*は、dq逆変換器9に入力される。 The q-axis current controller 8 calculates the q-axis voltage command Vq * on the dq coordinate axis so that the q-axis current deviation input from the subtractor 6b becomes zero. The q-axis voltage command Vq * calculated by the q-axis current controller 8 is input to the dq inverse converter 9.
 dq逆変換器9は、d軸電流制御器7およびq軸電流制御器8から入力されたdq座標軸(磁束軸-磁束軸直交軸)系のd軸電圧指令Vd*およびq軸電圧Vq*を、三相交流座標上の電圧指令に変換する回路である。このdq逆変換器9は、位置・速度推定部15から入力される位相角θdcに基づいて、d軸電圧指令Vd*およびq軸電圧指令Vq*を三相交流座標系の電圧指令Vu*、Vv*およびVw*に変換する。そして、変換後の各電圧指令、すなわちU相の電圧指令Vu*、V相の電圧指令Vv*およびW相の電圧指令Vw*を電圧補正部10に出力する。 The dq reverse converter 9 receives the d-axis voltage command Vd * and the q-axis voltage Vq * of the dq coordinate axis (magnetic flux axis-magnetic flux axis orthogonal axis) system inputted from the d-axis current controller 7 and the q-axis current controller 8. This is a circuit that converts the voltage command into a three-phase AC coordinate. The dq inverse converter 9 converts the d-axis voltage command Vd * and the q-axis voltage command Vq * into the voltage command Vu * of the three-phase AC coordinate system based on the phase angle θdc input from the position / velocity estimation unit 15. Convert to Vv * and Vw *. Each voltage command after conversion, that is, the U-phase voltage command Vu *, the V-phase voltage command Vv *, and the W-phase voltage command Vw * is output to the voltage correction unit 10.
 電圧指令発生部20では、以上説明したような各構成要素の動作により、三相の電圧指令Vu*、Vv*およびVw*を発生することができる。 The voltage command generator 20 can generate three-phase voltage commands Vu *, Vv *, and Vw * by the operation of each component as described above.
 電圧補正部10は、電圧指令発生部20から出力された三相の電圧指令Vu*、Vv*およびVw*をそれぞれ補正し、その補正結果を補正後の電圧指令Vu**、Vv**およびVw**としてPWM発生部11に出力する。なお、電圧補正部10により行われる電圧指令の補正方法については、後で詳細に説明する。 The voltage correction unit 10 corrects the three-phase voltage commands Vu *, Vv *, and Vw * output from the voltage command generation unit 20, respectively, and corrects the corrected voltage commands Vu **, Vv **, and Output to the PWM generator 11 as Vw **. The voltage command correction method performed by the voltage correction unit 10 will be described in detail later.
 PWM発生部11は、電圧補正部10から出力された補正後の三相の電圧指令Vu**、Vv**およびVw**を所定周期の三角波キャリアと比較し、その比較結果に基づいて三相のPWM波形を発生する。このPWM発生部11からは、三相のPWM波形として、U相、V相、W相の各相に対応するPWM信号UP、VP、WPと、これらの反転信号であるPWM信号UN、VN、WNとが発生される。PWM発生部11により発生されたこれらのPWM信号は、制御装置2からインバータ3に出力される。 The PWM generation unit 11 compares the corrected three-phase voltage commands Vu **, Vv **, and Vw ** output from the voltage correction unit 10 with a triangular wave carrier of a predetermined period, and based on the comparison result, Generate phase PWM waveform. From this PWM generator 11, as three-phase PWM waveforms, PWM signals UP, VP, WP corresponding to the U-phase, V-phase, and W-phase, and PWM signals UN, VN, WN is generated. These PWM signals generated by the PWM generator 11 are output from the control device 2 to the inverter 3.
 電流推定部21は、インバータ3から入力される直流電流信号IDCに基づいて、インバータ3において直流母線34に流れる直流電流の検出結果を取得し、その検出結果に基づいて交流電動機4の相電流およびdq座標上の電流を求める部分である。この電流推定部21は、サンプルホールド回路12、電流再現器13およびdq変換器14により構成される。 The current estimation unit 21 acquires a detection result of the DC current flowing in the DC bus 34 in the inverter 3 based on the DC current signal IDC input from the inverter 3, and based on the detection result, the phase current of the AC motor 4 and This is a part for obtaining the current on the dq coordinate. The current estimation unit 21 includes a sample and hold circuit 12, a current reproducer 13, and a dq converter 14.
 サンプルホールド回路12は、インバータ3から入力される直流電流信号IDCを所定のサンプリングタイミングでサンプリングすることで、インバータ3の直流母線34に流れる直流電流の検出結果を取得する。このサンプルホールド回路12によるサンプリングタイミングは、PWM発生部11において用いられる三角波キャリアに基づいて決定される。なお、この点については、後で詳しく説明する。 The sample hold circuit 12 acquires the detection result of the direct current flowing through the direct current bus 34 of the inverter 3 by sampling the direct current signal IDC input from the inverter 3 at a predetermined sampling timing. The sampling timing by the sample and hold circuit 12 is determined based on the triangular wave carrier used in the PWM generator 11. This point will be described in detail later.
 電流再現器13は、サンプルホールド回路12により取得された直流電流の検出結果に基づいて、交流電動機4のU相、V相、W相にそれぞれ流れる相電流Iu、Iv、Iwを再現し、これらの相電流の再現値Iuc、Ivc、Iwcを求める。電流再現器13により求められた相電流の再現値Iuc、Ivc、Iwcは、dq変換器14に対して出力される。 The current reproducer 13 reproduces the phase currents Iu, Iv, and Iw flowing in the U phase, V phase, and W phase of the AC motor 4 based on the detection result of the DC current acquired by the sample hold circuit 12, respectively. The reproduced values Iuc, Ivc, and Iwc of the phase current are obtained. The phase current reproduction values Iuc, Ivc, and Iwc obtained by the current reproducer 13 are output to the dq converter 14.
 dq変換器14は、電流再現器13から相電流の再現値Iuc、Ivc、Iwcを受けて、これらをdq座標上の電流値Id、Iqに変換する回路である。このdq変換器14は、位置・速度推定部15から入力される位相角θdcに基づいて、三相の相電流の再現値Iuc、Ivc、Iwcを、d軸電流Idおよびq軸電流Iqに変換する。変換されたd軸電流Idおよびq軸電流Iqの値は、減算器6a、6bにそれぞれ出力され、前述のようなd軸電流偏差、q軸電流偏差の計算に用いられる。 The dq converter 14 is a circuit that receives the phase current reproduction values Iuc, Ivc, and Iwc from the current reproduction unit 13 and converts them into current values Id and Iq on the dq coordinates. The dq converter 14 converts the reproduction values Iuc, Ivc, Iwc of the three-phase phase currents into the d-axis current Id and the q-axis current Iq based on the phase angle θdc input from the position / velocity estimation unit 15. To do. The converted values of the d-axis current Id and the q-axis current Iq are output to the subtracters 6a and 6b, respectively, and used for the calculation of the d-axis current deviation and the q-axis current deviation as described above.
 電流推定部21では、以上説明したような各構成要素の動作により、三相の相電流の再現値Iuc、Ivc、Iwcを求めると共に、d軸電流Idおよびq軸電流Iqの値を求めることができる。 The current estimation unit 21 obtains the reproduction values Iuc, Ivc, and Iwc of the three-phase phase currents and the values of the d-axis current Id and the q-axis current Iq by the operation of each component as described above. it can.
 位置・速度推定部15は、交流電動機4の回転子位置(交流電動機4が同期電動機の場合)または二次磁束位相(交流電動機4が誘導電動機の場合)を推定演算し、その演算結果を位相角θdcとして出力する。さらに、この位相角θdcの値に基づいて、交流電動機4の電気角速度ω1を推定演算し、その演算結果を出力する。なお、これらの推定演算の方法については、周知であるため、説明を省略する。 The position / speed estimation unit 15 estimates and calculates the rotor position of the AC motor 4 (when the AC motor 4 is a synchronous motor) or the secondary magnetic flux phase (when the AC motor 4 is an induction motor), and calculates the phase of the calculation result. Output as angle θdc. Further, based on the value of the phase angle θdc, the electric angular velocity ω1 of the AC motor 4 is estimated and calculated, and the calculation result is output. In addition, since these estimation calculation methods are well-known, description is abbreviate | omitted.
 速度変換部16は、交流電動機4の極数Pに基づいて、位置・速度推定部15から出力された電気角速度ω1を回転速度ωrに変換する。速度変換部16により求められた回転速度ωrは、q軸電流指令発生器1において入力され、前述のq軸電流指令Iq*を発生するのに用いられる。 The speed converter 16 converts the electrical angular velocity ω1 output from the position / speed estimator 15 into a rotational speed ωr based on the number P of poles of the AC motor 4. The rotational speed ωr obtained by the speed converter 16 is input to the q-axis current command generator 1 and used to generate the q-axis current command Iq *.
 インバータ3は、交流電動機4に接続されており、制御装置2のPWM発生部11から入力されたPWM信号に基づいて交流電圧を発生して交流電動機4に供給する。このインバータ3は、直流電源31、インバータ主回路32、出力プリドライバ33、直流母線34およびシャント電流検出器35により構成される。 The inverter 3 is connected to the AC motor 4, generates an AC voltage based on the PWM signal input from the PWM generator 11 of the control device 2, and supplies the AC voltage to the AC motor 4. The inverter 3 includes a DC power supply 31, an inverter main circuit 32, an output predriver 33, a DC bus 34, and a shunt current detector 35.
 直流電源31は、直流母線34を介してインバータ主回路32に接続されており、直流電圧VDCを発生する。 The DC power supply 31 is connected to the inverter main circuit 32 via the DC bus 34 and generates a DC voltage VDC.
 インバータ主回路32は、正極側にそれぞれ設けられた三相のスイッチング素子Sup、SvpおよびSwpと、負極側にそれぞれ設けられた三相のスイッチング素子Sun、SvnおよびSwnとを有している。スイッチング素子Sup、SvpおよびSwpは、直流母線34を介して直流電源31の正極側にそれぞれ接続されており、スイッチング素子Sun、SvnおよびSwnは、直流母線34を介して直流電源31の負極側にそれぞれ接続されている。正極側のスイッチング素子Sup、Svp、Swpと負極側のスイッチング素子Sun、Svn、Swnとの間には、交流電動機4のU相、V相、W相にそれぞれつながる交流線が接続されている。 The inverter main circuit 32 has three-phase switching elements Sup, Svp, and Swp provided on the positive electrode side, and three-phase switching elements Sun, Svn, and Swn provided on the negative electrode side, respectively. The switching elements Sup, Svp and Swp are connected to the positive electrode side of the DC power supply 31 via the DC bus 34, and the switching elements Sun, Svn and Swn are connected to the negative electrode side of the DC power supply 31 via the DC bus 34. Each is connected. Between the positive-side switching elements Sup, Svp, Swp and the negative-side switching elements Sun, Svn, Swn, AC lines connected to the U-phase, V-phase, and W-phase of the AC motor 4 are connected.
 出力プリドライバ33は、制御装置2のPWM発生部11から出力された前述のPWM信号UP、VP、WP、UN、VNおよびWNに基づいて、インバータ主回路32のスイッチング素子Sup、Svp、Swp、Sun、SvnおよびSwnのスイッチング状態をそれぞれ制御する。この出力プリドライバ33により各スイッチング素子のスイッチング状態が制御されることで、直流電源31から供給される直流電圧VDCが三相の交流電圧Vu、Vv、Vwに変換されて交流電動機4に出力される。 The output pre-driver 33 is based on the aforementioned PWM signals UP, VP, WP, UN, VN and WN output from the PWM generator 11 of the control device 2, and the switching elements Sup, Svp, Swp, The switching states of Sun, Svn and Swn are controlled respectively. By controlling the switching state of each switching element by the output pre-driver 33, the DC voltage VDC supplied from the DC power supply 31 is converted into a three-phase AC voltage Vu, Vv, Vw and output to the AC motor 4. The
 シャント電流検出器35は、直流母線34に流れる直流電流を検出する過電流保護用の回路であり、直流母線34上に設けられている。このシャント電流検出器35による直流電流の検出結果は、前述の直流電流信号IDCとして、制御装置2の電流推定部21に出力される。 The shunt current detector 35 is a circuit for overcurrent protection that detects a direct current flowing through the direct current bus 34, and is provided on the direct current bus 34. The detection result of the direct current by the shunt current detector 35 is output to the current estimation unit 21 of the control device 2 as the above-described direct current signal IDC.
 なお、以上説明した交流電動機駆動システムの構成は、センサレスベクトル制御系をベースにしているが、センサ付のベクトル制御にも本発明を適用可能である。 The configuration of the AC motor drive system described above is based on a sensorless vector control system, but the present invention can also be applied to vector control with a sensor.
 本実施形態による交流電動機駆動システムは、以上説明したような構成を有している。この構成における本発明の主な特徴部分は、電圧指令の補正を行う電圧補正部10と、サンプルホールド回路12による直流電流信号IDCのサンプリングタイミングである。以下では、これらについて詳細に説明する。 The AC motor drive system according to the present embodiment has the configuration as described above. The main characteristic part of the present invention in this configuration is the voltage correction unit 10 for correcting the voltage command and the sampling timing of the DC current signal IDC by the sample hold circuit 12. These will be described in detail below.
 最初に、本発明による電圧指令の補正を行わない場合のパルス幅変調(PWM)の様子について、図2~4の例を用いて説明する。図2は、電圧指令の補正を行わない場合のPWM波形およびスイッチ状態の例を示す図である。制御装置2において、電圧指令発生部20により、たとえば図2に示すような電圧指令Vu*、Vv*およびVw*が発生されたとする。電圧補正部10により電圧指令の補正を行わない場合、PWM発生部11は、これらの電圧指令を三角波キャリアとそれぞれ比較することで、図2に示すようなPWM信号UP、VP、WPと、これらをそれぞれ反転したPWM信号UN、VN、WNとを生成する。 First, the state of pulse width modulation (PWM) when voltage command correction according to the present invention is not performed will be described with reference to the examples of FIGS. FIG. 2 is a diagram illustrating an example of a PWM waveform and a switch state when the voltage command is not corrected. In the control device 2, it is assumed that, for example, voltage commands Vu *, Vv * and Vw * as shown in FIG. When the voltage command is not corrected by the voltage correction unit 10, the PWM generation unit 11 compares these voltage commands with a triangular wave carrier, respectively, to thereby generate PWM signals UP, VP, WP as shown in FIG. PWM signals UN, VN, and WN are generated.
 インバータ3では、上記のPWM信号UP、VP、WP、UN、VN、WNに応じて、出力プリドライバ33によりインバータ主回路32のスイッチング素子Sup、Svp、Swp、Sun、Svn、Swnのスイッチ状態がそれぞれ制御される。その結果、図2に示すように各スイッチング素子のスイッチ状態が変化する。なお、図2では、U相、V相、W相のスイッチング状態をそれぞれ括弧内の1または0の符号に対応付けて、電圧ベクトルとして表している。たとえば、電圧ベクトルV(1,0,0)は、U相では正極側のスイッチング素子Supがオン(負極側のスイッチング素子Sunがオフ)であり、V相、W相では負極側のスイッチング素子Svn、Swnがオン(正極側のスイッチング素子Svp、Swpがオフ)であることを表している。 In the inverter 3, the switching state of the switching elements Sup, Svp, Swp, Sun, Svn, Swn of the inverter main circuit 32 is changed by the output pre-driver 33 according to the PWM signals UP, VP, WP, UN, VN, WN. Each is controlled. As a result, the switch state of each switching element changes as shown in FIG. In FIG. 2, the switching states of the U phase, the V phase, and the W phase are respectively expressed as voltage vectors in association with 1 or 0 in parentheses. For example, in the voltage vector V (1, 0, 0), the switching element Sup on the positive electrode side is on (the switching element Sun on the negative electrode side is off) in the U phase, and the switching element Svn on the negative electrode side in the V phase and W phase. , Swn is ON (switching elements Svp and Swp on the positive electrode side are OFF).
 インバータ3は、スイッチング素子Sup、Svp、Swp、Sun、Svn、Swnの各スイッチ状態の組み合わせに応じて、8通りの電圧ベクトルを交流電動機4に出力可能である。すなわち、正極側の全てのスイッチング素子Sup、Svp、Swpをオンとし、負極側の全てのスイッチング素子Sun、Svn、Swnをオフとすることで、電圧ベクトルV(1,1,1)を出力することができる。また、これとは反対に、正極側の全てのスイッチング素子Sup、Svp、Swpをオフとし、負極側の全てのスイッチング素子Sun、Svn、Swnをオンとすることで、電圧ベクトルV(0,0,0)を出力することができる。これらの電圧ベクトルは、いずれも相電流が流れない零ベクトルである。さらに、U相、V相およびW相のいずれか少なくとも1相において、正極側のスイッチング素子をオンとし、負極側のスイッチング素子をオフとして、それ以外の相では、正極側のスイッチング素子をオフとし、負極側のスイッチング素子をオンとする。これにより、零ベクトル以外に、6種類の各電圧ベクトルV(1,0,0)、V(0,1,0)、V(0,0,1)、V(1,1,0)、V(1,0,1)、V(0,1,1)を出力することができる。 The inverter 3 can output eight kinds of voltage vectors to the AC motor 4 according to combinations of switch states of the switching elements Sup, Svp, Swp, Sun, Svn, and Swn. That is, by turning on all the switching elements Sup, Svp, Swp on the positive side and turning off all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (1, 1, 1) is output. be able to. On the contrary, by turning off all the switching elements Sup, Svp, Swp on the positive side and turning on all the switching elements Sun, Svn, Swn on the negative side, the voltage vector V (0, 0 , 0) can be output. These voltage vectors are all zero vectors in which no phase current flows. Further, in at least one of the U phase, the V phase, and the W phase, the positive side switching element is turned on, the negative side switching element is turned off, and in the other phases, the positive side switching element is turned off. The negative side switching element is turned on. Thus, in addition to the zero vector, each of the six types of voltage vectors V (1, 0, 0), V (0, 1, 0), V (0, 0, 1), V (1, 1, 0), V (1, 0, 1) and V (0, 1, 1) can be output.
 図3は、図2に電圧ベクトルで示した各スイッチ状態において、インバータ主回路32のスイッチング素子Sup、Svp、Swp、Sun、Svn、Swnにそれぞれ流れる電流の様子を示す図である。図3(a)は、電圧ベクトルV(1,1,1)における電流の様子を示しており、図3(b)は、電圧ベクトルV(1,1,0)における電流の様子を示しており、図3(c)は、電圧ベクトルV(1,0,0)における電流の様子を示しており、図3(d)は、電圧ベクトルV(0,0,0)における電流の様子を示している。これらの図では、電流が流れる部分を太線で示している。 FIG. 3 is a diagram showing the states of currents flowing through the switching elements Sup, Svp, Swp, Sun, Svn, and Swn of the inverter main circuit 32 in the respective switch states indicated by voltage vectors in FIG. FIG. 3A shows the state of current in the voltage vector V (1,1,1), and FIG. 3B shows the state of current in the voltage vector V (1,1,0). 3 (c) shows the state of current in the voltage vector V (1, 0, 0), and FIG. 3 (d) shows the state of current in the voltage vector V (0, 0, 0). Show. In these drawings, the portion through which the current flows is indicated by a thick line.
 図3(b)、図3(c)において示すように、電圧ベクトルV(1,1,0)またはV(1,0,0)では、シャント電流検出器35を介してインバータ3の直流母線34に、W相の相電流IwまたはU相の相電流Iuに応じた直流シャント電流IDCが流れる。したがって、これらの電圧ベクトルに対応するスイッチ状態のときにシャント電流IDCを検出して得られた直流電流信号IDCを前述のようにサンプリングすることで、2つの相電流Iw、Iuの検出が可能となる。 As shown in FIGS. 3 (b) and 3 (c), the voltage vector V (1,1,0) or V (1,0,0) has a DC bus of the inverter 3 via the shunt current detector 35. A direct current shunt current I DC corresponding to the W-phase phase current Iw or the U-phase phase current Iu flows through 34. Therefore, by sampling as described above the direct current signal IDC obtained by detecting the shunt current I DC at switch state corresponding to these voltage vectors, two phase currents Iw, can be detected Iu It becomes.
 図4は、交流電動機4に流れる各相のモータ電流(相電流)Iu、IvおよびIwと、直流母線34に流れるシャント電流IDCおよびスイッチ状態との関係を示す図である。たとえば、図4(c)に示す各スイッチ状態に対して、図4(a)に示すような相電流が流れると共に、図4(b)に示すシャント電流IDCが直流母線34に流れる。したがって、図4(b)に示すように、スイッチ状態ごとにシャント電流IDCをサンプリングすることで、相電流Iw、Iuを検出することができる。 4, each phase of the motor current flowing to the AC motor 4 (phase currents) Iu, and Iv and Iw, a diagram showing the relationship between the shunt current I DC and the switch state flows through the DC bus 34. For example, for each switch state shown in FIG. 4 (c), together with the flow phase current as shown in FIG. 4 (a), the shunt current I DC shown in FIG. 4 (b) flows to the DC bus 34. Accordingly, as shown in FIG. 4 (b), by sampling the shunt current I DC for each switch state, it is possible to detect the phase current Iw, Iu.
 上記のような相電流の検出方法を採用した場合、キャリア信号の半周期内に2回の電流サンプリングとアナログデジタル変換を行う必要がある。また、スイッチ状態の変化に応じて電流サンプリングのタイミングを任意に設定する必要がある。しかし、これらの機能を実現可能なマイコンは、一般的に高価格であるという問題がある。また、スイッチ状態が変化した直後は、スイッチの切り替えに伴ってシャント電流IDCに多くの電流リプルが含まれるため、これをサンプリングすると電流検出誤差が大きくなってしまうという問題もある。特に、こうした電流検出誤差にはオフセット成分が含まれてしまう傾向があるため、トルク精度に大きな悪影響を与えてしまう。 When the phase current detection method as described above is employed, it is necessary to perform current sampling and analog-digital conversion twice within a half cycle of the carrier signal. In addition, it is necessary to arbitrarily set the current sampling timing in accordance with the change in the switch state. However, there is a problem that microcomputers that can realize these functions are generally expensive. Further, immediately after the switch state is changed, because it contains a lot of current ripple to shunt current I DC in accordance with the switching of the switch, there is also this problem current detection error is sampled increases. In particular, such a current detection error tends to include an offset component, which greatly affects torque accuracy.
 これらの問題点を解決する方法の一つとして、電圧指令Vu*、Vv*およびVw*を補正して、電流サンプリングのタイミングを固定化することが考えられる。その具体的な方法について、図5、6を参照して以下に説明する。図5は、従来の二相変調方式をベースとして電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図であり、図6は、図5とは別の方法で電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図である。 As one method for solving these problems, it is conceivable to correct the voltage commands Vu *, Vv * and Vw * to fix the current sampling timing. The specific method will be described below with reference to FIGS. FIG. 5 is a diagram showing an example of a PWM waveform and a switch state when a voltage command is corrected based on the conventional two-phase modulation method, and FIG. 6 is a diagram different from FIG. It is a figure which shows the example of the PWM waveform at the time of correct | amending and a switch state.
 図5の例では、図2に示した補正前の電圧指令Vu*、Vv*およびVw*に対して、これらの間隔を変えずに全体を上昇させ、最も高い電圧指令(ここではVu*)が三角波キャリアの頂点と一致するように補正することで、補正後の電圧指令Vu**、Vv**およびVw**を求めている。この場合、三角波キャリアの頂点のタイミングで電流サンプリングを行うと、U相の相電流Iuをシャント電流IDCとして検出できるが、他の相電流については検出することができない。 In the example of FIG. 5, the voltage command Vu *, Vv *, and Vw * before correction shown in FIG. 2 is raised without changing their interval, and the highest voltage command (here, Vu *). Is corrected so as to coincide with the apex of the triangular wave carrier to obtain corrected voltage commands Vu **, Vv ** and Vw **. In this case, when a current is sampled at the timing of the apexes of the triangular wave carrier, it can detect the phase currents Iu of the U phase as a shunt current I DC, can not be detected for the other phase currents.
 そこで図6に示すように、三角波キャリアの1周期ごとに、補正後の電圧指令Vu**、Vw**の大きさが三角波キャリアの頂点と交互に一致するように、各電圧指令Vu*、Vv*およびVw*を補正する。この補正によって得られた補正後の電圧指令Vu**、Vv**、Vw**は、その最大値と中間値の間隔(図6の例では、電圧指令Vu**が最大のときには電圧指令Vv**との間隔、電圧指令Vw**が最大のときには電圧指令Vu**との間隔)が、所定の最小電圧以上となるように設定されている。また、三角波キャリアの連続する2周期について、補正後の電圧指令Vu**、Vv**、Vw**をそれぞれ平均化すると、その各平均値の間隔は、元の補正前の電圧指令Vu*、Vv*、Vw*の間隔と等しくなるように設定されている。 Therefore, as shown in FIG. 6, each voltage command Vu *, Vw **, Vw ** is corrected so that the magnitude of the corrected voltage commands Vu ** and Vw ** alternately coincide with the apex of the triangular wave carrier. Correct Vv * and Vw *. The corrected voltage commands Vu **, Vv **, and Vw ** obtained by this correction are the intervals between the maximum value and the intermediate value (in the example of FIG. 6, when the voltage command Vu ** is the maximum, the voltage command The interval between Vv ** and the interval between voltage command Vu ** when voltage command Vw ** is maximum is set to be equal to or greater than a predetermined minimum voltage. Further, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged for two consecutive cycles of the triangular wave carrier, the interval between the average values is the original voltage command Vu * before correction. , Vv * and Vw * are set to be equal to each other.
 図6のように電圧指令Vu*、Vv*およびVw*を補正して、三角波キャリアの頂点のタイミングで電流サンプリングを行うと、U相の相電流IuとW相の相電流Iwを交互にシャント電流IDCとして検出することができる。なお、図5、6では、三角波キャリアの頂点部分に合わせて電圧指令Vu*、Vv*、Vw*を補正する例を説明したが、反対に、三角波キャリアの谷部分に合わせて電圧指令Vu*、Vv*、Vw*を補正することもできる。 When the voltage commands Vu *, Vv *, and Vw * are corrected as shown in FIG. 6 and current sampling is performed at the timing of the apex of the triangular wave carrier, the U-phase phase current Iu and the W-phase phase current Iw are shunted alternately. it can be detected as a current I DC. 5 and 6, the example in which the voltage commands Vu *, Vv *, and Vw * are corrected in accordance with the apex portion of the triangular wave carrier has been described. Conversely, the voltage command Vu * is adjusted in accordance with the valley portion of the triangular wave carrier. , Vv * and Vw * can be corrected.
 しかし、図6のような電圧指令の補正方法および相電流の検出方法を採用した場合、補正後の電圧指令Vu**、Vv**、Vw**は、三角波キャリアの正側ピーク方向に偏って補正される。このようにすると、シャント電流IDCに含まれる電流リプルが増大してしまう傾向がある。さらに、三角波キャリアの頂点のタイミングで電流サンプリングを行うため、多くの高調波リプルを含んだ電流をサンプリングしてしまうことになる。その結果、電流検出誤差が大きくなり、トルク精度が劣化してしまうという問題がある。 However, when the voltage command correction method and the phase current detection method as shown in FIG. 6 are adopted, the corrected voltage commands Vu **, Vv **, and Vw ** are biased toward the positive peak direction of the triangular wave carrier. Corrected. In this way, there is a tendency that the current ripples contained in the shunt current I DC increases. Furthermore, since current sampling is performed at the timing of the apex of the triangular wave carrier, a current containing many harmonic ripples is sampled. As a result, there is a problem that current detection error becomes large and torque accuracy is deteriorated.
 そこで本発明では、電圧補正部10において、以下に説明するような方法により、電圧指令Vu*、Vv*、Vw*を補正する。そして、補正後の電圧指令Vu**、Vv**およびVw**を基にしたPWM信号によってインバータ3の直流母線34に流れるシャント電流IDCに応じた直流電流信号IDCを、サンプルホールド回路12において、以下に説明するようなタイミングでサンプリングする。 Therefore, in the present invention, the voltage correction unit 10 corrects the voltage commands Vu *, Vv *, and Vw * by the method described below. Then, the voltage command Vu ** after correction, the DC current signal IDC in accordance with the shunt current I DC flowing through the DC bus 34 of inverter 3 by the PWM signal based on the Vv ** and Vw **, sample and hold circuit 12 The sampling is performed at the timing as described below.
 図7は、本発明による電圧指令の補正を行った場合のPWM波形およびスイッチ状態の例を示す図である。 FIG. 7 is a diagram showing an example of a PWM waveform and a switch state when the voltage command is corrected according to the present invention.
 図7において、三角波キャリアの最初の周期(図7中の[A]の期間)では、図2に示した補正前の電圧指令Vu*、Vv*、Vw*に対して、これを図のように変形させることで補正後の電圧指令Vu**、Vv**およびVw**を求め、PWM制御を行っている。ここで、補正前の元の電圧指令は、Vu*>Vv*>Vw*の関係にある。以下の説明では、補正前の電圧指令値が最大の相を最大相(ここではU相)、最小の相を最小相(ここではW相)、中間の大きさの相を中間相(ここではV相)と呼ぶことにする。 In FIG. 7, in the first period of the triangular wave carrier (period [A] in FIG. 7), the voltage commands Vu *, Vv *, and Vw * before correction shown in FIG. Thus, the corrected voltage commands Vu **, Vv ** and Vw ** are obtained and PWM control is performed. Here, the original voltage command before correction has a relationship of Vu *> Vv *> Vw *. In the following description, the phase with the maximum voltage command value before correction is the maximum phase (here, U phase), the minimum phase is the minimum phase (here, W phase), and the medium phase is the intermediate phase (here, V phase).
 上記の期間Aにおける各電圧指令の補正量は、前半部分、すなわち三角波キャリアの上り部分における補正量と、後半部分、すなわち三角波キャリアの下り部分における補正量とを合計すると、差し引きで0となるように設定されている。すなわち、補正後の電圧指令Vu**、Vv**、Vw**を期間Aについてそれぞれ平均化すると、その各平均値同士の間隔は、元の補正前の電圧指令Vu*、Vv*、Vw*同士の間隔と等しくなるように設定されている。 The correction amount of each voltage command in the period A is such that the sum of the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier, is 0 by subtraction. Is set to That is, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged for the period A, the intervals between the average values are the original voltage commands Vu *, Vv *, and Vw before correction. * It is set to be equal to the distance between each other.
 上記のような電圧指令の補正を電圧補正部10において行うことで、PWM発生部11において、図7に示すようなPWM信号UP、VP、WPと、これらをそれぞれ反転したPWM信号UN、VN、WNとが生成される。その結果、図7に示す順番に従って、インバータ3から出力される電圧ベクトルが変化する。すなわち、期間Aでは、零ベクトルV(1,1,1)、電圧ベクトルV(1,1,0)、電圧ベクトルV(1,0,0)、零ベクトルV(0,0,0)、電圧ベクトルV(0,0,1)、零ベクトルV(1,1,1)の順に、インバータ3から出力される。 By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in period A, zero vector V (1, 1, 1), voltage vector V (1, 1, 0), voltage vector V (1, 0, 0), zero vector V (0, 0, 0), The voltage vector V (0, 0, 1) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.
 零ベクトルV(1,1,1)またはV(0,0,0)以外の電圧ベクトルが出力されているときには、その電圧ベクトルに応じた相電流がシャント電流IDCとして直流母線34に流れ、シャント電流検出器35により検出される。図8は、期間Aにおける電圧ベクトルと検出される相電流の関係を示した一覧表である。なお、図8の表では、期間Aにおいて順に出力される零ベクトル以外の各電圧ベクトルを、それぞれ第1、第2、第3の電圧ベクトルとして表している。 When the voltage vector other than zero vectors V (1, 1, 1) or V (0,0,0) is output flows to the DC bus 34 phase current corresponding to the voltage vector as a shunt current I DC, It is detected by the shunt current detector 35. FIG. 8 is a table showing the relationship between the voltage vector in the period A and the detected phase current. In the table of FIG. 8, each voltage vector other than the zero vector output in order in the period A is represented as a first voltage vector, a second voltage vector, and a third voltage vector, respectively.
 図8の表から、第1の電圧ベクトルV(1,1,0)の出力時、および第3の電圧ベクトルV(0,0,1)の出力時には、W相の相電流Iwがシャント電流検出器35により検出されることが分かる。ただし、第1の電圧ベクトルV(1,1,0)と第3の電圧ベクトルV(0,0,1)とでは、各相のスイッチ状態を表す符号がそれぞれ反転しているため、相電流Iwが反対方向に流れる。なお、第2の電圧ベクトルV(1,0,0)の出力時には、U相の相電流Iuがシャント電流IDCとして流れるが、これはサンプルホールド回路12によるサンプリングの対象とはされない。 From the table of FIG. 8, when the first voltage vector V (1, 1, 0) is output and when the third voltage vector V (0, 0, 1) is output, the W-phase current Iw is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 1, 0) and the third voltage vector V (0, 0, 1), the signs representing the switch states of the respective phases are inverted, so that the phase current Iw flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I DC, which are not subject to sampling by the sample and hold circuit 12.
 図10は、第1の電圧ベクトルV(1,1,0)および第3の電圧ベクトルV(0,0,1)において、シャント電流IDCとして検出される相電流の様子を示す図である。図10(a)は、第1の電圧ベクトルV(1,1,0)におけるスイッチ状態と相電流の様子を示しており、図10(b)は、第3の電圧ベクトルV(0,0,1)におけるスイッチ状態と相電流の様子を示している。 10, in the first voltage vector V (1,1,0) and a third voltage vector V (0,0,1), is a diagram illustrating a state of a phase current is detected as a shunt current I DC . FIG. 10A shows the state of the switch state and the phase current in the first voltage vector V (1,1,0), and FIG. 10B shows the third voltage vector V (0,0). , 1) shows the state of the switch and the phase current.
 図10(a)に示すように、第1の電圧ベクトルV(1,1,0)では、交流電動機4からインバータ3の方向に相電流Iwが流れる。したがって、インバータ3から交流電動機4に向かう方向を正方向とすると、シャント電流検出器35では、符号が負である相電流-Iwがシャント電流IDCとして検出される。一方、図10(b)に示すように、第3の電圧ベクトルV(0,0,1)では、インバータ3から交流電動機4の方向に相電流Iwが流れる。したがって、インバータ3から交流電動機4に向かう方向を正方向とすると、シャント電流検出器35では、符号が正である相電流Iwがシャント電流IDCとして検出される。 As shown in FIG. 10A, the phase current Iw flows from the AC motor 4 to the inverter 3 in the first voltage vector V (1, 1, 0). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the phase current -Iw sign is negative is detected as a shunt current I DC. On the other hand, as shown in FIG. 10B, the phase current Iw flows from the inverter 3 to the AC motor 4 in the third voltage vector V (0, 0, 1). Therefore, when a direction from the inverter 3 to the AC motor 4 is the positive direction, the shunt current detector 35, the code is positive matrix phase current Iw is detected as a shunt current I DC.
 ここで、零ベクトル以外の電圧ベクトルがインバータ3から出力された場合、電流波形が大きく変化するため、各相の相電流には、制御に必要な基本波成分以外の脈動分が含まれてしまう。しかし、上記の第1の電圧ベクトルV(1,1,0)と第3の電圧ベクトルV(0,0,1)のように、互いに符号が反対の電圧ベクトルを三角波キャリアの1周期内で印加し、それぞれで流れる相電流を検出して平均値を求めれば、脈動成分をキャンセルできるため、高精度な検出が可能になる。 Here, when a voltage vector other than the zero vector is output from the inverter 3, the current waveform changes greatly, so that the phase current of each phase includes a pulsation component other than the fundamental wave component necessary for control. . However, like the first voltage vector V (1, 1, 0) and the third voltage vector V (0, 0, 1), voltage vectors having opposite signs are included in one period of the triangular wave carrier. If the average value is obtained by detecting the phase currents that are applied and flowing through each of them, the pulsation component can be canceled, so that highly accurate detection becomes possible.
 すなわち、図7に示したような補正後の電圧指令Vu**、Vv**およびVw**に従って生成されるPWM波形により交流電動機4を駆動することで、期間Aにおいて、第1、第3の電圧ベクトルに応じて相電流-Iw、Iwを検出することができる。これにより、脈動成分を排除した相電流Iwの検出を行うことができる。 In other words, the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG. The phase currents -Iw and Iw can be detected in accordance with the voltage vector. Thereby, it is possible to detect the phase current Iw excluding the pulsation component.
 さらに、図7に示すように、期間Aでは、中間相に対する補正後の電圧指令Vv**が常に零となるように補正している。これにより、三角波キャリアの零クロスのタイミング、すなわち最小値と最大値の中間値をとる時点で、インバータ3の直流母線34に流れるシャント電流IDCをそれぞれ検出すれば、第1の電圧ベクトルV(1,1,0)に対応する相電流-Iwと、第3の電圧ベクトルV(0,0,1)に対応する相電流Iwとを検出することができる。したがって、期間Aの前半にある三角波キャリアの上り区間と、期間Aの後半にある三角波キャリアの下り区間とで、反対符号の電圧ベクトルによる同一相の電流検出を、固定のタイミングで行うことができる。 Further, as shown in FIG. 7, in the period A, the corrected voltage command Vv ** for the intermediate phase is always corrected to zero. Thus, the zero cross timing of the triangular wave carrier, i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current −Iw corresponding to (1, 1, 0) and the phase current Iw corresponding to the third voltage vector V (0, 0, 1) can be detected. Accordingly, in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period A and the downstream section of the triangular wave carrier in the second half of period A. .
 また、図7において、三角波キャリアの次の周期(図7中の[B]の期間)では、図2に示した補正前の電圧指令Vu*、Vv*、Vw*に対して、これを前述の期間Aとは異なる手法により、図のように変形させている。これにより、補正後の電圧指令Vu**、Vv**およびVw**を求め、PWM制御を行っている。 In FIG. 7, in the next period of the triangular wave carrier (period [B] in FIG. 7), the voltage commands Vu *, Vv *, and Vw * before correction shown in FIG. It is deformed as shown in the figure by a method different from the period A. Thus, corrected voltage commands Vu **, Vv ** and Vw ** are obtained and PWM control is performed.
 なお、期間Bにおける各電圧指令の補正量についても、前述の期間Aと同様に、前半部分、すなわち三角波キャリアの上り部分における補正量と、後半部分、すなわち三角波キャリアの下り部分における補正量とを合計すると、差し引きで0となるように設定されている。すなわち、補正後の電圧指令Vu**、Vv**、Vw**を期間Bについてそれぞれ平均化すると、その各平均値同士の間隔は、元の補正前の電圧指令Vu*、Vv*、Vw*の同士の間隔と等しくなるように設定されている。 As for the correction amount of each voltage command in the period B, similarly to the period A described above, the correction amount in the first half portion, that is, the upward portion of the triangular wave carrier, and the correction amount in the second half portion, that is, the downward portion of the triangular wave carrier. The total is set to be 0 by subtraction. That is, when the corrected voltage commands Vu **, Vv **, and Vw ** are averaged over the period B, the intervals between the average values are the original voltage commands Vu *, Vv *, and Vw before correction. It is set to be equal to the interval between *.
 上記のような電圧指令の補正を電圧補正部10において行うことで、PWM発生部11において、図7に示すようなPWM信号UP、VP、WPと、これらをそれぞれ反転したPWM信号UN、VN、WNとが生成される。その結果、図7に示す順番に従って、インバータ3から出力される電圧ベクトルが変化する。すなわち、期間Bでは、零ベクトルV(1,1,1)、電圧ベクトルV(1,0,1)、電圧ベクトルV(1,0,0)、零ベクトルV(0,0,0)、電圧ベクトルV(0,1,0)、零ベクトルV(1,1,1)の順に、インバータ3から出力される。 By performing the voltage command correction as described above in the voltage correction unit 10, the PWM generation unit 11 performs PWM signals UP, VP, WP as shown in FIG. 7 and PWM signals UN, VN, WN is generated. As a result, the voltage vector output from the inverter 3 changes in the order shown in FIG. That is, in the period B, the zero vector V (1, 1, 1), the voltage vector V (1, 0, 1), the voltage vector V (1, 0, 0), the zero vector V (0, 0, 0), The voltage vector V (0, 1, 0) and the zero vector V (1, 1, 1) are output from the inverter 3 in this order.
 零ベクトルV(1,1,1)またはV(0,0,0)以外の電圧ベクトルが出力されているときには、その電圧ベクトルに応じた相電流がシャント電流IDCとして直流母線34に流れ、シャント電流検出器35により検出される。図9は、期間Bにおける電圧ベクトルと検出される相電流の関係を示した一覧表である。なお、図9の表では、図8と同様に、期間Bにおいて順に出力される零ベクトル以外の各電圧ベクトルを、それぞれ第1、第2、第3の電圧ベクトルとして表している。 When the voltage vector other than zero vectors V (1, 1, 1) or V (0,0,0) is output flows to the DC bus 34 phase current corresponding to the voltage vector as a shunt current I DC, It is detected by the shunt current detector 35. FIG. 9 is a list showing the relationship between the voltage vector in the period B and the detected phase current. In the table of FIG. 9, similarly to FIG. 8, the voltage vectors other than the zero vector output in order in the period B are represented as first, second, and third voltage vectors, respectively.
 図9の表から、第1の電圧ベクトルV(1,0,1)の出力時、および第3の電圧ベクトルV(0,1,0)の出力時には、V相の相電流Ivがシャント電流検出器35により検出されることが分かる。ただし、第1の電圧ベクトルV(1,0,1)と第3の電圧ベクトルV(0,1,0)とでは、各相のスイッチ状態を表す符号がそれぞれ反転しているため、相電流Ivが反対方向に流れる。なお、第2の電圧ベクトルV(1,0,0)の出力時には、U相の相電流Iuがシャント電流IDCとして流れるが、これはサンプルホールド回路12によるサンプリングの対象とはされない。 From the table of FIG. 9, when the first voltage vector V (1, 0, 1) is output and when the third voltage vector V (0, 1, 0) is output, the V-phase current Iv is the shunt current. It can be seen that it is detected by the detector 35. However, in the first voltage vector V (1, 0, 1) and the third voltage vector V (0, 1, 0), the signs representing the switch states of the respective phases are inverted, so that the phase current Iv flows in the opposite direction. At the time of output of the second voltage vector V (1, 0, 0), but the phase current Iu of the U phase flows as the shunt current I DC, which are not subject to sampling by the sample and hold circuit 12.
 以上説明したように、図7に示したような補正後の電圧指令Vu**、Vv**およびVw**に従って生成されるPWM波形により交流電動機4を駆動することで、期間Bにおいて、第1、第3の電圧ベクトルに応じて相電流-Iv、Ivを検出することができる。これにより、脈動成分を排除した相電流Ivの検出を行うことができる。 As described above, the AC motor 4 is driven by the PWM waveform generated according to the corrected voltage commands Vu **, Vv ** and Vw ** as shown in FIG. The phase currents -Iv and Iv can be detected according to the first and third voltage vectors. Thereby, it is possible to detect the phase current Iv excluding the pulsation component.
 さらに、図7に示すように、期間Bでは、最小相に対する補正後の電圧指令Vw**が常に零となるように補正している。これにより、三角波キャリアの零クロスのタイミング、すなわち最小値と最大値の中間値をとる時点で、インバータ3の直流母線34に流れるシャント電流IDCをそれぞれ検出すれば、第1の電圧ベクトルV(1,0,1)に対応する相電流-Ivと、第3の電圧ベクトルV(0,1,0)に対応する相電流Ivとを検出することができる。したがって、期間Bの前半にある三角波キャリアの上り区間と、期間Bの後半にある三角波キャリアの下り区間とで、反対符号の電圧ベクトルによる同一相の電流検出を、固定のタイミングで行うことができる。 Further, as shown in FIG. 7, in the period B, the corrected voltage command Vw ** for the minimum phase is always corrected to zero. Thus, the zero cross timing of the triangular wave carrier, i.e. at the time of taking an intermediate value between the minimum and maximum values, by detecting the shunt current I DC flowing through the DC bus 34 of the inverter 3, respectively, the first voltage vector V ( The phase current −Iv corresponding to (1, 0, 1) and the phase current Iv corresponding to the third voltage vector V (0, 1, 0) can be detected. Accordingly, in-phase current detection using voltage vectors of opposite signs can be performed at a fixed timing in the upstream section of the triangular wave carrier in the first half of period B and the downstream section of the triangular wave carrier in the second half of period B. .
 以上説明したように、期間A、Bにおいて、三角波キャリアが零(中間値)となる固定のタイミングでシャント電流IDCをそれぞれ検出することで、互いに異なる二相の相電流を検出することができる。したがって、サンプルホールド回路12では、三角波キャリアの半周期ごとに、固定のタイミングで直流電流信号IDCをサンプリングすればよい。そのため、安価なマイコンを用いて、サンプルホールド回路12を実現することが可能となる。 As described above, the period A, in B, by detecting the shunt current I DC respectively fixed timing the triangular wave carrier is zero (the middle value), it is possible to detect the phase currents of different two-phase . Therefore, the sample hold circuit 12 may sample the DC current signal IDC at a fixed timing every half cycle of the triangular wave carrier. Therefore, the sample hold circuit 12 can be realized by using an inexpensive microcomputer.
 上記の期間A、Bを繰り返し、サンプルホールド回路12においてシャント電流IDCに応じた直流電流信号IDCをサンプリングすることで、二相の相電流を検出することができる。こうして得られた相電流の検出結果に基づいて、電流再現器13により、各相の相電流の再現値Iuc、Ivc、Iwcを求めることができる。 The above periods A, Repeat B, the sample and hold circuit 12 to sample the DC current signal IDC in accordance with the shunt current I DC, it is possible to detect the phase current of two phases. Based on the detection result of the phase current obtained in this way, the current reproduction unit 13 can obtain the reproduction values Iuc, Ivc, and Iwc of the phase current of each phase.
 図11は、インバータ3から出力される各電圧ベクトルをαβ座標上に表した図である。符号の異なる電圧ベクトルは、αβ座標上で表記すると、図11に示すような関係になっている。 FIG. 11 is a diagram illustrating the voltage vectors output from the inverter 3 on the αβ coordinates. When voltage vectors having different signs are expressed on αβ coordinates, they have a relationship as shown in FIG.
 以上説明したように、制御装置2は、電圧補正部10およびPWM発生部11により、零ベクトルを除くいずれかの電圧ベクトルを第1の電圧ベクトルとして出力するようにインバータ3を制御する。そして、第1の電圧ベクトルを出力した後、零ベクトルを除いた、第1の電圧ベクトルとは異なるいずれかの電圧ベクトルを、第2の電圧ベクトルとして出力するようにインバータ3を制御する。さらに、第2の電圧ベクトルを出力した後、零ベクトルを出力するようにインバータ3を制御し、その後、第1の電圧ベクトルを反転した電圧ベクトルを第3の電圧ベクトルとして出力するようにインバータ3を制御する。このような制御を三角波キャリアの最初の周期に対応する期間Aにおいて行うと共に、次の周期に対応する期間Bでは、第1の電圧ベクトルおよび第3の電圧ベクトルとして、前回とは異なる種類の電圧ベクトルをそれぞれ出力するようにインバータ3を制御する。こうしたインバータ3の制御により交流電動機4の駆動を制御することで、サンプルホールド回路12における直流電流信号IDCのサンプリングを三角波キャリアの半周期ごとに固定のタイミングで行い、その結果を用いて、電流再現器13において三相の相電流の再現値Iuc、Ivc、Iwcを求めることができる。 As described above, the control device 2 controls the inverter 3 so that the voltage correction unit 10 and the PWM generation unit 11 output any voltage vector except the zero vector as the first voltage vector. Then, after outputting the first voltage vector, the inverter 3 is controlled so as to output any voltage vector different from the first voltage vector, excluding the zero vector, as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and then the inverter 3 is output so that a voltage vector obtained by inverting the first voltage vector is output as the third voltage vector. To control. Such control is performed in the period A corresponding to the first period of the triangular wave carrier, and in the period B corresponding to the next period, the first voltage vector and the third voltage vector are different types of voltages from the previous period. The inverter 3 is controlled so as to output each vector. By controlling the driving of the AC motor 4 by controlling the inverter 3, the sampling of the DC current signal IDC in the sample and hold circuit 12 is performed at a fixed timing every half cycle of the triangular wave carrier, and the result is used to reproduce the current. The reproduction value Iuc, Ivc, Iwc of the three-phase phase current can be obtained in the unit 13.
 なお、上記の説明において、零ベクトル以外で出力される各電圧ベクトルの内容は、補正前の元の電圧指令の大小関係に応じて変化する。したがって、期間A、Bにおいて第1、第2および第3の電圧ベクトルとしてそれぞれ出力される電圧ベクトルは、上記の例に限定されるものではない。 In the above description, the content of each voltage vector output other than the zero vector changes according to the magnitude relationship of the original voltage command before correction. Accordingly, the voltage vectors output as the first, second, and third voltage vectors in the periods A and B are not limited to the above example.
 次に、電圧補正部10の動作の詳細について説明する。図12は、電圧補正部10による電圧指令の補正の詳細を示す図である。 Next, details of the operation of the voltage correction unit 10 will be described. FIG. 12 is a diagram illustrating details of correction of the voltage command by the voltage correction unit 10.
 図7を用いて上記で説明したように、本実施形態の制御装置2では、三角波キャリアの連続する2周期、すなわち前述の期間Aおよび期間Bにおいて、インバータ3のシャント電流検出器35により2つの相電流を検出できるように、電圧補正部10において各相の電圧指令を補正する。図12では、この三角波キャリアの連続する2周期を半周期ごとに区切り、それぞれの期間をTc0、Tc1、Tc2、Tc3と表している。 As described above with reference to FIG. 7, in the control device 2 of the present embodiment, the two shunt current detectors 35 of the inverter 3 perform two periods in two consecutive periods of the triangular wave carrier, that is, the period A and the period B described above. The voltage correction unit 10 corrects the voltage command for each phase so that the phase current can be detected. In FIG. 12, two continuous cycles of the triangular wave carrier are divided into half cycles, and the respective periods are represented as Tc0, Tc1, Tc2, and Tc3.
 電圧補正部10は、三角波キャリアの連続する2周期のうち最初の周期(期間A)に対応する期間Tc0およびTc1では、各相の電圧指令同士の間隔を変えずに、中間相の電圧指令を常に0とする。一方、次の周期(期間B)に対応する期間Tc2およびTc3では、各相の電圧指令同士の間隔を変えずに、最小相の電圧指令を常に0とする。図12では、こうして決定された各相の電圧指令を、シフト前の最大相の電圧指令Vmax、中間相の電圧指令Vmidおよび最小相の電圧指令Vminとして示している。 In the periods Tc0 and Tc1 corresponding to the first period (period A) of the two consecutive periods of the triangular wave carrier, the voltage correction unit 10 outputs the intermediate phase voltage command without changing the interval between the voltage commands of each phase. Always 0. On the other hand, in the periods Tc2 and Tc3 corresponding to the next cycle (period B), the voltage command of the minimum phase is always set to 0 without changing the interval between the voltage commands of each phase. In FIG. 12, the voltage commands for the respective phases thus determined are shown as the maximum phase voltage command Vmax, the intermediate phase voltage command Vmid and the minimum phase voltage command Vmin before shifting.
 電圧補正部10は、以下の式(1)~(3)により、期間Tc0に対するシフト後の最大相、中間相および最小相の各電圧指令Vmax0、Vmid0およびVmin0を算出する。すなわち、最大相については、シフト前の最大相の電圧指令Vmaxと中間相の電圧指令Vmidの差分に基づいて、シフト前の電圧指令Vmaxからのシフト量を決定する。また、最小相については、シフト前の最小相の電圧指令Vminと中間相の電圧指令Vmidの差分および所定の必要シフト量Vshiftに基づいて、シフト前の電圧指令Vminからのシフト量を決定する。これらのシフト量に基づいて最大相および最小相の電圧指令をそれぞれシフトさせることで、シフト後の最大相の電圧指令Vmax0および最小相の電圧指令Vmin0を得ることができる。図12では、式(1)、(3)でそれぞれ表される最大相と最小相のシフト量を図中に矢印で示している。なお、式(2)は、中間相に対するシフト量が0であることを示している。 The voltage correction unit 10 calculates the voltage commands Vmax0, Vmid0, and Vmin0 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc0 by the following formulas (1) to (3). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax of the maximum phase before the shift and the voltage command Vmid of the intermediate phase. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the difference between the voltage command Vmin for the minimum phase before the shift and the voltage command Vmid for the intermediate phase and a predetermined required shift amount Vshift. By shifting the voltage command for the maximum phase and the minimum phase based on these shift amounts, the voltage command Vmax0 for the maximum phase and the voltage command Vmin0 for the minimum phase after the shift can be obtained. In FIG. 12, the shift amounts of the maximum phase and the minimum phase represented by the equations (1) and (3) are indicated by arrows in the drawing. Equation (2) indicates that the shift amount with respect to the intermediate phase is zero.
 最大相:Vmax0=2(Vmax-Vmid)        ・・・(1)
 中間相:Vmid0=Vmid=0              ・・・(2)
 最小相:Vmin0=2(Vmin-Vmid)-Vshift ・・・(3)
Maximum phase: Vmax0 = 2 (Vmax−Vmid) (1)
Intermediate phase: Vmid0 = Vmid = 0 (2)
Minimum phase: Vmin0 = 2 (Vmin−Vmid) −Vshift (3)
 式(3)におけるVshiftは、予め設定された必要シフト量を表している。この必要シフト量Vshiftは、電流検出に必要な電圧ベクトルの出力時間に基づいて設定することができる。たとえば、シャント電流IDCにおけるリンギングの大きさや、各スイッチング素子のデッドタイムの設定値などに基づいて、電圧ベクトルの変化からインバータ3の直流母線34に流れるシャント電流IDCを検出するまでに必要な時間を決定し、その時間に応じて必要シフト量Vshiftを決定することができる。あるいは、シャント電流IDCの波形を観測し、その観測結果に基づいて必要シフト量Vshiftを決定してもよい。式(3)から、図12に示すように、補正後の最小相の電圧指令Vmin0は、0から必要シフト量Vshiftを差し引いた値よりも小さいことが分かる。 Vshift in the equation (3) represents a necessary shift amount set in advance. This necessary shift amount Vshift can be set based on the output time of the voltage vector necessary for current detection. For example, required before the basis ringing and size of shunt current I DC, and the like set value of the dead time of the switching elements, to detect the shunt current I DC flowing from the change of the voltage vector to the DC bus 34 of the inverter 3 Time is determined, and the required shift amount Vshift can be determined according to the time. Alternatively, by observing the waveform of the shunt current I DC, it may determine the required shift amount Vshift based on the observation result. From equation (3), it can be seen that the corrected minimum phase voltage command Vmin0 is smaller than the value obtained by subtracting the required shift amount Vshift from 0, as shown in FIG.
 電圧補正部10は、以下の式(4)~(6)により、期間Tc1に対するシフト後の最大相、中間相および最小相の各電圧指令Vmax1、Vmid1およびVmin1を算出する。すなわち、最大相については、補正後の電圧指令が0となるようにシフト前の電圧指令Vmaxからのシフト量を決定する。また、最小相については、上記の必要シフト量Vshiftに基づいて、シフト前の電圧指令Vminからのシフト量を決定する。そして、これらのシフト量に基づいて最大相および最小相の電圧指令をそれぞれシフトさせることで、シフト後の最大相の電圧指令Vmax1および最小相の電圧指令Vmin1を得ることができる。図12では、式(4)、(6)でそれぞれ表される最大相と最小相のシフト量を図中に矢印で示している。なお、式(5)は、中間相に対するシフト量が0であることを示している。 The voltage correction unit 10 calculates the voltage commands Vmax1, Vmid1, and Vmin1 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc1 by the following equations (4) to (6). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the minimum phase, the shift amount from the voltage command Vmin before the shift is determined based on the necessary shift amount Vshift. The maximum phase voltage command Vmax1 and the minimum phase voltage command Vmin1 can be obtained by shifting the maximum phase and minimum phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the minimum phase represented by the equations (4) and (6) are indicated by arrows in the drawing. Equation (5) indicates that the shift amount with respect to the intermediate phase is zero.
 最大相:Vmax1=0                   ・・・(4)
 中間相:Vmid1=Vmid=0              ・・・(5)
 最小相:Vmin1=Vshift              ・・・(6)
Maximum phase: Vmax1 = 0 (4)
Intermediate phase: Vmid1 = Vmid = 0 (5)
Minimum phase: Vmin1 = Vshift (6)
 式(6)から、図12に示すように、補正後の最小相の電圧指令Vmin1は、必要シフト量Vshiftに等しいことが分かる。 From equation (6), it can be seen that the corrected minimum phase voltage command Vmin1 is equal to the required shift amount Vshift, as shown in FIG.
 電圧補正部10は、以下の式(7)~(9)により、期間Tc2に対するシフト後の最大相、中間相および最小相の各電圧指令Vmax2、Vmid2およびVmin2を算出する。すなわち、最大相については、シフト前の最大相の電圧指令Vmaxと最小相の電圧指令Vminの差分に基づいて、シフト前の電圧指令Vmaxからのシフト量を決定する。また、中間相については、上記の必要シフト量Vshiftに基づいて、シフト前の電圧指令Vmidからのシフト量を決定する。そして、これらのシフト量に基づいて最大相および中間相の電圧指令をそれぞれシフトさせることで、シフト後の最大相の電圧指令Vmax2および中間相の電圧指令Vmid2を得ることができる。図12では、式(7)、(8)でそれぞれ表される最大相と中間相のシフト量を図中に矢印で示している。なお、式(9)は、最小相に対するシフト量が0であることを示している。 The voltage correction unit 10 calculates the voltage commands Vmax2, Vmid2, and Vmin2 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc2 by the following formulas (7) to (9). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined based on the difference between the voltage command Vmax for the maximum phase before the shift and the voltage command Vmin for the minimum phase. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the necessary shift amount Vshift. Then, the maximum phase voltage command Vmax2 and the intermediate phase voltage command Vmid2 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the intermediate phase represented by the equations (7) and (8) are indicated by arrows in the drawing. Equation (9) indicates that the shift amount with respect to the minimum phase is zero.
 最大相:Vmax2=2(Vmax-Vmin)        ・・・(7)
 中間相:Vmid2=-Vshift             ・・・(8)
 最小相:Vmin2=Vmin=0              ・・・(9)
Maximum phase: Vmax2 = 2 (Vmax−Vmin) (7)
Intermediate phase: Vmid2 = −Vshift (8)
Minimum phase: Vmin2 = Vmin = 0 (9)
 式(8)から、図12に示すように、補正後の中間相の電圧指令Vmid2は、0から必要シフト量Vshiftを差し引いた値に等しいことが分かる。 From equation (8), as shown in FIG. 12, it can be seen that the corrected intermediate phase voltage command Vmid2 is equal to 0 minus the required shift amount Vshift.
 電圧補正部10は、以下の式(10)~(12)により、期間Tc3に対するシフト後の最大相、中間相および最小相の各電圧指令Vmax3、Vmid3およびVmin3を算出する。すなわち、最大相については、補正後の電圧指令が0となるようにシフト前の電圧指令Vmaxからのシフト量を決定する。また、中間相については、シフト前の中間相の電圧指令Vmidと最小相の電圧指令Vminの差分および上記の必要シフト量Vshiftに基づいて、シフト前の電圧指令Vmidからのシフト量を決定する。そして、これらのシフト量に基づいて最大相および中間相の電圧指令をそれぞれシフトさせることで、シフト後の最大相の電圧指令Vmax3および中間相の電圧指令Vmid3を得ることができる。図12では、式(10)、(11)でそれぞれ表される最大相と中間相のシフト量を図中に矢印で示している。なお、式(12)は、最小相に対するシフト量が0であることを示している。 The voltage correction unit 10 calculates the voltage commands Vmax3, Vmid3, and Vmin3 of the maximum phase, the intermediate phase, and the minimum phase after the shift with respect to the period Tc3 by the following equations (10) to (12). That is, for the maximum phase, the shift amount from the voltage command Vmax before the shift is determined so that the corrected voltage command becomes zero. For the intermediate phase, the shift amount from the voltage command Vmid before the shift is determined based on the difference between the voltage command Vmid for the intermediate phase before the shift and the voltage command Vmin for the minimum phase and the required shift amount Vshift. The maximum phase voltage command Vmax3 and the intermediate phase voltage command Vmid3 can be obtained by shifting the maximum phase and intermediate phase voltage commands based on these shift amounts. In FIG. 12, the shift amounts of the maximum phase and the intermediate phase represented by the expressions (10) and (11) are indicated by arrows in the drawing. Expression (12) indicates that the shift amount with respect to the minimum phase is zero.
 最大相:Vmax3=0                   ・・・(10)
 中間相:Vmid3=2(Vmid-Vmin)+Vshift ・・・(11)
 最小相:Vmin3=Vmin=0              ・・・(12)
Maximum phase: Vmax3 = 0 (10)
Intermediate phase: Vmid3 = 2 (Vmid−Vmin) + Vshift (11)
Minimum phase: Vmin3 = Vmin = 0 (12)
 式(10)から、図12に示すように、補正後の中間相の電圧指令Vmid3は、必要シフト量Vshiftよりも大きいことが分かる。 From equation (10), it can be seen that the corrected intermediate phase voltage command Vmid3 is larger than the required shift amount Vshift, as shown in FIG.
 本実施形態の制御装置2では、以上説明したような補正を電圧補正部10が行うことにより、サンプルホールド回路12において、Tc0~Tc3の各期間で符号の異なる最小相と中間相の電流検出を1回ずつ行うことができる。この電流検出の結果を基に、電流再現器13において、三相の相電流の再現値Iuc、Ivc、Iwcを求めることができる。なお、周知のように、PWM制御では、各相の線間電圧の関係が維持されている限り、各相の相電圧を変化させても、交流電動機4への基本波成分に影響はない。上記のTc0~Tc3の各期間では、これらの期間における線間電圧の平均値が、元の補正前の線間電圧と一致するように、電圧補正部10において各相の電圧指令を補正している。したがって、電圧指令の補正による交流電動機4の基本波への影響はない。 In the control device 2 of the present embodiment, the voltage correction unit 10 performs the correction as described above, whereby the sample and hold circuit 12 detects the currents of the minimum phase and the intermediate phase having different signs in each period of Tc0 to Tc3. It can be done once. Based on the result of this current detection, the current reproducer 13 can obtain the reproduced values Iuc, Ivc, and Iwc of the three-phase currents. As is well known, in PWM control, as long as the relationship of the line voltage of each phase is maintained, changing the phase voltage of each phase does not affect the fundamental wave component to AC motor 4. In each period of Tc0 to Tc3, the voltage correction unit 10 corrects the voltage command of each phase so that the average value of the line voltage in these periods matches the original line voltage before correction. Yes. Therefore, there is no influence on the fundamental wave of the AC motor 4 due to the correction of the voltage command.
 以上説明した本発明の第1の実施形態によれば、以下の作用効果を奏する。 According to the first embodiment of the present invention described above, the following operational effects are obtained.
(1)交流電動機4の制御装置2は、三相の交流電動機4と接続されたインバータ3に接続され、インバータ3において正極側と負極側にそれぞれ三相ずつ設けられたスイッチング素子Sup、Svp、Swp、Sun、SvnおよびSwnのスイッチング状態に応じて交流電動機4の駆動を制御する。インバータ3は、上記の各スイッチング素子のスイッチング状態に応じて、8通りの電圧ベクトルを交流電動機4に出力可能である。この電圧ベクトルは、正極側の全てのスイッチング素子Sup、Svp、Swpがオンであり、負極側の全てのスイッチング素子Sun、Svn、Swnがオフであるスイッチング状態と、正極側の全てのスイッチング素子Sup、Svp、Swpがオフであり、負極側の全てのスイッチング素子Sun、Svn、Swnがオンであるスイッチング状態とにそれぞれ対応する2種類の零ベクトルV(0,0,0)およびV(1,1,1)を含む。交流電動機4の制御装置2は、零ベクトルを除くいずれかの電圧ベクトルを第1の電圧ベクトルとして出力するようにインバータ3を制御し、第1の電圧ベクトルを出力した後、零ベクトルを除いた第1の電圧ベクトルとは異なるいずれかの電圧ベクトルを、第2の電圧ベクトルとして出力するようにインバータ3を制御する。さらに、第2の電圧ベクトルを出力した後、零ベクトルを出力するようにインバータ3を制御し、この零ベクトルを出力した後、第1の電圧ベクトルを反転した電圧ベクトルを第3の電圧ベクトルとして出力するようにインバータ3を制御する。これにより、交流電動機4の駆動を制御する。このようにしたので、安価なマイコンを用いて高精度に相電流を検出可能な交流電動機の制御装置を実現することができる。 (1) The control device 2 of the AC motor 4 is connected to an inverter 3 connected to the three-phase AC motor 4, and in the inverter 3, switching elements Sup, Svp, provided with three phases respectively on the positive electrode side and the negative electrode side, The drive of the AC motor 4 is controlled according to the switching states of Swp, Sun, Svn and Swn. The inverter 3 can output eight voltage vectors to the AC motor 4 in accordance with the switching state of each switching element. This voltage vector includes the switching state in which all the switching elements Sup, Svp, Swp on the positive side are on and all the switching elements Sun, Svn, Swn on the negative side are off, and all the switching elements Sup on the positive side. , Svp, Swp are off, and two types of zero vectors V (0,0,0) and V (1, corresponding to the switching states where all the switching elements Sun, Svn, Swn on the negative side are on, respectively. 1, 1). The control device 2 of the AC motor 4 controls the inverter 3 so as to output any voltage vector except the zero vector as the first voltage vector, and after outputting the first voltage vector, the zero vector is removed. The inverter 3 is controlled so as to output any voltage vector different from the first voltage vector as the second voltage vector. Further, after outputting the second voltage vector, the inverter 3 is controlled to output the zero vector, and after outputting the zero vector, a voltage vector obtained by inverting the first voltage vector is used as the third voltage vector. The inverter 3 is controlled to output. Thereby, the drive of the AC motor 4 is controlled. Since it did in this way, the control apparatus of the alternating current motor which can detect a phase current with high precision using an inexpensive microcomputer is realizable.
(2)制御装置2は、上記の第1の電圧ベクトル、第2の電圧ベクトル、零ベクトルおよび第3の電圧ベクトルを繰り返して順次出力するようにインバータ3を制御する。このとき、第1の電圧ベクトルおよび第3の電圧ベクトルとして、前回とは異なる種類の電圧ベクトルをそれぞれ出力するようにインバータ3を制御する。このようにしたので、互いに異なる二相の相電流を検出し、これに基づいて各相の相電流を求めることができる。 (2) The control device 2 controls the inverter 3 so as to repeatedly output the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector in order. At this time, the inverter 3 is controlled so that different types of voltage vectors from the previous time are output as the first voltage vector and the third voltage vector, respectively. Since it did in this way, the phase current of two phases which are different from each other can be detected, and the phase current of each phase can be obtained based on this.
(3)制御装置2は、電流推定部13により、第1の電圧ベクトルおよび第3の電圧ベクトルの出力期間中に、インバータ3の直流母線34に流れる直流シャント電流IDCの検出結果に基づいて、交流電動機4の相電流を求める。このようにしたので、電流推定部13において、各相の相電流を求め、その再現値Iuc、Ivc、Iwcを求めることができる。 (3) The control device 2 uses the current estimation unit 13 based on the detection result of the DC shunt current I DC flowing in the DC bus 34 of the inverter 3 during the output period of the first voltage vector and the third voltage vector. Then, the phase current of the AC motor 4 is obtained. Since it did in this way, in the electric current estimation part 13, the phase current of each phase can be calculated | required and the reproduction values Iuc, Ivc, and Iwc can be calculated | required.
(4)制御装置2は、三相の電圧指令Vu*、Vv*およびVw*を発生する電圧指令発生部20と、これらの電圧指令を補正する電圧補正部10と、PWM発生部11とを備える。PWM発生部11は、電圧補正部10により補正された補正後の電圧指令Vu**、Vv**およびVw**と、所定周期の三角波キャリアとを比較し、その比較結果に基づいて三相のPWM波形を発生する。こうしてPWM発生部11により発生されたPWM波形をインバータ3に出力することで、制御装置2は、スイッチング素子Sup、Svp、Swp、Sun、SvnおよびSwnのスイッチング状態を制御して、交流電動機4の駆動を制御する。このようにしたので、補正後の電圧指令に応じて、交流電動機4の駆動を確実に制御することができる。 (4) The control device 2 includes a voltage command generator 20 that generates three-phase voltage commands Vu *, Vv *, and Vw *, a voltage corrector 10 that corrects these voltage commands, and a PWM generator 11. Prepare. The PWM generator 11 compares the corrected voltage commands Vu **, Vv **, and Vw ** corrected by the voltage correction unit 10 with a triangular wave carrier having a predetermined period, and based on the comparison result, the three-phase PWM waveform is generated. By outputting the PWM waveform generated by the PWM generator 11 to the inverter 3 in this way, the control device 2 controls the switching state of the switching elements Sup, Svp, Swp, Sun, Svn and Swn, and the AC motor 4 Control the drive. Since it did in this way, according to the voltage command after correction | amendment, the drive of the alternating current motor 4 can be controlled reliably.
(5)電圧補正部10は、三角波キャリアの連続する2周期のうち最初の周期では、三相の電圧指令で中間の電圧指令Vmidを0とし、さらに三相の電圧指令で最大の電圧指令Vmaxおよび最小の電圧指令Vminをシフトさせることにより、各相の電圧指令を補正する。また、最初の周期に続く次の周期では、最小の電圧指令Vminを0とし、さらに最大の電圧指令Vmaxおよび中間の電圧指令Vmidをシフトさせることにより、各相の電圧指令を補正する。このようにしたので、上記の第1、第2および第3の電圧ベクトルを順次出力すると共に、これらの電圧ベクトルを三角波キャリアの周期ごとに変化させることができる。 (5) The voltage correction unit 10 sets the intermediate voltage command Vmid to 0 in the three-phase voltage command and the maximum voltage command Vmax in the three-phase voltage command in the first cycle out of the two consecutive cycles of the triangular wave carrier. And the voltage command of each phase is corrected by shifting the minimum voltage command Vmin. In the next cycle following the first cycle, the minimum voltage command Vmin is set to 0, and the maximum voltage command Vmax and the intermediate voltage command Vmid are shifted to correct the voltage command of each phase. Since it did in this way, while said 1st, 2nd and 3rd voltage vector is output sequentially, these voltage vectors can be changed for every period of a triangular wave carrier.
(6)電圧補正部10は、上記の最初の周期の前半期間Tc0では、前述の式(1)、(3)に従って、最大の電圧指令Vmaxと中間の電圧指令Vmidの差分に基づいて、最大の電圧指令Vmaxに対するシフト量を決定すると共に、最小の電圧指令Vminと中間の電圧指令Vmidの差分および所定の必要シフト量Vshiftに基づいて、最小の電圧指令Vminに対するシフト量を決定する。また、最初の周期の後半期間Tc1では、前述の式(4)、(6)に従って、補正後の電圧指令が0となるように最大の電圧指令Vmaxに対するシフト量を決定すると共に、必要シフト量Vshiftに基づいて最小の電圧指令Vminに対するシフト量を決定する。一方、上記の次の周期の前半期間では、前述の式(7)、(8)に従って、最大の電圧指令Vmaxと最小の電圧指令Vminの差分に基づいて最大の電圧指令Vmaxに対するシフト量を決定すると共に、必要シフト量Vshiftに基づいて中間の電圧指令Vmidに対するシフト量を決定する。また、次の周期の後半期間では、前述の式(10)、(11)に従って、補正後の電圧指令が0となるように最大の電圧指令Vmaxに対するシフト量を決定すると共に、中間の電圧指令Vmidと最小の電圧指令Vminの差分および必要シフト量Vshiftに基づいて、中間の電圧指令Vmidに対するシフト量を決定する。このようにしたので、上記のような各相の電圧指令の補正を適切かつ確実に行うことができる。 (6) In the first half period Tc0 of the first cycle, the voltage correction unit 10 determines the maximum value based on the difference between the maximum voltage command Vmax and the intermediate voltage command Vmid according to the above-described equations (1) and (3). The shift amount with respect to the minimum voltage command Vmin is determined based on the difference between the minimum voltage command Vmin and the intermediate voltage command Vmid and the predetermined required shift amount Vshift. Further, in the second half period Tc1 of the first cycle, the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (4) and (6), and the necessary shift amount. A shift amount with respect to the minimum voltage command Vmin is determined based on Vshift. On the other hand, in the first half period of the next cycle, the shift amount with respect to the maximum voltage command Vmax is determined based on the difference between the maximum voltage command Vmax and the minimum voltage command Vmin according to the above-described equations (7) and (8). At the same time, the shift amount for the intermediate voltage command Vmid is determined based on the necessary shift amount Vshift. In the second half of the next cycle, the shift amount with respect to the maximum voltage command Vmax is determined so that the corrected voltage command becomes 0 according to the above-described equations (10) and (11), and the intermediate voltage command Based on the difference between Vmid and the minimum voltage command Vmin and the required shift amount Vshift, the shift amount for the intermediate voltage command Vmid is determined. Since it did in this way, correction | amendment of the voltage command of each phase as mentioned above can be performed appropriately and reliably.
(7)上記の必要シフト量Vshiftは、各スイッチング素子のスイッチング状態が変化してからインバータ3の直流母線34に流れる直流シャント電流IDCを検出するまでの時間に応じて設定することができる。このようにすれば、相電流に応じたシャント電流IDCを正確に検出することができる。 (7) above needs shift Vshift can be set according to the time to detect a DC shunt current I DC flowing from the switching state is changed to the DC bus 34 of the inverter 3 of each switching element. In this way, it is possible to accurately detect the shunt current I DC corresponding to the phase current.
(8)制御装置2は、サンプルホールド回路12により、三角波キャリアが最小値と最大値の中間値をとる時点でインバータ3の直流母線34に流れる直流シャント電流IDCの検出結果を取得する。このようにしたので、異なる2つの相の相電流に応じたシャント電流IDCを一定のタイミングごとに取得することができる。 (8) The control device 2, the sample hold circuit 12, obtains the detection result of the DC shunt current I DC to when the triangular wave carrier takes an intermediate value between the minimum and maximum values flowing through the DC bus 34 of the inverter 3. Since this is done, it is possible to obtain the shunt current I DC corresponding to the phase currents of two different phases for each fixed timing.
(第2の実施形態)
 次に本発明の第2の実施形態について説明する。図13は、本発明の第2の実施形態に係る制御装置2Bの構成を示す図である。
(Second Embodiment)
Next, a second embodiment of the present invention will be described. FIG. 13 is a diagram showing a configuration of a control device 2B according to the second embodiment of the present invention.
 前述の第1の実施形態では、図12に示したように、正弦波状の電流で交流電動機4を駆動するセンサレス駆動制御に適用した例を示した。こうしたセンサレス駆動制御は様々な用途に展開されているが、低速域における性能劣化が問題となっている。その解決策として、たとえば、日本国特開2009-189176号公報に開示されているような、120度通電駆動をベースにした手法が周知である。この手法では、三相のスイッチング素子のうち二相のスイッチング素子のみスイッチング状態を切り替えることで120度通電駆動を行うため、安価なマイコンでの実現が可能である。その反面、120度通電駆動をベースにしているため、トルク脈動が大きく、また電流に含まれる高調波成分が多いため、損失が大きいという問題がある。 In the first embodiment described above, as shown in FIG. 12, an example is shown in which the present invention is applied to sensorless drive control in which the AC motor 4 is driven by a sinusoidal current. Such sensorless drive control has been developed for various uses, but performance degradation at low speeds is a problem. As a solution to this problem, for example, a method based on 120-degree energization driving as disclosed in Japanese Patent Application Laid-Open No. 2009-189176 is well known. In this method, 120-degree energization driving is performed by switching the switching state of only the two-phase switching elements among the three-phase switching elements, so that it can be realized with an inexpensive microcomputer. On the other hand, since it is based on 120-degree conduction drive, there is a problem that torque pulsation is large and there are many harmonic components included in the current, so that loss is large.
 そこで、本実施形態では、交流電動機が低速駆動している低速域では、上記のような手法を利用して、三相の各スイッチング素子のうち二相のスイッチング素子のみスイッチング状態を切り替えることで、120度通電駆動を行うようにインバータ3を制御する。また、それ以外の中高速域では、第1の実施形態で説明したようなセンサレス駆動制御により、三相の各スイッチング素子のスイッチング状態を切り替えて、三相通電駆動を行うようにインバータ3を制御する。これにより、両者の欠点を補うような構成とする。なお、本発明の正弦波センサレス駆動方法は、前述のように安価なマイコンに適した方式であるため、もともと安価なマイコンで実現できる120度通電駆動との組み合わせに非常に適している。 Therefore, in this embodiment, by switching the switching state of only the two-phase switching elements among the three-phase switching elements in the low-speed range where the AC motor is driven at a low speed, The inverter 3 is controlled to perform 120-degree energization driving. Further, in other medium and high speed ranges, the inverter 3 is controlled to perform the three-phase energization drive by switching the switching state of each of the three-phase switching elements by the sensorless drive control as described in the first embodiment. To do. Thereby, it is set as the structure which compensates both faults. Since the sine wave sensorless driving method of the present invention is a method suitable for an inexpensive microcomputer as described above, it is very suitable for a combination with 120-degree energization driving that can be realized originally by an inexpensive microcomputer.
 図13に示すように、本実施形態による制御装置2Bでは、図1に示した第1の実施形態による制御装置2の構成に、低速域のセンサレス駆動制御が可能な120度通電駆動を実現するための構成を加えて、これらを切り替えられるようにしている。本制御装置2Bを、図1の制御装置2の代わりに用いることで、本発明の第2の実施形態による交流電動機駆動システムが実現できる。 As shown in FIG. 13, in the control device 2B according to the present embodiment, 120-degree energization drive capable of sensorless drive control in the low speed region is realized in the configuration of the control device 2 according to the first embodiment shown in FIG. The structure for this is added, and these can be switched now. By using this control device 2B instead of the control device 2 of FIG. 1, an AC motor drive system according to the second embodiment of the present invention can be realized.
 図13において、図1と同じ構成には、同一の符号を付している。この図13に示した制御装置2Bには、図1の構成に加えて、120度通電部17と、120度通電駆動向けのPWMを行う120度PWM部18と、120度通電駆動と三相通電駆動を切り替えるための切替スイッチ19a~19eからなる。また、図1のd軸電流制御器7および位置・速度推定部15に替えて、d軸電流制御器7Bおよび位置・速度推定部15Bを有している。 In FIG. 13, the same components as those in FIG. 1 are denoted by the same reference numerals. In addition to the configuration of FIG. 1, the control device 2B shown in FIG. 13 includes a 120-degree energization unit 17, a 120-degree PWM unit 18 that performs PWM for 120-degree energization drive, a 120-degree energization drive, and a three-phase It comprises changeover switches 19a to 19e for switching energization drive. Further, in place of the d-axis current controller 7 and the position / speed estimation unit 15 of FIG. 1, a d-axis current controller 7B and a position / speed estimation unit 15B are provided.
 120度通電部17は、120度通電駆動を行うための位相値θdcBと、120度通電駆動を行った結果から演算により求められた速度推定値ω1cBとを出力する。位相値θdcBは、60度毎の離散的な値である。この位相値θdcBと、位置・速度推定部15Bから出力される連続的な値である位相値θdcSとを、切替スイッチ19cにて切替え、dq逆変換器9に出力する。なお、各切替スイッチ19a~19eは、120度通電駆動時には「0」側に切り替えられ、三相通電駆動時には「1」側に切り替えられる。 The 120-degree energization unit 17 outputs a phase value θdcB for performing 120-degree energization driving and a speed estimation value ω1cB obtained by calculation from the result of performing 120-degree energization driving. The phase value θdcB is a discrete value every 60 degrees. The phase value θdcB and the phase value θdcS which is a continuous value output from the position / velocity estimation unit 15B are switched by the changeover switch 19c and output to the dq inverse converter 9. Each of the change-over switches 19a to 19e is switched to the “0” side during 120-degree energization driving, and is switched to the “1” side during three-phase energization driving.
 切替スイッチ19eは、120度通電部17から出力される速度推定値ω1cBと、位置・速度推定部15Bから出力される速度推定値ω1cSとを切り替え、いずれか一方を交流電動機4の電気角速度ω1cとして速度変換部16に出力する。 The changeover switch 19e switches between the estimated speed value ω1cB output from the 120-degree energization unit 17 and the estimated speed value ω1cS output from the position / speed estimation unit 15B, and either one is used as the electrical angular velocity ω1c of the AC motor 4. Output to the speed converter 16.
 切替スイッチ19aは、PWM発生部11から出力される三相通電駆動用のPWM波形と、120度PWM部18から出力される120度通電駆動用のPWM波形とを切り替え、いずれか一方をインバータ3に出力する。 The changeover switch 19a switches between a PWM waveform for three-phase energization driving output from the PWM generator 11 and a PWM waveform for 120-degree energization driving output from the 120-degree PWM unit 18, and either one is switched to the inverter 3 Output to.
 切替スイッチ19dは、q軸電流制御器8へのフィードバック電流として、dq変換器14により座標変換されたq軸電流Iqと、サンプルホールド回路12によりサンプリングされたシャント電流IDCの検出結果とを切り替える。なお、120度通電駆動時には、電流制御は1つでよいので、シャント電流IDCの検出値をそのままフィードバックすればよい。 Changeover switch 19d is switched as a feedback current to the q-axis current controller 8, a q-axis current Iq which are coordinate transformation by dq converter 14, the sample-and-hold circuit 12 and a detection result of shunt current I DC sampled . At the time of 120-degree conduction drive, the current control may be one, or may be directly fed back to the detection value of the shunt current I DC.
 切替スイッチ19bは、0と、d軸電流制御器7Bから出力される値とを切り替え、いずれか一方をd軸電圧指令Vd*としてdq逆変換器9に出力する。120度通電駆動時には、「d軸」という概念がない。そのため、d軸電圧指令Vd*が0になるように、切替スイッチ19bにて切り替えている。なお、q軸電流制御器8から出力されるq軸電圧指令Vq*は、120度通電駆動時および三相通電駆動時において共通に用いられる。 The changeover switch 19b switches between 0 and the value output from the d-axis current controller 7B, and outputs either one to the dq inverse converter 9 as the d-axis voltage command Vd *. There is no concept of “d-axis” during 120-degree energization drive. Therefore, the changeover switch 19b switches so that the d-axis voltage command Vd * becomes zero. The q-axis voltage command Vq * output from the q-axis current controller 8 is commonly used during 120-degree energization driving and three-phase energization driving.
 d軸電流制御器7Bは、d軸電流Idのフィードバック制御は行わずに、三相通電駆動において用いられるd軸電圧指令Vd*をq軸電流指令Iq*から直接演算して求めている。ここで、120度通電駆動と三相通電駆動の切替を行う場合、d軸電流制御器7Bにおける値の処理が問題となる。たとえば、d軸電圧指令Vd*をPI制御(比例積分制御)により求める場合、積分器の初期値設定に関する問題や、120度通電駆動時の動作に関する問題が生じる。そこで、本実施形態では、d軸電流制御器7Bにおいて、q軸電流指令Iq*に基づくフィードフォワード演算によりd軸電圧指令Vd*を決定するようにして、フィードバック制御を行わないようにした。この結果、120度通電駆動と三相通電駆動の切替をスムーズに行うことができる。なお、q軸電流指令Iq*に替えて、電流推定部21により推定されたq軸電流Iqに基づくフィードフォワード演算により、d軸電圧指令Vd*を決定してもよい。 The d-axis current controller 7B calculates the d-axis voltage command Vd * used in the three-phase energization drive directly from the q-axis current command Iq * without performing feedback control of the d-axis current Id. Here, when switching between 120-degree energization drive and three-phase energization drive, processing of values in the d-axis current controller 7B becomes a problem. For example, when the d-axis voltage command Vd * is obtained by PI control (proportional integral control), there are problems relating to the initial value setting of the integrator and problems relating to the operation during 120-degree conduction drive. Therefore, in the present embodiment, the d-axis current controller 7B determines the d-axis voltage command Vd * by feedforward calculation based on the q-axis current command Iq *, and does not perform feedback control. As a result, switching between the 120-degree energization drive and the three-phase energization drive can be performed smoothly. Instead of the q-axis current command Iq *, the d-axis voltage command Vd * may be determined by a feedforward calculation based on the q-axis current Iq estimated by the current estimation unit 21.
 d軸電流制御器7Bは、一次遅れフィルタ71、乗算器72、q軸インダクタンス設定器73および符号反転器74からなる。これらの構成によって実現される演算処理は、交流電動機4のd軸電圧に対するフィードフォワード演算となるため、三相通電駆動を安定に行うことができる。なお、一次遅れフィルタ71の時定数Trは、たとえばq軸電流制御器8の応答設定値に応じて設定しておけばよい。 The d-axis current controller 7B includes a first-order lag filter 71, a multiplier 72, a q-axis inductance setting unit 73, and a sign inverter 74. Since the arithmetic processing realized by these configurations is a feedforward calculation for the d-axis voltage of the AC motor 4, three-phase energization driving can be performed stably. Note that the time constant Tr of the first-order lag filter 71 may be set according to the response set value of the q-axis current controller 8, for example.
 位置・速度推定部15Bは、軸誤差推定器151、零設定器152、PI制御器153、減算器6cおよび位相演算器154からなる。このような位置・速度推定部15Bの構成は、中高速センサレスの位置・速度推定器として周知である。 The position / velocity estimation unit 15B includes an axis error estimator 151, a zero setter 152, a PI controller 153, a subtractor 6c, and a phase calculator 154. Such a configuration of the position / speed estimation unit 15B is well known as a medium / high speed sensorless position / speed estimator.
 軸誤差推定器151は、永久磁石モータである交流電動機4の実際の回転子位置と、制御装置2Bで演算された回転子位置との偏差を推定演算し、その結果を軸誤差Δθdcとして出力する。 The axis error estimator 151 estimates and calculates the deviation between the actual rotor position of the AC motor 4 that is a permanent magnet motor and the rotor position calculated by the control device 2B, and outputs the result as an axis error Δθdc. .
 減算器6cは、零設定器152から出力される「0」と、軸誤差推定器151からの軸誤差Δθdcとの差分を求め、その演算結果をPI制御器153に出力する。 The subtractor 6 c calculates a difference between “0” output from the zero setting unit 152 and the axis error Δθdc from the axis error estimator 151, and outputs the calculation result to the PI controller 153.
 PI制御器153は、減算器6cからの差分が0となるように速度推定値ω1cSを演算することで、軸誤差Δθdcの値を0とするための制御を行う。このPI制御器153により、PLL(Phase Locked Loop)が構成される。なお、PI制御器153により求められた速度推定値ω1cSは、交流電動機4の電気角速度に相当するものである。 The PI controller 153 performs control for setting the value of the axis error Δθdc to 0 by calculating the speed estimated value ω1cS so that the difference from the subtractor 6c becomes 0. This PI controller 153 constitutes a PLL (Phase Locked Loop). The estimated speed value ω1cS obtained by the PI controller 153 corresponds to the electrical angular velocity of the AC motor 4.
 位相演算器154は、PI制御器153からの速度推定値ω1cSを積分処理することで、位相値θdcSを求める。この位相値θdcSは、位置・速度推定部15Bから切替スイッチ19cに出力される。 The phase calculator 154 integrates the estimated speed value ω1cS from the PI controller 153 to obtain the phase value θdcS. The phase value θdcS is output from the position / velocity estimation unit 15B to the changeover switch 19c.
 以上説明した本発明の第2の実施形態によれば、前述の第1の実施形態による(1)~(8)の各作用効果に加えて、さらに以下の(9)、(10)の作用効果を奏する。 According to the second embodiment of the present invention described above, in addition to the functions and effects (1) to (8) according to the first embodiment, the following functions (9) and (10) are further provided. There is an effect.
(9)制御装置2Bは、交流電動機4が低速駆動している低速域では、インバータ3のスイッチング素子のうち、二相のスイッチング素子のみスイッチング状態を切り替えて120度通電駆動を行うように、インバータ3を制御する。また、低速域以外の速度域では、三相のスイッチング素子のスイッチング状態を切り替えて三相通電駆動を行うように、インバータ3を制御する。このようにしたので、低速域から高速域までのあらゆる速度域において、高性能なセンサレス駆動を実現することができる。 (9) In the low speed region where the AC motor 4 is driven at a low speed, the control device 2B switches the switching state of only the two-phase switching elements among the switching elements of the inverter 3 so as to perform 120-degree energization driving. 3 is controlled. Further, in the speed range other than the low speed range, the inverter 3 is controlled so that the switching state of the three-phase switching element is switched to perform the three-phase energization drive. Since it did in this way, a high-performance sensorless drive is realizable in all the speed ranges from a low speed range to a high speed range.
(10)制御装置2Bは、三相通電駆動において用いられるd軸電圧指令Vd*を出力するd軸電流制御器7Bと、120度通電駆動および三相通電駆動において共通に用いられるq軸電圧指令Vq*を出力するq軸電流制御器8とを備える。d軸電流制御器7Bは、外部から入力されるq軸電流指令Iq*または交流電動機4に流れる電流の検出結果であるq軸電流Iqに基づくフィードフォワード演算により、d軸電圧指令Vd*を出力する。このようにしたので、120度通電駆動と三相通電駆動の切替をスムーズに行うことができる。 (10) The control device 2B includes a d-axis current controller 7B that outputs a d-axis voltage command Vd * used in three-phase current drive, and a q-axis voltage command commonly used in 120-degree current drive and three-phase current drive. A q-axis current controller 8 for outputting Vq *. The d-axis current controller 7B outputs a d-axis voltage command Vd * by a feedforward calculation based on a q-axis current command Iq * input from the outside or a q-axis current Iq that is a detection result of a current flowing in the AC motor 4. To do. Since it did in this way, switching with 120 degree energization drive and three phase energization drive can be performed smoothly.
 なお、本実施形態では、上記のd軸電流制御器7Bを用いずに、第1の実施形態で説明したようなd軸電流制御器7を用いたフィードバック演算により、d軸電圧指令Vd*を出力することもできる。ただし、本発明の特徴である、低機能マイコンによる処理の実現という観点からは、図13のような構成の方がより望ましい。 In the present embodiment, the d-axis voltage command Vd * is obtained by feedback calculation using the d-axis current controller 7 as described in the first embodiment without using the d-axis current controller 7B. It can also be output. However, the configuration as shown in FIG. 13 is more desirable from the viewpoint of realizing processing by a low-function microcomputer, which is a feature of the present invention.
(第3の実施形態)
 次に本発明の第3の実施形態について説明する。図14は、本発明の第3の実施形態に係る交流電動機駆動システム40の構成を示す図である。
(Third embodiment)
Next, a third embodiment of the present invention will be described. FIG. 14 is a diagram showing a configuration of an AC motor drive system 40 according to the third embodiment of the present invention.
 図14に示す交流電動機駆動システム40は、第1の実施形態で説明したq軸電流指令発生器1、制御装置2、インバータ3および交流電動機4を一体化して構成されている。q軸電流指令発生器1、制御装置2およびインバータ3は、1つの回路基板上に実装された駆動制御基板41として、交流電動機4の内部にパッケージされている。 The AC motor drive system 40 shown in FIG. 14 is configured by integrating the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 described in the first embodiment. The q-axis current command generator 1, the control device 2, and the inverter 3 are packaged inside the AC motor 4 as a drive control board 41 mounted on one circuit board.
 一体化された交流電動機駆動システム40への配線は、たとえば図14に示す電源線および通信線のみとすることができる。電源線は、インバータ3に接続され、スイッチング素子への直流電源等を供給するために用いられる。通信線は、q軸電流指令発生器1や制御装置2に接続され、q軸電流指令発生器1に入力される回転速度指令の送信や、制御装置2から出力される動作状態の送信などに用いられる。 The wiring to the integrated AC motor drive system 40 can be, for example, only the power line and the communication line shown in FIG. The power supply line is connected to the inverter 3 and is used to supply a DC power supply or the like to the switching element. The communication line is connected to the q-axis current command generator 1 and the control device 2 to transmit a rotational speed command input to the q-axis current command generator 1 and to transmit an operation state output from the control device 2. Used.
 以上説明した本発明の第3の実施形態によれば、交流電動機駆動システム40として、q軸電流指令発生器1、制御装置2、インバータ3および交流電動機4が全て一体化されているため、小型化を実現できる。また、各構成間の配線を無くすことができるため、取扱性の向上を図ることができる。 According to the third embodiment of the present invention described above, since the q-axis current command generator 1, the control device 2, the inverter 3, and the AC motor 4 are all integrated as the AC motor drive system 40, a small size is achieved. Can be realized. In addition, since the wiring between the respective components can be eliminated, the handling property can be improved.
 なお、交流電動機駆動システム40において、制御装置2に替えて制御装置2Bを用いることで、前述の第2の実施形態で説明したような構成を適用してもよい。 In the AC motor drive system 40, the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2.
(第4の実施形態)
 次に本発明の第4の実施形態について説明する。図15は、本発明の第4の実施形態に係る油圧制御システムの構成を示す図である。
(Fourth embodiment)
Next, a fourth embodiment of the present invention will be described. FIG. 15 is a diagram showing a configuration of a hydraulic control system according to the fourth embodiment of the present invention.
 図15に示す油圧制御システムは、たとえば自動車内部のトランスミッションの油圧制御や、ブレーキの油圧制御などに用いられるものである。この油圧制御システムは、油圧制御ユニット25および油圧回路50により構成されている。 The hydraulic control system shown in FIG. 15 is used for, for example, hydraulic control of a transmission inside a vehicle, hydraulic control of a brake, and the like. This hydraulic control system includes a hydraulic control unit 25 and a hydraulic circuit 50.
 油圧制御ユニット25は、第1の実施形態で説明したq軸電流指令発生器1、制御装置2、インバータ3および交流電動機4と、交流電動機4に取り付けられたオイルポンプ26により構成されている。オイルポンプ26は、交流電動機4の駆動に応じて動作し、油圧回路50の油圧を制御する。 The hydraulic control unit 25 includes the q-axis current command generator 1, the control device 2, the inverter 3 and the AC motor 4 described in the first embodiment, and an oil pump 26 attached to the AC motor 4. The oil pump 26 operates according to the driving of the AC motor 4 and controls the hydraulic pressure of the hydraulic circuit 50.
 油圧回路50は、作動油を貯蔵するタンク51と、油圧を設定値以下に保つリリーフバルブ52と、作動油の吐出先を切り替えるソレノイドバルブ53と、油圧アクチュエータとして動作するシリンダ54で構成される。 The hydraulic circuit 50 includes a tank 51 that stores hydraulic oil, a relief valve 52 that keeps the hydraulic pressure below a set value, a solenoid valve 53 that switches a discharge destination of the hydraulic oil, and a cylinder 54 that operates as a hydraulic actuator.
 オイルポンプ26は、交流電動機4の駆動に応じて動作することで油圧を生成し、油圧アクチュエータであるシリンダ54を駆動する。このとき、油圧回路50において、ソレノイドバルブ53により作動油の吐出先が切り替えられると、オイルポンプ26の負荷が変化し、油圧制御ユニット25に対する負荷外乱が発生する。その結果、定常状態の圧力に比べて数倍以上の負荷が生じることで、交流電動機4が停止してしまうことがある。このように交流電動機4が停止しても、本実施形態による油圧駆動システムでは、第1の実施形態で説明したような方法で交流電動機4の駆動制御を行うことにより、停止状態での回転子位置を推定可能である。したがって、何ら問題が生じない。 The oil pump 26 operates according to the driving of the AC motor 4 to generate hydraulic pressure, and drives the cylinder 54 that is a hydraulic actuator. At this time, when the hydraulic oil discharge destination is switched by the solenoid valve 53 in the hydraulic circuit 50, the load of the oil pump 26 changes, and a load disturbance to the hydraulic control unit 25 occurs. As a result, the AC motor 4 may stop due to a load that is several times greater than the steady-state pressure. Thus, even if the AC motor 4 is stopped, the hydraulic drive system according to the present embodiment performs the drive control of the AC motor 4 by the method described in the first embodiment, so that the rotor in the stopped state is obtained. The position can be estimated. Therefore, no problem occurs.
 なお、従来の油圧制御システムでは、リリーフバルブ52により、交流電動機4の多大な負荷となる油圧を逃がすことが必須であった。しかし、本実施形態によれば、以下で説明するように、リリーフバルブ52を排除してもよい。 In the conventional hydraulic control system, it has been essential to release the hydraulic pressure that is a great load of the AC motor 4 by the relief valve 52. However, according to the present embodiment, the relief valve 52 may be eliminated as described below.
 図16は、本発明の第4の実施形態に係る油圧制御システムの構成の変形例を示す図である。この油圧制御システムには、図15では存在しているリリーフバルブ52が設けられていない。すなわち、本実施形態による油圧制御システムでは、交流電動機4への全速度域の制御性能が向上できるため、過大負荷を避けるための機械的な保護装置であるリリーフバルブなしで、油圧のコントロールが可能となる。 FIG. 16 is a diagram showing a modification of the configuration of the hydraulic control system according to the fourth embodiment of the present invention. This hydraulic control system is not provided with the relief valve 52 which is present in FIG. That is, in the hydraulic control system according to the present embodiment, the control performance in the entire speed range for the AC motor 4 can be improved, so that the hydraulic pressure can be controlled without a relief valve which is a mechanical protection device for avoiding an excessive load. It becomes.
 以上説明した本発明の第4の実施形態によれば、様々な用途の油圧制御において本発明を適用することができる。 According to the fourth embodiment of the present invention described above, the present invention can be applied in hydraulic control for various uses.
 なお、上記第4の実施形態では、作動油の圧力を制御する油圧制御への適用例を説明したが、たとえば水など、他の流体の圧力制御において適用してもよい。すなわち本発明は、交流電動機の駆動によりポンプを動作させることで流体圧を制御する様々な流体圧制御システムにおいて適用可能である。 In the fourth embodiment, the application example to the hydraulic control for controlling the pressure of the hydraulic oil has been described. However, the fourth embodiment may be applied to the pressure control of other fluids such as water. That is, the present invention can be applied to various fluid pressure control systems that control fluid pressure by operating a pump by driving an AC motor.
 また、油圧制御ユニット25において、制御装置2に替えて制御装置2Bを用いることで、前述の第2の実施形態で説明したような構成を適用してもよい。さらに、第3の実施形態で説明したような構成としてもよい。 In the hydraulic control unit 25, the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.
(第5の実施形態)
 次に本発明の第5の実施形態について説明する。図17は、本発明の第5の実施形態に係る位置決めシステムの構成を示す図である。
(Fifth embodiment)
Next, a fifth embodiment of the present invention will be described. FIG. 17 is a diagram showing a configuration of a positioning system according to the fifth embodiment of the present invention.
 図17に示す位置決めシステムは、第1の実施形態で説明した制御装置2、インバータ3および交流電動機4と、q軸電流指令発生器1Cと、位置決め装置70により構成されている。 The positioning system shown in FIG. 17 includes the control device 2, the inverter 3, the AC motor 4, the q-axis current command generator 1C, and the positioning device 70 described in the first embodiment.
 q軸電流指令発生器1Cは、位置制御器101、速度制御器102、減算器6dおよび減算器6eにより構成される。 The q-axis current command generator 1C includes a position controller 101, a speed controller 102, a subtracter 6d, and a subtractor 6e.
 減算器6dは、外部から入力される位置指令θ*と、制御装置2により求められた位相角θdcとの差分を演算し、その演算結果を位置制御器101に出力する。位置制御器101は、減算器6dからの演算結果に基づいて、速度指令ωr*を求める。減算器6eは、位置制御器101からの速度指令ωr*と、制御装置2により求められた交流電動機4の回転速度ωrとの差分を演算し、その演算結果を速度制御器102に出力する。速度制御器102は、減算器6eからの演算結果に基づいて、q軸電流指令Iq*を求め、制御装置2に出力する。 The subtractor 6 d calculates the difference between the position command θ * input from the outside and the phase angle θdc obtained by the control device 2, and outputs the calculation result to the position controller 101. The position controller 101 obtains the speed command ωr * based on the calculation result from the subtractor 6d. The subtractor 6 e calculates the difference between the speed command ωr * from the position controller 101 and the rotational speed ωr of the AC motor 4 obtained by the control device 2, and outputs the calculation result to the speed controller 102. The speed controller 102 obtains a q-axis current command Iq * based on the calculation result from the subtractor 6 e and outputs it to the control device 2.
 位置決め装置70は、交流電動機4の負荷として、交流電動機4の回転軸に接続されている。この位置決め装置70は、たとえばボールねじなどを利用した装置であり、交流電動機4の駆動に応じて動作することで対象物の位置を調整することができる。なお、位置決め装置70には位置センサは取り付けられておらず、位置制御器101では、制御装置2から出力される位相角θdcをそのまま用いて、交流電動機4の駆動制御を行う。これにより、位置決め装置70の動作を制御して、位置指令θ*に応じた対象物の位置調整を行うことができる。したがって、位置センサを用いることなく、対象物の位置制御が可能となる。 The positioning device 70 is connected to the rotating shaft of the AC motor 4 as a load of the AC motor 4. This positioning device 70 is a device using, for example, a ball screw and can adjust the position of the object by operating in accordance with the driving of the AC motor 4. Note that no position sensor is attached to the positioning device 70, and the position controller 101 performs drive control of the AC motor 4 using the phase angle θdc output from the control device 2 as it is. Thereby, the operation of the positioning device 70 can be controlled, and the position of the object can be adjusted according to the position command θ *. Therefore, the position of the object can be controlled without using a position sensor.
 以上説明した本発明の第5の実施形態によれば、様々な対象物の位置制御において本発明を適用することができる。 According to the fifth embodiment of the present invention described above, the present invention can be applied in position control of various objects.
 なお、上記の位置決めシステムにおいて、制御装置2に替えて制御装置2Bを用いることで、前述の第2の実施形態で説明したような構成を適用してもよい。さらに、第3の実施形態で説明したような構成としてもよい。 In the positioning system described above, the configuration described in the second embodiment may be applied by using the control device 2B instead of the control device 2. Furthermore, it is good also as a structure as demonstrated in 3rd Embodiment.
 以上、本発明の様々な実施形態を具体的に説明したが、本発明は、これらの実施形態に限定されるものではなく、その要旨を逸脱しない範囲で種々の変更が可能であることは言うまでもない。上記の各実施形態や変形例はあくまで一例であり、発明の特徴が損なわれない限り、本発明はこれらの内容に限定されるものではない。 Although various embodiments of the present invention have been specifically described above, the present invention is not limited to these embodiments, and it goes without saying that various modifications can be made without departing from the scope of the present invention. Yes. The above-described embodiments and modifications are merely examples, and the present invention is not limited to these contents as long as the features of the invention are not impaired.
 次の優先権基礎出願の開示内容は引用文としてここに組み込まれる。
 日本国特許出願2013年第172202号(2013年8月22日出願)
The disclosure of the following priority application is hereby incorporated by reference.
Japanese patent application 2013 No. 172202 (filed on August 22, 2013)
 1:q軸電流指令発生器、2:制御装置、3:インバータ、4:交流電動機、5:d軸電流指令発生器、6a~6e:減算器、7:d軸電流制御器、8:q軸電流制御器、9:dq逆変換器、10:電圧補正部、11:PWM発生部、12:サンプルホールド回路、13:電流再現器、14:dq変換器、15:位置・速度推定部、16:速度変換部、17:120度通電部、18:120度PWM部、19a~19e:切替スイッチ、20:電圧指令発生部、21:電流推定部、31:直流電源、32:インバータ主回路、33:出力プリドライバ、34:直流母線、35:シャント電流検出器 1: q-axis current command generator, 2: control device, 3: inverter, 4: AC motor, 5: d-axis current command generator, 6a to 6e: subtractor, 7: d-axis current controller, 8: q Axis current controller, 9: dq inverse converter, 10: voltage correction unit, 11: PWM generation unit, 12: sample hold circuit, 13: current reproduction unit, 14: dq converter, 15: position / speed estimation unit, 16: Speed conversion unit, 17: 120-degree energization unit, 18: 120-degree PWM unit, 19a to 19e: changeover switch, 20: voltage command generation unit, 21: current estimation unit, 31: DC power supply, 32: inverter main circuit 33: Output pre-driver, 34: DC bus, 35: Shunt current detector

Claims (13)

  1.  三相の交流電動機と接続されたインバータに接続され、前記インバータにおいて正極側と負極側にそれぞれ三相ずつ設けられたスイッチング素子のスイッチング状態に応じて前記交流電動機の駆動を制御する交流電動機の制御装置であって、
     前記インバータは、前記スイッチング素子のスイッチング状態に応じて、8通りの電圧ベクトルを前記交流電動機に出力可能であり、
     前記電圧ベクトルは、正極側の全ての前記スイッチング素子がオンであり、負極側の全ての前記スイッチング素子がオフであるスイッチング状態と、正極側の全ての前記スイッチング素子がオフであり、負極側の全ての前記スイッチング素子がオンであるスイッチング状態とにそれぞれ対応する2種類の零ベクトルを含み、
     前記交流電動機の制御装置は、
     前記零ベクトルを除くいずれかの電圧ベクトルを第1の電圧ベクトルとして出力するように前記インバータを制御し、
     前記第1の電圧ベクトルを出力した後、前記零ベクトルを除いた前記第1の電圧ベクトルとは異なるいずれかの電圧ベクトルを、第2の電圧ベクトルとして出力するように前記インバータを制御し、
     前記第2の電圧ベクトルを出力した後、前記零ベクトルを出力するように前記インバータを制御し、
     前記零ベクトルを出力した後、前記第1の電圧ベクトルを反転した電圧ベクトルを第3の電圧ベクトルとして出力するように前記インバータを制御することにより、前記交流電動機の駆動を制御する交流電動機の制御装置。
    Control of an AC motor that is connected to an inverter connected to a three-phase AC motor and controls the driving of the AC motor according to the switching state of switching elements provided on the positive side and the negative side in the inverter, respectively. A device,
    The inverter can output eight voltage vectors to the AC motor according to the switching state of the switching element,
    The voltage vector includes a switching state in which all the switching elements on the positive electrode side are on and all the switching elements on the negative electrode side are off, and all the switching elements on the positive electrode side are off, Two types of zero vectors respectively corresponding to switching states in which all the switching elements are on,
    The control device for the AC motor is:
    Controlling the inverter to output any voltage vector except the zero vector as a first voltage vector;
    After outputting the first voltage vector, the inverter is controlled to output any voltage vector different from the first voltage vector excluding the zero vector as a second voltage vector;
    Controlling the inverter to output the zero vector after outputting the second voltage vector;
    After the zero vector is output, the inverter is controlled so that a voltage vector obtained by inverting the first voltage vector is output as a third voltage vector, thereby controlling the drive of the AC motor. apparatus.
  2.  請求項1に記載の交流電動機の制御装置において、
     前記第1の電圧ベクトル、前記第2の電圧ベクトル、前記零ベクトルおよび前記第3の電圧ベクトルを繰り返して順次出力するように前記インバータを制御し、
     前記第1の電圧ベクトルおよび前記第3の電圧ベクトルとして、前回とは異なる種類の電圧ベクトルをそれぞれ出力するように前記インバータを制御する交流電動機の制御装置。
    In the control apparatus for an AC motor according to claim 1,
    Controlling the inverter to repeatedly output the first voltage vector, the second voltage vector, the zero vector, and the third voltage vector sequentially;
    A control apparatus for an AC motor that controls the inverter to output different types of voltage vectors from the previous time as the first voltage vector and the third voltage vector, respectively.
  3.  請求項1または2に記載の交流電動機の制御装置において、
     前記第1の電圧ベクトルおよび前記第3の電圧ベクトルの出力期間中に、前記インバータの直流母線に流れる直流電流の検出結果に基づいて、前記交流電動機の相電流を求める電流推定部を備える交流電動機の制御装置。
    In the control apparatus of the alternating current motor according to claim 1 or 2,
    An AC electric motor comprising a current estimating unit that obtains a phase current of the AC electric motor based on a detection result of a direct current flowing through the DC bus of the inverter during an output period of the first voltage vector and the third voltage vector Control device.
  4.  請求項1乃至3のいずれか一項に記載の交流電動機の制御装置において、
     三相の電圧指令を発生する電圧指令発生部と、
     前記電圧指令を補正する電圧補正部と、
     前記電圧補正部により補正された補正後の電圧指令と、所定周期の三角波キャリアとを比較し、その比較結果に基づいて三相のPWM波形を発生するPWM発生部と、を備え、
     前記PWM発生部により発生されたPWM波形を前記インバータに出力することで、前記スイッチング素子のスイッチング状態を制御して、前記交流電動機の駆動を制御する交流電動機の制御装置。
    In the control apparatus of the alternating current motor according to any one of claims 1 to 3,
    A voltage command generator for generating a three-phase voltage command;
    A voltage correction unit for correcting the voltage command;
    Comparing the corrected voltage command corrected by the voltage correction unit with a triangular wave carrier of a predetermined period, and a PWM generation unit that generates a three-phase PWM waveform based on the comparison result,
    A control apparatus for an AC motor that controls a driving state of the AC motor by controlling a switching state of the switching element by outputting a PWM waveform generated by the PWM generator to the inverter.
  5.  請求項4に記載の交流電動機の制御装置において、
     前記電圧補正部は、
     前記三角波キャリアの連続する2周期のうち最初の周期では、前記三相の電圧指令で中間の電圧指令を0とし、さらに前記三相の電圧指令で最大の電圧指令および最小の電圧指令をシフトさせることにより、前記電圧指令を補正し、
     前記最初の周期に続く次の周期では、前記最小の電圧指令を0とし、さらに前記最大の電圧指令および前記中間の電圧指令をシフトさせることにより、前記電圧指令を補正する交流電動機の制御装置。
    In the control device for an AC motor according to claim 4,
    The voltage correction unit is
    In the first of two consecutive cycles of the triangular wave carrier, the intermediate voltage command is set to 0 by the three-phase voltage command, and the maximum voltage command and the minimum voltage command are shifted by the three-phase voltage command. By correcting the voltage command,
    In the next period following the first period, the minimum voltage command is set to 0, and the maximum voltage command and the intermediate voltage command are shifted to correct the voltage command.
  6.  請求項5に記載の交流電動機の制御装置において、
     前記電圧補正部は、
     前記最初の周期の前半期間では、前記最大の電圧指令と前記中間の電圧指令の差分に基づいて前記最大の電圧指令に対するシフト量を決定すると共に、前記最小の電圧指令と前記中間の電圧指令の差分および所定の必要シフト量に基づいて前記最小の電圧指令に対するシフト量を決定し、
     前記最初の周期の後半期間では、補正後の電圧指令が0となるように前記最大の電圧指令に対するシフト量を決定すると共に、前記必要シフト量に基づいて前記最小の電圧指令に対するシフト量を決定し、
     前記次の周期の前半期間では、前記最大の電圧指令と前記最小の電圧指令の差分に基づいて前記最大の電圧指令に対するシフト量を決定すると共に、前記必要シフト量に基づいて前記中間の電圧指令に対するシフト量を決定し、
     前記次の周期の後半期間では、補正後の電圧指令が0となるように前記最大の電圧指令に対するシフト量を決定すると共に、前記中間の電圧指令と前記最小の電圧指令の差分および前記必要シフト量に基づいて前記中間の電圧指令に対するシフト量を決定する交流電動機の制御装置。
    In the control device for an AC motor according to claim 5,
    The voltage correction unit is
    In the first half of the first cycle, a shift amount for the maximum voltage command is determined based on a difference between the maximum voltage command and the intermediate voltage command, and the minimum voltage command and the intermediate voltage command Determining a shift amount for the minimum voltage command based on the difference and a predetermined required shift amount;
    In the latter half of the first cycle, the shift amount for the maximum voltage command is determined so that the corrected voltage command is 0, and the shift amount for the minimum voltage command is determined based on the required shift amount. And
    In the first half of the next cycle, a shift amount for the maximum voltage command is determined based on a difference between the maximum voltage command and the minimum voltage command, and the intermediate voltage command is determined based on the required shift amount. Determine the shift amount for
    In the second half of the next cycle, the shift amount for the maximum voltage command is determined so that the corrected voltage command becomes 0, and the difference between the intermediate voltage command and the minimum voltage command and the necessary shift A control apparatus for an AC motor that determines a shift amount with respect to the intermediate voltage command based on the amount.
  7.  請求項6に記載の交流電動機の制御装置において、
     前記必要シフト量は、前記スイッチング素子のスイッチング状態が変化してから前記インバータの直流母線に流れる直流電流を検出するまでの時間に応じて設定される交流電動機の制御装置。
    The control apparatus for an AC motor according to claim 6,
    The required shift amount is a control device for an AC motor that is set according to a time from when a switching state of the switching element is changed until a DC current flowing through a DC bus of the inverter is detected.
  8.  請求項4乃至7のいずれか一項に記載の交流電動機の制御装置において、
     前記三角波キャリアが最小値と最大値の中間値をとる時点で前記インバータの直流母線に流れる直流電流の検出結果を取得する交流電動機の制御装置。
    In the control apparatus of the alternating current motor according to any one of claims 4 to 7,
    A control apparatus for an AC motor that acquires a detection result of a DC current flowing in a DC bus of the inverter when the triangular wave carrier takes an intermediate value between a minimum value and a maximum value.
  9.  請求項1乃至8のいずれか一項に記載の交流電動機の制御装置において、
     前記交流電動機が低速駆動している低速域では、前記スイッチング素子のうち、二相のスイッチング素子のみスイッチング状態を切り替えて120度通電駆動を行うように、前記インバータを制御し、
     前記低速域以外の速度域では、三相の前記スイッチング素子のスイッチング状態を切り替えて三相通電駆動を行うように、前記インバータを制御する交流電動機の制御装置。
    In the control apparatus of the alternating current motor according to any one of claims 1 to 8,
    In the low speed range where the AC motor is driven at a low speed, the inverter is controlled such that only the two-phase switching elements among the switching elements are switched to perform a 120-degree conduction drive.
    An AC motor control device that controls the inverter so as to perform three-phase energization driving by switching a switching state of the three-phase switching element in a speed region other than the low-speed region.
  10.  請求項9に記載の交流電動機の制御装置において、
     前記三相通電駆動において用いられる第1の電圧指令を出力する第1の電流制御器と、
     前記120度通電駆動および前記三相通電駆動において共通に用いられる第2の電圧指令を出力する第2の電流制御器と、を備え、
     前記第1の電流制御器は、外部から入力される電流指令または前記交流電動機に流れる電流の検出結果に基づくフィードフォワード演算により、前記第1の電圧指令を出力する交流電動機の制御装置。
    The control apparatus for an AC motor according to claim 9,
    A first current controller that outputs a first voltage command used in the three-phase energization drive;
    A second current controller that outputs a second voltage command commonly used in the 120-degree energization drive and the three-phase energization drive,
    The first current controller is a control device for an AC motor that outputs the first voltage command by a feedforward calculation based on a current command input from the outside or a detection result of a current flowing in the AC motor.
  11.  請求項1乃至10のいずれか一項に記載の交流電動機の制御装置と、
     前記交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、
     前記交流電動機の制御装置により、前記スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、を備え、
     前記交流電動機の制御装置、前記インバータおよび前記交流電動機は、一体化されている交流電動機駆動システム。
    A control device for an AC motor according to any one of claims 1 to 10,
    An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
    A three-phase AC motor that is driven and controlled in accordance with the switching state of the switching element by the AC motor control device,
    The AC motor control device, the inverter, and the AC motor are integrated with each other.
  12.  請求項1乃至10のいずれか一項に記載の交流電動機の制御装置と、
     前記交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、
     前記交流電動機の制御装置により、前記スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、
     前記交流電動機の駆動に応じて動作し、流体圧を制御するポンプとを備える流体圧制御システム。
    A control device for an AC motor according to any one of claims 1 to 10,
    An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
    A three-phase AC motor that is driven and controlled according to the switching state of the switching element by the AC motor control device;
    A fluid pressure control system comprising: a pump that operates according to driving of the AC motor and controls fluid pressure.
  13.  請求項1乃至10のいずれか一項に記載の交流電動機の制御装置と、
     前記交流電動機の制御装置に接続され、正極側と負極側にそれぞれ三相ずつスイッチング素子が設けられたインバータと、
     前記交流電動機の制御装置により、前記スイッチング素子のスイッチング状態に応じて駆動制御される三相の交流電動機と、
     前記交流電動機の駆動に応じて動作し、対象物の位置を調整する位置決め装置とを備える位置決めシステム。
    A control device for an AC motor according to any one of claims 1 to 10,
    An inverter connected to the control device of the AC motor and provided with switching elements of three phases on each of the positive electrode side and the negative electrode side;
    A three-phase AC motor that is driven and controlled according to the switching state of the switching element by the AC motor control device;
    A positioning system comprising a positioning device that operates according to driving of the AC motor and adjusts the position of an object.
PCT/JP2014/067664 2013-08-22 2014-07-02 Ac electric motor control device, ac electric motor drive system, fluid pressure control system, and positioning system WO2015025622A1 (en)

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