JP2003198341A - Current-amplifying comparator - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a current-amplifying comparator for suppressing offset current mΔ occurred at a current mirror circuit by amplifying a minute optical current through the use of a subthreshold area of a transistor. <P>SOLUTION: The comparator comprises transistors M1-M5, wherein a source of the transistor M1 is connected to an electric voltage Vdd, sources of the transistors M2 and M4 are commonly connected to a gate of the transistor M1 to be driven by a constant current, a reference voltage Vref is supplied to a gate of the transistor M4 from the voltage Vdd. The comparator comprises a current mirror circuit in which a source of the transistor M3 is connected to a GND (ground) voltage, a drain and a gate of the transistor M3 are short- circuited to be connected to a drain of the transistor M2, the source and the gate of the transistor M3 are connected to a source and a gate of the transistor M5, and a drain of the transistor M5 is connected to a drain of the transistor M4 to output. Consequently, the transistor M1 operates in the substhreshold area. <P>COPYRIGHT: (C)2003,JPO

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current type comparator which determines the magnitude of a current and outputs the result as a digital signal.

[0002]

2. Description of the Related Art In FIG. 4, third and fifth field-effect transistors (hereinafter simply referred to as third and fifth transistors with field effect terms omitted) whose sources are connected to a power supply Vdd are referred to as M23 and M25. The gate of is connected in common to the drain of the third transistor M23 to form a current mirror circuit. In the example shown in the figure, the photodiode P1 is connected to the power source 0V (GND) A minute photocurrent Iph proportional to the light hν flows as a drain current (input signal).

Further, the drain of the fourth transistor M24 is connected to the drain of the fifth transistor M25, the source thereof is connected to the power source 0V (GND), and the reference voltage Vref is connected between the power source 0V (GND) and this gate. The fourth transistor M2
Comparator output OUT is output from the drain of 4.
In such a configuration, the size of the third transistor M23 (m
= 1), the size of the fifth transistor M25 (m = k, k>
Select 1) and select the 5th voltage for the same source-gate voltage.
The drain current I25 of the transistor M25 is selected to be m times the drain current (input signal) Iph of the third transistor M23.

By such selection, the fourth transistor M2
Reference current Ir by the reference voltage Vref applied to the gate of 4
By comparing with ef, (1) When the input signal Iph is (m · Iph) ≧ Iref
From the transistor M25 to the comparator output OUT direction (m ・ Ip
The current of (h − Iref) flows, and the electric charge is accumulated in the parasitic capacitance C.

(2) When the input signal Iph is (m · Iph) <Iref,
Instantaneously from the parasitic capacitance C to the direction of the fourth transistor M24
A current of (Iref−m · Iph) flows. Therefore, the magnitude of the input signal Iph can be determined by the inflow / outflow of the output current to / from the parasitic capacitance C of the comparator output OUT. next,
A conventional WTA circuit will be described with reference to FIGS. In FIG. 5, two sets of transistors (M3
The WTA circuit composed of 1, M33) and (M32, M34) will be described. The sources of the transistors M31 and M32 are connected to 0V, the gates thereof are connected to a common line Vc, and the sources of the respective transistors M33 and M34 are connected thereto and driven by a constant current Ic. Then, the drain of the transistor M31 and the gate of the transistor M33 are connected in common so that the input signal I31
The drain of M32 and the gate of transistor M34 are connected in common and driven by input signal I32.
Output currents I33 and I34 are output to the drain of 4.

With such a configuration, the transistors M31 and M32 which operate with the input signals I31 and I32 are the same as the minute input signals I31 and I32.
6 operates in the sub-threshold region shown in FIG. 6, and a slight change in the drain currents I31, I32 appears as a large change in the drain-source voltages Vds1, Vds2. The drain-source voltages Vds1 and Vds2 differ from the common line potential Vc by the drain current of the transistors M33 and M34.
The drain-source voltages Vds1 and Vds2 are simultaneously amplified while setting the potential Vc so that the sum of I33 and I34 becomes equal to the constant current Ic. Generally, in such a circuit, when the difference between the drain-source voltages Vds1 and Vds2 is about several tens of mV, the larger input signal I31, I32 corresponds to the corresponding transistor M33, M34.
It is possible to operate so that the drain current of is fully output (Ic).

[0007]

In the current discriminating method according to the prior art, for example, when comparing currents of a few pA to several nA at a minute level such as a photocurrent, the fifth side on the reference side of the current mirror shown in FIG. 4 is used. Number of parallel stages of transistors (m = k, k>
1) is increased to equivalently reduce the current flowing through the fifth transistor on the reference side from a fraction to a few tens of minutes,
The current comparison with the current flowing through the third transistor on the comparison side is made possible.

However, in such a current determination method, (1) a plurality of fifth transistors on the reference side of the current mirror must be arranged in parallel, and the physical area of the layout becomes large. (2) It is difficult to adjust a minute current due to the offset of the reference-side fifth transistor arranged in parallel. There is a problem to say. That is, in the prior art of FIG. 4, the optical input h
The photocurrent Iph generated by ν flows through the third transistor M23. Third transistor M23 and fifth transistor M25
Constitute a current mirror circuit. For example, assuming that the transistors M23 and M25 have the same size, a plurality of transistors M25 are arranged in parallel.

Now, the transistors M23 and M25
Offset does not occur if and are exactly the same, but the source-gate voltage Vgs0 of the transistor M23 corresponding to the drain current Iph of the transistor M23 and the individual drain of the transistor M25 when the same source-gate voltage Vgs0 is added to the transistor M25. If only the offset current Δ is generated in the current, the drain current I25 of the transistor M25
= M (Iph + Δ), which is compared with the reference current Iref at the point shifted by the offset current mΔ.

The present invention has been made in view of the above points, and an object thereof is to solve the above-mentioned problems and to amplify a minute photocurrent by utilizing the subthreshold region of the transistor M23 to obtain a current mirror. It is to provide a current amplification type comparator that suppresses an offset current mΔ generated in a circuit.

[0011]

According to the present invention, there is provided a first field effect transistor having a source connected to one potential of a direct current voltage, a minute input current to be judged being inputted to the drain, and both sources. Second and fourth field effect transistors, which are commonly connected to the gate of the first field effect transistor and driven by a predetermined constant current, and the drain of the first field effect transistor and the gate of the second field effect transistor. And a predetermined reference voltage is applied to the gate of the fourth field effect transistor, the source is connected to the other potential of the DC voltage, and the drain and gate are short-circuited to make the drain of the second field effect transistor. Third to connect with
A field effect transistor, a fifth field effect transistor having a source and a gate of the third field effect transistor connected to corresponding sources and gates, and a drain connected to a drain of the fourth field effect transistor to serve as a current output terminal; And a current mirror circuit consisting of.

With such a configuration, the minute input current to be judged inputted to the drain of the first field effect transistor drives the first field effect transistor to the operating point in the subthreshold region. As a result, a minute change in the input current input to the drain of the first field effect transistor in the subthreshold region can cause a large change in the source-drain voltage Vds1 of the first field effect transistor. .

As a result, the currents can be compared by using the voltage amplified by the first field effect transistor as one input signal of a commonly used comparator. Further, one potential of the DC voltage is set to + potential and the other potential is set to GND potential, and the minute input current to be determined is received by the photosensor connected between the GND potential and the drain of the first field effect transistor. Is the reverse sensor current flowing through the
The second and fourth field effect transistors can be P-channel field effect transistors, and the third and fifth field effect transistors can be N-channel field effect transistors.

Further, one potential of the DC voltage is the GND potential and the other potential is the + potential, and the minute input current to be judged is
The photosensor connected between the + potential and the drain of the first field effect transistor receives and flows in the reverse direction, and the first, second, and fourth field effect transistors are N-channel field effect transistors. The third and fifth field effect transistors can be P-channel field effect transistors.

[0015]

1 is a circuit diagram of a current amplification type comparator according to an embodiment of the present invention, FIG. 2 is an operation explanatory diagram for explaining a subthreshold region of a P-type MOSFET, and FIG. 3 is another embodiment. It is a circuit diagram of a current amplification type comparator according to an example. In FIG. 1, a current amplification type comparator using a part of a WTA circuit according to the present invention has a source connected to one potential Vdd of a DC voltage, and a first minute input current (Iph) to be judged is inputted to a drain. The transistor M1 and the second and fourth transistors M2 and M4, which are connected between both sources and the gate of the first transistor M1 in common and are driven by the predetermined constant current Ic of the constant current circuit M6, and the DC voltage A predetermined reference voltage Vref is applied between one of the potentials Vdd and the gate of the fourth transistor M4, the source is connected to the other potential GND of the DC voltage, and the drain and the gate are short-circuited to connect the second transistor M2. Connecting the source and gate of the third transistor M3 for inputting the drain current of the third transistor M3 to the corresponding source and gate, and connecting the drain to the drain of the fourth transistor M4. A current mirror circuit composed of a fifth transistor M5 for output and a current mirror circuit can be provided.

When the switch is off, the gate potential of the second transistor M2 is high, and the second transistor M2 is off. On the other hand, the fourth transistor M4 has the reference voltage Vr.
ef is applied and a predetermined current is passed. At this time, since the current I2 of the second transistor M2 does not flow, the third and fifth currents
No current flows through the transistors M3 and M5. Therefore, the current
Iref flows into the parasitic capacitance C of output OUT. When the switch is turned on, a small input current (Iph) is input, and the gate potential of the fourth transistor M4 becomes higher than the gate potential of the second transistor M2, the current I2 of the second transistor M2 is increased.
Is greater than the current Iref, I2 = Iref + I5, and the current I5
Flows out from the parasitic capacitance C.

The minute input current to be judged (Iph) inputted to the drain of the first transistor M1 drives the first transistor M1 at the operating point of the subthreshold region shown in FIG. 2 described later. As a result, a minute change (I1 → I2) in the input current (Iph) input to the drain of the first transistor M1 in the subthreshold region causes a large change in the source-drain voltage Vds1 of the first transistor M1. (Vds1 → Vds2) is generated. As a result, the source-drain voltage Vds1 of the first transistor M1 and the reference voltage
The comparison with Vref can be performed by a comparator.

[0018]

(Embodiment 1) In FIG. 1, the current amplification type comparator shown in the drawing is configured such that one potential Vdd of a DC voltage is a positive potential and the other potential OV is a GND potential, and a minute input current (Iph) to be judged is detected. ) Is the reverse sensor current Iph that flows when the photosensor P1 connected between the GND potential and the drain of the first transistor M1 receives the light hν and flows through the first, second, and fourth transistors M1, M2, and M4. Is a P-channel transistor, the third and fifth transistors M3.M5 are N-channel transistors, and the comparator output current can flow from the drain of the fourth transistor M4 or flow out to the drain of the fifth transistor M5.

With this configuration, when the switch is turned on and the optical signal hν is incident on the photosensor (photodiode) P1, a photocurrent Iph depending on this light amount flows through the photo diode P1. It is known that this photocurrent Iph is at a very minute level of at most 100 pA to several nA. The drain of the first transistor M1 and the photodiode
Since P1 is connected, this photocurrent Iph becomes the drain current of the first transistor M1. The first transistor (Pch-FET) M1 must exist in the sub-threshold region shown in FIG. 2 in order for the first transistor M1 to pass a minute current of several nA level as the drain current.
In this region, a large voltage change can be generated even with a minute current change.

FIG. 2 shows the drain current-drain-source voltage of the transistor in the subthreshold region.
Shows Vds characteristics. In FIG. 2, the horizontal axis represents the drain-source voltage Vds of the P-channel transistor, and the transistor drain-source voltage Vds is shown on the 0V side from the potential of the power supply voltage Vdd. Further, the drain current Id shows the characteristics on a logarithmic scale in the illustrated example.
(I1, Vds1), (I2, Vds2) when drain current Id is several nA level
The operating point is in the subthreshold region on the solid line shown in FIG.

As a result, the drain of the first transistor M1
Source-to-source voltage Vds (≈gate potential of second transistor M2)
Since the value of is greatly changed, the maximum current capacity that the second transistor M2 can flow changes depending on the gate potential of the second transistor M2. The circuit composed of the first transistor M1 and the second transistor M2 is Winn described in the prior art.
er-Take-All (WTA circuit) Part of the "winning circuit" and is used to determine the magnitude of minute current. That is,
A minute photocurrent Iph generated by the optical signal hν can be used for judgment by current amplification in a circuit forming a part of the WTA circuit.

Next, the current amplifying means by the first transistor M1 and the second transistor M2, and the second transistor M2
The operation of the fourth transistor M4 will be described. Returning to FIG. 1, in order to simplify the description, the second transistor M2 and the fourth transistor M2
The characteristics are exactly the same as those of the transistor M4. Now, it is assumed that the drain-source voltage Vds (Vds1) of the first transistor M1 is equal to the reference voltage Vref. At this time, the current Ic by the constant current circuit M6 that drives the second transistor M2 and the fourth transistor M4 is equal to the current Ic of the second transistor M2 and the fourth transistor M4.
Both source potentials Vc fluctuate as floating potentials, and the drain currents of both transistors are determined so that the drain currents are equally distributed to (1/2) Ic.

Next, when the drain-source voltage Vds (Vds1 → Vds2) of the first transistor M1 becomes slightly higher than the gate voltage (reference voltage Vref) of the fourth transistor M4, the second transistor
The drain current of the transistor M2 increases and the drain current of the fourth transistor M4 decreases. Now, tentatively second and fourth
Assuming that the current amplification factor Gm due to the increase / decrease in the drain current of the transistors (M2, M4) is almost constant, the fluctuation of the floating potential Vc is almost the same as the fluctuation width of the drain-source voltage Vds (Vds1 → Vds2) of the first transistor M1. It can be considered as half. That is, the change width of the comparator output current is (Gm / 2)
We can expect a change in (V2-V1). (Embodiment 2) Another embodiment of the current type comparator described in Embodiment 1 will be described with reference to FIG. In Figure 3, one of the DC voltages is the GND potential (0V) and the other is the + potential.
(Vdd) and the minute input current (Iph) to be judged is + potential (Vd
The photosensor (photodiode) P1 connected between d) and the drain of the first transistor M11 outputs an optical signal hν.
Is the reverse sensor current Iph that flows by receiving
The second and fourth transistors M11, M12, M14 are N-channel transistors, the third and fifth transistors M13, M15 are P-channel transistors, and the comparator output current is the fifth.
It can flow from the drain of the field effect transistor M15 or flow out to the drain of the fourth field effect transistor M14.

The difference from the first current type comparator described in FIG. 1 is that the insertion position of the photosensor (photodiode) P1 is changed from the GND side to the potential Vdd.
The fourth transistors M11, M12, M14 are changed from P-channel transistors to N-channel transistors, and the third and fifth transistors M13, M15 are changed from N-channel transistors to P-channel transistors.

This operation is performed as the optical signal hν increases.
Except that the operation direction of the comparator output current with respect to the GND level is reversed, the operation is the same as that described for the first current type comparator, and therefore detailed description is omitted. According to the present invention, the following features can be obtained. That is, (1) Since a minute photocurrent is once converted into a voltage signal, the reference voltage can be adjusted very easily.

(2) Further, since it is possible to strengthen the fluctuation of a minute current, the accuracy of the current comparator can be improved. (3) Since the magnitude of the current can be determined at a relatively large current level, the magnitude determination can be speeded up. (4) Since it is not necessary to arrange a plurality of transistors on the reference side of the current mirror in parallel, the layout is simplified and the chip area can be reduced.

[0027]

According to the present invention, the sub-threshold region of the first transistor is used to detect a minute photocurrent as a voltage change, and this voltage change is compared by a comparator to obtain the current of the prior art. It is possible to provide a current amplification type comparator that can suppress the offset current mΔ generated in the mirror circuit.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a current type comparator according to an embodiment of the present invention.

FIG. 2 is an operation explanatory diagram illustrating a subthreshold region of a WTA circuit.

FIG. 3 is a circuit diagram of a current type comparator according to another embodiment.

FIG. 4 is a circuit diagram of a current type comparator using a current mirror circuit according to a conventional technique.

FIG. 5 is a prior art 2-cell WTA circuit diagram.

FIG. 6 is an operation explanatory diagram illustrating a subthreshold region of a conventional WTA circuit.

[Explanation of symbols]

M1 to M6, M11 to M16, M23 to M25, M31 to M34 Transistor P1 Photo sensor hν Optical signal Iph Photocurrent I2, I5, Iref Current Vref Reference voltage Vdd, GND DC voltage

─────────────────────────────────────────────────── ─── Continuation of front page (51) Int.Cl. ⁷ Identification code FI theme code (reference) H03K 17/78 H03K 17/78 L 5J500 F term (reference) 5J039 DA17 DB18 DC05 KK17 LL03 MM00 NN06 5J050 AA01 BB24 CC14 DD18 EE02 EE32 FF10 5J066 AA01 AA12 CA13 FA06 HA10 HA17 HA19 HA29 HA38 HA44 KA05 KA09 MA21 MD05 ND01 ND14 ND22 ND23 PD01 TA02 5J091 AA01 AA12 CA13 FA06 HA10 HA01 HA17 HA12 A02 HA01 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 HA12 TA12 HA44 KA05 KA09 MA21 TA02 UL02 5J500 AA01 AA12 AC13 AF06 AH10 AH17 AH19 AH29 AH38 AH44 AK05 AK09 AM21 AT02 DM05 DN01 DN14 DN22 DN23 DP01 LU02

Claims (3)

[Claims] 1. A source is connected to one potential of a DC voltage,
A second field effect transistor having a drain to which a minute input current to be determined is input and a source and a gate of the first field effect transistor are commonly connected to drive with a predetermined constant current. A fourth field effect transistor, a drain of the first field effect transistor and a gate of the second field effect transistor are connected to each other, and
A third reference for applying a predetermined reference voltage to the gate of the field effect transistor, connecting the source to the other potential of the direct current voltage, short-circuiting the drain and gate, and connecting to the drain of the second field effect transistor. A field effect transistor, and a fifth field effect transistor in which the source and gate of the third field effect transistor are connected to the corresponding source and gate, and the drain is connected to the drain of the fourth field effect transistor to serve as a current output terminal. And a current mirror circuit including the current mirror circuit, wherein the first field effect transistor operates in a subthreshold region. 2. The current amplification type comparator according to claim 1, wherein one potential of the DC voltage is set to + potential and the other potential is set to + potential.
The ground potential is set to the ground potential, and the minute input current to be determined is a reverse direction sensor current flowing by receiving light by a photosensor connected between the ground potential and the drain of the first field effect transistor. A current amplification type comparator characterized in that the fourth field effect transistor is a P-channel field effect transistor, and the third and fifth field effect transistors are N-channel field effect transistors. 3. The current amplification type comparator according to claim 1, wherein one potential of the DC voltage is a GND potential and the other potential is a + potential, and the minute input current to be judged is the + potential and the first potential. The reverse sensor current flows when a photosensor connected between the drain and the field effect transistor receives light, and the first, second, and fourth field effect transistors are N-channel field effect transistors, and the third, The fifth field-effect transistor is a P-channel field-effect transistor, A current amplification type comparator characterized in that.