EP3483983A1 - Receiving antenna for satellite navigation on a vehicle - Google Patents

Abstract

An antenna for receiving circularly polarized satellite radio signals comprises a conductor loop, which is designed as a loop emitter and forms a resonant structure. The circumference of the loop is shorter than the free-space wavelength "and there are at least three vertical radiators, which are coupled via reactance circuits with a base. The input resistance of two ports of a distribution and phase shifter network for electromagnetic excitation of the line shaft on the loop is in each case a real resistance and each of the gates is coupled to a respective ring line cross point.

Description

The invention relates to an antenna for receiving circularly polarized satellite radio signals, in particular for satellite radio navigation.

Satellite radio signals are transmitted due to polarization rotations in the transmission path usually with circularly polarized electromagnetic waves and are used in all known satellite navigation systems. Modern navigation systems provide, in particular for the global accessibility in connection with a high navigation accuracy in the mobile navigation, to evaluate the simultaneously received radio signals of several satellite navigation systems. Such composite-receiving systems are collectively referred to as GNSS (Global Navigation Satellite System) and include known systems such as GPS (Global Positioning System), GLONASS, Galileo and Beidou, etc. Satellite antennas for vehicle navigation typically become available the electrically conductive outer skin of the vehicle body constructed. Circularly polarized satellite receiving antennas are used, such as those described, for example, in US Pat DE 10 2009 040 910 A or the DE 40 08 505 A are known. For the construction on vehicles are particularly those antennas, which are characterized by a low overall height in conjunction with cost-effective manufacturability. This includes, for example, especially from the DE10 2009 040 910 A known, designed as a resonant structure ring line radiator with a small volume, which is imperative especially for mobile applications. The antenna has a small footprint and is very low with a height of less than one tenth of the free space wavelength.

As a further receiving antennas for satellite navigation on vehicles patch antennas are known in the prior art, which, however, compared to stamped sheet metal antennas are more complex in construction. A challenge to satellite antennas for GNSS is the demand for a large one Frequency bandwidth, which is given for example in GPS by the frequency band L1 with the center frequency 1575 MHz (required bandwidth about 80 MHz) and the frequency band L2 with the center frequency 1227 MHz (required bandwidth about 53 MHz). This requirement is covered, for example, by separate antennas assigned to one of the frequency bands L1 and L2, respectively, or by a broadband antenna comprising both frequency bands. Systems for the simultaneous evaluation of signal contents in the frequency bands L1 and L2 place particularly high demands on the antennas. And this with a small amount of available space, as it is always given especially in vehicle construction. The use of separate, in close proximity antennas for the two frequency bands involves the problem of mutual electromagnetic coupling with the effect of influencing the directional diagrams and the polarization purity and in particular the cross polarization. Due to the signals of the positioning satellites which are incident at low elevation angles, suppression of the opposite direction of polarization - cross polarization (LHCP) - is of decisive importance with regard to correct location results, even with sufficient gain in the desired, mostly right-handed circular polarization direction (RHCP). The accuracy of the location result is thus particularly affected by the ratio of the desired direction of polarization to the cross polarization of the satellite receiving antenna, so the Kreuzpolarisationsabstand. On the other hand, the realization of a satellite navigation antenna is technically difficult, which covers both frequency bands with a bandwidth of about 360 MHz at a center frequency of about 1385 MHz and still meets the sometimes stringent requirements for the Kreuzpolarisationsabstand and the antenna gain. Satellite receiving antennas for satellite navigation are particularly intended for installation on horizontal surfaces of the electrically conductive vehicle body. With respect to the antenna properties, the substantially horizontal vehicle roof acts as a conductive base.

For use on vehicles are particularly satellite receiving antennas with a small volume. Antennas of this type according to the prior art are known as patch antennas. However, these are less efficient in terms of low elevation angle reception and more complex in construction. This disadvantage is partially solved by loop antennas, such as those in the DE 10 2009 040 910 A are described. However, even for such antennas, it is desirable to improve the cross-polarization distance over the full bandwidth of the above-described frequency bands L1, L2 or L5.

The object of the present invention is therefore to provide an antenna for receiving circularly polarized satellite radio signals for satellite navigation, which has a sufficient gain with a small volume and a high cross-polarization distance over a large frequency range and thus is particularly accurate for the acquisition Locating results in a vehicle is suitable.

This object is solved by the features of claim 1.

The particular advantage of the invention is given by the fact that the strict requirement of the frequency bandwidth of the Kreuzpolarisationsabstands in conjunction with a low overall volume of the antenna, as it exists for current and especially for future cars, can be met.

An antenna according to the invention further has the advantage that it can be produced in a particularly cost-effective manner and is thus particularly suitable for mass production and use in the mass production of vehicles.

The inventive idea is, inter alia, to realize the antenna with a shortened loop and still comply with the strict requirements in terms of Kreuzpolarisationsabstands in a wide frequency range. This is u.a. achieved in that the resonance of the ring line at the frequency f0 is made by the vertical resonant radiators on their reactance circuits with capacitive reactance X without the involvement of the arrangement for exciting the antenna.

The loop emitter is then in self-resonance at the resonance frequency f0 when the excitation is not involved in the resonance formation is. The antenna impedance 37 present at this frequency is a high impedance real resonance resistor and is often between 300 ohms to 500 ohms. Thus - due to the inventive demand of a real input resistance 43 of the distribution and phase shifter network 8 at the gates T2 and T3 - the natural resonance of the ring line emitter 2 is not affected by the excitation 3. This ensures that no resonant currents of the ring line radiator 2 flow through the ports T2 and T3 of the distribution and phase shifter network 8. The separate adjustment of the resonance of the ring line radiator 2 and the distribution and phase shifter network 8 with respect to the real input resistance 43 of the gates T2 and T3 to the resonance frequency f0 leads to the invention according to the large frequency bandwidth of the suppression of the cross polarization of the antenna. 1

The inventive requirement of a real input resistance of the distribution and phase shifter network 8 at the gates T2 and T3 - by which the natural resonance of the loop emitter 2 is not affected by the excitation 3 - can naturally only at a frequency, i. for the resonant frequency f0 are exactly and in the frequency environment of the resonant frequency f0 only approximately fulfilled. Compliance with this requirement at or near the resonance frequency f0 makes it possible to achieve the widest possible bandwidth in the frequency environment of the resonance frequency f0.

According to the invention, the large frequency bandwidth of the cross-polarization spacing in the vicinity of the resonant frequency of the loop emitter can be achieved if the capacitive and the inductive reactive power of the loop emitter are balanced without a contribution from the means for exciting the loop emitter. The particular frequency bandwidth of the Kreuzpolarisationsabstands to the resonant frequency f0 is inventively achieved in that the two gates T2, T3 of the distribution and phase shifter network with a corresponding phase difference ΔΦ the two - this phase difference of a current wave on the ring line of the ring line radiator representing - vertical resonance excitation radiator in the Way, that no reactive power exchange takes place over these gates at resonance. This is due to the demand for the real input resistance of the gates T2, T3 of the distribution and phase shifter network achieved. This shows that the achievable frequency bandwidth of the cross-polarization distance is particularly large when this real input resistance of the gates corresponds approximately to the large resonance resistance of the loop emitter at the frequency f0 of up to 500 ohms.

Advantageous embodiments of the invention will be described below.

The distribution and phase shifting network (8) may be associated with two of the vertical resonance radiators (4a, 4b) as vertical resonance excitation radiators (10a, 10b) in such a way that each of the ports T2, T3 of the distribution and phase shifting network (8) respectively is coupled via one of the two vertical resonance excitation emitters (10a, 10b) with its ring line coupling point (7a, 7b) and the other vertical resonance emitters (4c, 4d, ..) are active as passive vertical resonance emitters (9a, 9b, etc.).

Each of the ports T2, T3 can each be directly connected via one of the resonance excitation emitters 10a, 10b to the ring line cross-over point 7 and the input resistance of the gates T2, T3 in the high impedance real, that between this and the high impedance resonance resistance of the loop emitter 2 resistance adjustment consists.

By virtue of the vertical resonant radiators 4a, 4b, 4c loaded with the reactive resistance circuit 13 with capacitive reactance X, the stretched length L of the ring line 14 of the resonant ring conductor 2 can be shortened from approximately the free space wavelength λ to approximately one third of the free space wavelength λ ,

The reactance circuit 13 with which each of the passive vertical resonant radiators 9, 9a, 9b, and also the resonance excitation radiator 10a, 10b is coupled to the base 6 can each be formed by a capacitance.

The reactance circuit 13 with which the vertical resonance excitation radiators 10a, 10b are coupled to the base 6 can each be referred to as a series circuit of a first reactance circuit 20a, 20b and a second reactance resistor circuit 21a, 21b may be formed with a connecting point 19 connecting them, to each of which one of the ports T2, T3 is connected.

The reactance circuit 13 to which the vertical resonance excitation emitters 10a, 10b are coupled to the base 6 and the first reactance circuit 20a, 20b of each vertical resonance excitation emitter 10a, 10b may pass through a first capacitance 22a, 22b and the second reactance circuit 21a, 21b a second capacitance 23a, 23b may be formed, and the size and ratio of the first and second capacitances may each be selected such that resistance matching is achieved between the high impedance resonant resistance of the loop emitter 2 and a real input impedance of the lower impedance level gates T2, T3 given is.

The reactance circuits 13c, etc., with which the passive vertical resonant radiators 9c, etc. are coupled to the electrically conductive base 6, can each be formed from the series connection of a capacitor 28 and a parallel oscillation circuit consisting of a parallel capacitor 45 and a parallel inductor 46 and the first reactance circuit 20a, 20b of the vertical resonance excitation emitters 10a, 10b may each be formed as a first capacitance 22a, 22b, and their second reactance circuit 21a, 21b may be respectively formed as a parallel oscillation circuit consisting of a second capacitance 23a, 23b and a shunt inductance 46a , 46b and the resonant frequency of all the parallel arcing circuits can each be selected in the frequency range between the upper frequency band L1 and the lower frequency band L2 in such a way that the reactance of these resonant circuits in the frequency band L1 is capacitive and in the frequency band L2 in is productive.

However, it may be that between the capacitance 28 and the parallel oscillation circuit of the parallel capacitance 45 and the parallel inductance 46 of the reactance circuits 13c, etc. of the passive vertical resonant radiator 9c, etc. in each case a further parallel resonant circuit 44, etc. is connected in series and the first capacitance 22a, 22b in the first reactance circuit 20a, 20b, the vertical resonance excitation emitters 10a, 10b to the junction point 19a, 19b is also followed by a further parallel resonant circuit 44a, 44b and the parallel resonant circuits in the first reactance circuit 20a, 20b and second reactance circuit 21a, 21b of the vertical resonance excitation emitters 10a, 10b are tuned such that their reactance Xa, Xb is capacitive in both frequency bands L1, L2 and their relation to each other is according to the optimum measure to match and the reactance circuits 13c , etc. of the passive vertical resonant radiator 9c, etc. and the vertical resonant exciter radiators 10a, 10b are matched with respect to each other in their frequency response.

The first capacitance 22a, 22b of the first reactance circuit 20a, 20b of the vertical resonance excitation emitters 10a, 10b and the capacitances 28 in the reactance circuits 13c, etc. of the passive vertical resonance emitter 9c, etc. may be formed such that all the vertical resonance emitters 4a, 4b, etc. are formed at their lower end to individually shaped area capacitance electrodes 32a, 32b, 32c, 32d, that the capacitances 22a, 22b, 28 by interposing a dielectric plate 33 between the sheet capacitance electrodes 32a, 32b, 32c, 32d and the electrically conductive base 6 designed as an electrically conductive printed circuit board 35 for coupling the passive vertical resonant radiators 9c, 9d, etc. to the electrically conductive base 6, and capacitive coupling of the vertical resonance excitation radiators 10a, 10b on the electrically conductive base 6 can each one of the senior Layer of the coated printed circuit board 35 isolated, planar counter electrode 34 for connecting the second reactance circuit 21b be designed.

The vertical radiators may be formed for capacitive coupling at its lower end to individually shaped area capacitance electrodes 32a, 32b, 32c, 32d, but it may be for coupling the passive vertical resonant radiators 9, 9a, 9b each one of the conductive layer of the coated circuit board 35th insulated, flat counter-electrode 34 for connecting the reactance circuit 13c, 13d formed from concentrated reactive elements may be formed.

The conductive structure consisting of the ring line 14 and the vertical resonance radiators 4, 4a-d connected thereto can be fixed by a dielectric support structure 36 in such a way that the dielectric plate 33 is realized in the form of an air gap.

There may be a total of three vertical resonant radiators 4a, 4b, etc. having azimuthally equally distributed loop coupling points 7, two of which are excited as vertical resonant excitation emitters 10a, 10b via the ports T2, T3 of a distribution and phase shifting network 8 Phase angle ΔΦ = 120 degrees.

There may be a multiplicity of N> 4 vertical resonance radiators 4, 4a, 4b, 4c,... With coupling points 7 distributed identically over the extended length L of the ring line 14, so that N ring line sections 30a, 30b, 30c, 30d are formed are and between the crosspoints 7 of the two vertical resonance excitation emitters 10a, 10b, .. n ring line crosspoints are selected and the phase angle ΔΦ = (n + 1) * 2π / N <180 ° is selected.

The loop can be designed square. There may be four vertical resonance radiators 4a, 4b, 4c, 4d with azimuthally equal length ring line sections 30a, 30b, of which in particular two azimuthally consecutive as vertical resonance excitation radiators 10a, 10b, .. over the gates T2, T3 of a distribution - And phase shifter network 8 are energized whose phase angle ΔΦ = 90 degrees.

However, the ring line can be rectangular in shape with four ring line crosspoints 7, 7a,... 7d at the corners and with different lengths of the ring line sections 30a, 30b and with different ring conductor widths 15a, 15b, whereby by appropriately selecting the ring conductor widths 15a, 15b are set for the ring line sections 30a, 30b respectively the same characteristic impedance ZL and the same effective electrical length.

A distribution and phase shifting network 8 for a phase shift ΔΦ = 90 ° can contain a hybrid ring 38 which, in addition to the two ports T2 and T3, has a fourth gate T4 closed with an ohmic terminating resistor 40 and which ends with corrugation resistance of the gates T2 and T3 is decoupled from the gate T1.

For a phase shift ΔΦ = 90 °, the distribution and phase shifting network 8 may comprise a Wilkinson divider 39 with an ohmic balancing resistor 41, the input at the port 1 and one of the two output branches comprising a phase rotation element 17 with the phase angle ΔΦ = 90 ° to the gate T3 is connected and at the other of the two output branches the gate T2 is formed.

There may be a first and a second distribution and phase shifter network 8 with phase angle ΔΦ = 90 ° and four designed as excitation radiators vertical radiators azimuthally distributed evenly on the circumference of the ring conductor, wherein the first distribution and phase shifter network 8 with its gates T2, T3 with a first pair of vertical resonance excitation emitters 10a, 10b, and the second distribution and phase shifting network 8 is coupled with its ports T2a, T3a to a second pair of vertical resonance excitation emitters 10c, 10d opposite the first pair, and another distribution - And phase shifter network 8 with ΔΦ = 180 ° is present, which is connected to the excitation of the rotating line shaft with its gates T2c, T3c to the input ports T1, T1 in such a way that adjusts the rotating shaft on the ring line.

The loss-free matching circuit 18 for transforming the low-impedance level Z0 into the high-impedance real input resistance in both output branches of the distribution and phase-shifting network 8 can be essentially by a lambda / 4-transformation line 12a, 28b and a series circuit comprising an inductance 27a, 27b and a series circuit Capacity 28a, 28b for fine adjustment of the high impedance input resistance of the gates T2 and T3 included.

All vertical resonant radiators (4a, 4b, 4c, etc.) along the ring line (14) can be approximately equally distributed so that none of the distances between adjacent ring line crosspoints (7a, 7b, 7c) on the circumference of the ring line radiator (2) is less than half the distance with equidistant distribution over the extended length of the loop (14).

By the action of the vertical resonant radiators (4, 4a, 4b, 4c) loaded with the reactive resistance circuit (13, 13a, ... 13c, ...), the stretched length L of the ring line (14) of the in-line resonant radiator (4) can be determined Resonant ring line radiator (2) starting from about the free space wavelength λ can be shortened to about one third of the free space wavelength λ.

The vertical radiators (4a,... 4d) for capacitive coupling at their lower end can be formed into individually designed areal capacitive electrodes (32a, 32b, 32c, 32d), but for coupling the passive vertical resonant radiators (9c, 9d) in each case one of the conductive layer of the coated printed circuit board (35) isolated, planar counter electrode (34) for connecting the parallel capacitor (45a, 45b) and the parallel inductance (46a, 46b) formed parallel resonant circuit to the ground terminal point (11) is formed. ( Fig. 13a . b . 14 . Fig. 5 )

The conductive structure, consisting of the ring line (14) and the associated vertical radiators (4, 4a-d) can be fixed by a dielectric support structure (36) so that the dielectric plate (33) is realized in the form of an air gap. ( Fig. 13a . b . 14 )

A multiplicity of N> 4 vertical resonance radiators (4a, 4b, 4c,...) With coupling points (7) equally distributed over the extended length L of the ring line (14) can be present, so that N ring line sections (30a,... 30d ) are formed and between the crosspoints (7a, 7b) of the two vertical resonance excitation emitters (10a, 10b) n loop coupling points are selected and the Phase angle ΔΦ = (n + 1) * 2π / N <180 ° is selected. ( Fig. 7 )

The ring line (14) can be designed rectangular with four loop coupling points (7a, .. 7d) at the corners and with different lengths of the loop portions (30a, 30b) and with different ring conductor widths (15a, 15b), wherein by accordingly Choice of the ring conductor widths (15a, 15b) for the ring line sections (30a, 30b) are each set the same characteristic impedance ZL and the same effective electrical length. ( Figure 9 . 10a, 10b, 10c )

The phase shifting phase shift distribution and phase shifting network (8) may include a Wilkinson divider (39) having an ohmic balancing resistor (41), the input at port 1 and one of the two output branches of the Wilkinson divider (39) is followed by a phase rotation member (17) with the phase angle ΔΦ = 90 ° to the gate T3 and at the other of the two output branches Tor T2 is formed. ( Fig. 11c )

All vertical resonant radiators (4a, 4b, 4c, etc.) can be approximately equally distributed along the ring line (14), so that none of the distances between adjacent ring line crosspoints (7a, 7b, 7c) on the circumference of the ring line radiator (2) smaller is greater than half the equidistant spacing over the stretched length of the loop (14).

A first and a second distribution and phase shifting network (8) with phase angle ΔΦ = 90 ° and four vertical radiators designed as excitation radiators may be present azimuthally distributed in the same circumference of the ring line (14), the first distribution and phase shifting network (8). with its ports T2, T3 is coupled to a first pair of resonant excitation emitters 10a, 10b and the second distribution and phase shifting network (8) is coupled with its ports T2a, T3a to a second pair of excitation emitters 10c, 10d opposite the first pair and another distribution and phase shifter network (8) with ΔΦ = 180 ° is present, which is connected to the energizing of the circulating line shaft with its gates T2c, T3c with the gates T1, T1a in such a way that on the ring line, the rotating shaft established.

According to a further aspect, the invention relates to an antenna arrangement in which a plurality of respective frequency band associated antennas of the type described above are provided, the ring line emitters (2) are arranged concentrically with each other in such a way that the ring line radiator (2) for the highest in frequency Frequency band is arranged at the innermost and each further ring line emitter (2) surrounds a ring line emitter for the next higher frequency band.

• Fig. 1:
  Antenne 1 nach der Erfindung, bestehend im Beispiel aus dem Ringleitungsstrahler 2 mit dem Ringleiter 14 in der Höhe h < 0,15λ über der leitenden Grundfläche 6 mit N = 3 vertikalen Resonanzstrahlern 4a, 4b, 4c an azimutal gleich verteilten Ringleitungs-Koppelpunkten 7a, 7b, 7c und der elektromagnetischen Erregung 3 durch das Verteil- und Phasenschiebernetzwerk 8. Die vertikalen Resonanzstrahler 4a, 4b, 4c sind jeweils über eine kapazitiv wirkende Blindwiderstandsschaltung 13a,13b,13c über den Masse-Anschlusspunkt 11 mit der leitenden Grundfläche 6 verkoppelt. Die elektromagnetische Erregung 3 erfolgt über das Verteil- und Phasenschiebernetzwerk 8, welches eingangsseitig mit seinem Tor T1 mit dem Antennenanschluss 5 verbunden ist und dessen Tore T2 und T3 mit zwei von den vertikalen Resonanzstrahlern 4a, 4b, 4c als vertikale Resonanz-Erregungsstrahler 10a und 10b in der Weise verbunden sind, dass die zugehörigen Ringleitungs-Koppelpunkte 7a, 7b direkt mit den im Verteilnetzwerk 16 und im Phasendrehglied 17 bewirkten Ausgangssignalen an den Toren T2 und T3 mit Phasenwinkelunterschied ΔΦ = 360°/N = 120° angesteuert sind, sodass sich auf dem Ringleiter 14 die laufende Leitungswelle einstellt. Hierbei ist erfindungsgemäß vorausgesetzt, dass die Tore T2 und T3 jeweils einen reellen Eingangswiderstand 43 besitzen, sodass die Resonanz des Ringleitungsstrahlers 2 durch Anschluss der elektromagnetischen Erregung 3 nicht beeinflusst ist. Zur Impedanzanpassung zwischen dem Verteil- und Phasenschiebernetzwerk 8 an den Toren T2, T3 und der hochohmigen Antennenimpedanz 37 bei der Resonanzfrequenz f0 des Ringleitungsstrahlers 2 (sh. Fig. 2b) an den Ringleitung-Koppelpunkten 7a und 7b ist das Verteil- und Phasenschiebernetzwerk 8 hochohmig gestaltet und befindet sich auf einem Impedanzniveau von etwa 300 - 500 Ohm.

• Fig. 2:
  Figur a) zeigt den Ringleitungsstrahler 2 nach der Erfindung mit drei azimutal gleich verteilten vertikalen Resonanzstrahlern 4a - 4c als Resonanzstruktur. Die Erregung 3 ist nicht dargestellt. Die Blindwiderstandsschaltungen 13 a- 13c sind durch die Kapazitäten 28a, 28b, 28c realisiert. Bei Rotationssymmetrie der Anordnung sind alle Kapazitäten gleich groß. Bei Abstimmung des Ringleitungsstrahlers 2 auf eine Resonanzfrequenz- zum Beispiel f0= 1392 MHz - stellt sich an den vertikalen Resonanzstrahlern 4a- 4c jeweils an der mit 42 bezeichneten Messstrecke der in Figur 2b) dargestellte Verlauf der Antennenimpedanz 37 mit ihrem durch die Strahlung bedingten reellen Resonanzwiderstand von ca. 340 Ohm ein. Bei Anschluss der Erregung an die Resonanzstrahler 4a und 4b - wie in Figur 1 - wird das Resonanzverhalten der Struktur in der Frequenz nicht verändert. Im Beispiel der 3 azimutal gleich verteilten vertikalen Resonanzstrahler 4a, 4b, 4c beträgt der Phasendrehwinkel des Phasendrehglieds ΔΦ= 120°.
• Fig.3:
  Antenne 1 nach der Erfindung wie in Figur 1 jedoch mit einem Verteilnetzwerk 16 und einem Phasendrehglied 17 auf dem Impedanzniveau Z0 üblicher koaxialer Leitungen (Z0 = 50 Ohm). Zur Anpassung an das hochohmige Impedanzniveau der Antennenimpedanz 37 (sh. Figur 2b) sind dem Phasendrehglied 17 und einem Arm des Verteilnetzwerks 16 jeweils ein Anpassnetzwerk 18 nachgeschaltet. Der Eingangswiderstand 43 an den Toren T2 und T3 ist - ebenso wie in Figur 1 - bei der Resonanzfrequenz des Ringleitungsstrahlers 2 hochohmig und reell.
• Fig.3a:
  Verteil- und Phasenschiebernetzwerk 8 wie in Figur 3 mit beispielhafter Ausführung der Anpassnetzwerke 18 an den beiden Ausgängen. Die Anpassung bei Resonanz des Ringleitungsstrahlers 2 ist durch die λ/4-Transformationsleitungen 12a, 12b, durch welche die Transformation von dem niederohmigen Impedanzniveau Z0 zu dem hochohmigen Impedanzniveau der Antennenimpedanz 37 gegeben ist, bewirkt. Die Serienresonanzkreise, bestehend aus den Kapazitäten 28a, 28b und den Induktivitäten 12a und 12b ermöglichen eine Feinkorrektur der Anpassung über einen erweiterten Frequenzbereich.
• Fig.4:
  Antenne nach der Erfindung wie in Figur 3 jedoch mit besonderer Gestaltung der Blindwiderstandsschaltungen 13a und 13b in den beiden vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils unterteilt in eine erste Blindwiderstandsschaltung 20a, 20b und eine zweite Blindwiderstandsschaltung 21a, 21b mit einem dazwischen liegenden Verknüpfungspunkt 19a, 19b zum Anschluss der Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8.
• Fig.5:
  Erfindungsgemäße Ausführungsformen der insgesamt kapazitiv wirkenden Blindwiderstandsschaltungen 13 zur Ankopplung der vertikalen Resonanzstrahler an die leitende Grundfläche 6 in Figur 4
  1. a) Blindwiderstandsschaltung 13a bzw. 13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b, unterteilt in die erste Blindwiderstandsschaltung 20a bzw. 20b und die über den Verknüpfungspunkt 19a bzw. 19b verbundene zweite Blindwiderstandsschaltung 21a bzw. 21b, jeweils realisiert durch eine erste Kapazität 22a bzw. 22b und entsprechend durch die zweite Kapazität 23a bzw.23b.
  2. b) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert durch eine Kapazität 28. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur a) beschaltet.
  3. c) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur a) jedoch mit einem Parallelresonanzkreis, bestehend aus der Parallelinduktivität 46a bzw. 46b und der Parallelkapazität 23a bzw.23b in der zweiten Blindwiderstandsschaltung 21a bzw. 21b zur Erweiterung der Frequenzbandbreite der Antenne.
  4. d) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert, wie unter Figur b) jedoch mit einem seriellen Parallelresonanzkreis bestehend aus der Parallelinduktivität 46 und der Parallelkapazität 45 in Serienschaltung zur Kapazität 28 zur Erweiterung der Frequenzbandbreite der Antenne in Entsprechung der Gestaltung der Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 4a bzw. 4b wie unter c). Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur c) beschaltet.
  5. e) Darstellung der Frequenzbänder L1 und L2 für die Satelliten-Navigation mit den Mittenfrequenzen fm1 und fm2 und den unteren und oberen Grenzfrequenzen fu1, fo1 bzw.fu2, fo2. Die Frequenz fm beschreibt die Mittenfrequenz zwischen fu1 und fo2.
  6. f) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur c) jedoch mit einem weiteren Parallelresonanzkreis 44a bzw. 44b in Serienschaltung zur ersten Kapazität 22a bzw. 22b hin zum Verknüpfungspunkt 19 zur gesonderten Optimierung der Schaltung für jeweils einen der beiden Frequenzbereiche L1 und L2 zur Gestaltung einer Zweibandantenne.
  7. g) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert wie unter Figur d), jedoch mit einem weiteren Parallelresonanzkreis 44 in Serienschaltung zur Kapazität 28 in der Blindwiderstandsschaltung 13c. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur f) beschaltet zur Gestaltung einer Zweibandantenne.
  8. h) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur f) jedoch mit einer Zusatz-Parallelkapazität 47, gebildet durch die, die isolierte Gegenelektrode 34 überkragende Fläche der Kapazitätselektrode 32a, 32b in den Figuren 13c und 13d mit der elektrisch leitenden beschichteten Leiterplatte 35. Die Zusatz-Parallelkapazität 47 ermöglicht die Erweiterung des Bereichs der Impedanzanpassung jeweils am Verknüpfungspunkt 19a, 19b bei ebenso gegebener Einhaltung der Resonanzbedingung für den Ringleitungsstrahler 2. Die Resonanzbedingung und die Impedanzanpassung werden durch Abstimmung der Größen der Kapazitätselektroden 32a, 32b und der isolierten Gegenelektroden 34 aufeinander hergestellt.
• Fig.6:
  Anpassungsverhältnisse bei der Resonanzfrequenz f0 in Abhängigkeit vom Maß "t" für die Unterteilung in die erste Kapazität 22a, 22b und die zweite Kapazität 23a, 23b am Beispiel des Ringleitungsstrahlers 2 in Figur 4.
  1. a) Reflexionsfaktor an der Antennenanschlusstelle 5 als Eingangstor T1 eines verlustfreien Verteil- und Phasenschiebernetzwerks 8 der Antenne 1 in Abhängigkeit vom Teilungsmaß t . Anpassung ist mit topt = 0,13 erreicht.
  2. b) Transformationsfaktor bei der Resonanzfrequenz f0 zwischen dem Eingangswiderstand 43 mit Z0 = 50 Ohm des Phasendrehglieds 17 und dem Resonanzwiderstand bei f0 am Ringleitungsstrahler-Koppelpunkt 7 in Abhängigkeit vom Teilungsmaß t.
  3. c) Relative Leistung P/Pmax bei f0 in Abhängigkeit vom Teilungsmaß t.
• Fig.7:
  Antenne nach der Erfindung mit Widerstands-Anpassung durch kapazitive Unterteilung wie im Beispiel der Figur 4 jedoch mit insgesamt 6 azimutal um jeweils 60° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a .. 4f, von denen vier als vertikale passive Resonanzstrahler 9a..9d gestaltet sind. Die Erregung 3 erfolgt beispielhaft an den beiden um 120° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 120° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• Fig.8:
  Antenne nach der Erfindung wie in Figur 7, jedoch mit insgesamt 8 azimutal um jeweils 45° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a..4h. Die Erregung 3 erfolgt beispielhaft an den beiden um 90° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 90° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• Fig.9:
  Antenne nach der Erfindung mit rechteckig ausgeführter Ringleitung 14 mit vier vertikalen Resonanzstrahler 4a bis 4d im Bereich der Ringleitungs-Ecken. Zwei der vertikalen Resonanzstrahler sind als vertikale Resonanz-Erregungsstrahler 10a, 10b und die beiden übrigen Strahler als passive vertikale Resonanzstrahler 9a, 9b ausgeführt. Impedanzanpassung an das niederohmig (z.B. Z= 50 Ohm) ausgeführte Verteil- und Phasenschiebernetzwerk 8 ist durch kapazitive Unterteilung, ähnlich wie in den Figuren 4, 7 und 8 erreicht. Die Ringleitung 14 ist vorzugsweise quadratisch gestaltet mit gleichen Ringleiterbreiten 15a, 15b und gleichen Ringleitungs-Abschnitten 30a, 30b. Sowohl die Ringleiterbreite 15a, 15b als auch die Ringleitungs-Abschnitte 30a, 30b können jedoch abgestimmt aufeinander in Grenzen unterschiedlich gewählt sein.
• Fig. 10:
  Rechteckförmiger Ringleitungsstrahler 2 einer Antenne 1 nach der Erfindung ohne Darstellung der elektromagnetischen Erregung 3 zur Erläuterung der Resonanzstruktur des Ringleitungsstrahlers 2.
  1. a) geometrischer Aufbau des Ringleitungsstrahlers 2 mit großen unterschiedlich wählbaren Ringleiterbreiten 15a und 15b sowie den unterschiedlich wählbaren Ringleitungs-Abschnitten 30a und 30b. Die vertikalen Resonanzstrahler 4a - 4d sind mit ihren Ringleitungs-Koppelpunkten 7a - 7d im weiten Bereich der Ecken des rechteckförmigen Ringleiters 14 vorgesehen. Die unterbrochene Linie kennzeichnet etwa den Verlauf der Schwerlinie 24 der Stromdichteverteilung der Leitungswelle bei Erregung des Ringleitungsstrahlers 2. Aufgrund der Stromverdrängung verdichtet sich die Stromverteilung hin zum Rand des Ringleiters 14. Der für die Funktion bei der Erregung des Ringleitungsstrahlers 2 relevante Strom der Leitungswelle fließt demnach - gekennzeichnet durch die als unterbrochene Linie gezeichnete Schwerlinie der Stromverteilung 24 - auch bei großen Leiterbreiten 15a, 15b mehr zum Rand hin gedrängt. Dies gilt insbesondere für sehr große Ringleiterbreiten 15a, 15b bis hin zur vollkommenen Schließung der inneren Öffnung zum Zentrum hin, welches praktisch stromlos ist, wenn die Ringleitung 14 als eine geschlossene leitende Fläche realisiert ist.
  2. b) gemäß dem durch die Schwerpunktlinie der Stromdichteverteilung 24 repräsentierten Strom kann die Resonanzstruktur des Ringleitungsstrahlers 2 durch ein grob angenähertes Ersatzschaltbild allgemein wiedergegeben werden. Die einzelnen Ringleitungs-Abschnitte 30a -30d sind jeweils durch die induktive und kapazitive Wirkung des zugehörigen Abschnitts der Ringleitung 14 unter Einbeziehung der kapazitiven Wirkung der Blindwiderstandsschaltung 13 als konzentrierte induktive Elemente (Ln) und kapazitive Elemente (Cn) dargestellt. Jeder n-te Ringleitungsabschnitt 30a - 30b ist durch eine π-Struktur gemäß Figur c), bestehend aus einer Längsinduktivität 2*Ln und jeweils einer Querkapazität Cn an deren beiden Enden, dargestellt. Die Strahlungsdämpfung jedes horizontal orientierten Ringleitungsabschnitt 30a - 30b ist durch den Dämpfungsfaktor d der konzentrierten Induktivität einbezogen. Die Aneinanderreihung einander benachbarter Ringleitungs-Abschnitte 30a-30d erfolgt jeweils über einen gemeinsamen vertikalen Resonanzstrahler 4a..4d unter Zusammenziehung der Querkapazitäten Cn der benachbarten Ringleitungs-Abschnitte, wie in Figur c) dargestellt. Die geringfügige induktive Wirkung der vertikalen Resonanzstrahler ist bei dieser Grundsatzbetrachtung aufgrund der geringen Antennenhöhe h vernachlässigt. Erfindungsgemäß vorteilhafte Voraussetzungen für den Ringleitungsstrahler 2 sind für die Resonanzfrequenz f0=ωo/2π dann erreicht, wenn jeweils alle Wellenwiderstände $\mathit{ZLn}=\sqrt{\frac{\mathit{Ln}}{\mathit{Cn}}}$
     
     und jeweils alle $\sqrt{\mathit{Ln}\square \mathit{Cn}}={}^1{/}_{\mathit{<figure-callout\; id="1"\; label="\omega on"\; filenames="imgf0006.png,imgf0013.png"\; state="\{\{state\}\}">\omega on</figure-callout>}},$
     
     welche die elektrische Länge des Ringleitungs-Abschnitts repräsentieren, für alle n gleich groß sind.
     Bei rotationssymmetrischen Ringleitungsstrahlern 2 ist dies stets gegeben. Andernfalls kann diese Voraussetzung zum Beispiel bei einer rechteckförmigen Struktur des Ringleitungsstrahlers 2 durch individuelle Gestaltung der Ringleiterbreiten 15a-15d in den einzelnen Ringleitungs-Abschnitten 30a-30b erreicht werden.
• Fig.11:
  1. a) Gestaltung des Verteil- und Phasenschiebernetzwerks 8 - zum Beispiel wie in den Figuren 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Hybridring 3, durch welchen sowohl die Leistungsteilung als auch die Phasenverschiebung erfolgt. Bei Einspeisung am Tor T1 wird der Ringleitungsstrahler 2 über die Tore T2 und T3 mit dem Phasenunterschied von 90° erregt. Der Abschluss von Tor T4 mit dem ohmschen Abschlusswiderstand 40 bewirkt bei Abweichung der Frequenz von der Resonanzfrequenz f0 breitbandig die teilweise Absorption der Leistung der unerwünschten Polarisation bei Einspeisung am Tor T1.
  2. b) Idealisierte Streumatrix zur Beschreibung des allgemein bekannten grundsätzlichen Wellenverhaltens eines Hybridrings 38 in Figur a) an den Toren T1 bis T4. Die wichtige Entkopplung der Tore T1 und T4 voneinander ist durch die Streuparameter S14 = 0, S41= 0 in der Matrix fett gekennzeichnet.
  3. c) Gestaltung des Verteil- und Phasenschiebernetzwerks 8 - zum Beispiel wie in den Figuren 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Wilkinson-Teilers 16 mit nachgeschaltetem λ/4 Leitungs-Phasenschieber 17 an einem Tor T3* zur Erzeugung des Phasenunterschieds von 90°. Der ohmsche Symmetrierungswiderstand 40 absorbiert bei unsymmetrischer Belastung des Wilkinson-Teilers 16 bei Abweichung von der Resonanzfrequenz f0 teilweise die Ströme, welche die unerwünschte Polarisation auf dem Ringleitungsstrahler 2 hervorrufen.
• Fig.12:
  Antenne nach der Erfindung z. B. wie in Figur 9 mit rechteckig geformter Ringleitung 14. Die Kapazitäten 22a, 22b und 28 sind in der Weise gebildet, dass die vertikalen Resonanzstrahler 4a - 4d an ihrem unteren Ende zu individuell gestalteten flächigen Kapazitätselektroden 32a, 32b, 32c, 32d ausgeformt sind. Durch Zwischenlage zwischen diesen und die als elektrisch leitend beschichtete Leiterplatte ausgeführte elektrisch leitende Grundfläche 6 befindliche dielektrische Platte 33 sind die Kapazitäten 28 zur Ankopplung der passiven vertikalen Resonanzstrahler 4c, 4d an die elektrisch leitende Grundfläche 6 gestaltet. Zur kapazitiven Ankopplung der aktiven vertikalen Resonanzstrahler 10a, 10b an die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8 ist dieser Anschluss als jeweils eine von der leitenden Schicht isolierte, flächige Gegenelektrode 34 gestaltet. Die Gegenelektroden 34 können somit als Verknüpfungspunkte 19 der Antenne in Figur 9 ausgeführt sein und können als Anschlusspunkte für die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8, wie in Figur 13 angedeutet, dienen.
• Fig.13:
  Antenne 1 ähnlich wie in Figur 11, jedoch ist die dielektrische Wirkung der dielektrischen Platte 33 durch einen Luftspalt realisiert. Typische Abmessungen eines Ringleitungsstrahlers 2 für den Frequenzbereich L1 sind für quadratische Antennen ein Abmessung von 30mm bis 40mm und für die Höhe h = 8mm. Zur Widerstandsanpassung des vorzugsweise auf niedrigem Impedanzniveau (Z0) ausgeführten Verteil- und Phasenschiebernetzwerks 8 ist die kapazitive Unterteilung wie in Figur 9 vorgesehen.
  1. a) Die beispielhaft zugehörigen Blindwiderstandsschaltungen 13a, 13b und 13c sind in den Figuren 5a und 5b dargestellt. Die Kapazität 28 der Kapazitätselektrode 32c, 32d gegen die elektrisch leitende Grundfläche 6 bzw. die Kapazität 22a, 22b gegen die isolierte Gegenelektrode 34a, 34b beträgt jeweils ca. 0.3pF. Die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8 mit ihrem reellen Eingangswiderstand 43 sind jeweils an eine Gegenelektrode 34a, 34b als Verknüpfungspunkt 19a, 19b angeschlossen. Die kapazitive Unterteilung zur Impedanzanpassung ist jeweils durch die Kapazität 22a, 22b und der Kapazität 23a, 23b gegeben. Diese zwischen dem Verknüpfungspunkt 19a, 19b und dem Massepunkt 11 eingebrachte Kapazität 23a, 23b ist auf der Rückseite der beschichteten Leiterplatte 35 als SMD-Bauteil angebracht. Hierzu sind auf der Rückseite der Leiterplatte isolierte Pads 29 als Kontaktstützpunkte gestaltet, welche jeweils über eine Durchkontaktierung 26 mit der Gegenelektrode 34 einerseits und andererseits z.B. über die Kapazität 23a, 23b (sh. Fig. 4 u. 5a) mit Masse verbunden sind.
  2. b) die Blindwiderstandsschaltungen 13a, 13b der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur a) gemäß Figur 5a bzw. wie in 5c gestaltet. Bei Gestaltung nach Figur 5c ist der auf der Rückseite der beschichteten Leiterplatte 35 eingebrachten Kapazität 23a, 23b eine Induktivität 46a, 46b als SMD-Bauteil parallel geschaltet. Im Gegensatz zur Figur a) sind zur Bildung der Blindwiderstandsschaltungen der beiden passiven Resonanzstrahler 9a, 9b den Kapazitätselektroden 32c, 32d auf der beschichteten Leiterplatte 35 isolierte Gegenelektroden 34 gegenübergestellt. Ausgehend von der Gegenelektrode 34 als Kontaktstützpunkt sind auf der Rückseite der Leiterplatte 35 die Parallelinduktivität 46 und die Parallelkapazität 45 zum Massepunkt 11 auf der Leiterplatte 35 geschaltet, sodass die Blindwiderstandsschaltung 13 in Figur 5d realisiert ist.
     Bei Ausgestaltung aller Blindwiderstandsschaltungen 13 aller vertikalen Resonanzstrahler mit jeweils einer Kapazitätselektrode 32 und einer gegenüberliegenden isolierten Gegenelektrode 34 können alle Schaltungen in den Figuren 5c und 5f für die Resonanz-Erregungsstrahler 10a, 10b durch Einsatz von SMD-Bauteilen auf der Rückseite der Leiterplatte 35 realisiert werden. Auf gleiche Weise können die Blindwiderstandsschaltungen für die passiven Resonanzstrahler 9a, 9b durch Beschaltung der isolierten Gegenelektroden 34 mit SMD-Bauteilen auf der Rückseite der Leiterplatte 35 gemäß den in den Figuren 5d und 5g angegebenen Schaltungen realisiert werden.
  3. c) Wie unter Figur 5h beschrieben dient die Zusatz-Parallelkapazität 47 zur freien Gestaltung der Impedanzanpassung bei Beibehaltung der Resonanzeigenschaften des Ringleitungsstrahlers 2. Die Figur zeigt die großflächige Überdeckung der isolierten Gegenelektrode 34a, 34b mit der Kapazitätselektrode 32a, 32b der Resonanz-Erregungsstrahler 10a, 10b. Der in der vertikalen Projektion der Kapazitätselektrode 32a, 32b gegenüber der isolierten Gegenelektrode 34 bestehende flächige Überstand bildet mit der elektrisch leitenden Schicht der beschichteten Leiterplatte 35 die Zusatz-Parallelkapazität 47. Die Blindwiderstandsschaltung der beiden passiven Resonanzstrahler 9a, 9b sind wie in Figur a) gestaltet.
  4. d) die Blindwiderstandsschaltungen 13 der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur c) mit einer Zusatz-Parallelkapazität 47 gestaltet. Die beispielhaft möglichen Blindwiderstandsschaltungen 13 der beiden passiven Resonanzstrahler 9a, 9b können wie in Figur b) beschrieben, gestaltet werden.
• Fig.14:
  Im Bild ist die Oberseite der Leiterplatte 35 einer Antenne 1 nach der Erfindung dargestellt, auf welche der elektrische Ringleitungsstrahler 2 aufgesetzt wird. Für die breitbandige Gestaltung einer Antenne 1 nach der Erfindung - welche zum Beispiel beide Frequenzbereiche L1 und L2 mit einer zwischen den beiden Frequenzbereichen liegenden Mittenfrequenz fm (sh. Figur 5e) - ist die Blindwiderstandsschaltung 13, wie in den Figuren 5c bzw. 5d gestaltet. Hierfür ist für die Blindwiderstandsschaltung in den beiden aktiven vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils die Blindwiderstandsschaltung 13a, 13b in Figur 5c und für die beiden passiven vertikalen Resonanzstrahler 9a, 9b die Blindwiderstandsschaltung 13c in Figur 5d vorgesehen. Dies ist bei dem Beispiel im Figur 13a dadurch erreicht, dass für alle Kapazitätselektroden 32 jeweils eine Gegenelektrode 34 vorhanden ist und, dass der durch die Kapazitätselektroden 32 bewirkten Kapazität 22a, 22b bzw. 28 an allen vertikalen Resonanzstrahlern 4a - 4d eine Parallelschaltung aus einer Parallelkapazität 23a, 23b bzw. 45 und einer Parallelinduktivität 46a, 46b, bzw. 46 - dargestellt als SMD- Bauteile - zwischen der Gegenelektrode 34 und der elektrisch leitenden Grundfläche 6 in Serie geschaltet ist. Figur 13a zeigt die Oberseite der Leiterplatte 35 mit Durchkontaktierungen 26 auf den Gegenelektroden 34 unter den Kapazitätselektroden 32. In Figur 14b ist die Unterseite der Leiterplatte 35 mit den Pads 29 dargestellt, welche mit den isolierten Gegenelektroden 34 über die Durchkontaktierungen 26 verbunden sind,. Die Blindelemente sind als SMD-Bauelemente gemäß den Figuren 5c und 5d angebracht. Zusätzlich sind die beiden Verknüpfungspunkte 19 mit der jeweils daran angeschlossenen Serienschaltung aus der Kapazität 28a, 28b und der Induktivität 27a, 27b als Teile der in Figur 3a enthaltenen Anpassungsschaltung 18 des Verteil-und Phasenschiebernetzwerks 8 zur Bildung der Tore T2 und T3 dargestellt.
• Fig. 15
  Zeigt eine Antenne nach der Erfindung mit vier azimutal am Umfang der Ringleitung 14 gleich verteilten Resonanz-Erregungsstrahlern 10a bis 10d. Alle Strahler werden entsprechend einer umlaufenden Welle mit jeweils 90°-Phasenunterschied zwischen benachbarten Strahlern erregt. Hierzu sind ein erstes und ein zweites Verteil- und Phasenschiebernetzwerk (8) mit Phasenwinkel ΔΦ = 90° vorhanden, wobei das erste Verteil- und Phasenschiebernetzwerk (8) mit seinen Toren T2, T3 mit einem ersten Paar der Resonanz-Erregerstrahler 10a, 10b verkoppelt ist und das zweite Verteil-und Phasenschiebernetzwerk (8) mit seinen Toren T2a, T3a mit einem dem ersten Paar gegenüberliegendem zweiten Paar der Erregerstrahler 10c, 10d verkoppelt ist. Zur Anregung einer laufenden Welle werden einander gegenüberliegende Resonanz-Erregungsstrahler 10a, 10c bzw. 10b, 10d jeweils mit einem Phasenunterschied von ΔΦ = 180° erregt. Hierzu ist ein weiteres Verteil- und Phasenschiebernetzwerk 31, 39 mit einer Phasenverschiebung von ΔΦ = 180° vorhanden, welches mit seinem Tor T2c mit dem Tor T1 des ersten Verteil- und Phasenschiebernetzwerks 8, 36 und mit seinem Tor T3c mit dem Tor T1b des weiteren Verteil- und Phasenschiebernetzwerks 8, 38b verbunden ist. Auch hier kommt erfindungsgemäß vorzugsweise jeweils ein Hybridring 38 zum Einsatz. Das 180° Phasennetzwerk 31 kann beispielhaft als Wilkinson-Teiler 39 mit einer λ/2-langen Verzögerungsleitung 17 (sh. Figur 11c) eingesetzt werden. Alle Anschlüsse sind in der Weise gewählt, dass sich auf der Ringleitung die leitende Welle in der gewünschten Umlaufrichtung einstellt. Mit dieser Anordnung ist der besondere Vorteil der azimutal symmetrischen Anregung verbunden, sodass der Abgleich der Anordnung besonders problemfrei erfolgen kann. Als Einschränkung ist jedoch der erhöhte Aufwand anzumerken.

The invention will be further explained below with reference to exemplary embodiments. The accompanying figures show in detail:

• Fig. 1 :
  Antenna 1 according to the invention, consisting in the example of the ring line radiator 2 with the ring conductor 14 in the height h <0.15λ above the conductive base 6 with N = 3 vertical resonant radiators 4a, 4b, 4c at azimuthally equally distributed loop coupling points 7a, 7b, 7c and the electromagnetic excitation 3 through the distribution and phase shifter network 8. The vertical resonance radiators 4a, 4b, 4c are each coupled via a capacitive reactance circuit 13a, 13b, 13c via the ground terminal 11 with the conductive base 6. The electromagnetic excitation 3 takes place via the distribution and phase shifting network 8, which is connected on the input side to the antenna terminal 5 with its gate T1 and whose ports T2 and T3 are connected to two of the vertical resonance radiators 4a, 4b, 4c as vertical resonance excitation emitters 10a and 10b are connected in such a way that the associated loop coupling points 7a, 7b are controlled directly with the output signals caused in the distribution network 16 and in the phase rotation member 17 at the gates T2 and T3 with phase angle difference ΔΦ = 360 ° / N = 120 °, so that on the ring conductor 14 adjusts the current line wave. In this case, the invention assumes that the ports T2 and T3 each have a real input resistance 43, so that the resonance of the ring line radiator 2 is not influenced by the connection of the electromagnetic excitation 3. For impedance matching between the Distribution and phase shifter network 8 at the gates T2, T3 and the high-impedance antenna impedance 37 at the resonant frequency f0 of the loop emitter 2 (see FIG. Fig. 2b ) at the ring line crosspoints 7a and 7b, the distribution and phase shifting network 8 is designed high impedance and is at an impedance level of about 300 - 500 ohms.

• Fig. 2 :
  Figure a) shows the ring line emitter 2 according to the invention with three azimuthally equally distributed vertical resonant radiators 4a - 4c as a resonant structure. The excitement 3 is not shown. The reactance circuits 13a-13c are realized by the capacitances 28a, 28b, 28c. With rotational symmetry of the arrangement, all capacities are equal. When the ring line emitter 2 is tuned to a resonant frequency, for example f0 = 1392 MHz, the vertical resonance emitter 4a-4c in each case adjoins the measuring path designated 42 in FIG FIG. 2b ) course of the antenna impedance 37 with their due to the radiation real resonance resistance of about 340 ohms. Upon connection of the excitation to the resonant radiators 4a and 4b - as in FIG. 1 - The resonance behavior of the structure is not changed in frequency. In the example of the three azimuthally uniformly distributed vertical resonant radiators 4a, 4b, 4c, the phase rotation angle of the phase rotation element ΔΦ = 120 °.
• Figure 3 :
  Antenna 1 according to the invention as in FIG. 1 but with a distribution network 16 and a phase shifter 17 on the impedance level Z0 usual coaxial lines (Z0 = 50 ohms). To adapt to the high impedance impedance level of the antenna impedance 37 (see FIG. FIG. 2b ) are the phase shift member 17 and an arm of the distribution network 16 each have a matching network 18 connected downstream. The input resistor 43 at the gates T2 and T3 is - as well as in FIG. 1 - At the resonant frequency of the ring line radiator 2 high impedance and real.
• 3a :
  Distribution and phase shifting network 8 as in FIG. 3 exemplary implementation of matching networks 18 at the two outputs. The matching at resonance of the loop antenna 2 is effected by the λ / 4 transform lines 12a, 12b, which cause the transformation from the low impedance level Z0 to the high impedance level of the antenna impedance 37. The series resonant circuits, consisting of the capacitances 28a, 28b and the inductors 12a and 12b, allow a fine correction of the adaptation over an extended frequency range.
• Figure 4 :
  Antenna according to the invention as in FIG. 3 However, with particular design of the reactance circuits 13a and 13b in the two vertical resonance excitation emitters 10a, 10b each divided into a first reactance circuit 20a, 20b and a second reactance circuit 21a, 21b with an intermediate node 19a, 19b for connecting the gates T2 and T3 of the distribution and phase shifter network 8.
• Figure 5 :
  Inventive embodiments of the total capacitive reactance circuits 13 for coupling the vertical resonant radiator to the conductive base 6 in FIG. 4
  1. a) reactance circuit 13a and 13b of the two vertical active resonance excitation emitters 10a and 10b, respectively, divided into the first reactance circuit 20a and 20b and connected via the node 19a and 19b second reactance circuit 21a and 21b, respectively realized by a first Capacitors 22a and 22b, respectively, and second capacitors 23a and 23b, respectively.
  2. b) reactance circuit 13c of the remaining passive vertical resonance radiator 4c, realized by a capacitance 28. The two active resonance excitation radiators are connected as in FIG.
  3. c) reactance circuit 13a bzw.13b of the two vertical active resonance excitation emitters 10a and 10b as shown in Figure a) but with a parallel resonant circuit consisting of the parallel inductance 46a and 46b and the parallel capacitance 23a and 23b in the second reactance circuit 21a and 21b for expanding the frequency bandwidth of the antenna.
  4. d) reactance circuit 13c of the rest of the passive vertical resonant radiator 4c, realized as in Figure b) but with a serial parallel resonant circuit consisting of the parallel inductance 46 and the parallel capacitance 45 in series with the capacitance 28 for expanding the frequency bandwidth of the antenna in accordance with the design of the reactance circuit 13a or 13b of the two vertical active resonance excitation emitters 4a and 4b as in c). The two active resonance excitation emitters are connected as in FIG.
  5. e) Representation of the frequency bands L1 and L2 for satellite navigation with the center frequencies fm1 and fm2 and the lower and upper limit frequencies fu1, fo1 and fu2, fo2, respectively. The frequency fm describes the center frequency between fu1 and fo2.
  6. f) reactance circuit 13a bzw.13b of the two vertical active resonance excitation emitters 10a and 10b as in Figure c) but with a further parallel resonant circuit 44a and 44b connected in series to the first capacitor 22a and 22b to the node 19 for separate optimization of the circuit for each one of the two frequency ranges L1 and L2 for the design of a dual-band antenna.
  7. g) reactance circuit 13c of the remaining passive vertical resonant radiator 4c, realized as in FIG. d), but with a further parallel resonant circuit 44 connected in series with the capacitor 28 in the reactance circuit 13c. The two active resonance excitation emitters are connected as in FIG. 1f to form a dual-band antenna.
  8. h) reactance circuit 13a bzw.13b of the two vertical active resonance excitation emitters 10a and 10b as shown in Figure f) but with an additional parallel capacitance 47, formed by the isolated counter electrode 34 projecting surface of the capacitance electrode 32a, 32b in the FIGS. 13c and 13d The auxiliary parallel capacitance 47 allows the extension of the range of impedance matching at the node 19a, 19b, as well as the resonance condition for the loop antenna 2. The resonance condition and the impedance matching are made by tuning the sizes of the capacitance electrodes 32a , 32b and the insulated counterelectrodes 34 are made to each other.
• Figure 6 :
  Adaptation ratios at the resonant frequency f0 as a function of the measure "t" for the division into the first capacitance 22a, 22b and the second capacitance 23a, 23b using the example of the ring line radiator 2 in FIG FIG. 4 ,
  1. a) Reflection factor at the Antennenanschlusstelle 5 as the input port T1 of a lossless distribution and phase shifter network 8 of the antenna 1 as a function of the pitch t. Adjustment is achieved with topt = 0.13.
  2. b) transformation factor at the resonance frequency f0 between the input resistance 43 with Z0 = 50 ohms of the phase rotation member 17 and the resonance resistance at f0 at the ring line radiating crosspoint 7 as a function of the pitch t.
  3. c) Relative power P / Pmax at f0 as a function of the pitch t.
• Figure 7 :
  Antenna according to the invention with resistance adaptation by capacitive division as in the example of FIG. 4 However, with a total of 6 azimuthal by 60 ° offset from each other distributed vertical resonant radiators 4a .. 4f, of which four are designed as vertical passive resonant radiators 9a..9d. The excitation 3 takes place by way of example at the two by 120 ° azimuthally offset from each other and by the difference angle ΔΦ = 120 ° excited vertical resonance excitation emitters 10a, 10b.
• Figure 8 :
  Antenna according to the invention as in FIG. 7 , but with a total of 8 azimuthal vertical resonance emitters 4a..4h distributed at 45 ° to each other. The excitation 3 takes place, for example, at the two azimuthally offset by 90 ° azimuthal and by the difference angle ΔΦ = 90 ° excited vertical resonance excitation radiators 10a, 10b.
• Figure 9 :
  Antenna according to the invention with rectangular executed ring line 14 with four vertical resonant radiators 4a to 4d in the region of the ring line corners. Two of the vertical resonance radiators are designed as vertical resonance excitation emitters 10a, 10b and the other two emitters as passive vertical resonance emitters 9a, 9b. Impedance matching to the low impedance (eg Z = 50 ohm) executed distribution and phase shifter network 8 is by capacitive subdivision, similar as in FIGS. 4 . 7 and 8th reached. The ring line 14 is preferably of a square design with identical ring conductor widths 15a, 15b and identical ring line sections 30a, 30b. However, both the ring conductor widths 15a, 15b and the ring line portions 30a, 30b may be chosen to be different from one another within limits.
• Fig. 10 :
  Rectangular ring line emitter 2 of an antenna 1 according to the invention without representation of the electromagnetic excitation 3 for explaining the resonant structure of the ring line emitter 2.
  1. a) geometric structure of the ring line radiator 2 with large differently selectable ring conductor widths 15a and 15b and the different selectable ring line sections 30a and 30b. The vertical resonance radiators 4a-4d are with their ring line coupling points 7a-7d in the wide region of the corners of the rectangular ring conductor fourteenth intended. Due to the current displacement, the current distribution is compressed towards the edge of the ring conductor 14. The relevant for the function in the excitation of the loop emitter 2 current of the line shaft flows accordingly - Characterized by the drawn as a broken line line of gravity of the power distribution 24 - even with large conductor widths 15a, 15b pushed more towards the edge. This is especially true for very large annular widths 15a, 15b to the complete closure of the inner opening towards the center, which is virtually de-energized when the loop 14 is realized as a closed conductive surface.
  2. b) according to the current represented by the centroid of the current density distribution 24, the resonant structure of the loop emitter 2 can be generally represented by a roughly approximated equivalent circuit. The individual ring line sections 30a-30d are each represented by the inductive and capacitive action of the associated section of the ring line 14, taking into account the capacitive effect of the reactance circuit 13 as concentrated inductive elements (Ln) and capacitive elements (Cn). Each n-th ring line section 30a-30b is represented by a π-structure according to FIG. C), consisting of a longitudinal inductance 2 * Ln and in each case a transverse capacitance Cn at both ends thereof. The radiation attenuation of each horizontally oriented loop portion 30a-30b is included by the attenuation factor d of the concentrated inductance. The juxtaposition of adjacent ring line sections 30a-30d takes place in each case via a common vertical resonant radiator 4a..4d with contraction of the transverse capacitances Cn of the adjacent loop line sections, as shown in FIG. The slight inductive effect of the vertical resonance radiators is neglected in this basic consideration due to the small antenna height h. According to the invention advantageous conditions for the ring line emitter 2 are then reached for the resonant frequency f0 = ωo / 2π, if in each case all characteristic impedance $\mathit{ZLn}=\sqrt{\frac{\mathit{ln}}{\mathit{cn}}}$
     
     and all of them $\sqrt{\mathit{ln}\square \mathit{cn}}={}^1{/}_{\mathit{<figure-callout\; id="1"\; label="\omega on"\; filenames="imgf0006.png,imgf0013.png"\; state="\{\{state\}\}">\omega on</figure-callout>}}.$
     
     which represent the electrical length of the loop section for all n are the same size.
     In rotationally symmetrical ring line emitters 2, this is always the case. Otherwise, this requirement can be achieved, for example, in the case of a rectangular structure of the ring conduit radiator 2 by individual design of the ring conductor widths 15a-15d in the individual ring line sections 30a-30b.
• Figure 11 :
  1. a) Design of the distribution and phase shifter network 8 - for example, as in the FIGS. 7 and 8th - However, as matched to the resonant frequency f0 hybrid ring 3, through which both the power division and the phase shift occurs. When fed to the gate T1 of the loop emitter 2 is excited via the gates T2 and T3 with the phase difference of 90 °. The termination of gate T4 with the ohmic terminator 40 causes in the case of deviation of the frequency of the resonant frequency f0 wideband the partial absorption of the power of the unwanted polarization when fed to the gate T1.
  2. b) Idealized scattering matrix for describing the well-known fundamental wave behavior of a hybrid ring 38 in Figure a) at the gates T1 to T4. The important decoupling of the gates T1 and T4 from each other is indicated in bold by the scattering parameters S14 = 0, S41 = 0 in the matrix.
  3. c) Design of the distribution and phase shifter network 8 - for example as in the FIGS. 7 and 8th - However, as adapted to the resonant frequency f0 Wilkinson divider 16 with downstream λ / 4 line phase shifter 17 to a gate T3 * to generate the phase difference of 90 °. The ohmic balancing resistor 40 partially absorbs the currents that cause the unwanted polarization on the loop emitter 2 in asymmetric loading of the Wilkinson divider 16 in deviation from the resonant frequency f0.
• Figure 12 :
  Antenna according to the invention z. B. as in FIG. 9 with rectangular shaped loop 14. The capacitances 22a, 22b and 28 are formed in such a way that the vertical Resonant emitters 4a - 4d are formed at its lower end to individually shaped surface capacitance electrodes 32a, 32b, 32c, 32d. By means of an intermediate layer between these and the electrically conductive base surface 6 embodied as an electrically conductive printed circuit board, the capacitors 28 for coupling the passive vertical resonance radiators 4c, 4d to the electrically conductive base 6 are designed. For capacitive coupling of the active vertical resonance radiators 10a, 10b to the ports T2 and T3 of the distribution and phase shifting network 8, this connection is designed as a respective planar counter electrode 34 isolated from the conductive layer. The counterelectrodes 34 can thus be used as connection points 19 of the antenna in FIG FIG. 9 and can be designed as connection points for the ports T2 and T3 of the distribution and phase shifting network 8, as in FIG FIG. 13 indicated, serve.
• Figure 13 :
  Antenna 1 similar to in FIG. 11 However, the dielectric effect of the dielectric plate 33 is realized by an air gap. Typical dimensions of a loop antenna 2 for the frequency range L1 are for square antennas a dimension of 30mm to 40mm and for the height h = 8mm. In order to match the resistance of the distribution and phase-shifting network 8, which is preferably implemented at a low impedance level (Z0), the capacitive subdivision is as in FIG FIG. 9 intended.
  1. a) The exemplary associated reactance circuits 13a, 13b and 13c are in the FIGS. 5a and 5b shown. The capacitance 28 of the capacitance electrode 32c, 32d against the electrically conductive base area 6 and the capacitance 22a, 22b against the isolated counterelectrode 34a, 34b is in each case approximately 0.3pF. The gates T2 and T3 of the distribution and phase shifter network 8 with their real input resistance 43 are each connected to a counter electrode 34a, 34b as the connection point 19a, 19b. The capacitive division for impedance matching is given by the capacitance 22a, 22b and the capacitance 23a, 23b, respectively. This introduced between the node 19 a, 19 b and the ground point 11 capacitance 23 a, 23 b is mounted on the back of the coated circuit board 35 as an SMD component. For this purpose, isolated pads 29 are designed as contact bases on the back of the circuit board, which each have a through-connection 26 with the Counter electrode 34 on the one hand and on the other hand, for example via the capacitance 23a, 23b (sh. Fig. 4 u. 5a) are connected to ground.
  2. b) the reactance circuits 13a, 13b of the two resonance excitation emitters 10a, 10b are as shown in Figure a) according to FIG. 5a or as in 5c designed. When designing after FIG. 5c is the introduced on the back of the coated circuit board 35 capacitance 23a, 23b, an inductance 46a, 46b connected in parallel as an SMD component. In contrast to FIG. 1 a), counterelectrodes 34 isolated from the capacitive electrodes 32 c, 32 d on the coated printed circuit board 35 are used to form the reactance circuits of the two passive resonant radiators 9 a, 9 b. Starting from the counter electrode 34 as a contact support 35, the parallel inductance 46 and the parallel capacitance 45 are connected to the ground point 11 on the circuit board 35 on the back of the circuit board 35, so that the reactance circuit 13 in FIG. 5d is realized.
     In embodiment of all reactance circuits 13 of all vertical resonant radiators, each with a capacitance electrode 32 and an opposing insulated counter electrode 34, all circuits in the FIGS. 5c and 5f for the resonance excitation emitters 10a, 10b by use of SMD components on the back of the circuit board 35 can be realized. Similarly, the reactance circuits for the passive resonant radiators 9a, 9b by wiring the isolated counter electrodes 34 with SMD components on the back of the circuit board 35 according to those in the FIGS. 5d and 5g specified circuits can be realized.
  3. c) As under FIG. 5h The figure shows the large area coverage of the isolated counter electrode 34a, 34b with the capacitance electrode 32a, 32b of the resonance excitation emitters 10a, 10b. The planar projection which exists in the vertical projection of the capacitance electrode 32a, 32b with respect to the isolated counterelectrode 34 forms, together with the electrically conductive layer of the coated printed circuit board 35, the auxiliary parallel capacitance 47 Reactance resistor circuit of the two passive resonant radiators 9a, 9b are designed as in Figure a).
  4. d) the reactance circuits 13 of the two resonance excitation emitters 10a, 10b are designed with an additional parallel capacitance 47 as in FIG. The exemplary possible reactance circuits 13 of the two passive resonant radiators 9a, 9b can be designed as described in FIG.
• Figure 14 :
  The picture shows the upper side of the printed circuit board 35 of an antenna 1 according to the invention, on which the electric ring line emitter 2 is placed. For the broadband design of an antenna 1 according to the invention - which, for example, both frequency ranges L1 and L2 with a lying between the two frequency ranges center frequency fm (sh. FIG. 5e ) - is the reactance circuit 13, as in the FIGS. 5c or 5d designed. For this, for the reactance circuit in the two active vertical resonance excitation radiators 10a, 10b, respectively, the reactance circuit 13a, 13b in FIG FIG. 5c and for the two passive vertical resonance radiators 9a, 9b, the reactance circuit 13c in FIG FIG. 5d intended. This is the example in FIG. 13a achieved in that for each of the capacitance electrodes 32 each have a counter electrode 34 is present and that caused by the capacitance electrodes 32 capacitance 22a, 22b and 28 on all vertical resonant radiators 4a - 4d a parallel connection of a parallel capacitance 23a, 23b and 45 and a Parallel inductance 46a, 46b, and 46 - shown as SMD components - between the counter electrode 34 and the electrically conductive base 6 is connected in series. FIG. 13a shows the upper side of the printed circuit board 35 with vias 26 on the counter electrodes 34 under the capacitance electrodes 32 FIG. 14b the underside of the circuit board 35 is shown with the pads 29, which are connected to the isolated counter electrodes 34 via the vias 26,. The dummy elements are as SMD components according to the FIGS. 5c and 5d appropriate. In addition, the two connection points 19 are connected to the respectively connected series circuit of the capacitance 28a, 28b and the inductance 27a, 27b as parts of in FIG. 3a contained matching circuit 18 of the distribution and phase shifter network 8 to form the gates T2 and T3.
• Fig. 15
  Shows an antenna according to the invention with four azimuthally on the circumference of the loop 14 equally distributed resonance excitation emitters 10a to 10d. All radiators are excited in accordance with a rotating wave, each with 90 ° phase difference between adjacent radiators. For this purpose, a first and a second distribution and phase shifter network (8) with phase angle ΔΦ = 90 ° are present, wherein the first distribution and phase shifter network (8) coupled with its gates T2, T3 with a first pair of the resonance excitation emitters 10a, 10b and the second distribution and phase shifter network (8) is coupled with its ports T2a, T3a to a first pair of the second pair of excitation emitters 10c, 10d. For exciting a traveling wave, opposing resonance excitation radiators 10a, 10c and 10b, 10d are energized respectively with a phase difference of Δφ = 180 °. For this purpose there is another distribution and phase shifting network 31, 39 with a phase shift of ΔΦ = 180 °, which with its gate T2c to the gate T1 of the first distribution and phase shifting network 8, 36 and with its gate T3c to the gate T1b further Distribution and phase shifter network 8, 38b is connected. Again, according to the invention preferably each a hybrid ring 38 is used. The 180 ° phase network 31 may be exemplified as a Wilkinson divider 39 having a λ / 2-long delay line 17 (see FIG. FIG. 11c ) are used. All connections are selected in such a way that sets the conductive wave in the desired direction of rotation on the loop. With this arrangement, the particular advantage of the azimuthally symmetrical excitation is connected, so that the adjustment of the arrangement can be carried out particularly problem-free. As a limitation, however, the increased effort is noted.

For the design of a two-band multiband antenna according to the invention - for example for the frequency ranges L1 and L2 - the reactance circuit 13 is designed multi-frequency in each case in such a way that resonance of the ring line radiator 2 in the separate frequency bands L1 and L2 separately and correspondingly as frequency f01 and given as frequency f02. For this purpose, starting from the pads 29 in FIG. 14b the back of the coated circuit board 35 in such a way that the reactance circuits 13a, 13b, and 13c are realized.

In the following, the invention will be described again in connection with its advantageous embodiments.

In a basic form, an antenna 1 according to the invention comprises a ring line emitter 2 already described above, as described, for example, in US Pat FIG. 2a for a rotationally symmetrical arrangement with three along the electrically short ring conductor 14 equidistantly distributed vertical resonant radiators 4a, 4b, 4c is shown. From the ring conductor width 15 and the length of a loop portion 30a, 30b, etc. results in an inductive effect, which together with the capacitive reactance circuit 13 in each of the vertical resonant radiator causes a resonance behavior with a resonant angular frequency ω0 = 2π * f0, which is characterized by the frequency characteristic of the antenna impedance 37 with the impedance maximum 42 'in FIG. 2b expresses.

According to the invention, this resonant structure is capable of self resonant frequency f0, with corresponding excitation via two vertical resonant radiators 4a, 4b - referred to herein as vertical resonant excitation radiators 10a, 10b - with signals corresponding to the azimuthal position of vertical resonant radiators 4a, 4b. for example like in FIG. 1 represented - to convey a current current wave in exactly one direction. The structure of the ring line radiator 2 in FIG. 2a itself thus does not yet have a feature for the preference of one of the two possible directions of movement of the current wave. The direction of travel is determined solely by connecting the distribution and phase shifting network 8 according to the sign of the phase angle ΔΦ, as in FIG. 1 , set. The phase difference .DELTA..PHI. Between these signals is therefore selected in such a way that it corresponds to the phase difference .DELTA..PHI. Of a current wave running on the loop line between the ring line crosspoints 7a and 7b of the two resonance excitation emitters 10a and 10b at the resonant frequency f0 of the ring line radiator 2. The direction of rotation of the current wave is thus due to the sign of the phase difference ΔΦ of the exciter signals in conjunction with the position of the two loop coupling points 7a, 7b given. The direction of travel is thus exclusively by connecting the distribution and phase shifter network 8 according to the sign of the phase angle ΔΦ, as in FIG. 1 , set.

Naturally, the condition for the excitation of a running direction can be fulfilled exactly only at the resonance frequency f0 and the proportion of the undesired running direction and thus the cross polarization increase with increasing deviation from this frequency. As a particular advantage of the invention, however, results in the very large bandwidth for a sufficiently large suppression of Kreuzpolarisation from the claim according to the invention for a real input resistance 43 of both gates T2, T3 of the distribution and phase shifter network 8, wherein the resonant frequency f0 of the loop emitter 2 by the excitation 3 is not changed at the resonance frequency f0. A reactive power exchange between the ring line emitter 2 and the excitation 3 thus does not take place at the resonance frequency f0. Particularly high bandwidth of the suppression of the cross polarization is achieved if - as in FIG. 1 the ports T2 and T3 are directly connected to the vertical resonance excitation emitters 10a, 10b so as to be effective at the corresponding loop coupling points 7a, 7b.

According to the invention, it is therefore provided to tune the ring line emitter 2 and the phase shift by the angle ΔΦ in the distribution and phase shifter network 8 in such a way that this angle at the resonant frequency f0 of the electrical length of the loop portion 30 as an angle measure the proportion of the electrical length 2π corresponds to the ring line 14 between the two vertical resonance excitation emitters 10a, 10b, to which the gates T2 and T3 are connected. In this case, between the vertical resonance excitation radiators 9a, 9b on the ring conductor 14 more, but passive vertical resonant radiator 4 - as in the FIGS. 7 and 8th shown - be arranged.

This results in total of N> 3 vertical resonant radiators 4a, 4b, 4c, 4d and thus the same number of loop portions 30a, 30b, 30c, 30d and over the ring line 14 equally distributed crosspoints 7 of the present invention set phase angle of the distribution and phase shifter network 8 to ΔΦ = n * 2π / N <180 °, if between the crosspoints 7 of the two vertical resonance excitation emitters 10a, 10b ring line sections 30 are selected ,

For rotationally symmetric arrangements with a high number of N ring line sections in total, it is advantageously possible to achieve a large frequency bandwidth for the suppression of the cross polarization.

For special applications, such as for the superposition of circularly polarized radiation with azimuthal phase distribution of 2π and a circularly polarized radiation with azimuthal phase distribution of 4π in an antenna diversity system with two antennas, the invention can advantageously also a ring line emitter 2 with a electrical length of 4π be applied as follows. In a simple special case with a polygonal ring line emitter 2 with N = 8 ring line sections 30 azimuthally distributed over the electrical length 4π, two mutually adjacent vertical resonance excitation emitters 10a, 10b (n = 1) are connected via a distribution and phase shifting network 8 with ΔΦ = π / 2 excited. By analogy with this, with N = 16 vertical resonance radiators and n = 2 ring line sections between the two vertical resonance excitation radiators 10a, 10b, the phase angle ΔΦ = n * 4π / N should also be set ΔΦ = π / 2. In an expedient and space-saving manner, two ring line emitters 2 with different phase distribution can be arranged concentrically to one another in such a way that the emitter with azimuthal 2π phase distribution is surrounded by the emitter with 4π phase distribution.

In an advantageous manner, the excitation of the ring line 14 with the electrical length 2π of the tuned to the resonant frequency f0 ring line radiator 2 in FIG. 1 in such a way that the gates T2, T3 have a real high-impedance input resistance 43 and are connected directly to the ring line coupling points 7a, 7b via the two vertical resonance exciter radiators 10a, 10b. In this case, both the distribution network 16 and the Phase shifter 17 high impedance designed according to the high resonance resistance.

In an advantageous embodiment of the invention are in FIG. 3 the distribution network 16 and the phase shifter 17 low impedance, ie designed for the impedance level Z0 (eg Z0 = 50 ohms). For transformation into the high impedance level of the ring line radiator 2, a matching network 18 is connected downstream of the two output branches. For this purpose, for example, in each case the in FIG. 3a illustrated λ / 4 transformation line 12a, 12b with the series resonant circuit of the inductance 27a, 27b and the capacitance 28a, 28b.

In an advantageous embodiment of the invention, the impedance match between the low impedance - designed at the impedance level Z0 - designed gates T 2, T3 of the distribution and phase shifter network 8 and at the ring line crosspoints 7a, 7b each present high impedance resonance resistance of the ring line radiator 2 - FIG. 4 represented by serial division of the capacitive reactance circuit 13a, 13b with design of a node 19a, 19b therebetween for connection of the two ports T2, T3 with their respective real input impedance 43 at the impedance level Z0. The frequency bandwidth of cross-polarization suppression in circular polarization antennas depends to a large extent on the bandwidth of the antenna and thus on its electrical size. In the case of the electrically small ring-shaped line radiators 2 considered here having dimensions, as described for example in connection with FIG FIG. 13 are described, it is possible with the frequently prescribed requirement of the cross-polarization distance of 15db and above to detect each of the frequency bands L1 and L2 separately.

For this purpose, the invention is in the example of the FIG. 4 shown subdivision of the reactance circuit 13a, 13b of the two vertical resonance excitation emitters 10a, 10b, which also in FIG. 5a is shown. The illustrated subdivision is given purely capacitively by the first capacitance 22a, 22b and the second capacitance 23a, 23b. In this case, it is important according to the invention that in the vertical resonance excitation emitters 10a, 10b effective resulting capacitance each of those in the other passive vertical resonant radiators 9a, 9b - represented by the capacitance 28 in the FIGS. 4 . 7 . 8th and in FIG. 5b - corresponds and thus the resonance of the ring line radiator 2 at the resonant frequency f0 is maintained by the capacitive subdivision. An exchange of reactive power between the ring line radiator 2 and the electromagnetic excitation 3 at the node 19 is excluded at the resonant frequency f0. However, this feature no longer fully complies with increasing frequency deviation from the resonant frequency f0 due to the frequency dependence of the loop antenna 2 and the frequency dependence of the distribution and phase shifting network 8 with respect to impedance matching and phase shift. Advantageously, however, the inventive adjustment described with increasing frequency deviation leads to a comparatively particularly small increase in the cross polarization. The antenna according to the invention is thus advantageous broadband with respect to the suppression of the cross polarization.

On the other hand, with a small antenna described here, it is hardly possible to cover the entire frequency band of both frequency ranges L1, and L2 between the frequencies fu2 and fo1 (see FIG. FIG. 5e ) with an above-described cover antenna without additional measures.

According to the invention allows the design of the reactance circuit 13a, 13b - shown in FIG. 5c in the two vertical resonance excitation radiators 10a, 10b and the reactance circuit 13c in FIG FIG. 5d a single-band antenna which captures both frequency bands L1 and L2.
This is inventively achieved in that both in the second reactance circuits 21a, 21b (sh. FIG. 5c ) of the two vertical resonance excitation emitters 10a, 10b by parallel connection of the parallel inductance 46a, 46b to the existing second capacitance 23a, 23b as well as in the reactance circuits 13c (see FIG. FIG. 5d ) of the passive vertical resonant radiator 9c are each a parallel resonant circuit with the inductor 46 and Capacity 45 is added. In this case, the resonance frequency of all parallel oscillation circuits is in each case in the frequency range between the upper frequency band L1 and the lower frequency band L2 (eg, fm in FIG FIG. 5e ) in such a way that the reactance of the resonant circuits in the frequency band L1 is capacitive and in the frequency band L2 is inductive. However, the reactance of all reactance circuits 13a, 13b, 13c as a whole is nevertheless capacitive in both frequency bands L1 and L2.

An advantageous development of the invention according to the above design of the reactance circuits takes place in the design of a dual-band antenna in such a way that both the first capacitance 22a, 22b of the first reactance circuit 13a 13b of the vertical resonance excitation emitters 10a, 10b and the capacitance 28 of the reactance circuit 13c of the passive vertical resonant radiator 9c in each case a further parallel resonant circuit 44, 44a, 44b is connected in series. Thereby, it is advantageously possible, for example, for both the reactance X1a, X1b of the first reactance circuit 20a, 20b and also the reactance X2a, X2b of the second reactance circuit 21a, 21b, and thus the reactance X of the reactance circuit 13a, 13b in both frequency bands L2 and L2 is capacitive in each case and favorable values for the pitch t for the capacitive division can also be selected separately for both frequency bands. The respective reactance circuits 13a, 13b and the reactance circuit 13c are incorporated in FIGS Figures 5f and 5g shown. Proper sizing of all the dummy elements in the reactance circuit 13a, 13b makes it possible to achieve the capacitive division described above by selecting the resonant frequency of the parallel resonant circuits with respect to the center frequencies fm1 and fm2 in view of the above-described optimization with respect to t -topt.

In a particularly advantageous embodiment of the invention, the broadband suppression of the unwanted polarization LHCP a designated RHCP antenna with a hybrid ring 38 as a distribution and phase shifter network 8 for excitation of the square ring line radiator 2 in FIG. 13 reached. In a conventional, known form, a hybrid ring 38 consists of λ / 4-long pipe sections, as in FIG. 11a shown. It is also known that the pipe sections can be simulated by concentrated dummy elements. In principle, a hybrid ring 38 can be manufactured for different impedance levels. However, production is particularly economical for the low impedance level Z0.

However, a hybrid ring 38 can be perfectly balanced only for a particular frequency - generally about the center frequency of a frequency band (fm1, fm2) - and by the idealized scattering matrix with respect to its ports T1 to T4 in FIG FIG. 11b to be discribed. At deviating frequencies arises at the antenna in FIG. 13 naturally when excited at the gate T1 and upon excitation of the ring line radiator 2 via the ports T2 and T3 in addition to the desired radiation in RHCP mode proportionally unwanted radiation in the opposite direction, ie the LHCP mode. Added to this is the increasing reactive frequency of the resonance frequency f0 increasing reactive component of the antenna impedance, which also increases the LHCP mode. An optimal interaction between the hybrid ring 38 and the ring line emitter 2 with respect to the suppression of the cross polarization is inventively achieved when the hybrid ring 38 is matched with respect to its real input resistance at the gates T2 and T3 and its required phase of 90 ° to the frequency f0, which also forms the resonance frequency f0 of the corresponding ring line radiator 2. As a special feature of a hybrid ring 38 causes the wiring of the gate T4 decoupled from the gate T1 with a resistance of size Z0 according to the invention, that the unwanted LHCP content in the radiation even at frequency deviation of the frequency f0, for which both the ring line emitters 2 and the hybrid ring 38 is tuned, is largely absorbed.

In an advantageous further development of the invention, the distribution and phase shifting network 8 of the Wilkinson divider 39 acts as a distribution network 16 with a downstream phase-shifting element 17 FIG. 11 for use. The signal path, starting from the gate T1, branches into two λ / 4-long lines whose live ends with the ohmic balancing resistor 41 as Compensation resistance are connected together. The gate T3 *, for example, a λ / 4-long line is connected downstream as a phase shifter 17 for a phase rotation of 90 °. This line can also be simulated by concentrated components. The system is only adjusted if the doors T2 and T3 have a resistance-resistant finish. Similar to the use of the hybrid ring 38 described above, this only applies to the resonance frequency f0. With increasing deviation from this frequency, the symmetry of the Wilkinson divider 39 is disturbed. According to the invention, the resulting undesired LHCP content via the ohmic balancing resistor 41 is equally strongly attenuated.

To meet particularly high demands on the suppression of the cross polarization in the two frequency bands L1 and L2, it may be provided to use a separate antenna 1 according to the invention for each frequency band. In an extremely advantageous continuation of the invention, the two ring line radiators 2 are arranged concentrically in an antenna arrangement with the proviso that the ring line radiator 2 for the higher frequency in the band L2 is surrounded by the larger ring line radiator for the lower frequency in the band L1. Of course, this is the increased cost of two separate and matched to the respective frequency band distribution and phase shifter network 8 to accept. Utilizing the frequency spacing between different frequency bands, a plurality of antennas 1 according to the invention assigned to one frequency band can be formed in this way, the ring line emitters 2 being arranged concentrically with each other in such a way that the ring line emitter 2 is arranged innermost for the highest frequency band in the frequency and each further ring line emitter 2 surrounds a ring line emitter 2 for the next higher frequency band in frequency. Such an arrangement has the particular advantage that in addition to the two frequency bands L2 and L1, the lower frequency band L5, which also serves for satellite navigation, can be detected hereby. From the summary and the parallel evaluation of all signals from these frequency bands in the navigation receiver, the quality of the location results can be extremely improved even under difficult reception conditions.

Such a concentric arrangement of the ring line emitters 2 moreover has the advantage that a linear, vertically polarized antenna for other radio services can be arranged in the center of the arrangement without impairing the properties of the ring line emitters 2.

List of terms

• Antenna 1
• Ring line radiator 2
• electromagnetic excitation 3
• vertical resonance radiators 4, 4a, 4b, 4c, ...
• Antenna connection point 5
• Conductive base 6
• Ring line coupling points 7, 7a, .. 7d, ...
• Distribution and Phase Shift Network 8
• vertical passive resonant radiators 9, 9a, 9b
• vertical resonance excitation emitters 10a, 10b, ..
• Ground connection point 11
• λ / 4 transformation line 12
• Reactive resistance circuit 13, 13a, ... 13c, ...
• Ring line 14
• Ring conductor width 15, 15a, .., 15c ,.
• Distribution network 16
• Phase turn member 17
• Matching network 18
• Junction point 19
• first reactance circuit 20a, 20b
• second reactance circuit 21a, 21b
• first capacity 22a, 22b
• second capacitance 23a, 23b
• Heavy line of the current density distribution 24
• Serial parallel resonant circuit 25
• Through-connection 26
• Inductance 27a, 27b
• Capacity 28a, 28b
• Pad 29
• Ring line section 30a, 30b, 30c, 30d
• 180 ° phase shift network 31
• Capacitance electrode 32a, 32b, 32c, 32d,
• dielectric plate 33
• isolated counter electrode 34
• Coated printed circuit board 35
• Support structure 36
• Antenna impedance 37
• Hybrid ring 38
• Wilkinson divider 39
• Ohmic terminator 40
• ohmic balancing resistor 41
• Measuring section 42
• Impedance maximum 42 '
• real input resistance 43
• another parallel resonant circuit 44
• Parallel capacity 45
• Parallel inductance 46, 46a, 46b
• Additional parallel capacity 47
• Elongated length of the ring line radiator L
• Reactance X, X1a, X1b, X2a, X2b, Xc
• Height h
• Goals T1, T2, T3, T4
• Characteristic impedance ZL
• Center Z

Claims (15)

Antenna (1) for receiving circularly polarized satellite radio signals, comprising at least one conductor loop arranged above a conductive base (6), with an arrangement for electromagnetic excitation (3) of the conductor loop connected to an antenna connection point (5), comprising the following features: - The conductor loop is designed as a ring line radiator (2) by a polygonal or circular closed loop line (14) running at a height h <0.15 of the free space wavelength λ over the conductive base surface (6), - The ring line radiator (2) forms a resonant structure and by the electromagnetic excitation (3) excitable in such a way that adjusts the current distribution of a current line wave in a single direction of rotation on the ring line (14), the phase difference over a cycle is just 2π . - There are distributed at the circumference of the ring line radiator (2), at ring line crosspoints (7, 7a, 7b, 7c, 7d) with the ring line radiator (2) galvanically coupled and the conductive base (6) extending vertical radiator (4) available . there is at least one distribution and phase-shifting network (8) with a gate T1 coupled on the input side with the antenna connection point (5) and with two ports T2, T3 whose transmission coefficients S ₁₂ and S ₁₃ differ from each other by a phase angle ΔΦ, comprising the following features: the circumference (L) of the ring line (14) is shorter than the free space wavelength λ, there are at least three vertical radiators (4, 4a, 4b, 4c, ...) which are coupled via reactance circuits (13) with capacitive reactance X to the base area (6), by which a resonance is produced, the input resistance of each of the ports T2, T3 of the distribution and phase shifter network (8) for the electromagnetic excitation of the line shaft on the ring line (14) is in each case a real resistance, - Each of the ports T2, T3 is coupled to a respective ring line cross-point (7, 7a, 7b). Antenna (1) according to claim 1,
characterized in that
the distribution and phase shift network (8) are associated with two of the vertical resonance radiators (4a, 4b) as vertical resonance excitation emitters (10a, 10b) in such a way that each of the gates T2, T3 of the distribution and phase shifter network (8) via one of the two vertical resonance excitation emitters (10a, 10b) is coupled to its ring line coupling point (7a, 7b) and the other vertical resonance emitters (4c, 4d, ..) are active as passive vertical resonance emitters (9a, 9b, etc.). (Fig. 1) Antenna (1) according to claim 1 or 2,
characterized in that
each of the ports T2, T3 is respectively directly connected via one of the vertical resonance excitation emitters (10a, 10b) to its ring line crosspoint (7a, 7b) and the input resistance of the gates T2, T3 is high impedance real in that between this and the high-impedance resonance resistance of the ring line radiator (2) resistance matching exists. (Fig. 2) Antenna (1) according to at least one of claims 1 to 3,
characterized in that
in the distribution and phase shifter network (8) a distribution network (16) and a downstream in one of its output branches phase shifter (17) are provided, which are both designed for a low impedance Z _{0 level} and in each of the two output branches a lossless matching circuit (18) Transformation of the low impedance level Z ₀ in the high impedance real input resistance of the gates T2, T3 to adapt to the high impedance resonance resistance of the ring line radiator (2) is present. (Fig. 3) Antenna (1) according to at least one of claims 1 to 4,
characterized in that
the reactance circuit (13) with which each of the passive vertical resonance radiators (9, 9a, 9b) and also the vertical resonance excitation radiators (10a, 10b) is coupled to the base (6) is formed by a respective capacitance (28). (Fig. 3) Antenna (1) according to at least one of claims 1 to 5,
characterized in that
the reactance circuit (13) with which the vertical resonance excitation emitters (10a, 10b) are coupled to the base (6) respectively as a series circuit of a first reactance circuit (20a, 20b) and a second reactance circuit (21a, 21b) this connecting point of connection (19) is formed, with which in each case one of the ports T2, T3 is connected. (Fig. 4) Antenna (1) according to at least one of claims 1 to 6,
characterized in that
the reactance circuit (13) to which the vertical resonance excitation emitters (10a, 10b) are coupled to the base (6), and the first reactance circuit (20a, 20b) of each vertical resonance excitation emitter (10a, 10b) is replaced by a first capacitance (22a 22b) and the second reactance resistor circuit (21a, 21b) is formed by a second capacitance (23a, 23b) and the size and the ratio of the first and the second capacitance are each selected such that between the high-impedance resonance resistance of the loop emitter ( 2) and a lower impedance real input resistance of the gates T2, T3 each resistance adjustment is given. (Fig. 4) Antenna (1) according to at least one of claims 1 to 7,
characterized in that
the reactance circuits (13c, etc.) with which passive vertical resonant radiators (9c, etc.) are coupled to the electrically conductive base (6), each of the series connection of a capacitor (28) and a Parallel vibration circuit - consisting of a parallel capacitor (45) and a parallel inductance (46) - are formed and the first reactance circuit (20a, 20b) of the vertical resonance excitation emitter (10a, 10b) each as a first capacitor (22a, 22b) is formed and whose second reactance circuit (21a, 21b) is designed in each case as a parallel resonant circuit consisting of a second capacitance (23a, 23b) and a parallel inductance (46a, 46b) and the resonant frequency of all parallel oscillation circuits in the frequency range between the upper frequency band L1 and the lower frequency band L2 is chosen in such a way that the reactance of the resonant circuits in the frequency band L1 capacitive and in the frequency band L2 is inductive. (FIGS. 5c, 5d, 5e) Antenna (1) according to claim 8,
characterized in that
however, between the capacitance (28) and the parallel oscillation circuit of the parallel capacitance (45) and the parallel inductance (46) of the reactance circuits (13c, etc.) of the passive vertical resonant radiator (9c, etc.) in each case a further parallel resonant circuit (44) connected in series is and the first capacitance (22a, 22b) in the first reactance circuit (20a, 20b) of the vertical resonance excitation emitters (10a, 10b) to the connection point (19a, 19b) is also followed by a further parallel resonant circuit (44a, 44b) and the parallel oscillation circuits in the first reactance circuit (20a, 20b) and the second reactance circuit (21a, 21b) of the vertical resonance excitation radiators (10a, 10b) are tuned such that the reactance (Xa, Xb) in both frequency bands L1, L2 is capacitive and their relationship to each other according to the optimal measure topt for adaptation and the reactance circuits ( 13c, etc.) of the passive vertical resonant radiator (9c, etc.) and the vertical resonant exciter radiators (10a, 10b) are matched with respect to their frequency response. Antenna (1) according to one of claims 7 to 9,
characterized in that
the first capacitance (22a, 22b) of the first reactance circuit (20a, 20b) of the vertical resonance excitation emitters (10a, 10b) and the capacitances (28) in the Reactance circuits (13c, etc.) of the passive vertical resonant radiators (9c, etc.) are formed in such a way that all vertical resonant radiators (4a, 4b, 4c, etc.) are provided at their lower end to form individualized capacitive capacitive electrodes (32a, 32b, 32c, 32d) that the capacitances (22a, 22b, 28) by the interposition of a dielectric plate (33) between the two-dimensional capacitance electrodes (32a, 32b, 32c, 32d) and designed as electrically conductive coated circuit board (35) electrically conductive base (6) for coupling the passive vertical resonant radiator (9c, 9d, etc.) are designed to the electrically conductive base (6), and that for capacitive coupling of the vertical resonance excitation emitter (10a, 10b) on the electrically conductive base (6) each one of the conductive layer of the coated printed circuit board (35) isolated, planar counter electrode (34) for connecting the second reactance resistor circuit (2 1a, 21b) is designed. (FIGS. 13a, b, 5) Antenna (1) according to at least one of claims 1 to 10,
characterized in that
a total of three vertical resonant radiators (4a, 4b) with azimuthally equally distributed loop coupling points (7) are present, of which two as vertical resonance excitation emitters (10a, 10b) via the gates T2, T3 of a distribution and phase shifter network (8) excited are whose phase angle ΔΦ = 120 degrees. (Figs. 1, 3, 4) Antenna (1) according to at least one of claims 1 to 11,
characterized in that
the ring conduit (14) is square in shape with four loop coupling points at the corners and there are four vertical resonance radiators (4a, 4b, 4c, 4d) with azimuthally equal length loop portions (30a, 30b), two of which are azimuthally consecutive as vertical resonance Excitation radiators (10a, 10b) are energized via the ports T2, T3 of a distribution and phase shift network (8) whose phase angle ΔΦ = 90 degrees. Antenna (1) according to at least one of claims 1 to 12,
characterized in that
the distribution and phase shift network (8) for a phase shift the phase angle ΔΦ = 90 ° includes a hybrid ring (38) which next to the two ports T2 and T3 a fourth, with a resistive terminator (40) completed Tor T4 and which is decoupled at Wellendenstandsrichtigem completion of the gates T2 and T3 from the gate T1 , (Figures 11a, 11b) Antenna (1) according to at least one of claims 1 to 13,
characterized in that
the lossless matching circuit (18) for transforming the low impedance level Z0 into the high resistance real input resistance (43) in both output branches (T2, T3) of the distribution and phase shift network (8) each substantially by a lambda / 4 transformation line (12) a series circuit of an inductance (27a, 27b) and a capacitance (28a, 28b) for fine adjustment of the high-impedance real input resistance (43) of the gates T2 and T3 is formed. (Figure 3a) Antenna arrangement in which a plurality of respective frequency band associated antennas are provided according to at least one of the preceding claims, the ring line emitters (2) are arranged concentrically to each other in such a way that the ring line emitter (2) for the highest frequency in the frequency band is arranged innernst and each further ring line emitter (2) surrounds a ring line emitter for the next higher frequency band.

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