DE112017004788T5 - ANTENNA DEVICE AND ELECTRONIC EQUIPMENT - Google Patents

Abstract

An antenna device (101) comprises: a radiating element (10); a coupling circuit (30) and a nonradiative resonant circuit (20). The coupling circuit (30) comprises a first coupling element (31) and a second coupling element (32), wherein the first coupling element (31) between a supply circuit (1) and the radiation element (10) is connected, wherein the second coupling element (32) the first coupling element (31) is coupled. One end of the second coupling element (32) is grounded and the other end thereof is connected to the nonradiative resonant circuit (20). A frequency characteristic of a return loss of the radiating element (10) seen from the feeding circuit (1) is set by a resonance frequency characteristic of the non-radiating resonant circuit (20). With this design, an interference problem with respect to the radiation of two radiation elements is achieved.

Description

Technical area

The present invention relates to an antenna device corresponding to a wide frequency band and to electronic equipment comprising the antenna device.

Background Art

To widen a frequency band or to correspond to multiple frequency bands, an antenna device is used with two radiating elements which are directly or indirectly coupled to one another. In addition, Patent Document 1 represents an antenna device having two radiating elements and a coupling degree adjusting circuit that controls the power supply to the two radiating elements.

List of quoted publications

Patent document

Patent Document 1: International Publication No. 2012/153690

Brief description of the invention

Technical problem

In the antenna device shown in Patent Document 1, the first radiating element and the second radiating element are coupled to each other via the transformer, and a feeding circuit and the antenna device are adjusted by adjusting the coupling. Since the first radiating element and the second radiating element need not be arranged in parallel in the antenna device shown in Patent Document 1, there is a great deal of freedom in designing patterns for them. In addition, even if the first radiation element and the second radiation element are provided closer to each other, a predetermined degree of coupling may be selected. This allows easy adaptation of the supply circuit and a multi-resonant antenna.

However, in a frequency band in which one of the radiating elements substantially contributes to the radiation, a radiation characteristic of radiation from the other radiating element may not achieve a desired radiation characteristic of radiation of the one radiating element, which deteriorates a radiation characteristic of an antenna.

An object of the present invention is to provide an antenna device which avoids an interference problem with respect to the radiation of two radiating elements to widen the tape and electronic equipment comprising the antenna device.

the solution of the problem

• ein Strahlungselement;
• eine Kopplungsschaltung, die ein erstes Kopplungselement und ein zweites Kopplungselement umfasst, wobei das erste Kopplungselement zwischen das Strahlungselement und einen Speisekreis geschaltet ist, wobei das zweite Kopplungselement an das erste Kopplungselement gekoppelt ist; und
• einen nichtstrahlenden Resonanzkreis, der mit dem zweiten Kopplungselement verbunden ist,
• wobei eine Frequenzcharakteristik einer Rückflussdämpfung des Strahlungselements durch eine Resonanzfrequenzcharakteristik des nichtstrahlenden Resonanzkreises eingestellt ist.

(1) An antenna device according to the present invention comprises:

• a radiating element;
• a coupling circuit comprising a first coupling element and a second coupling element, the first coupling element being connected between the radiating element and a feeding circuit, the second coupling element being coupled to the first coupling element; and
• a nonradiative resonant circuit connected to the second coupling element,
• wherein a frequency characteristic of a return loss of the radiating element is set by a resonance frequency characteristic of the non-radiating resonant circuit.

In the above embodiment, the radiating element and the nonradiative resonant circuit do not interfere with each other with respect to the radiation, the radiating element being connected to the first coupling element of the coupling circuit, the nonradiative resonant circuit being connected to the second coupling element of the coupling circuit, and a radiation characteristic of the radiating element being not adversely affected. In addition, the frequency characteristic of a return loss of the radiating element seen from the feeding circuit is adjusted by a resonance characteristic of the non-radiating resonant circuit, and a pole position is generated in a desired frequency band to widen the band of the frequency characteristic of the antenna.

(2) Preferably, a direction of a magnetic field generated when current in the first coupling element flows in a direction from a terminal connected to the feeding circuit to a terminal connected to the radiating element is opposite to a direction of a magnetic field that is generated when current in the second coupling element flows in a direction from a terminal connected to the nonradiative resonant circuit to a grounded terminal. Thus, due to the coupling between the first coupling element and the second coupling element, a mutual inductance reduces the inductances of the first coupling element and the second coupling element, with a small influence on a circuit characteristic and the radiation characteristic of the radiation element 10 ,

(3) Preferably, the first coupling element and the second coupling element are multilayered coil conductor patterns, and the coupling circuit forms a transformer in which the first coupling element and the second coupling element are electromagnetically coupled to each other. Thus, a coupling circuit having a high coupling coefficient is formed between the first coupling element and the second coupling element, and the resonance characteristic of the nonradiative resonant circuit seen from the sense circuit is likely to be exhibited.

(4) Preferably, in plan view of the radiating element, one half or more of the nonradiative resonant circuit is contained in a formation region of the radiating element. Thus, the nonradiative resonant circuit is shielded by the radiating element. This increases a nonradiative property of the nonradiative resonant circuit seen from a distance.

(5) Preferably, the radiating element is constituted by a conductive member forming three sides in plan view, and the nonradiative resonant circuit is surrounded in plan view from the three sides. Thus, the nonradiative resonant circuit is shielded by the radiating element. This increases the nonradiative property of the nonradiative resonant circuit seen from a distance.

(6) Preferably, the non-radiative resonance circuit is formed by a linear conductor pattern having a feedback part at a center. Thus, the sharpness of the resonance of the nonradiative resonant circuit is reduced, and the nonradiative resonant circuit can attenuate a reflection coefficient in a relatively wide band including the band in which the pole location is generated in the frequency characteristic of the antenna and its periphery. In addition, the nonradiative resonant circuit may be provided in a small area.

(7) Preferably, the conductor pattern includes a first linear conductor pattern part extending from the coupling circuit, and a second conductor pattern part returning at the return part to be away from the radiating element. This suppresses unnecessary coupling between the nonradiative resonant circuit and the radiating element.

(8) It is also preferable to include a phase shifter connected between the feed circuit and the first coupling element and having a frequency dependency. This enables the formation of an antenna device that performs impedance matching in a wide band.

(9) Preferably, a second terminal of the second coupling element is grounded, the second terminal being opposite to a first terminal to which the nonradiative resonant circuit is connected, and a length of a lead between the first coupling element and the feeding circuit and a length of one Line between the second terminal of the second coupling element and ground are each less than 1/8 wavelength of a resonant frequency.

Since the coupling circuit mainly uses magnetic field coupling, the strength of the coupling is increased when the coupling circuit is inserted in a portion where a large current flows. The strong coupling can increase the influence of resonance obtained by adding the coupling circuit and the parasitic element, and since a resonance bandwidth is widened, a frequency band in which communication is possible is widened. In addition, a signal strength is increased and a communication characteristic is improved.

(10) The antenna device may include an inductor connected between the second coupling element and the nonradiative resonance circuit. Since the inductor is inserted in a portion where the current is low while suppressing a change in the coupling (a change in the impedance matching is suppressed), the resonant frequency on the side of the nonradiative resonant circuit can be reduced and a desired communication band can be achieved. Alternatively, while maintaining the resonant frequency, the length of the nonradiative resonant circuit can be reduced, thereby reducing the area used.

(11) The antenna device may include an inductor connected between the first terminal of the second coupling element and ground. Thus, reactance caused by a parasitic capacitance between ground and the coupling circuit by the insertion of the coupling circuit can be suppressed, and a change from a matching state in which the coupling circuit is not mounted can be suppressed. In addition, the resonance frequency of the nonradiative resonant circuit can be reduced and a desired communication band or a desired communication characteristic can be obtained. Alternatively, while maintaining the resonant frequency, the length of the Antenna can be reduced, thereby reducing the area used.

(12) The antenna device may further include a capacitor connected between the second coupling element and the nonradiative resonance circuit. Thus, the resonant frequency on the side of the nonradiative resonant circuit can be increased and a desired communication band can be achieved.

(13) The antenna device may further include a capacitor connected between the first terminal of the second coupling element and ground. Thus, a parasitic capacitance generated between ground and the coupling circuit by the insertion of the coupling circuit can be reduced, and a change from a matching state in which the coupling circuit is not mounted can be suppressed. In addition, the resonance frequency on the side of the non-radiative resonance circuit can be increased, and a desired communication band or a desired communication characteristic can be obtained.

(14) The antenna device may be configured to further include: a second coupling circuit including a third coupling element and a fourth coupling element, the third coupling element being connected between the first coupling element and the feeding circuit, the fourth coupling element being connected to the third coupling element Coupling element is coupled; and a second nonradiative resonant circuit connected to the fourth coupling element. Thus, the number of the resonances to be added can be increased, and a bandwidth is widened, and accordingly, a domain in which communication is possible is broadened. At the same resonant frequency, the impedance matching is improved.

(15) The antenna device may further include: a second coupling circuit including a third coupling element and a fourth coupling element, the third coupling element being connected between the second coupling element and the nonradiative resonant circuit, the fourth coupling element being coupled to the third coupling element; and a second nonradiative resonant circuit connected to the fourth coupling element. With this structure, a plurality of nonradiative resonant circuits can be used, and a communication characteristic is improved.

(16) The antenna device may further include a switch connected between the nonradiative resonant circuit and ground. This may change a resonant frequency added by the provision of the coupling circuit and the nonradiative resonant circuit and may alter the matching to improve the impedance matching. In addition, the resonance frequency or the matching can be changed in such a manner that the coupling circuit and the nonradiative resonant circuit are easily coupled to each other, thereby improving the impedance matching.

(17) In the case that the coupling circuit includes a parasitic capacitance, the antenna device preferably further includes an inductor connected to the coupling circuit and suppressing a reactance component generated in the coupling circuit by parallel resonance with the parasitic capacitance. Thus, a reactance component added by the insertion of the coupling circuit is canceled, and a change from a matching state in which the coupling circuit is not mounted can be suppressed.

• ein Strahlungselement, mit dem ein Speisekreis verbunden ist;
• eine Kopplungsschaltung, die ein erstes Kopplungselement und ein zweites Kopplungselement umfasst, wobei das erste Kopplungselement zwischen das Strahlungselement und Masse geschaltet ist, wobei das zweite Kopplungselement an das erste Kopplungselement gekoppelt ist; und
• einen nichtstrahlenden Resonanzkreis, der mit dem zweiten Kopplungselement verbunden ist,
• wobei eine Frequenzcharakteristik einer Rückflussdämpfung des Strahlungselements durch eine Resonanzfrequenzcharakteristik des nichtstrahlenden Resonanzkreises eingestellt ist.

(18) An antenna device according to the present invention comprises:

• a radiating element to which a feeding circuit is connected;
• a coupling circuit comprising a first coupling element and a second coupling element, the first coupling element being connected between the radiating element and ground, the second coupling element being coupled to the first coupling element; and
• a nonradiative resonant circuit connected to the second coupling element,
• wherein a frequency characteristic of a return loss of the radiating element is set by a resonance frequency characteristic of the non-radiating resonant circuit.

In the above embodiment, the radiation element and the nonradiative resonance circuit do not interfere with each other with the radiation element connected to the first coupling element of the coupling circuit, the nonradiative resonance circuit being connected to the second coupling element of the coupling circuit and the radiation characteristic of the radiation element not adversely affected , In addition, the frequency characteristic of a return loss of the radiating element seen from the feeding circuit is adjusted by the resonance characteristic of the non-radiating resonant circuit, and a pole point is generated in a desired frequency band to widen the band of the frequency characteristic of the antenna. Since a current in a section connected to ground, is particularly high, the radiating element and the non-radiative resonance circuit can be coupled to each other via the coupling circuit. In addition, there is greater freedom in the arrangement of the coupling circuit and the non-radiative resonance circuit.

(19) Electronic equipment according to the present invention comprises the above antenna device according to any one of (1) to (18); the supply circuit connected to the coupling circuit; and a housing in which the feeding circuit is received, wherein a part of the radiation element or the entire radiation element is a part of the housing.

In the above embodiment, it is not necessary to provide a conductive member or pattern specifically for the radiating element, and size reduction can be achieved. In addition, in electronic equipment with a metal housing, electromagnetic waves are not blocked by the metal housing.

Advantageous Effects of the Invention

According to the present invention, the antenna device which avoids the interference problem with respect to the radiation of two radiating elements to widen the band and the electronic equipment comprising the antenna device can be achieved.

list of figures

• 1 FIG. 15 is a perspective view showing a principal configuration of an antenna device. FIG 101 according to a first embodiment as well as electronic equipment representing the antenna device 101 includes.
• 2 Fig. 10 is a plan view of a main part of the antenna device 101 ,
• 3 is a plan view of a location where a non-radiative resonant circuit 20 is formed.
• 4 represents an embodiment of a coupling circuit 30 and an associated circuit.
• 5 (A) Fig. 10 is an equivalent circuit diagram of the antenna device 101 in a high band. 5 (B) Fig. 10 is an equivalent circuit diagram of the antenna device 101 in a low band.
• 6 represents a frequency characteristic of a return loss of the antenna device 101 and an antenna device of a comparative example.
• 7 FIG. 4 is a conceptual diagram illustrating a difference in impedance matching depending on the coupling strength of the coupling circuit. FIG.
• 8th is a perspective view of the coupling circuit 30 ,
• 9 10 are exploded plan views illustrating some conductor patterns formed on layers of the coupling circuit.
• 10 is a circuit diagram of the coupling circuit 30 with four coil conductor patterns.
• 11 illustrates a circuit design of an antenna device 102 according to a second embodiment.
• 12 illustrates a circuit design of an antenna device 103 according to a third embodiment.
• 13 illustrates a circuit design of an antenna device 104 according to a fourth embodiment.
• 14 illustrates a circuit design of an antenna device 105 according to a fifth embodiment.
• 15 illustrates a circuit design of an antenna device 106A according to a sixth embodiment.
• 16 illustrates a circuit design of an antenna device 106B according to the sixth embodiment.
• 17 illustrates a circuit design of an antenna device 106C according to the sixth embodiment.
• 18 illustrates a circuit design of an antenna device 106D according to the sixth embodiment.
• 19 (A) illustrates a circuit design of an antenna device 107A according to a seventh embodiment. 19 (B) illustrates a circuit design of an antenna device 107B according to the seventh embodiment.
• 20 Figure 11 is an exploded plan view illustrating conductor patterns formed on layers of the coupling circuit 30 are formed according to the seventh embodiment.
• 21 is a sectional view of the coupling circuit 30 according to the seventh embodiment.
• 22 is a plan view, which is an overlap between a ladder pattern L12 and a ladder pattern L21 in particular in the coupling circuit 30 represents according to the seventh embodiment.
• 23 Figure 11 is an exploded plan view illustrating conductor patterns formed on layers of another coupling circuit 30 are formed according to the seventh embodiment.
• 24 illustrates a circuit design of an antenna device 108 according to an eighth embodiment.
• 25 illustrates a circuit design of an antenna device 109 according to a ninth embodiment.
• 26 illustrates a circuit design of an antenna device 110 according to a tenth embodiment.
• 27 (A) and 27 (B) provide circuit designs of antenna devices 111A and 111B according to an eleventh embodiment.
• 28 illustrates a circuit design of an antenna device 112 according to a twelfth embodiment.
• 29 illustrates a circuit design of an antenna device 113 according to a thirteenth embodiment.
• 30 illustrates a circuit design of an antenna device 114 according to a fourteenth embodiment.
• 31 (A) illustrates a circuit design of an antenna device 115 according to a fifteenth embodiment. 31 (B) represents a frequency characteristic of a return loss of in 31 (A) illustrated antenna device 115 and an antenna device according to a comparative example.
• 32 illustrates a circuit design of an antenna device 116 according to a sixteenth embodiment.
• 33 illustrates a circuit design of an antenna device 117 according to a seventeenth embodiment.
• 34 represents a frequency characteristic of a return loss of the antenna device 117 represents.
• 35 Fig. 10 is a circuit diagram of an antenna device 118A according to an eighteenth embodiment.
• 36 is a circuit diagram of another antenna device 118B according to the eighteenth embodiment.
• 37 is a circuit diagram turn another antenna device 118C according to the eighteenth embodiment.
• 38 is a plan view of a main part of an antenna device 119 according to a nineteenth embodiment.
• 39 is a perspective view of the coupling circuit 30 according to the nineteenth embodiment.
• 40 represents an embodiment of another coupling circuit 30 according to the nineteenth embodiment, and is an exploded plan view, which is based on layers of the coupling circuit 30 represents formed conductor patterns.
• 41 Fig. 10 is a circuit diagram of an antenna device 120 according to a twentieth embodiment, in which a feeding circuit 1 connected is.
• 42 is an equivalence circuit diagram showing a phase shifter 50 according to the twentieth embodiment, in which an ideal transformer IT and parasitic inductance components are shown separately.
• 43 represents a frequency characteristic of a phase shift amount of the phase shifter 50 represents.
• 44 (A) FIG. 12 is a circuit diagram of the antenna device shown in the first embodiment which is not the phase shifter 50 includes, and 44 (B) represents impedance locations on a Smith chart with impedances from the supply circuit 1 represented antenna device.
• 45 (A) Fig. 10 is a circuit diagram of an antenna device with added phase shifter 50 , 45 (B) represents impedance locations on a Smith chart with impedances from the supply circuit 1 represented antenna device.
• 46 (A) FIG. 12 is a circuit diagram of an antenna device including an impedance matching capacitor. FIG C5 includes. 46 (B) represents an impedance location which, on a Smith chart, imparts impedance from the sense circuit 1 represents viewed antenna device.
• 47 represents a frequency characteristic of a return loss of in 44 (A) and 46 (A) shown antenna devices and an antenna device according to a comparative example.
• 48 is an external perspective view of the phase shifter 50 ,
• 49 is a plan view of layers in the phase shifter 50 ,
• 50 is a sectional view of the phase shifter 50 ,
• 51 FIG. 10 is a plan view illustrating a part of a metal case of electronic equipment according to a twenty-first embodiment. FIG.
• 52 (A) and 52 (B) FIG. 15 is perspective views illustrating parts of metal cases of different electronic equipment according to the twenty-first embodiment.

Description of the embodiments

In the following, several embodiments for the implementation of the present invention will now be described by way of specific examples with reference to the drawings. Like reference numerals designate like parts throughout the drawings. In view of the explanation of the main points and the intelligibility, the embodiments are shown separately for the sake of simplicity. However, the parts shown in the different embodiments may be replaced or combined. In a second embodiment and further embodiments, a description of the common with a first embodiment will be omitted, and only different points will be described. In particular, the same or substantially the same effects achieved by the same or substantially the same education, not repeated in the embodiments.

The "antenna device" shown in the embodiments is applicable to one that transmits signals, or one that receives signals. Also, in the case that the "antenna device" is described as an antenna that radiates electromagnetic waves, the antenna device is not limited to a source that generates the electromagnetic waves. Also, in the case of receiving an electromagnetic wave generated by a communication partner antenna apparatus, that is, even if transmission and reception are reversed, the same or substantially the same effects are produced.

First embodiment

1 FIG. 15 is a perspective view showing a principal configuration of an antenna device. FIG 101 according to the first embodiment and of electronic equipment comprising the antenna device 101 includes. 2 Fig. 10 is a plan view of a main part of the antenna device 101 ,

A metal housing of the electronic equipment comprises a radiating element 10 which is an end portion of the metal shell, and a metal shell main body 40 , The metal housing body 40 is made up of a planned part 41 and side panel parts 42 and 43 together.

The antenna device 101 includes the radiating element 10 , a non-radiative resonant circuit 20 and the coupling circuit 30 ,

The radiation element 10 is the end portion of the metal housing and is made up of an end surface part 11 and side panel parts 12 and 13 together. An end portion of the side surface part 12 is about an inductor 8th with a ground of a circuit substrate 6 connected (is grounded). Although an end portion of the side surface part 13 is open, between this open end and ground is a parasitic capacitance C generated. It is noted that on the circuit substrate 6 a connector 7 such as a USB is mounted and in the Endflächenteil 11 an opening for the connector 7 is formed. However, the connector is 7 no component of the antenna device 101 ,

The circuit substrate 6 includes a mass region GZ in which a ground electrode GND is formed, and a non-mass region NGE in which no ground electrode is formed. The end portion of the metal housing, which is the radiating element 10 is located on the side of the non-mass region. In the non-mass region NGE of the circuit substrate 6 is by means of a conductor pattern of the nonradiative resonant circuit 20 educated. Also in the non-mass region NGE of the circuit substrate 6 is a feed line 9 formed, which is the coupling circuit 30 and the radiating element 10 connects with each other.

As in 2 is shown, the non-radiative resonant circuit 20 formed by a linear conductor pattern, which in the middle of a return part 20FB having. In this way, since the linear conductor pattern is used with a return part in the middle, the nonradiative resonant circuit is used 20 is provided in a small area, and an electrical length necessary for resonance can be obtained. In addition, in this embodiment, a first linear conductor pattern part 21 that is different from the coupling circuit 30 extends, and a second linear conductor pattern part 22 included, that to one of the radiating element 10 the remote side. As a part near the radiating element 10 (in particular the end surface part 11 ) is short and the directions of extension are opposite to each other, in this structure is a substantial coupling with the radiating element 10 (especially the end face part 11 ) weak. This suppresses an unnecessary Coupling between the non-radiating resonant circuit 20 and the radiating element 10 ,

It is noted that the second linear conductor pattern part 22 wider than the first linear conductor pattern part 21 is. Thus, a resonance bandwidth can be widened.

3 is a plan view illustrating a location at which the nonradiative resonant circuit 20 is formed. The radiation element 10 is formed by a conductive member (the end face part 11 and the side panel parts 12 and 13 ), which in plan view forms three sides, and the nonradiative resonant circuit 20 is of a radiating element formation region 10Z surrounded in plan view by the three sides of the radiating element 10 is surrounded. It does not need the entire non-radiative resonant circuit 20 within the radiating element formation region 10Z and preferably one half or more of the nonradiative resonant circuit is provided 20 in the radiating element formation region 10Z contain. Since the non-radiating resonant circuit 20 is not used as a radiating element is the non-radiative resonant circuit 20 preferably "non-radiative". In the case that the nonradiative resonant circuit 20 is surrounded in plan view from the three sides of the conductive member and one half or more of the nonradiative resonant circuit 20 in the radiating element formation region 10Z is included, is thus the non-radiative resonant circuit 20 through the radiating element 10 shielded. This increases a non-radiative property of the nonradiative resonant circuit 20 , seen from a distance.

4 represents an embodiment of the coupling circuit 30 and an associated circuit. The coupling circuit 30 comprises a first coupling element 31 and a second coupling element 32 connected to the first coupling element 31 is coupled, and by the first coupling element 31 and the second coupling element 32 is a transformer formed. The first coupling element 31 and the second coupling element 32 have small inductances of 10 nH each or less. The radiation element 10 and the nonradiative resonant circuit 20 are via the coupling circuit 30 with a coupling coefficient of 0.5 or more, preferably with a coupling coefficient of 0.8 or more. With a smaller inductance of a coupling element, the influence on a circuit characteristic and a radiation characteristic of the radiation element 10 be suppressed to a greater degree. With higher coupling coefficients are the radiating element 10 and the nonradiative resonant circuit 20 more electrically connectable to each other, and a resonance point can only be added to a frequency at which the nonradiative resonant circuit 20 contributes to a greater extent to the resonance. By forming a transformer in which an electrical magnetic field coupling between the first coupling element 31 and the second coupling element 32 is formed, in this way, a coupling circuit with a high coupling coefficient between the first coupling element 31 and the second coupling element 32 formed, and it shows with some probability a resonance characteristic of the non-radiative resonant circuit 20 at view of the radiation element 10 from a feeding circuit 1 ,

The first coupling element 31 is between the radiating element 10 and the food circle 1 connected. A first end of the second coupling element 32 is with the nonradiative resonant circuit 20 and a second end thereof is connected to ground of the circuit substrate 6 connected (is grounded).

In the electronic equipment according to this embodiment, the metal part of the housing that receives the feeding circuit is used as the radiating element, and thus it is unnecessary to provide a conductive member or a conductor pattern specifically for the radiating element, thereby achieving size reduction. In addition, even with electronic equipment with a metal housing, electromagnetic waves are not blocked by the metal housing.

5 (A) Fig. 10 is an equivalent circuit diagram of the antenna device 101 in a high band. In a high band (eg 1.6 GHz to 2.3 GHz) has the inductor 8th (please refer 2 and 4 ) a predetermined high impedance, and a tip of the radiating element 10 is equivalent open. In this state, the radiating element serves 10 as a monopole type antenna resonating at 3/4 wavelengths or (2n + 1) / 4 wavelengths (n is a natural number).

5 (B) Fig. 10 is an equivalent circuit diagram of the antenna device 101 in a low band. In a low band (eg 700 MHz to 900 MHz) has the inductor 8th a predetermined inductance, and the tip of the radiating element 10 is about the inductor 8th grounded. In this state, the radiating element serves 10 as a loop antenna with a wavelength or its integer multiple.

A series connection, which is an inductor L20 and a capacitor C20 includes, which in 5 (A) and 5 (B) is an element for representing an equivalent circuit, wherein the nonradiative resonant circuit 20 simply represented as a concentrated constant circuit. The non-radiating resonant circuit 20 serves as open stub which resonates at a predetermined frequency at 3/4 wavelengths or (2n + 1) / 4 wavelengths (n is a natural number). Thus, for illustration in 5 (A) and 5 (B) the inductor L20 and the capacitor C20 used. The non-radiating resonant circuit 20 For example, it resonates in a frequency band whose center is 2.1 GHz. It is noted that in this embodiment, since the nonradiative resonant circuit 20 has a shape in which the linear conductor pattern is returned in the middle, compared with a simple linear conductor pattern no clear standing wave is formed in the linear conductor pattern and a Q value of the resonance as a resonance circuit is small.

6 represents a frequency characteristic of a return loss of the antenna device 101 and an antenna device of a comparative example. In 6 For example, a return loss characteristic RL1 is a return loss of the antenna device 101 According to this embodiment, and a return loss characteristic RL2 is a return loss of the antenna device according to the comparative example. The antenna device according to the comparative example is an antenna device in which the coupling circuit 30 and the nonradiative resonant circuit 20 not included. In both antenna devices is at a central frequency F1 of a low band (700 MHz to 900 MHz) produces a pole. The reason for this is the resonance characteristic of in 5 (B) illustrated loop antenna. Another pole is at one frequency F2 generated (at 1.75 GHz). The reason for this is the 3/4 wavelength resonance of the in 5 (A) shown monopole antenna. Again, another pole is at one frequency F3 generated (by 2.3 GHz). The reason for this is the 5/4 wavelength resonance of the in 5 (A) shown monopole antenna.

It is noted that preferably one length " r1 "A line between the first coupling element 31 and the food circle 1 , as in 4 represented, and a length " r2 A line between an end portion of the second coupling element 32 and mass less than 1/8 wavelength of the resonant frequency. The wavelength here may mean an effective wavelength taking into account a wavelength-shortening effect of a magnetic body or a dielectric. The threshold is set to " 1 / 8th Wavelength "set, because this is practical until a condition is met, in which a current of 1/8 wavelength 1 / √2, in other words, until a power that can be transmitted is halved.

7 Here is a conceptual diagram of a difference in impedance matching depending on the strength of the coupling 7 are the places T1 . T2 and T3 Impedance Places on a Smith Chart Impedances on View of the Antenna Device 101 from the feeding circle 1 represent. The place T1 is a characteristic in a state where the coupling circuit 30 and the nonradiative resonant circuit 20 not provided, the place T2 is a characteristic in a state where the first coupling element 31 and the second coupling element 32 the coupling circuit 30 suitably coupled together, and the location T3 is a characteristic in a state where the coupling between the first coupling element 31 and the second coupling element 32 the coupling circuit 30 too strong.

When the coupling between the first coupling element 31 and the second coupling element 32 the coupling circuit 30 In this way, the input impedance, as seen from the supply circuit, deviates from the impedance (eg 50 Ω) on the side of the supply circuit (and the transmission line). Therefore, it is important that the first coupling element 31 and the second coupling element 32 the coupling circuit 30 are appropriately connected. The length " r1 "The line between the first coupling element 31 and the food circle 1 as well as the length " r2 "The line between the end portion of the second coupling element 32 and ground are set to a range of less than 1/8 wavelength of the resonant frequency, and thereby the coupling through the coupling circuit 30 be set appropriately.

In the antenna device 101 According to this embodiment, another pole is at one frequency F0 generated (by 2.1 GHz). The reason for this is the resonance characteristic of the nonradiative resonance circuit 20 , This means that the nonradiative resonant circuit 20 is resonant in a frequency band whose center frequency is 2.1 GHz, the pole at 2.1 GHz is in the frequency characteristic of a return loss of the antenna device 101 , from the food circle 1 seen, generated. With the antenna device 101 According to this embodiment, a high band application frequency band is broadened from 1.6 GHz to 2.3 GHz.

In the low band is the nonradiative resonant circuit 20 not in resonance, and the return loss characteristic in the low band is not affected. This means the non-radiative resonance circuit 20 affects the return loss characteristic of the supply circuit 1 seen, for example, in a frequency band of 1.6 GHz or above, and the nonradiative resonant circuit 20 has essentially no influence in a lower frequency band.

The return loss characteristic around the frequency F0 is due to the resonance characteristic of the nonradiative resonance circuit 20 determined, and accordingly, the return loss characteristic at about the frequency F0 be suitably influenced by the shape of the conductor pattern forming the non-radiative resonant circuit. Since the non-radiating resonant circuit 20 is formed by the linear conductor pattern having a feedback part in the middle, in this embodiment, the sharpness of the resonance of the nonradiative resonance circuit 20 reduced, and the non-radiative resonance circuit 20 can attenuate a reflection coefficient in a wide band, including the band in which the pole at the frequency F0 is generated, and its periphery.

It is noted that the nonradiative resonant circuit 20 acting as an open stub, substantially independent of the radiating element 10 is provided. Thus, there is no influence on a low band, unlike, for example, the case where a stub is formed in the radiating element.

Next, an embodiment of the coupling circuit will be described 30 described. 8th is a perspective view of the coupling circuit 30 , and 9 Figure 11 is an exploded plan view illustrating conductor patterns formed on layers of the coupling circuit.

The coupling circuit included in the antenna device according to this embodiment 30 is a rectangular paralleielipiped-shaped chip component applied to the circuit substrate 6 to assemble. In 8th are an external form of the coupling circuit 30 and an inner structure thereof is shown separately. The outer shape of the coupling circuit 30 is represented by a two-dot chain line. On an outer surface of the coupling circuit 30 are a supply circuit connection terminal PF , a radiating element connecting terminal PA , a ground connection PG and a non-radiating resonant circuit connecting terminal PS educated. In addition, the coupling circuit includes 30 a first surface MS1 and a second area MS2 which is a surface opposite to the first surface. In this embodiment, the first surface is MS1 a mounting surface, and this surface faces a circuit substrate. On an upper surface (second surface), one towards the mounting surface (first surface) MS1 is opposite surface, a direction discrimination mark DDM is provided. This direction discrimination mark DDM is used for detecting a direction of a chip component when, for example, the coupling circuit 30 is mounted as a chip component in a mounting device on a circuit substrate.

Within the coupling circuit 30 are a first ladder pattern L11 , a second ladder pattern L12 , a third ladder pattern L21 and a fourth conductor pattern L22 educated. The first ladder pattern L11 and the second conductor pattern L12 are over a layer connection conductor V1 connected with each other. The third conductor pattern L21 and the fourth conductor pattern L22 are over a layer connection conductor V2 connected with each other. It is noted that 8th insulating materials S11 . S12 . S21 and S22 on which the respective conductor patterns are formed, shown separately in a stacking direction. These insulating materials S11 . S12 . S21 and S22 may be a non-magnetic ceramic multilayer body formed of LTCC (low-temperature buried ceramic) or the like, or may be a synthetic resin multi-layer body formed of a synthetic resin material such as polyimide or liquid crystal polymer. In this way, since the material layers are nonmagnetic (no magnetic ferrite), it is possible to use the material layers for a coupling circuit also in a high frequency band exceeding hundreds of MHz.

The above conductor patterns and the layer connection conductors are each formed of a conductor material containing Ag or Cu as a main component and having a low resistivity. For example, in the case that the material layers are ceramic, the conductor patterns and the layer connection conductors are formed by screen printing and firing a conductive paste containing Ag or Cu as a main component. For example, in the case where the material layers are made of synthetic resin, the conductor patterns and the layer connection conductors are patterned by etching or the like a metal foil such as an Al foil or a Cu foil.

As in 9 shown are the first conductor pattern L11 , the second ladder pattern L12 , the third ladder pattern L21 and the fourth conductor pattern L22 from a layer near the mounting surface formed in this order. A first end of the first conductor pattern L11 is with the radiating element connecting terminal PA and a second end thereof is over the layer connection conductor V1 with a first end of the second conductor pattern L12 connected. A second end of the second conductor pattern L12 is with the supply circuit connection terminal PF connected. In addition, a first end of the third conductor pattern L21 with a non-radiating resonant circuit connection terminal PS connected, and a second end of the third conductor pattern L21 is over the layer connection conductor V2 with a first End of the fourth conductor pattern L22 connected. A second end of the fourth conductor pattern L22 is with the ground connection PG connected.

In addition, a winding direction of the first coupling element 31 from the supply circuit connection terminal PF to the radiating element connecting terminal PA and a winding direction of the second coupling element 32 from the non-radiating resonant circuit connecting terminal PS to the ground connection PG opposite to each other. That is, a magnetic field (magnetic flux) generated when in the first coupling element 31 from the supply circuit connection terminal PF to the radiating element connecting terminal PA Current flows, and a magnetic field (magnetic flux), which is generated when in the second coupling element 32 from the non-radiating resonant circuit connecting terminal PS to the ground connection PG Electricity flows, weaken each other. When the radiating element connecting terminal PA As a monopole antenna is in resonance, here have the first coupling element 31 and the second coupling element 32 in the coupling circuit 30 that go beyond the food circle 1 and the ground electrode GND is connected, opposite polarities. In the first coupling element 31 Current flows from the supply circuit connection terminal PF to the radiating element connecting terminal PA , and in the second coupling element 32 Current flows from the non-radiating resonant circuit connection terminal PS to the ground connection PG , Magnetic fields (magnetic fluxes) that are generated weaken each other. Thus, mutual inductance decreases due to the coupling between the first coupling element 31 and the second coupling element 32 the inductances of the first coupling element 31 and the second coupling element 32 with a small influence on the circuit characteristic and the radiation characteristic of the radiating element 10 ,

10 is a circuit diagram of the coupling circuit 30 with the four above coil conductor patterns. The second conductor pattern L12 and the first ladder pattern L11 are connected in series to the first coupling element 31 to build. Likewise, the fourth ladder pattern L22 and the third conductor pattern L21 connected in series to the second coupling element 32 to build. The second conductor pattern L12 and the third conductor pattern L21 are adjacent to each other in the thickness direction, and the magnetic field coupling between the second conductor pattern L12 and the third conductor pattern L21 is especially strong. Thus, the second conductor pattern L12 and the third conductor pattern L21 in 10 adjacent to each other. As can be seen, magnetic field coupling is between the second conductor pattern L12 and the fourth conductor pattern L22 as well as between the first conductor pattern L11 and the third conductor pattern L21 manufactured.

At the in 9 The example shown is a capacitor formation conductor pattern C11 in a part of the second conductor pattern L12 formed, and a capacitor formation conductor pattern C12 is in a part of the third ladder pattern L21 educated. As in 10 is accordingly between a center of the second conductor pattern L12 and the nonradiative resonant circuit connection terminal PS a capacitor C1 educated. The capacitor C1 serves as an impedance matching circuit between the supply circuit 1 and the nonradiative resonant circuit 20 ,

Second embodiment

11 illustrates a circuit design of an antenna device 102 according to the second embodiment. In the antenna device 102 is an inductor 35 between the second coupling element 32 the coupling circuit 30 and the nonradiative resonant circuit 20 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

Because the inductor 35 is inserted in a section where the current is low while changing the coupling of the coupling circuit 30 is suppressed, according to this embodiment, the resonance frequency of the nonradiative resonance circuit 20 can be reduced and a desired communication band can be achieved. Alternatively, while maintaining the resonant frequency, the length of the nonradiative resonant circuit 20 be reduced and thereby reduce the area used.

It is noted that the above inductor 35 also with the coupling circuit 30 can be integrated. The inductor is preferred 35 however, designed so that it does not contact the first coupling element 31 is coupled.

Third embodiment

12 illustrates a circuit design of an antenna device 103 according to a third embodiment. In the antenna device 103 is the inductor 35 between the second coupling element 32 the coupling circuit 30 and ground switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

When the coupling circuit 30 is added to the antenna device is between ground and the coupling circuit 30 a parasitic Capacity generated. According to this embodiment, resonance between the above inductor 35 and the parasitic capacitance suppress a reactance component. In a frequency band in which an antenna characteristic is added by adding the coupling circuit 30 to the antenna device should not be changed, therefore, by the insertion of the inductor 35 with such an inductance that it resonates with the parasitic capacitance, a change from a matching state is suppressed, in which the coupling circuit 30 not mounted.

In addition, the insertion of the inductor 35 the resonant frequency of the nonradiative resonant circuit 20 can be reduced, and a desired communication band or a desired communication characteristic can be achieved. Alternatively, while maintaining the resonant frequency, the length of the antenna can be reduced and the range used reduced.

It is noted that the above inductor 35 also with the coupling circuit 30 It may be integrated. The inductor is preferred 35 however, formed so as not to be the same to the first coupling element 31 is coupled.

Fourth embodiment

13 illustrates a circuit design of an antenna device 104 according to a fourth embodiment. In the antenna device 104 is a capacitor 36 between the second coupling element 32 the coupling circuit 30 and the nonradiative resonant circuit 20 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

According to this embodiment, the resonant frequency on the side of the nonradiative resonant circuit can be increased and a desired communication band can be achieved.

It is noted that the above capacitor 36 with the coupling circuit 30 can be integrated.

Fifth embodiment

14 illustrates a circuit design of an antenna device 105 according to a fifth embodiment. In the antenna device 105 is the capacitor 36 between the second coupling element 32 the coupling circuit 30 and ground switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

According to this embodiment, a parasitic capacitance between the ground and the coupling circuit 30 by inserting the coupling circuit 30 is generated, can be reduced (the combined capacitance can be reduced), and a change from a matching state in which the coupling circuit 30 is not mounted, can be suppressed. In addition, the resonant frequency of the nonradiative resonant circuit 20 can be increased and a desired communication band or a desired communication characteristic can be achieved.

It is noted that the capacitor 36 with the coupling circuit 30 can be integrated.

Sixth embodiment

15 illustrates a circuit design of an antenna device 106A according to a sixth embodiment. In the antenna device 106A is the inductor 35 between the first coupling element 31 the coupling circuit 30 and the radiating element 10 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

In the formation of the antenna device 106A is the first coupling element 31 closer to the feeding circuit 1 where the current is strong, as the inserted inductor 35 , Thus, while maintaining a portion of power for delivery to the nonradiative resonant circuit 20 the resonant frequency of the radiating element 10 be changed and set a level of impedance matching. In addition, the reduction of a self-resonant frequency unlikely due to the inductances of the first coupling element 31 and the second coupling element 32 and the parasitic capacitance is determined between the first coupling element 31 and the second coupling element 32 is generated, and thus the self-resonant frequency does not adversely affect the use in a communication frequency band. That is, in a state of self-resonance, the energy in the frequency band comes to ground and is not radiated. In a state where the natural resonance frequency is higher than the communication frequency band, however, such a problem does not arise.

16 illustrates a circuit design of an antenna device 106B according to the sixth embodiment. In the antenna device 106B is the inductor 35 between the first coupling element 31 the coupling circuit 30 and the food circle 1 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

Because the first coupling element 31 the coupling circuit 30 is inserted on a side where the current is weaker than at the location of the inductor 35 is compared with a case where the inductor is between the radiating element 10 and the first coupling element 31 is inserted in the formation of the antenna device 106B a suitable adjustment of the level of impedance matching at a resonance (resonant frequency) possible by the coupling circuit 30 and the nonradiative resonant circuit 20 is added. Specifically, it is possible to avoid a situation where an input impedance excessively changes and the impedance is no longer matched.

In addition, the insertion of the inductor 35 the self-resonant frequency of the coupling circuit 30 Thus, by setting the self-resonant frequency to a frequency band which is not to be radiated, unnecessary radiation can be suppressed.

17 illustrates a circuit design of an antenna device 106C according to the sixth embodiment. In the antenna device 106C is the capacitor 36 between the first coupling element 31 the coupling circuit 30 and the radiating element 10 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

In the formation of the antenna device 106C can by the capacity of the inserted capacitor 36 the resonant frequency of the radiating element 10 can be adjusted and the level of impedance matching can be set.

18 illustrates a circuit design of an antenna device 106D according to the sixth embodiment. In the antenna device 106D is the capacitor 36 between the first coupling element 31 the coupling circuit 30 and the food circle 1 switched (inserted). The further training is the same as in the 4 shown circuit in the first embodiment.

In the formation of the antenna device 106D can by the capacity of the inserted capacitor 36 the resonant frequency of the radiating element 10 can be adjusted and the level of impedance matching can be set. Because the capacitor 36 between the feeding circuit 1 and the first coupling element 31 In addition, there is a parasitic capacitance between the first coupling element 31 and the second coupling element 32 is generated, and the capacitor 36 connected in series in the structure. Accordingly, a combined capacitance included in a self-resonant circuit system is reduced, and the natural resonance frequency is increased. Thus, the self-resonance frequency can be excluded from the communication band to be used.

Seventh embodiment

19 (A) illustrates a circuit design of an antenna device 107A according to a seventh embodiment, and 19 (B) illustrates a circuit design of an antenna device 107B according to the seventh embodiment. The formation of these antenna devices 107A and 107B is the same as the one in 4 shown circuit in the first embodiment. If a self-inductance of the first coupling element 31 the coupling circuit 30 when L1 is represented and a self-inductance of the second coupling element 32 when L2 is represented have the first coupling element 31 and the second coupling element 32 the coupling circuit 30 in the antenna device 107A a ratio of L2 > L1 and have in the antenna device 107B a ratio of L2 <L1. Compared with a case in which L1 = L2, with the ratio of L2> L1, the resonant frequency of the nonradiative resonant circuit 20 be reduced. In a comparison with the same resonant frequency, alternatively, the non-radiative resonant circuit 20 be shortened.

If L2 > L1 also contributes compared to a design in which the inductor with the second coupling element 32 outside the coupling circuit 30 connected (added) is the entire second coupling element 32 with a relatively large self-inductance to the coupling with the first coupling element 31 at. Thus, a non-radiative resonant circuit 20 be increased to be supplied power component.

If L2 <L1, compared to a design in which the inductor bears with the first coupling element 31 outside the coupling circuit 30 connected (added) is also the entire first coupling element 31 with a relatively large self-inductance to the coupling with the second coupling element 32 at. Thus, a non-radiative resonant circuit 20 be increased to be supplied power component.

20 Figure 11 is an exploded plan view illustrating conductor patterns formed on layers of the coupling circuit 30 are formed according to this embodiment. The coupling circuit 30 included in an antenna device according to this embodiment is a rectangular parallelepiped-shaped chip component to be mounted on a circuit substrate.

On insulating materials S11 . S12 . S21 . S22 and S23 each are ladder patterns L11 . L12 . L21 . L22 and L23 educated. A first end of the ladder pattern L11 is with the radiating element connecting terminal PA and a second end thereof is over the layer connection conductor V1 with a first end of the conductor pattern L12 connected. A second end of the conductor pattern L12 is with the supply circuit connection terminal PF connected. A first end of the ladder pattern L21 is with the non-radiating resonant circuit connection terminal PS and a second end thereof is via a layer connection conductor V21 with a first end of the conductor pattern L22 connected. A second end of the conductor pattern L22 is via a layer connection conductor V22 with a first end of the conductor pattern L23 connected. A second end of the conductor pattern L23 is with the ground connection PG connected.

21 is a sectional view of the coupling circuit 30 , 22 is a plan view, in particular an overlap between the conductor pattern L12 and the conductor pattern L21 represents. A coil opening or a coil diameter of the conductor patterns L11 and L12 that is the first coupling element 31 is smaller than a coil opening or a coil diameter of the conductor patterns L21 . L22 and L23 that the second coupling element 32 form. In addition, parts of the line width of the conductor patterns overlap each other L11 and L12 as well as the conductor pattern L21 . L22 and L23 , At the in 21 and 22 example shown is overlapped approximately 1/2 width along the entire circumference.

23 Figure 11 is an exploded plan view illustrating conductor patterns formed on layers of another coupling circuit 30 are formed according to the seventh embodiment. The shape and size of the conductor patterns are different from those in the 20 illustrated example. Among the line patterns of the coupling circuit, which in 23 is shown, a coil outer diameter of the conductor pattern L11 and L12 that is the first coupling element 31 smaller than a coil inner diameter of the conductor patterns L21 . L22 and L23 that the second coupling element 32 form.

At the in 20 to 23 As shown, a parasitic capacitance is suppressed between the conductor patterns (FIG. L11 and L12 ), which is the first coupling element 31 form and conductor patterns ( L21 . L22 and L23 ), which is the second coupling element 32 form, is generated. Accordingly, the self-resonant frequency is increased by the inductances of the first coupling element 31 and the second coupling element 32 and the above parasitic capacitance is determined, and the natural resonance frequency can be excluded from the communication band to be used. Even if the conductor patterns ( L11 and L12 ), which is the first coupling element 31 form, and the conductor patterns ( L21 . L22 and L23 ), which is the second coupling element 32 form, are misaligned in a plane direction (the direction of the XY Level, as in 22 In addition, the portion where the coil opening of the first coupling member is continuously maintained is constantly maintained 31 and the coil opening of the second coupling element 32 overlap each other. Accordingly, a change in the coupling degree of the magnetic field coupling between the first coupling element 31 and the second coupling element 32 small, the change being caused by a misalignment of the conductor patterns ( L11 and L12 ), which is the first coupling element 31 form, and the conductor pattern ( L21 . L22 and L23 ), which is the second coupling element 32 form, in the plane direction.

Both examples in 20 and 23 are examples of a coupling circuit in which the ratio L1 <L2 described above is satisfied. If L1> L2, the first coupling element 31 be formed by conductor pattern whose coil opening is relatively large.

It is noted that 20 and 23 Illustrate examples in which the influence of a misalignment of the conductor patterns ( L11 and L12 ), which is the first coupling element 31 form, and the conductor pattern ( L21 . L22 and L23 ), which is the second coupling element 32 form, is reduced. Likewise, the influence on a misalignment of the conductor patterns, which may be the first coupling element 31 form, in the plane direction and the influence of misalignment of the conductor patterns, the second coupling element 32 to be reduced in the plane direction. For example, coil openings or coil diameter of the conductor pattern L11 and L12 , which are adjacent to each other in the stacking direction, are different from each other, and parts of their line width may overlap one another in the structure. Likewise, for example, coil openings or coil diameter of the conductor pattern L21 . L22 and L23 that are adjacent to each other in the stacking direction, are different from each other, and parts of the line width thereof may overlap each other in the structure.

Eighth embodiment

24 illustrates a circuit design of an antenna device 108 according to an eighth embodiment. The antenna device 108 comprises a first coupling circuit 30A . a second coupling circuit 30B , a first nonradiative resonant circuit 20A and a second nonradiative resonant circuit 20B , The second coupling circuit 30B comprises a third coupling element 33 and a fourth coupling element 34 which are connected to each other. The third coupling element 33 the second coupling circuit 30B is between the first coupling element 31 and the food circle 1 connected. The first non-radiative resonance circuit 20A is with the second coupling element 32 connected, and the second non-radiative resonant circuit 20B is with the fourth coupling element 34 connected. The further training is the same as in the 4 shown circuit in the first embodiment.

A resonant frequency of the first nonradiative resonant circuit 20A and a resonance frequency of the second nonradiative resonance circuit 20B are different from each other, and thus multiple poles are generated corresponding to these resonance frequencies, and a communication bandwidth is widened. When the resonant frequency of the first nonradiative resonant circuit 20A and the resonant frequency of the second nonradiative resonant circuit 20B In addition, the poles produced in the two nonradiative resonant circuits become deeper, and the impedance matching in this frequency band is improved.

Ninth embodiment

25 illustrates a circuit design of an antenna device 109 according to a ninth embodiment. The antenna device 109 includes the first coupling circuit 30A , the second coupling circuit 30B , the first nonradiative resonant circuit 20A and the second nonradiative resonant circuit 20B , The second coupling circuit 30B includes the third coupling element 33 and the fourth coupling element 34 which are connected to each other.

The third coupling element 33 is between the second coupling element 32 and the first nonradiative resonant circuit 20A connected. The first non-radiative resonance circuit 20A is with the second coupling element 32 connected, and the second non-radiative resonant circuit 20B is with the fourth coupling element 34 connected. The further training is the same as in the 4 shown circuit in the first embodiment.

In this embodiment, the resonant frequency of the first nonradiative resonant circuit 20A and the resonant frequency of the second nonradiative resonant circuit 20B substantially equal, and thus the poles generated in the two nonradiative resonant circuits become deeper, and the impedance matching in this frequency band is improved.

Tenth embodiment

26 Fig. 10 is a circuit diagram of an antenna device 110 according to a tenth embodiment. The antenna device 110 includes a switch 37 that is between the nonradiative resonant circuit 20 and ground is switched. The antenna device 110 also includes a switch 38 between the radiating element 10 and mass. The further training is the same as in the 4 shown circuit in the first embodiment.

The switches 37 and 38 are switched independently or in conjunction with each other. By changing the frequency of a pole by the provision of the coupling circuit 30 and the nonradiative resonant circuit 20 is generated, according to the state of the switch 37 , or by changing an adaptation state, the impedance matching can be improved. By changing the resonant frequency of the non-radiating resonant circuit 20 or by changing the impedance matching state between the coupling circuit 30 and the nonradiative resonant circuit 20 to a simple coupling of the non-radiating resonant circuit 20 to the feeding circuit 1 via the coupling circuit 30 In addition, the impedance matching can be improved.

In addition, according to the state of the switch 38 the frequency of a by resonance of the radiating element 10 generated pole are changed.

Eleventh embodiment

27 (A) and 27 (B) provide circuit designs of antenna devices 111A and 111B according to an eleventh embodiment. In the examples in both 27 (A) as well as 27 (B) are parasitic capacitances caused by capacitors cs1 and cs2 and the like are represented between the first coupling element 31 and the second coupling element 32 the coupling circuit 30 contain. In addition, the coupling circuit includes 30 an inductor L3 which is between the first coupling element 31 and the second coupling element 32 is switched.

The inductor L3 and the capacitors cs1 and cs2 parasitic capacitances are in parallel resonance. Accordingly, one in the coupling circuit 30 generated reactance component in this parallel resonant frequency band suppressed. Thus, a reactance component is formed by inserting the coupling circuit 30 is added, and a change from an adaptation state in which the coupling circuit 30 is not mounted, can be suppressed.

Twelfth embodiment

28 illustrates a circuit design of an antenna device 112 according to a twelfth embodiment.

The antenna device 112 According to this embodiment, the radiating element comprises 10 , the coupling circuit 30 and the nonradiative resonant circuit 20 , The feeding circle 1 is with the radiating element 10 connected. The coupling circuit 30 includes the first coupling element 31 that is between the radiating element 10 and ground, and the second coupling element 32 connected to the first coupling element 31 is coupled. The non-radiating resonant circuit 20 is with the second coupling element 32 connected. In addition, the inductor 35 in this example between the first coupling element 31 and mass inserted.

In the above embodiment, the radiating element interfere 10 and the nonradiative resonant circuit 20 not with respect to the radiation, and a radiation characteristic of the radiating element 10 is not influenced unfavorably. In addition, a frequency characteristic of a return loss of the radiating element becomes 10 , from the food circle 1 seen by a resonance characteristic of the nonradiative resonance circuit 20 is set, and in a desired frequency band, a pole is generated to broaden the band of the frequency characteristic of the antenna. Since a current is particularly high in a ground connected portion, the radiating element 10 and the nonradiative resonant circuit 20 via the coupling circuit 30 be coupled to each other. In addition, there is greater freedom in the arrangement of the coupling circuit 30 and the nonradiative resonant circuit 20 ,

Thirteenth embodiment

29 illustrates a circuit design of an antenna device 113 according to a thirteenth embodiment.

The antenna device 113 According to this embodiment, a substrate comprises 5 on which the coupling circuit 30 and the nonradiative resonant circuit 20 are each formed using conductor patterns. The further training is the same as in the 4 shown circuit in the first embodiment.

The substrate 5 is formed of a resin multi-layer substrate or a ceramic multi-layer substrate. For example, in the case of a resin multilayer substrate, a plurality of thermoplastic synthetic resin materials are stacked on surfaces of which copper foil patterns are formed, and pressed with heat. In the case of a ceramic multilayer substrate, a plurality of ceramic green sheets are stacked and fired on surfaces of which conductive pattern patterns are formed.

It is noted that in the case that the coupling circuit 30 and the nonradiative resonant circuit 20 are formed on different substrates, the non-radiative resonant circuit 20 may be formed by the resin multi-layer substrate or the ceramic multi-layer substrate.

Because the coupling circuit 30 and the nonradiative resonant circuit 20 are integrated with each other, the area used is reduced according to this embodiment.

Fourteenth embodiment

A twelfth embodiment is an antenna device comprising a PIFA (planar inverted F-shaped antenna) and a parasitic radiating element.

30 illustrates a circuit design of an antenna device 114 according to the fourteenth embodiment. The antenna device 114 According to this embodiment comprises a feeding radiation element 10A , a feed line 10AF , a parasitic radiation element 10B and the coupling circuit 30 , The feeding circle 1 is between the feeders 10AF and ground switched. Training and effect of the coupling circuit 30 are those described in the above embodiments.

The first coupling element 31 the coupling circuit 30 is between a connection point ps between the feed radiating element 10A and the feed line AF and ground switched. The power radiating element 10A , the feeders 10AF and the first coupling element 31 form a PIFA. This means the first coupling element 31 the coupling circuit 30 is provided on a section of a short pen of the PIFA. The short pen here connects the connection point ps and mass with each other. In this section, a capacitor or an inductor may be provided.

The parasitic radiation element 10B is a parasitic radiating element of the monopole type. The second coupling element 32 the coupling circuit 30 is near a ground end of the parasitic radiating element 10B inserted.

A resonant current iA of the feeding radiating element flows between an open end of the feeding radiating element 10A and a ground end of the first coupling element 31 , In addition, a resonance current flows iB between an open end of the parasitic radiating element 10B and a ground end of the second coupling element 32 , A phase of the in the feed radiating element 10A flowing electricity iA and a phase of in the parasitic radiating element 10B flowing electricity iB differ from each other.

When the phase of resonance of the feeding radiating element and the phase of resonance of the parasitic radiating element are the same, generally, a notch between the two resonance frequencies is in a frequency characteristic of the antenna device. Therefore, the band can not be widened even if the parasitic radiating element is provided. That is, the parasitic radiating element may not be provided adjacent to the feeding radiating element to widen the band.

In contrast, in this embodiment, in the first coupling element 31 the coupling circuit 30 flowing current and that in the second coupling element 32 flowing current a phase difference. The phase of resonance of the feed radiating element 10A and the phase of resonance of the parasitic radiating element 10B are therefore not the same, and thus there is no notch between the two resonance frequencies. The phase difference between the first coupling element 31 and the second coupling element 32 is at most 180 °, and a parasitic component produces a phase difference of less than or equal to 180 °. This means, by an effect of the parasitic capacitance between the first coupling element 31 and the second coupling element 32 is the phase difference between that in the first coupling element 31 flowing current and that in the second coupling element 32 flowing current greater than 0 ° and less than 180 °.

There, as in 30 represented, the resonance current iA between the open end and the short point in the PIFA flows, the phase of differ in the supply circuit 1 flowing current and the phase of the resonance current iA from each other. When the first coupling element 31 the coupling circuit 30 in the feed line 10AF is inserted and the parasitic radiation element 10B with the second coupling element 32 Accordingly, since there is no correlation between the phase of the parasitic radiation element 10B flowing electricity iB and the phase of in the feed radiating element 10A flowing electricity iA As described above, the resonance of the feeding radiating element 10A and the resonance of the parasitic radiating element 10B have the same phase, in which case the above notch is present. In this embodiment, such a problem does not arise and the parasitic radiating element 10B and the feeding radiating element 10A may be provided adjacent to each other.

Although this embodiment is an example of the formation of the feed radiating element as PIFA, the feed radiating element is not limited to a PIFA and may also be a typical inverted F-shaped antenna. The same or substantially the same effect can be achieved.

Fifteenth embodiment

A fifteenth embodiment illustrates an example of an antenna device comprising a plurality of parasitic radiating elements.

31 (A) illustrates a circuit design of an antenna device 115 according to the fifteenth embodiment. The antenna device 115 According to this embodiment, the feed radiating element comprises 10A , the feeders 10AF , the parasitic radiation element 10B , a parasitic radiation element 10C and the coupling circuit 30 , The feeding circle 1 is between the feeders 10AF and ground switched.

The parasitic radiation element 10C is to a ground end of the same mainly to the feed line 10AF to the feed radiating element 10A coupled. The further training is the same as in the 30 illustrated antenna device 114 ,

31 (B) represents a frequency characteristic of a return loss of in 31 (A) illustrated antenna device 115 and an antenna device according to a comparative example. In 31 (B) For example, a return loss characteristic RL1 is a return loss of the antenna device 115 According to this embodiment, and a return loss characteristic RL2 is a return loss of the antenna device according to the comparative example. The antenna device according to the comparative example is an antenna device in which the coupling circuit 30 and the parasitic radiating element 10B are not included and the first coupling element 31 only serves as a short pen of a PIFA. In both antenna devices, a pole is at a center frequency F1 a low band generated. The reason for this is a resonance of the feeding radiation element 10A of 1/4 wavelength. Another pole is at one frequency F2 generated. The reason for this is a resonance of the feeding radiation element 10A of 3/4 wavelengths. Again, another pole is at one frequency F3 generated. The reason for this is a resonance of the parasitic radiating element of the monopole type 10C of 1/4 wavelength.

In the antenna device 115 according to this embodiment is also at a frequency F0 creates a pole. The reason for this is a resonance characteristic of the parasitic radiating element 10B , In this way it is possible to form an antenna device which is the parasitic radiating element 10B that with the coupling circuit 30 is connected, and the parasitic radiation element 10C includes, that no coupling of the coupling circuit 30 between switches.

In addition, in this embodiment, the feed radiating element is not limited to a PIFA and may also be a typical inverted F-shaped antenna. The same or substantially the same effect is achieved.

Sixteenth embodiment

A sixteenth embodiment illustrates an example of an antenna device comprising a plurality of parasitic radiating elements.

32 illustrates a circuit design of an antenna device 116 according to the sixteenth embodiment. The antenna device 116 According to this embodiment, the feed radiating element comprises 10A , the feeders 10AF , a short pen 10AS , the parasitic radiating elements 10B and 10C and the coupling circuit 30 , The power radiating element 10A is a radiating element of a PIFA.

In this embodiment, the first coupling element 31 the coupling circuit 30 around the ground end of the parasitic radiating element 10B inserted, and the second coupling element 32 the coupling circuit 30 is around the ground end of the parasitic radiating element 10C inserted. The parasitic radiation element 10B is at the ground end of the same mainly to the feed line 10AF to the feed radiating element 10A coupled.

According to this embodiment, the two parasitic radiating elements 10B and 10C be formed so that the same via the coupling circuit 30 are coupled together.

It is noted that in this embodiment, the feeding radiating element is not limited to a PIFA or an inverted F-shaped antenna and, for example, is a monopole-type radiating element. That is, each feed radiating element connected to the parasitic radiating element 10B or the like, may be used, and the same or substantially the same effect is achieved.

Seventeenth embodiment

33 illustrates a circuit design of an antenna device 117 according to a seventeenth embodiment. The antenna device 117 according to this embodiment comprises feed radiation elements 10U and 10V , the feeders 10AF , the parasitic radiation element 10B , the parasitic radiation element 10C and the coupling circuit 30 , The feeding circle 1 is between the feeders 10AF and ground switched. Training and effect of the coupling circuit 30 are those described in the above embodiments.

The power radiating elements 10U and 10V and the feeders 10AF Form a branch feed monopole antenna or a branch feed PIFA. The parasitic radiation element 10C is mainly with the feeders 10AF coupled to serve as a monopole type antenna or an inverted L-shaped type antenna.

34 represents a frequency characteristic of a return loss of the antenna device 117 in 34 creates a pole by a frequency F1 is indicated, mainly by a fundamental wave, in the feed radiating element 10U and the feedline 10AF is generated in a branch antenna, which by the feed radiation elements 10U and 10V and the feeders 10AF is formed. A pole by a frequency F2 is indicated by a fundamental wave generated in the parasitic radiating element 10C is generated. One by one frequency F3 For example, the indicated pole location is primarily caused by a 3/4 wavelength harmonic present in the feed radiating element 10U and the feedline 10AF is generated. A pole by a frequency F4 is indicated, is formed by a in the parasitic radiation element 10B generated fundamental wave. A pole by a frequency F5 indicated, is mainly due to resonance occurring in the feed radiating element 10V in through the feed radiating elements 10U and 10V and the feeders 10AF formed branch antenna is generated.

It is noted that a parasitic capacitance between the feed radiating element 10V and the parasitic radiating element 10B is actively generated so that a phase difference of resonance current between the feeding radiation element 10V and the parasitic radiating element 10B designed with about 90 °. Thus, one by the frequency F4 indicated pole of the power radiating element 10V and one by the frequency F5 indicated pole of the parasitic radiation element 10B generated.

By receiving the branch antenna with the feed radiating element 10V For example, in the antenna device according to this embodiment, a communication band widened to 2700 MHz can be covered, and a broadband antenna covering a low band of 700 MHz to 900 MHz and a high band of 1700 MHz to 2700 MHz can be formed.

Eighteenth embodiment

35 Fig. 10 is a circuit diagram of an antenna device 118A according to an eighteenth embodiment. In the antenna device 118A is the parasitic radiation element 10B formed on a side surface portion of the metal housing. The second coupling element 32 the coupling circuit 30 is with the parasitic radiating element 10B connected. The further training is the same as in the 4 shown circuit in the first embodiment.

In the structure of the antenna device 118A can the parasitic radiation element 10B from the radiating element 10 be separated, and it can be achieved a good radiation characteristic at a resonant frequency by the coupling circuit 30 and the parasitic radiating element 10B is added. Furthermore, the radiation characteristic of the radiation element 10 not deteriorated at frequencies other than the resonant frequency.

36 is a circuit diagram of another antenna device 118B according to the eighteenth embodiment. In the antenna device 118B is the parasitic radiation element 10B formed on a side surface portion of the metal housing. An end portion of the parasitic radiating element 10B is about the inductor 8th connected to ground of a circuit substrate or the like (grounded). The parasitic radiation element 10B serves as a 1/2-wavelength resonant antenna.

Since the tip of the side surface portion of the metal shell is grounded, in the structure of the antenna device 118B Variations in the antenna characteristic due to a change in the environment are suppressed. Also, in the case where there is a side face part of another metal case grounded through a slot in front of the tip of the side face part of the metal case, since the tip of the side face part of the metal case is grounded, a field maximum point moves from the tip of the parasitic one radiating element 10B to a center, and a good radiation characteristic can be obtained at a resonance frequency generated by the coupling circuit 30 is added. Furthermore, the resonance frequency can be easily controlled by the inductance of the inductor 8th be set.

37 is a circuit diagram turn another antenna device 118C according to the eighteenth embodiment. In the antenna device 118C is the feed radiating element 10A from an end face part of the metal shell to a side face part thereof. Likewise, the parasitic radiation element 10B from an end face part of the metal shell to a side face part thereof. In this way, the main part of the parasitic radiation element 10B are located on the end surface portion of the metal housing. In addition, the parasitic radiation element 10B near a ground end of the power radiating element 10A lie. As a field maximum point of the feeding radiation element 10A moving from the ground to a center, with this structure, unnecessary interference between the feed radiating element 10A and the parasitic radiating element 10B be suppressed.

Nineteenth embodiment

38 is a plan view of a main part of an antenna device 119 according to a nineteenth embodiment.

A metal housing of electronic equipment includes the radiating element 10 which is an end portion of the metal housing. A connection position of the feed line 9 for the radiating element 10 and a location of the nonradiative resonant circuit 20 differ from those at the in 2 illustrated antenna device 101 in the first embodiment.

In this embodiment, in plan view of the circuit substrate 6 , the feeders 9 with the left side panel part 13 of the radiating element 10 connected. Accordingly, the nonradiative resonant circuit 20 on the right side of the coupling circuit 30 arranged. This Positional relationship is to that in 2 example shown in reverse (symmetric ratio). The further embodiment is the same as that shown in the first embodiment.

39 is a perspective view of the coupling circuit 30 according to this embodiment. The outer shape of the coupling circuit 30 is represented by a two-dot chain line. On an outer surface of the coupling circuit 30 are the supply circuit connection terminal PF , the radiating element connecting terminal PA , the ground connection PG and the non-radiating resonant circuit connection terminal PS educated. The coupling circuit 30 is essentially the same as the one in 1 illustrated coupling circuit 30 in the first embodiment. However, the second surface is MS2 the mounting surface, and this surface faces the circuit substrate. On an upper surface (first surface), one towards the mounting surface (second surface) MS2 is opposite surface, the direction discrimination mark DDM is provided. Thus, the location of the terminals differs from that in FIG 1 illustrated coupling circuit 30 in plan view. In the in 1 illustrated coupling circuit 30 are, in plan view, the ground connection PG , the feed connection terminal PF and the nonradiative resonant circuit connection terminal PS from the radiating element connecting terminal PA made in this order in a clockwise direction. In the nineteenth embodiment, as in 39 shown are the ground connection PG , the feed connection terminal PF and the nonradiative resonant circuit connection terminal PS from the radiating element connecting terminal PA made out in this order counterclockwise.

As described above, since the first end and the second end of the first coupling element and the first end and the second end of the second coupling element are on both the first surface MS1 as well as the second surface MS2 are formed, either the first surface or the second surface can serve as a mounting surface. Accordingly, either the first surface MS1 or the second surface MS2 the coupling circuit 30 be selected as a mounting surface for mounting on a circuit substrate, so that the terminals are arranged at locations that are suitable for the location of a circuit or an element, with those on the coupling circuit 30 formed first coil and second coil are connected.

In the 8th and 39 Examples shown represent examples in which layer connection conductors, which are the four on the first surface MS1 formed connections and the four on the second surface MS2 connect formed terminals together, are formed on end surfaces of the multi-layer body. However, multiple via conductors may be formed within the multilayer body, and the four on the first surface MS1 formed connections and the four on the second surface MS2 formed connections can be connected to each other via these via conductors.

In addition to the formation of the above via conductors, on the mounting surface of the coupling circuit 30 LGA connections (land grid array) may be formed.

40 represents an embodiment of another coupling circuit 30 according to this embodiment and is an exploded plan view, the on layers of the coupling circuit 30 represents formed conductor patterns.

As in 40 shown are the first conductor pattern L11 , the second ladder pattern L12 , the third ladder pattern L21 and the fourth conductor pattern L22 each on the insulating material S11 , the insulating material S12 , the insulating material S21 and the insulating material S22 educated. The insulating materials S11 . S12 . S21 and S22 are stacked so that these coil conductor patterns are arranged from a layer near the mounting surface in the following order: first conductor pattern L11 , second ladder pattern L12 , third ladder pattern L21 and fourth conductor pattern L22 ,

A first end of the first conductor pattern L11 is with the radiating element connecting terminal PA and a second end thereof is over the layer connection conductor V1 with a first end of the second conductor pattern L12 connected. A second end of the second conductor pattern L12 is with the supply circuit connection terminal PF connected. A first end of the third conductor pattern L21 is with the non-radiating resonant circuit connection terminal PS connected, and a second end of the third conductor pattern L21 is over the layer connection conductor V2 with a first end of the fourth conductor pattern L22 connected. A second end of the fourth conductor pattern L22 is with the ground connection PG connected.

The conductor patterns on the in 40 layers shown are in the 9 illustrated conductor patterns in a symmetrical relationship. Thus, in the coupling circuit comprising these conductor patterns, in plan view, the ground terminal PG , the feed connection terminal PF and the non-radiating resonant circuit connection terminal PS from the radiating element connecting terminal PA made out in this order counterclockwise.

According to this example, the terminals may be located at locations suitable for the location of a circuit or element with those in the coupling circuit 30 formed first coil and second coil are connected.

Twentieth Embodiment

A twentieth embodiment is an antenna device further comprising a phase shifter.

41 Fig. 10 is a circuit diagram of an antenna device 120 according to the twentieth embodiment, in which the feeding circuit 1 connected is. In the antenna device 120 is between the food circle 1 and the first coupling element 31 the coupling circuit 30 a phase shifter 50 connected. The phase shifter 50 is a phase shifter by which a phase shift amount changes depending on the frequency (has frequency dependency). The phase shifter 50 includes a first coil Lp , a second coil ls and a capacitor C3 which are connected to each other.

It is noted that in this example capacitors C4 and C5 for impedance matching between the supply circuit 1 and the phase shifter 50 are switched.

The formation of the coupling circuit 30 , of the radiating element 10 and the nonradiative resonant circuit 20 is the same as that shown in the first embodiment.

42 Fig. 12 is an equivalent circuit diagram showing the above phase shifter 50 represents, being an ideal transformer IT and parasitic inductance components (serial parasitic inductance components La and Lc and parallel parasitic inductance component lb ) are shown separately.

The coupling coefficient between the first coil Lp and the second coil ls , in the 41 is lower than in a conventional high-frequency converter, and accordingly, the serial parasitic inductance component Lc large. However, because the capacitance of the capacitor C3 is also large, an impedance matching is guaranteed. In addition, because the capacity of the capacitor C3 is large, is a proportion of a high-band signal that the capacitor C3 bypasses, higher than the one through the first coil Lp and the second coil ls formed transformer bypasses, and a phase shift effect of the transformer is small. With respect to the low band, on the other hand, the amount that is the capacitor C3 bypasses, relatively small, and the phase shift effect of the transformer is large. Thus, the coupling coefficient is determined so that the phase shift amount with respect to a low-band signal is substantially 180 degrees and the phase shift amount with respect to a high-band signal is substantially 90 degrees.

43 represents a frequency characteristic of the phase shift amount of the phase shifter 50 In this example, the phase shift amount in a low band (700 MHz to 900 MHz band) is substantially 180 °, and the phase shift amount in a high band (1.7 GHz to 2.7 GHz band). Band) is essentially 90 °.

Next, effects caused by the provision of the phase shifter will be described 50 together with the coupling circuit 30 be achieved. 44 (A) Fig. 12 is a circuit diagram of the antenna device shown in the first embodiment which is not the phase shifter 50 includes, and 44 (B) represents impedance locations on a Smith chart with impedances from the supply circuit 1 represented antenna device.

45 (A) Fig. 10 is a circuit diagram of an antenna device with added phase shifter 50 , and 45 (B) represents impedance locations on a Smith chart with impedances from the supply circuit 1 represented antenna device. This antenna device is a circuit that is not the capacitor C5 in the in 41 illustrated circuit comprises.

46 (A) FIG. 12 is a circuit diagram of an antenna device including the impedance matching capacitor. FIG C5 includes (the same as in 41 shown), and 46 (B) represents an impedance location which, on a Smith chart, imparts impedance from the sense circuit 1 represented antenna device represents.

In 44 (B) is the place T0 an impedance location of an antenna device according to a comparative example, wherein the coupling circuit 30 and the nonradiative resonant circuit 20 are not provided, and the place T1 is an impedance location of the antenna device used in 44 (A) is shown. Both are results from a run of 1.7 GHz to 2.7 GHz. How out 44 (B) is apparent by the inclusion of the coupling circuit 30 and the nonradiative resonant circuit 20 As described above, in the frequency characteristic of the antenna, a pole is generated (small loop in the diagram), and accordingly, the resonance frequency band moves to the center of the diagram. It is noted that a higher frequency band is still in a peripheral area of the Diagram is located and an adjustment in the high frequency band turns out to be difficult.

In 45 (B) is the place T2 an impedance location of the antenna device containing the phase shifter 50 , the coupling circuit 30 and the nonradiative resonant circuit 20 includes, and the place T1 is the same as the one in 44 (A) illustrated place T1 , Both are results from a run of 1.7 GHz to 2.7 GHz. How out 45 (B) shows, moves by the inclusion of the phase shifter 50 the phase continues in a low band by substantially 180 °, and in a high band, the phase advances substantially 90 °. Accordingly, the high frequency band also moves to the center of the diagram.

In 46 (B) is the place T3 an impedance location of the antenna device used in 46 (A) is shown, and is a result of a run from 1.7 GHz to 2.7 GHz. As from a comparison with the in 45 (B) place shown T2 As can be seen, the high frequency band rotates through a function of the capacitor C5 which is shunted, clockwise. Thus, the adaptation is improved in all frequency bands.

47 represents a frequency characteristic of a return loss of in 44 (A) and 46 (A) shown antenna devices and the antenna device according to the comparative example. In 47 is a return loss characteristic RL1 a return loss characteristic of the antenna device according to the comparative example, in which the coupling circuit 30 and the nonradiative resonant circuit 20 not included, a return loss characteristic RL2 is a return loss characteristic of in 44 (A) shown antenna device, and a return loss characteristic RL3 is a return loss characteristic of in 46 (A) illustrated antenna device. The return loss characteristics RL1 and RL2 in 47 are the same as the ones in 6 shown. In a comparison of the return loss characteristics RL2 and RL3 proves that the return loss in all bands is small and that the high band is widened with the same radiating element to a broad band of, for example, 1.4 GHz to 2.6 GHz.

48 is an external perspective view of the phase shifter 50 , and 49 is a plan view of layers in the phase shifter 50 , Besides that is 50 a sectional view of the phase shifter 50 ,

An upper surface made of a material S1 corresponds to a mounting surface (lower surface) of a multilayer body 100 , On the material S1 are a connection T1 as the first goal P1 , a connection T2 as a second goal P2 , a ground connection G and an open port NC formed.

The material layers of the multilayer body 100 may be a nonmagnetic ceramic multilayer body formed of LTCC or the like, or may be a synthetic resin multilayer body formed of a synthetic resin material such as polyimide or liquid crystal polymer. In this way, since the material layers are nonmagnetic (no magnetic ferrite), it is possible to use the material layers as a transformer and phase shifter having a predetermined inductance and a predetermined coupling coefficient even in a high frequency band exceeding several hundreds MHz.

The above conductor patterns and the layer connection conductors are each formed of a conductor material containing Ag or Cu as a main component and having a low resistivity. For example, in the case that the material layers are ceramic, the conductor patterns and the layer connection conductors are formed by screen printing and firing a conductive paste containing Ag or Cu as a main component. For example, in the case where the material layers are made of synthetic resin, the conductor patterns and the layer connection conductors are patterned by etching or the like a metal foil such as an Al foil or a Cu foil.

The phase shifter 50 includes several insulating materials S1 to S9 , On the materials S1 to S9 Different conductor patterns are formed. The "different conductor patterns" include not only conductor patterns formed on surfaces of the materials, but also layer connection conductors. The layer connection conductors include not only via conductors but also end surface electrodes formed on end surfaces of the multilayer body.

The upper surface of the material S1 corresponds to the mounting surface (lower surface) of the multi-layer body. On the material S1 are the connection T1 as the first goal P1 , the connection T2 as a second goal P2 , the ground connection G and the open port NC formed.

On the materials S5 and S4 are the leaders L1A1 respectively L1A2 educated. On the material S3 are the leaders L1A3 and L1B1 educated. On the material S2 are the leaders L1B2 and L1C educated.

A first end of the ladder L1A1 is with the connection T1 connected as the first gate. A second end of the ladder L1A1 is about one Layers connecting conductors V11 with a first end of the conductor L1A2 connected. A second end of the ladder L1A2 is via a layer connection conductor V12 with a first end of the conductor L1A3 connected. A second end of the ladder L1A3 is with a first end of the ladder L1B1 connected. The second end of the ladder L1A3 and the first end of the ladder L1B1 are over a layer connection conductor V13 with a first end of the conductor L1B2 connected. A second end of the ladder L1B1 is via a layer connection conductor V14 with a second end of the conductor L1B2 connected. The second end of the ladder L1B2 is with a first end of the ladder L1C connected. A second end of the ladder L1C with the ground connection G connected.

On the materials S6 and S7 are the leaders L2A1 respectively L2A2 educated. On the material S8 are the leaders L2A3 and L2B1 educated. On the material S9 are the leaders L2B2 and L2C educated.

A first end of the ladder L2A1 is with the connection T2 connected as a second goal. A second end of the ladder L2A1 is via a layer connection conductor V21 with a first end of the conductor L2A2 connected. A second end of the ladder L2A2 is via a layer connection conductor V22 with a first end of the conductor L2A3 connected. A second end of the ladder L2A3 is with a first end of the ladder L2B1 connected. The second end of the ladder L2A3 and the first end of the ladder L2B1 are over a layer connection conductor V23 with a first end of the conductor L2B2 connected. A second end of the ladder L2B1 is via a layer connection conductor V24 with a second end of the conductor L2B2 connected. The second end of the ladder L2B2 is with a first end of the ladder L2C connected. A second end of the ladder L2C is with the ground connection G connected.

The ladder L1A1 . L1A2 . L1A3 . L1B1 . L1B2 and L1C as well as the layer connection conductors V11 . V12 . V13 and V14 form the first coil Lp , In addition, the ladder form L2A1 . L2A2 . L2A3 . L2B1 . L2B2 and L2C as well as the layer connection conductors V21 . V22 . V23 and V24 the second coil ls , Both the first coil Lp as well as the second coil ls are right-angled helical coils.

Twenty-first embodiment

A twenty-first embodiment represents a radiating element having a structure different from that of the radiating element shown in the first embodiment.

51 is a plan view of part of a metal housing of electronic equipment. The metal housing of electronic equipment includes the radiating element 10 which is an end portion of the metal shell and the metal shell body 40 , Although in the first embodiment, an example has been shown in which the end portion of the metal housing with three sides in plan view as a radiation element 10 is used as in 51 shown, the radiating element 10 be formed by a planar end portion of the metal housing.

52 (A) and 52 (B) FIG. 15 are perspective views of parts of metal housings of various electronic equipment. FIG. At an in 52 (A) The example shown has the radiating element 10 which is the end portion of the metal housing, one to the XY Level parallel plane and one to the Y Z Plane parallel plane. At an in 52 (B) The example shown has the radiating element 10 which is the end portion of the metal housing, one to the XY Level parallel plane, one to the Y Z Plane parallel plane and two to the XZ Level parallel planes.

As in 52 (A) and 52 (B) shown, the end portion of the metal housing may have different shapes.

Further embodiments

In the above-described embodiments, examples were shown in which the end portion of the metal shell is used as the radiating element. However, a part of the radiating element or the entire radiating element may be a conductor pattern formed on a circuit substrate or the like, or may be a component other than the housing.

Although that in 4 illustrated example illustrates a case in which a parasitic capacitance between one end of the radiating element 10 and ground, a low impedance capacitance in a high band may be actively provided at that location to cause the radiating element 10 serves as PIFA. In addition, the location where the capacitance is generated may be grounded to form a PIFA.

The linear conductor pattern of the nonradiative resonant circuit 20 is not limited to a shape that returns in the middle, and may extend only in one direction. Alternatively, the non-radiative resonant circuit 20 may be bent to an L-shape or may be curved. Furthermore, the non-radiative resonance circuit 20 include a ladder pattern that is divided into several Branches divides. Thus, multiple poles can be generated.

In addition, at the non-radiative resonant circuit 20 a tip of the linear conductor pattern may be connected to ground to form a circuit similar to a short stub.

In the above-described examples, mainly examples of the use of a fundamental wave resonance of the non-radiative resonance circuit became 20 described. However, any harmonic resonance of the nonradiative resonant circuit 20 also be used, such as double-wave resonance (secondary resonance), three-wave resonance (tertiary resonance) or 3/2-wave resonance. In addition, both the fundamental resonance and the harmonic resonance can be used, or several harmonic resonances can be used.

Likewise, with respect to the radiating element 10 also any harmonic resonance such as double-wave resonance (secondary resonance), three-wave resonance (tertiary resonance) or 3/2-wave resonance can be used. In addition, both the fundamental resonance and the harmonic resonance can be used, or several harmonic resonances can be used.

The embodiments described above have described a so-called smartphone or electronic equipment with the same shape as a smartphone. However, the embodiments are similarly applicable to various pieces of electronic equipment such as a mobile phone including a feature phone, a wearable terminal including a smart watch and smart glasses, a laptop PC, a tab-end terminal, a camera, and the like Game console, a toy or the like.

Finally, the embodiments described above are in all respects exemplary embodiments and not restrictive. Any transformation and modification may be made by a person skilled in the art. The scope of the present invention is defined by the claims and not by the embodiments described above. In addition, the scope of the present invention further includes modifications of the embodiments within a scope equivalent to the claims.

LIST OF REFERENCE NUMBERS

C, C1

capacity

C3,

C4 capacitor

C5

Impedance matching capacitor

C11, C12

Capacity Formations conductor pattern

C20

capacitor

Cs1, Cs2

Capacitor (parasitic capacitance)

G

ground connection

GND

ground electrode

GZ

mass region

IT

ideal transformer

L1A1, L1A2, L1A3, L1B1, L1B2, L1C

ladder

L2A1, L2A2, L2A3, L2B1, L2B2, L2C

ladder

La, Lc

serial parasitic inductance component

lb

parallel parasitic inductance component

Lp

first coil

ls

second coil

L11

first ladder pattern

L12

second conductor pattern

L20

inductor

L21

third conductor pattern

L22

fourth conductor pattern

L21 to L23

conductor pattern

L3

inductor

MS1

first surface

MS2

second surface

NGE

Non-mass region

PA

Radiating element connection port

PF

Feed circuit connecting terminal

PG

ground connection

PS

Not bright resonant circuit connecting terminal

ps

junction

P1

first goal

P2

second goal

S1 to S9

material

S11, S12, S21, S22, S23

insulating material

T1, T2

connection

V1, V2

Layers connecting conductors

V11, V12, V13, V14, V21, V22, V23, V24

Layers connecting conductors

1

supply circuit

5

substratum

6

circuit substrate

7

Interconnects

8th

inductor

9

feeder

10

radiating element

10A

Feeding radiation element

10AF

feeder

10AS

short pen

10B, 10C

parasitic radiation element

10U, 10V

Feeding radiation element

10Z

Radiating element formation region

11

end surface portion

12, 13

Side surface portion

20

Not bright resonant circuit connecting terminal

20A

first nonradiative resonant circuit

20B

second nonradiative resonant circuit

20FB

Return member

21

first linear conductor pattern part

22

second linear conductor pattern part

30

coupling circuit

30A

first coupling circuit

30B

second coupling circuit

31

first coupling element

32

second coupling element

33

third coupling element

34

fourth coupling element

35

inductor

36

capacitor

37, 38

counter

40

Metal housing main part

41

planner part

42, 43

Side surface portion

50

phase shifter

100

Multi-layer body

101 to 105

antenna device

106A to 106D

antenna device

107A, 107B

antenna device

108 to 110

antenna device

111A, 111B

antenna device

112 to 117

antenna device

118A to 118C

antenna device

119, 120

antenna device

Claims (19)

Antenna device having the following features: a radiating element; a coupling circuit comprising a first coupling element and a second coupling element, the first coupling element being connected between the radiating element and a feeding circuit, the second coupling element being coupled to the first coupling element; and a nonradiative resonant circuit connected to the second coupling element, wherein a frequency characteristic of a return loss of the radiating element is set by a resonance frequency characteristic of the non-radiating resonant circuit. Antenna device according to Claim 1 further comprising a mass, wherein a direction of a magnetic field generated when current in the first coupling element flows in a direction from a terminal connected to the feeding circuit to a terminal connected to the radiating element is opposite to a direction of a magnetic field, generated when current in the second coupling element flows in a direction from a terminal connected to the nonradiative resonant circuit to a grounded terminal. Antenna device according to Claim 1 or 2 wherein the first coupling element and the second coupling element are multi-layered coil conductor patterns and the coupling circuit forms a transformer in which the first coupling element and the second coupling element are electromagnetically coupled to each other. Antenna device according to one of Claims 1 to 3 in which one half or more of the nonradiative resonant circuit is included in plan view of the radiating element in a formation region of the radiating element. Antenna device according to one of Claims 1 to 3 in which the radiating element is formed by a conductive component, which forms three sides in plan view, and the nonradiative resonant circuit is surrounded in plan view by the three sides. Antenna device according to one of Claims 1 to 5 in which the nonradiative resonant circuit is formed by a linear conductor pattern having a return part at a center. Antenna device according to Claim 6 wherein the conductor pattern comprises a first linear conductor pattern portion extending from the coupling circuit and a second conductor pattern portion returning at the return portion to be distant from the radiating element. Antenna device according to one of Claims 1 to 7 further comprising a phase shifter connected between the feed circuit and the first coupling element and having a frequency dependency. Antenna device according to Claim 1 further comprising a ground, wherein a second terminal of the second coupling element is grounded, the second terminal facing a first terminal to which the nonradiative resonant circuit is connected, and a length of a lead between the first coupling element and the feeding circuit and a length of a line between the second terminal of the second coupling element and ground are each less than 1/8 wavelength of a resonance frequency. Antenna device according to one of Claims 1 to 9 further comprising an inductor connected between the second coupling element and the nonradiative resonant circuit. Antenna device according to Claim 9 further comprising an inductor connected between the first terminal of the second coupling element and ground. Antenna device according to one of Claims 1 to 11 further comprising a capacitor connected between the second coupling element and the nonradiative resonant circuit. Antenna device according to Claim 9 further comprising a capacitor connected between the first terminal of the second coupling element and ground. Antenna device according to one of Claims 1 to 13 further comprising: a second coupling circuit comprising a third coupling element and a fourth coupling element, the third coupling element being connected between the first coupling element and the supply circuit, the fourth coupling element being coupled to the third coupling element; and a second nonradiative resonant circuit connected to the fourth coupling element. Antenna device according to one of Claims 1 to 13 further comprising: a second coupling circuit comprising a third coupling element and a fourth coupling element, the third coupling element being connected between the second coupling element and the nonradiative resonant circuit, the fourth coupling element being coupled to the third coupling element; and a second nonradiative resonant circuit connected to the fourth coupling element. Antenna device according to Claim 1 further comprising: a ground and a switch connected between the non-radiative resonant circuit and ground. Antenna device according to one of Claims 1 to 16 in which the coupling circuit comprises a parasitic capacitance and wherein the antenna device further comprises an inductor connected to the coupling circuit and which suppresses a reactance component generated in the coupling circuit by parallel resonance with the parasitic capacitance. An antenna device comprising: a radiating element to which a feeding circuit is connected; a coupling circuit comprising a first coupling element and a second coupling element, the first coupling element being connected between the radiating element and ground, the second coupling element being coupled to the first coupling element; and a nonradiative resonant circuit connected to the second coupling element, wherein a frequency characteristic of a return loss of the radiating element is set by a resonance frequency characteristic of the non-radiating resonant circuit. Electronic equipment comprising the antenna device according to one of the Claims 1 to 18 , the supply circuit connected to the coupling circuit and a housing in which the supply circuit is accommodated, wherein a part of the radiation element or the entire radiation element is a part of the housing.

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