DE102017010514A1 - Reception antenna for satellite navigation on a vehicle - Google Patents

Abstract

An antenna (1) for receiving circularly polarized satellite radio signals, comprising at least one horizontally oriented conductor loop arranged above a conductive base (6), with an arrangement for electromagnetic excitation (3) of the conductor loop connected to an antenna connection point (5), comprising the following features :
the conductor loop is designed as a ring line radiator (2) running through a polygonal or circular closed ring line (14) in a horizontal plane with the height h <0.15 of the free space wavelength λ over the conductive base surface (6),
- The ring line radiator (2) forms a resonant structure and is excitable by electromagnetic excitation in such a way that adjusts the current distribution of a current line wave in a single direction of rotation on the ring line (14), the phase difference over a cycle is just 2π
- There are distributed on the circumference of the ring line radiator (2), at ring line crosspoints (7) with the ring line radiator (2) galvanically coupled, vertical and the conductive base (6) extending vertical radiator (4) available
there is a distribution and phase shifting network (8) with a gate T1 coupled on the input side with the antenna connection point (5) and with two ports T2, T3 whose transmission coefficients S ₁₂ and S ₁₃ differ from each other by a phase angle ΔΦ
characterized by the following features
- The ring line radiator (2) is designed as a short ring line (14) whose circumference (L) is shorter than the free space wavelength λ.
there are at least three vertical radiators (4) as vertical resonance radiators (4, 4a, 4b, 4c,...), which are coupled via reactance circuits (13) with capacitive reactance X to the base (6), through which the Resonance of the short loop (14) is made.
- The input resistance of each of the ports T2, T3 of the distribution and phase shifter network (8) for electromagnetic excitation of the line shaft on the ring line (14) is in each case a substantially real resistance.
- The distribution and phase shifter network (8) two of the ring line coupling points (7a, 7b) are assigned in such a way that each of the gates T2, T3 of the distribution and phase shifter network (8) coupled to its associated ring line cross-point (7a, 7b) is.

Description

The invention relates to an antenna 1 for the reception of circularly polarized satellite radio signals, in particular for satellite radio navigation.

Satellite radio signals are transmitted due to polarization rotations in the transmission path usually with circularly polarized electromagnetic waves and are used in all known satellite navigation systems. Modern navigation systems provide, in particular for the global accessibility in connection with a high navigation accuracy in the mobile navigation, to evaluate the simultaneously received radio signals of several satellite navigation systems. Such composite-receiving systems are collectively referred to as GNSS (Global Navigation Satellite System) and include known systems such as GPS (Global Positioning System, GLONASS, Galileo and Beidou, etc.) Satellite antennas for vehicle navigation are typically used on the Circularly polarized satellite receiving antennas are used, as described, for example, in DE102009040910.6, the DE-A-4008505 which are known from A-1016379. For the construction on vehicles are particularly those antennas, which are characterized by a low overall height in conjunction with cost-effective manufacturability. These include, for example, in particular the known from DE102009040910 designed as a resonant structure ring line radiator with a small volume, which is imperative especially for mobile applications. The antenna has a small footprint and is very low with a height of less than one tenth of the free space wavelength.

As further receiving antennas for satellite navigation on vehicles according to the prior art patch antennas 14 known, however, which are complex compared to punched from sheet metal antennas in construction. A challenge to the satellite antennas for GNSS is the requirement of a large frequency bandwidth, which, for example, in GPS by the frequency band L1 with the center frequency 1575 MHz (required bandwidth about 80 MHz) and the frequency band L2 with the center frequency 1227 MHz (required bandwidth about 53 MHz) is given. This need is for example by separate, one of the frequency bands L1 or. L2 assigned antenna, or a broadband antenna comprising both frequency bands covered. Systems for the simultaneous evaluation of signal contents in the frequency bands L1 and L2 place particularly high demands on the antennas. And this with a small amount of available space, as it is always given especially in vehicle construction. The use of separate, in close proximity antennas for the two frequency bands involves the problem of mutual electromagnetic coupling with the effect of influencing the directional diagrams and the polarization purity and in particular the cross polarization. Due to the signals of the positioning satellites, which are incident at low elevation angles, suppression of the opposite direction of polarization - the cross polarization (LHCP) - is of decisive importance with regard to correct location results, even with sufficient gain in the desired, mostly right-handed circular polarization direction (RHCP). The accuracy of the location result is thus particularly affected by the ratio of the desired direction of polarization to the cross polarization of the satellite receiving antenna, so the Kreuzpolarisationsabstand. On the other hand, the realization of a satellite navigation antenna is technically difficult, which covers both frequency bands with a bandwidth of about 360 MHz at a center frequency of about 1385 MHz and still meets the sometimes stringent requirements for the Kreuzpolarisationsabstand and the antenna gain. Satellite receiving antennas for satellite navigation are particularly intended for installation on horizontal surfaces of the electrically conductive vehicle body. With respect to the antenna properties, the substantially horizontal vehicle roof acts as a conductive base.

For use on vehicles are particularly satellite receiving antennas with a small volume. Antennas of this type according to the prior art are known as patch antennas. However, these are less efficient in terms of reception at a low elevation angle and more complex in construction. This disadvantage is partially remedied by loop antennas, as described for example in DE 102009040910. However, even for such antennas, it is desirable to have the cross-polarization spacing over the full bandwidth of the frequency bands described above L1 . L2 or L5 to improve.

The object of the present invention is therefore to provide an antenna for the reception of circularly polarized satellite radio signals for satellite navigation, which has a sufficient gain with a small volume and a high gain over a large frequency range Having cross polarization distance and thus is suitable for obtaining particularly accurate detection results in a vehicle.

• Antenne (1) für den Empfang zirkular polarisierter Satellitenfunksignale, umfassend wenigstens eine horizontal orientierte, über einer leitenden Grundfläche (6) angeordnete Leiterschleife, mit einer mit einer Antennenanschlussstelle (5) verbundenen Anordnung zur elektromagnetischen Erregung (3) der Leiterschleife, umfassend die folgenden Merkmale:
  • - die Leiterschleife ist als Ringleitungsstrahler (2) durch eine polygonale oder kreisförmige geschlossene Ringleitung (14) in einer horizontalen Ebene mit der Höhe h < 0,15 der Freiraumwellenlänge λ über der leitenden Grundfläche (6) verlaufend gestaltet,
  • - der Ringleitungsstrahler (2) bildet eine Resonanzstruktur und ist durch elektromagnetische Erregung in der Weise erregbar, dass sich auf der Ringleitung (14) die Stromverteilung einer laufenden Leitungswelle in einer einzigen Umlaufrichtung einstellt, deren Phasenunterschied über einen Umlauf gerade 2π beträgt
  • - es sind am Umfang des Ringleitungsstrahlers (2) verteilt, an Ringleitungs-Koppelpunkten (7) mit dem Ringleitungsstrahler (2) galvanisch verkoppelte, vertikale und zur leitenden Grundfläche (6) hin verlaufende vertikale Strahler (4) vorhanden
  • - es ist ein Verteil-und Phasenschiebernetzwerk (8) mit einem eingangsseitig mit der Antennenanschlussstelle (5) verkoppeltem Tor T1 und mit zwei Toren T2, T3 vorhanden, deren Transmissionskoeffizienten S₁₂ und S₁₃ sich voneinander um einen Phasenwinkel ΔΦ unterscheiden gekennzeichnet durch die folgenden Merkmale
    • - der Ringleitungsstrahler (2) ist als kurze Ringleitung (14) ausgeführt, dessen Umfang (L) kürzer ist als die Freiraum-Wellenlänge λ.
    • - es sind mindestens drei vertikale Strahler (4) als vertikale Resonanzstrahler (4, 4a, 4b, 4c,...) vorhanden, welche über Blindwiderstandsschaltungen (13) mit kapazitiver Reaktanz X mit der Grundfläche (6) verkoppelt sind, durch welche die Resonanz der kurzen Ringleitung (14) hergestellt ist.
    • - der Eingangswiderstand jedes der Tore T2, T3 des Verteil-und Phasenschiebernetzwerks (8) zur elektromagnetischen Erregung der Leitungswelle auf der Ringleitung (14) ist jeweils ein im Wesentlichen reeller Widerstand.
    • - dem Verteil-und Phasenschiebernetzwerk (8) sind zwei der Ringleitungskoppelpunkte (7a, 7b) in der Weise zugeordnet, dass jedes der Tore T2, T3 des Verteil-und Phasenschiebernetzwerks (8) jeweils mit dem ihm zugeordneten Ringleitungskoppelpunkt (7a, 7b) verkoppelt ist.

This object is achieved in an antenna according to the preamble of the main claim by the characterizing features of the main claim and the measures proposed in the further claims:

• Antenna ( 1 ) for receiving circularly polarized satellite radio signals, comprising at least one horizontally oriented, over a conductive base ( 6 ) arranged conductor loop, with one with an antenna connection point ( 5 ) associated arrangement for electromagnetic excitation ( 3 ) of the conductor loop, comprising the following features:
  • - the conductor loop is used as a ring line radiator ( 2 ) by a polygonal or circular closed loop ( 14 ) in a horizontal plane with the height h <0.15 of the free space wavelength λ over the conductive base ( 6 ) running,
  • - the ring tube emitter ( 2 ) forms a resonant structure and is excitable by electromagnetic excitation in such a way that on the loop ( 14 ) adjusts the current distribution of a current line wave in a single direction of rotation, whose phase difference over a cycle is just 2π
  • - it is on the circumference of the ring line radiator ( 2 ), at ring line crosspoints ( 7 ) with the ring line emitter ( 2 ) galvanically coupled, vertical and conductive base ( 6 ) extending vertical radiator ( 4 ) available
  • it is a distribution and phase shifting network ( 8th ) with an input side with the antenna connection point ( 5 ) coupled gate T1 and with two goals T2 . T3 present whose transmission coefficients S ₁₂ and S ₁₃ from each other by a phase angle ΔΦ distinguished by the following features
    • - the ring tube emitter ( 2 ) is a short loop ( 14 ) whose scope ( L ) is shorter than the free space wavelength λ.
    • - there are at least three vertical emitters ( 4 ) as vertical resonance radiators ( 4 . 4a . 4b . 4c , ...), which are connected via reactance circuits ( 13 ) with capacitive reactance X with the base area ( 6 ) are coupled by which the resonance of the short loop ( 14 ) is made.
    • - the input resistance of each of the gates T2 . T3 of the distribution and phase shifting network ( 8th ) for the electromagnetic excitation of the line shaft on the loop ( 14 ) is in each case a substantially real resistance.
    • - the distribution and phase-shifting network ( 8th ) are two of the loop coupling points ( 7a . 7b) assigned in the way that each of the gates T2 . T3 of the distribution and phase shifting network ( 8th ) in each case with its associated ring line cross point ( 7a . 7b) is coupled.

The particular advantage of the invention is given by the fact that the strict requirement of the frequency bandwidth of the Kreuzpolarisationsabstands in conjunction with a low overall volume of the antenna 1 , as existing for current and in particular future automobiles, can be met. With an antenna 1 According to the invention, the advantage is further associated that it is particularly inexpensive to produce and thus is particularly suitable for mass production and use in the mass production of vehicles.

The inventive idea consists of, among others, the antenna 1 To realize according to the preamble of claim 1 as a shortened loop and still comply with the strict requirements regarding the Kreuzpolarisationsabstands in a wide frequency range. This is achieved by the features of the present invention in particular in that the resonance of the loop at the frequency f0 through the vertical resonance radiators via their reactance circuits 13 with capacitive reactance X is made without involvement of the arrangement for exciting the antenna.

The ring line emitter 2 is then at the resonance frequency f0 in self-resonance when the arousal 3 it is not drawn into the resonance formation. The antenna impedance present at this frequency 37 is a high impedance real resonance resistor 42 and is often between 300 ohms to 500 ohms. This is - due to the demand according to the invention of a real input resistance 43 of the distribution and phase shifter network 8th at the gates T2 and T3 - The natural resonance of the ring line radiator 2 through the excitement 3 unaffected. This ensures that no resonant currents of the ring line radiator 2 over the gates T2 and T3 of the distribution and phase shifter network 8th flow. The separate adjustment of the resonance of the loop emitter and the distribution and phase shifter network 8th with respect to the real input resistance 43 the gates T2 and T3 to the resonant frequency f0 leads to the inventively large frequency bandwidth of the suppression of the cross polarization of the antenna 1 ,

• - diesen Phasenunterschied einer Stromwelle auf der Ringleitung 14 des Ringleitungsstrahlers 2 repräsentierenden - vertikalen Resonanz-Erregungsstrahler 10a, 10b in der Weise erregen, dass über diese Tore bei Resonanz kein Blindleistungsaustausch erfolgt. Dies ist durch die Forderung nach dem reellen Eingangswiderstand der Tore T2, T3 des Verteil-und Phasenschiebernetzwerks 8 erreicht. Hierbei zeigt sich, dass die erreichbare Frequenzbandbreite des Kreuzpolarisationsabstands besonders groß ist, wenn dieser reelle Eingangswiderstand der Tore, etwa dem großen Resonanzwiderstand des Ringleitungsstrahlers bei der Frequenz f0 von bis zu 500 Ohm entspricht.

According to the invention, the large frequency bandwidth of the cross-polarization spacing in the vicinity of the resonant frequency of the loop emitter can be achieved when the capacitive and the inductive reactive power of the loop emitter without a contribution of the means for excitation 3 of the ring line radiator are balanced. The particular frequency bandwidth of the cross-polarization distance around the resonant frequency f0 is inventively achieved in that the two goals T2 . T3 a distribution and phase shifting network 8th with corresponding phase difference ΔΦ the two

• - This phase difference of a current wave on the loop 14 of the ring line radiator 2 representative - vertical resonance excitation emitter 10a . 10b in such a way that no reactive power exchange takes place over these gates at resonance. This is due to the demand for the real input resistance of the gates T2 . T3 of the distribution and phase shifter network 8th reached. This shows that the achievable frequency bandwidth of the cross-polarization distance is particularly large if this real input resistance of the gates, such as the large resonance resistance of the loop emitter at the frequency f0 of up to 500 ohms.

The invention will be explained in more detail below on the basis of exemplary embodiments:

The distribution and phase shifting network ( 8th ), two of the vertical resonance radiators ( 4a . 4b) as vertical resonance excitation emitters ( 10a . 10b) be assigned in the way that each of the gates T2 . T3 of the distribution and phase shifting network ( 8th ) via one of the two vertical resonance excitation emitters ( 10a . 10b) with its loop connection point ( 7a . 7b) is coupled and the other vertical resonant radiators ( 4c . 4d ..) as a passive vertical resonance radiator ( 9a . 9b , etc) are effective.

Each of the gates T2 . T3 can each directly via one of the resonance excitation emitters 10a . 10b with its ring line cross point 7 be connected and the input resistance of the gates T2 . T3 be in the way high impedance real, that between this and the high impedance resonant resistance of the loop emitter 2 Resistance adjustment exists.

By the effect of the reactance circuit 13 with capacitive reactance X loaded vertical resonance radiator 4a . 4b . 4c can the stretched length L the ring line 14 of the resonant ring tube radiator 2 be shortened from about the free space wavelength λ to about one third of the free space wavelength λ.

The reactance circuit 13 with which each of the passive vertical resonant radiators 9 . 9a . 9b , and also the resonance excitation emitter 10a . 10b with the base area 6 is coupled, each may be formed by a capacity.

The reactance circuit 13 with which the vertical resonance excitation emitters 10a . 10b with the base area 6 can each be connected as a series circuit of a first reactance circuit 20a . 20b and a second reactance resistor circuit 21a . 21b with a linking point connecting them 19 be formed, with each one of the gates T2 . T3 connected is.

The reactance circuit 13 with which the vertical resonance excitation emitters 10a . 10b with the base area 6 are coupled and the first reactance circuit 20a . 20b each vertical resonance excitation emitter 10a . 10b can through a first capacity 22a . 22b and the second reactance circuit 21a . 21b through a second capacity 23a . 23b may be formed and the size and the ratio of the first and the second capacitance may each be selected in such a way that between the high impedance resonance resistance of the loop emitter 2 and a real input resistance of the gates T2 . T3 is given at a lower impedance level each resistance adjustment.

The reactance circuits 13c , etc., with which the passive vertical resonance emitters 9c , etc. with the electrically conductive base 6 can be coupled, each from the series connection of a capacity 28 and a parallel oscillation circuit - consisting of a parallel capacitance 45 and a parallel inductance 46 - Be formed and the first reactance circuit 20a . 20b the vertical resonance excitation emitter 10a . 10b can each be considered a first capacity 22a . 22b be formed and their second reactance circuit 21a . 21b can each be used as a parallel oscillation circuit - consisting of a second capacity 23a . 23b and a parallel inductance 46a . 46b be designed and the resonant frequency of all Parallel vibration circuits can each be in the frequency range between the upper frequency band L1 and the lower frequency band L2 be chosen in such a way that the reactance of these resonant circuits in the frequency band L1 capacitive and in the frequency band L2 is inductive.

It may be that, however, between the capacity 28 and the parallel oscillation circuit from the parallel capacitance 45 and the parallel inductance 46 the reactance circuits 13c , etc. of the passive vertical resonant radiator 9c , etc. in each case a further parallel resonant circuit 44c , etc. is connected in series and the first capacity 22a . 22b in the first reactance circuit 20a . 20b the vertical resonance excitation emitter 10a . 10b towards the point of connection 19a . 19b likewise in each case a further parallel resonant circuit 44a . 44b is connected downstream and the parallel resonant circuits in the first reactance circuit 20a . 20b and the second reactance circuit 21a . 21b the vertical resonance excitation emitter 10a . 10b are tuned in the way that their reactance Xa , Xb in both frequency bands L1 . L2 is capacitive and their relationship to each other according to the optimal measure topt exists for adaptation and the reactance circuits 13c , etc. of the passive vertical resonant radiator 9c , etc., and the vertical resonance excitation radiator 10a . 10b are matched with respect to their frequency behavior.

The first capacity 22a . 22b the first reactance circuit 20a . 20b the vertical resonance excitation emitter 10a . 10b as well as the capacities 28 in the reactance circuits 13c , etc. of the passive vertical resonant radiator 9c , etc can be formed in such a way that all vertical resonance emitters 4a . 4b , etc. at their lower end to individually designed capacitive capacitive electrodes 32a . 32b . 32c . 32d are formed that capacity 22a . 22b 28 by interposing a dielectric plate 33 between the two-dimensional capacitance electrodes 32a . 32b . 32c . 32d and the electrically conductive coated circuit board 35 executed electrically conductive base 6 for coupling the passive vertical resonance radiators 9c . 9d , etc to the electrically conductive base 6 be designed, and capacitive coupling of the vertical resonance excitation emitters 10a . 10b on the electrically conductive base 6 may each one of the conductive layer of the coated circuit board 35 isolated, flat counter electrode 34 for connection of the second reactance resistor circuit 21b be designed.

The vertical radiators can for capacitive coupling at their lower end to individually designed flat capacitance electrodes 32a . 32b . 32c . 32d be formed, but it can be used to couple the passive vertical resonance radiator 9 . 9a . 9b one each from the conductive layer of the coated circuit board 35 isolated, flat counter electrode 34 for connecting the reactance circuit formed of concentrated reactive elements 13c , 13d be formed.

It may be the conductive structure consisting of the loop 14 and the associated vertical resonant radiators 4 . 4a-d through a dielectric support structure 36 be so fixed that the dielectric plate 33 realized in the form of an air gap.

There can be a total of three vertical resonance emitters 4a . 4b , etc. with azimuthally equally distributed loop crosspoints 7 be present, two of which are called vertical resonance excitation emitters 10a . 10b over the gates T2 . T3 a distribution and phase shifting network 8th are excited whose phase angle ΔΦ = 120 degrees.

It can be a variety of N> 4 vertical resonant radiators 4 . 4a . 4b . 4c , ... with the stretched length L the ring line 14 equally distributed crosspoints 7 be present, so that N loop sections 30a . 30b . 30c . 30d are formed and between the crosspoints 7 the two vertical resonance excitation emitters 10a . 10b n Ring line sections are selected and the phase angle ΔΦ = (n + 1) * 2π / N <180 ° is selected.

There can be four vertical resonance emitters 4a . 4b . 4c . 4d with azimuthal equal length loop sections 30a . 30b of which two are azimuthal consecutive as vertical resonance excitation emitters 10a . 10b .. over the gates T2 . T3 a distribution and phase shifting network 8th are excited whose phase angle ΔΦ = 90 degrees.

However, it can be the ring line rectangular with four loop crosspoints 7 . 7a .. 7d at the corners and with different lengths of loop sections 30a . 30b and with different ring conductor widths 15a . 15b be designed, wherein by appropriate choice of the ring conductor widths 15a . 15b for the loop sections 30a . 30b each the same characteristic impedance ZL and the same effective electrical length are set.

It can be a distribution and phase shifting network 8th for a phase shift by the phase angle ΔΦ = 90 ° a hybrid ring 38 included, which next to the two gates T2 and T3 a fourth, with an ohmic terminator 40 completed gate T4 owns and which at wave resistance correct completion of the gates T2 and T3 from the gate T1 is decoupled.

It can be the distribution and phase shifting network 8th for a phase shift by the phase angle ΔΦ = 90 ° a Wilkinson divider 39 with an ohmic balancing resistor 41 included, with the entrance at the gate 1 and one of the two output branches a phase shifter 17 with the phase angle ΔΦ = 90 ° towards the goal T3 is connected downstream and at the other of the two output branches the gate T2 is formed.

It can be a first and a second distribution and phase shifter network 8th are provided with a phase angle ΔΦ = 90 ° and four designed as excitation emitters azimuthal vertical radiator evenly distributed on the circumference of the ring conductor, wherein the first distribution and phase shifter network 8th with his gates T2 . T3 with a first pair of vertical resonance excitation emitters 10a . 10b , is coupled and the second distribution and phase shifter network 8th with his gates T2a . T3a with a second pair of vertical resonance excitation emitters opposing the first pair 10c . 10d is coupled and another distribution and phase shifter network 8th with ΔΦ = 180 °, which is used to excite the circulating line shaft with its gates T 2c . t3c with the entrance gates T1 . T1 is connected in such a way that adjusts the rotating shaft on the ring line.

It can be the lossless matching circuit 18 for transforming the low impedance level Z0 in the high impedance real input resistance in both output branches of the distribution and phase shifter network 8th each essentially by a lambda / 4 transformation line 12a . 28b and a series circuit of an inductance 27a , 27b and a capacity 28a , 28b for fine adjustment of the high impedance input resistance of the gates T2 and T3 contain.

All vertical resonance radiators ( 4a . 4b . 4c , etc.) along the loop ( 14 ), so that none of the distances between adjacent ring line crosspoints ( 7a . 7b . 7c) at the circumference of the ring line radiator ( 2 ) is less than half of the distance with equidistant distribution over the extended length of the loop ( 14 ).

• 1: Antenne 1 nach der Erfindung, bestehend im Beispiel aus dem Ringleitungsstrahler 2 mit dem Ringleiter 14 in der Höhe h < 0,15λ über der leitenden Grundfläche 6 mit N = 3 vertikalen Resonanzstrahlern 4a, 4b, 4c an azimutal gleich verteilten Ringleitungs-Koppelpunkten 7a, 7b, 7c und der elektromagnetischen Erregung 3 durch das Verteil-und Phasenschiebernetzwerk 8. Die vertikalen Resonanzstrahler 4a, 4b, 4c sind jeweils über eine kapazitiv wirkende Blindwiderstandsschaltung 13a,13b,13c über den Masse-Anschlusspunkt 11 mit der leitenden Grundfläche 6 verkoppelt. Die elektromagnetische Erregung 3 erfolgt über das Verteil-und Phasenschiebernetzwerk 8, welches eingangsseitig mit seinem Tor T1 mit dem Antennenanschluss 5 verbunden ist und dessen Tore T2 und T3 mit zwei von den vertikalen Resonanzstrahlern 4a, 4b, 4c als vertikale Resonanz-Erregungsstrahler 10a und 10b in der Weise verbunden sind, dass die zugehörigen Ringleitungs-Koppelpunkte 7a, 7b direkt mit den im Verteilnetzwerk 16 und im Phasendrehglied 17 bewirkten Ausgangssignalen an den Toren T2 und T3 mit Phasenwinkelunterschied ΔΦ = 360°/N = 120° angesteuert sind, sodass sich auf dem Ringleiter 14 die laufende Leitungswelle einstellt. Hierbei ist erfindungsgemäß vorausgesetzt, dass die Tore T2 und T3 jeweils einen reellen Eingangswiderstand 43 besitzen, sodass die Resonanz des Ringleitungsstrahlers 2 durch Anschluss der elektromagnetischen Erregung 3 nicht beeinflusst ist. Zur Impedanzanpassung zwischen dem Verteil-und Phasenschiebenetzwerk 8 an den Toren T2, T3 und der hochohmigen Antennenimpedanz 42 bei der Resonanzfrequenz f0 des Ringleitungsstrahlers 2 (sh. 2b) an den Ringleitung-Koppelpunkten 7a und 7b ist das Verteil-und Phasenschiebenetzwerk 8 hochohmig gestaltet und befindet sich auf einem Impedanzniveau von etwa 300 - 500 Ohm.
• 2: Figur a) zeigt den Ringleitungsstrahler 2 nach der Erfindung mit drei azimutal gleich verteilten vertikalen Resonanzstrahlern 4a - 4c als Resonanzstruktur. Die Erregung 3 ist nicht dargestellt. Die Blindwiderstandsschaltungen 13 a- 13c sind durch die Kapazitäten 28a, 28b, 28c realisiert. Bei Rotationssymmetrie der Anordnung sind alle Kapazitäten gleich groß. Bei Abstimmung des Ringleitungsstrahlers 2 auf eine Resonanzfrequenz- zum Beispiel f0= 1392 MHz - stellt sich an den vertikalen Resonanzstrahlern 4a- 4c jeweils an der mit 42 bezeichneten Messstrecke der in Figur b) dargestellte Verlauf der Antennenimpedanz mit ihrem durch die Strahlung bedingten reellen Resonanzwiderstand von ca. 340 Ohm ein. Bei Anschluss der Erregung an die Resonanzstrahler 4a und 4b - wie in 1 - wird das Resonanzverhalten der Struktur in der Frequenz nicht verändert. Im Beispiel der 3 azimutal gleich verteilten vertikalen Resonanzstrahler 4a, 4b, 4c beträgt der Phasendrehwinkel des Phasendrehglieds ΔΦ= 120°.
• 3: Antenne 1 nach der Erfindung wie in 1 jedoch mit einem Verteilnetzwerk 16 und einem Phasendrehglied 17 auf dem Impedanzniveau Z0 üblicher koaxialer Leitungen (Z0 = 50 Ohm). Zur Anpassung an das hochohmige Impedanzniveau der Antennenimpedanz 42 (sh. 2b) sind dem Phasendrehglied 17 und einem Arm des Verteilnetzwerks 16 jeweils ein Anpassnetzwerk 18 nachgeschaltet. Der Eingangswiderstand 43 an den Toren T2 und T3 ist - ebenso wie in 1 - bei der Resonanzfrequenz des Ringleitungsstrahlers 2 hochohmig und reell.
• 3a: Verteil-und Phasenschiebernetzwerk 8 wie in 3 mit beispielhafter Ausführung der Anpassnetzwerke 18 an den beiden Ausgängen. Die Anpassung bei Resonanz des Ringleitungsstrahlers 2 ist durch die λ/4-Transformationsleitungen 12a, 12b, durch welche die Transformation von dem niederohmigen Impedanzniveau Z0 zu dem hochohmigen Impedanzniveau der Antennenimpedanz 42 gegeben ist, bewirkt. Die Serienresonanzkreise, bestehend aus den Kapazitäten 28a, 28b und den Induktivitäten 12a und 12b ermöglichen eine Feinkorrektur der Anpassung über einen erweitertenFrequenzbereich.
• 4: Antenne nach der Erfindung wie in 3 jedoch mit besonderer Gestaltung der Blindwiderstandsschaltungen 13a und 13b in den beiden vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils unterteilt in eine erste Blindwiderstandsschaltung 20a, 20b und eine zweite Blindwiderstandsschaltung 21a, 21b mit einem dazwischen liegenden Verknüpfungspunkt 19a, 19b zum Anschluss der Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8.
• 5: Erfindungsgemäße Ausführungsformen der insgesamt kapazitiv wirkenden Blindwiderstandsschaltungen 13 zur Ankopplung der der vertikalen Resonanzstrahler an die leitende Grundfläche 6 in 4
  1. a) Blindwiderstandsschaltung 13a bzw. 13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b, unterteilt in die erste Blindwiderstandsschaltung 20a bzw. 20b und die über den Verknüpfungspunkt 19a bzw. 19b verbundene zweite Blindwiderstandsschaltung 21a bzw. 21b, jeweils realisiert durch eine erste Kapazität 22a bzw. 22b und entsprechend durch die zweite Kapazität 23a bzw.23b.
  2. b) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert durch eine Kapazität 28. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur a) beschaltet.
  3. c) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur a) jedoch mit einem Parallelresonanzkreis, bestehend aus der Parallelinduktivität 46a bzw. 46b und der Parallelkapazität 23a bzw.23b in der zweiten Blindwiderstandsschaltung 21a bzw. 21b zur Erweiterung der Frequenzbandbreite der Antenne.
  4. d) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert, wie unter Figur b) jedoch mit einem seriellen Parallelresonanzkreis 25c,25d bestehend aus der Parallelinduktivität 46c, 46d und der Parallelkapazität 45c, 45d in Serienschaltung zur Kapazität 28 zur Erweiterung der Frequenzbandbreite der Antenne in Entsprechung der Gestaltung der Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 4a bzw. 4b wie unter c). Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur c) beschaltet.
  5. e) Darstellung der Frequenzbänder L1 und L2 für die Satelliten-Navigation mit den Mittenfrequenzen fm1 und fm2 und den unteren und oberen Grenzfrequenzen fu1, fo1 bzw.fu2, fo2. Die Frequenz fm beschreibt die Mittenfrequenz zwischen fu1 und fo2.
  6. f) Blindwiderstandsschaltung 13a bzw. 13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur c) jedoch mit einem weiteren Parallelresonanzkreis 44a bzw 44b in Serienschaltung zur ersten Kapazität 22a bzw. 22b hin zum Verknüpfungspunkt 19 zur gesonderten Optimierung der Schaltung für jeweils einen der beiden Frequenzbereiche L1 und L2 zur Gestaltung einer Zweibandantenne.
  7. g) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert wie unter Figur d), jedoch mit einem weiteren Parallelresonanzkreis 44c in Serienschaltung zur Kapazität 28 in der Blindwiderstandsschaltung 13c. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur f) beschaltet zur Gestaltung einer Zweibandantenne.
  8. h) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur f) jedoch mit einer Zusatz-Parallelkapazität 47, gebildet durch die, die isolierte Gegenelektrode 34 überkragende Fläche der Kapazitätselektrode 32a, 32b in den 13c und 13d mit der elektrisch leitenden beschichteten Leiterplatte 35. Die Zusatz-Parallelkapazität 47 ermöglicht die Erweiterung des Bereichs der Impedanzanpassung jeweils am Verknüpfungspunkt 19a, 19b bei ebenso gegebener Einhaltung der Resonanzbedingung für den Ringleitungsstrahler 2. Die Resonanzbedingung und die Impedanzanpassung werden durch Abstimmung der Größen der Kapazitätselektroden 32a, 32b und der isolierten Gegenelektroden 34 aufeinander hergestellt.
• 6: Anpassungsverhältnisse bei der Resonanzfrequenz f0 in Abhängigkeit vom Maß „t“ für die Unterteilung in die erste Kapazität 22a, 22 b und die zweite Kapazität 23a, 23c am Beispiel des Ringleitungsstrahlers 2 in 4.
  1. a) Reflexionsfaktor an der Antennenanschlusstelle 5 als Eingangstor T1 eines verlustfreien Verteil-und Phasenschiebernetzwerks 8 der Antenne 1 in Abhängigkeit vom Teilungsmaß t. Anpassung ist mit topt = 0,13 erreicht.
  2. b) Transformationsfaktor bei der Resonanzfrequenz f0 zwischen dem Eingangswiderstand 43 mit Z0 = 50 Ohm des Phasendrehglieds 17 und dem Resonanzwiderstand 42 bei f0 am Ringleitungsstrahler-Koppelpunkt 7 in Abhängigkeit vom Teilungsmaß t.
  3. c) Relative Leistung P/Pmax bei f0 in Abhängigkeit vom Teilungsmaß t.
• 7:
  • Antenne nach der Erfindung mit Widerstands-Anpassung durch kapazitive Unterteilung wie im Beispiel der 4 jedoch mit insgesamt 6 azimutal um jeweils 60° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a .. 4f, von denen 4 als vertikale passive Resonanzstrahler 9a..9d gestaltet sind.
  • Die Erregung 3 erfolgt beispielhaft an den beiden um 120° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 120° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• 8: Antenne nach der Erfindung wie in 7, jedoch mit insgesamt 8 azimutal um jeweils 45° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a..4h. Die Erregung 3 erfolgt beispielhaft an den beiden um 90° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 90° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• 9: Antenne nach der Erfindung mit rechteckig ausgeführter Ringleitung 14 mit vier vertikalen Resonanzstrahler 4a bis 4d im Bereich der Ringleitungs-Ecken. Zwei der vertikalen Resonanzstrahler sind als vertikale Resonanz-Erregungsstrahler 10a, 10b und die beiden übrigen Strahler als passive vertikale Resonanzstrahler 9a, 9b ausgeführt. Impedanzanpassung an das niederohmig (Z= 50 Ohm) ausgeführte Verteil-und Phasenschiebernetzwerk 8 ist durch kapazitive Unterteilung, ähnlich wie in den 4, 7 und 8 erreicht. Die Ringleitung 14 ist vorzugsweise quadratisch gestaltet mit gleichen Ringleiterbreiten 15a, 15b und gleichen Ringleitungs-Abschnitten 30a, 30b. Sowohl die Ringleiterbreite 15a, 15b als auch die Ringleitungs-Abschnitte 30a, 30b können jedoch abgestimmt aufeinander in Grenzen unterschiedlich gewählt sein.
• 10: Rechteckförmiger Ringleitungsstrahler 2 einer Antenne 1 nach der Erfindung ohne Darstellung der elektromagnetischen Erregung 3 zur Erläuterung der Resonanzstruktur des Ringleitungsstrahlers 2.
  1. a) geometrischer Aufbau des Ringleitungsstrahlers 2 mit großen unterschiedlich wählbaren Ringleiterbreiten 15a und 15b sowie den unterschiedlich wählbaren Ringleitungs-Abschnitten 30a und 30b. Die vertikalen Resonanzstrahler 4a - 4d sind mit ihren Ringleitungs-Koppelpunkten 7a - 7d im weiten Bereich der Ecken des rechteckförmigen Ringleiters 14 vorgesehen. Die unterbrochene Linie kennzeichnet etwa den Verlauf der Schwerlinie 24 der Stromdichteverteilung der Leitungswelle bei Erregung des Ringleitungsstrahlers 2. Aufgrund der Stromverdrängung verdichtet sich die Stromverteilung hin zum Rand des Ringleiters 14. Der für die Funktion bei der Erregung des Ringleitungsstrahlers 2 relevante Strom der Leitungswelle fließt demnach - gekennzeichnet durch die als unterbrochene Linie gezeichnete Schwerlinie der Stromverteilung 24 - auch bei großen Leiterbreiten 15a, 15b mehr zum Rand hin gedrängt. Dies gilt insbesondere für sehr große Ringleiterbreiten 15a, 15b bis hin zur vollkommenen Schließung der inneren Öffnung zum Zentrum hin, welches praktisch stromlos ist, wenn die Ringleitung 14 als eine geschlossene leitende Fläche realisiert ist.
  2. b) gemäß dem durch die Schwerlinie der Stromdichteverteilung 24 repräsentierten Strom kann die Resonanzstruktur des Ringleitungsstrahlers 2 durch ein grob angenähertes Ersatzschaltbild allgemein wiedergegeben werden. Die einzelnen Ringleitungs-Abschnitte 30a - 30d sind jeweils durch die induktive und kapazitive Wirkung des zugehörigen Abschnitts der Ringleitung 14 unter Einbeziehung der kapazitiven Wirkung der Blindwiderstandsschaltung 13 als konzentrierte induktive Elemente (Ln) und kapazitive Elemente (Cn) dargestellt. Jeder n-te Ringleitungsabschnitt 30a - 30b ist durch eine π-Struktur gemäß Figur c), bestehend aus einer Längsinduktivität 2*Ln und jeweils einer Querkapazität Cn an deren beiden Enden, dargestellt. Die Strahlungsdämpfung jedes horizontal orientierten Ringleitungsabschnitt 30a - 30b ist durch den Dämpfungsfaktor d der konzentrierten Induktivität einbezogen. Die Aneinanderreihung einander benachbarter Ringleitungs-Abschnitte 30a-30d erfolgt jeweils über einen gemeinsamen vertikalen Resonanzstrahler 4a..4d unter Zusammenziehung der Querkapazitäten Cn der benachbarten Ringleitungs-Abschnitte, wie in Figur c) dargestellt. Die geringfügige induktive Wirkung der vertikalen Resonanzstrahler ist bei dieser Grundsatzbetrachtung aufgrund der geringen Antennenhöhe h vernachlässigt. Erfindungsgemäß vorteilhafte Voraussetzungen für den Ringleitungsstrahler 2 sind für die Resonanzfrequenz f0=ωo/2π dann erreicht, wenn jeweils alle Wellenwiderstände $ZLn=\sqrt{\frac{Ln}{Cn}}$
     
     und jeweils alle $\sqrt{Ln·Cn}=\raisebox{1ex}{$1$}\!\left/ \!\raisebox{-1ex}{$\omega on$}\right.,$
     
     welche die elektrische Länge des Ringleitungs-Abschnitts repräsentieren, für alle n gleich groß sind.
  Bei rotationssymmetrischen Ringleitungsstrahlern 2 ist dies stets gegeben. Andernfalls kann diese Voraussetzung zum Beispiel bei einer rechteckförmigen Struktur des Ringleitungsstrahlers 2 durch individuelle Gestaltung der Ringleiterbreiten 15a-15d in den einzelnen Ringleitungs-Abschnitten 30a-30b erreicht werden.
• 11:
  1. a) Gestaltung des Verteil-und Phasenschiebernetzwerk 8 - zum Beispiel wie in den 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Hybridring 3, durch welchen sowohl die Leistungsteilung als auch die Phasenverschiebung erfolgt. Bei Einspeisung am Tor T1 wird der Ringleitungsstrahler 2 über die Tore T2 und T3 mit dem Phasenunterschied von 90° erregt. Der Abschluss von Tor T4 mit dem ohmschen Abschlusswiderstand 40 bewirkt bei Abweichung der Frequenz von der Resonanzfrequenz f0 breitbandig die teilweise Absorption der Leistung der unerwünschten Polarisation bei Einspeisung am Tor T1.
  2. b) Idealisierte Streumatrix zur Beschreibung des allgemein bekannten grundsätzlichen Wellenverhaltens eines Hybridrings 38 in Figur a) an den Toren T1 bis T4. Die wichtige Entkopplung der Tore T1 und T4 voneinander ist durch die Streuparameter S14 = 0, S41= 0 in der Matrix fett gekennzeichnet.
  3. c) Gestaltung des Verteil-und Phasenschiebernetzwerk 8 - zum Beispiel wie in den 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Wilkinson-Teilers 16 mit nachgeschaltetem λ/4 Leitungs-Phasenschieber 17 an einem Tor T3* zur Erzeugung des Phasenunterschieds von 90°. Der ohmsche Symmetrierungswiderstand 40 absorbiert bei unsymmetrischer Belastung des Wilkinson-Teilers 16 bei Abweichung von der Resonanzfrequenz f0 teilweise die Ströme, welche die unerwünschte Polarisation auf dem Ringleitungsstrahler 2 hervorrufen.
• 12: Antenne nach der Erfindung z. B. wie in 9 mit rechteckig geformter Ringleitung 14. Die Kapazitäten 22a, 22b und 28 sind in der Weise gebildet, dass die vertikalen Resonanzstrahler 4a - 4d an ihrem unteren Ende zu individuell gestalteten flächigen Kapazitätselektroden 32a, 32b, 32c, 32d ausgeformt sind. Durch Zwischenlage zwischen diesen und die als elektrisch leitend beschichtete Leiterplatte ausgeführte elektrisch leitende Grundfläche 6 befindliche dielektrische Platte 33 sind die Kapazitäten 28 zur Ankopplung der passiven vertikalen Resonanzstrahler 4c, 4d an die elektrisch leitende Grundfläche 6 gestaltet. Zur kapazitiven Ankopplung der aktiven vertikalen Resonanzstrahler 10a, 10b an die Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8 ist dieser Anschluss als jeweils eine von der leitenden Schicht isolierte, flächige Gegenelektrode 34 gestaltet. Die Gegenelektroden 34 können somit als Verknüpfungspunkte 19 der Antenne in 9 ausgeführt sein und können als Anschlusspunkte für die Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8, wie in 13 angedeutet, dienen.
• 13: Antenne 1 ähnlich wie in 11, jedoch ist die dielektrische Wirkung der dielektrischen Platte 33 durch einen Luftspalt realisiert. Typische Abmessungen eines Ringleitungsstrahlers 2 für den Frequenzbereich L1 sind für quadratische Antennen ein Abmessung von 30mm bis 40mm und für die Höhe h = 8mm. Zur Widerstandsanpassung des vorzugsweise auf niedrigem Impedanzniveau (Z0) ausgeführten Verteil-und Phasenschiebernetzwerks 8 ist die kapazitive Unterteilung wie in 9 vorgesehen.
  1. a) Die beispielhaft zugehörigen Blindwiderstands-schaltungen 13a, 13b und 13c sind in den 5a und 5b dargestellt. Die Kapazität 28 der Kapazitätselektrode 32b, 32c gegen die elektrisch leitende Grundfläche 6 bzw. die Kapazität 22a, 22b gegen die isolierte Gegenelektrode 34a, 34b beträgt jeweils ca. 0.3pF. Die Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8 mit ihrem reellen Eingangswiderstand 43 sind jeweils an eine Gegenelektrode 34a, 34b als Verknüpfungspunkt 19a, 19b angeschlossen. Die Kapazitive Unterteilung zur Impedanzanpassung ist jeweils durch die Kapazität 22a, 22b und der Kapazität 23a, 23b gegeben. Diese zwischen dem Verknüpfungspunkt 19a, 19b und dem Massepunkt 11 eingebrachte Kapazität 23a, 23b ist auf der Rückseite der beschichteten Leiterplatte 35 als SMD-Bauteil angebracht. Hierzu sind auf der Rückseite der Leiterplatte isolierte Pads 29 als Kontaktstützpunkte gestaltet, welche jeweils über eine Durchkontaktierung 26 mit der Gegenelektrode 34 einerseits und andererseits z.B. über die Kapazität 23a, 23b (sh. 4 u. 5a) mit Masse verbunden sind.
  2. b) die Blindwiderstandsschaltungen 13a, 13b der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur a) gemäß 5a bzw. wie in 5c gestaltet. Bei Gestaltung nach 5c ist der auf der Rückseite der beschichteten Leiterplatte 35 eingebrachten Kapazität 23a, 23b eine Induktivität 46a, 46b als SMD-Bauteil parallel geschaltet. Im Gegensatz zur Figur a) sind zur Bildung der Blindwiderstandsschaltungen der beiden passiven Resonanzstrahler 9a, 9b den Kapazitätselektroden 32b, 32c auf der beschichteten Leiterplatte 35 isolierte Gegenelektroden 34 gegenübergestellt. Ausgehend von der Gegenelektrode 34 als Kontaktstützpunkt sind auf der Rückseite der Leiterplatte 35 die Parallelinduktivität 46 und die Parallelkapazität 45 zum Massepunkt 11 auf der Leiterplatte 35 geschaltet, sodass die Blindwiderstandsschaltung 13 in 5d realisiert ist. Bei Ausgestaltung aller Blindwiderstandsschaltungen 13 aller vertikalen Resonanzstrahler mit jeweils einer Kapazitätselektrode 32 und einer gegenüberliegenden isolierten Gegenelektrode 34 können alle Schaltungen in den 5c und 5f für die Resonanz-Erregungsstrahler 10a, 10b durch Einsatz von SMD-Bauteilen auf der Rückseite der Leiterplatte 35 realisiert werden. Auf gleiche Weise können die Blindwiderstandsschaltungen für die passiven Resonanzstrahler 9a, 9b durch Beschaltung der isolierten Gegenelektroden 34 mit SMD-Bauteilen auf der Rückseite der Leiterplatte 35 gemäß den in den 5d und 5g angegebenen Schaltungen realisiert werden.
  3. c) Wie unter 5h beschrieben dient die Zusatz-Parallelkapazität 47 zur freien Gestaltung der Impedanzanpassung bei Beibehaltung der Resonanzeigenschaften des Ringleitungsstrahlers 2. Die Figur zeigt die großflächige Überdeckung der isolierten Gegenelektrode 34a, 34b mit der Kapazitätselektrode 32a, 32b der Resonanz-Erregungsstrahler 10a, 10b. Der in der vertikalen Projektion der Kapazitätselektrode 32a, 32b gegenüber der isolierten Gegenelektrode 34 bestehende flächige Überstand bildet mit der elektrisch leitenden Schicht der beschichteten Leiterplatte 35 die Zusatz-Parallelkapazität 47. Die Blindwiderstandsschaltung der beiden passiven Resonanzstrahler 9a, 9b sind wie in Figur a) gestaltet.
  4. d) die Blindwiderstandsschaltungen 13 der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur c) mit einer Zusatz-Parallelkapazität 47 gestaltet. Die beispielhaft möglichen Blindwiderstandsschaltungen 13 der beiden passiven Resonanzstrahler 9a, 9b können wie in Figur b) beschrieben, gestaltet werden.
• 14: Im Bild ist die Oberseite der Leiterplatte 35 einer Antenne 1 nach der Erfindung dargestellt, auf welche der elektrische Ringleitungsstrahler 2 aufgesetzt wird. Für die breitbandige Gestaltung einer Antenne 1 nach der Erfindung - welche zum Beispiel beide Frequenzbereiche L1 und L2 mit einer zwischen den beiden Frequenzbereichen liegenden Mittenfrequenz fm (sh. 5e) - ist die Blindwiderstandsschaltung 13, wie in den 5c bzw. 5d gestaltet. Hierfür ist für die Blindwiderstandsschaltung in den beiden aktiven vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils die Blindwiderstandsschaltung 13a, 13b in 5c und für die beiden passiven vertikalen Resonanzstrahler 9a, 9b die Blindwiderstandsschaltung 13c in 5d vorgesehen. Dies ist bei dem Beispiel im 13a dadurch erreicht, dass für alle Kapazitätselektroden 32 jeweils eine Gegenelektrode 34 vorhanden ist und, dass der durch die Kapazitätselektroden 32 bewirkten Kapazität 22a, 22b bzw. 28 an allen vertikalen Resonanzstrahlern 4a - 4d eine Parallelschaltung aus einer Parallelkapazität 23a, 23b bzw. 45 und einer Parallelinduktivität 46a, 46b, bzw. 46 - dargestellt als SMD- Bauteile - zwischen der Gegenelektrode 34 und der elektrisch leitenden Grundfläche 6 in Serie geschaltet ist. 13a zeigt die Oberseite der Leiterplatte 35 mit Durchkontaktierungen 26 auf den Gegenelektroden 34 unter den Kapazitätselektroden 32. In 14b ist die Unterseite der Leiterplatte 35 mit den Pads 29 dargestellt, welche mit den isolierten Gegenelektroden 34 über die Durchkontaktierungen 26 verbunden sind,. Die Blindelemente sind als SMD-Bauelemente gemäß den 5c und 5d angebracht. Zusätzlich sind die beiden Verknüpfungspunkte 19 mit der jeweils daran angeschlossenen Serienschaltung aus der Kapazität 28a, 28b und der Induktivität 27a, 27b als Teile der in 3a enthaltenen Anpassungsschaltung 18 des Verteil-und Phasenschiebernetzwerks 8 zur Bildung der Tore T2 und T3 dargestellt.
• 15 Zeigt eine Antenne nach der Erfindung mit vier azimutal am Umfang der Ringleitung 14 gleich verteilten Resonanz-Erregungsstrahlern 10a bis 10d. Alle Strahler werden entsprechend einer umlaufenden Welle mit jeweils 90°-Phasenunterschied zwischen benachbarten Strahlern erregt. Hierzu sind ein erstes und ein zweites Verteil-und Phasenschiebernetzwerk (8) mit Phasenwinkel ΔΦ = 90° vorhanden, wobei das erste Verteil-und Phasenschiebernetzwerk (8) mit seinen Toren T2, T3 mit einem ersten Paar der Resonanz-Erregerstrahler 10a, 10b verkoppelt ist und das zweite Verteil-und Phasenschiebernetzwerk (8) mit seinen Toren T2a, T3a mit einem dem ersten Paar gegenüberliegendem zweiten Paar der Erregerstrahler 10c, 10d verkoppelt ist. Zur Anregung einer laufenden Welle werden einander gegenüberliegende Resonanz-Erregungsstrahler 10a, 10c bzw. 10b, 10d jeweils mit einem Phasenunterschied von ΔΦ = 180° erregt. Hierzu ist ein weiteres Verteil-und Phasenschiebernetzwerk 31, 39 mit einer Phasenverschiebung von ΔΦ = 180° vorhanden, welches mit seinem Tor T2c mit dem Tor T1 des ersten Verteil-und Phasenschiebernetzwerks 8, 36 und mit seinem Tor T3c mit dem Tor T1b des weiteren Verteil-und Phasenschiebernetzwerks 8, 38b verbunden ist. Auch hier kommt erfindungsgemäß vorzugsweise jeweils ein Hybridring 38 zum Einsatz. Das 180° Phasennetzwerk 31 kann beispielhaft als Wilkinson-Teiler 39 mit einer λ/2-langen Verzögerungsleitung 17 (sh. 11c) eingesetzt werden. Alle Anschlüsse sind in der Weise gewählt, dass sich auf der Ringleitung die leitende Welle in der gewünschten Umlaufrichtung einstellt. Mit dieser Anordnung ist der besondere Vorteil der azimutal symmetrischen Anregung verbunden, sodass der Abgleich der Anordnung besonders problemfrei erfolgen kann. Als Einschränkung ist jedoch der erhöhte Aufwand anzumerken.

The invention will be further explained below with reference to exemplary embodiments. The accompanying figures show in detail:

• 1 : Antenna 1 according to the invention, consisting in the example of the ring line emitter 2 with the ring conductor 14 in height h <0.15λ above the conductive base 6 with N = 3 vertical resonance radiators 4a . 4b . 4c at azimuthal equally distributed loop coupling points 7a . 7b . 7c and the electromagnetic excitement 3 through the distribution and phase shifting network 8th , The vertical resonance emitters 4a . 4b . 4c each have a capacitive reactance circuit 13a , 13b, 13c via the ground connection point 11 with the conductive base 6 coupled. The electromagnetic excitation 3 takes place via the distribution and phase shift network 8th , which on the input side with his gate T1 with the antenna connection 5 is connected and its gates T2 and T3 with two of the vertical resonance radiators 4a . 4b . 4c as vertical resonance excitation emitters 10a and 10b are connected in such a way that the associated loop coupling points 7a . 7b directly with the distribution network 16 and in the phase shifter 17 caused output signals at the gates T2 and T3 are controlled with phase angle difference ΔΦ = 360 ° / N = 120 °, so that on the ring conductor 14 sets the current line wave. In this case, the invention assumes that the gates T2 and T3 in each case a real input resistance 43 own, so that the resonance of the ring line radiator 2 by connection of the electromagnetic excitation 3 is not affected. For impedance matching between the distribution and phase shift network 8th at the gates T2 . T3 and the high impedance antenna impedance 42 at the resonant frequency f0 of the ring line radiator 2 (Sh. 2 B) at the ring line crosspoints 7a and 7b is the distribution and phase-shifting network 8th designed with high impedance and is at an impedance level of about 300 - 500 ohms.
• 2 Figure a) shows the ring line emitter 2 according to the invention with three azimuthally evenly distributed vertical resonance radiators 4a - 4c as a resonance structure. The excitement 3 is not shown. The reactance circuits 13 a- 13c are through the capacities 28a . 28b . 28c realized. With rotational symmetry of the arrangement, all capacities are equal. When tuning the ring line radiator 2 to a resonant frequency - for example f0 = 1392 MHz - arises at the vertical resonant radiators 4a - 4c in each case at the measuring path labeled 42 the course of the movement shown in FIG Antenna impedance with their caused by the radiation real resonance resistance of about 340 ohms. When connecting the excitation to the resonant radiator 4a and 4b - as in 1 - The resonance behavior of the structure is not changed in frequency. In the example of the 3 azimuthally uniformly distributed vertical resonance radiators 4a . 4b . 4c is the phase angle of rotation of the phase shifter ΔΦ = 120 °.
• 3 : Antenna 1 according to the invention as in 1 however with a distribution network 16 and a phase shifter 17 at the impedance level Z0 conventional coaxial cables ( Z0 = 50 ohms). To adapt to the high impedance impedance level of the antenna impedance 42 (Sh. 2 B) are the phase shifter 17 and one arm of the distribution network 16 each a matching network 18 downstream. The input resistance 43 at the gates T2 and T3 is - as well as in 1 - At the resonance frequency of the ring line radiator 2 high impedance and real.
• 3a : Distribution and Phase Shift Network 8th as in 3 with exemplary implementation of matching networks 18 at the two exits. The adaptation at resonance of the ring line radiator 2 is through the λ / 4 transformation lines 12a . 12b through which the transformation from the low impedance level Z0 to the high impedance impedance level of the antenna impedance 42 is given causes. The series resonant circuits, consisting of the capacities 28a . 28b and the inductors 12a and 12b allow a fine adjustment of the adaptation over an extended frequency range.
• 4 : Antenna according to the invention as in 3 but with special design of reactance circuits 13a and 13b in the two vertical resonance excitation emitters 10a . 10b each divided into a first reactance circuit 20a . 20b and a second reactance resistor circuit 21a . 21b with an intermediate point of connection 19a . 19b to connect the gates T2 and T3 of the distribution and phase shifter network 8th ,
• 5 Inventive embodiments of the total capacitive reactance circuits 13 for coupling the vertical resonance radiator to the conductive base 6 in 4
  1. a) reactance circuit 13a or. 13b the two vertical active resonance excitation emitters 10a or 10b, divided into the first reactance circuit 20a or. 20b and those about the node 19a or. 19b connected second reactance circuit 21a or. 21b , each realized by a first capacity 22a or. 22b and accordingly by the second capacity 23a or. 23b ,
  2. b) reactance circuit 13c of the remaining passive vertical resonance radiator 4c , realized by a capacity 28 , The two active resonance excitation emitters are connected as in FIG.
  3. c) reactance circuit 13a or. 13b the two vertical active resonance excitation emitters 10a or. 10b as in figure a) but with a parallel resonant circuit consisting of the parallel inductance 46a or. 46b and the parallel capacity 23a or. 23b in the second reactance resistor circuit 21a or. 21b for expanding the frequency bandwidth of the antenna.
  4. d) reactance circuit 13c of the remaining passive vertical resonance radiator 4c , realized as in Figure b) but with a serial parallel resonant circuit 25c , 25d consisting of the parallel inductance 46c . 46d and the parallel capacity 45c . 45d in series with the capacity 28 for expanding the frequency bandwidth of the antenna in accordance with the design of the reactance circuit 13a or 13b of the two vertical active resonance excitation emitters 4a or. 4b as under c). The two active resonance excitation emitters are connected as in FIG.
  5. e) representation of the frequency bands L1 and L2 for the satellite navigation with the center frequencies fm1 and fm 2 and the lower and upper cutoff frequencies fu1 . fo1 or. fu2 . fo2 , The frequency fm describes the center frequency between fu1 and fo2 ,
  6. f) reactance circuit 13a or. 13b the two vertical active resonance excitation emitters 10a or. 10b as in Figure c) but with a further parallel resonant circuit 44a or 44b in series with the first capacity 22a or. 22b towards the point of connection 19 for separately optimizing the circuit for one of the two frequency ranges L1 and L2 for designing a dual band antenna.
  7. g) reactance circuit 13c of the remaining passive vertical resonance radiator 4c , realized as in figure d), but with a further parallel resonant circuit 44c in series with the capacity 28 in the reactance circuit 13c , The two active resonance excitation emitters are connected as in FIG. 1f to form a dual-band antenna.
  8. h) reactance circuit 13a or 13b of the two vertical active resonance excitation emitters 10a bzw.10b as in Figure f) but with an additional parallel capacity 47 , formed by the, the isolated counter electrode 34 overhanging surface of the capacitance electrode 32a . 32b in the 13c and 13d with the electrically conductive coated printed circuit board 35 , The additional parallel capacity 47 allows the extension of the range of impedance matching at each node 19a . 19b at the same given compliance with the resonance condition for the ring line radiator 2 , The resonance condition and the impedance matching are made by tuning the sizes of the capacitance electrodes 32a . 32b and the isolated counter electrodes 34 made to each other.
• 6 : Adaptation ratios at the resonance frequency f0 depending on the measure "t" for the division into the first capacity 22a . 22 b and the second capacity 23a . 23c the example of the ring line radiator 2 in 4 ,
  1. a) Reflection factor at the antenna connection point 5 as an entrance gate T1 a lossless distribution and phase shifting network 8th the antenna 1 depending on the degree of division t , Adjustment is achieved with topt = 0.13.
  2. b) transformation factor at the resonant frequency f0 between the input resistance 43 With Z0 = 50 ohms of the phase shifter 17 and the resonance resistance 42 at f0 at the ring line radiator crosspoint 7 depending on the degree of division t ,
  3. c) relative power P / Pmax at f0 depending on the degree of division t ,
• 7 :
  • Antenna according to the invention with resistance adaptation by capacitive division as in the example of 4 However, with a total of 6 azimuthally offset by 60 ° to each other distributed vertical resonance radiators 4a .. 4f, of which 4 as a vertical passive resonant radiator 9a ..9d are designed.
  • The excitement 3 takes place, for example, at the two by 120 ° azimuthal staggered and excited by the difference angle ΔΦ = 120 ° vertical resonance excitation emitters 10a . 10b ,
• 8th : Antenna according to the invention as in 7 , but with a total of 8th azimuthally by 45 ° offset from each other distributed vertical resonance radiators 4a ..4h. The excitement 3 takes place, for example, at the two by 90 ° azimuthal staggered and excited by the difference angle ΔΦ = 90 ° vertical resonance excitation emitters 10a . 10b ,
• 9 : Antenna according to the invention with rectangular executed ring line 14 with four vertical resonance emitters 4a to 4d in the area of the ring line corners. Two of the vertical resonance radiators are called vertical resonance excitation radiators 10a . 10b and the other two radiators as passive vertical resonance radiators 9a . 9b executed. Impedance matching to the low impedance (Z = 50 ohm) distributed and phase shift network 8th is by capacitive subdivision, similar in the 4 . 7 and 8th reached. The ring line 14 is preferably square designed with the same ring conductor widths 15a . 15b and same loop sections 30a . 30b , Both the ring conductor width 15a . 15b as well as the loop sections 30a . 30b however, they can be chosen differently within limits.
• 10 : Rectangular ring line emitter 2 an antenna 1 according to the invention without representation of the electromagnetic excitation 3 to explain the resonant structure of the ring line radiator 2 ,
  1. a) geometric structure of the ring line radiator 2 with large differently selectable ring conductor widths 15a and 15b and the different selectable loop sections 30a and 30b , The vertical resonance emitters 4a - 4d are with their loop crosspoints 7a - 7d in the wide area of the corners of the rectangular ring conductor 14 intended. The broken line indicates approximately the course of the heavy line 24 the current density distribution of the line shaft upon excitation of the ring line radiator 2 , Due to the current displacement, the current distribution condenses towards the edge of the ring conductor 14 , The for the function in the excitation of the loop emitter 2 relevant current of the line shaft flows accordingly - characterized by the drawn as a broken line line of gravity of the power distribution 24 - even with large ladder widths 15a . 15b pushed more to the edge. This is especially true for very large ring conductor widths 15a . 15b up to the complete closure of the inner opening towards the center, which is virtually de-energized when the loop 14 is realized as a closed conductive surface.
  2. b) according to the line of gravity of the current density distribution 24 Current represented can be the resonant structure of the loop emitter 2 are generally represented by a roughly approximated equivalent circuit diagram. The individual loop sections 30a - 30d are each due to the inductive and capacitive effect of the associated section of the loop 14 including the capacitive effect of the reactance circuit 13 shown as concentrated inductive elements (Ln) and capacitive elements (Cn). Every nth loop section 30a - 30b is by a π-structure according to FIG c), consisting of a longitudinal inductance 2 * Ln and in each case a transverse capacitance Cn at the two ends thereof. The radiation damping of each horizontally oriented loop section 30a - 30b is included by the attenuation factor d of the concentrated inductance. The juxtaposition of adjacent ring line sections 30a - 30d takes place in each case via a common vertical resonant radiator 4a ..4d with contraction of the transverse capacitances Cn of the adjacent loop portions, as shown in Figure c). The slight inductive effect of the vertical resonance radiators is neglected in this basic consideration due to the small antenna height h. According to the invention advantageous conditions for the ring line radiator 2 are then reached for the resonant frequency f0 = ωo / 2π, if in each case all characteristic impedances $ZLn=\sqrt{\frac{Ln}{Cn}}$
     
     and all of them $\sqrt{Ln·Cn}=\raisebox{1ex}{$1$}\!\left/ \!\raisebox{-1ex}{$\omega On$}\right..$
     
     which represent the electrical length of the loop portion, are the same for all n.
  For rotationally symmetrical ring line radiators 2 This is always the case. Otherwise, this condition may be met, for example, by a rectangular structure of the loop emitter 2 by individual design of the ring conductor widths 15a - 15d in the individual loop sections 30a - 30b be achieved.
• 11 :
  1. a) Design of the distribution and phase shifter network 8th - for example, as in the 7 and 8th - but as to the resonant frequency f0 coordinated hybrid ring 3 through which both the power division and the phase shift takes place. When feeding at the gate T1 becomes the ring tube radiator 2 over the gates T2 and T3 excited with the phase difference of 90 °. The conclusion of Tor T4 with the ohmic terminator 40 causes at deviation of the frequency of the resonance frequency f0 broad band the partial absorption of the power of the unwanted polarization when fed to the gate T1 ,
  2. b) Idealized scattering matrix to describe the well-known fundamental wave behavior of a hybrid ring 38 in figure a) at the gates T1 to T4 , The important decoupling of the gates T1 and T4 from each other is indicated in bold by the scattering parameters S14 = 0, S41 = 0 in the matrix.
  3. c) Design of the distribution and phase shift network 8th - for example, as in the 7 and 8th - but as to the resonant frequency f0 coordinated Wilkinson divider 16 with downstream λ / 4 line phase shifter 17 at a gate T3 * for generating the phase difference of 90 °. The ohmic balancing resistor 40 Absorbs in the case of asymmetrical loading of the Wilkinson divider 16 at deviation from the resonance frequency f0 partly the currents which the unwanted polarization on the loop emitter 2 cause.
• 12 : Antenna according to the invention z. B. as in 9 with rectangular shaped loop 14 , The capacities 22a . 22b and 28 are formed in such a way that the vertical resonance emitters 4a - 4d at its lower end to individually designed capacitive capacitive electrodes 32a . 32b . 32c . 32d are formed. By interposition between these and the electrically conductive coated circuit board designed as an electrically conductive base 6 located dielectric plate 33 are the capacities 28 for coupling the passive vertical resonance radiators 4c . 4d to the electrically conductive base 6 designed. For the capacitive coupling of the active vertical resonance radiators 10a . 10b to the gates T2 and T3 of the distribution and phase shifter network 8th this connection is in each case one of the conductive layer insulated, flat counter electrode 34 designed. The counter electrodes 34 can thus be used as connection points 19 the antenna in 9 be executed and can serve as connection points for the gates T2 and T3 of the distribution and phase shifter network 8th , as in 13 indicated, serve.
• 13 : Antenna 1 similar to in 11 However, the dielectric effect of the dielectric plate 33 realized by an air gap. Typical dimensions of a loop emitter 2 for the frequency range L1 are for square antennas a dimension of 30mm to 40mm and for the height h = 8mm. For resistance adaptation of the preferably at a low impedance level ( Z0 ) carried out distribution and phase shifter network 8th is the capacitive subdivision as in 9 intended.
  1. a) The exemplary associated reactance circuits 13a . 13b and 13c are in the 5a and 5b shown. The capacity 28 the capacitance electrode 32b . 32c against the electrically conductive base 6 or the capacity 22a . 22b against the isolated counter electrode 34a . 34b is about 0.3pF each. The gates T2 and T3 of the distribution and phase shifter network 8th with its real input resistance 43 are each at a counter electrode 34a . 34b as a link point 19a . 19b connected. The capacitive subdivision for impedance matching is in each case by the capacity 22a . 22b and the capacity 23a . 23b given. This between the connection point 19a . 19b and the earth point 11 introduced capacity 23a . 23b is on the back of the coated circuit board 35 attached as an SMD component. For this purpose, on the back of the circuit board insulated pads 29 designed as contact bases, each via a via 26 with the counter electrode 34 on the one hand and on the other hand, for example, about capacity 23a . 23b (Sh. 4 and , 5a) are connected to ground.
  2. b) the reactance circuits 13a . 13b the two resonance excitation emitters 10a . 10b are as in Figure a) according to 5a or as in 5c designed. When designing after 5c is the one on the back of the coated circuit board 35 introduced capacity 23a . 23b an inductance 46a . 46b connected in parallel as an SMD component. In contrast to Figure a) are to form the reactance circuits of the two passive resonant radiator 9a . 9b the capacitance electrodes 32b . 32c on the coated circuit board 35 isolated counterelectrodes 34 compared. Starting from the counter electrode 34 as a contact base are on the back of the circuit board 35 the parallel inductance 46 and the parallel capacity 45 to the earth point 11 on the circuit board 35 switched so that the reactance circuit 13 in 5d is realized. In embodiment of all reactance circuits 13 all vertical resonant radiators, each with a capacitance electrode 32 and an opposing isolated counter electrode 34 can all circuits in the 5c and 5f for the resonance excitation emitter 10a . 10b by using SMD components on the back of the PCB 35 will be realized. Similarly, the reactance circuits for the passive resonant radiators 9a . 9b by wiring of the isolated counterelectrodes 34 with SMD components on the back of the PCB 35 according to the in the 5d and 5g specified circuits can be realized.
  3. c) As under 5h described describes the additional parallel capacity 47 for free design of the impedance matching while retaining the resonant properties of the ring line radiator 2 , The figure shows the large area coverage of the isolated counter electrode 34a . 34b with the capacitance electrode 32a . 32b the resonance excitation emitter 10a . 10b , The one in the vertical projection of the capacitance electrode 32a . 32b opposite the isolated counter electrode 34 existing planar projection forms with the electrically conductive layer of the coated printed circuit board 35 the additional parallel capacity 47 , The reactance circuit of the two passive resonant radiators 9a . 9b are designed as in Figure a).
  4. d) the reactance circuits 13 the two resonance excitation emitters 10a . 10b are as in Figure c) with an additional parallel capacity 47 designed. The exemplary possible reactance circuits 13 the two passive resonant radiators 9a . 9b can be designed as described in FIG.
• 14 : In the picture is the top of the circuit board 35 an antenna 1 illustrated according to the invention, to which the electrical loop emitter 2 is put on. For the broadband design of an antenna 1 according to the invention - which, for example, both frequency ranges L1 and L2 with a center frequency fm (sh. 5e) - is the reactance circuit 13 as in the 5c or. 5d designed. This is for the reactance circuit in the two active vertical resonance excitation emitters 10a . 10b each the reactance circuit 13a . 13b in 5c and for the two passive vertical resonance radiators 9a . 9b the reactance circuit 13c in 5d intended. This is the example in 13a achieved by that for all capacitance electrodes 32 one counterelectrode each 34 is present and that through the capacitance electrodes 32 caused capacity 22a . 22b or. 28 on all vertical resonance radiators 4a - 4d a parallel connection from a parallel capacity 23a . 23b or. 45 and a parallel inductance 46a . 46b , or. 46 - shown as SMD components - between the counter electrode 34 and the electrically conductive base 6 connected in series. 13a shows the top of the PCB 35 with vias 26 on the counter electrodes 34 under the capacitance electrodes 32 , In 14b is the bottom of the circuit board 35 with the pads 29 shown, which with the isolated counter electrodes 34 over the vias 26 are connected,. The dummy elements are as SMD components according to the 5c and 5d appropriate. In addition, the two join points 19 with the respective connected series circuit of the capacity 28a . 28b and the inductance 27a . 27b as parts of in 3a included matching circuit 18 of the distribution and phase shifter network 8th to form the gates T2 and T3 shown.
• 15 Shows an antenna according to the invention with four azimuthally on the circumference of the loop 14 equally distributed resonant excitation emitters 10a to 10d , All radiators are excited in accordance with a rotating wave, each with 90 ° phase difference between adjacent radiators. For this purpose, a first and a second distribution and phase shifter network ( 8th ) with phase angle ΔΦ = 90 °, the first distribution and phase shifting network ( 8th ) with his gates T2 . T3 with a first pair of resonance excitation emitters 10a . 10b is coupled and the second distribution and phase shifting network ( 8th ) with his gates T2a . T3a with a second pair of exciter radiators opposite the first pair 10c . 10d is coupled. To stimulate a current wave are each other opposite resonance excitation emitters 10a . 10c or. 10b . 10d each excited with a phase difference of ΔΦ = 180 °. For this purpose is another distribution and phase shift network 31 . 39 with a phase shift of ΔΦ = 180 °, which coincides with its gate T 2c with the gate T1 of the first distribution and phase shifter network 8th . 36 and with his gate t3c with the gate T1b further distribution and phase shifting network 8th . 38b connected is. Again, preferably according to the invention each a hybrid ring 38 for use. The 180 ° phase network 31 can be exemplified as a Wilkinson divider 39 with a λ / 2-long delay line 17 (Sh. 11c) be used. All connections are selected in such a way that sets the conductive wave in the desired direction of rotation on the loop. With this arrangement, the particular advantage of the azimuthally symmetrical excitation is connected, so that the adjustment of the arrangement can be carried out particularly problem-free. As a limitation, however, the increased effort is noted.

For the design of a dual-band multiband antenna according to the invention - for example for the frequency ranges L1 and L2 - is the reactance circuit 13 designed multi-frequency in each case in such a way that the resonance of the ring line radiator 2 in the separate frequency bands L1 and L2 each separately and correspondingly as frequency f01 and as a frequency f02 given is. This is done starting from the pads 29 in 14b the back of the coated circuit board 35 redesigned in such a way that the reactance circuits 13a . 13b , or. 13c are realized.

In the following, the invention will be described again in connection with its advantageous embodiments.

In a basic form contains an antenna 1 according to the invention a ring line radiator already described above 2 as he is for example in 2a for a rotationally symmetrical arrangement with three along the electrically short ring conductor 14 equidistantly distributed vertical resonance radiators 4a . 4b . 4c is shown. From the ring conductor width 15 and the length of a loop section 30a , 30b, etc. results in an inductive effect, which together with the capacitive reactance circuit 13 in each of the vertical resonant radiators causes a resonance behavior with a resonant angular frequency ω0 = 2π * f0, which is due to the frequency characteristic of the antenna impedance 42 in 2 B expresses.

According to the invention, this resonance structure is at the natural resonance frequency f0 capable, with appropriate excitement over two vertical resonance radiators 4a . 4b - here for identification as a vertical resonance excitation emitter 10a . 10b in the phase of the azimuthal position of the vertical resonance radiators 4a . 4b corresponding signals - for example as in 1 represented - to convey a current current wave in exactly one direction. The structure of the ring line radiator 2 in 2a itself thus does not yet have a feature for the preference of one of the two possible directions of movement of the current wave. The direction of travel is solely by connecting the distribution and phase shift network 8th according to the sign of the phase angle ΔΦ , as in 1 , set. The phase difference ΔΦ between these signals is therefore chosen according to the invention such that it at the resonant frequency f0 of the ring line radiator 2 the phase difference ΔΦ a running on the loop current shaft between the loop coupling points 7a and 7b the two resonance excitation emitters 10a and 10b equivalent. The direction of rotation of the current wave is thus by the sign of the phase difference ΔΦ the excitation signals in conjunction with the position of the two loop coupling points 7a . 7b specified. The direction of travel is thus exclusively by connecting the distribution and phase shifter network 8th according to the sign of the phase angle ΔΦ , as in 1 , set.

Naturally, the condition for the excitation of a running direction can be exactly only at the resonance frequency f0 be fulfilled and the proportion of unwanted running direction and thus the cross polarization increase with increasing deviation from this frequency. As a particular advantage of the invention, however, results in the very large bandwidth for a sufficiently large suppression of the cross polarization of the inventive requirement for a real input resistance 43 both goals T2 . T3 of the distribution and phase shifter network 8th , where the resonant frequency f0 the ring line emitter 2 by the excitation 3 at the resonant frequency f0 not changed. A reactive power exchange between the ring line radiator 2 and the excitement 3 thus finds at the resonant frequency f0 not happening. Particularly high bandwidth of the suppression of the cross polarization is achieved if - as in 1 represented - the gates T2 and T3 directly to the vertical resonance excitation emitters 10a . 10b connected so that they are connected to the corresponding loop crosspoints 7a . 7b are effective.

According to the invention is thus provided, the ring line radiator 2 and the phase shift around the angle ΔΦ in the distribution and phase shifting network 8th to co-ordinate in the way that this Angle at the resonant frequency f0 the electrical length of the loop section 30 as an angle measure the proportion of the electrical length 2π of the loop 14 between the two vertical resonance excitation emitters 10a . 10b corresponds to which the gates T2 and T3 are connected. In this case, between the vertical resonance excitation emitters 9a . 9b at the ring conductor 14 further, but passive vertical resonance emitters 4 - as in the 7 and 8th shown - be arranged.

This results in a total of N> 3 vertical resonance radiator 4a . 4b . 4c . 4d and thus the same number of loop sections 30a . 30b . 30c . 30d and over the loop 14 equally distributed crosspoints 7 the phase angle of the distribution and phase shift network set according to the invention 8th to ΔΦ = n * 2π / N <180 °, if between the crosspoints 7 the two vertical resonance excitation emitters 10a . 10b n loop sections 30 are selected.

For rotationally symmetric arrangements with a high number of N ring line sections in total, it is advantageously possible to achieve a large frequency bandwidth for the suppression of the cross polarization.

For special applications, such as for the superposition of circularly polarized radiation with azimuthal phase distribution of 2π and a circularly polarized radiation with azimuthal phase distribution of 4π in a Antennenendiversity system with two antennas, the invention can advantageously also on a loop emitter 2 with an electrical length of 4π be applied as follows. In a simple special case with a polygonal loop emitter 2 with N = 8 ring line sections distributed over the electrical length 4π azimuthally equal 30 become two mutually adjacent vertical resonance excitation emitters 10a . 10b (n = 1) via a distribution and phase shifting network 8th excited with ΔΦ = π / 2. By analogy, with N = 16 vertical resonant radiators and n = 2 loop sections between the two vertical resonant excitation radiators 10a . 10b Accordingly, the phase angle ΔΦ = n * 4π / N also ΔΦ = π / 2 set. In an expedient and space-saving way, two ring line emitters 2 be arranged with different phase distribution concentric to each other in such a way that the radiator is surrounded with azimuthal 2π- phase distribution of the radiator with 4π-phase distribution.

Advantageously, the excitation of the loop takes place 14 with the electrical length 2π of the resonant frequency f0 matched loop line radiator 2 in 1 in the way that the gates T2 . T3 a real high impedance input resistor 43 own and over the two vertical resonance excitation emitters 10a . 10b directly to the loop connection points 7a . 7b are connected. In this case, both the distribution network 16 as well as the phase shifter 17 high impedance corresponding to the high resonance resistance 42 designed.

In an advantageous embodiment of the invention are in 3 the distribution network 16 and the phase shifter 17 low impedance, ie for the impedance level Z0 (eg Z0 = 50 ohms) designed. For transformation into the high impedance level of the loop emitter 2 each of the two output branches is a matching network 18 downstream. For this purpose, for example, in each case the in 3a illustrated λ / 4 transformation line 12a . 12b with the series resonant circuit from the inductor 27a . 27b and the capacity 28a . 28b ,

In a particularly advantageous embodiment of the invention, the impedance matching takes place between the low-impedance - at the impedance level Z0 - designed gates T 2 . T3 of the distribution and phase shifter network 8th and at the loop connection points 7a . 7b respectively present high-impedance resonance resistor 42 of the ring line radiator 2 - as in 4 represented - by serial subdivision of the capacitive reactance circuit 13a . 13b with design of a connection point 19a . 19b in between to connect the two gates T2 . T3 with their respective real input resistance 43 at the impedance level Z0 , The frequency bandwidth of cross-polarization suppression in circular polarization antennas depends to a large extent on the bandwidth of the antenna and thus on its electrical size. In the case of the electrically small ring line emitters considered here 2 with dimensions, such as those related to 13 In the frequently prescribed requirement of the cross polarization pitch of 15db and above, each of the frequency bands is possible L1 and L2 to be recorded separately.

For this purpose, the invention is in the example of the 4 illustrated subdivision of the reactance circuit 13a . 13b the two vertical resonance excitation emitters 10a . 10b , which also in 5a is shown. The subdivision shown is purely capacitive by the first capacity 22a . 22b and the second capacity 23a . 23b given. In this case, it is important according to the invention that in the vertical resonance excitation emitters 10a . 10b effective resulting capacitance each of those in the other passive vertical resonant radiators 9a . 9b - represented by the capacity 28 in the 4 . 7 . 8th and in 5b - Corresponds and thus the resonance of the ring line radiator 2 at the resonant frequency f0 is retained by the capacitive subdivision. An exchange of reactive power between the loop emitter 2 and the electromagnetic excitement 3 at the point of connection 19 is at the resonant frequency f0 locked out. This feature, however, coincides with increasing frequency deviation from the resonant frequency f0 due to the frequency dependence of the loop emitter 2 and the frequency dependence of the distribution and phase shift network 8th with respect to the impedance matching and the phase shift is no longer completely closed. Advantageously, however, the inventive adjustment described with increasing frequency deviation leads to a comparatively particularly small increase in the cross polarization. The antenna according to the invention is thus advantageous broadband with respect to the suppression of the cross polarization.

On the other hand, it is hardly possible with a small antenna described here, the entire frequency band of both frequency ranges L1 , and L2 between the frequencies fu2 and fo1 (Sh. 5e) with a cover antenna described above to capture without additional measures.

According to the invention allows the design of the reactance circuit 13a . 13b - shown in 5c - In the two vertical resonance excitation emitters 10a . 10b and the reactance circuit 13c in 5d a single-band antenna, which both frequency bands L1 and L2 detected.
This is inventively achieved in that both in the second reactance circuits 21a . 21b (Sh. 5c) the two vertical resonance excitation emitters 10a . 10b by parallel connection of the parallel inductance 46a . 46b to the existing second capacity 23a . 23b as well as in the reactance circuits 13c (Sh. 5d) the passive vertical resonance radiator 9c each a parallel resonant circuit with the inductance 46 and the capacity 45 is added. Here, the resonance frequency of all parallel oscillation circuits is in the frequency range between the upper frequency band L1 and the lower frequency band L2 (eg fm in 5e) chosen in such a way that the reactance of the resonant circuits in the frequency band L1 capacitive and in the frequency band L2 is inductive. The reactance of all reactance circuits 13a . 13b . 13c Overall, however, is in both frequency bands L1 and L2 nevertheless capacitive.

An advantageous continuation of the invention according to the above design of the reactance circuits takes place in the design of a dual-band antenna in such a way that both the first capacity 22a . 22b the first reactance circuit 13a 13b the vertical resonance excitation emitter 10a . 10b as well as the capacity 28 the reactance circuit 13c of the passive vertical resonance radiator 9c in each case a further parallel resonant circuit 44a . 44b . 44c is connected in series. This makes it possible, for example, advantageously that both the reactance x1a . X 1b the first reactance circuit 20a . 20b and also the reactance x2a . x2b the second reactance circuit 21a . 21b and thus the reactance X the reactance circuit 13a . 13b in both frequency bands L2 and L2 is capacitive and also favorable values for the pitch t can be selected separately for the capacitive division for both frequency bands. The corresponding reactance circuits 13a . 13b and the reactance circuit 13c are in the 5f and 5g shown. A suitable dimensioning of all reactive elements in the reactance circuit 13a . 13b allows the capacitive division described above by choosing the resonant frequency of the parallel resonant circuits with respect to the center frequencies fm1 and fm 2 in view of the optimization described above t -topt to reach.

In a particularly advantageous embodiment of the invention, the broadband suppression of the unwanted polarization direction LHCP of a RHCP antenna provided with a hybrid ring 38 as a distribution and phase shifting network 8th to excite the square ring tube radiator 2 in 13 reached. In a usual, known form, there is a hybrid ring 38 from λ / 4-long pipe sections, as in 11a shown. It is also known that the pipe sections can be simulated by concentrated dummy elements. Basically, a hybrid ring 38 for different impedance levels. However, production is particularly economical for the low impedance level Z0 ,

A hybrid ring 38 but only for a certain frequency - generally about the center frequency of a frequency band ( fm1 . fm 2 ) - be perfectly balanced and through the idealized litter matrix regarding its gates T1 to T4 in 11b to be discribed. At deviating frequencies arises at the antenna in 13 naturally when excited at the gate T1 and upon excitation of the ring line radiator 2 over the gates T2 and T3 in addition to the desired radiation in RHCP mode proportionately the unwanted radiation in the opposite direction, ie the LHCP mode. In addition, with increasing frequency deviation from the resonant frequency f0 increasing reactive component of the antenna impedance, which also increases the LHCP mode. An optimal interaction between the hybrid ring 38 and the ring line radiator 2 with respect to the suppression of the cross polarization is achieved according to the invention when the hybrid ring 38 in terms of its real input resistance at the gates T2 and T3 and its required phase of 90 ° to the frequency f0 which is also the resonance frequency f0 the corresponding ring line radiator 2 forms. As a special feature of a hybrid ring 38 causes the wiring of the gate T1 decoupled gates T4 with a resistance of the size Z0 According to the invention, that the unwanted LHCP content in the radiation even at frequency offset from the frequency f0 for which both the loop emitter 2 as well as the hybrid ring 38 is tuned, is largely absorbed.

In an advantageous further development of the invention comes as distribution and phase shifter network 8th the Wilkinson divider 39 as a distribution network 16 with downstream phase shifter 17 in 11 for use. The signal path, starting from the gate T1 , branches into two λ / 4-long cables whose live ends with the ohmic Symmetrierungswiderstand 41 connected as compensation resistance. The gate T3 * is, for example, a λ / 4-long line as a phase shifter 17 downstream for a phase rotation of 90 °. This line can also be simulated by concentrated components. Only in the case of resistance-resistant completion of the goals T2 and T3 the system is balanced. Similar to using the hybrid ring described above 38 this only applies to the resonance frequency f0 to. With increasing deviation from this frequency is the symmetry of the Wilkinson divider 39 disturbed. According to the invention, the resulting undesired LHCP content is above the ohmic balancing resistance 41 equally muted.

To meet particularly high demands on the suppression of cross polarization in the two frequency bands L1 and L2 It is inventively provided for each frequency band a separate antenna according to the invention 1 use. In an extremely advantageous continuation of the invention, the two ring line radiators 2 arranged concentrically to each other with the proviso that the loop emitter 2 for the higher frequency in the band L2 from the larger ring channel radiator for the lower frequency in the band L1 is surrounded. Naturally, this is the increased cost of two separate and matched to the respective frequency band distribution and phase shifter network 8th to accept. By taking advantage of the frequency spacing between different frequency bands, a plurality of antennas according to the invention assigned to one frequency band can be used in this way 1 be formed, the ring line radiator 2 be arranged concentrically with each other in such a way that the loop emitter 2 for the highest frequency band in the frequency is arranged innermost and each further ring line emitter 2 a ring line radiator 2 for the frequency band nearest higher in frequency. Such an arrangement has the particular advantage that hereby besides the two frequency bands L2 and L1 also the frequency band lower in frequency L5 , which also serves for satellite navigation, can be detected. From the summary and the parallel evaluation of all signals from these frequency bands in the navigation receiver, the quality of the location results can be extremely improved even under difficult reception conditions.

Such a concentric arrangement of the ring line radiator 2 moreover has the advantage that in the center of the arrangement, a linear, vertically polarized antenna for other radio services can be arranged without the properties of the ring line emitters 2 to impair.

LIST OF REFERENCE NUMBERS

antenna 1 Loop radiator 2 electromagnetic excitation 3 vertical resonant radiators 4, 4a, 4b, 4c, ... Antenna connection point 5 Leading base area 6 Ring line coupling points 7, 7a, .. 7d, ... Distribution and phase shifting network 8th vertical passive resonance radiators 9, 9a, 9b vertical resonance excitation emitters 10a, 10b, .. Ground connection point 11 λ / 4 transformation line 12 Reactance circuit 13, 13a, ... 13c, ... loop 14 Ring conductor width 15, 15a, .., 15c ,. distribution network 16 Phase shifter 17 matching 18 junction 19 first reactance circuit 20a, 20b second reactance circuit 21a, 21b first capacity 22a, 22b second capacity 23a, 23b Heavy line of the current density distribution 24 Serial parallel resonant circuit 25 via 26 inductance 27a, 27b capacity 28a, 28b pad 29 Ring line section 30a, 30b, 30c, 30d 180 ° phase shift network 31 capacitance electrode 32a, 32b, 32c, 32d, dielectric plate 33 isolated counter electrode 34 Coated circuit board 35 support structure 36 37 antenna impedance hybrid ring 38 Wilkinson divider 39 ohmic terminating resistor 40 ohmic balancing resistance 41 Antenna impedance, resonance resistance 42 real input resistance 43 another parallel resonant circuit 44 parallel capacitance 45 parallel inductance 46, 46a, 46b Additional shunt capacitance 47 Stretched length of the ring line radiator L reactance X, X1a, X1b, X2a, X2b, Xc height H Gates T1, T2, T3, T4 impedance ZL center Z

QUOTES INCLUDE IN THE DESCRIPTION

This list of the documents listed by the applicant has been generated automatically and is included solely for the better information of the reader. The list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions.

Cited patent literature

• DE 4008505 A [0002]

Claims (23)

An antenna (1) for receiving circularly polarized satellite radio signals, comprising at least one horizontally oriented conductor loop arranged above a conductive base (6), with an arrangement for electromagnetic excitation (3) of the conductor loop connected to an antenna connection point (5), comprising the following features the conductor loop is designed as a ring line radiator (2) running through a polygonal or circular closed ring line (14) in a horizontal plane with the height h <0.15 of the free space wavelength λ over the conductive base surface (6), - the ring line radiator (2 ) forms a resonant structure and is excitable by electromagnetic excitation in such a way that adjusts the current distribution of a current line wave in a single direction of rotation on the ring line (14), the phase difference over a revolution is just 2π - are on the circumference of the ring line radiator (2 ), to ring line coupling points (7) with the ring line emitter (2) galvanically coupled, vertical and the conductive base (6) extending down vertical radiator (4) - there is a distribution and phase shifter network (8) with an input side to the antenna connection point (5 ) coupled gate T1 and with two ports T2, T3 present, whose transmission coefficients S ₁₂ and S ₁₃ differ from each other by a phase angle ΔΦ characterized by the following features - the ring line radiator (2) is designed as a short ring line (14) whose scope ( L) is shorter than the free space wavelength λ. there are at least three vertical radiators (4) as vertical resonance radiators (4, 4a, 4b, 4c,...), which are coupled via reactance circuits (13) with capacitive reactance X to the base (6), through which the Resonance of the short loop (14) is made. - The input resistance of each of the ports T2, T3 of the distribution and phase shifter network (8) for electromagnetic excitation of the line shaft on the ring line (14) is in each case a substantially real resistance. - The distribution and phase shifter network (8) two of the ring line coupling points (7a, 7b) are assigned in such a way that each of the gates T2, T3 of the distribution and phase shifter network (8) coupled to its associated ring line cross-point (7a, 7b) is. Antenna (1) to Claim 1 , characterized in that the distribution and phase shifting network (8) are associated with two of the vertical resonance radiators (4a, 4b) as vertical resonance excitation radiators (10a, 10b) in such a way that each of the gates T2, T3 of the distribution and phase shifting network (8) in each case via one of the two vertical resonance excitation emitters (10a, 10b) is coupled to the ring line coupling point (7a, 7b) and the other vertical resonance emitter (4c, 4d, ..) as a passive vertical resonance emitter (9a, 9b, etc ) are effective. ( 1 ) Antenna (1) to Claim 2 characterized in that each of the ports T2, T3 is respectively directly connected via one of the vertical resonance excitation emitters (10a, 10b) to its ring line crosspoint (7a, 7b) and the input resistance of the gates T2, T3 is high impedance real in the manner, that there is resistance matching between this and the high-impedance resonant resistance of the ring line radiator (2). ( 2 ) Antenna (1) to Claim 1 to 3 , characterized in that in the distribution and phase shifter network (8) a distribution network (16) and a downstream in one of its output branches phase shifter (17) are provided, which are both designed for the impedance level Z ₀ conventional coaxial and in each of the two output branches a loss-free matching circuit (18) for the transformation of the low impedance level Z ₀ in the high-impedance real input resistance of the gates T2, T3 to adapt to the high impedance resonance resistance of the ring line radiator (2) is present. ( 3 ) Antenna (1) to Claim 1 to 4 characterized in that by the action of the vertical resonant radiators (4, 4a, 4b, 4c) loaded with the reactance resistor circuit (13, 13a, ... 13c, ...), the stretched length L of the loop (14 ) of the ring line radiator (2) in resonance, starting from approximately the free space wavelength λ, is shortened to approximately one third of the free space wavelength λ. Antenna (1) to Claim 1 to 5 characterized in that the reactance circuit (13) with which each of the passive vertical resonance radiators (9, 9a, 9b) and also the vertical resonance excitation radiators (10a, 10b) is coupled to the base (6) is represented by a capacitance (28 ) is formed. ( 3 ) Antenna (1) to Claim 2 and 4 to 6 characterized in that the reactance circuit (13) with which the vertical resonance excitation emitters (10a, 10b) are coupled to the base (6) are each a series circuit of a first reactance circuit (20a, 20b) and a second reactance circuit (21a, 21b) is formed with a connecting point (19) connecting these, with which in each case one of the ports T2, T3 is connected. ( 4 ) Antenna (1) to Claim 2 and 4 to 7 characterized in that the reactance circuit (13) to which the vertical resonance excitation emitters (10a, 10b) are coupled to the ground plane (6) and the first reactance circuit (20a, 20b) of each vertical resonance excitation emitter (10a, 10b) is connected through one first capacitance (22a, 22b) and the second reactance resistor circuit (21a, 21b) are formed by a second capacitance (23a, 23b), and the size and ratio of the first and second capacitances are each selected to be between the high impedance Resonance resistance of the ring line radiator (2) and a lower impedance real input resistance of the gates T2, T3 each resistance adjustment is given. ( 4 ) Antenna (1) to Claim 3 and 5 to 8th , characterized in that the reactance circuits (13c, etc.), with which the passive vertical resonant radiators (9c, etc.) are coupled to the electrically conductive base (6), in each case from the series connection of a capacitance (28) and a parallel oscillation circuit, consisting of a parallel capacitor (45) and a parallel inductor (46) - are formed and the first reactance circuit (20a, 20b) of the vertical resonance excitation emitter (10a, 10b) is formed as a first capacitor (22a, 22b) and the second Reactive resistance circuit (21a, 21b) in each case as a parallel resonant circuit - consisting of a second capacitor (23a, 23b) and a parallel inductance (46a, 46b) is designed and the resonant frequency of all parallel arcing circuits respectively in the frequency range between the upper frequency band L1 and the lower frequency band L2 in the Way is chosen, that the reactance of the resonant circuits in the frequency band L1 kapaziti v and in the frequency band L2 is inductive. ( 5c . 5d . 5e) Antenna (1) to Claim 9 characterized in that, however, between the capacitance (28) and the parallel oscillation circuit of the parallel capacitance (45) and the parallel inductance (46) of the reactance circuits (13c, etc.) of the passive vertical resonance radiator (9c, etc.) each have a further parallel resonant circuit (44c , etc.) are connected in series and the first capacitance (22a, 22b) in the first reactance circuit (20a, 20b) of the vertical resonance excitation emitters (10a, 10b) towards the connection point (19a, 19b) also each have another parallel resonant circuit ( 44a, 44b) and the parallel oscillation circuits in the first reactance circuit (20a, 20b) and the second reactance circuit (21a, 21b) of the vertical resonance excitation radiators (10a, 10b) are tuned such that the reactance (Xa, Xb ) is capacitive in both frequency bands L1, L2 and their relationship to each other is according to the optimum measure topt for adaptation and the reactance circuits (13c, etc.) of the passive vertical resonance radiators (9c, etc.) and the vertical resonance excitation radiators (10a, 10b) are equalized in frequency characteristics. Antenna (1) to Claim 8 to 10 characterized in that the first capacitance (22a, 22b) of the first reactance circuit (20a, 20b) of the vertical resonance excitation emitters (10a, 10b) and the capacitances (28) in the reactance circuits (13c, etc.) of the passive vertical resonant emitter (10) 9c, etc.) are formed in such a way that all the vertical resonant radiators (4a, 4b, 4c, etc) are formed at their lower end into individually designed area capacitance electrodes (32a, 32b, 32c, 32d), that the capacitances (22a , 22b, 28) by interposing a dielectric plate (33) between the two-dimensional capacitance electrodes (32a, 32b, 32c, 32d) and the electrically conductive base (6) designed as an electrically conductive printed circuit board (35) for coupling the passive vertical resonance emitters ( 9 c, 9 d, etc.) to the electrically conductive base (6) are designed, and that for the capacitive coupling of the vertical resonance excitation emitter (10 a, 10 b) on the electrically conductive base (6) each one of the conductive layer of the coated printed circuit board (35) isolated, planar counter electrode (34) for connecting the second reactance resistor circuit (21a, 21b) is designed. ( 13a, b . 5 ) Antenna (1) to Claim 1 - 11 characterized in that the vertical radiators (4a, .. 4d) for capacitive coupling as in Claim 10 are formed at their lower end to individually designed area capacitance electrodes (32a, 32b, 32c, 32d), but that for coupling the passive vertical resonant radiator (9c, 9d) each one of the conductive layer of the coated printed circuit board (35) isolated, planar counter electrode (34) for connecting the parallel capacitor circuit (45a, 45b) and the parallel inductor (46a, 46b) formed parallel resonant circuit to the ground terminal point (11) is formed. ( 13a, b . 14 . 5 ) Antenna (1) to Claim 11 and 12 characterized in that the conductive structure consisting of the ring conduit (14) and the associated vertical radiators (4, 4a-d) is fixed by a dielectric support structure (36) such that the dielectric plate (33) is in the form of an air gap is realized. ( 13a, b . 14 ) Antenna (1) to Claim 1 to 13 , characterized in that a total of three vertical resonant radiators (4a, 4b), with azimuthally equally distributed loop coupling points (7) are present, of which two as vertical resonance excitation radiators (10a, 10b) via the gates T2, T3 of a distribution and phase shifter network (8) whose phase angle ΔΦ = 120 degrees. ( 1 . 3 . 4 ) Antenna (1) to Claim 1 to 14 , characterized in that a plurality of N> 4 vertical resonant radiators (4a, 4b, 4c, ...) with over the extended length L of the ring line (14) equally distributed crosspoints (7) is present, so that N loop portions (30a , 30d) are formed and between the coupling points (7a, 7b) of the two vertical resonance excitation emitters (10a, 10b) n ring line sections are selected and the phase angle ΔΦ = (n + 1) * 2π / N <180 ° is selected. ( 7 ) Antenna (1) to Claim 1 to 15 , characterized in that there are four vertical resonance radiators (4a, 4b, 4c, 4d) with azimuthally equal length ring line sections (30a, 30b), two of which azimuthally successive than vertical resonance excitation radiators (10a, 10b) over the gates T2, T3 of a distribution and phase shifter network (8) whose phase angle ΔΦ = 90 degrees. Antenna (1) as in Claim 16 , characterized in that the ring line (14) but rectangular with four loop cross-coupling points (7a, .. 7d) at the corners and with different lengths of the loop portions (30a, 30b) and with different ring conductor widths (15a, 15b) designed is, wherein by appropriate choice of the ring conductor widths (15a, 15b) for the ring line sections (30a, 30b) respectively the same characteristic impedance ZL and the same effective electrical length are set. ( 9 . 10a . 10b . 10c) Antenna (1) to Claim 1 to 17 characterized in that the distribution and phase shifter network (8) for a phase shift by the phase angle ΔΦ = 90 °, a hybrid ring (38) containing, in addition to the two gates T2 and T3, a fourth, with a resistive terminating resistor (40) closed gate T4 and which is decoupled from the gate T1 when the doors T2 and T3 are closed with respect to the shaft resistance. ( 11a . 11b) Antenna (1) to Claim 1 to 18 , characterized in that the distribution and phase shifting network (8) for a phase shift by the phase angle ΔΦ = 90 ° includes a Wilkinson divider (39) with an ohmic balancing resistor (41), the input at the gate 1 and one of the two output branches of the Wilkinson divider (39) has a phase shifter (17) connected to the phase angle ΔΦ = 90 ° to the gate T3 and at the other of the two output branches Tor T2 is formed. ( 11c) Antenna (1) to Claim 1 to 19 , characterized in that the lossless matching circuit (18) for transforming the low impedance level Z0 into the high impedance real input resistance (43) in both output branches (T2, T3) of the distribution and phase shift network (8) each substantially by a lambda Transformation line (12) and a series circuit of an inductance (27a, 27b) and a capacitance (28a, 28b) for fine adjustment of the high-impedance real input resistance (43) of the gates T2 and T3 is formed. ( 3a) Antenna (1) to Claim 1 to 20 , characterized in that all vertical resonant radiators (4a, 4b, 4c, etc.) along the ring line (14) are approximately equally distributed, so that none of the distances between adjacent ring line crosspoints (7a, 7b, 7c) on the circumference of the ring line radiator (2) is less than half the equidistant spacing over the extended length of the loop (14). Antenna (1) to Claim 1 to 21 Characterized in that there are for sufficiently large frequency spacing between different frequency bands more respectively assigned a frequency band antenna 1 according to the invention, the ring line emitters 2 are arranged concentrically to each other in such a way that the ring main radiator 2 arranged to be the highest in the frequency band intimately and each further ring line emitter 2 surrounds a ring line emitter 2 for the next higher frequency band. Antenna (1) to Claim 1 to 22 , characterized in that a first and a second distribution and phase shifter network (8) with phase angle ΔΦ = 90 ° and four designed as excitation emitters vertical radiators azimuthally distributed evenly on the circumference of the ring line (14) are provided, wherein the first distribution and phase shifter network (8) is coupled with its ports T2, T3 to a first pair of resonant excitation emitters 10a, 10b, and the second distributing and phase shifting network (8) having its ports T2a, T3a with a second pair of exciter emitters 10c opposite the first pair, 10d is coupled and another distribution and phase shifting network (8) with ΔΦ = 180 ° is present, which is connected to the excitation of the circulating line shaft with its gates T2c, T3c with the gates T1, T1a in such a way that on the loop adjusts the rotating shaft.

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