EP3483983B1 - Receiving antenna for satellite navigation on a vehicle - Google Patents

Description

The invention relates to an antenna for receiving circularly polarized satellite radio signals, in particular for satellite radio navigation, according to the preamble of claim 1 ( EP 2 296 227 A2 ).

Due to polarization rotations on the transmission path, satellite radio signals are usually transmitted using circularly polarized electromagnetic waves and are used in all known satellite navigation systems. Modern navigation systems provide for global accessibility in conjunction with high navigation accuracy in mobile navigation to evaluate the simultaneously received radio signals from several satellite navigation systems. Such systems that receive in a network are summarized under the term GNSS (Global Navigation Satellite System) and include well-known systems such as GPS (Global Positioning System), GLONASS, Galileo and Beidou etc. Satellite antennas for navigation on vehicles are usually on the electrically conductive outer skin of the vehicle body. Circularly polarized satellite reception antennas are used, such as those from EP 2 296 227 A2 or the DE 40 08 505 A are known. Antennas that are characterized by a low overall height combined with cost-effective manufacturing are particularly suitable for installation on vehicles. This particularly includes, for example, the one from the EP 2 296 227 A2 well-known ring line radiators designed as a resonance structure with a small construction volume, which is particularly required for mobile applications. The antenna has a small footprint and is very low, with a height of less than a tenth of the free space wavelength.

Patch antennas are known as other receiving antennas for satellite navigation on vehicles, but they are more complex to construct than antennas stamped from sheet metal. A challenge of the satellite antennas for GNSS is the requirement for a large frequency bandwidth, which for GPS, for example, is specified by the frequency band L1 with the center frequency 1575 MHz (required bandwidth approx. 80 MHz) and the frequency band L2 with the center frequency 1227 MHz (required bandwidth approx. 53 MHz). This requirement is met, for example, by a separate antenna assigned to one of the frequency bands L1 or L2, or by a broadband antenna covering both frequency bands. Systems for the simultaneous evaluation of signal content in the frequency bands L1 and L2 place particularly high demands on the antennas. And this with little available installation space, as is always the case, especially in vehicle construction. The use of separate antennas in close proximity to one another for the two frequency bands entails the problem of mutual electromagnetic coupling with the effect of influencing the directional diagrams and the polarization purity and, in particular, cross-polarization. Due to the signals from the positioning satellites arriving at low elevation angles, even with sufficient gain in the desired, mostly right-handed circular polarization direction (RHCP), the suppression of the opposite polarization direction - the cross polarization (LHCP) - is of crucial importance with regard to correct positioning results. The accuracy of the positioning result is therefore particularly influenced by the ratio of the desired polarization direction to the cross polarization of the satellite receiving antenna, i.e. the cross polarization distance. On the other hand, it is technically difficult to create a satellite navigation antenna that covers both frequency bands with a bandwidth of approx. 360 MHz at a center frequency of around 1385 MHz and still meets the sometimes strict requirements for the cross polarization distance and the antenna gain. Satellite receiving antennas for satellite navigation are designed in particular for installation on horizontal surfaces of the electrically conductive vehicle body. In terms of the antenna properties, the essentially horizontal vehicle roof acts as a conductive base surface.

Satellite reception antennas with a small construction volume are particularly suitable for use on vehicles. State-of-the-art antennas of this type are known as patch antennas. However, these are less efficient in terms of reception at low elevation angles and more complex to set up. This disadvantage is partly overcome by ring line antennas, as used for example in the EP 2 296 227 A2 However, even for such antennas it is desirable to improve the cross-polarization distance over the full bandwidth of the frequency bands L1, L2 or L5 described above.

From the US 2014/0028512 A1 A loop antenna for vertically polarized signals is known, in which reactance circuits are equipped with series or parallel capacitances and inductances. In the GB 1 105 354 A An antenna arrangement for several frequency bands with two concentric ring line radiators is disclosed.

The present invention is therefore associated with the task of specifying an antenna for receiving circularly polarized satellite radio signals for satellite navigation, which has sufficient gain with a small construction volume and a high cross-polarization distance over a large frequency range and is therefore particularly precise for acquisition Location results in a vehicle is suitable.

This object is solved by the features of claim 1.

The particular advantage of the invention is that the strict requirement of the frequency bandwidth of the cross-polarization distance in conjunction with a small construction volume of the antenna, as exists for current and especially for future automobiles, can be met.

An antenna according to the invention also has the advantage that it can be manufactured particularly cost-effectively and is therefore particularly suitable for series production and use in the series production of vehicles.

The inventive idea here is, among other things, to realize the antenna with a shortened ring line and still meet the strict requirements regarding the cross-polarization distance in a wide frequency range This is achieved, among other things, by the fact that the resonance of the ring line at the frequency f0 is produced by the vertical resonance radiators via their reactance circuits with capacitive reactance X without involving the arrangement for exciting the antenna.

The ring line radiator is then in natural resonance at the resonance frequency f0 if the excitation is not included in the resonance formation is. The antenna impedance 37 present at this frequency is a high-ohm real resonance resistance and is often between 300 ohms and 500 ohms. This means that - due to the requirement according to the invention of a real input resistance 43 of the distribution and phase shifter network 8 at the gates T2 and T3 - the natural resonance of the ring line radiator 2 is not influenced by the excitation 3. This ensures that no resonance-forming currents of the ring line radiator 2 flow via the gates T2 and T3 of the distribution and phase shifter network 8. The separate adjustment of the resonance of the ring line radiator 2 and the distribution and phase shifter network 8 with respect to the real input resistance 43 of the gates T2 and T3 to the resonance frequency f0 leads to the large frequency bandwidth according to the invention for suppressing the cross polarization of the antenna 1.

The inventive requirement of a real input resistance of the distribution and phase shift network 8 at the gates T2 and T3 - through which the natural resonance of the ring line radiator 2 is not influenced by the excitation 3 - can naturally only be met at one frequency, i.e. exactly for the resonance frequency f0 and only approximately in the frequency environment of the resonance frequency f0. Compliance with this requirement at or close to the resonance frequency f0 enables the achievement of the largest possible bandwidth in the frequency environment of the resonance frequency f0.

According to the invention, the large frequency bandwidth of the cross-polarization distance can be achieved in the vicinity of the resonance frequency of the ring line radiator if the capacitive and inductive reactive power of the ring line radiator are balanced without a contribution from the device to excitation of the ring line radiator. The special frequency bandwidth of the cross-polarization distance around the resonance frequency f0 is achieved according to the invention in that the two gates T2, T3 of the distribution and phase shifter network with a corresponding phase difference ΔΦ represent the two vertical resonance excitation radiators in the - representing this phase difference of a current wave on the ring line of the ring line radiator Way to ensure that no reactive power exchange takes place via these gates in the event of resonance. This is due to the requirement for the real input resistance of the gates T2, T3 of the distribution and phase shifter network reached. This shows that the achievable frequency bandwidth of the cross-polarization distance is particularly large if this real input resistance of the gates corresponds approximately to the large resonance resistance of the ring line radiator at the frequency f0 of up to 500 ohms.

Advantageous embodiments of the invention are described below.

Two of the vertical resonance radiators (4a, 4b) can be assigned to the distribution and phase shifter network (8) as vertical resonance excitation radiators (10a, 10b) in such a way that each of the gates T2, T3 of the distribution and phase shifter network (8) is coupled to its ring line coupling point (7a, 7b) via one of the two vertical resonance excitation radiators (10a, 10b) and the other vertical resonance radiators (4c, 4d,..) act as passive vertical resonance radiators (9a, 9b, etc.).

Each of the gates T2, T3 can be connected directly via one of the resonance excitation radiators 10a, 10b to its ring line coupling point 7 and the input resistance of the gates T2, T3 can be high-impedance real in such a way that there is resistance matching between this and the high-impedance resonance resistance of the ring line radiator 2.

Due to the effect of the vertical resonance radiators 4a, 4b, 4c loaded with the reactance circuit 13 with capacitive reactance .

The reactance circuit 13 with which each of the passive vertical resonance radiators 9, 9a, 9b, and also the resonance excitation radiator 10a, 10b is coupled to the base surface 6, can each be formed by a capacitance.

The reactance circuit 13, with which the vertical resonance excitation radiators 10a, 10b are coupled to the base area 6, can each be used as a series circuit can be formed from a first reactance circuit 20a, 20b and a second reactance circuit 21a, 21b with a connection point 19 connecting them, to which one of the gates T2, T3 is connected.

The reactance circuit 13 with which the vertical resonance excitation radiators 10a, 10b are coupled to the base area 6 and the first reactance circuit 20a, 20b of each vertical resonance excitation radiator 10a, 10b can be formed by a first capacitance 22a, 22b and the second reactance circuit 21a, 21b by a second capacitance 23a, 23b and the size and the ratio of the first and the second capacitance can each be selected in such a way that resistance matching is provided between the high-ohmic resonance resistance of the ring line radiator 2 and a real input resistance of the gates T2, T3 at a lower impedance level.

The reactance circuits 13c, etc., with which the passive vertical resonance radiators 9c, etc. are coupled to the electrically conductive base area 6, can each be formed from the series connection of a capacitance 28 and a parallel oscillation circuit - consisting of a parallel capacitance 45 and a parallel inductance 46 and the first reactance circuit 20a, 20b of the vertical resonance excitation radiators 10a, 10b can each be formed as a first capacitance 22a, 22b and their second reactance circuit 21a, 21b can each be formed as a parallel oscillation circuit - consisting of a second capacitance 23a, 23b and a parallel inductance 46a , 46b and the resonance frequency of all parallel oscillation circuits can be selected in the frequency range between the upper frequency band L1 and the lower frequency band L2 in such a way that the reactance of these resonance circuits is capacitive in the frequency band L1 and inductive in the frequency band L2.

However, it may be that between the capacitance 28 and the parallel resonant circuit comprising the parallel capacitance 45 and the parallel inductance 46 of the reactance circuits 13c, etc. of the passive vertical resonance radiators 9c, etc., a further parallel resonance circuit 44, etc. is connected in series and the first capacitance 22a, 22b in the first reactance circuit 20a, 20b of the vertical resonance excitation radiators 10a, 10b is also followed by a further parallel resonance circuit 44a, 44b upstream of the vertical resonance excitation radiators 10a, 10b towards the connection point 19a, 19b and the parallel oscillation circuits in the first reactance circuit 20a, 20b and the second reactance circuit 21a, 21b of the vertical resonance excitation radiators 10a, 10b are tuned in such a way that their reactance Xa, Xb is capacitive in both frequency bands L1, L2 and their relationship to one another is in accordance with the optimal measure topt for adaptation and the reactance circuits 13c, etc. of the passive vertical resonance radiators 9c, etc. and the vertical resonance excitation radiators 10a, 10b are matched to one another with regard to their frequency behavior.

The first capacitance 22a, 22b of the first reactance circuit 20a, 20b of the vertical resonance excitation radiators 10a, 10b as well as the capacitances 28 in the reactance circuits 13c, etc. of the passive vertical resonance radiators 9c, etc. can be formed in such a way that all vertical resonance radiators 4a, 4b, etc. are formed at their lower end into individually designed flat capacitance electrodes 32a, 32b, 32c, 32d, that the capacitances 22a, 22b, 28 are connected to the electrically conductive base surface 6 by interposing a dielectric plate 33 between the flat capacitance electrodes 32a, 32b, 32c, 32d and the electrically conductive base surface 6 designed as an electrically conductive coated circuit board 35 for coupling the passive vertical resonance radiators 9c, 9d, etc. to the electrically conductive Base area 6 can be designed, and for the capacitive coupling of the vertical resonance excitation radiators 10a, 10b on the electrically conductive base area 6, a flat counter electrode 34 insulated from the conductive layer of the coated circuit board 35 can be designed for connecting the second reactance circuit 21b.

The vertical radiators can be formed into individually designed flat capacitance electrodes 32a, 32b, 32c, 32d for capacitive coupling at their lower end, but for coupling the passive vertical resonance radiators 9, 9a, 9b, a flat counter electrode 34 insulated from the conductive layer of the coated circuit board 35 can be formed for connecting the reactance circuit 13c, 13d formed from concentrated reactive elements.

The conductive structure, consisting of the ring line 14 and the vertical resonance radiators 4, 4a-d connected to it, can be fixed by a dielectric support structure 36 in such a way that the dielectric plate 33 is realized in the form of an air gap.

There can be a total of three vertical resonance radiators 4a, 4b, etc. with azimuthally equally distributed ring line coupling points 7, two of which are excited as vertical resonance excitation radiators 10a, 10b via the gates T2, T3 of a distribution and phase shifter network 8, the phase angle of which is ΔΦ = 120 degrees.

There can be a plurality of N>4 vertical resonance radiators 4, 4a, 4b, 4c,... with coupling points 7 evenly distributed over the extended length L of the ring line 14, so that N ring line sections 30a, 30b, 30c, 30d are formed are and between the coupling points 7 of the two vertical resonance excitation radiators 10a, 10b,... n ring line coupling points are selected and the phase angle ΔΦ = (n + 1) * 2π / N < 180 ° is selected.

The ring line can be designed square. There can be four vertical resonance radiators 4a, 4b, 4c, 4d with ring line sections 30a, 30b of equal azimuthal length, of which in particular two azimuthally successive as vertical resonance excitation radiators 10a, 10b,... via the gates T2, T3 of a distribution - and phase shifter network 8 are excited, the phase angle ΔΦ = 90 degrees.

However, the ring line can be designed rectangularly with four ring line coupling points 7, 7a,..7d at the corners and with different lengths of the ring line sections 30a, 30b and with different ring conductor widths 15a,15b, whereby the same characteristic impedance ZL and the same effective electrical length are set for the ring line sections 30a, 30b by selecting the ring conductor widths 15a,15b accordingly.

A distribution and phase shifter network 8 for a phase shift by the phase angle ΔΦ = 90° can contain a hybrid ring 38 which, in addition to the two gates T2 and T3, has a fourth gate T4 terminated with an ohmic terminating resistor 40 and which is decoupled from the gate T1 when the gates T2 and T3 are terminated with the correct characteristic impedance.

The distribution and phase shifter network 8 can contain a Wilkinson divider 39 with an ohmic balancing resistor 41 for a phase shift by the phase angle ΔΦ = 90°, wherein the input at gate 1 and one of the two output branches is followed by a phase shifter 17 with the phase angle ΔΦ = 90° towards the gate T3 and the gate T2 is formed at the other of the two output branches.

There can be a first and a second distribution and phase shifter network 8 with phase angle ΔΦ = 90 ° and four vertical radiators designed as excitation radiators, azimuthally equally distributed on the circumference of the ring conductor, the first distribution and phase shifter network 8 with its gates T2, T3 a first pair of vertical resonance excitation radiators 10a, 10b, and the second distribution and phase shifter network 8 with its gates T2a, T3a is coupled to a second pair of vertical resonance excitation radiators 10c, 10d opposite the first pair and a further distribution - and phase shifter network 8 with ΔΦ = 180 ° is present, which is connected with its gates T2c, T3c to the input gates T1, T1 in order to excite the rotating line wave in such a way that the rotating wave is established on the ring line.

It can contain the lossless adaptation circuit 18 for transforming the low-impedance impedance level Z0 into the high-impedance real input resistance in both output branches of the distribution and phase shift network 8, each essentially through a lambda/4 transformation line 12a, 28b and a series circuit comprising an inductance 27a, 27b and a capacitor 28a, 28b for fine adjustment of the high-impedance real input resistance of the gates T2 and T3.

All vertical resonance radiators (4a, 4b, 4c, etc.) can be distributed approximately equally along the ring line (14), so that none of the distances between adjacent ring line coupling points (7a, 7b, 7c) on the circumference of the ring line radiator ( 2) is less than half the distance with an equidistant distribution over the extended length of the ring line (14).

Due to the effect of the vertical resonance radiators (4, 4a, 4b, 4c) loaded with the reactance circuit (13, 13a,...13c,...) with capacitive reactance X, the extended length L of the ring line (14) of the ring line radiator (2) in resonance can be shortened from approximately the free space wavelength λ to approximately one third of the free space wavelength λ.

The vertical radiators (4a,..4d) can be formed at their lower end into individually designed flat capacitance electrodes (32a, 32b, 32c, 32d) for capacitive coupling, but for coupling the passive vertical resonance radiators (9c, 9d) in each case a flat counter electrode (34) insulated from the conductive layer of the coated circuit board (35) is formed for connecting the parallel resonance circuit formed from the parallel capacitance (45a, 45b) and the parallel inductance (46a, 46b) to the ground connection point (11). ( Fig. 13a , b , 14 , Fig. 5 )

The conductive structure, consisting of the ring line (14) and the vertical radiators (4, 4a-d) connected to it, can be fixed by a dielectric support structure (36) in such a way that the dielectric plate (33) is realized in the form of an air gap. ( Fig. 13a , b , 14 )

A plurality of N>4 vertical resonance radiators (4a, 4b, 4c,...) with coupling points (7) evenly distributed over the extended length L of the ring line (14) can be present, so that N ring line sections (30a,..30d) are formed and n ring line coupling points are selected between the coupling points (7a, 7b) of the two vertical resonance excitation radiators (10a, 10b) and the Phase angle ΔΦ = (n+1)*2π/N < 180° is selected. ( Fig.7 )

The ring line (14) can be rectangular in shape with four ring line coupling points (7a,..7d) at the corners and with different lengths of the ring line sections (30a, 30b) and with different ring line widths (15a, 15b), whereby correspondingly Selection of the ring conductor widths (15a, 15b) for the ring conductor sections (30a, 30b) each have the same characteristic impedance ZL and the same effective electrical length. ( Fig.9 , 10a, 10b, 10c )

The distribution and phase shifter network (8) for a phase shift by the phase angle ΔΦ = 90 ° can contain a Wilkinson divider (39) with an ohmic balancing resistor (41), with the input at gate 1 and one of the two output branches of the Wilkinson divider (39) a phase rotating element (17) with the phase angle ΔΦ = 90 ° is connected downstream of the gate T3 and is formed on the other of the two output branches gate T2. ( Fig. 11c )

All vertical resonance radiators (4a, 4b, 4c, etc.) can be distributed approximately equally along the ring line (14), so that none of the distances between adjacent ring line coupling points (7a, 7b, 7c) on the circumference of the ring line radiator (2) is smaller than half the distance with equidistant distribution over the extended length of the ring line (14).

There can be a first and a second distribution and phase shifter network (8) with phase angle ΔΦ = 90° and four vertical radiators designed as excitation radiators, equally azimuthally distributed on the circumference of the ring line (14), the first distribution and phase shifter network (8) is coupled with its gates T2, T3 to a first pair of resonance exciter radiators 10a, 10b and the second distribution and phase shifter network (8) is coupled with its gates T2a, T3a to a second pair of exciter radiators 10c, 10d opposite the first pair and a further distribution and phase shifter network (8) with ΔΦ = 180 ° is present, which is connected with its gates T2c, T3c to the gates T1, T1a in order to excite the rotating line wave in such a way that the rotating wave is on the ring line sets.

According to a further aspect, the invention relates to an antenna arrangement in which a plurality of antennas of the type described above are provided, each assigned to a frequency band, the ring line radiators (2) of which are arranged concentrically to one another in such a way that the ring line radiator (2) for the frequency band with the highest frequency is arranged innermost and each further ring line radiator (2) surrounds a ring line radiator for the next higher frequency band.

• Fig. 1:
  Antenne 1 nach der Erfindung, bestehend im Beispiel aus dem Ringleitungsstrahler 2 mit dem Ringleiter 14 in der Höhe h < 0,15λ über der leitenden Grundfläche 6 mit N = 3 vertikalen Resonanzstrahlern 4a, 4b, 4c an azimutal gleich verteilten Ringleitungs-Koppelpunkten 7a, 7b, 7c und der elektromagnetischen Erregung 3 durch das Verteil- und Phasenschiebernetzwerk 8. Die vertikalen Resonanzstrahler 4a, 4b, 4c sind jeweils über eine kapazitiv wirkende Blindwiderstandsschaltung 13a,13b,13c über den Masse-Anschlusspunkt 11 mit der leitenden Grundfläche 6 verkoppelt. Die elektromagnetische Erregung 3 erfolgt über das Verteil- und Phasenschiebernetzwerk 8, welches eingangsseitig mit seinem Tor T1 mit dem Antennenanschluss 5 verbunden ist und dessen Tore T2 und T3 mit zwei von den vertikalen Resonanzstrahlern 4a, 4b, 4c als vertikale Resonanz-Erregungsstrahler 10a und 10b in der Weise verbunden sind, dass die zugehörigen Ringleitungs-Koppelpunkte 7a, 7b direkt mit den im Verteilnetzwerk 16 und im Phasendrehglied 17 bewirkten Ausgangssignalen an den Toren T2 und T3 mit Phasenwinkelunterschied ΔΦ = 360°/N = 120° angesteuert sind, sodass sich auf dem Ringleiter 14 die laufende Leitungswelle einstellt. Hierbei ist erfindungsgemäß vorausgesetzt, dass die Tore T2 und T3 jeweils einen reellen Eingangswiderstand 43 besitzen, sodass die Resonanz des Ringleitungsstrahlers 2 durch Anschluss der elektromagnetischen Erregung 3 nicht beeinflusst ist. Zur Impedanzanpassung zwischen dem Verteil- und Phasenschiebernetzwerk 8 an den Toren T2, T3 und der hochohmigen Antennenimpedanz 37 bei der Resonanzfrequenz f0 des Ringleitungsstrahlers 2 (sh. Fig. 2b) an den Ringleitung-Koppelpunkten 7a und 7b ist das Verteil- und Phasenschiebernetzwerk 8 hochohmig gestaltet und befindet sich auf einem Impedanzniveau von etwa 300 - 500 Ohm.
• Fig. 2: Figur a) zeigt den Ringleitungsstrahler 2 nach der Erfindung mit drei azimutal gleich verteilten vertikalen Resonanzstrahlern 4a - 4c als Resonanzstruktur. Die Erregung 3 ist nicht dargestellt. Die Blindwiderstandsschaltungen 13 a- 13c sind durch die Kapazitäten 28a, 28b, 28c realisiert. Bei Rotationssymmetrie der Anordnung sind alle Kapazitäten gleich groß. Bei Abstimmung des Ringleitungsstrahlers 2 auf eine Resonanzfrequenz- zum Beispiel f0= 1392 MHz - stellt sich an den vertikalen Resonanzstrahlern 4a- 4c jeweils an der mit 42 bezeichneten Messstrecke der in Figur 2b) dargestellte Verlauf der Antennenimpedanz 37 mit ihrem durch die Strahlung bedingten reellen Resonanzwiderstand von ca. 340 Ohm ein. Bei Anschluss der Erregung an die Resonanzstrahler 4a und 4b - wie in Figur 1 - wird das Resonanzverhalten der Struktur in der Frequenz nicht verändert. Im Beispiel der 3 azimutal gleich verteilten vertikalen Resonanzstrahler 4a, 4b, 4c beträgt der Phasendrehwinkel des Phasendrehglieds ΔΦ= 120°.
• Fig.3:
  Antenne 1 nach der Erfindung wie in Figur 1 jedoch mit einem Verteilnetzwerk 16 und einem Phasendrehglied 17 auf dem Impedanzniveau Z0 üblicher koaxialer Leitungen (Z0 = 50 Ohm). Zur Anpassung an das hochohmige Impedanzniveau der Antennenimpedanz 37 (sh. Figur 2b) sind dem Phasendrehglied 17 und einem Arm des Verteilnetzwerks 16 jeweils ein Anpassnetzwerk 18 nachgeschaltet. Der Eingangswiderstand 43 an den Toren T2 und T3 ist - ebenso wie in Figur 1 - bei der Resonanzfrequenz des Ringleitungsstrahlers 2 hochohmig und reell.
• Fig.3a:
  Verteil- und Phasenschiebernetzwerk 8 wie in Figur 3 mit beispielhafter Ausführung der Anpassnetzwerke 18 an den beiden Ausgängen. Die Anpassung bei Resonanz des Ringleitungsstrahlers 2 ist durch die λ/4-Transformationsleitungen 12a, 12b, durch welche die Transformation von dem niederohmigen Impedanzniveau Z0 zu dem hochohmigen Impedanzniveau der Antennenimpedanz 37 gegeben ist, bewirkt. Die Serienresonanzkreise, bestehend aus den Kapazitäten 28a, 28b und den Induktivitäten 12a und 12b ermöglichen eine Feinkorrektur der Anpassung über einen erweiterten Frequenzbereich.
• Fig.4:
  Antenne nach der Erfindung wie in Figur 3 jedoch mit besonderer Gestaltung der Blindwiderstandsschaltungen 13a und 13b in den beiden vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils unterteilt in eine erste Blindwiderstandsschaltung 20a, 20b und eine zweite Blindwiderstandsschaltung 21a, 21b mit einem dazwischen liegenden Verknüpfungspunkt 19a, 19b zum Anschluss der Tore T2 und T3 des Verteil-und Phasenschiebernetzwerks 8.
• Fig.5:
  Erfindungsgemäße Ausführungsformen der insgesamt kapazitiv wirkenden Blindwiderstandsschaltungen 13 zur Ankopplung der vertikalen Resonanzstrahler an die leitende Grundfläche 6 in Figur 4
  1. a) Blindwiderstandsschaltung 13a bzw. 13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b, unterteilt in die erste Blindwiderstandsschaltung 20a bzw. 20b und die über den Verknüpfungspunkt 19a bzw. 19b verbundene zweite Blindwiderstandsschaltung 21a bzw. 21b, jeweils realisiert durch eine erste Kapazität 22a bzw. 22b und entsprechend durch die zweite Kapazität 23a bzw.23b.
  2. b) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert durch eine Kapazität 28. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur a) beschaltet.
  3. c) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur a) jedoch mit einem Parallelresonanzkreis, bestehend aus der Parallelinduktivität 46a bzw. 46b und der Parallelkapazität 23a bzw.23b in der zweiten Blindwiderstandsschaltung 21a bzw. 21b zur Erweiterung der Frequenzbandbreite der Antenne.
  4. d) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert, wie unter Figur b) jedoch mit einem seriellen Parallelresonanzkreis bestehend aus der Parallelinduktivität 46 und der Parallelkapazität 45 in Serienschaltung zur Kapazität 28 zur Erweiterung der Frequenzbandbreite der Antenne in Entsprechung der Gestaltung der Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 4a bzw. 4b wie unter c). Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur c) beschaltet.
  5. e) Darstellung der Frequenzbänder L1 und L2 für die Satelliten-Navigation mit den Mittenfrequenzen fm1 und fm2 und den unteren und oberen Grenzfrequenzen fu1, fo1 bzw.fu2, fo2. Die Frequenz fm beschreibt die Mittenfrequenz zwischen fu1 und fo2.
  6. f) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur c) jedoch mit einem weiteren Parallelresonanzkreis 44a bzw. 44b in Serienschaltung zur ersten Kapazität 22a bzw. 22b hin zum Verknüpfungspunkt 19 zur gesonderten Optimierung der Schaltung für jeweils einen der beiden Frequenzbereiche L1 und L2 zur Gestaltung einer Zweibandantenne.
  7. g) Blindwiderstandsschaltung 13c des übrigen passiven vertikalen Resonanzstrahlers 4c, realisiert wie unter Figur d), jedoch mit einem weiteren Parallelresonanzkreis 44 in Serienschaltung zur Kapazität 28 in der Blindwiderstandsschaltung 13c. Die beiden aktiven Resonanz-Erregungsstrahler sind wie in Figur f) beschaltet zur Gestaltung einer Zweibandantenne.
  8. h) Blindwiderstandsschaltung 13a bzw.13b der beiden vertikalen aktiven Resonanz-Erregungsstrahler 10a bzw.10b wie unter Figur f) jedoch mit einer Zusatz-Parallelkapazität 47, gebildet durch die, die isolierte Gegenelektrode 34 überkragende Fläche der Kapazitätselektrode 32a, 32b in den Figuren 13c und 13d mit der elektrisch leitenden beschichteten Leiterplatte 35. Die Zusatz-Parallelkapazität 47 ermöglicht die Erweiterung des Bereichs der Impedanzanpassung jeweils am Verknüpfungspunkt 19a, 19b bei ebenso gegebener Einhaltung der Resonanzbedingung für den Ringleitungsstrahler 2. Die Resonanzbedingung und die Impedanzanpassung werden durch Abstimmung der Größen der Kapazitätselektroden 32a, 32b und der isolierten Gegenelektroden 34 aufeinander hergestellt.
• Fig.6:
  Anpassungsverhältnisse bei der Resonanzfrequenz f0 in Abhängigkeit vom Maß "t" für die Unterteilung in die erste Kapazität 22a, 22b und die zweite Kapazität 23a, 23b am Beispiel des Ringleitungsstrahlers 2 in Figur 4.
  1. a) Reflexionsfaktor an der Antennenanschlusstelle 5 als Eingangstor T1 eines verlustfreien Verteil- und Phasenschiebernetzwerks 8 der Antenne 1 in Abhängigkeit vom Teilungsmaß t . Anpassung ist mit topt = 0,13 erreicht.
  2. b) Transformationsfaktor bei der Resonanzfrequenz f0 zwischen dem Eingangswiderstand 43 mit Z0 = 50 Ohm des Phasendrehglieds 17 und dem Resonanzwiderstand bei f0 am Ringleitungsstrahler-Koppelpunkt 7 in Abhängigkeit vom Teilungsmaß t.
  3. c) Relative Leistung P/Pmax bei f0 in Abhängigkeit vom Teilungsmaß t.
• Fig.7:
  Antenne nach der Erfindung mit Widerstands-Anpassung durch kapazitive Unterteilung wie im Beispiel der Figur 4 jedoch mit insgesamt 6 azimutal um jeweils 60° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a .. 4f, von denen vier als vertikale passive Resonanzstrahler 9a..9d gestaltet sind. Die Erregung 3 erfolgt beispielhaft an den beiden um 120° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 120° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• Fig.8:
  Antenne nach der Erfindung wie in Figur 7, jedoch mit insgesamt 8 azimutal um jeweils 45° gegeneinander versetzt verteilten vertikalen Resonanzstrahlern 4a..4h. Die Erregung 3 erfolgt beispielhaft an den beiden um 90° azimutal gegeneinander versetzten und um den Differenzwinkel ΔΦ= 90° erregten vertikalen Resonanz-Erregungsstrahlern 10a, 10b.
• Fig.9:
  Antenne nach der Erfindung mit rechteckig ausgeführter Ringleitung 14 mit vier vertikalen Resonanzstrahler 4a bis 4d im Bereich der Ringleitungs-Ecken. Zwei der vertikalen Resonanzstrahler sind als vertikale Resonanz-Erregungsstrahler 10a, 10b und die beiden übrigen Strahler als passive vertikale Resonanzstrahler 9a, 9b ausgeführt. Impedanzanpassung an das niederohmig (z.B. Z= 50 Ohm) ausgeführte Verteil- und Phasenschiebernetzwerk 8 ist durch kapazitive Unterteilung, ähnlich wie in den Figuren 4, 7 und 8 erreicht. Die Ringleitung 14 ist vorzugsweise quadratisch gestaltet mit gleichen Ringleiterbreiten 15a, 15b und gleichen Ringleitungs-Abschnitten 30a, 30b. Sowohl die Ringleiterbreite 15a, 15b als auch die Ringleitungs-Abschnitte 30a, 30b können jedoch abgestimmt aufeinander in Grenzen unterschiedlich gewählt sein.
• Fig. 10:
  • Rechteckförmiger Ringleitungsstrahler 2 einer Antenne 1 nach der Erfindung ohne Darstellung der elektromagnetischen Erregung 3 zur Erläuterung der Resonanzstruktur des Ringleitungsstrahlers 2.
    1. a) geometrischer Aufbau des Ringleitungsstrahlers 2 mit großen unterschiedlich wählbaren Ringleiterbreiten 15a und 15b sowie den unterschiedlich wählbaren Ringleitungs-Abschnitten 30a und 30b. Die vertikalen Resonanzstrahler 4a - 4d sind mit ihren Ringleitungs-Koppelpunkten 7a - 7d im weiten Bereich der Ecken des rechteckförmigen Ringleiters 14 vorgesehen. Die unterbrochene Linie kennzeichnet etwa den Verlauf der Schwerlinie 24 der Stromdichteverteilung der Leitungswelle bei Erregung des Ringleitungsstrahlers 2. Aufgrund der Stromverdrängung verdichtet sich die Stromverteilung hin zum Rand des Ringleiters 14. Der für die Funktion bei der Erregung des Ringleitungsstrahlers 2 relevante Strom der Leitungswelle fließt demnach - gekennzeichnet durch die als unterbrochene Linie gezeichnete Schwerlinie der Stromverteilung 24 - auch bei großen Leiterbreiten 15a, 15b mehr zum Rand hin gedrängt. Dies gilt insbesondere für sehr große Ringleiterbreiten 15a, 15b bis hin zur vollkommenen Schließung der inneren Öffnung zum Zentrum hin, welches praktisch stromlos ist, wenn die Ringleitung 14 als eine geschlossene leitende Fläche realisiert ist.
    2. b) gemäß dem durch die Schwerpunktlinie der Stromdichteverteilung 24 repräsentierten Strom kann die Resonanzstruktur des Ringleitungsstrahlers 2 durch ein grob angenähertes Ersatzschaltbild allgemein wiedergegeben werden. Die einzelnen Ringleitungs-Abschnitte 30a -30d sind jeweils durch die induktive und kapazitive Wirkung des zugehörigen Abschnitts der Ringleitung 14 unter Einbeziehung der kapazitiven Wirkung der Blindwiderstandsschaltung 13 als konzentrierte induktive Elemente (Ln) und kapazitive Elemente (Cn) dargestellt. Jeder n-te Ringleitungsabschnitt 30a - 30b ist durch eine π-Struktur gemäß Figur c), bestehend aus einer Längsinduktivität 2*Ln und jeweils einer Querkapazität Cn an deren beiden Enden, dargestellt. Die Strahlungsdämpfung jedes horizontal orientierten Ringleitungsabschnitt 30a - 30b ist durch den Dämpfungsfaktor d der konzentrierten Induktivität einbezogen. Die Aneinanderreihung einander benachbarter Ringleitungs-Abschnitte 30a-30d erfolgt jeweils über einen gemeinsamen vertikalen Resonanzstrahler 4a..4d unter Zusammenziehung der Querkapazitäten Cn der benachbarten Ringleitungs-Abschnitte, wie in Figur c) dargestellt. Die geringfügige induktive Wirkung der vertikalen Resonanzstrahler ist bei dieser Grundsatzbetrachtung aufgrund der geringen Antennenhöhe h vernachlässigt. Erfindungsgemäß vorteilhafte Voraussetzungen für den Ringleitungsstrahler 2 sind für die Resonanzfrequenz f0=ωo/2π dann erreicht, wenn jeweils alle Wellenwiderstände $\mathit{ZLn}=\sqrt{\frac{\mathit{Ln}}{\mathit{Cn}}}$
       
       und jeweils alle $\sqrt{\mathit{Ln}\square \mathit{Cn}}=\raisebox{1ex}{$1$}\!\left/ \!\raisebox{-1ex}{$\mathit{\omega on}$}\right.$
       
       , welche die elektrische Länge des Ringleitungs-Abschnitts repräsentieren, für alle n gleich groß sind.
  • Bei rotationssymmetrischen Ringleitungsstrahlern 2 ist dies stets gegeben. Andernfalls kann diese Voraussetzung zum Beispiel bei einer rechteckförmigen Struktur des Ringleitungsstrahlers 2 durch individuelle Gestaltung der Ringleiterbreiten 15a-15d in den einzelnen Ringleitungs-Abschnitten 30a-30b erreicht werden.
• Fig.11:
  1. a) Gestaltung des Verteil- und Phasenschiebernetzwerks 8 - zum Beispiel wie in den Figuren 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Hybridring 3, durch welchen sowohl die Leistungsteilung als auch die Phasenverschiebung erfolgt. Bei Einspeisung am Tor T1 wird der Ringleitungsstrahler 2 über die Tore T2 und T3 mit dem Phasenunterschied von 90° erregt. Der Abschluss von Tor T4 mit dem ohmschen Abschlusswiderstand 40 bewirkt bei Abweichung der Frequenz von der Resonanzfrequenz f0 breitbandig die teilweise Absorption der Leistung der unerwünschten Polarisation bei Einspeisung am Tor T1.
  2. b) Idealisierte Streumatrix zur Beschreibung des allgemein bekannten grundsätzlichen Wellenverhaltens eines Hybridrings 38 in Figur a) an den Toren T1 bis T4. Die wichtige Entkopplung der Tore T1 und T4 voneinander ist durch die Streuparameter S14 = 0, S41 = 0 in der Matrix fett gekennzeichnet.
  3. c) Gestaltung des Verteil- und Phasenschiebernetzwerks 8 - zum Beispiel wie in den Figuren 7 und 8 - jedoch als auf die Resonanzfrequenz f0 abgestimmten Wilkinson-Teilers 16 mit nachgeschaltetem λ/4 Leitungs-Phasenschieber 17 an einem Tor T3* zur Erzeugung des Phasenunterschieds von 90°. Der ohmsche Symmetrierungswiderstand 40 absorbiert bei unsymmetrischer Belastung des Wilkinson-Teilers 16 bei Abweichung von der Resonanzfrequenz f0 teilweise die Ströme, welche die unerwünschte Polarisation auf dem Ringleitungsstrahler 2 hervorrufen.
• Fig.12:
  Antenne nach der Erfindung z. B. wie in Figur 9 mit rechteckig geformter Ringleitung 14. Die Kapazitäten 22a, 22b und 28 sind in der Weise gebildet, dass die vertikalen Resonanzstrahler 4a - 4d an ihrem unteren Ende zu individuell gestalteten flächigen Kapazitätselektroden 32a, 32b, 32c, 32d ausgeformt sind. Durch Zwischenlage zwischen diesen und die als elektrisch leitend beschichtete Leiterplatte ausgeführte elektrisch leitende Grundfläche 6 befindliche dielektrische Platte 33 sind die Kapazitäten 28 zur Ankopplung der passiven vertikalen Resonanzstrahler 4c, 4d an die elektrisch leitende Grundfläche 6 gestaltet. Zur kapazitiven Ankopplung der aktiven vertikalen Resonanzstrahler 10a, 10b an die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8 ist dieser Anschluss als jeweils eine von der leitenden Schicht isolierte, flächige Gegenelektrode 34 gestaltet. Die Gegenelektroden 34 können somit als Verknüpfungspunkte 19 der Antenne in Figur 9 ausgeführt sein und können als Anschlusspunkte für die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8, wie in Figur 13 angedeutet, dienen.
• Fig.13:
  Antenne 1 ähnlich wie in Figur 11, jedoch ist die dielektrische Wirkung der dielektrischen Platte 33 durch einen Luftspalt realisiert. Typische Abmessungen eines Ringleitungsstrahlers 2 für den Frequenzbereich L1 sind für quadratische Antennen ein Abmessung von 30mm bis 40mm und für die Höhe h = 8mm. Zur Widerstandsanpassung des vorzugsweise auf niedrigem Impedanzniveau (Z0) ausgeführten Verteil- und Phasenschiebernetzwerks 8 ist die kapazitive Unterteilung wie in Figur 9 vorgesehen.
  1. a) Die beispielhaft zugehörigen Blindwiderstandsschaltungen 13a, 13b und 13c sind in den Figuren 5a und 5b dargestellt. Die Kapazität 28 der Kapazitätselektrode 32c, 32d gegen die elektrisch leitende Grundfläche 6 bzw. die Kapazität 22a, 22b gegen die isolierte Gegenelektrode 34a, 34b beträgt jeweils ca. 0.3pF. Die Tore T2 und T3 des Verteil- und Phasenschiebernetzwerks 8 mit ihrem reellen Eingangswiderstand 43 sind jeweils an eine Gegenelektrode 34a, 34b als Verknüpfungspunkt 19a, 19b angeschlossen. Die kapazitive Unterteilung zur Impedanzanpassung ist jeweils durch die Kapazität 22a, 22b und der Kapazität 23a, 23b gegeben. Diese zwischen dem Verknüpfungspunkt 19a, 19b und dem Massepunkt 11 eingebrachte Kapazität 23a, 23b ist auf der Rückseite der beschichteten Leiterplatte 35 als SMD-Bauteil angebracht. Hierzu sind auf der Rückseite der Leiterplatte isolierte Pads 29 als Kontaktstützpunkte gestaltet, welche jeweils über eine Durchkontaktierung 26 mit der Gegenelektrode 34 einerseits und andererseits z.B. über die Kapazität 23a, 23b (sh. Fig. 4 u. 5a) mit Masse verbunden sind.
  2. b) die Blindwiderstandsschaltungen 13a, 13b der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur a) gemäß Figur 5a bzw. wie in 5c gestaltet. Bei Gestaltung nach Figur 5c ist der auf der Rückseite der beschichteten Leiterplatte 35 eingebrachten Kapazität 23a, 23b eine Induktivität 46a, 46b als SMD-Bauteil parallel geschaltet. Im Gegensatz zur Figur a) sind zur Bildung der Blindwiderstandsschaltungen der beiden passiven Resonanzstrahler 9a, 9b den Kapazitätselektroden 32c, 32d auf der beschichteten Leiterplatte 35 isolierte Gegenelektroden 34 gegenübergestellt. Ausgehend von der Gegenelektrode 34 als Kontaktstützpunkt sind auf der Rückseite der Leiterplatte 35 die Parallelinduktivität 46 und die Parallelkapazität 45 zum Massepunkt 11 auf der Leiterplatte 35 geschaltet, sodass die Blindwiderstandsschaltung 13 in Figur 5d realisiert ist.
     Bei Ausgestaltung aller Blindwiderstandsschaltungen 13 aller vertikalen Resonanzstrahler mit jeweils einer Kapazitätselektrode 32 und einer gegenüberliegenden isolierten Gegenelektrode 34 können alle Schaltungen in den Figuren 5c und 5f für die Resonanz-Erregungsstrahler 10a, 10b durch Einsatz von SMD-Bauteilen auf der Rückseite der Leiterplatte 35 realisiert werden. Auf gleiche Weise können die Blindwiderstandsschaltungen für die passiven Resonanzstrahler 9a, 9b durch Beschaltung der isolierten Gegenelektroden 34 mit SMD-Bauteilen auf der Rückseite der Leiterplatte 35 gemäß den in den Figuren 5d und 5g angegebenen Schaltungen realisiert werden.
  3. c) Wie unter Figur 5h beschrieben dient die Zusatz-Parallelkapazität 47 zur freien Gestaltung der Impedanzanpassung bei Beibehaltung der Resonanzeigenschaften des Ringleitungsstrahlers 2. Die Figur zeigt die großflächige Überdeckung der isolierten Gegenelektrode 34a, 34b mit der Kapazitätselektrode 32a, 32b der Resonanz-Erregungsstrahler 10a, 10b. Der in der vertikalen Projektion der Kapazitätselektrode 32a, 32b gegenüber der isolierten Gegenelektrode 34 bestehende flächige Überstand bildet mit der elektrisch leitenden Schicht der beschichteten Leiterplatte 35 die Zusatz-Parallelkapazität 47. Die Blindwiderstandsschaltung der beiden passiven Resonanzstrahler 9a, 9b sind wie in Figur a) gestaltet.
  4. d) die Blindwiderstandsschaltungen 13 der beiden Resonanz-Erregungsstrahler 10a, 10b sind wie in Figur c) mit einer Zusatz-Parallelkapazität 47 gestaltet. Die beispielhaft möglichen Blindwiderstandsschaltungen 13 der beiden passiven Resonanzstrahler 9a, 9b können wie in Figur b) beschrieben, gestaltet werden.
• Fig.14:
  Im Bild ist die Oberseite der Leiterplatte 35 einer Antenne 1 nach der Erfindung dargestellt, auf welche der elektrische Ringleitungsstrahler 2 aufgesetzt wird. Für die breitbandige Gestaltung einer Antenne 1 nach der Erfindung - welche zum Beispiel beide Frequenzbereiche L1 und L2 mit einer zwischen den beiden Frequenzbereichen liegenden Mittenfrequenz fm (sh. Figur 5e) - ist die Blindwiderstandsschaltung 13, wie in den Figuren 5c bzw. 5d gestaltet. Hierfür ist für die Blindwiderstandsschaltung in den beiden aktiven vertikalen Resonanz-Erregungsstrahlern 10a, 10b jeweils die Blindwiderstandsschaltung 13a, 13b in Figur 5c und für die beiden passiven vertikalen Resonanzstrahler 9a, 9b die Blindwiderstandsschaltung 13c in Figur 5d vorgesehen. Dies ist bei dem Beispiel im Figur 13a dadurch erreicht, dass für alle Kapazitätselektroden 32 jeweils eine Gegenelektrode 34 vorhanden ist und, dass der durch die Kapazitätselektroden 32 bewirkten Kapazität 22a, 22b bzw. 28 an allen vertikalen Resonanzstrahlern 4a - 4d eine Parallelschaltung aus einer Parallelkapazität 23a, 23b bzw. 45 und einer Parallelinduktivität 46a, 46b, bzw. 46 - dargestellt als SMD- Bauteile - zwischen der Gegenelektrode 34 und der elektrisch leitenden Grundfläche 6 in Serie geschaltet ist. Figur 13a zeigt die Oberseite der Leiterplatte 35 mit Durchkontaktierungen 26 auf den Gegenelektroden 34 unter den Kapazitätselektroden 32. In Figur 14b ist die Unterseite der Leiterplatte 35 mit den Pads 29 dargestellt, welche mit den isolierten Gegenelektroden 34 über die Durchkontaktierungen 26 verbunden sind,. Die Blindelemente sind als SMD-Bauelemente gemäß den Figuren 5c und 5d angebracht. Zusätzlich sind die beiden Verknüpfungspunkte 19 mit der jeweils daran angeschlossenen Serienschaltung aus der Kapazität 28a, 28b und der Induktivität 27a, 27b als Teile der in Figur 3a enthaltenen Anpassungsschaltung 18 des Verteil-und Phasenschiebernetzwerks 8 zur Bildung der Tore T2 und T3 dargestellt.
• Fig. 15
  Zeigt eine Antenne nach der Erfindung mit vier azimutal am Umfang der Ringleitung 14 gleich verteilten Resonanz-Erregungsstrahlern 10a bis 10d. Alle Strahler werden entsprechend einer umlaufenden Welle mit jeweils 90°-Phasenunterschied zwischen benachbarten Strahlern erregt. Hierzu sind ein erstes und ein zweites Verteil- und Phasenschiebernetzwerk (8) mit Phasenwinkel ΔΦ = 90° vorhanden, wobei das erste Verteil- und Phasenschiebernetzwerk (8) mit seinen Toren T2, T3 mit einem ersten Paar der Resonanz-Erregerstrahler 10a, 10b verkoppelt ist und das zweite Verteil-und Phasenschiebernetzwerk (8) mit seinen Toren T2a, T3a mit einem dem ersten Paar gegenüberliegendem zweiten Paar der Erregerstrahler 10c, 10d verkoppelt ist. Zur Anregung einer laufenden Welle werden einander gegenüberliegende Resonanz-Erregungsstrahler 10a, 10c bzw. 10b, 10d jeweils mit einem Phasenunterschied von ΔΦ = 180° erregt. Hierzu ist ein weiteres Verteil- und Phasenschiebernetzwerk 31, 39 mit einer Phasenverschiebung von ΔΦ = 180° vorhanden, welches mit seinem Tor T2c mit dem Tor T1 des ersten Verteil- und Phasenschiebernetzwerks 8, 36 und mit seinem Tor T3c mit dem Tor T1b des weiteren Verteil- und Phasenschiebernetzwerks 8, 38b verbunden ist. Auch hier kommt erfindungsgemäß vorzugsweise jeweils ein Hybridring 38 zum Einsatz. Das 180° Phasennetzwerk 31 kann beispielhaft als Wilkinson-Teiler 39 mit einer λ/2-langen Verzögerungsleitung 17 (sh. Figur 11c) eingesetzt werden. Alle Anschlüsse sind in der Weise gewählt, dass sich auf der Ringleitung die leitende Welle in der gewünschten Umlaufrichtung einstellt. Mit dieser Anordnung ist der besondere Vorteil der azimutal symmetrischen Anregung verbunden, sodass der Abgleich der Anordnung besonders problemfrei erfolgen kann. Als Einschränkung ist jedoch der erhöhte Aufwand anzumerken.

The invention is explained in more detail below using exemplary embodiments. The accompanying figures show in detail:

• Fig.1 :
  Antenna 1 according to the invention, consisting in the example of the ring line radiator 2 with the ring conductor 14 at the height h < 0.15λ above the conductive base surface 6 with N = 3 vertical resonance radiators 4a, 4b, 4c at azimuthally equally distributed ring line coupling points 7a, 7b, 7c and the electromagnetic excitation 3 through the distribution and phase shifter network 8. The vertical resonance radiators 4a, 4b, 4c are each coupled to the conductive base surface 6 via a capacitive reactance circuit 13a, 13b, 13c via the ground connection point 11. The electromagnetic excitation 3 is carried out via the distribution and phase shifter network 8, which is connected on the input side with its gate T1 to the antenna connection 5 and whose gates T2 and T3 are connected to two of the vertical resonance radiators 4a, 4b, 4c as vertical resonance excitation radiators 10a and 10b in such a way that the associated ring line coupling points 7a, 7b are directly controlled with the output signals generated in the distribution network 16 and in the phase shifter 17 at the gates T2 and T3 with a phase angle difference ΔΦ = 360°/N = 120°, so that the current line wave is set on the ring conductor 14. In this case, it is assumed according to the invention that the gates T2 and T3 each have a real input resistance 43, so that the resonance of the ring line radiator 2 is not influenced by the connection of the electromagnetic excitation 3. For impedance matching between the Distribution and phase shifting network 8 at the gates T2, T3 and the high-resistance antenna impedance 37 at the resonance frequency f0 of the ring line radiator 2 (see Fig. 2b ) at the ring line coupling points 7a and 7b, the distribution and phase shifting network 8 is designed to be high-impedance and is at an impedance level of approximately 300 - 500 ohms.
• Fig.2 : Figure a) shows the ring line radiator 2 according to the invention with three azimuthally equally distributed vertical resonance radiators 4a - 4c as a resonance structure. The excitation 3 is not shown. The reactance circuits 13 a - 13 c are realized by the capacitors 28a, 28b, 28c. If the arrangement is rotationally symmetrical, all capacitors are of the same size. When the ring line radiator 2 is tuned to a resonance frequency - for example f0 = 1392 MHz - the vertical resonance radiators 4a - 4c each have the measurement section designated 42. Figure 2b ) with its real resonance resistance of about 340 Ohms caused by the radiation. When the excitation is connected to the resonance radiators 4a and 4b - as in Figure 1 - the resonance behavior of the structure is not changed in frequency. In the example of the three azimuthally equally distributed vertical resonance radiators 4a, 4b, 4c, the phase rotation angle of the phase shifter is ΔΦ= 120°.
• Fig.3 :
  Antenna 1 according to the invention as in Figure 1 but with a distribution network 16 and a phase shifter 17 at the impedance level Z0 of usual coaxial lines (Z0 = 50 Ohm). To adapt to the high-impedance level of the antenna impedance 37 (see Figure 2b ), the phase shifter 17 and an arm of the distribution network 16 are each followed by a matching network 18. The input resistance 43 at the gates T2 and T3 is - as in Figure 1 - at the resonance frequency of the ring line radiator 2, high-impedance and real.
• Fig.3a :
  Distribution and phase shifting network 8 as in Figure 3 with exemplary design of the matching networks 18 at the two outputs. The matching at resonance of the ring line radiator 2 is effected by the λ/4 transformation lines 12a, 12b, through which the transformation from the low-resistance impedance level Z0 to the high-resistance impedance level of the antenna impedance 37 is provided. The series resonance circuits, consisting of the capacitances 28a, 28b and the inductances 12a and 12b, enable fine correction of the matching over an extended frequency range.
• Fig.4 :
  Antenna according to the invention as in Figure 3 but with a special design of the reactance circuits 13a and 13b in the two vertical resonance excitation radiators 10a, 10b, each divided into a first reactance circuit 20a, 20b and a second reactance circuit 21a, 21b with a connection point 19a, 19b in between for connecting the gates T2 and T3 of the distribution and phase shifter network 8.
• Fig.5 :
  Inventive embodiments of the overall capacitive reactance circuits 13 for coupling the vertical resonance radiators to the conductive base surface 6 in Figure 4
  1. a) Reactance circuit 13a or 13b of the two vertical active resonance excitation radiators 10a or 10b, divided into the first reactance circuit 20a or 20b and the second reactance circuit 21a or 21b connected via the connection point 19a or 19b, each realized by a first capacitance 22a or 22b and correspondingly by the second capacitance 23a or 23b.
  2. b) Reactance circuit 13c of the remaining passive vertical resonance radiator 4c, realized by a capacitor 28. The two active resonance excitation radiators are connected as in Figure a).
  3. c) Reactance circuit 13a or 13b of the two vertical active resonance excitation radiators 10a or 10b as in figure a) but with a parallel resonance circuit consisting of the parallel inductance 46a or 46b and the parallel capacitance 23a or 23b in the second reactance circuit 21a or 21b to extend the frequency bandwidth of the antenna.
  4. d) Reactance circuit 13c of the remaining passive vertical resonance radiator 4c, implemented as in Figure b), but with a serial parallel resonance circuit consisting of the parallel inductance 46 and the parallel capacitance 45 in series with the capacitance 28 to extend the frequency bandwidth of the antenna in accordance with the design of the reactance circuit 13a or 13b of the two vertical active resonance excitation radiators 4a or 4b as in c). The two active resonance excitation radiators are wired as in Figure c).
  5. e) Representation of the frequency bands L1 and L2 for satellite navigation with the center frequencies fm1 and fm2 and the lower and upper limit frequencies fu1, fo1 and fu2, fo2 respectively. The frequency fm describes the center frequency between fu1 and fo2.
  6. f) Reactance circuit 13a or 13b of the two vertical active resonance excitation radiators 10a or 10b as in figure c) but with a further parallel resonance circuit 44a or 44b in series connection to the first capacitor 22a or 22b towards the connection point 19 for separate optimization of the circuit for each of the two frequency ranges L1 and L2 to design a two-band antenna.
  7. g) Reactance circuit 13c of the remaining passive vertical resonance radiator 4c, implemented as in Figure d), but with a further parallel resonance circuit 44 in series with the capacitance 28 in the reactance circuit 13c. The two active resonance excitation radiators are connected as in Figure f) to form a two-band antenna.
  8. h) Reactance circuit 13a or 13b of the two vertical active resonance excitation radiators 10a or 10b as in Figure f) but with an additional parallel capacitance 47, formed by the surface of the capacitance electrode 32a, 32b projecting over the insulated counter electrode 34 in the Figures 13c and 13d with the electrically conductive coated circuit board 35. The additional parallel capacitance 47 enables the range of the impedance matching to be extended at the connection point 19a, 19b while also maintaining the resonance condition for the ring line radiator 2. The resonance condition and the impedance matching are established by matching the sizes of the capacitance electrodes 32a, 32b and the insulated counter electrodes 34 to one another.
• Fig.6 :
  Adaptation ratios at the resonance frequency f0 depending on the dimension "t" for the division into the first capacity 22a, 22b and the second capacity 23a, 23b using the example of the ring line radiator 2 in Figure 4 .
  1. a) Reflection factor at the antenna connection point 5 as input port T1 of a loss-free distribution and phase shift network 8 of the antenna 1 as a function of the pitch t . Adaptation is achieved with topt = 0.13.
  2. b) Transformation factor at the resonance frequency f0 between the input resistance 43 with Z0 = 50 Ohm of the phase shifter 17 and the resonance resistance at f0 at the ring line radiator coupling point 7 as a function of the pitch t.
  3. c) Relative power P/Pmax at f0 depending on the pitch t.
• Fig.7 :
  Antenna according to the invention with resistance matching by capacitive subdivision as in the example of Figure 4 but with a total of 6 vertical resonance radiators 4a .. 4f, each azimuthally offset by 60°, of which four are designed as vertical passive resonance radiators 9a..9d. The excitation 3 takes place, for example, at the two azimuthally offset by 120° vertical resonance excitation radiators 10a, 10b offset from each other and excited by the difference angle ΔΦ= 120°.
• Fig.8 :
  Antenna according to the invention as in Figure 7 , but with a total of 8 vertical resonance radiators 4a..4h, each azimuthally offset by 45° from each other. The excitation 3 takes place, for example, at the two vertical resonance excitation radiators 10a, 10b, which are azimuthally offset by 90° from each other and excited by the difference angle ΔΦ= 90°.
• Fig.9 :
  Antenna according to the invention with rectangular ring line 14 with four vertical resonance radiators 4a to 4d in the area of the ring line corners. Two of the vertical resonance radiators are designed as vertical resonance excitation radiators 10a, 10b and the two remaining radiators as passive vertical resonance radiators 9a, 9b. Impedance matching to the low-impedance (eg Z= 50 Ohm) distribution and phase shift network 8 is achieved by capacitive subdivision, similar to the Figures 4 , 7 and 8th The ring line 14 is preferably designed to be square with the same ring line widths 15a, 15b and the same ring line sections 30a, 30b. However, both the ring line width 15a, 15b and the ring line sections 30a, 30b can be selected to be different within limits in order to match one another.
• Fig.10 :
  • Rectangular ring line radiator 2 of an antenna 1 according to the invention without representation of the electromagnetic excitation 3 to explain the resonance structure of the ring line radiator 2.
    1. a) geometric structure of the ring line radiator 2 with large, differently selectable ring line widths 15a and 15b as well as the differently selectable ring line sections 30a and 30b. The vertical resonance radiators 4a - 4d are with their ring line coupling points 7a - 7d in the wide area of the corners of the rectangular ring line 14 provided. The broken line roughly indicates the course of the center line 24 of the current density distribution of the line wave when the ring line radiator 2 is excited. Due to the current displacement, the current distribution is concentrated towards the edge of the ring conductor 14. The current of the line wave relevant for the function when the ring line radiator 2 is excited therefore flows - indicated by the center line of the current distribution 24 drawn as a broken line - more towards the edge even with large conductor widths 15a, 15b. This applies in particular to very large ring conductor widths 15a, 15b up to the complete closure of the inner opening towards the center, which is practically currentless if the ring line 14 is implemented as a closed conductive surface.
    2. b) according to the current represented by the center of gravity of the current density distribution 24, the resonance structure of the ring line radiator 2 can be generally represented by a roughly approximated equivalent circuit diagram. The individual ring line sections 30a - 30d are each represented by the inductive and capacitive effect of the associated section of the ring line 14, including the capacitive effect of the reactance circuit 13, as concentrated inductive elements (Ln) and capacitive elements (Cn). Each n-th ring line section 30a - 30b is represented by a π structure according to figure c), consisting of a longitudinal inductance 2*Ln and a transverse capacitance Cn at both ends. The radiation attenuation of each horizontally oriented ring line section 30a - 30b is included by the attenuation factor d of the concentrated inductance. The connection of adjacent ring line sections 30a-30d is carried out via a common vertical resonance radiator 4a..4d by contracting the transverse capacitances Cn of the adjacent ring line sections, as shown in Figure c). The slight inductive effect of the vertical resonance radiator is neglected in this basic consideration due to the low antenna height h. Advantageous conditions according to the invention for the ring line radiator 2 are achieved for the resonance frequency f0=ωo/2π when all wave impedances $\mathit{ZLn}=\sqrt{\frac{\mathit{Ln}}{\mathit{Cn}}}$
       
       and all $\sqrt{\mathit{Ln}\square \mathit{Cn}}=\raisebox{1ex}{$1$}\!\left/ \!\raisebox{-1ex}{$\mathit{\omega on}$}\right.$
       
       , which represent the electrical length of the ring line section, for all n are the same size.
  • This is always the case with rotationally symmetrical ring line radiators 2. Otherwise, this requirement can be met, for example with a rectangular structure of the ring line radiator 2, by individually designing the ring line widths 15a-15d in the individual ring line sections 30a-30b.
• Fig.11 :
  1. a) Design of the distribution and phase shifting network 8 - for example as in the Figures 7 and 8th - but as a hybrid ring 3 tuned to the resonance frequency f0, through which both the power division and the phase shift occur. When fed in at gate T1, the ring line radiator 2 is excited via gates T2 and T3 with a phase difference of 90°. The termination of gate T4 with the ohmic termination resistor 40 causes the partial absorption of the power of the undesired polarization over a broadband band when fed in at gate T1 if the frequency deviates from the resonance frequency f0.
  2. b) Idealized scattering matrix to describe the generally known fundamental wave behavior of a hybrid ring 38 in Figure a) at the gates T1 to T4. The important decoupling of the gates T1 and T4 from each other is marked in bold in the matrix by the scattering parameters S14 = 0, S41 = 0.
  3. c) Design of the distribution and phase shifting network 8 - for example as in the Figures 7 and 8th - but as a Wilkinson divider 16 tuned to the resonance frequency f0 with a downstream λ/4 line phase shifter 17 at a gate T3* to generate the phase difference of 90°. The ohmic balancing resistor 40 partially absorbs the currents that cause the undesired polarization on the ring line radiator 2 when the Wilkinson divider 16 is loaded asymmetrically and deviates from the resonance frequency f0.
• Fig.12 :
  Antenna according to the invention, for example as in Figure 9 with rectangular shaped ring line 14. The capacities 22a, 22b and 28 are formed in such a way that the vertical Resonance radiators 4a - 4d are formed at their lower end into individually designed flat capacitance electrodes 32a, 32b, 32c, 32d. By interposing between these and the electrically conductive base surface 6, which is designed as an electrically conductively coated circuit board, the capacitances 28 are designed to couple the passive vertical resonance radiators 4c, 4d to the electrically conductive base surface 6. For the capacitive coupling of the active vertical resonance radiators 10a, 10b to the gates T2 and T3 of the distribution and phase shift network 8, this connection is designed as a flat counter electrode 34 insulated from the conductive layer. The counter electrodes 34 can thus be used as connection points 19 of the antenna in Figure 9 and can be used as connection points for the gates T2 and T3 of the distribution and phase shifting network 8, as shown in Figure 13 indicated, serve.
• Fig.13 :
  Antenna 1 similar to Figure 11 , however, the dielectric effect of the dielectric plate 33 is realized by an air gap. Typical dimensions of a ring line radiator 2 for the frequency range L1 are a dimension of 30mm to 40mm for square antennas and a height of h = 8mm. To adjust the resistance of the distribution and phase shift network 8, which is preferably designed at a low impedance level (Z0), the capacitive subdivision is as in Figure 9 intended.
  1. a) The exemplary reactance circuits 13a, 13b and 13c are shown in the Figures 5a and 5b The capacitance 28 of the capacitance electrode 32c, 32d against the electrically conductive base surface 6 and the capacitance 22a, 22b against the insulated counter electrode 34a, 34b is approximately 0.3pF in each case. The gates T2 and T3 of the distribution and phase shift network 8 with their real input resistance 43 are each connected to a counter electrode 34a, 34b as a connection point 19a, 19b. The capacitive subdivision for impedance matching is given by the capacitance 22a, 22b and the capacitance 23a, 23b. This capacitance 23a, 23b introduced between the connection point 19a, 19b and the ground point 11 is attached to the back of the coated circuit board 35 as an SMD component. For this purpose, insulated pads 29 are designed as contact points on the back of the circuit board, each of which is connected to the Counter electrode 34 on the one hand and on the other hand e.g. via the capacitance 23a, 23b (see Fig.4 and 5a) are connected to ground.
  2. b) the reactance circuits 13a, 13b of the two resonance excitation radiators 10a, 10b are as shown in Figure a) according to Figure 5a or as in 5c. When designed according to Figure 5c an inductance 46a, 46b is connected in parallel as an SMD component to the capacitance 23a, 23b introduced on the back of the coated circuit board 35. In contrast to Figure a), insulated counter electrodes 34 are placed opposite the capacitance electrodes 32c, 32d on the coated circuit board 35 to form the reactance circuits of the two passive resonance radiators 9a, 9b. Starting from the counter electrode 34 as a contact support point, the parallel inductance 46 and the parallel capacitance 45 are connected to the ground point 11 on the circuit board 35 on the back of the circuit board 35, so that the reactance circuit 13 in Figure 5d is realized.
     When designing all reactance circuits 13 of all vertical resonance radiators with a capacitance electrode 32 and an opposite insulated counter electrode 34, all circuits can be Figures 5c and 5f for the resonance excitation radiators 10a, 10b by using SMD components on the back of the circuit board 35. In the same way, the reactance circuits for the passive resonance radiators 9a, 9b can be realized by connecting the insulated counter electrodes 34 with SMD components on the back of the circuit board 35 according to the Figures 5d and 5g specified circuits can be realized.
  3. c) As under Figure 5h As described above, the additional parallel capacitance 47 serves to freely design the impedance matching while maintaining the resonance properties of the ring line radiator 2. The figure shows the large-area overlap of the insulated counter electrode 34a, 34b with the capacitance electrode 32a, 32b of the resonance excitation radiators 10a, 10b. The flat projection in the vertical projection of the capacitance electrode 32a, 32b compared to the insulated counter electrode 34 forms the additional parallel capacitance 47 with the electrically conductive layer of the coated circuit board 35. The Reactance circuit of the two passive resonance radiators 9a, 9b are designed as in figure a).
  4. d) the reactance circuits 13 of the two resonance excitation radiators 10a, 10b are designed as in Figure c) with an additional parallel capacitance 47. The exemplary reactance circuits 13 of the two passive resonance radiators 9a, 9b can be designed as described in Figure b).
• Fig.14 :
  The picture shows the top side of the circuit board 35 of an antenna 1 according to the invention, onto which the electrical ring line radiator 2 is placed. For the broadband design of an antenna 1 according to the invention - which, for example, has both frequency ranges L1 and L2 with a center frequency fm lying between the two frequency ranges (see Figure 5e ) - is the reactance circuit 13, as shown in the Figures 5c or 5d. For this purpose, the reactance circuit 13a, 13b in each case is designed for the reactance circuit in the two active vertical resonance excitation radiators 10a, 10b Figure 5c and for the two passive vertical resonance radiators 9a, 9b the reactance circuit 13c in Figure 5d This is the case in the example in Figure 13a This is achieved by providing a counter electrode 34 for each capacitance electrode 32 and by connecting a parallel circuit comprising a parallel capacitance 23a, 23b or 45 and a parallel inductance 46a, 46b or 46 - shown as SMD components - in series between the counter electrode 34 and the electrically conductive base surface 6 to the capacitance 22a, 22b or 28 caused by the capacitance electrodes 32 on all vertical resonance radiators 4a - 4d. Figure 13a shows the top side of the circuit board 35 with vias 26 on the counter electrodes 34 under the capacitance electrodes 32. In Figure 14b the underside of the circuit board 35 is shown with the pads 29, which are connected to the insulated counter electrodes 34 via the through-holes 26. The dummy elements are SMD components according to the Figures 5c and 5d In addition, the two connection points 19 with the series circuit connected to them, consisting of the capacitance 28a, 28b and the inductance 27a, 27b as parts of the Figure 3a included adaptation circuit 18 of the distribution and phase shift network 8 for forming the gates T2 and T3.
• Fig. 15
  Shows an antenna according to the invention with four resonance excitation radiators 10a to 10d distributed azimuthally on the circumference of the ring line 14. All radiators are excited according to a rotating wave with a 90° phase difference between adjacent radiators. For this purpose, a first and a second distribution and phase shifter network (8) with a phase angle ΔΦ = 90° are present, the first distribution and phase shifter network (8) being coupled with its gates T2, T3 to a first pair of resonance excitation radiators 10a, 10b and the second distribution and phase shifter network (8) being coupled with its gates T2a, T3a to a second pair of excitation radiators 10c, 10d opposite the first pair. To excite a traveling wave, opposing resonance excitation radiators 10a, 10c and 10b, 10d are each excited with a phase difference of ΔΦ = 180°. For this purpose, a further distribution and phase shifter network 31, 39 with a phase shift of ΔΦ = 180° is provided, which is connected with its gate T2c to the gate T1 of the first distribution and phase shifter network 8, 36 and with its gate T3c to the gate T1b of the further distribution and phase shifter network 8, 38b. Here too, according to the invention, a hybrid ring 38 is preferably used. The 180° phase network 31 can, for example, be designed as a Wilkinson divider 39 with a λ/2-long delay line 17 (see Figure 11c ). All connections are selected in such a way that the conductive wave on the ring line is aligned in the desired direction of rotation. This arrangement has the particular advantage of azimuthally symmetrical excitation, so that the adjustment of the arrangement can be carried out particularly easily. One limitation, however, is the increased effort.

For the design of a dual-band multi-band antenna according to the invention - for example for the frequency ranges L1 and L2 - the reactance circuit 13 is designed to be multi-frequency in such a way that resonance of the ring line radiator 2 in the separate frequency bands L1 and L2 is given separately and accordingly as frequency f01 and as frequency f02. For this purpose, starting from the pads 29 in Figure 14b the back of the coated circuit board 35 is redesigned in such a way that the reactance circuits 13a, 13b, and 13c are realized.

The invention is described again below in connection with its advantageous embodiments.

In a basic form, an antenna 1 according to the invention contains a ring line radiator 2 already described above, as shown, for example, in Figure 2a is shown for a rotationally symmetrical arrangement with three vertical resonance radiators 4a, 4b, 4c distributed equidistantly along the electrically short ring conductor 14. From the ring conductor width 15 and the length of a ring conductor section 30a, 30b, etc. This results in an inductive effect which, together with the capacitive reactance circuit 13, causes a resonance behavior with a resonance circular frequency ω0 =2π*f0 in each of the vertical resonance radiators, which is determined by the frequency curve of the antenna impedance 37 with the impedance maximum 42 'in Figure 2b expresses.

According to the invention, this resonance structure at the natural resonance frequency f0 is capable, with appropriate excitation via two vertical resonance radiators 4a, 4b - here referred to as vertical resonance excitation radiators 10a, 10b - with signals corresponding to the phase of the azimuthal position of the vertical resonance radiators 4a, 4b - for example as in Figure 1 shown - to convey a running current wave in exactly one direction. The structure of the ring line radiator 2 in Figure 2a itself therefore does not yet have any feature for preferring one of the two possible directions of travel of the current wave. The running direction is determined exclusively by connecting the distribution and phase shifter network 8 according to the sign of the phase angle ΔΦ, as in Figure 1 , fixed. The phase difference ΔΦ between these signals is therefore selected according to the invention in such a way that, at the resonance frequency f0 of the ring line radiator 2, it corresponds to the phase difference ΔΦ of a current wave running on the ring line between the ring line coupling points 7a and 7b of the two resonance excitation radiators 10a and 10b. The direction of rotation of the current wave is therefore determined by the sign of the phase difference ΔΦ of the excitation signals in conjunction with the position of the two ring line coupling points 7a, 7b specified. The running direction is therefore determined exclusively by connecting the distribution and phase shifter network 8 according to the sign of the phase angle ΔΦ, as in Figure 1 , fixed.

Naturally, the condition for the excitation of a running direction can only be fulfilled exactly at the resonance frequency f0 and the proportion of the undesired running direction and thus the cross-polarization increase with increasing deviation from this frequency. However, a particular advantage of the invention is the very large bandwidth for a sufficiently large suppression of the cross-polarization resulting from the inventive requirement for a real input resistance 43 of both gates T2, T3 of the distribution and phase shift network 8, whereby the resonance frequency f0 of the ring line radiator 2 is not changed by the excitation 3 at the resonance frequency f0. A reactive power exchange between the ring line radiator 2 and the excitation 3 therefore does not take place at the resonance frequency f0. A particularly high bandwidth of the suppression of the cross-polarization is achieved if - as in Figure 1 shown - the gates T2 and T3 are directly connected to the vertical resonance excitation radiators 10a, 10b so that they are effective at the corresponding ring line coupling points 7a, 7b.

According to the invention, it is therefore provided that the ring line radiator 2 and the phase shift by the angle ΔΦ in the distribution and phase shifter network 8 are coordinated with one another in such a way that this angle at the resonance frequency f0 of the electrical length of the ring line section 30 as an angle measure corresponds to the proportion of the electrical length 2π corresponds to the ring line 14 between the two vertical resonance excitation radiators 10a, 10b, to which the gates T2 and T3 are connected. In this case, between the vertical resonance excitation radiators 9a, 9b on the ring conductor 14, further, but passive, vertical resonance radiators 4 - as in the Figures 7 and 8th shown - arranged.

This results in a total of N>3 vertical resonance radiators 4a, 4b, 4c, 4d and thus the same number of ring line sections 30a, 30b, 30c, 30d and above the ring line 14 equally distributed coupling points 7, the phase angle of the distribution and phase shifter network 8 set according to the invention to ΔΦ = n * 2π / N < 180 °, if n ring line sections 30 are selected between the coupling points 7 of the two vertical resonance excitation radiators 10a, 10b .

For rotationally symmetrical arrangements with a large number of N ring line sections in total, a large frequency bandwidth for suppressing cross polarization can advantageously be achieved.

For special applications, such as for the superposition of circularly polarized radiation with an azimuthal phase distribution of 2π and a circularly polarized radiation with an azimuthal phase distribution of 4π in an antenna diversity system with two antennas, the invention can also advantageously be applied to a ring line radiator 2 with one electrical length of 4π can be applied as follows. In a simple special case with a polygon-shaped ring line radiator 2 with N = 8 ring line sections 30 equally distributed azimuthally over the electrical length 4π, two mutually adjacent vertical resonance excitation radiators 10a, 10b (n = 1) are connected via a distribution and phase shifter network 8 with ΔΦ = π/2 excited. In analogy to this, with N = 16 vertical resonance radiators and n = 2 ring line sections between the two vertical resonance excitation radiators 10a, 10b, the phase angle ΔΦ = n*4π/N must also be set to ΔΦ = π/2. In an expedient and space-saving manner, two ring line radiators 2 with different phase distributions can be arranged concentrically to one another in such a way that the radiator with an azimuthal 2π phase distribution is surrounded by the radiator with a 4π phase distribution.

The ring line 14 is advantageously excited with the electrical length 2π of the ring line radiator 2 in which is tuned to the resonance frequency f0 Figure 1 in such a way that the gates T2, T3 have a real high-resistance input resistance 43 and are connected directly to the ring line coupling points 7a, 7b via the two vertical resonance excitation radiators 10a, 10b. In this case, both the distribution network is 16 and that Phase rotating element 17 is designed with high resistance in accordance with the high resonance resistance.

In an advantageous embodiment of the invention, in Figure 3 the distribution network 16 and the phase shifter 17 are designed with low resistance, ie for the impedance level Z0 (eg Z0 = 50 ohms). For transformation into the high impedance level of the ring line radiator 2, a matching network 18 is connected downstream of the two output branches. For example, the in. is suitable for this Figure 3a λ/4 transformation line 12a, 12b shown with the series resonance circuit consisting of the inductance 27a, 27b and the capacitance 28a, 28b.

In an advantageous embodiment of the invention, the impedance matching takes place between the low-impedance gates T 2, T3 of the distribution and phase shifter network 8 - designed at the impedance level Z0 - and the high-impedance resonance resistance of the ring line radiator 2 present at the ring line coupling points 7a, 7b - as in Figure 4 - by serial division of the capacitive reactance circuit 13a, 13b with the design of a connection point 19a, 19b in between for connecting the two gates T2, T3 with their respective real input resistance 43 at the impedance level Z0. The frequency bandwidth of the suppression of the cross polarization in antennas with circular polarization is highly dependent on the bandwidth of the antenna and thus on its electrical size. In the case of the electrically small ring line radiators 2 considered here with dimensions such as those used in connection with Figure 13 As described above, it is possible to detect each of the frequency bands L1 and L2 separately, given the frequently specified requirement of a cross-polarization distance of 15 dB and more.

According to the invention, the one in the example is suitable for this Figure 4 shown subdivision of the reactance circuit 13a, 13b of the two vertical resonance excitation radiators 10a, 10b, which are also in Figure 5a is shown. The division shown is given purely capacitively by the first capacitance 22a, 22b and the second capacitance 23a, 23b. It is important according to the invention that the vertical resonance excitation radiators 10a, 10b effective resulting capacitance of that in the remaining passive vertical resonance radiators 9a, 9b - represented by the capacitance 28 in the Figures 4 , 7 , 8th and in Figure 5b - corresponds and thus the resonance of the ring line radiator 2 is maintained at the resonance frequency f0 through the capacitive subdivision. An exchange of reactive power between the ring line radiator 2 and the electromagnetic excitation 3 at the connection point 19 is excluded at the resonance frequency f0. However, as the deviation of the frequency from the resonance frequency f0 increases, this feature no longer applies completely due to the frequency dependence of the ring line radiator 2 and the frequency dependence of the distribution and phase shifter network 8 with regard to the impedance matching and the phase shift. However, the described adjustment according to the invention advantageously leads to a comparatively particularly small increase in cross polarization as the frequency deviation increases. The antenna according to the invention is therefore advantageously broadband with regard to the suppression of cross polarization.

However, with a small antenna described here, it is hardly possible to cover the entire frequency band of both frequency ranges L1 and L2 between the frequencies fu2 and fo1 (see. Figure 5e ) with a single band antenna described above without additional measures.

According to the invention, the design of the reactance circuit 13a, 13b - shown in Figure 5c - in the two vertical resonance excitation radiators 10a, 10b and the reactance circuit 13c in Figure 5d a single-band antenna that covers both frequency bands L1 and L2.

This is achieved according to the invention in that both in the second reactance circuits 21a, 21b (see Figure 5c ) of the two vertical resonance excitation radiators 10a, 10b by parallel connection of the parallel inductance 46a, 46b to the existing second capacitance 23a, 23b as well as in the reactance circuits 13c (see. Figure 5d ) of the passive vertical resonance radiators 9c each have a parallel resonance circuit with the inductance 46 and the capacity 45 is added. Here, the resonance frequency of all parallel oscillation circuits is in the frequency range between the upper frequency band L1 and the lower frequency band L2 (eg fm in Figure 5e ) is selected such that the reactance of the resonant circuits is capacitive in the frequency band L1 and inductive in the frequency band L2. However, the reactance of all reactance circuits 13a, 13b, 13c as a whole is still capacitive in both frequency bands L1 and L2.

An advantageous further development of the invention according to the above design of the reactance circuits takes place in the design of a two-band antenna in such a way that both the first capacitance 22a, 22b of the first reactance circuit 13a 13b of the vertical resonance excitation radiators 10a, 10b and the capacitance 28 of the reactance circuit 13c of the passive vertical resonance radiator 9c, a further parallel resonance circuit 44, 44a, 44b is connected in series. This makes it possible, for example, in an advantageous manner, for both the reactance X1a, X1b of the first reactance circuit 20a, 20b and also the reactance X2a, X2b of the second reactance circuit 21a, 21b and thus the reactance and L2 is each capacitive and favorable values for the division dimension t for the capacitive division can also be selected separately for both frequency bands. The corresponding reactance circuits 13a, 13b and the reactance circuit 13c are in the Figures 5f and 5g shown. A suitable dimensioning of all reactive elements in the reactance circuit 13a, 13b makes it possible to achieve the above-described capacitive subdivision by selecting the resonance frequency of the parallel resonant circuits in relation to the center frequencies fm1 and fm2 with a view to the optimization described above with respect to t -topt.

In a particularly advantageous embodiment of the invention, the broadband suppression of the undesirable polarization direction LHCP of an antenna intended for RHCP with a hybrid ring 38 as a distribution and phase shifter network 8 for exciting the square ring line radiator 2 in Figure 13 reached. In a usual, known form, a hybrid ring 38 consists of λ/4-long line pieces, as in Figure 11a shown. It is also known that the line sections can be reproduced using concentrated dummy elements. In principle, a hybrid ring 38 can be manufactured for different impedance levels. However, production for the low impedance level Z0 is particularly economical.

However, a hybrid ring 38 can only be completely balanced for a specific frequency - generally approximately the center frequency of a frequency band (fm1, fm2) - and can be adjusted by the idealized scattering matrix with respect to its gates T1 to T4 Figure 11b to be discribed. At frequencies that deviate from this, the antenna produces in Figure 13 Naturally, when excitation at the gate T1 and when the ring line radiator 2 is excited via the gates T2 and T3, in addition to the desired radiation in the RHCP mode, there is also proportionally the unwanted radiation in the opposite direction of rotation, i.e. the LHCP mode. In addition, the reactive component of the antenna impedance increases with increasing frequency deviation from the resonance frequency f0, which also increases the LHCP mode. An optimal interaction between the hybrid ring 38 and the ring line radiator 2 with regard to the suppression of cross polarization is achieved according to the invention when the hybrid ring 38 is adjusted to the frequency f0 with regard to its real input resistance at the gates T2 and T3 and its required phase of 90 °, which also forms the resonance frequency f0 of the corresponding ring line radiator 2. As a special property of a hybrid ring 38, the connection of the gate T4, which is decoupled from the gate T1, with an effective resistance of the size Z0 according to the invention causes the undesired LHCP component in the radiation to be eliminated even at a frequency offset from the frequency f0, for which both the ring line radiator 2 and the hybrid ring 38 is tuned, is largely absorbed.

In an advantageous further development of the invention, the Wilkinson divider 39 is used as a distribution and phase shifter network 8 as a distribution network 16 with a downstream phase shifter 17 Figure 11 for application. The signal path, starting from gate T1, branches into two λ/4-long lines, the live ends of which are connected to the ohmic balancing resistor 41 Compensating resistance are connected to each other. For example, a λ/4-long line is connected downstream of the gate T3* as a phase rotating element 17 for a phase rotation of 90°. This line can also be replicated using concentrated components. The system is only balanced when gates T2 and T3 are closed with the correct resistance. Similar to the use of the hybrid ring 38 described above, this only applies to the resonance frequency f0. As the deviation from this frequency increases, the symmetry of the Wilkinson divider 39 is disturbed. According to the invention, the resulting undesirable LHCP component is also strongly damped via the ohmic balancing resistor 41.

To meet particularly high demands on the suppression of cross-polarization in the two frequency bands L1 and L2, it can be provided to use a separate antenna 1 according to the invention for each frequency band. In an extremely advantageous further development of the invention, the two ring line radiators 2 are arranged concentrically to one another in an antenna arrangement with the proviso that the ring line radiator 2 for the higher frequency in the band L2 is surrounded by the larger ring line radiator for the lower frequency in the band L1. Naturally, the increased effort of two separate distribution and phase shifter networks 8 tuned to the respective frequency band must be accepted. By utilizing the frequency spacing between different frequency bands, several antennas 1 according to the invention can be formed, each assigned to a frequency band, the ring line radiators 2 of which are arranged concentrically to one another in such a way that the ring line radiator 2 for the highest frequency band is arranged at the innermost and each additional ring line radiator 2 surrounds a ring line radiator 2 for the next higher frequency band. Such an arrangement has the particular advantage that in addition to the two frequency bands L2 and L1, the lower frequency band L5, which is also used for satellite navigation, can also be detected. By combining and parallel evaluating all signals from these frequency bands in the navigation receiver, the quality of the positioning results can be greatly improved even under difficult reception conditions.

Such a concentric arrangement of the ring line radiators 2 also has the advantage that a linear, vertically polarized antenna for other radio services can be arranged in the center of the arrangement without impairing the properties of the ring line radiators 2.

List of designations

• Antenna 1
• Ring line radiator 2
• electromagnetic excitation 3
• vertical resonance radiators 4, 4a, 4b, 4c,...
• Antenna connection point 5
• Conductive base area 6
• Ring line crosspoints 7, 7a,...7d,...
• Distribution and phase shifting network 8
• vertical passive resonance radiators 9, 9a, 9b
• vertical resonance excitation radiators 10a, 10b,...
• Ground connection point 11
• λ/4 transformation line 12
• Reactance circuit 13, 13a,...13c,...
• Ring line 14
• Ring conductor width 15, 15a,..,15c,.
• Distribution network 16
• Phase rotating element 17
• Matching network 18
• Junction point 19
• first reactance circuit 20a, 20b
• second reactance circuit 21a,21b
• first capacity 22a,22b
• second capacity 23a,23b
• Center line of the current density distribution 24
• Serial parallel resonance circuit 25
• Through-hole plating 26
• Inductance 27a,27b
• Capacity 28a,28b
• Pad 29
• Ring line section 30a, 30b, 30c, 30d
• 180° phase shifter network 31
• capacitance electrode 32a, 32b, 32c, 32d,
• dielectric plate 33
• insulated counter electrode 34
• Coated circuit board 35
• Support structure 36
• Antenna impedance 37
• Hybrid ring 38
• Wilkinson divider 39
• ohmic terminating resistor 40
• ohmic balancing resistance 41
• Measuring section 42
• Impedance maximum 42'
• real input resistance 43
• additional parallel resonance circuit 44
• Parallel capacity 45
• Parallel inductance 46, 46a, 46b
• Additional parallel capacity 47
• Extended length of the ring line radiator L
• Reactance X, X1a, X1b, X2a, X2b, Xc
• Height h
• Goals T1, T2, T3, T4
• Wave impedance ZL
• Center Z

Claims (15)

1. An antenna (1) for the reception of circularly polarized satellite radio signals, comprising at least one conductor loop arranged above a conductive base surface (6); and an arrangement connected to an antenna connection point (5) for the electromagnetic excitation (3) of the conductor loop, comprising the following features:

   - the conductor loop is designed as a ring line radiator (2) by a polygonal or circular closed ring line (14) extending at a height h < 0.15 of the free-space wavelength λ above the conductive base surface (6),

   - the ring line radiator (2) forms a resonance structure and can be excited by the electromagnetic excitation (3) such that the current distribution of a running line wave in a single revolving direction is adopted on the ring line (14), the phase difference of said line wave over one revolution amounting to just 2π,

   - vertical resonance radiators (4) which are galvanically coupled to the ring line radiator (2) at ring line coupling points (7, 7a, 7b, 7c, 7d) and which extend toward the conductive base surface (6) are present distributed at the periphery of the ring line radiator (2),

   - at least one distribution and phase shifter network (8) is present which has one gate T1 coupled at the input side to the antenna connection point (5) and two gates T2, T3 whose transmission coefficients S ₁₂ and S ₁₃ differ from one another by a phase angle ΔΦ,

   comprising the following features:

   - the periphery (L) of the ring line (14) is shorter than the free-space wavelength λ,

   - at least three vertical radiators (4, 4a, 4b, 4c,...) are present which are coupled to the base surface (6), by which a resonance is produced, via reactance circuits (13) having a capacitive reactance X,
   each of the gates T2, T3 is coupled to a respective ring line coupling point (7, 7a, 7b),

   - characterized in that

   the input resistance of each of the gates T2, T3 of the distribution and phase shifter network (8) for the electromagnetic excitation of the line wave on the ring line (14) is a real resistance (43) in each case.
2. An antenna (1) according to claim 1,
   characterized in that
   two of the vertical resonance radiators (4a, 4b) are associated with the distribution and phase shifter network (8) as vertical resonant excitation radiators (10a, 10b) such that each of the gates T2, T3 of the distribution and phase shifter network (8) is coupled via a respective one of the two vertical resonant excitation radiators (10a, 10b) to its ring line coupling point (7a, 7b) and the other vertical resonance radiators (4c, 4d,..) act as passive vertical resonance radiators (9a, 9b, etc.).
3. An antenna (1) according to claim 2,
   characterized in that
   each of the gates T2, T3 is in each case connected directly via one of the vertical resonant excitation radiators (10a, 10b) to its ring line coupling point (7a, 7b) and the input resistance of the gates T2, T3 is real with high impedance between 300 ohms and 500 ohms such that there is resistance matching between said input resistance and a high-impedance resonance resistance of the ring line radiator (2).
4. An antenna (1) according to least one of the claims 1 to 3,
   characterized in that,
   in the distribution and phase shifter network (8), a distribution network (16) and a phase rotation element (17) connected downstream in one of its output branches are present which are both designed for a low-impedance impedance level Z₀ and, in each of the two output branches, a lossless matching circuit (18) is present for transforming the low-impedance impedance level Z₀ into an input resistance of the gates T2, T3 that is real with high impedance between 300 ohms and 500 ohms for matching to a high-impedance resonance resistance of the ring line radiator (2).
5. An antenna (1) according to at least one of the claims 2 to 4,
   characterized in that
   the reactance circuit (13) with which each of the passive vertical resonance radiators (9, 9a, 9b) and also each of the vertical resonant excitation radiators (10a, 10b) is coupled to the base surface (6) is in each case formed by a capacitor (28).
6. An antenna (1) according to at least one of the claims 2 to 5,
   characterized in that
   the reactance circuit (13) with which the vertical resonant excitation radiators (10a, 10b) are coupled to the base surface (6) is in each case formed as a serial connection of a first reactance circuit (20a, 20b) and a second reactance circuit (21a, 21b) with a connection point (19) which connects them and to which a respective one of the gates T2, T3 is connected.
7. An antenna (1) according to claim 6,
   characterized in that
   the reactance circuit (13) with which the vertical resonant excitation radiators (10a, 10b) are coupled to the base surface (6) and the first reactance circuit (20a, 20b) of each vertical resonant excitation radiator (10a, 10b) are formed by a first capacitor (22a, 22b) and the second reactance circuit (21a, 21b) is formed by a second capacitor (23a, 23b) and the size and the ratio of the first and the second capacitor are each selected such that resistance matching is in each case provided between the high-impedance resonance resistance of the ring line radiator (2) and a lower-impedance real input resistance of the gates T2, T3.
8. An antenna (1) according to at least one of the claims 6 to 7,
   characterized in that
   the reactance circuits (13c, etc.) with which the passive vertical resonance radiators (9c, etc.) are coupled to the electrically conductive base surface (6) are each formed from the serial connection of a capacitor (28) and a parallel oscillating circuit - consisting of a parallel capacitor (45) and a parallel inductor (46) - and the first reactance circuit (20a, 20b) of the vertical resonant excitation radiators (10a, 10b) is in each case formed as a first capacitor (22a, 22b) and the second reactance circuit (21a, 21b) of said vertical resonant excitation radiators (10a, 10b) is in each case designed as a parallel oscillating circuit - consisting of a second capacitor (23a, 23b) and a parallel inductor (46a, 46b) - and the resonant frequency of all the parallel oscillating circuits is in each case selected in the frequency range between the upper frequency band L1 and the lower frequency band L2 such that the reactance of the resonant circuits is capacitive in the frequency band L1 and inductive in the frequency band L2.
9. An antenna (1) according to claim 8,
   characterized in that
   a respective further parallel resonant circuit (44) is, however, connected in series between the capacitor (28) and the parallel oscillating circuit, consisting of the parallel capacitor (45) and the parallel inductor (46), of the reactance circuits (13c, etc.) of the passive vertical resonance radiators (9c, etc.) and a respective further parallel resonant circuit (44a, 44b) is likewise connected downstream of the first capacitor (22a, 22b) in the first reactance circuit (20a, 20b) of the vertical resonant excitation radiators (10a, 10b) toward the connection point (19a, 19b) and the parallel oscillating circuits in the first reactance circuit (20a, 20b) and the second reactance circuit (21a, 21b) of the vertical resonant excitation radiators (10a, 10b) are matched such that the reactance (Xa, Xb) in both frequency bands L1, L2 is capacitive in each case and their ratio to one another is present in accordance with the optimal dimension (topt) for matching and the reactance circuits (13c, etc.) of the passive vertical resonant radiators (9c, etc.) and the vertical resonant excitation radiators (10a, 10b) are matched to one another with respect to their frequency behavior.
10. An antenna (1) according to claim 8 or 9,
    characterized in that
    the first capacitor (22a, 22b) of the first reactance circuit (20a, 20b) of the vertical resonant excitation radiators (10a, 10b) and the capacitors (28) in the reactance circuits (13c, etc.) of the passive vertical resonance radiators (9c, etc.) are formed such that all the vertical resonance radiators (4a, 4b, 4c, 4d) are formed into individually designed areal capacitor electrodes (32a, 32b, 32c, 32d) at their lower ends; in that the capacitors (22a, 22b, 28) are designed for coupling the passive vertical resonance radiators (9c, 9d, etc.) to the electrically conductive base surface (6) by interposing a dielectric board (33) between the areal capacitor electrodes (32a, 32b, 32c, 32d) and the electrically conductive base surface (6) designed as an electrically conductively coated circuit board (35); and in that, for the capacitive coupling of the vertical resonant excitation radiators (10a, 10b) on the electrically conductive base surface (6), a respective areal counter-electrode (34) insulated from the conductive layer of the coated circuit board (35) is designed for connecting the second reactance circuit (21a, 21b).
11. An antenna (1) according to at least one of the claims 1 to 10,
    characterized in that
    a total of three vertical resonance radiators (4a, 4b) having azimuthally uniformly distributed ring line coupling points (7) are present, of which two are excited as vertical resonant excitation radiators (10a, 10b) via the gates T2, T3 of a distribution and phase shifter network (8) whose phase angle is ΔΦ = 120 degrees.
12. An antenna (1) according to at least one of the claims 1 to 10,
    characterized in that
    the ring line (14) is designed as square with four ring line coupling points at the corners and four vertical resonance radiators (4a, 4b, 4c, 4d) having azimuthally equally long ring line sections (30a, 30b) are present, of which two azimuthally consecutive ones are excited as vertical resonant excitation radiators (10a, 10b) via the gates T2, T3 of a distribution and phase shifter network (8) whose phase angle is ΔΦ = 90 degrees.
13. An antenna (1) according to claim 12,
    characterized in that
    the distribution and phase shifter network (8) includes a hybrid ring (38) for a phase shift by the phase angle ΔΦ = 90°, which hybrid ring, in addition to the two gates T2 and T3, has a fourth gate T4 which is terminated with an ohmic terminal resistance (40) and which is decoupled from the gate T1 when the gates T2 and T3 are terminated with the correct wave impedance.
14. An antenna (1) according to claim 4,
    characterized in that
    the lossless matching circuit (18) for transforming the low-impedance impedance level Z0 into the high-impedance real input resistance (43) is formed in both output branches (T2, T3) of the distribution and phase shifter network (8), in each case substantially by a lambda/4 transformation line (12) and a serial connection of an inductor (27a, 27b) and a capacitor (28a, 28b) for a fine adjustment of the high-impedance real input resistance (43) of the gates T2 and T3.
15. An antenna arrangement in which a plurality of antennas according to at least one of the preceding claims are provided which are each associated with a frequency band and whose ring line radiators (2) are arranged concentrically to one another such that the ring line radiator (2) for the frequency band with the highest frequency is arranged at the innermost point and each further ring line radiator (2) surrounds a ring line radiator for the next higher frequency band.

Images