EP2034557B1 - Antenna for satellite reception - Google Patents

Description

The invention relates to an antenna for receiving circularly polarized satellite radio signals.

In particular, in satellite broadcasting systems, it is particularly important to economize both with respect to the transmitted power emitted by the satellite and the efficiency of the receiving antenna. Satellite radio signals are transmitted due to polarization rotations in the transmission path usually with circularly polarized electromagnetic waves. In many cases, program contents are transmitted, for example, in frequency bands closely spaced separate frequency bands. This is done in the example of SDARS satellite broadcasting at a frequency of about 2.3 GHz in two adjacent frequency bands each with a bandwidth of 4 MHz with a spacing of the center frequencies of 8 MHz and 4 MHz. The signals are emitted by different satellites with a circularly polarized in one direction electromagnetic wave. As a result, circularly polarized antennas are used for reception in the corresponding direction. Such antennas are for example off DE-A-4008505 and DE-A-10163793 known. This satellite broadcasting system is additionally supported by the regional emission of terrestrial signals in another, arranged between the two satellite signals frequency band of the same bandwidth.

In a satellite broadcasting system, in which signals are transmitted in frequency closely adjacent frequency bands approximately the same width, the circularly polarized waves are radiated but with opposite directions of rotation, would therefore have to receive the two frequency bands differently circularly polarized antennas, for example according to Pattern out DE-A-4008505 and DE-A-10163793 known embodiments are used. In particular for reception in vehicles is the use of multiple antennas with separate Lines to the receiver or the use of a complicated switching device for selective reception of one or the other signal economically complex and thus disadvantageous. Separate processing based on frequency-selective measures of the two frequency bands within one and the same antenna is not achievable due to the high selection requirement with economic means.

Out US 5,751,252 For example, a satellite receiving antenna is known in which a dipole antenna whose antenna connection point lies in the plane of the loop antenna is arranged in the center of a circular loop antenna. The dipole antenna and loop antenna are connected via half-wave / half-balanced lines to a 90 ° hybrid coupler which combines the horizontally polarized signals received from the loop antenna and the vertically polarized signals received from the dipole antenna into a circularly polarized signal. The conductor loop of the loop antenna is interrupted on diametrically opposite sides, wherein one of the breaks contains a capacitive element and the other interruption forms an antenna junction, which is connected to the two halves of the conductor loop via a matching network of capacitors. The circumference of the conductor loop is half a wavelength long. Other antennas combining horizontal and vertical electric fields for receiving circularly polarized signals are off JP 2000-077934 A1 and DE 101 63 793 A1 known.

The object of the invention is therefore to provide an antenna which is suitable for receiving the radiated in both satellite frequency bands electromagnetic waves with both left-handed (LHCP) and with clockwise circular polarization (RHCP) and at its antenna connection point about the same, for the Satellite reception has suitable radiation characteristics. The antenna should also be able to be economically designed.

This object is achieved by the features specified in claims 1, 3 or 8.

According to the invention, the antenna for receiving circularly polarized satellite radio signals comprises at least one two- or three-dimensional antenna conductor structure connected to an antenna output terminal and is characterized in that the multidimensional antenna conductor pattern is designed to consist essentially of a plurality of antenna conductor sections which, relative to a space reference point common to the antenna conductor sections, arranged in pairs symmetrically and in the same direction, and that the multidimensional antenna conductor structure is further configured such that when the antenna is reciprocally operated as the transmitting antenna in the individual pairs of antenna conductor sections, antenna currents of at least approximately equal magnitude and the arithmetic mean of the current phases of these antenna currents respectively counted in the antenna conductor portions of each pair in the same direction at substantially seed individual pair of antenna conductor sections has at least approximately the same value with respect to a common phase reference point.

Such an antenna is capable of receiving and leaving both left-handed circularly polarized waves and right-handed circularly polarized waves Realize itself by relatively simple Antennenleitererstrukturen also suitable for receiving satellite signals elevation angle of the radiation pattern.

The distribution of the currents on an antenna in receive mode depends on the terminator at the antenna junction. In contrast, in the transmission mode, the distribution of the currents on the antenna conductors relative to the supply current at the antenna connection point is independent of the source resistance of the supplying signal source and is thus clearly linked to the directional diagram and the polarization of the antenna. Because of this uniqueness in conjunction with the law of reciprocity, according to which the radiation properties - such as directional diagram and polarization - are identical in transmission mode as in receiving mode, the object of the invention with respect to polarization and directional diagrams on the basis of the design of the antenna structure for generating corresponding currents in the transmission mode Antenna solved. Thus, the object of the invention for the receiving operation is solved. All considerations made below about currents on the antenna structure and their phases or their phase reference point thus refer to the reciprocal operation of the receiving antenna as a transmitting antenna, unless the receiving mode is specifically addressed.

• Fig. 1:
  Frequenzbänder zweier Satelliten-Rundfunksignale mit in unterschiedliche Drehrichtungen zirkular polarisierter Ausstrahlung in dichter Frequenznachbarschaft;
• Fig. 2:
  Darstellung des Zusammenhangs zwischen Strom durchflossenen, beliebig orientierten elektrisch sehr kurzen Leiterelementen und den zugehörigen elektrischen und magnetischen Feldstärkevektoren in einem fernen Aufpunkt;
• Fig. 3:
  1. a) Einen Monopol 7, der zur Gestaltung seines Vertikaldiagramms eine mit einem Blindelement 8 beschaltete Unterbrechungsstelle 5 aufweist,
  2. b) ein Vertikaldiagramm für den Empfang im Bereich von Elevationswinkeln zwischen 25° und 65°;
• Fig. 4:
  Satelliten-Empfangsantenne für den Empfang von Satellitensignalen kombiniert mit einer längeren Antenne für den Empfang von AM/FM-Rundfunksignalen;
• Fig. 5:
  1. a) Eine kreisförmige Schleifenantenne 14 mit Kapazitäten 16,
  2. b) eine kreisförmige Schleifenantenne 14 in konstanter Höhe h über einer leitenden Grundfläche 6 mit fiktivem Spiegelbild,
  3. c) ein Detail der Schleifantenne 14 zur Erläuterung der Berechnung des Wellenwiderstands Zw der umlaufenden Leitung über der leitenden Grundfläche 6;
• Fig. 6:
  Eine Variante der Schleifantenne 14 in Fig. 5b mit Auskopplung der Empfangssignale über eine symmetrische Zweidrahtleitung 26 außerhalb ihres Zentrums Z und mit einem Umsymmetrierglied 29 und einem Anpassnetzwerk 25;
• Fig. 7:
  Ein Vertikaldiagramm einer Schleifenantenne 14 nach Fig. 5b und Fig. 6 für
  1. a) links drehende zirkulare Polarisation und
  2. b) rechts drehende zirkulare Polarisation;
• Fig. 8:
  Eine weitere Ausführungsform der Schleifenantenne 14 bei welcher das Anpassnetzwerk 25 und das Umsymmetrierglied 29 auf der Ringebene angeordnet sind. Für die Verbindung zu einem Antenneneingangsanschluss 28 ist eine Zweidrahtleitung 26 im Zentrum Z zur leitenden Grundfläche 6 geführt, wo sie als Mikrostreifenleiter 30 über der Grundfläche 6 zur Anschlussstelle 28 weitergeführt ist;
• Fig. 9:
  Eine Ausführungsform der Schleifenantenne 14 gemäß der Erfindung mit einem als Stabantenne ausgebildeten Monopol 7 für den Empfang vertikal polarisierter Felder im Zentrum Z der horizontalen Schleifenantenne 14 mit einem Leistungsteiler- und Phasenschiebernetzwerk 31 zur phasengerechten Überlagerung der horizontal und vertikal polarisierten Feldanteile;
• Fig. 10:
  Eine Antenne ähnlich Fig. 9 jedoch mit einer vertikalen Zuleitung 26 zur Speisung der Schleifenantenne 14, wobei die Zuleitung 26 einen Monopol 7 und die Schleifenantenne 14 eine Dachkapazität 12 des Monopols 7 bildet;
• Fig. 11:
  Eine Schleifenantenne 14 mit zwei symmetrisch zueinander angeordneten Antennenanschlussstellen 3 und je einem Anpassnetzwerk 25 in der Schleifenebene sowie mit zentralem Anschluss an eine vertikale Zuleitung 26 als Alternative zu Figur 10;
• Fig. 12:
  Eine Ausführungsform mit zweigeteilter Zuführung zu der Schleifenantenne 14 in Form eines Bandleiters 34 mit durch Pfeile gekennzeichneten Stromwegen;
• Fig. 13:
  1. a) Eine symmetrische Ausführungsform einer Antenne mit vier in einem Quadrat angeordneten, horizontal liegenden Dipolen 21 und mit einem zentral im Phasenzentrum B angeordneten Verteilungsnetzwerk 10, dessen Ausgang 24 den Ausgangsanschluss 28 bildet.
  2. b) eine Antennenanordnung ähnlich Figur 13a jedoch mit einer Überlagerung empfangener horizontaler und vertikaler elektrischer Feldkomponenten wie an Hand der Figuren 10 und 11 erläutert. Das Dipolsystem wirkt als Dachkapazität des auf diese Weise gebildeten vertikalen Monopols;
• Fig. 14:
  Eine Antennenanordnung nach der Erfindung als Diversity-Empfangsantenne mit einem entsprechend gestalteten Verteilungsnetzwerk 10 zum Verfügbarmachen sowohl der Empfangssignale der Schleifenantenne 14 mit horizontal orientierten Leiterelementen als auch den Empfangssignalen des vertikalen Monopols 7;
• Fig. 15:
  Eine Antennenanordnung ähnlich Figur 10 mit einem extrem einfach als Blindwiderstand 41 realisierbaren Leistungsteiler- und Phasenschiebernetzwerk 31 über der Grundfläche 6;
• Fig. 16:
  Eine Antennenanordnung ähnlich den Beispielen in den Figuren 8 bis 15 mit einer in einer senkrecht zur Grundfläche 6 und symmetrisch bezüglich der Antennenanschlussstelle 3 orientierten Symmetrieebene SE der Antennenanordnung angeordneten linearen oder flächig gestalteten Antenne 24 für einen weiteren Funkdienst oder mehrere weitere Funkdienste;
• Fig. 17:
  1. a) Eine Antennenanordnung mit einem als Stabantenne ausgebildeten, vertikal polarisierten Monopol 7 und einer horizontal polarisierten Schleifenantenne 14 nach der Erfindung mit auf den Sendefall bezogen gemeinsamem Phasenzentrum B wie in Figur 9, jedoch mit getrennter Zuführung der Signale zum Anschluss für Vertikalpolarisation 49 beziehungsweise zum Anschluss für Horizontalpolarisation 48 eines Hybridkopplers 45 mit 90° positivem beziehungsweise negativem Phasenunterschied bezüglich des LHCP-Anschlusses 46 und des RHCP-Anschlusses 47 für getrennte Verfügbarkeit von LHCP- beziehungsweise RHCP-Signalen,
  2. b) eine Antennenanordnung wie in Figur a) jedoch mit einer Realisierung des Monopols 7 gemäß der Antennenanordnung in Figur 10 durch die Kombination der Wirkungen der Schleifenantenne 14 als Dachkapazität und der Zweidrahtleitung 26;
• Fig. 18:
  Eine Antennenanordnung mit phasengleicher Überlagerung der Empfangsspannungen aus den horizontalen und den vertikalen elektrischen Feldanteilen einer Schleifenantenne 14 und einer durch die vertikalen Zweidrahtleitung 26 gebildeten Monopolantenne 7. Mit Hilfe eines in einen der Leiter der Zweidrahtleitung 26 eingebrachten Netzwerks 53 erfolgt die Einstellung des Gleichtakt-zu-Gegentakt-Verhältnisses auf der vertikalen Zweidrahtleitung 26, womit das Verhältnis des Anteiles des vertikal polarisierten Feldes mit niederer Elevation der Hauptstrahlrichtung zu dem Anteil des horizontal polarisierten Feldes mit höherer Elevation der Hauptstrahlrichtung eingestellt wird. Dieses Netzwerk 53 kann im einfachsten Fall als eine Kapazität gestaltet werden;
• Fig. 19:
  1. a) Eine Antennenanordnung zur alternativen Auskopplung von RHCPbeziehungsweise LHCP-Signalen mit einer Schleifenantenne 14 mit zwei einander gegenüberliegenden Antennenanschlussstellen 3 und daran angeschlossenen Anpassnetzwerken 25 und einem im Zentrum der Schleifenantenne 14 befindlichen Monopol 7 in Form einer Stabantenne. Die Empfangssignale der beiden Antennen werden in einem 90°-Hybridkoppler 45 überlagert, an dessen Ausgänge ein LHCP/RHCP-Umschalter 55 angeschlossen ist. Angesteuert durch einen im Empfänger befindlichen Umschalter zwischen LHCP und RHCP - Satelliten-Empfangssignalen stehen die Signale der beiden Drehrichtungen der Polarisation alternierend zur Verfügung;
  2. b) Eine Variante der Antennenanordnung, die auch den Empfang elliptisch polarisierter Felder erlaubt;
• Fig. 20:
  Eine Antennenanordnung ähnlich der Variante der Fig. 19, bei der jedoch entsprechend der Antenne in Fig. 11 der Monopol 7 durch eine Zweidrahtleitung 26 gebildet ist, die die Schleifenantenne 14 mit der leitenden Grundfläche 6 verbindet.

The invention will be explained in more detail below with reference to exemplary embodiments. The accompanying figures show in detail:

• Fig. 1 :
  Frequency bands of two satellite broadcast signals with circularly polarized radiation in different directions of rotation in dense frequency neighborhood;
• Fig. 2 :
  Representation of the relationship between current traversed, arbitrarily oriented electrically very short conductor elements and the associated electric and magnetic field strength vectors in a distant Aufpunkt;
• Fig. 3 :
  1. a) a monopole 7 which has an interruption point 5 connected to a dummy element 8 for the purpose of designing its vertical diagram,
  2. b) a vertical diagram for reception in the range of elevation angles between 25 ° and 65 °;
• Fig. 4 :
  Satellite receiving antenna for receiving satellite signals combined with a longer antenna for receiving AM / FM broadcast signals;
• Fig. 5 :
  1. a) A circular loop antenna 14 with capacitances 16,
  2. b) a circular loop antenna 14 at a constant height h above a conductive base 6 with a fictitious mirror image,
  3. c) a detail of the grinding antenna 14 for explaining the calculation of the characteristic impedance Zw of the circulating line over the conductive base 6;
• Fig. 6 :
  A variant of the grinding antenna 14 in FIG Fig. 5b with coupling of the received signals via a symmetrical two-wire line 26 outside its center Z and with a Umsymmetrierglied 29 and a matching network 25;
• Fig. 7 :
  A vertical diagram of a loop antenna 14 after Fig. 5b and Fig. 6 For
  1. a) left rotating circular polarization and
  2. b) clockwise rotating circular polarization;
• Fig. 8 :
  Another embodiment of the loop antenna 14 in which the matching network 25 and the Umsymmetrierglied 29 are arranged on the ring plane. For the connection to an antenna input terminal 28, a two-wire line 26 is led in the center Z to the conductive base 6, where it as a microstrip conductor 30 is continued over the base 6 to the connection point 28;
• Fig. 9 :
  An embodiment of the loop antenna 14 according to the invention with a trained as a rod antenna monopole 7 for receiving vertically polarized fields in the center Z of the horizontal loop antenna 14 with a power splitter and phase shifter network 31 for phase-correct superposition of horizontally and vertically polarized field components;
• Fig. 10 :
  An antenna similar Fig. 9 but with a vertical feed line 26 for feeding the loop antenna 14, the feed line 26 forming a monopole 7 and the loop antenna 14 forming a roofing capacity 12 of the monopole 7;
• Fig. 11 :
  A loop antenna 14 with two antenna connection points 3 arranged symmetrically to one another and one matching network 25 each in the loop plane and with a central connection to a vertical feed line 26 as an alternative to FIG. 10 ;
• Fig. 12 :
  An embodiment with two-part supply to the loop antenna 14 in the form of a strip conductor 34 with current paths indicated by arrows;
• Fig. 13 :
  1. a) A symmetrical embodiment of an antenna with four arranged in a square, horizontally disposed dipoles 21 and with a centrally located in the phase center B distribution network 10, the output 24 forms the output terminal 28.
  2. b) an antenna arrangement similar FIG. 13a however, with a superposition of received horizontal and vertical electric field components as with the hand Figures 10 and 11 explained. The dipole system acts as a roofing capacity of the vertical monopole thus formed;
• Fig. 14 :
  An antenna arrangement according to the invention as a diversity receiving antenna with a correspondingly designed distribution network 10 for making available both the received signals of the loop antenna 14 with horizontally oriented conductor elements and the received signals of the vertical monopole 7;
• Fig. 15 :
  An antenna arrangement similar FIG. 10 with a power divider and phase shifter network 31, which can be realized extremely simply as a reactance 41, over the base surface 6;
• Fig. 16 :
  An antenna arrangement similar to the examples in FIGS FIGS. 8 to 15 with a linear or area-shaped antenna 24 arranged in a plane perpendicular to the base 6 and symmetrically with respect to the antenna connection point 3 symmetrical plane SE of the antenna arrangement for another radio service or several other radio services;
• Fig. 17 :
  1. a) An antenna arrangement with a rod antenna designed as a vertically polarized monopole 7 and a horizontally polarized loop antenna 14 according to the invention with respect to the transmission case common phase center B as in FIG. 9 but with separate supply of the signals to the vertical polarization terminal 49 and the horizontal polarization terminal 48 of a hybrid coupler 45 with 90 ° positive and negative phase difference with respect to the LHCP terminal 46 and the RHCP terminal 47 for separate availability of LHCP and RHCP signals, respectively .
  2. b) an antenna arrangement as in Figure a) but with a realization of the monopole 7 according to the antenna arrangement in FIG. 10 by combining the effects of the loop antenna 14 as the roof capacitance and the two-wire line 26;
• Fig. 18 :
  An antenna arrangement with in-phase superposition of the received voltages from the horizontal and the vertical electric field components of a loop antenna 14 and one through the vertical Two-wire line 26 formed monopole antenna 7. With the help of a introduced into one of the conductors of the two-wire line 26 network 53, the adjustment of the common mode to-push-pull ratio on the vertical two-wire line 26, whereby the ratio of the proportion of vertically polarized field with low elevation of the main beam direction is set to the proportion of the horizontal polarized field with higher elevation of the main beam direction. This network 53 can be designed as a capacity in the simplest case;
• Fig. 19 :
  1. a) An antenna arrangement for the alternative coupling of RHCP and LHCP signals with a loop antenna 14 with two opposing antenna connection points 3 and connected matching networks 25 and located in the center of the loop antenna 14 monopole 7 in the form of a rod antenna. The received signals of the two antennas are superimposed in a 90 ° hybrid coupler 45, to whose outputs an LHCP / RHCP switch 55 is connected. Triggered by a switch located in the receiver between LHCP and RHCP satellite received signals, the signals of the two directions of rotation of the polarization are alternately available;
  2. b) A variant of the antenna arrangement, which also allows the reception of elliptically polarized fields;
• Fig. 20 :
  An antenna arrangement similar to the variant of Fig. 19 in which, however, according to the antenna in Fig. 11 the monopole 7 is formed by a two-wire line 26 which connects the loop antenna 14 to the conductive base 6.

Although the object of the invention is directed to a receiving antenna, the properties of the antenna are described below for better traceability for the reciprocal operation of the antenna as a transmitting antenna, the transmission case, however, due to the naturally valid reciprocity relationship also applies to the reception case.

A particular advantage of an antenna according to the invention is the property that the electric field strength vector generated according to the law of reciprocity when operating the antenna as transmitting antenna in the far field is polarized in each point of the room at any time along a fixed, fixed line specific to that point of the room in that, however, there is no equality requirement with respect to the direction of this line in the space for the different spatial directions of the radiation pattern, as is known in the case of radio transmission with linear polarized antennas. Naturally, this line is always perpendicular to the propagation direction, but with respect to its other direction according to the invention is completely free formable. This results in a manageability variety that allows optimal adaptation to a required radiation pattern. For the inventive design of the antenna, it is only necessary to preclude a temporal change in the direction of the electrical and thus the magnetic field strength vector for the reciprocal operation as a transmitting antenna in each spatial direction over the period of high frequency oscillation. Spaces in which this condition is not met, always contribute to the support of one of the two satellite signals and thus forcibly to the attenuation of the other satellite signal and thus weaken the overall system.

In FIG. 1 illustrates the problem from which the invention proceeds. The problem arises from the fact that two satellite radio frequency bands with small bandwidth Bu or Bo closely adjacent at a high frequency in the L band or in the S band, in any case at a frequency of fm> 1 GHz with opposite directions, that is with right rotating circular polarization (RHCP) or left rotating circular polarization (LHCP) are radiated. With a bandwidth Bu or Bo of a few megahertz (typically about 4 -25 MHz), the relative frequency spacing between the center frequencies fmu and fmo is so small that a frequency-selective design of the antenna for left-rotating and right-rotating circular polarization is not possible at the same time.

In the following, the principles for the design of antennas are explained, which are based on the antenna according to the invention.

und $\underline{\overrightarrow{H}}$

erläutert werden. Die elektrisch sehr kurzen Leiterelemente sind als Vektoren Δ ₁ ...Δ ₅ dargestellt, deren Richtung sowohl durch die Richtung der Lage im Raum als auch durch die Zählpfeilrichtung des auf dem Leiterelement fließenden Stromes, welcher nach Betrag und Phase als konstant angesehen werden kann, gegeben ist. Die Koordinatenrichtungen des räumlichen Koordinatensystems sind mit x, y und z bezeichnet, sein Koordinatenursprung mit B. In einer allgemeinen Beschreibung des v-ten Leiterelementes mit dem komplexen Strom Iv und seiner durch den Positionsvektor p _ν beschriebenen Position im Raum kann sein Beitrag zum komplexen elektrischen Feldstärkevektor ${\underline{\overrightarrow{E}}}_y$

im - mit Abstand r_A vom Ursprung B des Koordinatensystems entfernten - Fernfeld-Aufpunkt P, dessen Position ferner durch den Einheits-Richtungsvektor r beschrieben ist, angegeben werden. Sind N solcher Leiterelemente vorhanden, dann lautet die elektrische Feldstärke summarisch: $${\overrightarrow{\underline{E}}}_v=\mathit{jexp}\left(-{\mathit{j\beta r}}_A\right)\cdot \frac{Z_0}{2\mathit{r\pi }}{\displaystyle \sum _{\upsilon =1}^N}\left\{\left({\overrightarrow{\mathrm{\Delta }}}_{\mathrm{v}}\times \overrightarrow{r}\right)\times \overrightarrow{r}\right\}\cdot {\mathrm{I}}_{\mathrm{\upsilon }}\cdot \mathit{exp}\left(j\cdot \beta \cdot {\overrightarrow{p}}_{\upsilon }+{\psi }_{\upsilon }\right)$$

Based on FIG. 2 should the relationship between current traversed, arbitrarily oriented in space electrically very short conductor elements or antenna conductor sections of length Δ1 ... Δ5 <λ / 20 and generated in the remote point P complex electric and magnetic field strength vectors $\underset{}{\overrightarrow{e}}$

and $\underset{}{\overrightarrow{H}}$

be explained. The electrically very short conductor elements are as vectors Δ ₁ ... Δ ₅ , whose direction is given both by the direction of the position in space and by the Zählpfeilrichtung of the current flowing on the conductor element current, which can be regarded as constant in magnitude and phase. The coordinate directions of the spatial coordinate system are denoted by x, y and z, its coordinate origin by B. In a general description of the vth conductor element with the complex current Iv and its by the position vector p _ν described position in space can be its contribution to the complex electric field strength vector ${\underset{}{\overrightarrow{e}}}_y$

in far-field point of view P, distant from the origin B of the coordinate system by a distance r _A , whose position is further divided by the unit direction vector r described. If there are N such conductor elements, then the electric field strength is summed up: $${\overrightarrow{\underset{}{e}}}_v=\mathit{JEXP}\left(-{\mathit{j\beta r}}_A\right)\cdot \frac{{\mathrm{<figure-callout\; id="0"\; label="Z"\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">Z</figure-callout>}}_0}{2\mathit{r\pi }}{\displaystyle \underset{\upsilon =1}{\overset{N}{\Sigma }}}\left\{\left({\overrightarrow{\mathrm{\Delta }}}_{\mathrm{v}}\times \overrightarrow{r}\right)\times \overrightarrow{r}\right\}\cdot {\mathrm{I}}_{\mathrm{\upsilon }}\cdot \mathit{exp}\left(j\cdot \beta \cdot {\overrightarrow{p}}_{\upsilon }+{\psi }_{\upsilon }\right)$$

zusammen, so lässt sich die Zeitfunktion der elektrischen Feldstärke bei willkürlich gewählter Grundphase wie folgt angeben: $${\overrightarrow{\underline{E}}}_v=c\cdot {\displaystyle \sum _{v=1}^N}{\mathrm{I}}_{\mathrm{\nu }}\cdot \left\{\left({\mathrm{\Delta }}_v\times \overrightarrow{r}\right)\times \overrightarrow{r}\right\}\cdot \cos \left(\mathit{wt}+\beta \cdot {\overrightarrow{p}}_v\cdot \overrightarrow{r}+{\psi }_v\right)$$

Herein: I _ν the current amplitude and ψ _ν the current phase of the ν-th conductor element; λ the wavelength; β = 2π / λ; Z _{0 is} the characteristic impedance of free space.
If one considers the factors, which act the same for all ladder elements, as a constant $$c=j\cdot e^{-j\cdot \beta \cdot r_A}\cdot \frac{{\mathrm{<figure-callout\; id="0"\; label="Z"\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">Z</figure-callout>}}_0}{2r_A\mathrm{\lambda }}$$

together, the time function of the electric field strength with arbitrarily chosen basic phase can be specified as follows: $${\overrightarrow{\underset{}{e}}}_v=c\cdot {\displaystyle \underset{v=1}{\overset{N}{\Sigma }}}{\mathrm{I}}_{\mathrm{\nu }}\cdot \left\{\left({\mathrm{\Delta }}_v\times \overrightarrow{r}\right)\times \overrightarrow{r}\right\}\cdot \cos \left(\mathit{wt}+\beta \cdot {\overrightarrow{p}}_v\cdot \overrightarrow{r}+{\psi }_v\right)$$

w is the angular frequency and t is the time parameter.
In equation (3), the expression in the curly bracket represents the spatial direction of the contribution of a conductor element to the resulting spatial direction of the resulting electric field strength vector.
Describing the vector Δ _ν by its components Δ _ν x, Δ y _{ν, ν} Δ z, then the direction vector of the v-th conductor element can be in the curly brackets indicate how follows: $$\widetilde{{\mathit{RV}}_v}=\left[\begin{array}{c}\mathrm{\Delta }{x}_v\\ \mathrm{\Delta }{y}_v\\ \mathrm{\Delta }{z}_v\end{array}\right]\times \left[\begin{array}{c}\mathrm{sin\theta }\cdot \cos \\ \mathrm{sin\theta }\cdot \mathrm{sin\phi }\\ \cos \end{array}\right]\times \left[\begin{array}{c}\mathrm{sin\theta }\cdot \cos \\ \mathrm{sin\theta }\cdot \mathrm{sin\phi }\\ \cos \end{array}\right]$$

In this case, θ is the elevation angle related to the vertical direction and φ is the azimuthal angle.
Substituted one obtains simplified instead of equation (3): $$\overrightarrow{e}=c\cdot {\displaystyle \underset{v=1}{\overset{N}{\Sigma }}}{\mathrm{I}}_{\mathrm{\nu }}\cdot \left[\begin{array}{c}{\mathit{RVX}}_{\nu }\\ {\mathit{RVY}}_{\nu }\\ {\mathit{RVZ}}_{\nu }\end{array}\right]\cdot \cos \left(\mathit{wt}+\beta \cdot {\overrightarrow{p}}_v\cdot \overrightarrow{r}+{\psi }_v\right)$$

From equation (4) shows that for different and randomly oriented conductor elements each having different components RVX _{ν, ν} RVY, RVZ _ν arise and these components contribute to a vibration having a different phase and amplitude to the total field strength. This turns the direction of the entire electric field strength vector e time-dependent in the point of view P The field strength vector thus oscillates over one period of the high frequency oscillation in the general case not along a line, as would be necessary to achieve the object of the invention.

In the following, antennas are presented, which partially solve the problem of the invention.

In the simplest form of the antenna, fictitious, equally long conductor elements can be arranged along a straight line and connected to one another in a conductive manner, so that essentially a rod-shaped conductor is formed and an interruption of the rod-shaped conductor forms an antenna connection point. Linear conductors have the property that all conductor elements have a same directional vector whose components in the x, y and z direction are in a common relationship to all the conductor elements. Thus, the expression in the curly brace in equation (5) can be subtracted from the summation, and the sum term alone leaves the superimposition of a series of frequency equal but different in amplitude and phase. This results in a resulting vibration, which with the following components of the E vector. $$\overrightarrow{e}=c\cdot \left[\begin{array}{c}{\mathit{RVX}}_{\nu }\\ {\mathit{RVY}}_{\nu }\\ {\mathit{RVZ}}_{\nu }\end{array}\right]{\displaystyle \underset{v=1}{\overset{N}{\Sigma }}}{\mathrm{I}}_{\mathrm{\nu }}\cdot \cos \left(\mathit{wt}+\beta \cdot {\overrightarrow{p}}_v\cdot \overrightarrow{r}+{\psi }_v\right)$$

in allen Raumrichtungen die gleiche Phase. Der elektrische Feldstärkevektor ist somit in jedem Punkt des Raumes zu jedem Zeitpunkt längs einer für diesen Punkt des Raumes spezifischen, feststehenden geraden Linie polarisiert, deren Raumrichtung durch den Richtungsvektor RV _v = RV gegeben ist.Thus have the vibration components of the electric field strength vector $\underset{}{\tilde{e}}$

the same phase in all spatial directions. The electric field strength vector is thus polarized in each point of the space at any time along a fixed straight line specific to that point of the space, its spatial direction through the direction vector RV _v = RV given is.

In particular, for the satellite broadcast reception in vehicles antennas are used with azimuthal omnidirectional, which are mounted on the electrically conductive vehicle skin. As follows from the hand Fig. 3a and 3b For this purpose, a substantially rod-shaped conductor 4 can be mounted substantially perpendicularly over a substantially horizontal, electrically conductive base surface 6. For the conductor elements or antenna conductor sections on the mirror image of the antenna formed vertically above the conductive base 6, the same spatial direction applies as for the antenna itself. This results in the desired for mobile reception Rundstrahl properties of the antenna. However, if the rod-shaped conductor 4 is inclined relative to the vertical line 2 on the base surface 6, this forms together with its mirror image a V-shaped antenna. Thus, not all conductor elements are oriented in the same direction and the object of the invention is not solved. It is therefore essential according to the invention that the deviation of the antenna from the vertical line on the base surface 6 is as small as possible.

In particular for the reception of geostationary satellites whose signals occur in northern latitudes with comparatively low elevation, it is provided that the conductors 4 forming a substantially perpendicular monopole 7 contain at least one interruption point 5 which connects at least one dummy element 8 to the design of the vertical diagram or bridged. In this way, the vertical diagram can be advantageously adapted to the requirements. In FIG. 3a an antenna junction 3 is formed at the base of the monopole 7, wherein the optimum reception in the range of the elevation angle between 25 ° and 65 °, as in FIG. 3b can be seen, is designed and it is the total length of the monopoly 7 about h2 = 5/8 λ designed to be received satellite signals. For this purpose, the interruption point 5 is mounted in the height of about h1 = 3/8 λ to 4/8 λ above the conductive base 6 and connected with an inductive resistance of about 200 ohms at the intended frequency f _m .

Vehicle antennas are often designed as combination antennas for multiple radio services. In particular, for the reception of AM / FM broadcast signals longer antennas are required. An antenna like in FIG. 3 with the height h2 can be advantageously extended to an AM / FM rod antenna with the total height hg, as in FIG. 4 is shown. In order to avoid the influence of the rod above the satellite receiving antenna on the radiation characteristics, a further interruption point 5 is provided at the upper end of the satellite receiving antenna, which is connected to a high impedance reactance, for example, with a parallel resonant circuit 39, the resonant frequency f _r on the center frequency f _{m of} the satellite frequency bands is tuned. A further interruption point 5 is connected at a distance 40, which is preferably smaller than 1 / 5λ to further secure the radiation characteristic also with a high-impedance reactance 39 connected. Already above the first parallel resonant circuit 39, the extension 32 of the rod antenna can be made largely free and in particular contain such series elements, which are high impedance at the satellite frequency.

In contrast to the previously presented antennas which are formed from a straight-line conductor or a plurality of mutually parallel rectilinear conductors, in the following more complex antenna structures according to the invention are to be considered.

In order to discuss the conditions required for this, FIG. 2 the vectors Δ ₁ and Δ _{2 of} the two equally long, very short conductor elements Δ ₁ = Δ ₂ considered, which are aligned parallel to each other and are symmetrically positioned with respect to the origin B of the coordinate system, so that the two position vectors p ₁ and p _{2 are} negatively equal to each other, ie p ₁ = - p ₂ and also the phase angles ψ ₁ and ψ _{2 are} negatively equal, ie ψ ₁ = -ψ ₂ . Due to the parallelism of the two conductor elements Δ ₁ and Δ ₂ applies Δ ₁ = Δ ₂ . This also applies to the two same direction vectors, that is to say: RV _1-2 = RV ₁ = RV ₂ . The contribution e _{1-2 of} the two current-carrying conductor elements to the electric field strength vector in the far point P is therefore according to equation (5): $${\overrightarrow{e}}_{1-2}=c\cdot {\mathrm{I}}_1\cdot \left[\begin{array}{c}{\mathit{<figure-callout\; id="1"\; label="RVX"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVY"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVZ"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_1\end{array}\right]\cdot \left[\cos \left(\mathit{wt}+\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)+\cos \left(\mathrm{wt}-\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)\right]$$

It follows immediately: $${\overrightarrow{e}}_{1-2}=c\cdot {\mathrm{I}}_1\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)\left[\begin{array}{c}{\mathit{<figure-callout\; id="1"\; label="RVX"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVY"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVZ"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_1\end{array}\right]\cdot \cos \left(\mathit{wt}\right)$$

enthalten ist. Bei willkürlicher Zuweisung der Nullphase für den Bezugspunkt - hier dem Ursprung des Koordinatensystems - ist die Cosinus-Schwingung in Gleichung (8) ohne Phasenverschiebung. Alle Komponenten des elektrischen Feldstärkevektors E _1-2 besitzen die gleiche Phase und die erfindungsgemäße Forderung der Polarisation ist erfüllt. Stellt man eine analoge Überlegung für das willkürlich orientierte Paar der in Fig. 2 dargestellten Leiterelemente Δ₃ = Δ₄ mit den Stromamplituden I₃ = I₄ mit den Phasenbeziehungen der Ströme ψ₃ = -ψ₄ an, so lautet der von diesem Part der Leiterelemente erzeugte Beitrag zur elektrischen Feldstärke in Analogie zu Gleichung (8), wie folgt: $${\overrightarrow{E}}_{3-4}=c\cdot {\mathrm{I}}_3\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_3\cdot \overrightarrow{r}+{\psi }_3\right)\left[\begin{array}{c}{\mathit{RVx}}_v\\ {\mathit{RVy}}_v\\ {\mathit{RVz}}_v\end{array}\right]\cdot \cos \left(\mathit{wt}\right)$$

From equation (8), for the ladder elements Δ ₁ and Δ ₂ , the phase of the cosine oscillations in equation (7) resulting from the internal product of the position vector p ₁ with the current phase ψ ₁ , due to the pairing symmetrical to the origin of the coordinate system both spatially and in terms of current phases now exclusively in the amplitude factor $$c\cdot {\mathrm{I}}_1\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)$$

is included. With arbitrary assignment of the zero phase for the reference point - here the origin of the coordinate system - the cosine oscillation in equation (8) is without phase shift. All components of the electric field strength vector e _1-2 have the same phase and the polarization requirement according to the invention is fulfilled. Imagine an analogous consideration for the arbitrarily oriented couple in Fig. 2 The conductor elements Δ ₃ = Δ ₄ with the current amplitudes I ₃ = I ₄ with the phase relationships of the currents ψ ₃ = -ψ ₄ indicate that the contribution to the electric field strength generated by this part of the conductor elements is analogous to equation (8) follows: $${\overrightarrow{e}}_{3-4}=c\cdot {\mathrm{I}}_3\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_3\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="3"\; label="\psi "\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_3\right)\left[\begin{array}{c}{\mathit{RVX}}_v\\ {\mathit{RVY}}_v\\ {\mathit{RVZ}}_v\end{array}\right]\cdot \cos \left(\mathit{wt}\right)$$

By superimposing the field strength contribution generated by both pairs of the conductor elements results: $${\overrightarrow{e}}_{1-2}+{\overrightarrow{e}}_{3-4}=c\cdot \left\{{\mathrm{I}}_1\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)\left[\begin{array}{c}{\mathit{<figure-callout\; id="1"\; label="RVX"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVY"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVZ"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_1\end{array}\right]+{\mathrm{I}}_3\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_3\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="3"\; label="\psi "\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_3\right)\left[\begin{array}{c}{\mathit{<figure-callout\; id="3"\; label="RVX"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_3\\ {\mathit{<figure-callout\; id="3"\; label="RVY"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_3\\ {\mathit{<figure-callout\; id="3"\; label="RVZ"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_3\end{array}\right]\right\}\cdot \cos \left(\mathit{wt}\right)$$

ergibt sich an Stelle von Gleichung (10) $${\overrightarrow{E}}_{1-2}+{\overrightarrow{E}}_{3-4}=c\cdot \left[\begin{array}{c}\mathit{SVx}\\ \mathit{SVy}\\ \mathit{SVz}\end{array}\right]\cdot \cos \left(\mathit{wt}\right)$$

The two directional vectors RV ₁ and RV ₃ of each arbitrarily oriented in space paired conductor elements are thus each with a factor of the current amplitude, the position vector p as well as the current phase ψ contains, weighted and summed. With the resulting sum vector SV : $$\overrightarrow{\mathit{SV}}=\{I_1\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_1\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="1"\; label="\psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_1\right)\left[\begin{array}{c}{\mathit{<figure-callout\; id="1"\; label="RVX"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVY"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_1\\ {\mathit{<figure-callout\; id="1"\; label="RVZ"\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_1\end{array}\right]+I_3\cdot \cos \left(\beta \cdot {\overrightarrow{p}}_3\cdot \overrightarrow{r}+{\mathrm{<figure-callout\; id="3"\; label="\psi "\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\psi </figure-callout>}}_3\right)\left[\begin{array}{c}{\mathit{<figure-callout\; id="3"\; label="RVX"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVX</figure-callout>}}_3\\ {\mathit{<figure-callout\; id="3"\; label="RVY"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVY</figure-callout>}}_3\\ {\mathit{<figure-callout\; id="3"\; label="RVZ"\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">RVZ</figure-callout>}}_3\end{array}\right]=\left[\begin{array}{c}\mathit{SVx}\\ \mathit{SVy}\\ \mathit{SVZ}\end{array}\right]$$

results in place of equation (10) $${\overrightarrow{e}}_{1-2}+{\overrightarrow{e}}_{3-4}=c\cdot \left[\begin{array}{c}\mathit{SVx}\\ \mathit{SVy}\\ \mathit{SVZ}\end{array}\right]\cdot \cos \left(\mathit{wt}\right)$$

Gilt dieser Zusammenhang für alle paarigen Leiterelemente, wie zum Beispiel den paarigen Leiterelementen Δ₃ und Δ₄, so gilt analog: $${\mathrm{\Psi }}_3={\mathrm{\Psi }}_0+\mathrm{\Delta }{\mathrm{\Psi }}_{34}\mathrm{und}{\mathrm{\Psi }}_4={\mathrm{\Psi }}_0-\mathrm{\Delta }{\mathrm{\Psi }}_{34},\mathrm{sodass\; gilt}:\left({\mathrm{\Psi }}_3+{\mathrm{\Psi }}_4\right)/2={\mathrm{\Psi }}_0\mathrm{usw}\mathrm{.}$$

The direction of the sum vector SV thus results not only from the directions of the two direction vectors of the paired conductor elements Δ ₁ , Δ ₂ , but also from their complex currents and is the ratio of the components SVx, SVy, SVz of the sum vector SV certainly. Each of these components changes in phase over the period of the cosine oscillation, so that according to the invention the polarization of the electric field strength vector takes place strictly along a line at any point in time. Naturally, this line is always perpendicular to the unit direction vector r oriented, but otherwise can take any direction. A component of the electric field strength perpendicular to this line does not exist at any time. This consideration can be extended to the superposition of any number of arbitrarily oriented in space paired conductor elements Δ _{ν of} this kind without changing the previous statements. For a more general representation, a common reference phase ψ ₀ is now introduced for the current phases of all the conductor elements and required that for the current phases of the paired conductor elements - eg ψ ₁ and ψ ₂ - that these by the same value Δψ ₁₂ but with different Signs deviate from this reference phase, that is: $${\mathrm{<figure-callout\; id="1"\; label="\Psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_1={\mathrm{<figure-callout\; id="0"\; label="\Psi "\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_0+\mathrm{\Delta }{\mathrm{\Psi }}_{12}\mathrm{and}{\mathrm{<figure-callout\; id="2"\; label="\Psi "\; filenames="imgf0003.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_2={\mathrm{\Psi }}_0-\mathrm{\Delta }{\mathrm{\Psi }}_{12}.\mathrm{so\; that\; applies}:\left({\mathrm{<figure-callout\; id="1"\; label="\Psi "\; filenames="imgf0005.png,imgf0013.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_1+{\mathrm{<figure-callout\; id="2"\; label="\Psi "\; filenames="imgf0003.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_2\right)/2={\mathrm{<figure-callout\; id="0"\; label="\Psi "\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_0$$

If this relationship applies to all paired conductor elements, such as the paired conductor elements Δ ₃ and Δ ₄ , the following applies analogously: $${\mathrm{<figure-callout\; id="3"\; label="\Psi "\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_3={\mathrm{<figure-callout\; id="0"\; label="\Psi "\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_0+\mathrm{\Delta }{\mathrm{\Psi }}_{34}\mathrm{and}{\mathrm{\Psi }}_4={\mathrm{\Psi }}_0-\mathrm{\Delta }{\mathrm{\Psi }}_{34}.\mathrm{so\; that\; applies}:\left({\mathrm{<figure-callout\; id="3"\; label="\Psi "\; filenames="imgf0003.png,imgf0004.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_3+{\mathrm{\Psi }}_4\right)/2={\mathrm{<figure-callout\; id="0"\; label="\Psi "\; filenames="imgf0003.png,imgf0007.png"\; state="\{\{state\}\}">\Psi </figure-callout>}}_0\mathrm{etc}\mathrm{,}$$

Under this condition, the field contributions of all conductor element pairs in equation (11) have the same basic phase ψ ₀ . Naturally, the choice of the basic phase of the time function ψ _{0 has} no influence on the sum vector SV ,

Thus, it can be summarized that an antenna consisting of a plurality of symmetrically to a common reference point B in space in the manner indicated in pairs and arranged identically aligned electrically very short conductor elements Δ ₁ , Δ ₂ and Δ ₃ , Δ ₄ , etc. as in FIG. 2 is achieved, it is achieved that - caused by the excitation of the antenna at the antenna connection point 3 - this pair act as radiating elementary antennas Δ _n , Δ _m and in both belonging to a pair of elementary antennas elemental antennas, eg. B. Δ ₁ , Δ ₂ in FIG. 2 flowing current is equal in magnitude and the spatial reference point for all elementary antenna pairs Δ _n , Δ _m forms a common phase center B in such a way that the arithmetic mean of the phases of the two, counted in the same direction currents of a pair of elementary antenna for all dipole pairs Δ _n , Δ _m .... has the same value (ψ0).

Electrically short antennas, which are antennas whose dimensions are <3/8 λ, have the property that the currents on these antennas have virtually constant phases over their extent. Thus, as follows, for. B. on hand of Fig. 5a and Fig. 5b is explained, by conductive juxtaposition of electrically very short conductor elements or antenna conductor sections about a common reference point, a loop antenna 14 - with an opening formed by an interruption of the loop 14 antenna connection point 3 - are formed. If the dimensions of the loop 14 are electrically sufficiently small so that the ring current is in magnitude at any point is equal to, exists for each very short conductor element Δ _n a pair of forming corresponding very short conductor element Δ _m such that the above conditions to the Apply loop 14. Such a loop 14 may, for example, be designed as a regular n-gon with the phase reference point B in the symmetry point of the n-gon. In another example, the loop antenna 14 is formed of a plurality of closed loops with a common phase reference point, but in one of the loops, the antenna junction 3 is formed by interruption. In a further advantageous embodiment, the loop antenna 14 consists of a plurality of loops arranged in a row, which are essentially in mutually parallel planes with the smallest possible distance from each other in the form of a coil or

Spiral are arranged. In this case, a substantially common central phase reference point is formed for all loops and the antenna connection point 3 is provided by the two ends of the coil.

In a particularly advantageous embodiment, as for example in the FIGS. 5a and 5b is shown, the loop antenna 14 is not electrically short and contains for effective electrical shortening several introduced at points of interruption 5 capacitors or capacitors. As a result, the constancy of the current is given by amount and phase on the conductor elements sufficiently.

Fig. 5a shows a circular loop antenna 14 with radius R, which may also be designed polygonal. At its center is the phase center B. The structure is subdivided into "z" line sections, each with the length Δs. The total orbital length is S. The antenna acts as a loop antenna with dimensions in the range of the wavelength, wherein nevertheless a homogeneous current distribution is achieved by dividing the structure and inserting capacitances 16 according to the invention. As a result, the length of the antenna is electrically shortened and creates a homogeneous, horizontally polarized electromagnetic field all around. In contrast to the one-dimensional structures described above, the loop is two-dimensional. According to the invention, for each of the electrically very short conductor elements Δ ₁ , Δ ₂ ,..., Which act as elemental antennas, there is a corresponding, equally aligned, very short conductor element, which is current-flowed in the opposite direction, so that the pair formation described above relates is given to the phase center B in the center. In FIG. 5a are exemplary two-pair electrically very short conductor elements, as vectors Δ ₁ , Δ ₂ , whose direction is given both by the direction of their position in space and by the counting arrow direction of the current flowing on the conductor element current, which can be regarded as constant in magnitude and phase.

In FIG. 5b the loop antenna 14 is arranged at a constant height h above the conductive base 6. Due to the reflection on the base 6 lies the common phase center B now on the base 6. By way of example again two paired electrically very short conductor elements, as vectors Δ ₁ , Δ ₂ , whose direction is given both by the direction of their position in space and by the counting arrow direction of the current flowing on the conductor element current, which can be regarded as constant in magnitude and phase. For each conductor element of the loop antenna 14 thus exists a corresponding paired conductor element on the virtual mirror image of the loop antenna 14. The vertical main beam direction can be adjusted by the choice of the height h and the radius of the line ring. It can be achieved a zero point in the vertical direction and in the horizontal direction.

The annular circumferential conductor length S is according to the invention again divided into z equal pieces of length Δs = S / z. The conductor characteristic impedance of the as shown in FIG. 5c circulating line over the conductive base 6 is Zw. The capacitive reactance ΔX per line section Δs and thus the capacitance value C = 1 / (ω * ΔX) to be inserted into this conductor section is defined assuming an elongated length Δs and for an approximately annular line with a large radius R of the annular loop antenna 14 with respect to the conductor height h by $$\mathrm{.DELTA.X}\mathrm{/}\mathrm{.DELTA.w}\mathrm{=}\tan \left(\mathrm{2}\mathrm{\pi \; \Delta s}\mathrm{/}\mathrm{\lambda }\right)\mathrm{,}$$

This results in a good approximation for the capacitance value C to be inserted into the line section Δs: $$\mathrm{C}\mathrm{=}\mathrm{1}\mathrm{/}\left(\mathrm{\omega }\mathrm{•}\mathrm{<figure-callout\; id="2"\; label="Twan"\; filenames="imgf0003.png"\; state="\{\{state\}\}">Twan</figure-callout>}\left(\mathrm{2}\mathrm{\pi \Delta s}\mathrm{/}\mathrm{\lambda }\right)\right)$$

Angular frequency of the satellite signals = w; Free space wavelength of the satellite signals = λ

In order to obtain a round diagram to a good approximation, the line of length S is to be divided into a sufficient number of sections by insertion of capacitances 16. For a meaningful subdivision: Δs / λ <1/8. If the sections .DELTA.s = S / z are chosen to be sufficiently small, the equality .DELTA.s of all sections is not absolutely necessary as long as a capacitance 16 is inserted only after each section, whose value is calculated according to the above-described criterion from the relative length Δs / λ of the relevant section.
As an example for the design of the reception in the range of an elevation angle between 25 ° and 65 ° with azimuthal omnidirectional characteristic, a horizontally arranged loop antenna 14 is placed at a distance of about 1/16 of the wavelength above the conductive base 6, as exemplified in FIG FIG. 5b is shown. The diameter of the loop antenna 14 is chosen to be slightly larger than 1/4 of the wavelength. Along the conductor guide is at intervals of about 1/8 of the wavelength each with a capacity 16 with a reactance of about -200 ohms connected interruption point 5 is introduced.

In FIG. 7 For example, the vertical diagram of such an antenna is shown for a) left-hand rotating circular polarization and b) right-hand rotating circular polarization. A possible small residual imbalance can be reduced by refinement of the circuit according to the above specifications with reactances and perfection of the symmetry of the antenna with respect to the antenna connection point 3. For the example of an annular loop antenna 14 in the frequency range around 1500 MHz, a radius R of about 4 cm, a height h of about 18 mm and a conductor diameter D of about 3 mm have to be favorable for realizing both the vertical directional pattern and a matching conductor characteristic impedance Zw proved.

FIG. 6 shows a further advantageous embodiment of a loop antenna 14 according to the invention with coupling 18 at the antenna connection point 3 via a symmetrical two-wire line 26 outside the center Z, a Umsymmetrierglied 29 and a matching network 25. The influence of not in the phase center located symmetrical vertical feed line in the form of symmetrical Two-wire line 26 does not reduce the polarization purity due to the symmetry property explained below. The connection of the one connection on the unbalanced side of the Umsymmetrierglieds 29 to the connection point 28 of the antenna arrangement is advantageously carried out using a guided over the conductive base 6 microstrip line 30. The other terminal on the unbalanced side of the Umsymmetrierglied 29 is connected to the electrically conductive base 6. Due to the Symmetry properties of the two-wire line 26 compensate for the effects of the currents flowing toward one another in the opposite direction on the conductors of the two-wire line 26, so that these also do not influence the radiation properties of the loop antenna 14. As will be explained below, the currents generated by the electromagnetic reception field on these conductors also have no influence on the effects at the antenna connection point 3.

An electrical conductor, which in a plane perpendicular to the base 6 and symmetrically with respect to the antenna connection point 3 oriented symmetry plane SE of the satellite antenna array, for example, as a planar or linear antenna 24 - FIG. 16 - Is guided, due to the symmetry of the antenna connection point 3 without influence on the operation of the satellite antenna. The effect of the currents produced by the electromagnetic reception field in the antenna 24 cancel each other out with respect to their effect at the antenna connection point 3. This also applies to the two electrical conductors of the two-wire line 26 in FIG. 6 to, which can be considered as guided in the plane of symmetry SE due to the small distance between the two conductors. From this the antenna 24 in FIG. 16 and the antenna connection point 3 decoupling property is used in an advantageous embodiment of the invention in the design of combination antennas for different radio services. Such an antenna can thus in addition to the satellite reception by arranging one or more separate and guided in the plane of symmetry SE antennas such. As the antenna 24 - for radio services such as AM / FM reception, cellular radio services, etc. are used.

When in the FIG. 8 illustrated advantageous embodiment of the loop antenna 14, the coupling takes place centrally and at the ring level. The matching network 25 and the Umsymmetrierglied 29 are also arranged on the ring plane. The two-wire line 26 is connected to the unbalanced side of the Umsymmetrierglied 29 and guided in the center Z to the base 6. There is their first conductor with the conductive base 6 and its second conductor with the guided over the base plate 6 microstrip conductor 30th connected. The latter establishes the connection to the connection point 28 of the antenna arrangement. Here, too, the effects of the currents flowing in the opposite direction compensate each other on the conductors of the two-wire line 26 so that they do not influence the radiation properties of the loop antenna 14.

In the event that the satellite broadcasting system is additionally supported by the regional radiation of vertically polarized terrestrial signals in another, closely adjacent in frequency frequency band of the same bandwidth, it is desirable to fill the vertical radiation pattern for these signals at low elevation angles out. This allows the antenna to compromise receiving both the satellite receive signals and the terrestrial signals. To achieve this, in an advantageous embodiment of the invention in the central phase reference point B of the loop antenna 14 in FIG. 9 an electrically short, vertically oriented monopole 7 attached. Furthermore, as a distribution or coupling network acting in reciprocal transmission case as a power distributor Leistungskopplungs- and phase shifter network 31 is provided to the separate terminals via the loop antenna 14 on the one hand and the monopoly 7 on the other is connected, and which is designed in such a way that in the reciprocal transmission case the phases of the currents flowing in the monopole 7 and in the loop antenna 14 are the same. Taking into account the reflection are due to the phase condition of the currents to the loop antenna 14 and the monopole antenna 7 with respect to the phase center B on the base 6, the above conditions required for the formation of paired conductor members Δ _n, Δ _m, thus meeting for the polarization of the electric field strength. The main beam direction in the vertical diagram of the loop antenna 14 is thereby drawn by adding the vertical monopole 7 to low elevation. The combination now allows to receive a vertical polarized electric field for additional terrestrial applications even at lower elevation. By varying the weighting of the superimposition of the two antennas, the vertical directional pattern can be filled to low elevation angles for these signals. Trained as a rod antenna monopoly 7 has in its vertical directional characteristic a similar main beam direction as the horizontal polarized loop antenna 14, but provides for low elevation angle a larger contribution than the loop antenna 14. With the aid of the unbalanced power coupling and phase shifter network 31, both the weighting of the properties of the two antennas can be set differently and additionally the alignment of the phase centers.

When arranged in FIG. 10 is the monopoly 7 different than the rod antenna in FIG. 9 realized. The vertical, provided for feeding the loop antenna 14 two-wire line 26 is utilized as a monopole 7, wherein the loop antenna 14 serves as a roof capacitor 12 of the monopoly 7. For this purpose, an additional coupling is provided, wherein the loop antenna 14 is used in a mode as a roofing capacity 12 of the monopole 7 for a vertically polarized field with. If necessary, a matching network 33 for the monopole mode is used, which is preferably designed such that the above power coupling and phase shifting network 31 can be connected thereto. Thus, here too, the weighting of the antennas can be adjusted differently with the aid of this asymmetrical power coupling and phase shifting network 31, and the phase center of gravity can be adjusted. The matching of the impedance of the loop antenna 14 can be done by means of the matching network 25, which can be realized in a simple embodiment as λ / 4-line transformer. Due to the vertically polarized receiving two-wire line 26 with the loop antenna 14 as a roof capacitance 12 relative to the base 6 and due to the horizontally polarized receiving loop antenna 14 between the two conductors of the two-wire line 26 signals from vertical and horizontal field components in acting in the transmission case as a power divider Leistungskopplungs- and Phase shifter network 31 superimposed. According to the invention, this property can advantageously be exploited for supporting the radiation properties at low elevation by phase-locked combination of the vertically and horizontally polarized antennas and choosing the same phase center of gravity (in analogy to the phase reference point in the origin of the coordinate system according to the above considerations). This can be a linearly polarized field produce, which at higher Elevation is preferably horizontal and preferably vertically polarized at low elevation.

In an advantageous embodiment of the invention according to FIG. 15 the unbalanced power coupling and phase shifting network 31 in the central base 19 of the antenna arrangement is realized in that one conductor of the two-wire line 26 is conductively connected to the conductive base 6 via a reactance 41 and the other conductor of the two-wire line 26 is led to the connection point 28 of the antenna arrangement , By selecting the reactance 41, the weighting of the reception of the horizontal and vertical polarized electric fields can be adjusted. At the in FIG. 15 As illustrated, the reactance 41 is realized by a capacitor whose size adjusts the desired weighting.

In the FIG. 10 described antenna is in FIG. 11 executed in a symmetrical embodiment with star-shaped multi-arm horizontal feed and central connection to a vertical feed as an alternative to the one-armed, "asymmetrical" feed. In this way the roundness of the azimuthal directional pattern is perfected. The example shows an embodiment with two-armed symmetrical feed to the two antenna connection points 3 formed in the loop antenna 14.

FIG. 12 shows a particularly advantageous two-armed feed via band conductor 34 of a loop antenna 14 and the current paths indicated by arrows. The central vertical feed takes place here, for example, in coaxial design, wherein the outer conductor of a coaxial line 35 is connected to one and the inner conductor to the other band of the strip conductor 34.

In a further advantageous embodiment of the invention as described below with reference to the Fig. 13a and 13b is described, is in each case a group of electrically very short, substantially in a horizontal plane extending conductor elements Δ _ν strung together electrically and thus an electrically short dipole 21 designed with almost the same phase of the currents on the conductor elements, which at one through a point of interruption formed Antenna connection point 3 can be coupled, or can be fed in reciprocal transmission case. In each case symmetrically to the common reference point B, an identically shaped and identically aligned electrically short dipole 21 is correspondingly present, so that for each electrically very short conductor element on the dipole 21, a corresponding corresponding conductor element extending essentially in the same plane exists on a corresponding dipole 21. Both dipoles 21 forming a pair are fed in the reciprocal transmission case at the antenna connection point 3 in each case with the same current in the amount. The arithmetic mean of the phases of the currents of a dipole pair counted in the same direction has the same value for all dipole pairs.

In an advantageous embodiment of the invention, the dipoles 21 are rectilinear and designed to be symmetrical to their antenna connection points 3 and extending in a horizontal plane, the antenna connection points 3 being arranged distributed equidistant on a horizontal circle whose center forms the common reference point B. The dipoles 21 are oriented perpendicular to the connecting line to the center of the circle. As a result, a circular array antenna system is given, as in a simplest form in FIG. 13a is shown. The figure shows a symmetrical embodiment of an antenna according to the invention with 4 arranged in a square dipoles 21 and with a centrally located in the phase center B coupling network 10, the output of which forms the connection point 28 and acts in reciprocal transmission case as a distribution network. The antenna connection points 3 are each connected via an electrical line 27 to one of the inputs 23 of the coupling network 10, wherein the dipole pairs are fed with equal signals according to amplitudes and phases. Adjacent ends of adjacent dipoles 21 may be interconnected via capacitors 16.

FIG. 13b shows a dipole arrangement similar FIG. 13a but with a superposition of the reception of horizontal and vertical electric field components, similar to the Figures 10 and 11 , The dipoles 21 additionally act as roofing capacitance 12 of the vertical monopole 7 thus formed by the two-wire line 26.

Likewise, as in FIG. 13b shown, in an advantageous embodiment of an antenna arrangement according to FIG. 13a the arranged in a square lying dipoles 21 on a conductive base 6 with a central coupling - similar to the antenna in FIG. 10 - be combined with a monopoly. In this arrangement, the vertical feed line in the form of the two-wire line 26 is used to power the dipoles 21 as a monopole 7 with the dipoles 21 as roof capacity 12. Thus, the weighting of the effects of the dipoles 21 and the monopole 7 formed in this way can be adjusted differently according to the requirements and the alignment of the phase centers can be done here with the help of the unbalanced in the reciprocal transmission case acting as a power divider power coupling and phase shifter network 31.

In a further, not shown, advantageous embodiment of the invention are similar to Fig. 13a and 13b , in the central phase reference point B of a circular array antenna system with horizontally oriented dipoles 21, an electrically short vertical monopole 7 and a distribution or coupling network 10 is present. The output 24 of the coupling network 10 is designed as a connection point 28 of the antenna arrangement and the antenna connection points 3 of the antennas in the circle group and the monopole 7 are fed in reciprocal transmission case via an electrical line 27 from the coupling network 10 in such a way that the phases of in the current supplied to the monopole 7 corresponds to the phase position of the currents fed into the circuit group antenna system with respect to the common phase reference point B. Finally, a plurality of electrically short vertical monopoles 7 can be arranged in pairs symmetrically to the central phase reference point and fed in the reciprocal transmission case via the coupling network 10 in such a way that the arithmetic mean of the current phases of the paired monopolies 7 and the phase of the current in the central Monopole 7 fed current relative to the phase reference point B are the same.

In a particularly advantageous embodiment of the invention, the coupling network 10, as in FIG. 14 shown adapted for the use of the antenna as a diversity receiving antenna in such a way that both the reception signals of the antennas with horizontally oriented conductor elements and those of the vertical monopole 7 are each alternatively alternatively available. This is done in the simplest case with the aid of a diversity switch 37, which is controlled by a diversity module 38. In this case, the received signals of both antennas are each received with their own - but same for both directions of the circular polarization - radiation characteristic.

In particular, in the automotive industry, the compatible expansion of simple devices to particularly powerful and thus more complex devices in an economical way is particularly important. A particular advantage of an antenna arrangement according to the invention is the possibility of a substantially horizontally polarized antenna and a substantially vertically polarized antenna combine to achieve separate connections for circularly polarized waves in both directions of rotation. Thus, for example, the loop antenna 14 with the vertical monopole 7 with common phase center B in FIG. 9 either combined in a low-cost manner using the power coupling and phase shifting network 31, as related to FIG. 9 or the antenna terminals 3 of the two antennas are combined in a more elaborate form by means of a 90 ° phase circuit with different signs so as to be available at an LHCP terminal 46 and a RHCP terminal 47 separated by LHCP and RHCP waves, respectively. It is particularly advantageous that in existing basic form of the loop antenna 14 and the monopole 7 combined construction of the antenna arrangement, both the low-cost operating form to achieve the object of the invention as well as the extension for separate representation for the operation for LHCPbeziehungsweise RHCP waves can be realized economically.

Antennas for circularly polarized waves are usually realized in the prior art by interconnecting similar antennas - such as two crossed dipoles or two crossed loop antennas - via 90 ° phase switching. In contrast, in the present case - as in Figure 17a illustrated a circularly polarized antenna of two different antennas according to the present invention, whose vertical directional patterns are congruent and their main direction is designed to receive the satellite signals suitable. The equality of the directional diagrams, for example, by choosing the structure of the trained as a rod antenna monopole 7 with dummy element 8 - similar to in connection with the in FIG. 3 described antenna - as well as by appropriate design of the loop antenna 14 - as related to FIG. 7 described - realized. The equality of the phase center B of both antennas can be accomplished by means of the matching network 25 for the loop antenna 14 and the matching network for the monopole mode, respectively. For the realization of such an antenna can - as in Figure 17a shown - the vertically and horizontally polarized antenna 7 and 14 at the common phase center B as in FIG. 9 , but with separate supply of the signals to the terminal for vertical polarization 49 or for connection for horizontal polarization 48 of a hybrid coupler 45 with 90 ° positive or negative phase difference with respect to the LHCP terminal 46 and the RHCP terminal 47 for the separate generation of LHCP or RHCP signals done.

A similar antenna arrangement is in FIG. 17b however, the realization of the monopole 7 is similar to the antenna arrangement in FIG FIG. 10 by the combination of acting as a roof capacitance loop antenna 14 and the two-wire line 26 takes place. By means of a combined matching circuit 50, both the adaptation of the loop antenna 14 and the adjustment of the monopole 7 as well as the adjustment of a common phase center B are ensured.

In a further advantageous antenna arrangement for the alternative decoupling of RHCP or LHCP signals, as in FIG. 19a shown, a loop antenna 14 - as in FIG. 11 - With two opposing antenna connection points 3 and connected thereto and located in the loop level matching networks 25, which are preferably implemented as λ / 4 transformation lines provided. The outputs of the matching networks 25 are connected in parallel in addition. The received signal is fed via the two-wire line 26 to a matching network 25 located on the base area 6, the output of which is in turn connected to one of the two inputs of a signal combination circuit designed in particular as a 90 ° hybrid coupler 45. At the antenna connection point 3 in the base of the located in the center of the arrangement, designed as a rod antenna monopole 7 is also a matching network 25 is connected, the output of the other feeds the two inputs of the 90 ° hybrid coupler 45. An LHCP / RHCP switch 55 connected to the outputs of the 90 ° hybrid coupler 45 alternatively provides satellite reception signals of the two directions of polarization at the connection point 28, driven by a switch located in a radio receiver module 52. When driven by a diversity control module 38, the antenna arrangement can also be advantageously used for polarization diversity by switching between reception for LHCP and RHCP waves.

Also, as in FIG. 19b in a variant of Fig. 19a 4, the axis ratio of the circularly elliptically polarized field is adjusted by introducing an attenuator 56 into the path of the monopole 7 from the loop antenna 14. Increasing attenuation is accompanied by the fact that the main beam direction of the antenna increases in elevation and the antenna can be optimized for optimal interference resistance to horizontally incident noise and temperature outside noise. By supplementing the attenuator 56 in Fig. 19b with a phase shifter (not shown) can be adjusted according to the invention by adjusting the phase with the attenuation of both the ellipticity, the sense of polarization and the elevation of the main beam direction of the antenna. The switch 55 may be omitted if necessary.

In a further particularly economical embodiment of such an antenna with circular or elliptically polarized field with reversible direction of rotation is in FIG. 20 - similar to the antenna in FIG. 11 - The separate monopoly 7 saved. For reception in vertical polarization, the two-wire line 26 is also utilized here. By inserting an appropriately designed network 53 into one of the strands of the vertical two-wire line 26, the difference of 90 ° between the phases of the horizontal field component picked up by the vertical two-wire line 26 with the loop antennas 14 as the roof capacitance 12 and the loop field 14 is set so that their combination with this phase difference is present at the microstrip conductor 30 to the matching network 54 and thus also at the junction 28. Thus, the antenna receives a circularly polarized field. A circuit combining the receive signals of the loop antenna 14 at the output of the matching networks 25 from the horizontally polarized electric field and the receiving signals of the vertical two-wire line 26 from the vertically polarized electric field comprises an LHCP / RHCP switch 55 for reversing the polarity of the receiving voltage of the loop antenna 14. The latter can be added in this way with different signs of the received voltage from the vertically polarized electric field, so that between the reception of LHCP field and RHCP field by switching the LHCP / RHCP switch 55 can be switched.

As already related to the antenna in FIG. 15 explained here can also correspond to the network 31 corresponding network 53 of reactances accordingly FIG. 18 to form the vertical directional pattern of the linearly-vertically polarized antenna into the ground-connected leg of the vertical two-wire line 26. With the aid of the network 53, the adjustment of the common mode to differential ratio on the vertical two-wire line 26 can be set. In contrast to the antennas described above in the Figures 19 and 20 the network 53 is to be designed in such a way that the reception voltages from the horizontal and the vertical electric field components are superimposed in phase. This network 53 can be designed in the simplest case as a capacity. By adjusting the common-mode to differential ratio on the vertical two-wire line 26, the ratio of the low-polarization vertically-polarized field of the main beam direction to the proportion of the higher polarization horizontally-polarized field of the main beam direction can be adjusted. By designing the network 53, the elevation of the main beam direction of the overall characteristic between the elevation angles 0 ° (horizontal) and 45 ° can thus be freely selected.

1. 1. Durch die Gestaltung elektrisch sehr kurzer Leiterelemente Δ₁, Δ₂,... der Antenne 1 ist sichergestellt, dass entsprechend dem zwischen Empfangsantennen und Sendeantennen geltenden Reziprozitätsgesetz bei Einspeisung von Sendeleistung an mindestens einer Antennenanschlussstelle der Antenne 3 der im Fernfeld erzeugte elektrische Feldstärkevektor E _ν in jedem Punkt P des Raumes zu jedem Zeitpunkt längs einer für diesen Punkt P des Raumes spezifischen, feststehenden geraden Linie polarisiert ist.
   Diese Bedingung lässt sich z.B. erfüllen, wenn alle Leiterelemente Δ1, Δ2 längs einer gestreckten Linie 2 angeordnet sind und leitend miteinander verbunden sind, so dass im Wesentlichen ein stabförmiger Leiter 4 gebildet ist und die Antennenanschlussstelle 3 durch eine Unterbrechung des stabförmigen Leiters 4 gebildet ist.
   Der im Wesentlichen stabförmige Leiter 4 ist vorzugsweise im Wesentlichen senkrecht über einer im Wesentlichen horizontalen leitenden Grundfläche 6 angebracht und weist eine Unterbrechungsstelle auf, durch welche die Antennenanschlussstelle 3 gebildet ist. Bevorzugt weist der so gebildete im Wesentlichen senkrechte Monopol 7 zur Gestaltung des Vertikaldiagramms mindestens eine Unterbrechungsstelle 5 auf, welche mit mindestens einem Blindelement 8 beschaltet ist. Die im Fußpunkt des Monopols 7 gebildete Antennenanschlussstelle 3 kann zur Gestaltung des optimalen Empfangs im Bereich eines Elevationswinkels zwischen 25° und 65° die Gesamtlänge h2 des Monopols 7 etwa 5/8 λ der zu empfangenden Satelliten-Signalen beitragen, wobei die Unterbrechungsstelle 5 in einer Höhe h1 von etwa 3/8 λ - 4/8 λ über einer leitenden Grundfläche 6 angebracht und mit einem bei dieser Frequenz induktiven Blindwiderstand 8 von circa 200 Ohm beschaltet ist ( Figur 3 ).
2. 2. Die Leiterelemente Δ₁, Δ₂,... können längs mehrerer zueinander paralleler, gestreckter gerader Linien angeordnet sein, so dass mehre stabförmige Leiter 4 gebildet sind, von denen in mindestens einem die Antennenanschlussstelle 3 ausgebildet ist. Die stabförmigen Leiter 4 können hierbei vertikal über der im Wesentlichen horizontalen leitenden Grundfläche 6 orientiert sein.
   Beispielsweise kann zur Gestaltung eines im Wesentlichen runden azimutalen Richtdiagramms eine Kreisgruppenantennenanlage 9 mit untereinander gleich ausgeführten stabförmigen Leitern 4 als parasitäre Strahler 11 vorgesehen sein, wobei im Zentrum Z der Kreisgruppenantennenanlage 9 eine Antenne gemäß der vorstehenden Ziffer 1 und eine gemäß den Anforderungen an die Rundheit des azimutalen Richtdiagramms hinreichend großen Anzahl von auf einem Kreis im gleichen Winkelabstand W voneinander angeordneten parasitären Strahlern 11 vorgesehen sind.
   Die Kreisgruppenantennenanlage 9 enthält ein Verteilungsnetzwerk bzw. ein Koppelungsnetzwerk mit mehreren Anschlüssen 23, wobei einer (24) der Anschlüsse als Antennenanschlussstelle 3 ausgeführt ist und die stabförmigen und gleichartig ausgeführten in der Kreisgruppe angeordneten stabförmigen Leiter 4 jeweils eine Unterbrechungsstelle 5 enthalten und somit als Strahler 7 ausgebildet sind, welche jeweils über eine gleichartige elektrische Leitung 27 an jeweils einen der übrigen Anschlüsse des Netzwerks 10 angeschlossen sind und im reziproken Sendefall mit gleichen Signalen nach Amplituden und Phasen speisbar sind, wobei der im Zentrum Z der Kreisgruppenantennenanlage 9 befindliche Strahler 7 zur Gestaltung des Richtdiagramms gleichfalls an einen der Anschlüsse des Netzwerks 10 angeschlossen ist und mit einem Signal mit gesonderter Amplitude und Phase speisbar ist. Alternativ kann im Zentrum Z der Kreisgruppe an Stelle des Strahlers 7 auch ein parasitärer Strahler 11 angebracht sein. Auch können die im Kreis angeordneten stabförmigen Leiter 4 zur Gestaltung des Vertikaldiagramms jeweils mindestens eine mit mindestens einem Blindelement 8 beschaltete Unterbrechungsstelle 5 enthalten. Entsprechendes gilt für den im Zentrum Z der Kreisgruppe angeordnete stabförmige Leiter, der zur Gestaltung des Vertikaldiagramms jeweils mindestens eine mit mindestens einem Blindelement 8 beschaltete Unterbrechungsstelle 5 enthalten kann. Zur Gestaltung möglichst niedriger stabförmiger Leiter können diese an ihrem oberen Ende eine Dachkapazität 12 enthalten und dadurch verlängert wirken. Darüber hinaus kann die Kreisgruppenantennenanlage 9 auch aus mehreren in konzentrischen Kreisen angeordneten und im jeweiligen Kreis aus gleichartig ausgeführten stabförmigen Leitern bestehen, die gegebenenfalls nach Betrag und Phase gleich erregt werden.
3. 3. In einer bevorzugten Ausgestaltung besteht die Antenne aus einer Vielzahl von jeweils symmetrisch zu einem gemeinsamen Bezugspunkt im Raum in der angegebenen Weise paarweise angeordneten und gleich ausgerichteten elektrisch sehr kurzen Leiterelementen Δ1, Δ2 bzw. Δ3, Δ4 bzw. Δ5, Δ6, wobei - bewirkt durch die Erregung der Antenne an der Antennenanschlussstelle 3 - diese paarweise als strahlende Elementarantennen Δ_n, Δ_m wirken und zwar so, dass der in beiden zu einem Elementarantennenpaar gehörigen Elementarantennen Δ_n, Δ_m fließende Strom der Größe nach gleich ist und der Bezugspunkt für alle Elementarantennenpaare Δ_n, Δ_m in der Weise ein gemeinsames Phasenzentrum B bilden, dass das arithmetische Mittel der Phasen der beiden, in der jeweils gleichen Richtung gezählten Ströme eines Elementarantennepaares für alle Elementarantennenpaare Δ_n, Δ_m den gleichen Wert besitzt.
   Bevorzugt ist durch leitende Aneinanderreihung von elektrisch sehr kurzen Leiterelementen um den gemeinsamen Bezugspunkt eine Schleifenantenne 14 mit einer an einer Stelle durch Unterbrechung der Schleife gestalteten Antennenanschlussstelle 3 gebildet ist, wobei die Abmessungen der Schleife elektrisch hinreichend klein sind, so dass der Ringstrom dem Betrag nach an jeder Stelle gleich ist und jedes sehr kurze Leiterelement durch ein korrespondierendes sehr kurzes Leiterelement zu einem Paar ergänzt ist. Alle Leiterelemente Δ₁, Δ₂.... verlaufen zweckmäßigerweise in einer Ebene wobei die Schleifenantenne 14 die Form eines regulären n-Ecks, dessen Phasen-Bezugspunkt durch den Symmetriepunkt des n-Ecks gegeben ist, oder die Form eines Kreisrings haben kann, wobei hier Bezugspunkt B durch den Mittelpunkt des Kreisrings gegeben ist. Die Schleifenantenne 14 kann auch aus mehreren geschlossenen Schleifen mit gemeinsamem Phasen-Bezugspunkt B gebildet sein, wobei jedoch in einer der Schleifen durch Unterbrechung die Antennen-anschlussstelle 3 auszubilden ist. Hierbei kann die Schleifenantenne 14 aus mehreren leitend aneinander gereihten Schleifen in im Wesentlichen zueinander parallelen Ebenen mit möglichst geringem Abstand voneinander in Form einer Spule gestaltet sein, so dass für alle Schleifen ein im Wesentlichen gemeinsamer Phasen-Bezugspunkt gebildet ist und die Antennenanschluss-stelle 3 durch die beiden Enden der Spirale gegeben ist.
   Soweit die Schleifenantenne 14 nicht elektrisch klein ist, kann sie zur wirksamen elektrischen Verkürzungen mehrere an Unterbrechungsstellen 5 eingebrachte Kapazitäten 16 enthalten, wodurch die Konstanz des Stromes nach Betrag und Phase auf den Leiterelementen Δ₁, Δ₂ hinreichend sichergestellt wird, ( Figur 5a ). Bevorzugt ist, dass die Schleifenantenne 14 kreisförmig oder angenähert quadratisch in einer Ebene parallel zu einer im Wesentlichen horizontalen leitenden Grundfläche 6 gestaltet ist, und an Unterbrechungsstellen eingebrachte Kapazitäten 16 aufweist, die sowohl die Konstanz des Stromes auf den Leiterelementen Δ₁, Δ₂ als auch das Vertikaldiagramm gestalten.
   Zur Gestaltung des Empfangs im Bereich eines Elevationswinkels zwischen 25° und 65° mit azimutaler Rundcharakteristik ist die Schleifenantenne 14 bevorzugt im Abstand von etwa 1/16 bis 1/8 der Wellenlänge über der leitenden Grundfläche 6 platziert ist, wobei die Seitenlänge der Schleifenantenne 14 etwa 1/4 der Wellenlänge gewählt ist und längs der Leiterführung in Abständen von etwa 1/8 der Wellenlänge jeweils mit einer Kapazität mit einem Blindwiderstand von etwa -200 Ohm beschaltete Unterbrechungsstelle eingebracht ist ( Figur 5b und c).
   In einer bevorzugten Ausgestaltung ist im zentralen Phasen-Bezugspunkt ein elektrisch kurzer vertikaler Monopol 7 sowie ein Verteilungsnetzwerk 10 vorgesehen, dessen Eingang als Antennenanschlussstelle 3 ausgeführt ist und die Schleifenantenne 14 und der Monopol 7 entsprechend dem zwischen Empfangsantennen und Sendeantennen geltenden Reziprozitätsgesetz jeweils über eine elektrische Leitung von einem Ausgang des Verteilungsnetzwerks in der Weise gespeist sind, dass die Phasen des in den Monopol 7 und in die Schleifenantenne eingespeisten Stromes jeweils gleich sind ( Figur 9 ). Das Verteilungsnetzwerk ist hierfür als ein Leistungsteiler- und Phasenschiebernetzwerk 31 mit separaten Anschlüssen für die Schleifenantenne 14 und den Monopol 7 in der Weise gestaltet, dass die Phasen des in den Monopol 7 und in die Schleifenantenne 14 eingespeisten Stromes zur Bildung des gemeinsamen Phasenzentrums B unter Berücksichtigung der Spiegelung an der Grundfläche 6 nahezu gleich sind und, dass die Gewichtung bei der Überlagerung der Wirkungen der Schleifenantenne 14 und des Monopols 7 in der Weise eingestellt ist, dass die Hauptrichtung des resultierenden vertikalen Richtdiagramms zwar für den Satellitenempfang eingestellt ist, dass das Richtdiagramm durch die Wirkung des Monopols 7 jedoch zu niedrigen Elevationswinkeln hin aufgefüllt ist ( Figur 9 ).
4. 4. In einer weiteren bevorzugten Variante ist jeweils eine Gruppe von elektrisch sehr kurzen, im Wesentlichen in einer horizontalen Ebene verlaufenden Leiterelementen Δ₁, Δ₂ elektrisch leitend aneinander gereiht, derart, dass sie mehrere elektrisch kurze Dipole 21 mit nahezu gleicher Phase der Ströme auf den Leiterelementen Δ₁, Δ₂ bilden, die an einer durch eine Unterbrechungsstelle gebildeten Dipol-Anschlussstelle 22 gespeist sind, wobei jeweils symmetrisch zum gemeinsamen Bezugspunkts B ein gleich geformter elektrisch kurzer Dipol 21 korrespondierend vorhanden ist, sodass zu jedem elektrisch sehr kurzen Leiterelement Δ₁ auf einem Dipol ein entsprechend korrespondierendes, im Wesentlichen in derselben Ebene verlaufendes Leiterelement Δ₂ auf dem korrespondieren Dipol 21 existiert und, wenn beide ein Paar bildende Dipole 21 an der Dipol-Anschlussstelle 22 jeweils mit dem gleichen Strom dem Betrag nach gespeist werden, das arithmetische Mittel der Phasen dieser in der jeweils gleichen Richtung gezählten Ströme eines Dipolpaares den gleichen Wert besitzt und dieser Wert für alle derart in derselben Ebene gebildeten Dipolpaare gleich ist.
   Die Dipole 21 sind bevorzugt geradlinig und zu Dipol-Anschlussstelle 22 symmetrisch und verlaufen in einer horizontalen Ebene, wobei die Dipol-Anschlussstellen mehrerer Dipolpaare äquidistant auf einem horizontalen Kreis, dessen Mittelpunkt den gemeinsamen Bezugspunkt B bildet, verteilt angeordnet und die Dipole 21 senkrecht zur Verbindungslinie zum Mittelpunkt des Kreises orientiert sind. Auf diese Weise wird eine Kreisgruppenantennenanlage 9 gebildet, die entsprechend dem Reziprozitätsgesetz ein Verteilungsnetzwerk 10 mit mehreren Ausgängen 23 enthält, dessen Eingang als Antennenanschlussstelle 3 ausgeführt ist, wobei die Dipol-Anschlussstellen jeweils über eine elektrische Leitung an einen der Ausgänge des Verteilungsnetzwerks 10 angeschlossen sind und die Dipolpaare mit gleichen Signalen nach Amplituden und Phasen gespeist sind ( Figur 13a ).
   Die Kreisgruppe sollte zur Erzeugung einer hinreichend runden azimutalen Strahlungscharakteristik eine hinreichende Anzahl von Dipolpaaren enthalten und über einer elektrisch leitenden horizontalen Grundfläche 6 in einem der Gestaltung der vertikalen Strahlungscharakteristik entsprechenden Abstand angeordnet sein ( Figur 13b ).
   Im zentralen Phasen-Bezugspunkt B kann ein elektrisch kurzer, vertikaler Monopol 7 vorhanden sein. Ferner ist ein Verteilungsnetzwerk 10 vorhanden, dessen dem Reiziprozitätsgesetz entsprechender Eingang die Antennenanschlussstelle 3 bildet, wobei die Kreisgruppenantennenanlage 9 und der Monopol 7 jeweils über eine elektrische Leitung 27 von einem Ausgang 23 des Verteilungsnetzwerks 10 in der Weise gespeist sind, dass die Phasen des in den Monopol 7 eingespeisten Stromes der Phasenlage der in der Kreisgruppenantennenanlage 9 eingespeisten Ströme bezüglich des gemeinsamen Phasen-Bezugspunkts B entsprechen. Hierbei sind zweckmäßigerweise mehrere elektrisch kurze vertikale Monopole 7 paarweise symmetrisch zum zentralen Phasen-Bezugspunkt B angeordnet vorhanden, wobei die Monopole entsprechend dem Reziprozitätsgesetz vom Verteilungsnetzwerk 10 in der Weise gespeist sind, dass das arithmetische Mittel der Stromphasen der paarweise angeordneten Monopole 7 und die Phase des in einen zentralen Monopol 7 eingespeisten Stromes bezogen auf den Phasen-Bezugspunkt B jeweils gleich sind.
5. 5. In einer bevorzugten Ausgestaltung ist das Verteilungsnetzwerk 10 für die Verwendung der Antenne als Diversity-Empfangsantenne in der Weise gestaltet ist, dass sowohl die Empfangssignale der vorangegangenen unter Ziffer 4 erläuterten Antenne als auch die des vertikalen Monopols 7 und die zusammengefassten Empfangssignale der Kreisgruppenantennenanlage 9 jeweils getrennt voneinander alternativ zur Verfügung stehen.
   Das Verteilungsnetzwerk 10 kann aber auch für die Verwendung der Antennenanordnung als Diversity-Empfangsantenne in der Weise gestaltet sein, dass sowohl die Empfangssignale der vorangegangenen unter Ziffer 3 erläuterten Antenne als auch die des vertikalen Monopols 7 und die Empfangssignale der Schleifenantenne 14 jeweils getrennt voneinander alternativ zur Verfügung stehen ( Figur 14 ).
6. 6. Die Auskopplung an der Antennenanschlussstelle 3 über eine daran angeschlossene symmetrische Zweidrahtleitung 26 wie sie unter Ziffer 3 erwähnt ist, kann auch so erfolgen, dass die Zweidrahtleitung innerhalb der senkrecht zur Grundfläche 6 und symmetrisch bezüglich der Antennenanschlussstelle 3 orientierten Symmetrieebene SE der Antennenanordnung zur leitenden Grundfläche 6 geführt ist ( Figur 6 ). Auch kann an Stelle des vertikalen Monopols 7 die Zuleitung zur Speisung der Schleifenantenne 14 als vertikal ausgerichtete Zweidrahtleitung 26 im Zentrum Z der Schleifenantenne 14 angeordnet sein, wodurch die Zweidrahtleitung zum einen die Funktion eines Monopols 7 mit der Schleifenantenne 14 als Dachkapazität 12 erhält und zum anderen die Speisung der Schleifenantenne 14 wahrgenommen ist, wobei im zentralen Fußpunkt auf der leitenden Grundfläche 6 zwei Auskopplungen für die beiden auf diese Weise gebildeten Antennen vorhanden sind (Figur 10). Hierbei kann (entsprechend dem Reziprozitätsgesetz) das unsymmetrische Leistungsteiler- und Phasenschiebernetzwerk 31 im Fußpunkt der Antennenanordnung dadurch realisiert werden, dass der eine Leiter der Zweidrahtleitung 26 über einen Blindwiderstand 41 mit der leitenden Grundfläche 6 leitend verbunden wird, und der andere Leiter der Zweidrahtleitung 26 zur Anschlussstelle 28 der Antennenanordnung geführt und durch die Wahl des Blindwiderstands 41 die Gewichtung des Empfangs des horizontal und des vertikal polarisierten elektrischen Feldes eingestellt wird ( Figur 15 ).
7. 7. Bei einer unter Ziffer 1 erwähnten Antenne kann zusätzlich für den Empfang von Signalen mit niedrigeren Frequenzen - wie zum Beispiel AM/FM-Rundfunksignalen - eine größere Gesamtlänge hg gestaltet sein, wobei die für den Satellitenempfang notwendige Länge h2 hinausgehende Teil der stabförmigen Antenne über eine Unterbrechungsstelle 5 abgetrennt ist und dieser Teil, abhängig von seiner Länge, mit einer oder mehreren Unterbrechungsstellen 5 in Abständen von weniger als 1/5λ versehen ist und wobei diese Unterbrechungsstellen jeweils mit einer auf die Mittelfrequenz f_m der Satelliten-Frequenzbänder abgestimmten Resonanzschaltung 39 beschaltet sind, welche bei dieser Frequenz hochohmig ist ( Figur 4 ).
   Innerhalb der senkrecht zur Grundfläche 6 und symmetrisch bezüglich der Antennenanschlussstelle 3 orientierten Symmetrieebene SE der Antennenanordnung kann mindestens eine linear beziehungsweise flächig gestaltete Antenne für einen oder mehrere andere Funkdienste gestaltet sein ( Figur 16 ).
8. 8. Bei der unter Ziffer 3 erwähnten Antenne sind die vertikalen Richtdiagramme das als Stabantenne ausgebildeten Monopols und der Schleifenantenne 14 bevorzugt deckungsgleich und bezüglich der Hauptrichtung für den Empfang der Satellitensignale eingestellt, wobei ein Anpassnetzwerk 25 für die Schleifenantenne 14 und ein Anpassnetzwerk 33 für den Monopol in der Form vorhanden sind, sodass ein gemeinsames Phasenzentrum B gebildet ist. Die beiden Ausgänge der Anpassnetzwerke 32, 33 können mit den Eingängen 48, 49 eines 90°-Hybridkopplers 45 verbunden sein, so dass ein Ausgang 46 für LHCP-Wellen und der andere Ausgang 47 für RHCP-Wellen gestaltet ist ( Figur 17a , Fig. 19 ).
9. 9. Die unter Ziffer 6 beschriebene Antenne ist bevorzugt so gestaltet, dass die Schleifenantenne 14 zwei aneinander gegenüberliegende Antennenanschlussstellen 3 und daran angeschlossene und in der Schleifenebene befindliche Anpassnetzwerke 25 aufweist, deren Ausgänge addierend parallel geschaltet sind, wobei das unsymmetrische Leistungsteiler- und Phasenschiebernetzwerk 31 im Fußpunkt der Antennenanordnung dadurch realisiert ist, dass der eine Leiter der Zweidrahtleitung 26 über einen Blindwiderstand 41 mit der leitenden Grundfläche 6 leitend verbunden ist und der andere Leiter der Zweidrahtleitung 26 zur Anschlussstelle 28 der Antennenanordnung geführt ist. Durch Wahl des Netzwerk 53 aus Blindwiderständen kann die Gewichtung des Empfangs des horizontal und des vertikal polarisierten elektrischen Feldes eingestellt werden. ( Figur 18 ). Zur Vertauschung der Polarität der Empfangsspannung der Schleifenantenne 14 kann vorgesehen sein, dass die Empfangsspannung der Schleifenantenne 14 mit unterschiedlichen Vorzeichen der Empfangsspannung aus dem vertikal polarisierten elektrischen Feld hinzugefügt werden kann, und der Empfang von LHC- und RHC-polarisiertem Feld durch Umschaltung der LHRCP/RHCP-Umschalter 55 wahlweise möglich ist ( Figur 20 ).

For the design of satellite receiving antennas, which are equally suitable for the reception of left-handed circularly polarized signals and for the reception of clockwise circularly polarized signals, the following features and combinations of features have proven to be preferred:

1. 1. The design of electrically very short conductor elements Δ ₁ , Δ ₂ ,... Of the antenna 1 ensures that the electric field strength vector generated in the far field is generated in accordance with the reciprocity law applicable between receiving antennas and transmitting antennas when transmitting power is supplied to at least one antenna connection point of the antenna 3 e _ν is polarized in every point P of space at any time along a fixed straight line specific to that point P of space.
   This condition can be met, for example, if all the conductor elements Δ1, Δ2 are arranged along a straight line 2 and are conductively connected to one another, so that substantially a rod-shaped conductor 4 is formed and the antenna connection point 3 is formed by an interruption of the rod-shaped conductor 4.
   The substantially rod-shaped conductor 4 is preferably mounted substantially perpendicularly over a substantially horizontal conductive base 6 and has an interruption point, through which the antenna connection point 3 is formed. Preferably, the thus formed substantially perpendicular monopole 7 for the design of the vertical diagram on at least one point of interruption 5, which is connected to at least one dummy element 8. The antenna connection point 3 formed at the base of the monopole 7 can contribute to the design of the optimal reception in the range of elevation angle between 25 ° and 65 ° the total length h2 of the monopole 7 about 5/8 λ of the satellite signals to be received, the interruption point 5 in one Height h1 of about 3/8 λ - 4/8 λ mounted above a conductive base 6 and connected to a reactive at this frequency reactance 8 of approximately 200 ohms ( FIG. 3 ).
2. 2. The conductor elements Δ ₁ , Δ ₂ ,... Can be arranged along a plurality of straight lines which are parallel to one another, so that a plurality of rod-shaped conductors 4 are formed, of which the antenna connection point 3 is formed in at least one. The rod-shaped conductors 4 can in this case be oriented vertically over the substantially horizontal conductive base surface 6.
   For example, to design a substantially round azimuthal directional diagram, a circular array antenna system 9 with mutually identical rod-shaped conductors 4 can be provided as parasitic radiators 11, wherein in the center Z of the circular array antenna system 9 an antenna according to the preceding paragraph 1 and one according to the requirements of the roundness of azimuthal Directional diagram sufficiently large number of arranged on a circle at the same angular distance W from each other parasitic radiators 11 are provided.
   The circular array antenna system 9 contains a distribution network or a coupling network with a plurality of terminals 23, wherein one (24) of the terminals is designed as antenna connection point 3 and the rod-shaped and identically designed arranged in the circle group rod-shaped conductor 4 each contain a point of interruption 5 and thus as a radiator. 7 are formed, which are each connected via a similar electrical line 27 to each one of the remaining terminals of the network 10 and reciprocally transmitting with the same signals according to amplitudes and phases are fed, said located in the center Z of the circular array antenna 9 emitters 7 for the design of Directional diagram is also connected to one of the terminals of the network 10 and can be fed with a signal with a separate amplitude and phase. Alternatively, in the center Z of the circle group instead of the radiator 7, a parasitic radiator 11 may be attached. The arranged in a circle rod-shaped conductors 4 for the design of the vertical diagram can each contain at least one connected to at least one dummy element 8 interruption point 5. The same applies to the rod-shaped conductor arranged in the center Z of the circle group, which may each contain at least one interruption point 5 connected to at least one dummy element 8 for designing the vertical diagram. To design the lowest possible rod-shaped conductors, these may contain at their upper end a roof capacity 12 and thereby act extended. In addition, the circular array antenna system 9 may also consist of a plurality of concentric circles arranged in the respective circle of similar executed rod-shaped conductors, which are optionally excited the same amount and phase.
3. 3. In a preferred embodiment, the antenna consists of a plurality of each symmetrical to a common reference point in space the manner indicated in pairs and arranged identically aligned very short conductor elements Δ1, Δ2 and Δ3, Δ4 and Δ5, Δ6, where - caused by the excitation of the antenna at the antenna connection point 3 - these act in pairs as radiating elementary antennas Δ _n , Δ _m in such a way that the associated both to one elementary antenna pair elementary antennas Δ _n, Δ _m current flowing according to size is the same and the reference point for all the elementary antenna pairs Δ _n, Δ _m B form a common phase center in such a way that the arithmetic mean of Phases of the two, in the same direction counted currents of an elementary antenna pair for all elementary antenna pairs Δ _n , Δ _{m has} the same value.
   Preferably, a loop antenna 14 is formed by a conductive juxtaposition of electrically very short conductor elements around the common reference point with an arranged at one point by interrupting the loop antenna connection point 3, wherein the dimensions of the loop are electrically sufficiently small, so that the ring current in the amount of Each location is the same and each very short conductor element is complemented by a corresponding very short conductor element to a pair. All the conductor elements Δ ₁ , Δ ₂ ... Suitably extend in a plane, wherein the loop antenna 14 may take the form of a regular n-corner whose phase reference point is given by the symmetry point of the n-corner, or the shape of a circular ring. where reference point B is given by the center of the annulus. The loop antenna 14 can also be formed of a plurality of closed loops with common phase reference point B, but in one of the loops by interruption, the antenna connection point 3 is to be formed. In this case, the loop antenna 14 can be configured from a plurality of loops arranged in a row in substantially parallel planes with the smallest possible distance from one another in the form of a coil, so that a substantially common phase reference point is formed for all loops and the antenna connection point 3 passes through the two ends of the spiral is given.
   As far as the loop antenna 14 is not electrically small, it may contain the effective electrical truncations several introduced at breakpoints 5 capacitances 16, whereby the constancy of the current is sufficiently ensured in accordance with the magnitude and phase on the conductor elements Δ _1, Δ _2, ( FIG. 5a ). It is preferred that the loop antenna 14 is circular or approximately square in shape in a plane parallel to a substantially horizontal conductive base 6 and has capacitances 16 introduced at break points, which will control both the continuity of the current on the conductor elements Δ ₁ , Δ ₂ and to shape the vertical diagram.
   To design the reception in the range of an elevation angle between 25 ° and 65 ° with azimuthal omnidirectional characteristic, the loop antenna 14 is preferably placed at a distance of approximately 1/16 to 1/8 of the wavelength above the conductive base 6, the side length of the loop antenna 14 being approximately 1/4 of the wavelength is selected and along the conductor guide at intervals of about 1/8 of the wavelength each with a capacity with a reactance of about -200 ohms connected interruption point is introduced ( FIG. 5b and c).
   In a preferred embodiment, an electrically short vertical monopole 7 and a distribution network 10 is provided in the central phase reference point, whose input is designed as an antenna connection point 3 and the loop antenna 14 and the monopole 7 in accordance with the reciprocity law between receiving antennas and transmitting antennas each via an electrical line are fed by an output of the distribution network in such a way that the phases of the current fed into the monopole 7 and into the loop antenna are the same ( FIG. 9 ) . The distribution network is designed for this purpose as a power divider and phase shifter network 31 with separate connections for the loop antenna 14 and the monopole 7 in such a way that the phases of the current fed into the monopole 7 and into the loop antenna 14 to form the common phase center B below Considering the reflection at the base 6 are almost equal and that the weighting is set in the superposition of the effects of the loop antenna 14 and the monopole 7 in such a way that the main direction of the resulting vertical radiation pattern is set for the satellite reception that the radiation pattern However, due to the effect of the monopole 7, it is filled up to low elevation angles ( FIG. 9 ).
4. 4. In a further preferred variant, in each case a group of electrically very short conductor elements Δ ₁ , Δ ₂ running substantially in a horizontal plane are connected to one another in an electrically conductive manner, such that they have a plurality of electrically short dipoles 21 with almost the same phase of the currents form the conductor elements Δ ₁ , Δ ₂ , which are fed to a dipole connection point 22 formed by an interruption point, wherein in each case symmetrically to the common reference point B an identically shaped electrically short dipole 21 is correspondingly present, so that for each electrically very short conductor element Δ ₁ on a dipole a corresponding corresponding, substantially in the same plane extending conductor element Δ ₂ on the corresponding dipole 21 exists and, if both a pair of dipoles 21 are fed at the dipole junction 22 in each case with the same amount of current, the arithmetic This means the phases of this in the jewe In the same direction counted currents of a dipole pair has the same value and this value is the same for all dipole pairs formed in the same plane.
   The dipoles 21 are preferably rectilinear and symmetrical to the dipole junction 22 and extend in a horizontal plane, the dipole junctions of several pairs of dipoles arranged equidistant on a horizontal circle whose center forms the common reference point B, and the dipoles 21 perpendicular to the connecting line are oriented to the center of the circle. In this way, a circular array antenna system 9 is formed, which according to the reciprocity law, a distribution network 10 having a plurality of outputs 23, whose input is designed as an antenna connection point 3, wherein The dipole connection points are each connected via an electrical line to one of the outputs of the distribution network 10 and the dipole pairs are supplied with equal signals by amplitudes and phases ( FIG. 13a ).
   The circle group should contain a sufficient number of dipole pairs to generate a sufficiently round azimuthal radiation characteristic and be arranged above an electrically conductive horizontal base surface 6 in a distance corresponding to the configuration of the vertical radiation characteristic ( FIG. 13b ).
   In the central phase reference point B, an electrically short, vertical monopole 7 may be present. Further, a distribution network 10 is provided, whose the Reiziprozitätsgesetz corresponding input forms the antenna connection point 3, the circular array antenna system 9 and the monopoly 7 are respectively fed via an electrical line 27 from an output 23 of the distribution network 10 in such a way that the phases of the in the Monopole 7 fed current of the phase angle of the fed in the group antenna array 9 streams with respect to the common phase reference point B correspond. Here are expediently several electrically short vertical monopoles 7 arranged in pairs symmetrically to the central phase reference point B, wherein the monopolies are fed according to the reciprocity law of the distribution network 10 in such a way that the arithmetic mean of the current phases of the paired monopolies 7 and the phase of in a central monopole 7 fed current relative to the phase reference point B are the same.
5. 5. In a preferred embodiment, the distribution network 10 is designed for the use of the antenna as a diversity receiving antenna in such a way that both the received signals of the previous explained under 4 antenna and those of the vertical monopoly 7 and the combined received signals of the circular array antenna system each separately as an alternative.
   However, the distribution network 10 can also be designed for the use of the antenna arrangement as a diversity receiving antenna in such a way that both the received signals of the previous explained in paragraph 3 antenna as well as the vertical monopole 7 and the received signals of the loop antenna 14 are each separated from each other alternatively Available ( FIG. 14 ).
6. 6. The decoupling at the antenna connection point 3 via a symmetrical two-wire line 26 connected thereto, as mentioned under point 3, can also take place in such a way that the two-wire line is conductive within the plane of symmetry SE of the antenna arrangement oriented perpendicular to the base surface 6 and symmetrically with respect to the antenna connection point 3 Base area 6 is guided ( FIG. 6 ). Also, in place of the vertical monopole 7, the supply line for feeding the loop antenna 14 may be arranged as a vertically aligned two-wire line 26 in the center Z of the loop antenna 14, whereby the two-wire line on the one hand the function of a monopoly 7 with the loop antenna 14 as roof capacity 12 receives and on the other the supply of the loop antenna 14 is perceived, wherein in the central base point on the conductive base surface 6 two outcouplings for the two antennas formed in this way are present ( FIG. 10 ). In this case (in accordance with the law of reciprocity), the unbalanced power divider and phase shifter network 31 can be realized in the base of the antenna arrangement in that one conductor of the two-wire line 26 is conductively connected to the conductive base 6 via a reactance 41, and the other conductor of the two-wire line 26 to Junction 28 of the antenna array out and adjusted by the choice of the reactance 41, the weighting of the reception of the horizontal and the vertical polarized electric field ( FIG. 15 ).
7. 7. In the case of an antenna mentioned under 1 above, it can also be used to receive signals with lower frequencies - such as AM / FM broadcast signals a greater total length hg be designed, wherein the necessary for the satellite reception length h2 part of the rod-shaped antenna is separated via an interruption point 5 and this part, depending on its length, with one or more interruption points 5 at intervals of less than 1 / 5λ is provided and wherein these interruption points are each connected to a tuned to the center frequency f _{m of} the satellite frequency bands resonant circuit 39 which is high impedance at this frequency ( FIG. 4 ).
   Within the symmetry plane SE of the antenna arrangement oriented perpendicular to the base surface 6 and symmetrically with respect to the antenna connection point 3, at least one antenna configured in a linear or planar manner can be designed for one or more other radio services ( FIG. 16 ).
8. 8. In the case of the antenna referred to in point 3, the vertical directional patterns of the monopole and loop antenna 14 are preferably congruent with respect to the main direction of reception of the satellite signals, with a matching network 25 for the loop antenna 14 and a matching network 33 for the monopole in the Form are present, so that a common phase center B is formed. The two outputs of the matching networks 32, 33 can be connected to the inputs 48, 49 of a 90 ° hybrid coupler 45, so that one output 46 is designed for LHCP waves and the other output 47 is designed for RHCP waves ( Figure 17a . Fig. 19 ).
9. 9. The antenna described in paragraph 6 is preferably designed so that the loop antenna 14 has two mutually opposite antenna connection points 3 and connected thereto and located in the loop plane matching networks 25 whose outputs are connected in addition in parallel, wherein the unbalanced power divider and phase shifter network 31 in Base point of the antenna arrangement is realized in that the one conductor of the two-wire line 26 is conductively connected via a reactance 41 to the conductive base 6 and the other conductor of the two-wire line 26 is guided to the connection point 28 of the antenna arrangement. By selecting the network 53 of reactances, the weighting of the reception of the horizontally and vertically polarized electric field can be adjusted. ( FIG. 18 ). For interchanging the polarity of the received voltage of the loop antenna 14, it may be provided that the receiving voltage of the loop antenna 14 can be added with different signs of the received voltage from the vertically polarized electric field, and the reception of LHC and RHC polarized field by switching the LHRCP / RHCP switch 55 is optional ( FIG. 20 ).

Claims (12)

1. Antenna for reception of circularly polarized satellite radio signals, comprising:

   - at least one two- or three-dimensional antenna conductor structure forming a loop antenna (14), the antenna conductor structure essentially comprised of a plurality of antenna conductor sections (Δ_v) which are arranged symmetrically in pairs with regard to a spatial reference point common to the antenna conductor sections (Δ_v) and extending in the same direction and are electrically connected to form a conductor loop arranged essentially in a horizontal plane, wherein at least one gap in the conductor loop forms an antenna connection point (3) of the loop antenna (14),

   - a linear antenna conductor structure, and

   - a matching and phase shift network (25, 31) connecting the at least one antenna connection point (3) of the loop antenna (14) and an antenna connection point (3) of the linear antenna structure to an antenna output port (28), characterized in that

   the loop antenna (14) is arranged such that, in case of reciprocal operation of the antenna as a transmitting antenna, antenna currents of at least approximately equal size flow in the individual pairs of antenna conductor sections (Δ_v) and the arithmetic mean of the current phases of said antenna currents, counted in the same direction in the antenna conductor sections (Δ_v) of each pair at essentially all pairs of antenna conductor sections (Δ_v) relative to a common phase reference point (B) has at least approximately the same value,
   the loop antenna (14) is arranged parallel to and spaced apart from a approximately horizontal electrically conductive base surface (6),
   the linear antenna structure is formed as a electrically short, vertical monopole (7) disposed in the phase reference point (B) of the loop antenna (14), and
   the matching and phase shift network (25, 31) connecting the antenna connection point (3) of the monopole (7) and the antenna connection point (3) of the loop antenna (14) to the antenna output port (28) is arranged such that, in reciprocal operation of the antenna as a transmitting antenna, it approximates the phases of the currents at the antenna connection points (3) of the monopole (7) and the loop antenna (14) to one another.
2. Antenna according to claim 1, characterized in that
   the matching and phase shift network (25, 31) is configured such that, in reciprocal operation of the antenna as a transmitting antenna, it overlaps the currents of the monopole (7) and the loop antenna (14) one another for influencing the vertical directional diagram.
3. Antenna for reception of circularly polarized satellite radio signals, comprising:

   - at least one two- or three-dimensional antenna conductor structure forming a loop antenna (14), the antenna conductor structure essentially comprised of a plurality of antenna conductor sections (Δ_v) which are arranged symmetrically in pairs with regard to a spatial reference point common to the antenna conductor sections (Δ_v) and extending in the same direction and are electrically connected to form a conductor loop arranged essentially in one horizontal plane, wherein at least one gap in the conductor loop forms an antenna connection point (3) of the loop antenna (14),

   - a linear antenna conductor structure, and

   - a matching and phase shift network (25, 31) connecting the at least one antenna connection point (3) of the loop antenna (14) and an antenna connection point (3) of the linear antenna structure to an antenna output port (28), characterized in that

   the loop antenna (14) is arranged such that, in case of reciprocal operation of the antenna as a transmitting antenna, antenna currents of at least approximately equal size flow in the individual pairs of antenna conductor sections (Δ_v) and the arithmetic mean of the current phases of said antenna currents, counted in the same direction in the antenna conductor sections (Δ_v) of each pair at essentially all pairs of antenna conductor sections (Δ_v) relative to a common phase reference point (B) has at least approximately the same value,
   the loop antenna (14) is arranged parallel to and spaced apart from a approximately horizontal conductive base surface (6),
   the at least one antenna connection point (3) of the loop antenna (14) is connected to the antenna output port (28) via a two-wire line (26) at least between the plane of the conductor loop and the electrically conductive base surface (6), wherein the two-wire line (26) and the antenna connection point (3) are symmetrically arranged to a vertical plane of symmetry (SE) comprising the spatial reference point and, in reciprocal operation of the antenna as a transmitting antenna, the formed phase reference point (B),
   the two-wire line (26) runs vertically through the spatial reference point and, in reciprocal operation of the antenna as a transmitting antenna, the formed phase reference point (B) and is used as a vertical monopole (7) with a top capacity (12) formed by the conductor loop, and
   the matching and phase shift network (33, 31) connects the two-wire line (26) to the antenna output port (28) and couples out both currents of the monopole (7) and the loop antenna (14) on the electrically conductive base surface (6).
4. Antenna according to claim 3, characterized in that one of the two conductors of the two-wire line (26) is conductively connected via a reactance (41) to the conductive base surface (6) for weighting of reception of the horizontally polarized and the vertically polarized electrical field and the other of the two conductors is connected to the antenna output port (28) via the matching and phase shift network (33, 31).
5. Antenna according to claim 3 or 4, characterized in that the loop antenna (14) has two antenna connection points (3) opposite each other in the plane of symmetry (SE), to which are connected matching networks (25) arranged in the loop plane, whose ports are connected in parallel for adding and connected to the two-wire line (26).
6. Antenna according to any of the claims 3 to 5, characterized in that at least one further linear or planar antenna (24) is located within the plane of symmetry (SE) for at least one additional radio service.
7. Antenna according to any of the claims 1 to 6, characterized in that the conductor loop comprises, for its electrically active shortening, at least one gap bridged by a capacitance (16), particularly a plurality of gaps spaced apart from each other and bridged by capacitances (16).
8. Antenna for reception of circularly polarized satellite radio signals, comprising:

   - at least one two- or three-dimensional antenna conductor structure (14), the antenna conductor structure essentially comprised of a plurality of antenna conductor sections (Δ_v) which are arranged symmetrically in pairs with regard to a spatial reference point common to the antenna conductor sections (Δ_v) and extending in the same direction essentially in a common horizontal plane,

   - a linear antenna conductor structure, and

   - a matching and phase shift network (31, 33) connecting the two- or three-dimensional antenna conductor structure and the linear antenna conductor structure to an antenna output port (28),

   characterized in that
   the two- or three-dimensional antenna conductor structure (14; 21; 42) is adapted such that, in case of reciprocal operation of the antenna as a transmitting antenna, antenna currents of at least approximately equal size flow in the individual pairs of antenna conductor sections (Δ_v) and the arithmetic mean of the current phases of said antenna currents, counted in the same direction in the antenna conductor sections (Δ_v) of each pair at essentially all pairs of antenna conductor sections (Δ_v) relative to a common phase reference point (B) has at least approximately the same value, wherein the antenna conductor sections are arranged in the form of a dipole group including a plurality of dipoles (21) arranged essentially in a common horizontal plane, wherein the dipoles (21) are arranged symmetrically in pairs with regard to the formed phase reference point (B), in reciprocal operation of the antenna as a transmitting antenna, or the spatial reference point, and the pairs of antenna conductor sections are each allocated to dipole pairs, and wherein the individual dipoles (21) are arranged such that the antenna currents, in reciprocal operation of the antenna in transmitting operation, on its dipole conductors have approximately equal phases, and the arithmetic mean of the phases of said antenna currents of each dipole pair, counted in the same direction, has the same value and the values for all dipole pairs in the common horizontal plane are the same,
   the dipoles (21) of the dipole group are straight dipoles each symmetrical to their dipole connection points (3), arranged parallel to and spaced apart from a approximately horizontal electrically conductive base surface (6), wherein the dipole connection points (3) are arranged in a circle around the phase reference point (B) or the spatial reference point in the common horizontal plane and connected to the antenna output port (28) via an interconnection network (10),
   the linear antenna conductor structure is disposed in the phase reference point (B) of the dipole group formed in reciprocal operation of the antenna as a transmitting antenna and forms an electrically short, vertical monopole (7), and that an antenna connection point of the monopole (7) and an output port of the interconnection network (10) are connected to the antenna output port (28) via the matching and phase shift network (31, 33), and that the matching and phase shift network (31, 33) matches the phases of the currents resulting from the reciprocal operation of the antenna as a transmitting antenna at the antenna connection point of the monopole and the output port of the interconnection network (10).
9. Antenna according to claim 8, characterized in that the matching and phase shift network (31, 33) is arranged such that it overlaps the currents of the monopole (7) and of the interconnection network (10) one another for influencing the vertical directional diagram.
10. Antenna according to any of the claims 1 to 9, characterized in that an antenna connection point of the monopole (7) and an antenna connection point of the two- or three-dimensional antenna conductor structure (14, 21) are each connected to a switch (37) of an antenna diversity system (38) connected to the antenna output port (28) either directly individually or via a matching network (25).
11. Antenna according to any of the claims 1 to 10, characterized in that an antenna connection point of the two- or three-dimensional antenna conductor structure (14) is each connected individually via a matching network (25, 33) to inputs of a signal combination circuit, in particular a 90° hybrid coupler (45), whose outputs separately from each other provide a left-hand circularly polarized reception signal and a right-hand circularly polarized reception signal.
12. Antenna according to claim 11, characterized in that an element (56) is connected between the antenna connection point of the monopole (7) and/or the two- or three-dimensional antenna conductor structure (14) and the respective associated input of the signal combination circuit (45) for adjusting the attenuation and/or the phase of the reception signal.

Images