WO2024069811A1

WO2024069811A1 - Motor drive device and refrigeration cycle instrument

Info

Publication number: WO2024069811A1
Application number: PCT/JP2022/036253
Authority: WO
Inventors: 和徳畠山; 慎也豊留; 亮一佐々木
Original assignee: 三菱電機株式会社
Priority date: 2022-09-28
Filing date: 2022-09-28
Publication date: 2024-04-04

Abstract

A motor drive device (100) drives a motor (40) that generates, in addition to a fundamental wave component of an induced voltage, at least one harmonic voltage of a fifth-order harmonic component and a seventh-order harmonic component of the induced voltage. The motor drive device (100) comprises: an inverter (30) that converts a DC voltage applied from a capacitor (22) to an AC voltage and applies the resulting voltage to the motor (40); and a control unit (60) that controls an inverter output voltage outputted by the inverter (30). The control unit (60) controls the inverter output voltage so as to suppress a torque variation component generated from the product of the fundamental wave component of a motor current which is a current flowing through the motor (40), and at least one harmonic induced voltage of a fifth-order harmonic component and a seventh-order harmonic component.

Description

Motor drive device and refrigeration cycle equipment

This disclosure relates to a motor drive device that drives a motor, and a refrigeration cycle device equipped with a motor drive device.

Patent Document 1 below discloses a technique for reducing noise caused by resonance between a three-phase fan motor and the rotor of the fan motor. Specifically, Patent Document 1 discloses a technique for adding the fifth and seventh order components of the induced voltage generated by the fan motor to the applied voltage of each phase of the fan motor in order to cancel out the sixth order sound component in the dq coordinate system.

JP 2014-87167 A

As described above, in the technology of Patent Document 1, the fifth and seventh order components of the induced voltage component are added to the voltage command of each phase of the fan motor. The fifth and seventh order components to be added have adjustment parameters such as the ratio G ( _G5 , _G7 ) to the amplitude of the induced voltage and the phase difference φ ( _φ5 , _φ7 ) to the induced voltage component. However, these adjustment parameters need to be adjusted depending on the operating state, and therefore there is a problem that the adjustment requires a lot of effort.

Furthermore, in the case of a method of uniquely setting adjustment parameters, it is expected that load fluctuations will occur, causing a mismatch between the voltage command frequency and the motor rotation speed frequency. Propeller fans and fan motors used in air conditioners and other appliances are placed in environments with large temperature changes, and are therefore devices that experience large load fluctuations. If a load fluctuation occurs and the voltage command frequency and the motor rotation speed frequency do not match, there is a risk that voltage will be applied at a phase different from the expected phase, which could result in worsening vibrations and noise.

The present disclosure has been made in consideration of the above, and aims to provide a motor drive device that can easily suppress the generation of vibrations and noise, even when applied to equipment with large load fluctuations.

In order to solve the above problems and achieve the objectives, the motor drive device according to the present disclosure is a motor drive device that drives a motor that generates at least one harmonic voltage of the fifth harmonic component and seventh harmonic component of the induced voltage in addition to the fundamental component of the induced voltage. The motor drive device includes an inverter that converts a DC voltage applied from a DC power source into an AC voltage and applies it to the motor, and a control unit that controls the inverter output voltage output by the inverter. The control unit controls the inverter output voltage so as to suppress the torque fluctuation component that is generated from the product of the fundamental component of the motor current, which is the current flowing through the motor, and at least one harmonic induced voltage of the fifth harmonic component and seventh harmonic component.

The motor drive device disclosed herein has the advantage of being able to easily suppress the generation of vibrations and noise, even when applied to equipment with large load fluctuations.

FIG. 1 is a diagram showing an example of the configuration of a motor drive device according to an embodiment; FIG. 1 is a block diagram showing an example of a hardware configuration for implementing the functions of a control unit according to an embodiment; FIG. 11 is a block diagram showing another example of a hardware configuration for implementing the functions of the control unit according to the embodiment; FIG. 1 is a diagram for explaining a method for generating a PWM signal according to an embodiment; FIG. 1 is a diagram showing an example of a waveform of an induced voltage that may occur in a typical permanent magnet motor. FIG. 6 is a diagram showing a change in power due to the fifth harmonic component of the induced voltage shown in FIG. 5 . FIG. 7 is a diagram showing the results of extracting only the power ripple components from the results shown in FIG. 6 . FIG. 6 is a diagram showing a change in power due to the seventh harmonic component of the induced voltage shown in FIG. 5 . FIG. 9 is a diagram showing the results of extracting only the power ripple components from the results shown in FIG. 8 . FIG. 6 is a diagram showing how power changes when the fifth harmonic component and the seventh harmonic component of the induced voltage shown in FIG. 5 exist simultaneously. FIG. 11 is a diagram showing the result of extracting only the power ripple component from the result shown in FIG. 10 . FIG. 2 is a diagram showing a configuration example of a control unit according to an embodiment; FIG. 13 is a diagram showing an example of the configuration of a torque fluctuation compensation amount calculation unit provided in the control unit shown in FIG. 12 . 1 is a flowchart showing a flow of processing by a control unit according to an embodiment. FIG. 13 is a diagram showing a configuration example of a control unit according to a modified example of the embodiment;

Below, the motor drive device and refrigeration cycle equipment according to the embodiment of the present disclosure are described in detail with reference to the drawings.

Embodiment
FIG. 1 is a diagram showing a configuration example of a motor drive device 100 according to an embodiment. The motor drive device 100 is connected to an AC power source 10 and a motor 40. The motor drive device 100 converts a first AC power based on a power source voltage supplied from the AC power source 10 into a second AC power having a desired amplitude and phase and supplies the second AC power to the motor 40. The motor drive device 100 includes a reactor 20, a rectifier 21, a capacitor 22, an inverter 30, a control unit 60, a current detection unit 74, and a DC voltage detection unit 76. The motor drive device 100 according to the embodiment can be used in a refrigeration cycle device. Examples of the refrigeration cycle device include an air conditioner, a heat pump water heater, a refrigerator, and a freezer.

A motor 40 is connected to the inverter 30, and a load 50 is connected to the motor 40. In this paper, the motor 40 is assumed to be a three-phase motor. An example of the load 50 is a propeller fan or a compressor of an outdoor unit provided in a refrigeration cycle device. Figure 1 illustrates an example in which the load 50 is a propeller fan.

The AC power source 10 may be, for example, a 50 Hz or 60 Hz commercial power source, but is not limited to these commercial power sources. The AC power source 10 may also be a power source system using distributed power sources that generate AC voltage using DC voltage output from a stationary storage battery, a solar power generation device, or the like.

In the motor drive device 100, the power supply voltage output from the AC power supply 10 is applied to the rectifier 21 via the reactor 20. The rectifier 21 rectifies the power supply voltage and converts it into a DC voltage. The capacitor 22 is connected in parallel to the rectifier 21 and the inverter 30 between the rectifier 21 and the inverter 30. The capacitor 22 smoothes the DC voltage output from the rectifier 21. The capacitor 22 outputs the smoothed DC voltage to the inverter 30. The capacitor 22 functions as a DC power supply for the inverter 30. The inverter 30 converts the DC voltage applied from the capacitor 22 into an AC voltage and applies it to the motor 40.

The reactor 20 may be an EI or EE type made of laminated electromagnetic steel sheets, or may use a ferrite or amorphous iron core. Copper or aluminum is used for the windings.

The rectifier 21 may be configured by arranging diodes in a bridge configuration, or may be configured by power semiconductor elements such as MOSFETs instead of diodes. The power semiconductor elements such as diodes and MOSFETs may be made of general silicon materials, or may be made using wide band gap semiconductors with lower loss. The capacitor 22 may be configured by using an aluminum electrolytic capacitor, a small capacity film capacitor, etc.

The inverter 30 has legs in which multiple switching elements 32 are connected in series, the number of which corresponds to the number of phases of the motor 40. The legs, the number of which corresponds to the number of phases, are connected in parallel to each other. The motor 40 is connected to the connection point between the switching elements 32 in each leg of the inverter 30. FIG. 1 illustrates an example in which each leg includes two switching elements 32, but the number may be any number other than two as long as it is more than one. FIG. 1 also illustrates an example in which the motor 40 is a three-phase motor, but this is not limited to this. The motor 40 may also be a multi-phase motor other than a three-phase motor.

As the switching element 32, an IGBT (Insulated Gate Bipolar Transistor), a MOSFET (Metal Oxide Semiconductor Field Effect Transistor), etc. are widely used. The example in FIG. 1 is an IGBT, to which a diode is connected in parallel. Note that in the case of a MOSFET, because of its structure it has a built-in parasitic diode, a configuration in which the diode is not connected in parallel may also be used.

Semiconductor elements made of silicon (Si) are widely used as the switching element 32, but in recent years, MOSFETs with superjunction structures have also been used in response to demands for higher efficiency. Recently, switching elements made of wide band gap semiconductors such as silicon carbide (SiC), gallium nitride (GaN), gallium oxide (Ga2O3), and diamond have also come to be used to achieve even higher efficiency. The inverter 30 may be configured using any of these switching elements as long as it is capable of performing a switching operation to apply an AC voltage to the motor 40.

The motor 40 can be, for example, an induction motor or a synchronous motor. These motors may be configured in any manner. For example, if the motor 40 is a synchronous motor, the stator may be configured with either concentrated winding or distributed winding. The windings may be made of any material, such as copper or aluminum wire, as long as they can pass current. If the motor 40 is a permanent magnet synchronous motor, the rotor may have any structure, including surface magnet and embedded magnet types, as long as it is capable of generating rotational force.

The current detection unit 74 detects the motor current, which is the current flowing through the motor 40. The DC voltage detection unit 76 detects the DC voltage applied to the inverter 30. The DC voltage applied to the inverter 30 can be detected by detecting the voltage across the capacitor 22. Representative examples of the current detection unit 74 include an ACCT (Alternating Current Transformer) that can detect only AC components, and a DCCT (Direct Current Transformer) that can detect both DC and AC components, but any device that can detect the motor current may be used.

The control unit 60 controls the inverter output voltage output by the inverter 30. Specifically, the control unit 60 generates a PWM (Pulse Width Modulation) signal, which is a signal that controls the switching of the switching element 32, based on the motor current detection value detected by the current detection unit 74 and the DC voltage detection value detected by the DC voltage detection unit 76, and sends it to the inverter 30.

Instead of the configuration of FIG. 1, the current detection unit 74 may detect the bus current flowing between the capacitor 22 and the inverter 30. In this configuration, the motor current can be calculated inside the control unit 60 by sampling the detected value of the bus current at a timing determined based on a reference signal that serves as a reference when generating a PWM signal.

FIG. 2 is a block diagram showing an example of a hardware configuration that realizes the functions of the control unit 60 according to the embodiment. FIG. 3 is a block diagram showing another example of a hardware configuration that realizes the functions of the control unit 60 according to the embodiment.

When implementing some or all of the functions of the control unit 60 according to the embodiment, the configuration can include a processor 300 that performs calculations, a memory 302 that stores programs read by the processor 300, and an interface 304 that inputs and outputs signals, as shown in FIG. 2.

When implementing the functions of the control unit 60, as shown in FIG. 2, the configuration can include a processor 300 that performs calculations, a memory 302 that stores programs read by the processor 300, and an interface 304 that inputs and outputs signals. The programs that execute the functions of the control unit 60 are held in the memory 302. The control unit 60 transmits and receives necessary information via the interface 304, and performs the control described below by having the processor 300 execute the programs held in the memory 302.

Processor 300 is an example of a computing means. Processor 300 may be a computing means called a microprocessor, a microcomputer, a CPU (Central Processing Unit), or a DSP (Digital Signal Processor). Memory 302 may be a non-volatile or volatile semiconductor memory such as a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Programmable ROM), or an EEPROM (registered trademark) (Electrically EPROM).

In addition, when implementing the functions of the control unit 60 according to the embodiment, the processing circuit 303 shown in FIG. 3 can be used. The processing circuit 303 can be a single circuit, a composite circuit, an ASIC (Application Specific Integrated Circuit), an FPGA (Field-Programmable Gate Array), or a combination of these. Information input to the processing circuit 303 and information output from the processing circuit 303 can be exchanged via an interface 304.

In addition, some of the processing in the control unit 60 may be performed by the processing circuit 303, and other processing that is not performed by the processing circuit 303 may be performed by the processor 300 and memory 302.

Next, the method of generating the PWM signal will be described. Figure 4 is a diagram used to explain the method of generating the PWM signal in the embodiment. The upper part of Figure 4 shows the waveforms of the carrier signal and the voltage command values Vu*, Vv*, and Vw* of each phase, and the lower part of Figure 4 shows the waveforms of the PWM signals UP, VP, WP, UN, VN, and WN of each phase.

The voltage command values Vu*, Vv*, Vw* are sinusoidal waveforms with phases differing by 2π/3, and the carrier signal is a triangular waveform signal that changes with a period that is the inverse of a frequency called the carrier frequency. In FIG. 1, the carrier signal is shown by a solid line, the U-phase voltage command value Vu* is shown by a dashed line, the V-phase voltage command value Vv* is shown by a dashed line, and the W-phase voltage command value Vw* is shown by a dashed line. The amplitudes of the voltage command values Vu*, Vv*, Vw* are determined by the voltage command value amplitude V*. Also, in FIG. 4, the amplitude of the carrier signal is set to 1/2 of the DC voltage Vdc applied to the inverter 30.

The control unit 60 generates PWM signals UP, VP, WP, UN, VN, and WN by comparing the amplitude of the carrier signal with the amplitudes of the voltage command values Vu*, Vv*, and Vw*. The switching elements 32 of each phase in the inverter 30 perform switching operations using these PWM signals UP, VP, WP, UN, VN, and WN, and AC voltages according to the voltage command values Vu*, Vv*, and Vw* are applied to the motor 40. Note that in FIG. 4, a high-level signal is output when the amplitudes of the voltage command values Vu*, Vv*, and Vw* are greater than the amplitude of the carrier signal, and a low-level signal is output when the amplitudes of the voltage command values Vu*, Vv*, and Vw* are smaller than the amplitude of the carrier signal, but the relationship between the high level and the low level may be reversed.

In recent years, permanent magnet motors with permanent magnets built into the rotor have been widely used, particularly for three-phase motors installed in refrigeration cycle equipment, due to the demand for higher efficiency. In such permanent magnet motors, when the rotor rotates, an induced voltage is generated, which is a generated voltage that corresponds to the rotation speed of the rotor. Figure 5 is a diagram showing an example of an induced voltage waveform that can be generated in a typical permanent magnet motor. The horizontal axis represents the phase θ of the fundamental wave component of the induced voltage, and the vertical axis represents the induced voltage.

This induced voltage is dominated by a fundamental component with a sine wave waveform, but in reality, the waveform is one in which harmonic components other than the fundamental component are superimposed. Note that in this article, harmonic components other than the fundamental component that may be included in the induced voltage are sometimes referred to as "harmonic induced voltage."

The main harmonic components other than the fundamental wave component are the 5th harmonic component, which is a frequency component five times the fundamental wave frequency f, which is the frequency of the fundamental wave component, and the 7th harmonic component, which is a frequency component seven times the fundamental wave frequency f. Figure 5 shows the combined waveform of the induced voltage, shown by the solid line, distorted by superimposing the 5th harmonic component (5f) shown by the dashed line and the 7th harmonic component (7f) shown by the dashed line on the fundamental wave component (1f) shown by the dashed line.

In typical motor control, the motor current flowing through the motor 40 is controlled to have a sinusoidal shape, thereby reducing noise and vibration of the motor 40, but this is affected by induced voltage. If the waveform of the induced voltage is distorted as shown in Figure 5, torque pulsation occurs in the motor 40, causing increased noise and vibration. Below, torque pulsation, which causes increased noise and vibration, is explained using several formulas.

In the motor 40, a U-phase induced voltage, a V-phase induced voltage, and a W-phase induced voltage are generated, each of which is 120 degrees out of phase with each other. In addition, the current driving the motor 40 can also be divided into a U-phase current, a V-phase current, and a W-phase current, taking into consideration the phase difference. Therefore, the power _Pm supplied to the motor 40 is the sum of the three-phase parts of the voltage-current products of the fundamental components of the three-phase induced voltages and the motor currents. The torque _τ1 for driving the motor 40 is determined by the power _Pm supplied to the motor 40 and acts on the motor 40.

Specifically, the above-mentioned power _Pm can be expressed as in the following equation (7) using _Eu1 , _Iu1 , _Ev1 , _Iv1 , _Ew1 , and _Iw1 represented by the following equations (1) to (6).

In the above formulas (1), (3), and (5), _Eu1 , _Ev1 , and _Ew1 respectively represent the fundamental wave component of the U-phase induced voltage, the fundamental wave component of the V-phase induced voltage, and the fundamental wave component of the W-phase induced voltage. In addition, _Iu1 , _Iv1 , and _Iw1 in the above formulas (2), (4), and (6) respectively represent the motor currents of the U-phase, V-phase, and W-phase, and θ represents the phase of the fundamental wave component of the induced voltage. In addition, _E1 represents the effective value of the fundamental wave component of the induced voltage, _I1 represents the effective value of the motor current, and φ represents the phase difference between the induced voltage and the motor current.

In the above formula (7), the effective value E ₁ , the effective value I ₁ , and the phase difference φ are constants. Therefore, it can be seen that the power P _m is only a DC component, and the AC component, which is a fluctuating component, does not appear in the above formula (7).

In this paper, the angular frequency of the rotation speed of the motor 40 is called the "mechanical angular frequency" and is represented by ω. The mechanical angular frequency ω and the rotation speed [rps] have the relationship ω = 2π × rotation speed [rps]. Using this mechanical angular frequency ω, the torque _τ1 can be expressed as the following equation (8) based on the above equation (7).

Here, if the amount of fluctuation in the mechanical angular frequency ω is represented as Δω, this amount of fluctuation Δω includes the torque τ ₁ in the above formula (8) and can be expressed by the following formula (9).

In the above formula (9), τ _load is the load torque of the motor 40, and J is the moment of inertia including the motor 40 and the load 50.

Here, when the load 50 is a propeller fan mounted on the outdoor unit of a refrigeration cycle device, the load torque τ _load is approximately constant and is balanced with the torque τ _1. Also, when the load 50 is a propeller fan, the moment of inertia J is very high and the fluctuation amount Δω of the mechanical angular frequency ω is extremely small, so the impact on the generation of vibration and noise is also extremely small. From the above, it can be seen that the torque τ ₁ generated by the fundamental wave component of the induced voltage and the fundamental wave component of the motor current has an extremely small impact on noise and vibration.

Next, let us consider the torque generated by the fifth harmonic component and the fundamental component of the motor current when the induced voltage waveform contains the fifth harmonic component described above.

First, in the same way as with the fundamental wave component, the fifth harmonic component of each phase of the induced voltage can be expressed mathematically together with the motor current of each phase as shown in the following equations (10) to (15).

In the above formulas (10), (12), and (14), _Eu5 , _Ev5 , and _Ew5 respectively represent the fifth harmonic component of the U-phase induced voltage, the fifth harmonic component of the V-phase induced voltage, and the fifth harmonic component of the W-phase induced voltage. _E5 represents the effective value of the fifth harmonic component of the induced voltage, and α represents the phase difference between the fifth harmonic component of the induced voltage and the motor current. The above formulas (11), (13), and (15) also reproduce the U-phase motor current _Iu1 , the V-phase motor current _Iv1 , and the W-phase motor current _Iw1 shown in the above formulas (2), (4), and (6).

Fig. 6 is a diagram showing the change in power due to the fifth harmonic component of the induced voltage shown in Fig. 5. Specifically, Fig. 6 shows the results of calculating power when the effective value _E5 , effective value _I1 , and phase difference α in the above formulas (10) to (15) are _E5 = 0.1, _I1 = 1.0, and α = 0, respectively. Fig. 7 is a diagram showing the results of extracting only the power ripple component from the results shown in Fig. 6. The horizontal axis in Figs. 6 and 7 represents the phase θ of the fundamental wave component of the induced voltage.

6 and 7 show that power pulsation occurs at a frequency six times the fundamental frequency f of the induced voltage. The value obtained by dividing this power pulsation component by the mechanical angular frequency ω of the motor 40 is the fluctuation in the torque output by the motor 40. For example, if the motor 40 is a 10-pole motor, the number of pole pairs (10÷2)×6=30, that is, torque pulsation occurs at a frequency 30 times the frequency of the rotational speed of the motor 40. This torque pulsation affects the rotational speed of the motor 40, and the speed fluctuation of the motor 40 makes the motor 40 a vibration source. If the frequency of the speed fluctuation of the motor 40 approaches the resonant frequency of the load 50, this will cause an increase in noise and vibration from the motor 40 and the load 50.

Next, we will explain the effect on noise and vibration when the induced voltage waveform contains the seventh harmonic component mentioned above.

Similar to the fifth harmonic component of the induced voltage, the seventh harmonic component of each phase of the induced voltage can be expressed mathematically together with the motor current of each phase as shown in the following equations (16) to (21).

In the above formulas (16), (18), and (20), _Eu7 , _Ev7 , and _Ew7 respectively represent the seventh harmonic component of the U-phase induced voltage, the seventh harmonic component of the V-phase induced voltage, and the seventh harmonic component of the W-phase induced voltage. _E7 represents the effective value of the seventh harmonic component of the induced voltage, and β represents the phase difference between the seventh harmonic component of the induced voltage and the motor current. The above formulas (17), (19), and (21) also reproduce the U-phase motor current _Iu1 , the V-phase motor current _Iv1 , and the W-phase motor current _Iw1 shown in the above formulas (2), (4), and (6).

Fig. 8 is a diagram showing the change in power due to the seventh harmonic component of the induced voltage shown in Fig. 5. Specifically, Fig. 8 shows the results of calculating power when the effective value E ₇ , the effective value I ₁ and the phase difference β in the above formulas (16) to (21) are E ₇ =0.1, I ₁ =1.0 and β=0, respectively. Fig. 9 is a diagram showing the results of extracting only the power ripple component from the results shown in Fig. 8. The horizontal axis in Figs. 8 and 9 represents the phase θ of the fundamental wave component of the induced voltage.

8 and 9 show that, as with the fifth harmonic component of the induced voltage, power pulsation occurs at a frequency six times the fundamental frequency f of the induced voltage. The value obtained by dividing this power pulsation component by the mechanical angular frequency ω of the motor 40 is the fluctuation in the torque output by the motor 40. For example, if the motor 40 is a 10-pole motor, the number of pole pairs (10÷2)×6=30, that is, torque pulsation occurs at a frequency 30 times the frequency of the rotational speed of the motor 40. This torque pulsation affects the rotational speed of the motor 40, and the speed fluctuation of the motor 40 makes the motor 40 a vibration source. If the frequency of the speed fluctuation of the motor 40 approaches the resonant frequency of the load 50, this will cause an increase in noise and vibration from the motor 40 and the load 50.

In Figs. 6 to 9, the calculation results are shown when either the 5th harmonic component or the 7th harmonic component of the induced voltage occurs alone, but in reality, the 5th harmonic component and the 7th harmonic component of the induced voltage occur simultaneously. Therefore, Figs. 10 and 11 show the calculation results when the 5th harmonic component and the 7th harmonic component of the induced voltage exist simultaneously. Fig. 10 is a diagram showing the change in power when the 5th harmonic component and the 7th harmonic component of the induced voltage shown in Fig. 5 exist simultaneously. Fig. 11 is a diagram showing the result of extracting only the power ripple component from the result shown in Fig. 10. Note that the horizontal axis in Figs. 10 and 11 represents the phase θ of the fundamental wave component of the induced voltage, as in Figs. 6 to 10.

10 shows the results of calculating the power when the effective value E ₅ , the effective value I ₁ and the phase difference α in the above formulas (10) to (15) are set to E ₅ =0.1, I ₁ =1.0 and α=0, respectively, and the effective value E ₇ , the effective value I ₁ and the phase difference β in the above formulas (16) to (21) are set to E ₇ =0.05, I ₁ =1.0 and β=0, respectively. Note that, in reality, the amplitude of the fifth harmonic component of the induced voltage is considered to be larger than the amplitude of the seventh harmonic component of the induced voltage, so the relationship E ₅ >E ₇ is set. Furthermore, in reality, the phase differences α and β are considered to be often α≠β, but since there is no loss of generality even if α=β, for convenience, α=β=0 is set.

10 and 11, as in the cases of Figs. 6 to 9, power pulsations occur at a frequency six times the fundamental frequency f of the induced voltage. Therefore, if the motor 40 is a 10-pole motor, torque pulsations occur at a frequency 30 times the frequency of the rotational speed of the motor 40, as in the cases of Figs. 6 to 9.

As described above, in motor control, if the motor current is controlled to be sinusoidal, it is possible to reduce noise and vibration of the motor 40. However, even if the motor current is controlled to be sinusoidal, if the fifth harmonic component or seventh harmonic component that may be included in the induced voltage is large, as described above, power pulsation with a frequency six times the fundamental frequency f occurs, which generates a pulsating component in the torque _τ1 output by the motor 40. Then, the difference between the torque _τ1 and the load torque _τload generates speed fluctuations, which ultimately lead to vibration and noise.

Therefore, to suppress vibration and noise in the motor 40 and the load 50, it is necessary to reduce the power pulsation that causes the vibration and noise, i.e., the power pulsation with a frequency six times the fundamental frequency f. In this paper, for the sake of convenience, the power pulsation with a frequency six times the fundamental frequency f is referred to as "sixth-order power pulsation."

There are two possible methods for reducing the sixth-order power pulsation. One is to bring the fifth-order and seventh-order harmonic components that may be included in the induced voltage of the motor 40 as close to zero as possible. The other is to control the motor current to cancel the sixth-order power pulsation. With the former method, it is difficult to reduce the harmonic components any further due to design constraints of the motor 40. Therefore, this paper proposes the latter method. Specifically, this is realized by the control systems shown in Figures 12 and 13. Figure 12 is a diagram showing an example of the configuration of a control unit 60 according to an embodiment. Figure 13 is a diagram showing an example of the configuration of a torque fluctuation compensation amount calculation unit 66 provided in the control unit 60 shown in Figure 12.

As shown in FIG. 12, the control unit 60 includes adders/

subtractors

61, 63, 64, and 65, a speed control unit 62, a torque fluctuation compensation amount calculation unit 66, a coordinate conversion unit 67, a d-axis current control unit 68, a q-axis current control unit 69, a speed and position estimation unit 70, and a PWM signal generation unit 71. Each adder/subtractor performs an addition or subtraction calculation according to the plus (+) or minus (-) sign next to it.

The motor currents Iu, Iv, and Iw of the UVW phases are input to the coordinate conversion unit 67 as detected values of the motor currents. The coordinate conversion unit 67 calculates the d-axis current Id and the q-axis current Iq by converting the motor currents Iu, Iv, and Iw into current values on the dq coordinates using the electrical angle phase θe generated by the speed and position estimation unit 70 described below.

The speed and position estimator 70 calculates a speed estimate ωe based on the d-axis current Id and the q-axis current Iq, as well as the d-axis voltage command value Vd* and the q-axis voltage command value Vq* described below. The speed estimate ωe is an estimate of the rotational speed of the motor 40. The speed of the motor 40 varies with load fluctuations. Therefore, the speed and position estimator 70 estimates the rotational speed of the motor 40, which varies with load fluctuations, and outputs a speed estimate ωe corresponding to the estimated rotational speed. The speed and position estimator 70 also calculates an electrical angle phase θe based on the speed estimate ωe. The electrical angle phase θe can be obtained by integrating the speed estimate ωe.

The torque fluctuation compensation amount calculation unit 66 generates a torque compensation current ΔIq, which is a compensation amount for making the torque fluctuation component zero, based on the motor currents Iu, Iv, Iw, the electrical angle phase θe, and the speed estimate ωe. The configuration and operation of the torque fluctuation compensation amount calculation unit 66 will be described in detail later.

Adder-subtracter 61 calculates the speed deviation, which is the deviation between the speed command value ω* and the speed estimate value ωe. Speed control unit 62 calculates the q-axis current command value Iq* based on the speed deviation. Adder-subtracter 63 adds the q-axis current command value Iq* and the torque compensation current ΔIq, and adder-subtracter 65 calculates the q-axis current deviation by subtracting the q-axis current Iq from the output of adder-subtracter 63. q-axis current control unit 69 generates a q-axis voltage command value Vq* that converges the q-axis current deviation to zero and outputs it to the PWM signal generation unit 71.

The adder/subtractor 64 calculates the d-axis current deviation by subtracting the d-axis current Id from the d-axis current command value Id*. The d-axis current control unit 68 generates a d-axis voltage command value Vd* that converges the d-axis current deviation to zero, and outputs it to the PWM signal generation unit 71.

The PWM signal generating unit 71 converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into voltage command values in a three-phase coordinate system using the electrical angle phase θe, and generates PWM signals UP, VP, WP, UN, VN, and WN based on the converted voltage command values in the three-phase coordinate system and the DC voltage Vdc, and outputs them to the inverter 30.

Next, we will explain the configuration and operation of the torque fluctuation compensation amount calculation unit 66. Figure 13 shows an example of the configuration of the torque fluctuation compensation amount calculation unit 66, and we will explain the reason for configuring it in this way.

First, assuming that the speed fluctuation of the motor 40 due to the power pulsation is small, the torque fluctuation of the motor 40 due to the power pulsation can be easily restored by grasping the parameters of the induced voltage described above, that is, the effective value _E1 of the fundamental wave component, the effective value _E5 of the fifth harmonic component, the effective value _E7 of the seventh harmonic component, and the phase differences α and β of the induced voltage. Therefore, as shown in Fig. 13, a processing unit for restoring the induced voltage is provided, and the restored induced voltages Eu, Ev, and Ew are multiplied by the motor currents Iu, Iv, and Iw for each phase, and the three phases are added to calculate the instantaneous power. Then, a band pass filter (BPF) is provided to extract only the 6-fold fluctuation component, and the BPF extracts the component of the 6-fold frequency of the fundamental wave frequency f of the induced voltage from the instantaneous power, thereby obtaining the component of the 6-fold power pulsation. The torque fluctuation component Δτ is obtained by dividing the sixth-order power pulsation component by the mechanical angular frequency ω. The phase θ used for the restoration induced voltages Eu, Ev, and Ew can be obtained by multiplying the electrical angle phase θe by the reciprocal of the number of pole pairs P of the motor 40. The mechanical angular frequency ω used in the calculation of the torque fluctuation component Δτ can be obtained by multiplying the speed estimate ωe by the reciprocal of the number of pole pairs P of the motor 40.

There are various possible control methods for suppressing the torque fluctuation component Δτ, but here we will explain one example.

Once the torque fluctuation component Δτ is known, the difference can be taken with zero as the target, proportional-integral control can be performed, and the output can be output as the torque compensation current ΔIq, as shown in Figure 13. Then, the torque compensation current ΔIq calculated in Figure 13 can be added to the q-axis current command value Iq*, as shown in Figure 12.

The above-mentioned process can be expressed in the form of a flowchart as shown in FIG. 14. FIG. 14 is a flowchart showing the flow of processing by the control unit 60 in the embodiment.

First, in a current detection step S001, the motor current is detected. In a position and speed estimation step S002, the phase θ, which is the position information of the motor 40, and the mechanical angular frequency ω, which is the information of the rotation speed of the motor 40, are estimated. In an induced voltage restoration step S003, the induced voltage is restored based on the effective value E ₁ of the fundamental wave component, the effective value E ₅ of the fifth harmonic component, the effective value E ₇ of the seventh harmonic component, and the phase differences α and β, which are previously measured, and the phase θ, which is the estimated position information of the motor 40. In a torque fluctuation component calculation step S004, the instantaneous power is calculated based on the restored induced voltage and the motor current, and the calculated instantaneous power is passed through a filter having a desired band to calculate the torque fluctuation component Δτ. In a q-axis current command value calculation step S005, a torque compensation current ΔIq for making the torque fluctuation component Δτ zero is calculated, and the calculated torque compensation current ΔIq is reflected in the q-axis current command value. By carrying out the above steps, vibration and noise in the motor 40 are reduced.

Next, the effects of the processing by the control unit 60 in the above-described embodiment will be described. First, by grasping in advance the fundamental wave component, the fifth harmonic component, and the seventh harmonic component of the induced voltage that may occur in the motor 40, it is possible to calculate the extent to which the torque fluctuation of the sixth harmonic component will occur in the motor 40. Then, by compensating the q-axis current command value Iq* so that the torque fluctuation becomes zero, it is possible to easily suppress the noise and vibration caused by the sixth harmonic component without complex control. This eliminates the need for measures such as vibration-proofing materials to prevent vibration of the motor 40 and sound-absorbing materials to absorb noise. This makes it possible to avoid an increase in the cost of the device and to obtain an inexpensive and highly reliable motor drive device.

Furthermore, when the load is an air conditioner or a heat pump water heater, the propeller fan mounted on the outdoor unit is a high inertia load because its radial length is longer than the direction of rotation of the motor 40. Furthermore, the propeller fan is thin, so vibrations are easily transmitted. Furthermore, since the propeller fan is made of resin, it has the characteristic that its hardness changes depending on the temperature. Since the outdoor unit is installed outdoors, not only is it easily affected by the outside temperature, but if it is installed in a place exposed to direct sunlight, the daytime temperature will be high. For this reason, the propeller fan mounted on the outdoor unit has the characteristic that its resonance point fluctuates.

As mentioned above, when the motor 40 is a 10-pole motor, the fifth and seventh harmonic components of the induced voltage generate torque pulsations 30 times the rotation speed. For this reason, conventionally, the operating point of the motor 40 was determined by considering the relationship between the resonance point due to temperature changes and the torque pulsations according to the rotation speed. However, with this method, the range of operating points that can be set is narrow, making it difficult to set an operating point that provides maximum efficiency. In addition, with this method, changing the operating point requires changing the actuator operation for other refrigeration cycle equipment with similar specifications, which increases the time required for equipment adjustment work.

In response to the above problems, by using the control method of the above embodiment, it is possible to suppress torque pulsation 30 times the rotational speed caused by the fifth and seventh harmonic components that may be included in the induced voltage of the motor 40, and to suppress the generation of vibration and noise in the motor 40. This eliminates the constraint of operating the motor 40 at a rotational speed other than the resonant frequency, making it possible to shorten the time required to adjust the equipment. Furthermore, even if the frequency of the maximum efficiency point of the motor 40 and the resonant frequency point of the motor 40 are close to each other, it is possible to operate the motor 40 at the maximum efficiency point while suppressing the generation of vibration and noise, thereby making it possible to increase the operating efficiency of the motor 40.

In the case of the configuration of FIG. 12, since only the q-axis current command value Iq* is compensated, only the q-axis voltage command value Vq* is manipulated, while the d-axis voltage command value Vd* is not manipulated, so there is a possibility that a voltage error will occur in the voltage applied to the motor 40. Therefore, in order to reduce the effect of this voltage error, the configuration of the control unit 60A shown in FIG. 15 is proposed. FIG. 15 is a diagram showing an example of the configuration of the control unit 60A according to a modified example of the embodiment.

Comparing the control unit 60A shown in FIG. 15 with the control unit 60 shown in FIG. 12, a non-interference control unit 80 and an adder/subtractor 72 have been added in FIG. 15. The other configurations are the same or equivalent to those of the control unit 60, and the same or equivalent components are denoted by the same reference numerals, and duplicated explanations will be omitted.

The non-interference control unit 80 includes a multiplier 80a. The non-interference control unit 80 calculates a compensation value Vdff* for the d-axis voltage command value Vd* based on the compensated q-axis current command value Iq* output from the adder/subtractor 63, that is, the q-axis current command value Iq* to which the torque compensation current ΔIq has been added, and the speed estimate ωe. The compensation value Vdff* for the d-axis voltage command value Vd* is a compensation value for suppressing mutual interference with the d-axis due to the compensated q-axis current command value Iq*. As shown in FIG. 15, the compensation value Vdff* for the d-axis voltage command value Vd* can be obtained by multiplying the compensated q-axis current command value Iq* by the q-axis inductance Lq of the motor 40 and the speed estimate ωe. By using the control unit 60A shown in FIG. 15, it is possible to reduce the effect of the voltage error on the motor applied voltage caused by the compensation of the q-axis current command value Iq*.

As described above, the motor drive device according to the embodiment includes an inverter that converts a DC voltage applied from a DC power supply into an AC voltage and applies it to a motor, and a control unit that controls the inverter output voltage output by the inverter. The control unit controls the inverter output voltage so as to suppress the torque fluctuation component generated from the product of the fundamental component of the motor current, which is the current flowing through the motor, and at least one harmonic induced voltage of the fifth harmonic component and the seventh harmonic component. This makes it possible for the motor drive device to easily suppress the generation of vibrations and noise, even when applied to equipment with large load fluctuations.

In the above configuration, the control unit can control the motor current to be sinusoidal, and can compensate the q-axis current command value based on a torque compensation current, which is a compensation amount generated based on the torque fluctuation component. The torque compensation current here is a compensation amount for making the torque fluctuation component zero when the motor is driven. When performing this control, the control unit may be equipped with a non-interference control unit that compensates for the d-axis voltage command value based on the compensated q-axis current command value and a speed estimation value that is an estimate of the motor rotation speed. By providing such a non-interference control unit, it is possible to reduce the effect of voltage error on the motor applied voltage caused by compensation of the q-axis current command value.

The configurations shown in the above embodiments are merely examples, and may be combined with other known technologies. Parts of the configurations may be omitted or modified without departing from the spirit of the invention.

10 AC power supply, 20 reactor, 21 rectifier, 22 capacitor, 30 inverter, 32 switching element, 40 motor, 50 load, 60, 60A control unit, 61, 63, 64, 65, 72 adder/subtractor, 62 speed control unit, 66 torque fluctuation compensation amount calculation unit, 67 coordinate conversion unit, 68 d-axis current control unit, 69 q-axis current control unit, 70 speed and position estimation unit, 71 PWM signal generation unit, 74 current detection unit, 76 DC voltage detection unit, 80 non-interference control unit, 80a multiplier, 100 motor drive device, 300 processor, 302 memory, 303 processing circuit, 304 interface.

Claims

1. A motor drive device that drives a motor that generates at least one harmonic voltage selected from a fifth harmonic component and a seventh harmonic component of an induced voltage in addition to a fundamental component of the induced voltage, comprising:
an inverter that converts a DC voltage applied from a DC power supply into an AC voltage and applies the AC voltage to the motor;
A control unit that controls an inverter output voltage output by the inverter;
Equipped with
The control unit controls the inverter output voltage so as to suppress a torque fluctuation component generated from a product of a fundamental wave component of a motor current, which is a current flowing through the motor, and at least one of the fifth harmonic component and the seventh harmonic component.
The motor drive device according to claim 1 , wherein the control unit controls the motor current to be sinusoidal, and compensates a q-axis current command value based on a compensation amount generated based on the torque fluctuation component.
The motor drive device according to claim 2 , wherein the control unit includes a non-interference control unit that compensates for the d-axis voltage command value based on a compensated q-axis current command value and a speed estimate that is an estimate of a rotation speed of the motor.
A refrigeration cycle device equipped with a motor drive device according to any one of claims 1 to 3.
The refrigeration cycle equipment is provided in an outdoor unit of an air conditioner or a heat pump water heater,
The refrigeration cycle equipment according to claim 4 , wherein the motor drives a propeller fan in the outdoor unit.