TWI282209B - Controller for synchronous motor, electric appliance and module - Google Patents

Controller for synchronous motor, electric appliance and module Download PDF

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Publication number
TWI282209B
TWI282209B TW093138591A TW93138591A TWI282209B TW I282209 B TWI282209 B TW I282209B TW 093138591 A TW093138591 A TW 093138591A TW 93138591 A TW93138591 A TW 93138591A TW I282209 B TWI282209 B TW I282209B
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TW
Taiwan
Prior art keywords
synchronous motor
controller
control device
periodic
component
Prior art date
Application number
TW093138591A
Other languages
Chinese (zh)
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TW200524265A (en
Inventor
Yoshitaka Iwaji
Tsunehiro Endo
Yasuo Notohara
Yuhachi Takakura
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Hitachi Appliances Inc
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Application filed by Hitachi Appliances Inc filed Critical Hitachi Appliances Inc
Publication of TW200524265A publication Critical patent/TW200524265A/en
Application granted granted Critical
Publication of TWI282209B publication Critical patent/TWI282209B/en

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Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/05Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for damping motor oscillations, e.g. for reducing hunting
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24Vector control not involving the use of rotor position or rotor speed sensors
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022Synchronous motors
    • H02P25/024Synchronous motors controlled by supply frequency
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/10Arrangements for controlling torque ripple, e.g. providing reduced torque ripple
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14Electronic commutators
    • H02P6/16Circuit arrangements for detecting position
    • H02P6/18Circuit arrangements for detecting position without separate position detecting elements
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02PCONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05Synchronous machines, e.g. with permanent magnets or DC excitation

Abstract

This invention provides the controller of a synchronous motor. It realizes a low vibration and low noise variable speed drive by suppressing a periodic interference in a driver. When a load unit generates the periodic interference in the driver, it calculates the rotor position of an AC synchronous motor, and by means of detecting a rotating speed directly. The resolving method is that: a sensorless drive is realized by calculating a difference (axial error) between the position of the magnetic flux axis of the AC synchronous motor. The position of the magnetic flux axis assumed in a controller, and correcting the rotating speed, such that the difference becomes zero. Further, according to the calculated value of this axial error, and by means of compensating, it is performed by providing a means for extracting the pulsation component of the torque generated from the motor or the load unit.

Description

1282209 (1) 九、發明說明 【發明所屬之技術領域】 本發明係關於同步電動機之控制裝置、電器及模組。 【先前技術】 交流電動機之速度或者不使用位置感測器之控制方 式’至目則爲止’公開有各種手法。例如,在以交流電動 機之代表例的永久磁鐵同步電動機爲對象之例子中,有曰 本專利特開2 00 1 -2 5 1 8 8 9號公報等之方式爲所周知。此控 制方式係不使用位置感測器,代之在控制器內部進行磁極 位置的推測運算的方式。 另外,電動機之負載裝置所產生之週期性轉矩干擾的 控制方法,則有日本專利特開平1 0 - 1 7 4 4 8 8號公報、日本 專利特開 2002-3 4290號公報等。日本專利特開平ιοί 7 4 4 8 8 號公報 所記載 之方式 ,係 抽出 包含在 電動機 之速度 檢測値之脈動成分,在換流器輸出電壓加上補正以將其抵 消之方式。在此方式之實現上,需要速度資訊。 日本專利特開 2002 -3 4290號公報之方式,係檢測包 含在轉矩電流成分之脈動成分,藉由在旋轉速度加上補 正,以穩定地控制電動機之方式。 [專利文獻1]日本專利特開2 0 0 1 -2 5 1 8 8 9號公報 [專利文獻2]日本專利特開平10- 1 744 8 8號公報 [專利文獻3]日本專利特開2 002- 3 4 290號公報 (2) 1282209 【發明內容】 [發明所欲解決之課題] 在曰本專利特開2001-251889號公報之方式中,雖可 實現無位置感測器’但是’在負載裝置連接壓縮機等之週 期性干擾所伴隨之負載的情形,無法抑制該週期性干擾。 其結果爲,產生旋轉脈動,而有變成裝置之振動' 噪音的 原因之課題。 曰本專利特開平1 0 - 1 7 4 4 8 8號公報之方式,雖可抑制 週期性干擾’但是,需要電動機之旋轉速度資訊。因此, 需要某種之速度檢測器。原理上,雖可安裝全1C等之位 置感測器,使用於電動機之速度檢測,但是,在負載裝置 爲空調等之壓縮機的情形,由於周圍環境之問題,感測器 之安裝有困難。 代替位置感測器,檢測電動機之中性點電位,由其之 變動成分以獲得速度資訊之方法雖也爲所周知,但是,速 度資訊以電氣角而言,只能每6 0度獲得,高速、高精度 之速度檢測有困難。特別是,基於驅動電動機之換流器之 導通延遲(空載期間)之影響所致之週期性干擾,對於電 動機之驅動頻率,係以6倍之頻率變動故,以電氣角60 度間隔之速度檢測,不可能抑制此干擾。另外,獲得中性 點電位用之配線,則有需要多1條線之課題。 日本專利特開2002 - 3 42 90號公報之方式係因應包含 在轉矩電流之脈動,改變旋轉速度本身,以提升控制裝置 整體之穩定性之方式。因此,旋轉脈動進一步增加,振 -6- (3) (3)1282209 動、噪苜的課題無法解決。另外,對象爲感應電動機故, 照這樣’則難於適用在同步電動機。 本發明之目的在於提供:可以抑制週期性干擾所引起 之振動、噪音之電動機之控制裝置。 [解決課題用手段] 本發明之特徵之一爲具有,在同步電動機之控制裝置 中’依據軸誤差推算値,求得前述電動機或者負載之某一 方’或者雙方所產生的週期性干擾成分之週期性干擾推算 器。 [發明之效果] 如依據本發明,可以實現能抑制週期性干擾所引起之 振動、噪音之電動機之控制裝置。 【貫施方式】 接著’參照第1圖至第1 5圖,說明依據本發明之交 流電動機之控制裝置的實施例。另外,在以下之實施例 中,電動機雖使用永久磁鐵型同步電動機(以下,省略爲 PM電動機)做說明,但是,關於其他之同步電動機(例 如,繞線型同步電動機、磁阻電動機等),也可同樣地加 以實現。 [實施例1] (4) 1282209 第ί 11係顯示依據本發明之交流電動機控制 施例1之系統構造方塊圖。本實施例!之控制裝 藉由上位控制裝置之指令1 0 0,對電動機給予旋 wr*之旋轉數指令產生器1,及運算電動機之交 壓’轉換爲脈波寬度調變波訊號(PWM訊號) 出之控制器2,及藉由此PWM訊號所驅動之換恭 對換流器3供給電力之轉換器4,及控制對象之 機5 ’及PM電動機之負載之壓縮機6,及檢測 對換流器3供給之電流1 〇之電流檢測器7所形成 控制器2係依據藉由電流檢測器7所檢測之 在控制器內部運算流經PM電動機5之三相交流 I v、I w而加以再現之電流再現器8,及將所再現 流電流 Iuc、Ivc、Iwc藉由相位角0 dc (在控制 假定之P Μ電動機的磁鐵磁通之位置)予以座標 軸上之成分Idc、Iqc之dq座標轉換器9,及對 之電流成分,給予指令I q *之I q *產生器1 〇,及 對於d軸上之電流成分,給予指令I d *之I d *產^ 及依據Id*、Iq*、以及電氣角頻率指令ω 1*,運 令Vdc*、Vqc*之電壓指令運算器12,及將Vdc* 換爲三相交流電壓指令 Vu*、Vv*、Vw*之dq 1 3,及依據三相交流電壓指令,產生開關換流器 波寬調變訊號(PWM訊號)之PWM脈波產生器 算相當於P Μ電動機5之磁鐵磁通位置β d與在 內部所假定之位置Θ dc之誤差的角度(軸誤差) 裝置之實 置係由: 轉數指令 流施加電 而予以輸 [器3,及 PM電動 轉換器4 ;〇 電流1 〇, 電流Iu、 之三相交 器內部所 轉換爲各 於q軸上 同樣地, 器 11, 算電壓指 、Vqc*轉 逆轉換器 3用之脈 1 4,及推 控制器2 △ β之 (8)1282209 此處, 期性成分。 週期性 通不均勻, 情形,等效 換流器之臂 等,也以換 另外, 空調等所使 載。在往復 期,負載激 爲了控 期性轉矩變 係以某種手 脈動成爲零 獲得速度資 獲得以電氣 但是, 之資訊,作 度份之延遲 於基於電動 爲比電氣角 其抑制有其 另外’ 考慮在電壓千擾VD或者負載轉矩TL包含有週 之電壓千擾V D,例如在p μ電動機之磁鐵磁 有導磁偏差之情形,或者捲線之相位間偏差之 性成爲週期性電壓干擾而造成影響。或者基於 短路防止期間(空載時間)之影響所致之干擾 流器之驅動頻率的6倍頻率而產生。 週期性負載轉矩干擾例如可以思考在冷凍庫或 φ 用之往復式壓縮機,或單旋轉式壓縮機等之負 式壓縮機之情形,以電動機之一旋轉爲一週 烈變動。 制性地抑制這些振動、噪音,如構成前述之週 動成爲零之控制系統即可。在習知的發明中, 段檢測旋轉速度資訊,控制施加電壓令該旋轉 而加以對應。在空調等之壓縮機中,難於直接 訊故,所以檢測電動機之中性點電位的變動, 角6 0度刻度之資訊,以推算速度。 在此方式中,對於電氣角週期,只能獲得6點 爲速度資訊並不充分。在此狀態下,產生6 0 的影響,或在速度檢測精度出現問題。或者對 機之感應電動勢電壓之畸變所產生之脈動,成 週期短之週期(主要爲1 / 6週期)故,要將 : 困難。 ; 雖也可考慮驅使控制理論,構築附在干擾觀測 -12- (9) 1282209 器等,以推算脈動轉矩之手法,但是,在此情形,觀測器 本身之響應頻率變成課題。脈動轉矩之頻率高之情形,因 應其,也需要提高觀測器之設定響應。脈動轉矩之頻率成 分變得愈高,則觀測器之高響應性更被要求,結果爲,需 要高速運算處理。因此,作爲目前爲止之週期性干擾之抑 制方法’一般在低速領域之振動抑制雖屬可能,但是,高 速旋轉時之抑制很難。 舉其一例’考察利用泛用微電腦以構成觀測器之情 形。在設觀測器響應時間爲1 m s ( 1 0 0 0 r a d / s —約1 5 0 Η z ) 之情形,可檢測之脈動轉矩爲3 0 Η ζ之程度。如將其設爲 4極之電動機時,則變成900 [r/min]。在壓縮機之情形, 最高旋轉數多數在3 000 [r/min]以上故,如不在30%程度之 速度以下,則變得無法適用。 在本發明中,著眼於第4圖之方塊圖,提出由軸誤差 △ 0以推算轉矩脈動成分△ Tm之手法。軸誤差△ 0係瞬 間可運算瞬間値故’可不受到運算延遲之影響,而做高精 度之推昇。另外’封於驅動頻率,對於高的頻率成分(例 如’ 6倍之振動成分),也可以檢測爲其特徵。此結果 爲’與習知之週期性干擾控制方法相比,大幅至高速領域 之推算變成可能。 在此種週期性干擾產生之情形,電動機轉矩Tm與負 載轉矩T L之差’係變成週期性之轉矩變動,成爲振動、 噪音之原因。在抑制此振動、噪音上,例如,需要以吸音 材包圍裝置整體等之對策’變成裝置之大型化,以及成本 -13- (10) 1282209 增加,係一種需要解決之課題。 在控制性地抑制振動、噪音上,如構成令前述之週期 性轉矩變動成爲零之控制系統即可。在習知的發明中,以 某種手段檢測旋轉速度,控制施加電壓以令該旋轉脈動成 爲零而加以對應。但是,在空調等之壓縮機中,電動機係 組裝在壓縮機內部故,難於簡單地獲得速度資訊,另外, 即使可以獲得’頂多只能獲得相當於電氣角6 0度刻度之 資訊.。因此,高精度化有其困難。 在本發明中,著眼於第4圖之方塊圖,提出由軸誤差 △ (9以推算轉矩脈動成分△ T m之手法。軸誤差△ 0係瞬 間可運算瞬間値故,可不受到運算延遲之影響,而做高精 度之推算。另外,對於驅動頻率,對於高的頻率成分(例 如,6倍之振動成分),也可以檢測爲其特徵。 第5圖係分別顯示負載轉矩TL以角頻率ω d而在正 弦波狀包含振動之成分的情形之轉矩脈動成分(△ Tm )、 旋轉速度變動(△ ω r )、軸誤差(△ 0 )。如考慮穩定 狀態時’ Tm與TL之平均値係一致,△ Tm只是振動成分 而已(第5(b)圖)。包含於旋轉速度之振動成分ΔωΓ 係積分此△ Tm者,與△ Tm相比,變成相位延遲9 0度之 波形。振動之大小本身雖依據慣量J而變化,但是,相位 可認爲幾乎延遲9 0度。軸誤差△ Θ係變成進而積分 △ ω r ’令符號反轉者(以第2圖所示定義之關係,反轉 符號)故,相位變成前進9 0度(以積分而延遲9 0度,符 號反轉故,變成前進9 0度)。即△ T m之變動成分,在 -14- (11) 1282209 △ 0中’係變成同相位之振動波形而被觀測到。如由方塊 線圖導出此關係時,變成如下。 第6(a)圖係顯示由△丁〇1至^0之方塊圖。藉由逆 轉換此方塊圖,可以求得由△ 0至△ 之傳達函數,變 成如同圖(c)般。 依據弟6(c)圖而求得△Tm時,由△ <9dc( △ (9之 推算値)可直接推算轉矩之脈動成分。但是,二階微分 △ 0dc實際上不可能。△ecu原本爲推算値,也多數包 φ 含有檢測値之雜訊等故,使用微分會增加推算誤差,另 外,也有來自運算週期之界限。 因此,著眼於「干擾成分爲週期函數」之點,將s = j ω d代入第 6 ( c )圖。如此一來,如第 6 ( d )圖所示 般’變成可推算將△ Θ予以常數倍者會成爲△ Tm。此結果 爲,與第5圖之(b )與(d )之波形的關係一致。 具體化第6 ( d )圖之構造係第7圖所示之△ Tm推算 器21 (週期性干擾推算器)°ΔΤγπ推算器21(週期性干 鲁 擾推算器)係由將△ <9 d c予以2 J7 Ρ倍之比例增益2 1 1,及 2個乘法器212所形成,實施第6(d)圖之運算。 依據第7圖,由△ 0 dc可以推算包含在△ Tm之角速 度ω d之週期性干擾成分。 接著,說明抑制此△ Tm之轉矩控制器 22 (第 8 圖)。 轉矩控制器所必要之條件爲, : (1 ) 對於週期性千擾成分,追從性高, -15- (16) 1282209 脈動成分△ Tmc在以(數學式1 )被座標轉換後,成 爲直流量故,可以積分控制器2 2 5去掉偏差。即此轉矩控 制器如來外部來看,在角頻率^3中’與增益變成無限大 之補償要素爲等效。即變成具有與實施例1之轉矩控制器 2 2同等之頻率特性。 在轉矩控制器22C之情形,與第8圖或第9圖之轉矩 控制器相比,調整處所成爲一次延遲濾波器之時間常數 TATr與積分控制器22 5之增益Ki ATR之2處。但是,TATR 對於ω d可以選擇充分大之時間常數故,調整方法並不特 別難。另外,KiATR之値係直接變成決定脈動成分抑制之 響應時間,控制響應時間對於KiATR之値,成爲線性。此 結果爲,可以獲得增益設定變得容易之效果。 [實施例4] 接著,利用第1 1圖,說明依據本發明之實施例4。 在實施例3中,提供對於振動頻率ω d,增益變成無 限大之轉矩控制器。此係動作上與實施例1之轉矩控制器 (第8圖)等效。因此’產生與實施例2中所記載者相同 之課題。即包含在轉矩脈動之ω d成分雖被去除,代之, PM電動機的驅動電流之畸變變大,容易產生pM電動機 之效率劣化,或者基於峰値電流之過電流跳脫等之不良。 因此’與實施例2相同’提出將角頻率ω d之增益由 無限大變成有限之方法。 第1 1圖係顯示實施例4之轉矩控制器2 2 D的構造。 -20- (17) 1282209 藉由代替第1圖之轉矩控制器2 2而使用本轉矩控制器 2 2 D,實施例4得以實現。 第1 1圖之轉矩控制器22D與第1 0圖之轉矩控制器 2 2C之差異爲,積分控制器22 5被變更爲不完全積分控制 器22 5 D之點。依據不完全積分器22 5 D內之時間常數Ti 與增益KiATR,峰値受到抑制。此結果爲,變成可調整ω d 成分之干擾抑制效果,在噪音、振動與P Μ電動機相電流 之畸變的最佳點之驅動變成可能。 φ [實施例5] 接著,利用第1 2圖,說明依據本發明之實施例5。 在實施例1〜4中,敘述了依據軸誤差△ 0之推算 値’以推算、抑制週期性轉矩脈動成分之方法。主要之脈 動成分雖出現於I q c或軸誤差推算値,但是,對於〗d ^也 會產生影響。 ^ d軸電流雖然對於轉矩沒有貢獻,但是,依據轉矩脈 馨 動,旋轉軸產生偏差’在d軸方向也產生基於脈動之電 流。實施例5便是利用此以進而降低轉矩脈動之實施例。 第1 2圖中’控制器2 E係與實施例1之控制器2幾乎 相同。新追加進行d軸(dc軸)之電流控制之^軸電流控 制器IdACR ( 22C ) 。22C例如係導入與第1〇圖所示之轉 矩控制器22C完全相同者(增益KiATR需要調整),輸入 / I d c以代替△ Tm c,將輸出加在I d *。在電壓指令運算器丨9 : 中’將I d * *當成新的指令値,進行電壓指令之運算。 -21 - (20) 1282209 另外,實施例雖以空調爲例做說明,但是,在其他之 電器,例如套裝空調或冷凍庫等之情形,也可以獲得同樣 的效果。 如前述般,如依據本發明,不使用檢測同步電動機之 旋轉速度或旋轉軸位置之感測器,可以實現抑制負載裝置 或電動機本身所產生之週期性轉矩干擾之高性能的電動機 驅動。另外’即使在有檢測同步電動機之旋轉速度或旋轉 軸位置之感測器的情形,也可以同樣地加以實現。 【圖式簡單說明】 第1圖係顯示依據本發明之同步電動機控制裝置之實 施例1的系統構造方塊圖。 第2圖係顯示依據本發明之同步電動機控制裝置之實 施例的軸誤差△ 0之定義向量圖。 第3圖係顯示依據本發明之同步電動機控制裝置之實 施例1的軸誤差推算器之內部構造方塊圖。 第4圖係說明依據本發明之交流電動機控制裝置之實 施例1之由對於電動機之施加電壓至軸誤差發生之原理的 方塊圖。 第5圖係顯示依據本發明之交流電動機控制裝置之實 施例1的週期性千擾轉矩,以及其所引起產生之旋轉脈 動、軸誤差變動的原理之波形圖。 第6圖係說明依據本發明之交流電動機控制裝置之實 施例1的脈動轉矩成分之推算原理方塊圖。 -24 - (21) 1282209 桌7圖係顯示依據本發明之同步電動機控制裝置之實 施例1的△ T m推算器之內部構造方塊圖。 第8圖係顯示依據本發明之同步電動機控制裝置之實 施例1的轉矩控制器之內部構造方塊圖。 第9圖係顯示依據本發明之同步電動機控制裝置之實 施例2之轉矩控制器的內部構造方塊圖。 第1 〇圖係顯不依據本發明之同步電動機控制裝置之 實施例3之轉矩控制器的內部構造方塊圖。 第1 1圖係顯示依據本發明之同步電動機控制裝置之 貫施例4之轉矩控制器的內部構造方塊圖。 第1 2圖係顯示依據本發明之同步電動機控制裝置之 實施例5的控制器之內部構造方塊圖。 第1 3圖係顯不適用依據本發明之同步電動機控制裝 置之實施例6的外觀構造圖。 第1 4圖係顯示將依據本發明之同步電動機控制裝置 適用於空調之實施例7之外觀構造圖。 第1 5圖係顯不在啓動依據本發明之空調的壓縮機, 令旋轉速度改變之情形,噪音的變化及電流波形之變化的 一例圖。 【主要元件符號說明】 1 :旋轉數指令產生器,2 :控制器,3 :換流器,4 : 轉換器,5 : P Μ電動機,6 :壓縮機,7 :電流檢測器, 8 :電流再現器,9 : dq座標轉換器,: 產生器, -25- (22) 1282209 1 1 : Id*產生器,12 :電壓指令運算器,13 : dq逆轉換 器,1 4 : P WM脈波產生器,1 5 : △ 0推算器,1 6 :加減 法器,1 7 :零指令產生器,1 8 :比例補償器,1 9 :轉換增 益,20:積分器,21 : ΔΤιη推算器,22:轉矩控制器, 41 :交流電源,42 :二極體橋,43 :平滑電容器.1282209 (1) Description of the Invention [Technical Field of the Invention] The present invention relates to a control device, an electric appliance and a module of a synchronous motor. [Prior Art] Various methods are disclosed for the speed of the AC motor or the control method of the position sensor without using it. For example, in the case of a permanent magnet synchronous motor which is a representative example of an AC motor, a method such as the one disclosed in Japanese Patent Laid-Open No. Hei 2 00 1 - 2 5 1 8 8 9 is known. This control method is a method in which the position sensor is not used and the speculative operation of the magnetic pole position is performed inside the controller. Further, a method of controlling the periodic torque disturbance generated by the load device of the electric motor is disclosed in Japanese Laid-Open Patent Publication No. Hei. No. Hei. No. Hei. No. 2002-3 4290. In the method described in Japanese Laid-Open Patent Publication No. PCT No. 4 4 4 8 No., the pulsation component included in the speed detection of the motor is extracted, and the inverter output voltage is corrected to cancel it. In this implementation, speed information is required. In the method of detecting the pulsation component included in the torque current component, the pulsation component included in the torque current component is corrected by the rotation speed to stably control the motor. [Patent Document 1] Japanese Patent Laid-Open Publication No. JP-A No. Hei. No. Hei. No. Hei. - Japanese Patent Application Publication No. 2001-251889 When the device is connected to a load accompanied by periodic disturbance of a compressor or the like, the periodic disturbance cannot be suppressed. As a result, there is a problem that the rotation pulsation is generated and the vibration of the device is caused. The method of the Japanese Patent Laid-Open Publication No. Hei No. Hei 10- 1 7 4 4 8 8 can suppress the periodic interference. However, the rotational speed information of the motor is required. Therefore, some kind of speed detector is needed. In principle, a position sensor such as a full 1C can be mounted for speed detection of the motor. However, in the case where the load device is a compressor such as an air conditioner, the sensor is difficult to install due to problems in the surrounding environment. Instead of the position sensor, the method of detecting the neutral point potential of the motor and the variable component thereof to obtain the speed information is well known. However, the speed information can only be obtained every 60 degrees in terms of the electrical angle, and the speed is high. High-precision speed detection is difficult. In particular, based on the periodic disturbance caused by the influence of the conduction delay (no-load period) of the inverter that drives the motor, the driving frequency of the motor is changed by a frequency of 6 times, and the speed is 60 degrees at an electrical angle. Detection, it is impossible to suppress this interference. In addition, when wiring for neutral potential is obtained, there is a problem that one more line is required. The method of Japanese Laid-Open Patent Publication No. 2002-3442 90 is based on the pulsation of the torque current to change the rotational speed itself to improve the stability of the overall control device. Therefore, the rotational pulsation is further increased, and the problem of vibration -6-(3) (3) 1282209 motion and noise cannot be solved. Further, the object is an induction motor, and as such, it is difficult to apply to a synchronous motor. SUMMARY OF THE INVENTION An object of the present invention is to provide a control device for an electric motor which can suppress vibration and noise caused by periodic disturbance. [Means for Solving the Problem] One of the features of the present invention is that, in the control device of the synchronous motor, the cycle of calculating the 干扰 according to the axis error and determining the periodic interference component generated by one or both of the motor or the load is obtained. Sexual interference estimator. [Effect of the Invention] According to the present invention, it is possible to realize a control device for an electric motor capable of suppressing vibration and noise caused by periodic disturbance. [Complex Mode] Next, an embodiment of a control device for an AC motor according to the present invention will be described with reference to Figs. 1 to 15 . Further, in the following embodiments, the motor is described using a permanent magnet type synchronous motor (hereinafter, abbreviated as a PM motor), but other synchronous motors (for example, a wound type synchronous motor, a reluctance motor, etc.) are also used. It can be implemented in the same way. [Embodiment 1] (4) 1282209 The following is a block diagram showing the system configuration of Embodiment 1 of the AC motor control according to the present invention. This embodiment! The control device is given a rotation number command generator 1 for rotating the motor by the command of the upper control device 100, and the voltage of the operation motor is converted into a pulse width modulated wave signal (PWM signal). The controller 2, and the converter 4 for supplying power to the inverter 3 by the PWM signal, and the compressor 5 for controlling the object 5' and the load of the PM motor, and the detecting pair converter 3, the current supplied by the current detector 7 is formed by the current detector 7 and is reproduced by the three-phase AC I v, I w of the PM motor 5 calculated by the current detector 7 . The current reproducer 8, and the dq coordinate converter which converts the reproduced currents Iuc, Ivc, Iwc by the phase angle 0 dc (the position of the magnet flux of the motor of the assumed P Μ motor) to the components Idc, Iqc on the coordinate axis 9, and the current component, the I q * generator 1 给予 of the instruction I q *, and the I d * yield of the instruction I d * for the current component on the d-axis and according to Id*, Iq*, And the electrical angular frequency command ω 1*, the voltage command operation of the Vdc*, Vqc* 12, and Vdc* is replaced by three-phase AC voltage command Vu*, Vv*, Vw* dq 1 3, and according to the three-phase AC voltage command, the PWM of the switching converter wave width modulation signal (PWM signal) is generated. The pulse generator is equivalent to the angle of the magnet flux position β d of the P Μ motor 5 and the error at the assumed position Θ dc (axis error). The device is implemented by: the number of revolutions command current is applied The inverter 3, the PM motor converter 4, the 〇 current 1 〇, the current Iu, and the internal phase of the three-phase converter are converted into the same on the q-axis, the device 11, the voltage finger, the Vqc* conversion converter 3 Use pulse 1 4, and push controller 2 △ β (8) 1282209 Here, the composition of the period. The periodicity is uneven, the situation, the equivalent of the arm of the inverter, etc., are also replaced by air conditioners. During the reciprocating period, the load is excited for a controlled torque variable. The torque is obtained by some kind of hand pulsation. The information is obtained by the electric power. However, the delay is based on the electric motor. Consider that the voltage disturbance VD or the load torque TL includes a weekly voltage disturbance VD, for example, when the magnet of the p μ motor has magnetic permeability deviation, or the phase deviation of the winding wire becomes periodic voltage interference. influences. Or it is generated based on 6 times the frequency of the drive frequency of the interferometer due to the influence of the short-circuit prevention period (no-load time). The periodic load torque disturbance can be considered, for example, in the case of a reciprocating compressor for a freezer or φ, or a negative compressor such as a single rotary compressor, in which one of the electric motors rotates for one week. These vibrations and noises are suppressed in a systematic manner, and it is sufficient to constitute a control system in which the above-described surroundings become zero. In the conventional invention, the segment detects the rotational speed information, and controls the applied voltage to cause the rotation to correspond. In a compressor such as an air conditioner, it is difficult to directly detect the cause, so the change of the neutral point potential of the motor and the information of the 60 degree scale are detected to estimate the speed. In this way, for the electrical angular period, only 6 points are obtained. The speed information is not sufficient. In this state, an influence of 60 is generated, or there is a problem in the speed detection accuracy. Or the pulsation caused by the distortion of the induced electromotive voltage of the machine, which is a period with a short period (mainly 1 / 6 cycles), so it is difficult to: In addition, it is also possible to consider the driving control theory and construct a method of estimating the pulsating torque by the interference observation -12-(9) 1282209, but in this case, the response frequency of the observer itself becomes a problem. In the case where the frequency of the pulsating torque is high, it is necessary to increase the set response of the observer. The higher the frequency component of the pulsating torque becomes, the higher the responsiveness of the observer is required, and as a result, high-speed arithmetic processing is required. Therefore, as a method of suppressing periodic interference so far, vibration suppression in the low-speed range is generally possible, but suppression at high-speed rotation is difficult. For example, consider the case of using a general-purpose microcomputer to form an observer. In the case where the observer response time is 1 m s (1 0 0 0 a d / s - approximately 1 50 Η z ), the detectable pulsating torque is 3 Η ζ. If it is set to a 4-pole motor, it becomes 900 [r/min]. In the case of a compressor, the maximum number of revolutions is usually more than 3 000 [r/min], and if it is not below 30%, it becomes unsuitable. In the present invention, focusing on the block diagram of Fig. 4, a method of estimating the torque ripple component ΔTm from the axis error Δ 0 is proposed. The axis error △ 0 is instantaneous and can be calculated as an instantaneous delay, which can be affected by the delay of the operation, and the high precision is pushed up. In addition, it is also characterized by a high frequency component (e.g., a vibration component of 6 times) that is sealed at the driving frequency. This result is made possible by the calculation of the large-scale high-speed field compared with the conventional periodic interference control method. In the case where such periodic disturbance occurs, the difference between the motor torque Tm and the load torque T L becomes a periodic torque fluctuation, which causes vibration and noise. In order to suppress this vibration and noise, for example, it is necessary to increase the size of the device by enclosing the entire device with a sound absorbing material, and the increase in cost -13-(10) 1282209 is a problem to be solved. In order to controlly suppress vibration and noise, it is sufficient to constitute a control system that causes the above-described cyclic torque fluctuation to become zero. In the conventional invention, the rotational speed is detected by some means, and the applied voltage is controlled so that the rotational pulsation becomes zero. However, in a compressor such as an air conditioner, the motor is incorporated in the compressor, so it is difficult to obtain speed information simply, and it is possible to obtain only the information equivalent to the electrical angle of 60 degrees. Therefore, it is difficult to achieve high precision. In the present invention, focusing on the block diagram of Fig. 4, a method of estimating the torque ripple component ΔT m by the axis error Δ (9) is proposed. The axis error Δ 0 is instantaneously operable, and can be delayed by the operation. In addition, for the driving frequency, for high frequency components (for example, 6 times the vibration component), it can also be detected as its characteristic. Figure 5 shows the load torque TL at the angular frequency. ω d and the torque ripple component (Δ Tm ), the rotational speed variation (Δ ω r ), and the axial error (Δ 0 ) in the case where the component of the vibration is included in the sinusoidal waveform. The average of Tm and TL when considering the steady state The 値Tm is the same, and ΔTm is only the vibration component (Fig. 5(b)). The vibration component ΔωΓ included in the rotational speed is integrated into this ΔTm, and becomes a waveform with a phase delay of 90 degrees compared with ΔTm. The size itself varies according to the inertia J, but the phase can be considered to be almost delayed by 90 degrees. The axis error Δ Θ becomes the integral Δ ω r 'the sign inversion (as defined in Fig. 2, Reverse sign) The phase becomes 90 degrees forward (90 degrees delay with integration, the sign is reversed, and becomes 90 degrees forward). That is, the variation component of ΔT m is changed to the same in -14-(11) 1282209 △ 0 The vibration waveform of the phase is observed. When the relationship is derived from the block diagram, it becomes as follows. Figure 6(a) shows a block diagram of Δ丁〇1 to ^0. By inversely transforming the block diagram, The transfer function from △ 0 to △ can be obtained as shown in Fig. (c). When ΔTm is obtained from the figure 6(c), it can be directly estimated from Δ <9dc( △ (calculation of 9) The pulsating component of the torque. However, the second-order differential Δ 0dc is practically impossible. △ecu is originally calculated as 推, and most of the package φ contains the noise of detecting 値. The use of differential will increase the estimation error, and also from the calculation cycle. Therefore, focusing on the point where the interference component is a periodic function, substituting s = j ω d into the sixth (c) diagram. As a result, as shown in Fig. 6(d), it becomes a deductible △ Θ A constant multiple will become Δ Tm. The result is the waveform of (b) and (d) of Fig. 5. The relationship is the same. The Δ Tm estimator 21 (periodic interference estimator) shown in Fig. 7 is embodied in the structure of Fig. 6 (d). The ΔΤγπ estimator 21 (periodic dry rudder estimator) is △ <9 dc is formed by 2 J1 Ρ times the proportional gain 2 1 1, and two multipliers 212, and the calculation of Fig. 6(d) is carried out. According to Fig. 7, it can be estimated from Δ 0 dc to be included in Δ Tm The periodic disturbance component of the angular velocity ω d Next, the torque controller 22 (Fig. 8) for suppressing this Δ Tm will be described. The necessary conditions for the torque controller are: (1) For the periodic disturbance component, the follow-up is high, -15- (16) 1282209 The pulsation component Δ Tmc is converted by (coordinate 1) The DC amount can be removed by integrating the controller 2 2 5 . That is to say, if the torque controller is externally viewed, it is equivalent to the compensation element in which the gain becomes infinite in the angular frequency ^3. That is, it has the same frequency characteristics as the torque controller 22 of the first embodiment. In the case of the torque controller 22C, the adjustment position becomes the time constant TATr of the primary delay filter and the gain Ki ATR of the integral controller 22 5 as compared with the torque controller of Fig. 8 or Fig. 9. However, TATR can choose a sufficiently large time constant for ω d, so the adjustment method is not particularly difficult. In addition, the A of the KiATR directly becomes the response time for determining the pulsation component suppression, and the control response time becomes linear after the KiATR. As a result, it is possible to obtain an effect that the gain setting becomes easy. [Embodiment 4] Next, a fourth embodiment according to the present invention will be described using Fig. 1 . In Embodiment 3, a torque controller for which the gain becomes infinitely large with respect to the vibration frequency ω d is provided. This operation is equivalent to the torque controller (Fig. 8) of the first embodiment. Therefore, the same problem as that described in the second embodiment is produced. In other words, the ω d component included in the torque ripple is removed, and the distortion of the drive current of the PM motor is increased, and the efficiency of the pM motor is likely to be deteriorated or the overcurrent trip due to the peak current is caused. Therefore, the same as in the second embodiment proposes a method of changing the gain of the angular frequency ω d from infinity to finite. Fig. 1 shows the construction of the torque controller 2 2 D of the fourth embodiment. -20- (17) 1282209 By using the torque controller 2 2 D instead of the torque controller 22 of Fig. 1, the embodiment 4 is realized. The difference between the torque controller 22D of Fig. 1 and the torque controller 2 2C of Fig. 10 is that the integral controller 22 5 is changed to the point of the incomplete integral controller 22 5 D. According to the time constant Ti and the gain KiATR in the incomplete integrator 22 5 D, the peak 値 is suppressed. As a result, it becomes possible to adjust the interference suppression effect of the ω d component, and it is possible to drive at the optimum point of distortion of noise, vibration, and P Μ motor phase current. φ [Embodiment 5] Next, a fifth embodiment according to the present invention will be described using Fig. 2 . In the first to fourth embodiments, a method of estimating and suppressing the periodic torque ripple component based on the estimated 値' of the axial error Δ 0 is described. The main pulsating component appears in I q c or the axis error estimation 但是, but it also has an effect on 〗 〖d ^. ^ Although the d-axis current does not contribute to the torque, the rotation axis produces a deviation according to the torque pulse, and a ripple-based current is also generated in the d-axis direction. Embodiment 5 is an embodiment in which this is used to further reduce torque ripple. In Fig. 12, the controller 2 E is almost the same as the controller 2 of the first embodiment. A new axis control device IdACR ( 22C ) for d-axis (dc axis) current control is added. For example, 22C is introduced in the same manner as the torque controller 22C shown in Fig. 1 (the gain KiATR needs to be adjusted), and / I d c is input instead of Δ Tm c , and the output is added to I d *. In the voltage command operator 丨9: 'I d * * is treated as a new command 値, and the voltage command is operated. -21 - (20) 1282209 In the embodiment, the air conditioner is taken as an example. However, the same effect can be obtained in other electric appliances such as an air conditioner or a freezer. As described above, according to the present invention, a high-performance motor drive that suppresses the periodic torque disturbance generated by the load device or the motor itself can be realized without using a sensor for detecting the rotational speed or the rotational axis position of the synchronous motor. Further, even in the case of a sensor having a rotational speed or a rotational axis position of the synchronous motor, the same can be realized. BRIEF DESCRIPTION OF THE DRAWINGS Fig. 1 is a block diagram showing the system configuration of a first embodiment of a synchronous motor control device according to the present invention. Fig. 2 is a view showing a definition vector of the axis error ? 0 of the embodiment of the synchronous motor control apparatus according to the present invention. Fig. 3 is a block diagram showing the internal configuration of the axis error estimator of the first embodiment of the synchronous motor control apparatus according to the present invention. Fig. 4 is a block diagram showing the principle of the application of voltage to the axis error to the motor of the embodiment 1 of the AC motor control device according to the present invention. Fig. 5 is a waveform diagram showing the principle of the periodic disturbance torque of the first embodiment of the AC motor control device according to the present invention, and the principle of the fluctuation of the rotation and the axis error caused thereby. Fig. 6 is a block diagram showing the principle of calculation of the ripple torque component of the first embodiment of the AC motor control device according to the present invention. - 24 - (21) 1282209 Table 7 is a block diagram showing the internal configuration of the ΔT m estimator of the first embodiment of the synchronous motor control device according to the present invention. Fig. 8 is a block diagram showing the internal configuration of a torque controller of the first embodiment of the synchronous motor control apparatus according to the present invention. Fig. 9 is a block diagram showing the internal configuration of a torque controller of the second embodiment of the synchronous motor control device according to the present invention. Fig. 1 is a block diagram showing the internal configuration of a torque controller of Embodiment 3 of a synchronous motor control device according to the present invention. Fig. 1 is a block diagram showing the internal configuration of a torque controller of the fourth embodiment of the synchronous motor control device according to the present invention. Fig. 12 is a block diagram showing the internal configuration of the controller of the fifth embodiment of the synchronous motor control device according to the present invention. Fig. 13 is a view showing an appearance configuration of the sixth embodiment of the synchronous motor control apparatus according to the present invention. Fig. 14 is a view showing the appearance configuration of a seventh embodiment in which the synchronous motor control device according to the present invention is applied to an air conditioner. Fig. 15 is a view showing an example of the change of the rotation speed and the change of the current waveform in the case where the compressor of the air conditioner according to the present invention is started. [Main component symbol description] 1 : Rotation number command generator, 2: controller, 3: inverter, 4: converter, 5: P Μ motor, 6: compressor, 7: current detector, 8: current Reproducer, 9: dq coordinate converter,: generator, -25- (22) 1282209 1 1 : Id* generator, 12: voltage command operator, 13: dq inverse converter, 1 4 : P WM pulse Generator, 1 5 : Δ 0 estimator, 1 6 : adder and subtracter, 1 7 : zero command generator, 1 8 : proportional compensator, 1 9 : conversion gain, 20: integrator, 21 : ΔΤιη estimator, 22: Torque controller, 41: AC power, 42: diode bridge, 43: smoothing capacitor.

-26--26-

Claims (1)

1282209 G ':=!:-/ y:/ : -r^ ..…9a_i'-s …一、… 十、申請專利範圍 第93 1 3 8 5 9 1號專利申請案 中文申請專利範圍修正本 民國9 6年1月5日修正 1 · 一種同步電動機之控制裝置,是針對具有藉由換 流器之輸出電壓以控制伴隨負載之同步電動機之控制器之 同步電動機之控制裝置,其特徵爲: 在前述控制器中’具有依據軸誤差推算値,以求得前 述電動機或負載之其中一方,或者雙方所產生之週期性干 擾成分之週期性干擾推算器。 2.如申請專利範圍第1項所記載之同步電動機之控 制裝置’其中’則述軸誤差推算値,係依據流經前述同步 電動機之交流電流,或者電源所供給之電流的至少其中一 方之檢測値所運算。 3 ·如申請專利範圍第1項所記載之同步電動機之控 制裝置,其中,具有藉由在週期性干擾之變動頻率或變動 頻率附近具有峰値之頻率特性補償器,以消除前述週期性 干擾之轉矩控制器。 4. 如申請專利範圍第1項所記載之同步電動機之控 制裝置,其中,前述軸誤差推算値係相當於前述同步電動 機之磁極軸的相位角,與前述同步電動機之磁極軸的推算 相位角之誤差的量。 5. 如申請專利範圍第1項所記載之同步電動機之控 制裝置,其中,前述負載係壓縮機。 -^ - Ί 年月曰修(更)正替換朽 ^ 96 1. : 6 ·如申請專利範圍第1項所記載之同步電動機之控 制裝置,其中,前述週期性千擾推算器係依據前述軸誤差 推算値,及週期性干擾之變動頻率,及前述同步電動機以 及前述負載裝置之常數,以運算前述週期性干擾成分。 7.如申請專利範圍第3項所記載之同步電動機之控 制裝置,其中,依據前述轉矩控制器之輸出,對前述控制 器之輸出電壓施加補正。1282209 G ':=!:-/ y:/ : -r^ .....9a_i'-s ... one,... X. Patent Application No. 93 1 3 8 5 9 No. 1 Patent Application Revision of Chinese Patent Application Range Amendment 1 January 5, 1996. 1. A control device for a synchronous motor is a control device for a synchronous motor having a controller for controlling a synchronous motor with a load by an output voltage of an inverter, and is characterized in that: In the controller described above, 'there is a periodic interference estimator that estimates the 依据 according to the axis error to obtain one of the aforementioned motor or load, or a periodic interference component generated by both. 2. The control device for a synchronous motor as described in the first paragraph of the patent application, wherein the axis error estimation is based on at least one of an alternating current flowing through the synchronous motor or a current supplied from the power source.値 calculated. 3. The control device for a synchronous motor according to claim 1, wherein the frequency characteristic compensator having a peak 附近 near a varying frequency or a varying frequency of the periodic disturbance is used to eliminate the periodic interference. Torque controller. 4. The control device for a synchronous motor according to claim 1, wherein the axis error estimation system corresponds to a phase angle of a magnetic pole axis of the synchronous motor and an estimated phase angle of a magnetic pole axis of the synchronous motor. The amount of error. 5. The control device for a synchronous motor according to claim 1, wherein the load is a compressor. -^ - Ί 曰 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 96 The error estimation 値, and the frequency of the periodic disturbance, and the constants of the synchronous motor and the load device are used to calculate the periodic interference component. 7. The control device for a synchronous motor according to claim 3, wherein the output voltage of the controller is corrected in accordance with an output of the torque controller. 8 ·如申請專利範圍第3項所記載之同步電動機之控 制裝置,其中,具備有,變更前述峰値,令週期性干擾之 抑制效果可以變更之手段。8. The control device for a synchronous motor according to the third aspect of the invention, wherein the peak enthalpy is changed, and the effect of suppressing the periodic disturbance can be changed. 9. 如申請專利範圍第3項所記載之同步電動機之控 制裝置,其中,前述轉矩控制器係以前述週期性干擾爲輸 入,乘上以前述週期性干擾之頻率改變之SIN函數以及 COS函數,求得個別之平均値,導出前述週期性干擾之 SIN成分、COS成分,對於前述控制器之輸出電壓施加令 前述SIN成分以及前述COS成分成爲零之藉由積分控制 或者不完全積分控制之補正。 10. 如申請專利範圍第1項所記載之同步電動機之控 制裝置,其中,具備有,對於前述同步電動機之磁極軸相 位,運算與其同步之電流成分的激磁電流成分之手段; 具備有,去除包含在前述激磁電流成分之脈動份的手 段。 11. 一種冷凍庫,具備有藉由換流器之輸出電壓以控 制同步電動機的控制器之同步電動機的控制裝置;其特徵 -2- 1282202_ 年月曰修(更)正替換頁I 9fi Ϊ 5, 爲: 在前述控制器中,具有依據軸誤差推算値,以求得前 述電動機或負載之其中一方,或者雙方所產生之週期性干 擾成分之週期性干擾推算器; 藉由前述同步電動機之控制裝置,以驅動壓縮機。9. The control device for a synchronous motor according to claim 3, wherein the torque controller receives the periodic disturbance as an input, multiplies the SIN function and the COS function which are changed by the frequency of the periodic disturbance. And obtaining an average 値 of the individual, and deriving the SIN component and the COS component of the periodic interference, and applying a correction to the output voltage of the controller to make the SIN component and the COS component zero by integral control or incomplete integral control . 10. The control device for a synchronous motor according to the first aspect of the invention, further comprising: means for calculating a field component of a current component synchronized with the phase of the magnetic pole of the synchronous motor; A means of pulsing a part of the aforementioned excitation current component. 11. A freezer having a control device for a synchronous motor having a controller for controlling a synchronous motor by an output voltage of the inverter; characterized in that it is replaced by a page I 9fi Ϊ 5, In the foregoing controller, there is a periodic interference estimator that estimates 値 according to the axis error to obtain one of the aforementioned motor or load, or a periodic interference component generated by both; and the control device of the synchronous motor To drive the compressor. 12· —種空調機,具備有藉由換流器之輸出電壓以控 制同步電動機的控制器之同步電動機的控制裝置;其特徵 爲· 在前述控制器中,具有依據軸誤差推算値,以求得前 述電動機或負載之其中一方,或者雙方所產生之週期性干 擾成分之週期性干擾推算器; 藉由前述同步電動機之控制裝置,以驅動壓縮機。 13. —種模組,是針對具備有,12. An air conditioner comprising: a control device for a synchronous motor having a controller for controlling a synchronous motor by an output voltage of an inverter; wherein the controller has a factor of 依据 according to an axis error A periodic interference estimator of one of the aforementioned electric motor or load, or a periodic interference component generated by both sides; and a control device of the synchronous motor to drive the compressor. 13. The type of module is for possession, 對連接於負載之同步電動機施加電壓之換流器,及 對換流器供給電流之轉換器,及 控制前述電壓之控制器之模組,其特徵爲: 在前述控制器中,具有依據軸誤差推算値,以求得前 述電動機或負載之其中一方,或者雙方所產生之週期性干 擾成分之週期性干擾推算器。 -3 -An inverter for applying a voltage to a synchronous motor connected to a load, a converter for supplying current to the inverter, and a module for controlling the voltage of the voltage, wherein: the controller has a basis error The 値 is calculated to obtain one of the aforementioned motor or load, or a periodic interference estimator of the periodic interference component generated by both parties. -3 -
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