JP2013150498A - Controller and control method of synchronous motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide practical controller and control method of a synchronous motor capable of reducing torque ripple due to manufacturing variations of a synchronous motor and a load connected therewith or the environmental variations.SOLUTION: The controller of a synchronous motor which is supplied with a three-phase AC power from a power inverter circuit including an inverter includes dq-axis current calculation means for determining the dq-axis current of a synchronous motor, a q-axis current command generator for obtaining a q-axis current command value from the output of the dq-axis current calculation means, a voltage command value generator for giving a q-axis voltage command value by using the q-axis current command value, a conversion circuit obtaining a three-phase AC voltage command value for driving the inverter from the output of the voltage command value generator, and a torque ripple reduction controller obtaining a compensation signal for reducing torque ripple, by estimating the induction voltage harmonic and torque ripple of a synchronous motor. The q-axis voltage command value is corrected by the compensation signal.

Description

  The present invention relates to a control device and a control method for a synchronous motor, and more particularly to a control device and a control method for a synchronous motor that can reduce vibration caused by torque ripple.

  Synchronous motors are widely used as motors that can contribute to a wide range of variable speed control and power consumption. For example, it is used as a fan motor for recent air conditioners and water heaters. In these applications, low noise and low vibration are required, and in order to achieve this, it is designed to have a sinusoidal induced voltage waveform.

  However, the fifth-order and seventh-order harmonic components are superimposed due to manufacturing variations of the synchronous motor. Further, the fan connected to the synchronous motor also causes torque vibration because the load torque is not constant due to manufacturing variations of the fan.

  Many studies have been made on these so-called torque ripple reduction measures. For example, Non-Patent Documents 1 and 2 introduce a simple vector control method that takes into account a reduction measure of torque ripple when the synchronous motor is applied to a relatively small capacity device such as a home appliance.

  In Non-Patent Document 3, the fifth-order and seventh-order harmonic components of the induced voltage constant of the motor are measured, and the current control is performed to cancel the torque ripple. When variations in the operating environment such as the above occur, there is a problem that the torque ripple is amplified and leads to noise and vibration.

“Simple vector control of a position sensorless permanent magnet synchronous motor for home appliances”, D. Vol. 124, no11, pages 1133 to 1140, 2004 “Position sensorless simple vector control of permanent magnet synchronous motor using adaptive adjustment of induced voltage constant”, D. Vol. 129, no 4, pages 406-414, 2009 "Study on torque ripple reduction method for IPMSM with distorted induced voltage", Master's thesis, Yokohama National University, 2008

  As described above, various analyzes and proposals have been made for reducing torque ripple when using a synchronous motor.

  However, not only a synchronous motor but also few studies have been made in consideration of the behavior of a load driven by the synchronous motor (for example, a fan such as an air conditioner or a water heater) and manufacturing variations. In addition, when the synchronous motor is applied to a relatively small-capacity device such as for home appliances, it is necessary to be able to cope with a position sensorless in order to realize a simple device configuration.

  From the above, the present invention provides a practical synchronous motor control device and control method capable of reducing torque ripple caused by manufacturing variations and environmental variations of the load connected to the synchronous motor and the synchronous motor. Objective.

  In the present invention, in a control apparatus for a synchronous motor that is supplied with a three-phase alternating current from a power conversion circuit including an inverter, a dq-axis current calculation means for obtaining a dq-axis current of the synchronous motor, and an output from the dq-axis current calculation means Q-axis current command generator for obtaining the axis current command value, voltage command value generator for giving q-axis voltage command value using the q-axis current command value, and driving the inverter from the output of the voltage command value generator A conversion circuit that obtains a three-phase AC voltage command value, and a torque ripple reduction controller that estimates the induced voltage harmonics and torque ripple of the synchronous motor and obtains a compensation signal for reducing the torque ripple. It is characterized by correcting the value.

  The compensation signal is added to the q-axis voltage command value obtained from the voltage command value generator.

  The compensation signal is reduced from the q-axis current command value obtained from the q-axis current command generator.

  The compensation signal is added to the q-axis voltage command value obtained from the voltage command value creator, and the compensation signal is subtracted from the q-axis current command value obtained from the q-axis current command creator.

  The dq-axis current calculating means calculates the dq-axis current using the alternating current of the synchronous motor.

  The dq-axis current calculating means calculates the dq-axis current using a direct current flowing into the inverter of the power conversion circuit.

  In the present invention, in a control method for a synchronous motor that is supplied with three-phase alternating current from a power conversion circuit including an inverter, a q-axis voltage command value is obtained using a q-axis current command value of the synchronous motor, and a q-axis voltage command value is obtained. A three-phase AC voltage command value for driving the inverter is obtained from the motor, and a compensation signal for reducing the torque ripple is calculated from the induced voltage harmonics and torque ripple of the synchronous motor, thereby correcting the q-axis voltage command value. It is characterized by.

  According to the present invention, since torque ripple can be reduced, it is possible to obtain a practical synchronous motor control device and control method that are not affected by manufacturing variations of the synchronous motor and loads connected to the synchronous motor and environmental variations. it can.

The figure which shows an example of a structure of the position sensorless simple vector control apparatus which concerns on this invention. The figure which shows an example of the position sensorless simple vector control apparatus of a permanent-magnet synchronous motor. The figure which showed the typical circuit structure of the power converter device 6. FIG. The figure which displayed the relationship of (alpha) and (beta) coordinate system and d and q coordinate system stationary. The figure which shows the specific numerical value of the various coefficients set for effect verification. The figure which showed the actual value and estimated value of the torque ripple measured on the conditions of FIG. The figure which showed the torque ripple observed at the time of control by FIG. The figure which showed the observed torque ripple at the time of control by FIG. The figure which showed the magnitude | size according to the generation order of the torque ripple at the time of control by FIG. The figure which showed the magnitude | size according to the generation order of the torque ripple at the time of control by FIG. The figure which contrasted and showed each part signal waveform before and after implementation of this invention. The figure which shows a flowchart when implement | achieving an electric motor control apparatus in software. The figure which shows the synchronous motor of the type which detects a direct current with a direct current detector. The figure which shows the alternative of this invention using a direct current. The figure which shows the alternative of this invention using a direct current.

  Examples of the present invention will be described in detail below.

  In short, the present invention is based on the discovery that the waveform of the induced voltage harmonic and the waveform of the torque ripple are very close. From this, the induced voltage harmonic and the torque ripple are estimated from theoretical equations, respectively, and the torque ripple is reduced by reflecting the ratio of the estimated values in the existing control device.

  For this reason, in the following description, the behavior in the conventional apparatus will be described first, and then the behavior in the apparatus of the present invention will be described. Further, several embodiments from the viewpoint of position sensorless will be described later.

  First, the behavior in the conventional apparatus will be described. First, the synchronous motor will be described. The synchronous motor in this case is a permanent magnet synchronous motor (PMSM).

  Equation (1) shows the dq axis equation of the permanent magnet synchronous motor PMSM in consideration of the fifth and seventh order magnetic flux harmonics that are considered to have a great influence on the torque ripple.

  However, in the first term on the right side of equation (1), vd is a d-axis voltage, vq is a q-axis voltage, R is a winding resistance, Ld is a d-axis inductance, Lq is a q-axis inductance, id is a d-axis current, iq Represents a q-axis current, and ω represents an electrical angular velocity.

In the second term of (1) the right side, K _E is the induced voltage constant. Figures 1, 5, and 7, which were subjected to K _E represents 1, fifth, and seventh order of degree. "C" also were subjected to K _E and "s" means that the sin component and cos component. Therefore, “K _E1s ” represents an induced voltage primary sine component. θ is the phase difference between voltage and current. Although not specifically described below, symbols and numerical values attached as subscripts to various symbols shall follow the above notation.

  Further, regarding the torque of the permanent magnet synchronous motor PMSM, the formula (2) is established. However, Te is main torque and Pn is the number of pole pairs.

Note (2) below, the current id, symbols marked with dash superscript iq ^{i 'd, i'} q is expressed by (3) (4). Iα and iβ in the equation mean a biaxial current obtained by converting a three-phase alternating current into a two-phase composed of orthogonal α-axis and β-axis components.

  Next, a conventional configuration of the position sensorless simple vector control device of the permanent magnet synchronous motor PMSM will be described. FIG. 2 shows an example of a position sensorless simple vector control device of a typical permanent magnet synchronous motor PMSM.

  In FIG. 2, 8 is a permanent magnet synchronous motor PMSM, and 2 is a motor control device. Reference numeral 6 denotes a power conversion circuit composed of, for example, a converter and an inverter. The power conversion circuit 6 and the synchronous motor 8 constitute a so-called power main circuit. The three-phase voltage values Vu, Vv, and Vw given to the synchronous motor 8 by the output of the motor control unit 4 are controlled to voltages whose magnitude and frequency are variable.

FIG. 3 shows a typical circuit configuration of the power converter 6. The power conversion device 6 includes an inverter main circuit 3, a DC power source 1 using a converter, and a gate drive circuit 11. Among them, the inverter main circuit 3 constitutes an arm by serially connecting two sets of circuits composed of conversion elements T connected in parallel with the antiparallel diode D. And three sets of arms are connected in parallel between the positive and negative terminals of the DC power source 1 by the converter. The three phases of the synchronous motor 8 are connected to the connection point h of the three sets of arms. The gate drive circuit 11 has six conversion circuits T of the inverter main circuit 3 according to the command values V ^* u, V ^* v, V ^* w of the three-phase voltage values Vu, Vv, Vw applied to the synchronous motor 8. Drive the ignition phase.

  As shown in FIGS. 2 and 3, the synchronous motor 8 used here is of a type that does not include a position sensor, and for controlling the synchronous motor 8, out of the three phases from the current detectors 7 a and 7 b. The appropriate two-phase currents iu and iv are input to the motor control unit 4.

Returning to FIG. 2, the motor control unit 4 controls the synchronous motor 8 in addition to the three-phase AC currents iu and iv, the inverter angular velocity command value ω ^* 1, and the d-axis current command generator 49 output d-axis. Input the current command value i ^* d. Finally, the three-phase AC voltage command values V ^* u, V ^* v, and V ^* w of the power main circuit are output to the synchronous motor 8.

  In the motor control unit 4, the three-phase / dq conversion circuit 41 obtains the detected values idc and iqc of the d-axis current and the q-axis current from the three-phase alternating currents iu and iv. In this conversion, conversion into a biaxial component (α, β axis) is performed in the conversion process, and the conversion is performed using the dc axis angle θdc viewed from the α axis at this time as a reference phase. This concept will be described later.

Next, the q-axis current command generator 44 calculates the q-axis current command value i ^* q using the detected value iqc of the q-axis current. Here, the q-axis current command generator 44 is composed of, for example, a first-order lag circuit, and is derived by a first-order lag calculation of equation (5). T1q is a delay time constant.

The voltage command value generator 48 receives the input inverter angular velocity command value ω ^* 1, the d-axis current command value i ^* d which is the output of the d-axis current command generator 49, and the q-axis current command value i ^* obtained by calculation ^. dq-axis voltage command values V ^* dc and V ^* qc are calculated from q using equation (6).

  In addition, the symbol showing the information used for calculation of (6) Formula has already been demonstrated. The only difference is that these symbols are marked with an asterisk “*” as a superscript, but this is a set value.

In addition, the d-axis current set value i ^* d is used in the calculation of the equation (6). Since the target is a surface-permanent permanent magnet synchronous motor (SPMSM), ( 7) Treat as zero as shown in the equation.

The dq axis voltage command values V ^* dc and V ^* qc obtained by the voltage command value generator 49 are the three-phase AC voltage command values V ^* u, V ^* v, Converted to V ^* w. In this way, the three-phase voltage values Vu, Vv, Vw given to the synchronous motor 8 by the output of the motor control unit 4 are controlled to voltages whose magnitude and frequency are variable. In the conversion process in the dq / three-phase conversion circuit 50, the conversion is performed using the dc axis angle θdc viewed from the α axis as a reference phase.

The dc axis angle θdc viewed from the α axis is obtained by the following process. First, in the axis error calculator 45, the axis error Δθ is calculated by the equation (8). For this calculation, the detected values idc and iqc of the d-axis current and the q-axis current obtained by the three-phase / dq conversion circuit 41 are used. The dq axis voltage command values V ^* dc and V ^* qc obtained by the voltage command value generator 49 are used. Here, the axis error Δθ is derived from the equation (8).

  Here, before entering the following description, the axis error Δθ will be described with reference to FIG. In FIG. 4, the relationship between the α and β coordinate systems, which are orthogonal coordinates obtained by two-phase conversion of the three-phase current flowing in the stator winding of the power main circuit, and the d and q coordinate systems is displayed statically. In this figure, d, q, α, and β indicate the d-axis, q-axis, α-axis, and β-axis. In addition, dc and qc are an estimated d-axis and an estimated q-axis, respectively, and represent a state in which the d-axis that should originally be at the position d is shifted to the position dc. The angle formed by the d-axis (d) and the estimated d-axis (dc) is the axial error Δθ, and this axial error Δθ should be zero. The angle formed by the α axis (α) and the estimated d axis (dc) is the dc axis angle θdc viewed from the α axis.

A PLL controller 46 (phase lock loop) is a transmitter, and an angular frequency adjustment value Δω1 is given as an output thereof. The adjustment value Δω1 of the angular frequency is added to the input inverter angular velocity command value (frequency command value) ω ^* 1 by the addition circuit ad1 to obtain the inverter frequency ω1.

  The dc axis angle θdc viewed from the α axis used in the dq / three-phase conversion circuit 50 or the three-phase / dq conversion circuit 41 is obtained by integrating the inverter frequency ω1 in the integrator 47.

  These circuits constituted by the axis error calculator 45, the PLL controller 46, the adder ad1, and the integrator 47 are obtained by adjusting the dc axis angle θdc viewed from the α axis so that the axis error Δθ is zero. is there. This circuit controls to increase the speed of the virtual axis (inverter frequency ω1) when the axis error Δθ is positive, and to decrease the speed of the virtual axis (inverter frequency ω1) when the axis error Δθ is negative. ing.

  In order to realize this, the PLL controller 46 calculates an angular frequency adjustment value Δω1 proportional to the axis error Δθ by the equation (9). Kps is a proportional gain.

  The adder ad1 executes equation (10).

  The integrator 47 executes equation (11) to obtain the dc axis phase θdc necessary for vector control.

  The proportional gain Kps in equation (9) is determined by equation (12).

The conventional permanent magnet synchronous motor PMSM and the motor control device 2 are configured as described above. As a countermeasure against torque ripple reduction in this conventional apparatus, the main torque Te is controlled to be constant by controlling the q-axis current iq by paying attention to the equation (2) which is the torque equation of the permanent magnet synchronous motor PMSM described above. Specifically, torque ripple is suppressed by correcting the q-axis current setting i ^* q to i ^{* ′} q as shown in the equation (13). The idea of this equation is that the torque ripple generated by the magnetic flux harmonic component is controlled by current control to cancel the torque ripple caused by the magnetic flux.

Incidentally, (13) ⁱ in the formula 'd5, ^i' q5 is ^{(14), i 'd7,} i' q7 is represented in (15).

  On the other hand, the torque ripple reduction in the present invention is based on the discovery that the waveform of the induced voltage harmonic and the waveform of the torque ripple are very close. For this reason, before describing the embodiment of the present invention, it will be described that the waveform of the induced voltage harmonic and the waveform of the torque ripple are very approximate.

In the dq axis equation (1) of the permanent magnet synchronous motor PMSM described above, d
The q-axis induced voltage harmonic components ed6 and eq6 are obtained by the equation (16).

  Further, in the torque equation (2) of the permanent magnet synchronous motor PMSM, the torque ripple component T6 is obtained by the equation (17).

  In the following description, the equations (18), (19), and (20) are established between the dq axis currents id and iq and the biaxial currents iα and iβ.

  Here, since the q-axis current iq is usually very large with respect to the d-axis current id, the equation (18) can be expressed as the equation (21). Further, by organizing the equations (19) and (20) using this result, it can be expressed by the equation (22).

  By substituting equation (22) into equation (17), the torque ripple equation can be expressed as equation (23).

  When the equation (23) is compared with the term of the induced voltage harmonic component eq6 of the q axis in the equation (16), the shape is very close. If the seventh-order cosine component is ignored, the phases are completely matched.

  From this, it can be understood that the waveform of the induced voltage harmonic and the waveform of the torque ripple are very close as described above. Therefore, by using this q-axis induced voltage harmonic component eq6, a torque ripple component can be approximately obtained by the following equation. Further, torque ripple can be reduced by controlling the direction to cancel the torque ripple component.

  Torque ripple T6 is expressed by equation (24) using q-axis induced voltage harmonic component eq6.

  The q-axis induced voltage harmonic component eq6 can be obtained by equation (25).

  Further, the current correction value iq6T can be obtained by equation (25).

  Hereinafter, the configuration of the position sensorless simple vector control apparatus according to the present invention reflecting the above knowledge will be described with reference to FIG. However, this example shows an example in which an alternating current is detected.

The configuration of FIG. 1 basically follows the conventional configuration of FIG. 2, and is different in that a circuit portion indicated by a thick line is added. Among these, the torque ripple reduction controller 42 calculates the current correction value iq6T of the equation (25). The current correction value iq6T is obtained from the approximation of the q-axis induced voltage harmonic component eq6 and the torque ripple, and finally corrects the q-axis voltage command value V ^* qc.

Most of the data used here for calculating the current correction value iq6T is an existing value given in advance as a set value. The data actually used for every moment is the d-axis current and q-axis current detection values idc and iqc obtained in the three-phase / dq conversion circuit 41, and the q-axis voltage command from the voltage command value generator 48. The value V ^* qc and the inverter angular velocity command value ω ^* 1.

In the present invention, the first method of correcting the current correction value iq6T obtained from the approximation of the q-axis induced voltage harmonic component eq6 and the torque ripple to the q-axis voltage command value V ^* qc from the upstream side. And there is a second method of correcting on the downstream side. Furthermore, as shown in FIG. 1, the first method and the second method are used in combination.

Specifically, the first method upstream is to add the current correction value iq6T to the q-axis current command value i ^* q given to the voltage command value generator 48. The subtracting circuit sb subtracts iq6T from i ^* q to obtain i ^{* ′} q, which is used for the calculation processing of the equation (6) in the voltage command value generator 48.

However, since the simple sensorless voltage command calculation uses an approximation that ignores the current differential term, the response sensitivity is low only by correcting the q-axis current command value i ^* q from the upstream. Therefore, the second method of correcting on the downstream side is effective.

In the second method, the torque ripple reduction voltage compensator 43 derives the voltage correction value ΔV ^* qc from the current correction value iq6T. The voltage correction value ΔV ^* qc is added to the q-axis voltage command value V ^* qc in the adder circuit ad2 to be used as V ^{* ′} qc for conversion in the dq / three-phase conversion circuit 50. Since the torque ripple reduction voltage compensator 43 has a proportional differential function, the response is improved.

  When the first method and the second method are used in combination, both transient response by the second method and long-term stability by the first method can be achieved.

  The voltage command calculation is performed by equation (27).

  Further, the current correction value is obtained by the equation (28).

  Next, the results of concrete verification of the effects of the present invention will be described. First, FIG. 5 is a diagram showing specific numerical values of various set coefficients. FIG. 6 is a diagram showing the actual value Ta of the torque ripple measured under these conditions and the estimated value Tb. Looking at this waveform, it can be seen that the estimated value is able to estimate the amplitude and phase almost exactly.

  FIG. 7 shows the observed torque ripple during the control according to FIG. 2, and FIG. 8 shows the observed torque ripple during the control according to FIG. FIG. 7 shows the result of reflecting the conventional torque ripple reduction control. However, it can be seen that a large torque ripple still occurs as compared with the result of FIG.

  FIG. 9 and FIG. 10 compare the magnitudes of the torque ripples for each generation order. Although both are 6th harmonics (rated frequency: 56 Hz), it can be seen that the present invention of FIG. 10 reduces the amplitude to 50% compared to the conventional case of FIG.

  FIG. 11 is a diagram showing the signal waveforms of the respective parts before and after implementation of the present invention. The left side shows the signal waveform after the present invention is implemented, and the right side is before the implementation. Note that the portion on the left side where the corresponding signal waveform does not exist is left blank.

First, the q-axis current command values i ^* q from the q-axis current command generator 44 are the same and are substantially constant values. After implementation of the present invention, the q-axis current command value i ^{* ′} q corrected as an upstream measure is used, but this value includes a signal simulating torque ripple, and thus includes a periodic fluctuation component. As a result, after execution, the q-axis voltage command value V ^* qc obtained by the voltage command value generator 49 itself includes a periodic fluctuation component.

Next, the q-axis voltage command value V ^* qc obtained by the voltage command value generator 49 will be described. This is a constant value before implementation. After implementation, the q-axis current command value i ^{* ′} q is used as a downstream countermeasure. The q-axis current command value i ^{* ′} q thus corrected includes a further emphasized periodic fluctuation component.

  As a result, the fluctuation of the torque Te is greatly suppressed.

  FIG. 12 is a diagram showing a processing flowchart when the motor control device of FIG. 1 is realized by software in a digital circuit. Here, first, in processing step S101, it is confirmed that the inverter starting condition is satisfied. The processing after processing step S102 is entered when the activation condition is satisfied.

  In process step S102, a speed deviation Δω is calculated. In process step S103, only when the speed deviation Δω is within the predetermined range, the subsequent torque ripple reduction process is entered, and otherwise, the torque ripple reduction process is not executed. This determination is based on the fact that the operation of the synchronous motor is unstable when the speed deviation Δω is outside the predetermined range, and the step-out may occur due to careless control.

  Processing step S104 and subsequent steps are torque ripple reduction processing. First, processing steps S104 and S105 correspond to processing in the torque ripple limiting processor 42. Here, the current correction value iq6T of the equation (26) is finally calculated. For this purpose, the torque ripple T6 is calculated by using the induced voltage harmonic component eq6 of the q axis by the equation (24). The q-axis induced voltage harmonic component eq6 is expressed by equation (25).

In processing step S104, torque ripple T6 is obtained from equations (24) and (25). Most of the data used here is an existing value given in advance as a set value, and the data actually used for every moment is the d-axis current and q-axis obtained in the three-phase / dq conversion circuit 41. The detected current values idc and iqc, the q-axis voltage command value V ^* qc from the voltage command value generator 48, and the inverter angular velocity command value ω ^* 1.

  In processing step S105, a current correction value iq6T is obtained from equation (26). In the above description, the examples (24), (25), and (26) are sequentially executed. However, the expression (28) may be directly executed.

Processing step S106 corresponds to the processing of the subtractor sb, and subtracts the current correction value iq6T from i ^* q. This result is reflected in the voltage command value generator 48 and used for upstream compensation.

  Processing step S107 and processing step S108 correspond to downstream compensation. The processing step S107 corresponds to the processing of the torque ripple reduction voltage compensator 43, and the processing step S108 corresponds to the processing of the adder ad2. The series of processes described above is executed at every control cycle in the computer.

  Finally, several embodiments from the viewpoint of position sensorless will be described. In the embodiment shown in FIG. 1, an AC current is detected to obtain a dq-axis current and is used for phase control. According to this technique of the present invention, torque ripple can be reduced only with internal information that has been used conventionally, without requiring special external information, and thus without adding a special detector.

  Therefore, as long as the dq-axis current can be reproduced and used for phase control, a position sensorless method based on another principle can be used. What is introduced here is the DC current detection shown in FIG. This is a synchronous motor of the type in which a direct current supplied from the converter 1 to the inverter 3 is detected by a direct current detector 61.

  14 and FIG. 15 differ from the configuration of FIG. 1 only in the circuit portion that reproduces the dq-axis currents idc and iqc from the direct current. In FIG. 14, a current estimation unit is provided that estimates the dq-axis currents idc and iqc using the DC current i and the dq-axis voltages vdc and vqc from the voltage command value generator 48.

  In FIG. 15, the current reproducer 42 reproduces the three-phase alternating current from the direct current i. Then, the three-phase / dq converter 41 obtains dq axis currents idc and iqc.

  The methods shown in FIGS. 14 and 15 are both well-known in the technical field and do not require any special change or contrivance for application to the present invention. It is stated that the present invention can be realized even with such a position sensorless synchronous motor without requiring any special additional parts.

  The present invention can be widely applied to position sensorless synchronous motors.

1: DC power supply by converter 2: Motor controller 3: Inverter main circuit 4: Motor controller 6: Power conversion circuits 7a, 7b: Current detector 8: Permanent magnet synchronous motor PMSM
11: Gate drive circuit 41: Three-phase / dq conversion circuit 42: Torque ripple reduction controller 43: Torque ripple reduction voltage compensator 44: q-axis current command generator 45: Axis error calculator 46: PLL controller 47: integrator 48 : Voltage command value generator 49: d-axis current command generator 50: dq / three-phase converter circuit vd: d-axis voltage vq: q-axis voltage R: winding resistance Ld: d-axis inductance Lq: q-axis inductance id: d Axial current iq: q-axis current ω: electrical angular velocity KE: induced voltage constant θ: phase difference Te: main torque Pn: pole pair number iα, iβ: two-phase AC currents Vu, Vv, Vw: three-phase AC voltages iu, iv, iw: three-phase AC current D: antiparallel diode T: conversion element ω ^* 1: inverter angular velocity command value V ^* u, V ^* v, V ^* w: three-phase AC voltage command value θdc: dc axis angle ad1: addition circuit ed6: d axis Induced voltage harmonic component EQ6: Induction of q-axis voltage harmonic component T6: the torque ripple component Iq6T: current correction value sb: subtraction circuit

Claims (10)

In the control device for a synchronous motor to which three-phase alternating current is supplied from a power conversion circuit including an inverter,
Dq-axis current calculation means for obtaining the dq-axis current of the synchronous motor, a q-axis current command generator for obtaining a q-axis current command value from the output of the dq-axis current calculation means, and a q-axis voltage using the q-axis current command value A voltage command value generator for giving a command value, a conversion circuit for obtaining a three-phase AC voltage command value for driving the inverter from the output of the voltage command value generator, an induced voltage harmonic and a torque ripple of the synchronous motor A control apparatus for a synchronous motor, comprising: a torque ripple reduction controller for estimating and calculating a compensation signal for reducing torque ripple, and correcting the q-axis voltage command value by the compensation signal. In the synchronous motor control device according to claim 1,
A control apparatus for a synchronous motor, wherein the compensation signal is added to a q-axis voltage command value obtained from the voltage command value generator. In the synchronous motor control device according to claim 1 or 2,
A control apparatus for a synchronous motor, wherein the compensation signal is subtracted from a q-axis current command value obtained from the q-axis current command generator. In the synchronous motor control device according to claim 1,
The compensation signal is added to the q-axis voltage command value obtained from the voltage command value creator, and the compensation signal is subtracted from the q-axis current command value obtained from the q-axis current command creator. A control device for a synchronous motor. In the synchronous motor control device according to any one of claims 1 to 4,
The dq-axis current calculation means calculates a dq-axis current using an alternating current of the synchronous motor, and controls the synchronous motor. In the synchronous motor control device according to any one of claims 1 to 4,
The dq axis current calculation means calculates a dq axis current using a direct current flowing into an inverter of the power conversion circuit. In a control method of a synchronous motor supplied with a three-phase alternating current from a power conversion circuit including an inverter,
A q-axis voltage command value is obtained using the q-axis current command value of the synchronous motor, a three-phase AC voltage command value for driving the inverter is obtained from the q-axis voltage command value, and an induced voltage harmonic of the synchronous motor is obtained. A control method for a synchronous motor, wherein a compensation signal for reducing torque ripple is calculated from a wave and torque ripple, and thereby the q-axis voltage command value is corrected. In the control method of the synchronous motor according to claim 7,
A method for controlling a synchronous motor, wherein the compensation signal is added to the q-axis voltage command value. In the control method of the synchronous motor according to claim 7 or 8,
A method for controlling a synchronous motor, wherein the compensation signal is subtracted from the q-axis current command value. In the control method of the synchronous motor according to claim 7,
A method for controlling a synchronous motor, wherein the compensation signal is added to the q-axis voltage command value, and the compensation signal is subtracted from the q-axis current command value.

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