JP2008206323A - Motor driving device - Google Patents

Motor driving device Download PDF

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JP2008206323A
JP2008206323A JP2007040224A JP2007040224A JP2008206323A JP 2008206323 A JP2008206323 A JP 2008206323A JP 2007040224 A JP2007040224 A JP 2007040224A JP 2007040224 A JP2007040224 A JP 2007040224A JP 2008206323 A JP2008206323 A JP 2008206323A
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motor
voltage
temperature
current
value
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Mitsuo Kawachi
光夫 河地
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Panasonic Holdings Corp
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Matsushita Electric Industrial Co Ltd
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Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor driving device that improves transient characteristics of a motor by increasing a phase difference of a motor current with respect to a preset induced voltage when an operating temperature of a motor is low. <P>SOLUTION: The motor driving device includes an inverter, which has a plurality of switching-element pairs, respectively composed of an upper-arm switching element arranged on the high-voltage side and a lower-arm switching element arranged on the low-voltage side, and converts a DC voltage into an AC voltage having a desired frequency and a desired voltage by operation of each switching element so as to supply the AC voltage to a motor having a plurality of phases as its drive voltage, an inverter control means for giving a drive signal to the inverter so as to flow a motor current having a prescribed phase difference to an induced current of a motor, and a temperature detection means for detecting at least one of temperatures of a winding temperature of a motor, a rotor-magnet temperature, and an atmospheric temperature. When a temperature detection value detected by the temperature detection means is smaller than a prescribed value, a preset phase difference is increased. <P>COPYRIGHT: (C)2008,JPO&INPIT

Description

本発明は、ブラシレスDCモータなどの電動機を任意の回転数で駆動する電動機駆動装置に関するものである。   The present invention relates to an electric motor drive device that drives an electric motor such as a brushless DC motor at an arbitrary rotation speed.

近年、空気調和機における圧縮機などの電動機を駆動する装置においては、地球環境保護の観点から消費電力を低減する必要性が大きくなっている。   In recent years, in an apparatus for driving an electric motor such as a compressor in an air conditioner, there is an increasing need to reduce power consumption from the viewpoint of protecting the global environment.

その中で、省電力の技術の一つとして、ブラシレスDCモータのような高効率な電動機を任意の周波数で駆動するインバータなどが広く一般に使用されている。さらに、駆動する技術としては、矩形波状の電流により駆動を行う矩形波駆動に対して、より効率が高く、騒音も低くすることが可能な正弦波駆動技術が注目されている。   Among them, as one of the power saving techniques, an inverter that drives a highly efficient electric motor such as a brushless DC motor at an arbitrary frequency is widely used. Furthermore, as a driving technique, a sine wave driving technique that is more efficient and can reduce noise is attracting attention as compared with the rectangular wave driving that is driven by a rectangular wave current.

空気調和機における圧縮機に備えられた電動機を駆動する場合、電動機の回転子の位置を検出するセンサを取りつけることが困難であるため、回転子の位置を何らかの方法で推定しながら駆動を行う位置センサレス正弦波駆動の技術が考案されている。回転子の位置を推定する方法としては、電動機の固定子巻線に生ずる誘起電圧を推定することにより行う方法がある(例えば、特許文献1参照)。   When driving an electric motor provided in a compressor of an air conditioner, it is difficult to attach a sensor for detecting the position of the rotor of the electric motor. A sensorless sinusoidal drive technology has been devised. As a method for estimating the position of the rotor, there is a method in which an induced voltage generated in a stator winding of an electric motor is estimated (see, for example, Patent Document 1).

図5に特許文献1の電動機駆動装置のシステム構成を示す。図5に示すように、インバータ制御部7は、複数のスイッチング素子4a〜4fと対をなす還流ダイオード5a〜5fからなるインバータ4と、電流検出部8、正弦波駆動部9、PWM信号生成部10、誘起電圧推定部11、位置速度推定部12から構成される。   FIG. 5 shows a system configuration of the electric motor drive device of Patent Document 1. As shown in FIG. 5, the inverter control unit 7 includes an inverter 4 including freewheeling diodes 5 a to 5 f that are paired with a plurality of switching elements 4 a to 4 f, a current detection unit 8, a sine wave drive unit 9, and a PWM signal generation unit. 10, an induced voltage estimation unit 11, and a position / speed estimation unit 12.

交流電源1からの入力電圧は整流回路2で直流に整流され、その直流電圧はインバータ4により3相の交流電圧に変換され、それによりブラシレスDCモータである電動機3が駆動される。   The input voltage from the AC power supply 1 is rectified to DC by the rectifier circuit 2, and the DC voltage is converted into a three-phase AC voltage by the inverter 4, thereby driving the electric motor 3 which is a brushless DC motor.

ここで、誘起電圧推定手段である誘起電圧推定部11は、電流検出値(iu、iv、iw)と電圧指令値(vu*、vv*、vw*)とに基づいて、電動機3の固定子巻線の各相に生じる誘起電圧を推定し、誘起電圧推定値(eu、ev、ew)として出力する。   Here, the induced voltage estimation unit 11 serving as the induced voltage estimation means is based on the detected current value (iu, iv, iw) and the voltage command value (vu *, vv *, vw *). The induced voltage generated in each phase of the winding is estimated and output as an induced voltage estimated value (eu, ev, ew).

位置速度推定手段である位置速度推定部11は、誘起電圧推定値(eu、ev、ew)を用いて、電動機の回転子の磁極位置および回転速度を推定し、それぞれ磁極位置推定値θ、回転速度推定値ωとして出力する。電流検出器6a、6bおよび電流検出部8は、電動機3の電流検出値(iu、iv、iw)を出力する電流検出手段を構成する。   The position / speed estimation unit 11 serving as a position / speed estimation means estimates the magnetic pole position and the rotational speed of the rotor of the motor using the induced voltage estimated values (eu, ev, ew). Output as the estimated speed value ω. The current detectors 6 a and 6 b and the current detection unit 8 constitute current detection means for outputting a current detection value (iu, iv, iw) of the electric motor 3.

正弦波駆動部9は、目標回転速度ω*と回転速度推定値ωとの差を速度誤差として算出し、その速度誤差と、電流検出値(iu、iv、iw)と、磁極位置推定値θとから、電動機3の電圧指令値(vu*、vv*、vw*)を演算する。PWM信号生成部10は、電圧指令値(vu*、vv*、vw*)に基づいてインバータ4のスイッチング素子4a〜4fを動作させるためのドライブ信号を生成する。   The sine wave drive unit 9 calculates the difference between the target rotational speed ω * and the rotational speed estimated value ω as a speed error, the speed error, the current detection value (iu, iv, iw), and the magnetic pole position estimated value θ. From these, voltage command values (vu *, vv *, vw *) of the electric motor 3 are calculated. The PWM signal generation unit 10 generates a drive signal for operating the switching elements 4a to 4f of the inverter 4 based on the voltage command values (vu *, vv *, vw *).

なお、正弦波駆動部9の動作は次の通りである。外部から与えられる電動機3の目標回転速度ω*と回転速度推定値ωとの速度誤差がゼロになるように、式(1)で表されるPI制御により電流指令振幅I*を求める。   The operation of the sine wave drive unit 9 is as follows. The current command amplitude I * is obtained by PI control represented by Expression (1) so that the speed error between the target rotational speed ω * of the motor 3 given from the outside and the rotational speed estimated value ω becomes zero.

I*=KPW・(ω*−ω)+KIW・Σ(ω*−ω) …(1)
式(1)の演算により求められる電流指令振幅I*と、0〜90[deg](0〜π/2[rad])の範囲における予め設定された電流位相βTとを用い、式(2)、(3)の演算により、dq軸電流指令(id*、iq*)を求める。
I * = KPW · (ω * −ω) + KIW · Σ (ω * −ω) (1)
Using the current command amplitude I * obtained by the calculation of Expression (1) and a preset current phase βT in the range of 0 to 90 [deg] (0 to π / 2 [rad]), Expression (2) , (3) to obtain the dq axis current command (id *, iq *).

電流位相βTは、例えば、図4に示すように電動機3の回転速度ωがωs0よりも小さい低速〜中速域では、システム効率が最大となるような最適位相βs0とし、回転速度がωs0〜ωe0の中速〜高速域では、一次関数的に電流位相をβs0からβe0まで増加させ、回転速度がωe以上の高速域では最大位相βe0となるように設定している。   For example, as shown in FIG. 4, the current phase βT is set to the optimum phase βs0 that maximizes the system efficiency in the low to medium speed range where the rotational speed ω of the electric motor 3 is smaller than ωs0, and the rotational speed is ωs0 to ωe0. In the medium speed to high speed range, the current phase is increased from βs0 to βe0 in a linear function, and the maximum phase βe0 is set in the high speed range where the rotational speed is equal to or higher than ωe.

id*=−I*・sin(βT) …(2)
iq*=I*・cos(βT) …(3)
また、固定子巻線の相電流指令(iu*、iv*、iw*)は、dq軸電流指令(id*、iq*)と磁極位置推定値θとを用い、式(4)〜(6)の演算により、2相−3相変換を行うことで求められる。2相−3相変換については説明を省略する。
id * = − I * · sin (βT) (2)
iq * = I * · cos (βT) (3)
Further, the phase current command (iu *, iv *, iw *) of the stator winding uses the dq axis current command (id *, iq *) and the magnetic pole position estimated value θ, and the equations (4) to (6) ) To obtain two-phase to three-phase conversion. Description of the two-phase to three-phase conversion is omitted.

iu*={√(2/3)}・{id*・cosθ}
−iq*・sinθ} …(4)
iv*={√(2/3)}・{id*・cos(θ−2π/3)
−iq*・sin(θ−2π/3)} …(5)
iw*={√(2/3)}・{id*・cos(θ+2π/3)
−iq*・sin(θ+2π/3)} …(6)
ここで、相電流指令値(iu*、iv*、iw*)と電流検出器20a,20bおよび電流検出部8から得られる相電流検出値(iu、iv、iw)との電流誤差がゼロとなるように、電流制御ゲイン(KPKn:比例ゲイン、KIKn:積分ゲイン、n=1、2、3(3相分))を用い、式(7)〜式(9)で表されるPI制御により相電圧指令(vu*、vv*、vw*)を演算する。
iu * = {√ (2/3)} · {id * · cos θ}
−iq * · sin θ} (4)
iv * = {√ (2/3)} · {id * · cos (θ-2π / 3)
−iq * · sin (θ−2π / 3)} (5)
iw * = {√ (2/3)} · {id * · cos (θ + 2π / 3)
−iq * · sin (θ + 2π / 3)} (6)
Here, the current error between the phase current command value (iu *, iv *, iw *) and the phase current detection value (iu, iv, iw) obtained from the current detectors 20a, 20b and the current detection unit 8 is zero. In this way, the current control gain (KPKn: proportional gain, KIKn: integral gain, n = 1, 2, 3 (for three phases)) is used, and PI control represented by Expression (7) to Expression (9) is performed. Phase voltage commands (vu *, vv *, vw *) are calculated.

vu*=KPK1・(iu*−iu)+KIK1・Σ(iu*−iu) …(7)
vv*=KPK2・(iv*−iv)+KIK2・Σ(iv*−iv) …(8)
vw*=KPK3・(iw*−iw)+KIK3・Σ(iw*−iw) …(9)
なお、相電流検出値(iu、iv、iw)を3相−2相変換してdq軸電流検出値(id、iq)を求め、dq軸電流指令(id*、iq*)とdq軸電流検出値(id、iq)との電流誤差がゼロとなるよう、PI制御によってdq軸電圧指令(vd*、vq*)を求め、dq軸電圧指令(vd*、vq*)を2相−3相変換して相電圧指令(vu*、vv*、vw*)を求めても良い。3相−2相変換についても説明を省略する。
vu * = KPK1 · (iu * −iu) + KIK1 · Σ (iu * −iu) (7)
vv * = KPK2 · (iv * −iv) + KIK2 · Σ (iv * −iv) (8)
vw * = KPK3 · (iw * −iw) + KIK3 · Σ (iw * −iw) (9)
The phase current detection values (iu, iv, iw) are three-phase to two-phase converted to obtain the dq axis current detection values (id, iq), the dq axis current command (id *, iq *) and the dq axis current. The dq-axis voltage command (vd *, vq *) is obtained by PI control so that the current error from the detected value (id, iq) becomes zero, and the dq-axis voltage command (vd *, vq *) is determined as two-phase-3. The phase voltage command (vu *, vv *, vw *) may be obtained by phase conversion. The description of the three-phase to two-phase conversion is also omitted.

具体的には、dq軸電流検出値(id、iq)は、式(10)、(11)の演算により求められる。   Specifically, the dq axis current detection value (id, iq) is obtained by the calculations of equations (10) and (11).

id={√(2)}・{iu・sin(θ+π/3)+iv・sinθ}…(10)
iq={√(2)}・{iu・cos(θ+π/3)+iv・cosθ}…(11)
dq軸電圧指令(vd*、vq*)は、電流制御ゲイン(KPD:d軸電流比例ゲイン、KID:d軸電流積分ゲイン、KPQ:q軸電流比例ゲイン、KIQ:q軸電流積分ゲイン)を用い、式(12)、(13)の演算により求められる。
id = {√ (2)} · {iu · sin (θ + π / 3) + iv · sin θ} (10)
iq = {√ (2)} · {iu · cos (θ + π / 3) + iv · cos θ} (11)
The dq-axis voltage command (vd *, vq *) is a current control gain (KPD: d-axis current proportional gain, KID: d-axis current integral gain, KPQ: q-axis current proportional gain, KIQ: q-axis current integral gain). And obtained by the calculations of equations (12) and (13).

vd*=KPD・(id*−id)+KID・Σ(id*−id) …(12)
vq*=KPQ・(iq*−iq)+KIQ・Σ(iq*−iq) …(13)
そこで、相電圧指令(vu*、vv*、vw*)は、dq軸電圧指令(vd*、vq*)を2相−3相変換することにより、式(14)〜(16)の演算により求められる。
vd * = KPD · (id * −id) + KID · Σ (id * −id) (12)
vq * = KPQ · (iq * −iq) + KIQ · Σ (iq * −iq) (13)
Therefore, the phase voltage commands (vu *, vv *, vw *) are obtained by calculating the equations (14) to (16) by performing two-phase to three-phase conversion on the dq axis voltage commands (vd *, vq *). Desired.

vu*={√(2/3)}・{vd*・cosθ
−vq*・sinθ} …(14)
vv*={√(2/3)}・{vd*・cos(θ−2π/3)
−vq*・sin(θ−2π/3)} …(15)
vw*={√(2/3)}・{vd*・cos(θ+2π/3)
−vq*・sin(θ+2π/3)} …(16)
次に、誘起電圧推定部11による電動機3の誘起電圧の推定方法について説明する。まず、各相の誘起電圧推定値(eu、ev、ew)は、相電流検出値(iu、iv、iw)と、相電圧指令(vu*、vv*、vw*)を用い、式(17)〜(19)の演算により求められる。
vu * = {√ (2/3)} · {vd * · cos θ
−vq * · sin θ} (14)
vv * = {√ (2/3)} · {vd * · cos (θ−2π / 3)
-Vq * · sin (θ-2π / 3)} (15)
vw * = {√ (2/3)} · {vd * · cos (θ + 2π / 3)
−vq * · sin (θ + 2π / 3)} (16)
Next, a method for estimating the induced voltage of the electric motor 3 by the induced voltage estimation unit 11 will be described. First, the induced voltage estimated value (eu, ev, ew) of each phase is obtained by using the phase current detection value (iu, iv, iw) and the phase voltage command (vu *, vv *, vw *) and the equation (17). ) To (19).

eu=vu*−R・iu−L・d(iu)/dt …(17)
ev=vv*−R・iv−L・d(iv)/dt …(18)
ew=vw*−R・iw−L・d(iw)/dt …(19)
ここで、Rは電動機3の巻線1相あたりの抵抗、Lはそのインダクタンスである。d(iu)/dt、d(iv)/dt、d(iw)/dtはそれぞれiu、iv、iwの時間微分である。式(17)〜(19)を展開すると、式(20)〜(22)が得られる。
eu = vu * −R · iu−L · d (iu) / dt (17)
ev = vv * −R · iv−L · d (iv) / dt (18)
ew = vw * −R · iw−L · d (iw) / dt (19)
Here, R is the resistance per phase of the winding of the electric motor 3, and L is its inductance. d (iu) / dt, d (iv) / dt, and d (iw) / dt are time derivatives of iu, iv, and iw, respectively. When formulas (17) to (19) are expanded, formulas (20) to (22) are obtained.

eu=vu*−R・iu
−(la+La)・d(iu)/dt
−Las・cos(2θ)・d(iu)/dt
−Las・iu・d{cos(2θ)}/dt
+0.5・La・d(iv)/dt
−Las・cos(2θ−2π/3)・d(iv)/dt
−Las・iv・d{cos(2θ−2π/3)}/dt
+0.5・La・d(iw)/dt
−Las・cos(2θ+2π/3)・d(iw)/dt
−Las・iw・d{cos(2θ+2π/3)}/dt …(20)
ev=vv*−R・iv
−(la+La)・d(iv)/dt
−Las・cos(2θ+2π/3)・d(iv)/dt
−Las・iv・d{cos(2θ+2π/3)}/dt
+0.5・La・d(iw)/dt
−Las・cos(2θ)・d(iw)/dt
−Las・iw・d{cos(2θ)}/dt
+0.5・La・d(iu)/dt
−Las・cos(2θ−2π/3)・d(iu)/dt
−Las・iu・d{cos(2θ−2π/3)}/dt …(21)
ew=vw*−R・iw
−(la+La)・d(iw)/dt
−Las・cos(2θ−2π/3)・d(iw)/dt
−Las・iw・d{cos(2θ−2π/3)}/dt
+0.5・La・d(iu)/dt
−Las・cos(2θ+2π/3)・d(iu)/dt
−Las・iu・d{cos(2θ+2π/3)}/dt
+0.5・La・d(iv)/dt
−Las・cos(2θ)・d(iv)/dt
−Las・iv・d{cos(2θ)}/dt …(22)
ここで、Rは巻線1相あたりの抵抗、laは巻線1相あたりの漏れインダクタンス、L
aは巻線1相あたりの有効インダクタンスの平均値、Lasは巻線1相あたりの有効インダクタンスの振幅である。d(iu)/dt、d(iv)/dt、d(iw)/dtは、1次オイラー近似で求める。
eu = vu * -R · iu
− (La + La) · d (iu) / dt
-Las · cos (2θ) · d (iu) / dt
-Las · iu · d {cos (2θ)} / dt
+ 0.5 · La · d (iv) / dt
−Las · cos (2θ−2π / 3) · d (iv) / dt
-Las · iv · d {cos (2θ-2π / 3)} / dt
+ 0.5 · La · d (iw) / dt
−Las · cos (2θ + 2π / 3) · d (iw) / dt
-Las · iw · d {cos (2θ + 2π / 3)} / dt (20)
ev = vv * −R · iv
− (La + La) · d (iv) / dt
-Las · cos (2θ + 2π / 3) · d (iv) / dt
−Las · iv · d {cos (2θ + 2π / 3)} / dt
+ 0.5 · La · d (iw) / dt
−Las · cos (2θ) · d (iw) / dt
-Las · iw · d {cos (2θ)} / dt
+ 0.5 · La · d (iu) / dt
−Las · cos (2θ−2π / 3) · d (iu) / dt
-Las · iu · d {cos (2θ-2π / 3)} / dt (21)
ew = vw * -R · iw
− (La + La) · d (iw) / dt
-Las · cos (2θ-2π / 3) · d (iw) / dt
-Las · iw · d {cos (2θ-2π / 3)} / dt
+ 0.5 · La · d (iu) / dt
−Las · cos (2θ + 2π / 3) · d (iu) / dt
-Las · iu · d {cos (2θ + 2π / 3)} / dt
+ 0.5 · La · d (iv) / dt
−Las · cos (2θ) · d (iv) / dt
-Las · iv · d {cos (2θ)} / dt (22)
Where R is the resistance per phase of the winding, la is the leakage inductance per phase of the winding, L
a is the average value of the effective inductance per phase of the winding, and Las is the amplitude of the effective inductance per phase of the winding. d (iu) / dt, d (iv) / dt, and d (iw) / dt are obtained by first-order Euler approximation.

u相電流iuは、v相電流ivとw相電流iwとの和を求め、その符号を変えることにより得られる値とする。ここで、相電流検出値(iu、iv、iw)が正弦波であると仮定し、電流指令振幅I*と電流位相βTとを用い、相電流検出値(iu、iv、iw)を簡略化すると、以下に示す式(23)〜(25)が得られる。   The u-phase current iu is a value obtained by calculating the sum of the v-phase current iv and the w-phase current iw and changing the sign thereof. Here, assuming that the phase current detection values (iu, iv, iw) are sine waves, the phase current detection values (iu, iv, iw) are simplified using the current command amplitude I * and the current phase βT. Then, the following formulas (23) to (25) are obtained.

eu=vu*+R・I*・sin(θ+βT)
+1.5・(la+La)・cos(θ+βT)
−1.5・Las・cos(θ−βT) …(23)
ev=vv*+R・I*・sin(θ+βT−2π/3)
+1.5・(la+La)・cos(θ+βT−2π/3)
−1.5・Las・cos(θ−βT−2π/3) …(24)
ew=vw*+R・I*・sin(θ+βT+2π/3)
+1.5・(la+La)・cos(θ+βT+2π/3)
−1.5・Las・cos(θ−βT+2π/3) …(25)
これらの式を用いて、誘起電圧推定部11は、誘起電圧推定値(eu、ev、ew)を求める。
eu = vu * + R · I * · sin (θ + βT)
+1.5 ・ (la + La) ・ cos (θ + βT)
-1.5 · Las · cos (θ−βT) (23)
ev = vv * + R · I * · sin (θ + βT−2π / 3)
+ 1.5 · (la + La) · cos (θ + βT-2π / 3)
−1.5 · Las · cos (θ−βT−2π / 3) (24)
ew = vw * + R · I * · sin (θ + βT + 2π / 3)
+ 1.5 · (la + La) · cos (θ + βT + 2π / 3)
−1.5 · Las · cos (θ−βT + 2π / 3) (25)
Using these equations, the induced voltage estimation unit 11 calculates an induced voltage estimated value (eu, ev, ew).

位置速度推定部12は、各相の誘起電圧基準値(eum、evm、ewm)を、次に示す3つの式によりそれぞれ求める。ここで、誘起電圧振幅値emは、eu、ev、ewの振幅値と一致させることで求められる。   The position / velocity estimation unit 12 obtains the induced voltage reference values (eum, evm, ewm) of each phase according to the following three equations. Here, the induced voltage amplitude value em is obtained by matching the amplitude values of eu, ev, and ew.

eum=em・sin(θ+βT) …(26)
evm=em・sin(θ+βT−2π/3) …(27)
ewm=em・sin(θ+βT+2π/3) …(28)
このようにして求めた誘起電圧基準値esmと、誘起電圧推定値esとの偏差εを求める。ここで、esmは、eum、evm、または、ewmを意味し、esは、eu、ev、または、ewを意味する。
eum = em · sin (θ + βT) (26)
evm = em · sin (θ + βT−2π / 3) (27)
ewm = em · sin (θ + βT + 2π / 3) (28)
A deviation ε between the induced voltage reference value esm thus obtained and the induced voltage estimated value es is obtained. Here, esm means eu, evm, or ewm, and es means eu, ev, or ew.

ε=es−esm (s=u、v、または、w) …(29)
この偏差εが0になれば磁極位置推定値θが真値となるので、偏差εを0に収束させるように、偏差εを用いたPI演算などを行って、磁極位置推定値θが求められる。また、磁極位置推定値θの変動値を演算することにより、回転速度推定値ωが求められる。
ε = es-esm (s = u, v, or w) (29)
When this deviation ε becomes 0, the magnetic pole position estimated value θ becomes a true value. Therefore, PI calculation using the deviation ε is performed so that the deviation ε converges to 0, and the magnetic pole position estimated value θ is obtained. . Further, the rotational speed estimated value ω is obtained by calculating the fluctuation value of the magnetic pole position estimated value θ.

このように、従来の電動機駆動装置は、相電圧方程式に基づいたモデルにより導出された誘起電圧推定値と誘起電圧基準値との偏差εを用いて磁極位置推定値θを生成し、正弦波状の相電流を流すことで位置センサレス正弦波駆動を実現している。
特許第3419725号公報
As described above, the conventional electric motor drive device generates the magnetic pole position estimated value θ using the deviation ε between the induced voltage estimated value derived from the model based on the phase voltage equation and the induced voltage reference value, and has a sinusoidal shape. Position sensorless sine wave drive is realized by flowing phase current.
Japanese Patent No. 3419725

しかしながら、従来の構成では、電動機3の動作温度(巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度)によって電流位相βTを変更する手段を備えていないため、定常状態の動作温度を考慮してシステム効率が最大となるような電流位相βTを予め設定すると、特に電動機3の雰囲気温度が極めて低い場合で、長時間電動機3を停止させた状態から電動機3を高速まで立ち上げる際に、回転子磁石温度が低いことで電動機3の誘起電圧が大きくなり、電圧飽和となることで電動機3の過渡特性(立ち上げ
性能)が低下するという課題を有していた。
However, since the conventional configuration does not include means for changing the current phase βT according to the operating temperature of the electric motor 3 (winding temperature, rotor magnet temperature, or ambient temperature), the operation in the steady state If the current phase βT is set so that the system efficiency is maximized in consideration of the temperature, the motor 3 is started up to a high speed from a state in which the motor 3 is stopped for a long time, particularly when the ambient temperature of the motor 3 is extremely low. However, the rotor magnet temperature is low, the induced voltage of the electric motor 3 is increased, and the voltage saturation causes a problem that the transient characteristic (startup performance) of the electric motor 3 is reduced.

なお、一般的に、ブラシレスDCモータの永久磁石には、フェライト磁石や希土類磁石が採用されているが、永久磁石による磁束が飽和しない限りは、回転速度が一定の場合でも永久磁石の温度が高くなるにつれて誘起電圧は小さくなるという関係が知られている。   In general, a ferrite magnet or a rare earth magnet is used as a permanent magnet of a brushless DC motor. However, as long as the magnetic flux generated by the permanent magnet is not saturated, the temperature of the permanent magnet is high even when the rotational speed is constant. It is known that the induced voltage decreases with time.

本発明は、前記従来の課題を解決するもので、電動機3の動作温度(巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度)によって予め設定された誘起電圧に対する電動機電流の位相差(電流位相βT)を変更し、電動機3の過渡特性(立ち上げ性能)を向上させる電動機駆動装置を提供することを目的とする。   The present invention solves the above-mentioned conventional problems, and the motor current is detected with respect to the induced voltage set in advance by the operating temperature of the motor 3 (winding temperature, rotor magnet temperature, or ambient temperature). An object of the present invention is to provide an electric motor drive device that changes the phase difference (current phase βT) and improves the transient characteristics (startup performance) of the electric motor 3.

上記従来の課題を解決するために、本発明の電動機駆動装置は、高圧側に配置された上アームスイッチング素子と低圧側に配置された下アームスイッチング素子からなるスイッチング素子対を複数有し、各スイッチング素子の動作により直流電圧を所望の周波数、電圧の交流電圧に変換し、複数相の電動機にその駆動電圧として供給するインバータと、電動機の誘起電圧に対して所定の位相差の電動機電流が流れるようにインバータに駆動信号を与えるインバータ制御手段と、電動機の巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度を検出する温度検出手段を備え、温度検出手段によって検出された温度検出値が所定値よりも小さい場合には予め設定された位相差を大きくするものである。   In order to solve the above-described conventional problems, the electric motor drive device of the present invention has a plurality of switching element pairs each composed of an upper arm switching element arranged on the high voltage side and a lower arm switching element arranged on the low voltage side, An inverter that converts a DC voltage into an AC voltage of a desired frequency and voltage by the operation of the switching element and supplies it as a drive voltage to a multi-phase motor, and a motor current having a predetermined phase difference with respect to the induced voltage of the motor flows Inverter control means for supplying a drive signal to the inverter and temperature detection means for detecting at least one of the winding temperature of the motor, the rotor magnet temperature, and the ambient temperature, and the temperature detected by the temperature detection means When the detected value is smaller than the predetermined value, the preset phase difference is increased.

これによって、電動機の動作温度が所定値よりも低い場合には、予め設定された誘起電圧に対する電動機電流の位相差(電流位相)を大きくすることで、弱め界磁制御の効果を強め、電動機の過渡特性(立ち上げ性能)を向上させることができる。   As a result, when the operating temperature of the motor is lower than the predetermined value, the effect of field weakening control is enhanced by increasing the phase difference (current phase) of the motor current with respect to the preset induced voltage, and the transient characteristics of the motor (Start-up performance) can be improved.

本発明の電動機駆動装置によれば、電動機の動作温度(巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度)が低い場合に予め設定された誘起電圧に対する電動機電流の位相差(電流位相)を大きくすることで、電動機の過渡特性(立ち上げ性能)を向上させることができる。   According to the motor driving device of the present invention, the phase difference of the motor current with respect to the preset induced voltage when the operating temperature of the motor (at least one of the winding temperature, the rotor magnet temperature, and the ambient temperature) is low. By increasing the (current phase), it is possible to improve the transient characteristics (startup performance) of the electric motor.

第1の発明は、高圧側に配置された上アームスイッチング素子と低圧側に配置された下アームスイッチング素子からなるスイッチング素子対を複数有し、各スイッチング素子の動作により直流電圧を所望の周波数、電圧の交流電圧に変換し、複数相の電動機にその駆動電圧として供給するインバータと、電動機の誘起電圧に対して所定の位相差の電動機電流が流れるようにインバータに駆動信号を与えるインバータ制御手段と、電動機の巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度を検出する温度検出手段とを備え、温度検出手段によって検出された温度検出値が所定値よりも小さい場合には予め設定された位相差を大きくするものである。   The first invention has a plurality of switching element pairs each composed of an upper arm switching element arranged on the high voltage side and a lower arm switching element arranged on the low voltage side, and a DC voltage is converted to a desired frequency by the operation of each switching element. An inverter that converts the voltage into an AC voltage and supplies the drive voltage to the motors of a plurality of phases; and an inverter control means that supplies a drive signal to the inverter so that a motor current having a predetermined phase difference with respect to the induced voltage of the motor flows. And a temperature detection means for detecting at least one of the winding temperature of the motor, the rotor magnet temperature, and the ambient temperature, and when the temperature detection value detected by the temperature detection means is smaller than a predetermined value The preset phase difference is increased.

これにより、電動機の動作温度が所定値よりも低い場合には、予め設定された誘起電圧に対する電動機電流の位相差(電流位相)かを大きくすることで、弱め界磁制御の効果を強め、電動機の過渡特性(立ち上げ性能)を向上させることができる。   As a result, when the operating temperature of the motor is lower than a predetermined value, the effect of field-weakening control is increased by increasing the phase difference (current phase) of the motor current with respect to the preset induced voltage, and the transient of the motor The characteristics (startup performance) can be improved.

第2の発明は、特に第1の発明において、温度検出手段は、電動機に印加される電動機電圧、電動機の固定子巻線に流れる電動機電流、電動機の回転子の磁極位置および回転速度に基づいて、電動機の巻線抵抗値もしくは誘起電圧定数の少なくともいずれか1つの電動機定数の温度変化を推定する定数推定手段を備え、定数推定手段によって推定された電
動機定数推定値と予め設定された電動機定数基準値との比率から、電動機の巻線温度もしくは回転子磁石温度の少なくともいずれか1つを算出するものである。
According to a second aspect of the invention, particularly in the first aspect of the invention, the temperature detecting means is based on the motor voltage applied to the motor, the motor current flowing in the stator winding of the motor, the magnetic pole position and the rotation speed of the rotor of the motor. , Comprising constant estimation means for estimating a temperature change of at least one of the winding resistance value or the induced voltage constant of the motor, the motor constant estimated value estimated by the constant estimation means and a preset motor constant reference At least one of the winding temperature of the electric motor and the rotor magnet temperature is calculated from the ratio to the value.

これにより、サーミスタや熱電対等の、温度を検出する温度センサが不要となるため、コスト低減と信頼性向上とを図ることができる。   This eliminates the need for a temperature sensor that detects temperature, such as a thermistor or a thermocouple, thereby reducing costs and improving reliability.

第3の発明は、特に第1の発明において、インバータ制御手段は、電動機の固定子巻線に流れる電動機電流を検出する電流検出手段と、電動機に印加される電動機電圧と電動機電流とから電動機に発生する誘起電圧を推定する誘起電圧推定手段と、誘起電圧に基づいて、電動機の回転子の磁極位置および回転速度を推定する位置速度推定手段とを備えるものである。   According to a third aspect of the present invention, in the first aspect of the invention, the inverter control means is provided in the motor from current detection means for detecting the motor current flowing in the stator winding of the motor, the motor voltage applied to the motor, and the motor current. An induced voltage estimating means for estimating the generated induced voltage and a position speed estimating means for estimating the magnetic pole position and the rotational speed of the rotor of the electric motor based on the induced voltage are provided.

これにより、エンコーダやレゾルバ等の、回転子の磁極位置を検出する位置センサが不要となるため、コスト低減と信頼性向上とを図ることができる。   This eliminates the need for position sensors that detect the magnetic pole position of the rotor, such as encoders and resolvers, thereby reducing costs and improving reliability.

以下、本発明の実施の形態について、図面を参照しながら説明する。なお、この実施の形態によって本発明が限定されるものではない。   Hereinafter, embodiments of the present invention will be described with reference to the drawings. Note that the present invention is not limited to the embodiments.

(実施の形態1)
図1は、本発明の実施の形態1における電動機駆動装置のシステム構成図を示すものである。図5に示す従来の電動機駆動装置と同じ構成要素は同一符号で示してあり、その説明は重複するため省略し、ここでは異なる部分についてのみ述べる。
(Embodiment 1)
FIG. 1 shows a system configuration diagram of an electric motor drive device according to Embodiment 1 of the present invention. The same components as those of the conventional electric motor drive device shown in FIG. 5 are denoted by the same reference numerals, and the description thereof is omitted because it is duplicated. Only different portions will be described here.

図1において、温度検出器15および温度検出部14は、電動機3の動作温度検出値(巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度)を出力する温度検出手段を構成する。   In FIG. 1, the temperature detector 15 and the temperature detector 14 constitute a temperature detector that outputs an operating temperature detection value (at least one of winding temperature, rotor magnet temperature, and ambient temperature) of the electric motor 3. To do.

電流位相補正部13は、電動機3の動作温度検出値が所定値よりも小さい場合には予め設定された誘起電圧に対する電動機電流の位相差(電流位相)が大きくなるように電流位相を補正し、電流位相補正値として出力する。正弦波駆動部9は、電流位相補正部13からの出力された電流位相補正値に基づいて、式(2)、式(3)および式(23)〜式(28)の演算を行う。   The current phase correction unit 13 corrects the current phase so that the phase difference (current phase) of the motor current with respect to the preset induced voltage becomes large when the operating temperature detection value of the motor 3 is smaller than a predetermined value, Output as current phase correction value. Based on the current phase correction value output from the current phase correction unit 13, the sine wave driving unit 9 performs calculations of Expression (2), Expression (3), and Expression (23) to Expression (28).

次に、電流位相補正部13の具体的な動作について図3を用いて説明する。図3において、電動機3の動作温度が所定値(例えば、20℃)よりも高い場合には、図4と同様に、電動機3の回転速度ωがωs0よりも小さい低速〜中速域では、システム効率が最大となるような最適位相β0sとし、回転速度がωs0〜ωe0の中速〜高速域では、一次関数的に電流位相をβs0からβe0まで増加させ、回転速度がωe0以上の高速域では最大位相βe0となるように設定し、電動機3の動作温度が所定値(例えば、20℃)以下となる場合には、回転速度に対する電流位相の設定値を、動作温度が高い場合よりも値が大きくなるように設定している。   Next, a specific operation of the current phase correction unit 13 will be described with reference to FIG. In FIG. 3, when the operating temperature of the electric motor 3 is higher than a predetermined value (for example, 20 ° C.), as in FIG. 4, in the low to medium speed range where the rotational speed ω of the electric motor 3 is smaller than ωs0, the system The optimum phase β0s that maximizes the efficiency is set, the current phase is increased linearly from βs0 to βe0 in the medium speed to high speed range of ωs0 to ωe0, and the maximum in the high speed range where the rotation speed is ωe0 or higher. When the phase βe0 is set and the operating temperature of the electric motor 3 is a predetermined value (for example, 20 ° C.) or less, the set value of the current phase with respect to the rotational speed is larger than that when the operating temperature is high. It is set to be.

すなわち、電動機3の回転速度ωがωs1よりも小さい低速〜中速域では、電流位相をβs1(βs0よりも値が大)とし、回転速度がωs1〜ωe1の中速〜高速域では、一次関数的に電流位相をβs1からβe1まで増加させ、回転速度がωe1以上の高速域では最大位相βe1(βe0よりも値が大)となるように設定するものである。   That is, the current phase is βs1 (value is larger than βs0) in the low to medium speed range where the rotational speed ω of the electric motor 3 is smaller than ωs1, and the linear function is in the medium speed to high speed range of the rotational speed ωs1 to ωe1. In particular, the current phase is increased from βs1 to βe1, and the maximum phase βe1 (a value larger than βe0) is set in the high speed region where the rotational speed is equal to or higher than ωe1.

ここで、電動機3の動作温度が所定値よりも高い状態から低い状態に変化した場合、動作温度が高い場合の電流位相の設定(回転速度ωに対して電流位相がβs0〜βe0となる設定)から、動作温度が低い場合の電流位相の設定(回転速度ωに対して電流位相がβ
s1〜βe1となる設定)と等しくなるまで、所定時間毎に電流位相の変化量Δβだけ増加させ続ける(逆の場合には、所定時間毎に電流位相の変化量Δβだけ減少させ続ける)。
Here, when the operating temperature of the electric motor 3 is changed from a state higher than a predetermined value to a low state, the current phase is set when the operating temperature is high (setting where the current phase is βs0 to βe0 with respect to the rotational speed ω). To set the current phase when the operating temperature is low (the current phase is β
The current phase change amount Δβ is continuously increased every predetermined time until it becomes equal to (s1 to βe1) (in the opposite case, the current phase change amount Δβ is continuously decreased every predetermined time).

このように、本発明の電動機駆動装置は、電動機3の動作温度が低い場合に、予め設定された誘起電圧に対する電動機電流の位相差(電流位相)を大きくすることで、電動機の過渡特性(立ち上げ性能)を向上させることができる。   As described above, when the operating temperature of the electric motor 3 is low, the electric motor driving device of the present invention increases the phase difference (current phase) of the electric motor current with respect to the preset induced voltage, so that the transient characteristics of the electric motor (standing Increase performance).

なお、図3における電流位相の設定は、電動機3の動作温度が所定値よりも高い場合と低い場合の2通りで記載しているが、特に限定しているわけでなく、複数の閾値を持たせて、各々の温度領域における電流位相を設定しても良いことは言うまでもない。   In addition, although the setting of the current phase in FIG. 3 is described in two ways when the operating temperature of the electric motor 3 is higher than a predetermined value and when it is lower, it is not particularly limited and has a plurality of threshold values. Needless to say, the current phase in each temperature region may be set.

(実施の形態2)
図2は、本発明の実施の形態2における電動機駆動装置のシステム構成図を示すものである。図5に示す従来の電動機駆動装置と同じ構成要素は同一符号で示してあり、その説明は重複するため省略し、ここでは異なる部分についてのみ述べる。
(Embodiment 2)
FIG. 2 shows a system configuration diagram of an electric motor drive device according to Embodiment 2 of the present invention. The same components as those of the conventional electric motor drive device shown in FIG. 5 are denoted by the same reference numerals, and the description thereof is omitted because it is duplicated. Only different portions will be described here.

図2において、定数推定部16は、相電圧指令(vu*、vv*、vw*)、相電流検出値(iu、iv、iw)、磁極位置推定値θおよび回転速度推定値ωに基づいて、電動機3の巻線抵抗値もしくは誘起電圧定数の少なくともいずれか1つの電動機定数の温度変化を推定し、電動機定数推定値と予め設定された電動機定数基準値(例えば、20℃時における電動機定数の値)との定数比率を出力する。   In FIG. 2, the constant estimation unit 16 is based on a phase voltage command (vu *, vv *, vw *), a phase current detection value (iu, iv, iw), a magnetic pole position estimation value θ, and a rotation speed estimation value ω. The temperature change of at least one of the winding resistance value or the induced voltage constant of the motor 3 is estimated, and the estimated motor constant value and a preset motor constant reference value (for example, the motor constant at 20 ° C. Value) and a constant ratio.

温度検出部14は、定数比率から電動機3の動作温度を推定し、温度推定値を温度検出値として出力する。電流位相補正部13は、電動機3の動作温度検出値が所定値よりも小さい場合には予め設定された電流位相から値が大きくなるように電流位相を補正し、電流位相補正値として出力する。   The temperature detector 14 estimates the operating temperature of the electric motor 3 from the constant ratio, and outputs the estimated temperature value as the detected temperature value. When the detected operating temperature value of the electric motor 3 is smaller than a predetermined value, the current phase correction unit 13 corrects the current phase so that the value becomes larger than a preset current phase, and outputs the current phase correction value.

正弦波駆動部9は、電流位相補正部13からの出力された電流位相補正値に基づいて、式(2)、式(3)および式(23)〜式(28)の演算を行う。   Based on the current phase correction value output from the current phase correction unit 13, the sine wave driving unit 9 performs calculations of Expression (2), Expression (3), and Expression (23) to Expression (28).

次に、定数推定部16の具体的な動作について、一例として電動機の巻線抵抗値の温度変化による推定値を導出する場合について説明する。   Next, a specific operation of the constant estimating unit 16 will be described as an example in which an estimated value based on a temperature change of the winding resistance value of the motor is derived.

はじめに、電動機の巻線抵抗値をR、d軸インダクタンスをLd、q軸インダクタンスをLq、誘起電圧定数をKeとすると、定常状態におけるdq軸上での電圧方程式は式(30)、式(31)のように表される。   First, assuming that the winding resistance value of the motor is R, the d-axis inductance is Ld, the q-axis inductance is Lq, and the induced voltage constant is Ke, the voltage equation on the dq axis in the steady state is expressed by Equations (30) and (31 ).

Vd=R・id−ωLq・iq …(30)
Vq=R・iq+ωLd・id+ωKe …(31)
ここで、巻線抵抗値Rの推定値をRsとすると、式(30)は式(32)のように式変形できる。
Vd = R · id−ωLq · iq (30)
Vq = R · iq + ωLd · id + ωKe (31)
Here, when the estimated value of the winding resistance value R is Rs, the equation (30) can be transformed into the equation (32).

(Rs−R)・id・id=Rs・id・id
−(Vd・id+ωLq・id・iq) …(32)
すなわち、巻線抵抗値の推定値Rsが実際の巻線抵抗値Rよりも大きい場合には、右辺は正となり、逆の場合には負となるため、式(33)のように巻線抵抗値の推定値Rsを導出する(idの2乗としていることで、idの符号の影響は考慮する必要が無くなる)。
(Rs-R) .id.id = Rs.id.id
− (Vd · id + ωLq · id · iq) (32)
That is, when the estimated value Rs of the winding resistance value is larger than the actual winding resistance value R, the right side is positive, and in the opposite case, it is negative. Therefore, the winding resistance is as shown in Expression (33). An estimated value Rs of the value is derived (because the id is squared, it is not necessary to consider the influence of the sign of id).

Rs[nT]=Rs[(n−1)T]
−Kr・∫{Rs[(n−1)T]・id・id
−(Vd・id+ωLq・id・iq)}dt
…(33)
ここで、式(33)はマイコンなどで演算を行うため離散時間系を取っており、nTは現在のサンプリング時間、(n−1)Tは1時刻前のサンプリング時間である。また、Krは積分ゲインである。
Rs [nT] = Rs [(n−1) T]
−Kr · ∫ {Rs [(n−1) T] · id · id
− (Vd · id + ωLq · id · iq)} dt
... (33)
Here, the equation (33) takes a discrete time system in order to perform calculation by a microcomputer or the like, where nT is the current sampling time, and (n−1) T is the sampling time one time before. Kr is an integral gain.

なお、式(33)では、式(32)の右辺の積分演算のみ行っているが、比例項を付加してPI補償を行っても良いことは言うまでもない。   In Expression (33), only the integral operation on the right side of Expression (32) is performed, but it goes without saying that PI compensation may be performed by adding a proportional term.

さらに、idの符号が変化しない場合は、式(32)を辺々idで除算することができるため、式(33)において演算時間の短縮を図ることが可能である。   Furthermore, when the sign of id does not change, the equation (32) can be divided by ids from side to side, so that the calculation time can be reduced in the equation (33).

本発明の定数推定部16では、このようにして導出した巻線抵抗値の推定値Rsと、予め設定された巻線抵抗の基準値R0(例えば、20℃時の巻線抵抗値)との比率(=Rs/R0)を出力する。   In the constant estimation unit 16 of the present invention, the estimated value Rs of the winding resistance value derived as described above and a preset reference value R0 (for example, the winding resistance value at 20 ° C.) of the winding resistance are set. The ratio (= Rs / R0) is output.

次に、温度検出部14では、定数推定部16から出力される定数の比率に基づいて、巻線温度を推定する。ここで、一般的に電動機の巻線は銅線が用いられており、銅の温度係数を考慮すると、巻線温度推定値Tsは式(34)のように表される。   Next, the temperature detection unit 14 estimates the winding temperature based on the ratio of the constants output from the constant estimation unit 16. Here, a copper wire is generally used for the winding of the electric motor, and the winding temperature estimated value Ts is expressed as in Equation (34) in consideration of the temperature coefficient of copper.

Ts=Rs/R0・(234.5+20)−234.5 …(34)
なお、巻線抵抗値の推定値Rsの導出方法について説明したが、誘起電圧Keに関しては、巻線抵抗値の推定値Rsを導出した上で、式(31)に基づいて温度変化を推定することができる。ただし、この場合は、磁石の温度係数を考慮して、磁石温度推定値を求める必要がある。
Ts = Rs / R0 · (234.5 + 20) −234.5 (34)
Although the method for deriving the estimated value Rs of the winding resistance value has been described, with respect to the induced voltage Ke, the estimated value Rs of the winding resistance value is derived and the temperature change is estimated based on the equation (31). be able to. However, in this case, it is necessary to obtain the estimated magnet temperature value in consideration of the temperature coefficient of the magnet.

このようにして、電動機の巻線温度もしくは磁石温度を推定することができ、サーミスタや熱電対等の、温度を検出する温度センサが不要となるため、コスト低減と信頼性向上とを図ることができる。   In this way, the winding temperature or magnet temperature of the electric motor can be estimated, and a temperature sensor such as a thermistor or a thermocouple that detects the temperature is not necessary, so that cost reduction and reliability improvement can be achieved. .

また、実施の形態1および2において、2つの電流検出器6a、6bにより電動機3の電流を検出しているが、インバータ4の直流電流、すなわち、整流回路2とインバータ4との間に流れる電流から電動機3の電流を検出するなどの手段を用いても良い。   In the first and second embodiments, the current of the motor 3 is detected by the two current detectors 6a and 6b, but the DC current of the inverter 4, that is, the current flowing between the rectifier circuit 2 and the inverter 4 For example, a means for detecting the current of the motor 3 may be used.

さらに、実施の形態1および2において、正弦波駆動部9へ与えられる目標回転速度ω*に電動機3の回転速度ωが追従するように、回転速度制御を行うものとしたが、例えば、電動機3のトルクを制御するようにしても同様の効果が得られる。   Further, in the first and second embodiments, the rotation speed control is performed so that the rotation speed ω of the electric motor 3 follows the target rotation speed ω * given to the sine wave driving unit 9. Even if the torque is controlled, the same effect can be obtained.

加えて、実施の形態1および2において、整流回路12を用いているが、力率改善型昇圧コンバータなどを用いても良い。   In addition, although the rectifier circuit 12 is used in the first and second embodiments, a power factor improving type boost converter or the like may be used.

以上のように、本発明の電動機駆動装置によれば、電動機の動作温度が低い場合に予め設定された誘起電圧に対する電動機電流の位相差(電流位相)を大きくすることで、電動機の過渡特性(立ち上げ性能)を向上させることができる。従って、本発明の電動機駆動装置は、ルームエアコンなどの圧縮機の電動機駆動装置に応用することができる。   As described above, according to the motor drive device of the present invention, when the operating temperature of the motor is low, the phase difference (current phase) of the motor current with respect to the preset induced voltage is increased, whereby the transient characteristics ( (Startup performance) can be improved. Therefore, the electric motor drive device of the present invention can be applied to an electric motor drive device of a compressor such as a room air conditioner.

本発明の実施の形態1による電動機駆動装置のシステム構成図1 is a system configuration diagram of an electric motor drive device according to Embodiment 1 of the present invention. 本発明の実施の形態2による電動機駆動装置のシステム構成図The system block diagram of the electric motor drive device by Embodiment 2 of this invention 本発明の電動機駆動装置における電流位相の一例を示す説明図Explanatory drawing which shows an example of the electric current phase in the electric motor drive device of this invention 予め設定された電流位相の一例を示す説明図Explanatory diagram showing an example of preset current phase 従来の電動機駆動装置のシステム構成図System configuration diagram of a conventional motor drive device

符号の説明Explanation of symbols

1 交流電源
2 整流回路
3 電動機
4 インバータ
4a〜4f スイッチング素子
5a〜5f 還流ダイオード
6a、6b 電流検出器
7 インバータ制御部
8 電流検出部
9 正弦波駆動部
10 PWM信号生成部
11 誘起電圧推定部
12 位置速度推定部
13 電流位相補正部
14 温度検出部
15 温度検出器
16 定数推定部
DESCRIPTION OF SYMBOLS 1 AC power supply 2 Rectifier circuit 3 Electric motor 4 Inverter 4a-4f Switching element 5a-5f Free-wheeling diode 6a, 6b Current detector 7 Inverter control part 8 Current detection part 9 Sine wave drive part 10 PWM signal generation part 11 Induced voltage estimation part 12 Position speed estimation unit 13 Current phase correction unit 14 Temperature detection unit 15 Temperature detector 16 Constant estimation unit

Claims (3)

高圧側に配置された上アームスイッチング素子と低圧側に配置された下アームスイッチング素子からなるスイッチング素子対を複数有し、各スイッチング素子の動作により直流電圧を所望の周波数、電圧の交流電圧に変換し、複数相の電動機にその駆動電圧として供給するインバータと、前記電動機の誘起電圧に対して所定の位相差の電動機電流が流れるように前記インバータに駆動信号を与えるインバータ制御手段と、前記電動機の巻線温度、回転子磁石温度、雰囲気温度の少なくともいずれか1つの温度を検出する温度検出手段とを備え、前記温度検出手段によって検出された温度検出値が所定値よりも小さい場合には予め設定された前記位相差を大きくすることを特徴とする電動機駆動装置。 It has multiple switching element pairs consisting of upper arm switching elements arranged on the high voltage side and lower arm switching elements arranged on the low voltage side, and the DC voltage is converted to the AC voltage of the desired frequency and voltage by the operation of each switching element. And an inverter for supplying a driving signal to the inverter so that a motor current having a predetermined phase difference flows with respect to the induced voltage of the electric motor, Temperature detecting means for detecting at least one of winding temperature, rotor magnet temperature, and ambient temperature, and preset if the temperature detection value detected by the temperature detecting means is smaller than a predetermined value. An electric motor drive device characterized by increasing the phase difference. 前記温度検出手段は、前記電動機に印加される電動機電圧、前記電動機の固定子巻線に流れる電動機電流、前記電動機の回転子の磁極位置および回転速度に基づいて、前記電動機の巻線抵抗値もしくは誘起電圧定数の少なくともいずれか1つの電動機定数の温度変化を推定する定数推定手段を備え、前記定数推定手段によって推定された電動機定数推定値と予め設定された電動機定数基準値との比率から、前記電動機の巻線温度もしくは回転子磁石温度の少なくともいずれか1つを算出することを特徴とする請求項1に記載の電動機駆動装置。 The temperature detecting means is based on the motor voltage applied to the motor, the motor current flowing in the stator winding of the motor, the magnetic pole position and the rotation speed of the rotor of the motor, or the winding resistance value of the motor or Constant estimation means for estimating a temperature change of at least one of the induced voltage constants of the motor constant is provided, and from the ratio between the motor constant estimated value estimated by the constant estimation means and a preset motor constant reference value, The motor drive device according to claim 1, wherein at least one of a winding temperature and a rotor magnet temperature of the motor is calculated. 前記インバータ制御手段は、前記電動機の固定子巻線に流れる電動機電流を検出する電流検出手段と、前記電動機に印加される電動機電圧と前記電動機電流とから前記電動機に発生する誘起電圧を推定する誘起電圧推定手段と、前記誘起電圧に基づいて、前記電動機の回転子の磁極位置および回転速度を推定する位置速度推定手段とを備えることを特徴とする請求項1に記載の電動機駆動装置。 The inverter control means includes: current detection means for detecting a motor current flowing in a stator winding of the motor; and an induction for estimating an induced voltage generated in the motor from the motor voltage applied to the motor and the motor current. The motor drive apparatus according to claim 1, further comprising: a voltage estimation unit; and a position / speed estimation unit that estimates a magnetic pole position and a rotation speed of a rotor of the motor based on the induced voltage.
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