WO2023098218A1 - 一种宽范围谐振式软开关双向直流变换器及其控制方法 - Google Patents

一种宽范围谐振式软开关双向直流变换器及其控制方法 Download PDF

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WO2023098218A1
WO2023098218A1 PCT/CN2022/117907 CN2022117907W WO2023098218A1 WO 2023098218 A1 WO2023098218 A1 WO 2023098218A1 CN 2022117907 W CN2022117907 W CN 2022117907W WO 2023098218 A1 WO2023098218 A1 WO 2023098218A1
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Prior art keywords
conversion unit
switch tube
power supply
resonant
bridge
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PCT/CN2022/117907
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English (en)
French (fr)
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刘斌
李玲
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刘三英
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Publication of WO2023098218A1 publication Critical patent/WO2023098218A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/38—Means for preventing simultaneous conduction of switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the technical field of DC converters, in particular to a wide-range resonant soft-switching bidirectional DC converter and a control method thereof.
  • FIG. 1 there are usually two ways to make a conversion circuit for a low-voltage battery pack: one is to use two stages, usually through a step-up or step-down scheme first, and then through a DC/DC regulator transform. The cost of the two-stage scheme is higher, and the efficiency of the two-stage transformation will decrease at the same time.
  • the other is to use a switch to change the turns ratio of the transformer, by changing the turns ratio of the transformer, or using a similar circuit to increase or decrease the transformer coil, the implementation method mentioned in the patent of the patent authorization number CN107733236B, as shown in Figure 2, Its essence is to increase and decrease through the additional transformer conversion circuit, so as to achieve different voltage ratios.
  • the control principle is simple and direct, but the change of high turn ratio will cause higher switch tube stress, and will also change the original main transformer. Inductance, leakage inductance parameters, the introduction of new current loop interference, including sudden voltage mutations may bring about another series of control parameter changes, and step-wise duty cycle adjustments are prone to oscillations and other problems.
  • the soft-switching coordination condition of the two converters is relatively poorly realizable; therefore, additional conversion circuits and transformers must be added, and the structure of the entire converter is complex and difficult to popularize and apply.
  • the object of the present invention is to provide a wide-range resonant type soft-switching bidirectional DC converter and its control method, which can not only realize soft-switching high-efficiency conversion, but also can be relatively simple and meet the bidirectional conversion of a wide range of voltages, so as to solve the problems existing in the prior art.
  • the technical problems that require two-stage converters to perform multiple conversions, many diversion path devices, and soft switching that cannot realize full conversion lead to large losses, making it unsuitable for applications in places with limited volume or relatively high cost requirements.
  • the technical solution adopted by the present invention is: a wide-range resonant soft-switching bidirectional DC converter, including a first DC power supply, an input energy storage filter capacitor, a primary bridge conversion unit, a series resonance unit, an isolation transformer, a secondary conversion unit, a resonant buffer unit, an output energy storage filter unit, and a second DC power supply;
  • the input energy storage filter capacitor is connected in parallel with the primary bridge conversion unit, and the primary bridge conversion unit is also connected to the first DC power supply connection;
  • the primary side of the isolation transformer is connected in series with the series resonant unit, and then connected to the primary bridge conversion unit, and the secondary side of the isolation transformer is connected to the secondary conversion unit;
  • the resonance buffer unit is connected to the The secondary conversion unit is connected in parallel, the output energy storage filter unit is connected in parallel with the resonance buffer unit, and the second DC power supply is connected to the output energy storage filter unit;
  • the primary bridge conversion unit is a full-bridge conversion unit or a half-bridge conversion unit;
  • the secondary conversion unit is a full-bridge conversion unit or a full-wave rectifier converter;
  • the series resonant unit includes a resonant capacitor connected in series and a resonant inductance;
  • the resonant capacitor is connected to the primary bridge conversion unit, and the resonant inductance is connected to the primary side of the isolation transformer;
  • the resonant buffer unit includes a buffer switch tube and a buffer capacitor connected in series;
  • the energy storage filter unit includes an energy storage inductor and an output energy storage filter capacitor connected in series;
  • the second DC power supply is connected to the output energy storage filter capacitor;
  • the primary bridge conversion unit When the primary bridge conversion unit is a full bridge conversion unit, the primary bridge conversion unit includes a first switch tube, a second switch tube, a third switch tube and a fourth switch tube; the first switch tube and the second switch tube Three switch tubes are connected in series to form a first bridge arm, the second switch tube and the fourth switch tube are connected in series to form a second bridge arm, and the first bridge arm and the second bridge arm are connected in parallel; the first switch tube and the second switch tube are connected in parallel.
  • the drains of the second switch tubes are connected to the positive pole of the first DC power supply and one end of the input energy storage filter capacitor, and the sources of the third switch tube and the fourth switch tube are connected to the first DC power supply.
  • the negative pole is connected to the other end of the input energy storage filter capacitor; the resonant capacitor is connected to the drain of the third switching tube, and the primary side of the isolation transformer is connected to the drain of the fourth switching tube; when the primary When the bridge conversion unit is a half-bridge conversion unit, the primary bridge conversion unit includes a first switch tube and a second switch tube connected in series, and the drain of the first switch tube is connected to the first DC power supply; The positive pole of the switch tube is connected to one end of the input energy storage filter capacitor, the source of the second switch tube is connected to the negative pole of the first DC power supply and the other end of the input energy storage filter capacitor; the resonant capacitor is connected to the The drain of the second switching tube is connected, and the primary side of the isolation transformer is connected to the source of the second switching tube;
  • the secondary conversion unit When the secondary conversion unit is a full-bridge conversion unit, the secondary conversion unit includes a fifth switching tube, a sixth switching tube, a seventh switching tube, and an eighth switching tube; the fifth switching tube and the seventh switching tube
  • the tubes are connected in series to form a third bridge arm, the sixth switch tube and the eighth switch tube are connected in series to form a fourth bridge arm, and the third bridge arm and the fourth bridge arm are connected in parallel; the secondary side of the isolation transformer is connected to the The drain connection of the seventh switch tube and the eighth switch tube;
  • the secondary conversion unit when the secondary conversion unit is a full-wave rectifier converter, the secondary conversion unit includes a fifth switch tube and a sixth switch tube; the fifth switch After the drain of the tube is connected to the drain of the sixth switch tube, it is then connected to the source of the buffer switch tube and one end of the energy storage inductance; the sources of the fifth switch tube and the sixth switch tube are isolated from the The secondary side of the transformer is connected, and the secondary side of the isolation transformer is also connected to one end of
  • first DC power supply and the second DC power supply are DC power supplies, rectified AC power supplies, step power supplies with switch control, or loads that can provide power supply voltages.
  • the first to eighth switching tubes may be diodes, or high-frequency switching tubes provided with anti-parallel diodes , the anti-parallel diode is an integrated diode, a parasitic diode or an external diode.
  • the input energy storage filter capacitor and the output energy storage filter capacitor are non-polar capacitors or polar capacitors; when the first DC power supply or the second DC power supply is a step-change power supply, the The input energy storage filter capacitor and the output energy storage filter capacitor are equivalent capacitors in series with a controllable switch and a capacitor; the resonant inductance is an external inductance, a coupling leakage inductance inside a transformer, or a coupling between an external inductance and an internal leakage inductance of a transformer inductance.
  • the power state setting circuit sampling or the external communication detection DC circuit device needs to output the voltage to determine whether the working state of the DC converter is the forward working state or the reverse working state;
  • the forward working state refers to the first DC The power supply is input, and the second DC power supply is output;
  • the reverse working state means that the second DC power supply is input, and the first DC power supply is output;
  • S200 Determine whether the working state of the primary bridge conversion unit and the secondary conversion unit is the inverter state or the rectification state, determine whether the resonance buffer unit is in the rectification buffer resonance state or the inverter resonance state; and perform corresponding sequential logic configuration and PWM drive configuration ;
  • the maximum duty cycle of the switching tube in the primary bridge conversion unit and the secondary conversion unit is not more than 0.5, and there is enough dead time;
  • the frequency of the driving signal applied to the buffer switch tube Q9 is the primary side bridge conversion unit or
  • the frequency of the switching tube driving signal in the secondary conversion unit is twice that of the primary side bridge conversion unit and the secondary conversion unit has the same operating frequency;
  • the operating frequencies of the PWM driving signals of the primary bridge conversion unit and the switching tubes of the secondary bridge converter are the same, and the frequency range is 95%-115% of the natural resonant frequency.
  • steps S300-S500 when the DC converter is working in the forward working state, if the PWM drive applied to the primary bridge conversion unit increases the duty cycle to the maximum limit value, it still cannot reach the second DC power supply voltage value, then fix the duty cycle, adjust the operating frequency to the optimum operating frequency point to enter the boost mode, and rectify and conduct the bridge arm of the secondary conversion unit before the next rectification conduction period.
  • a switching tube increases the PWM drive, otherwise it gradually reduces the PWM drive duty cycle applied to the primary bridge conversion unit and exits the boost mode; when the DC converter is working and the DC converter is working in the reverse working state, if the If the PWM drive applied by the secondary conversion unit increases the duty cycle to the maximum limit value and still cannot meet the requirement of the first DC power supply voltage value, then the non-current cycle of the primary bridge conversion unit is performed before the next rectification conduction cycle is about to start.
  • the switching tube of the internal rectification conduction bridge arm increases the PWM drive to boost the voltage, otherwise, the duty cycle of the PWM drive applied to the primary bridge conversion unit is gradually reduced according to the control and exits the boost mode.
  • steps S300-S500 when the DC converter is working in the forward working state, if the secondary conversion unit is a full-bridge conversion unit, only for the off-cycle period immediately before the next rectification conduction period PWM driving is applied to one of the switching tubes of the rectifying and conducting bridge arm, or PWM driving is applied to both switching tubes of the rectifying and conducting bridge arm in the off-cycle period; if the secondary conversion unit is a full-wave rectifying converter, then in the next Before the rectification conduction period is about to start, only PWM drive is applied to the non-rectification conduction switch tube in this period; when the DC converter is working in the reverse working state, if it is in boost mode, and the primary bridge conversion unit is a full bridge When converting the unit, only apply PWM drive to one of the switching tubes of the rectifying and conducting bridge arm in the off-cycle period before the next rectifying and conducting period, or apply PWM drive to the two switching tubes of the rectifying and conducting bridge arm in the off-cycle period Both increase the PWM drive;
  • steps S300-S500 by adjusting the duty ratio of the buffer switch tube, the adjustment of the output voltage in a certain range in the corresponding conversion mode and the soft switching state of the secondary conversion unit are realized; when the DC converter is working in the forward working state , the PWM drive applied to the snubber switch is delayed from the PWM drive of the primary bridge conversion unit, that is, there is a certain turn-on dead zone, and at the same time, the PWM drive applied to the buffer switch and the PWM drive of the primary bridge conversion unit are closed.
  • the PWM drive applied to the snubber switch is delayed from the PWM drive of the secondary conversion unit, that is, there is a certain turn-on dead zone, and the PWM drive applied to the snubber switch is turned off
  • the final minimum dead time is consistent with the minimum dead time of the secondary conversion unit PWM drive; if the DC converter works in the boost mode of the reverse working state, the buffer switch tube must not be earlier than the primary bridge conversion unit. The switch tube connected to the terminal is turned off.
  • the driving signal applied to the switching tube for boosting work of the primary bridge conversion unit is earlier than that of the secondary conversion unit.
  • the driving signal of the unit, applied to the switching tube driving signal in the primary bridge conversion unit that acts as a boost, is the delayed signal of the synchronous rectification signal in the previous period, that is, the period of the delayed signal is the synchronous rectification duty cycle, the boost duty cycle.
  • the traditional series resonant conversion needs wide-range frequency modulation to realize the voltage control mode.
  • the present invention mainly realizes voltage regulation by adjusting the duty cycle of each conversion unit switch tube, which is different from the traditional bridge converter.
  • the pressure control principle is close and relatively simple;
  • the input DC power supply can be a stepwise power supply controlled by a switch.
  • FIG. 1 is a schematic block diagram of an existing DC converter
  • Fig. 2 is the schematic circuit diagram of the existing bidirectional DC conversion implementation scheme
  • Fig. 3 is a schematic block diagram of an embodiment of the present invention.
  • Fig. 4 is the schematic circuit diagram of the embodiment of the present invention.
  • Fig. 5 is a schematic diagram of the specific implementation of the connection of the primary bridge conversion unit according to the embodiment of the present invention.
  • FIG. 6 is a schematic diagram of a specific implementation of the connection of secondary conversion units according to an embodiment of the present invention.
  • Fig. 7 is a circuit diagram of an embodiment of the present invention in a forward rectification working state
  • FIG. 8 is a circuit diagram of an embodiment of the present invention in a reverse rectification working state
  • FIG. 9 is a schematic waveform diagram of an embodiment of the present invention in a forward rectification working state
  • FIG. 10 is a schematic waveform diagram of an embodiment of the present invention in a reverse rectification working state.
  • a wide-range resonant soft-switching bidirectional DC converter includes a first DC power supply DC1, an input energy storage filter capacitor C1, a primary bridge conversion unit, a series resonance unit, and an isolation transformer Tra , a secondary conversion unit, a resonant buffer unit, an output energy storage filter unit and a second DC power supply DC2; the input energy storage filter capacitor C1 is connected in parallel with the primary bridge conversion unit, and the primary bridge conversion unit is also connected with the The first DC power supply DC1 is connected; the primary side of the isolation transformer Tra is connected in series with the series resonant unit, and then connected to the primary bridge conversion unit, and the secondary side of the isolation transformer Tra is connected to the secondary conversion unit.
  • the unit is connected; the resonance buffer unit is connected in parallel with the secondary conversion unit, the output energy storage filter unit is connected in parallel with the resonance buffer unit, and the second DC power supply DC2 is connected with the output energy storage filter unit;
  • the primary bridge conversion unit is a full-bridge conversion unit or a half-bridge conversion unit;
  • the secondary conversion unit is a full-bridge conversion unit or a full-wave rectifier converter;
  • the series resonant unit includes a resonant capacitor connected in series Cr and a resonant inductance Lr;
  • the resonant capacitor Cr is connected to the primary bridge conversion unit, and the resonant inductance Lr is connected to the primary side of the isolation transformer Tra;
  • the resonant snubber unit includes a series connection snubber switch tube Q9 and buffer capacitor Cs;
  • the output energy storage filter unit includes an energy storage inductor L1 and an output energy storage filter capacitor C2 connected in series;
  • the second DC power supply DC2 is connected to the output energy storage filter capacitor C2;
  • the primary bridge conversion unit When the primary bridge conversion unit is a full bridge conversion unit, the primary bridge conversion unit includes a first switching tube Q1, a second switching tube Q2, a third switching tube Q3 and a fourth switching tube Q4;
  • the switch tube Q1 and the third switch tube Q3 are connected in series to form a first bridge arm, the second switch tube Q2 and the fourth switch tube Q4 are connected in series to form a second bridge arm, and the first bridge arm and the second bridge arm are connected in parallel;
  • the drains of the first switching tube Q1 and the second switching tube Q2 are connected to the anode of the first DC power supply DC1 and one end of the input energy storage filter capacitor C1, and the third switching tube Q3 and the fourth switching tube
  • the source of the tube Q4 is connected to the negative pole of the first DC power supply DC1 and the other end of the input energy storage filter capacitor C1;
  • the resonant capacitor Cr is connected to the drain of the third switching tube Q3, and the isolation
  • the primary side of the transformer Tra is connected to the drain
  • the drain of the first switching tube Q1 is connected to the anode of the first DC power supply DC1 and one end of the input energy storage filter capacitor C1, the source of the second switching tube Q2 is connected to The negative pole of the first DC power supply DC1 is connected to the other end of the input energy storage filter capacitor C1; the resonant capacitor Cr is connected to the drain of the second switching tube Q2, and the primary side of the isolation transformer Tra is connected to the Describe the source connection of the second switching tube Q2;
  • the secondary conversion unit When the secondary conversion unit is a full-bridge conversion unit, the secondary conversion unit includes a fifth switching tube Q5, a sixth switching tube Q6, a seventh switching tube Q7, and an eighth switching tube Q8; the fifth switching tube Q5 and the seventh switch tube Q7 are connected in series to form a third bridge arm, the sixth switch tube Q6 and the eighth switch tube Q8 are connected in series to form a fourth bridge arm, and the third bridge arm and the fourth bridge arm are connected in parallel;
  • the secondary side of the isolation transformer Tra is connected to the drains of the seventh switch tube Q7 and the eighth switch tube Q8;
  • the secondary conversion unit when the secondary conversion unit is a full-wave rectifier converter, the secondary conversion unit includes a fifth switch tube Q5 and the sixth switching tube Q6; the drain of the fifth switching tube Q5 is connected to the drain of the sixth switching tube Q6, and then connected to the source of the buffer switching tube Q9 and one end of the energy storage inductor L1;
  • the first DC power supply DC1 and the second DC power supply DC2 are DC power supplies, rectified AC power supplies, step power supplies with switch control or loads that can provide power supply voltages.
  • the rectified three-phase AC is input as a DC source, the voltage of each phase is different.
  • the series switch is used for control switching, the rectified three-phase AC will be a step input power supply.
  • the embodiment of the present invention is also applicable to Switch controlled step power supply.
  • the input energy storage filter capacitor C1 and the output energy storage filter capacitor C2 are non-polar capacitors or polar capacitors; when the first DC power supply DC1 or the second DC power supply DC2 is a step-change power supply, The input energy storage filter capacitor C1 and the output energy storage filter capacitor C2 are equivalent capacitors in series with a controllable switch and a capacitor; the resonant inductance Lr is an external inductor, a coupling leakage inductance inside a transformer, or an external inductor and an internal transformer Leakage inductance of coupled inductance.
  • the first to eighth switching tubes Q8 may be diodes, or high-frequency switching tubes provided with anti-parallel diodes, so The aforementioned anti-parallel diodes are integrated diodes, parasitic diodes or external diodes.
  • the primary bridge conversion unit can be either a full bridge conversion unit or a half bridge conversion unit.
  • Figure 5(a) is the circuit diagram of the full-bridge conversion unit.
  • the full-bridge converter is composed of the switch tube A QA, the switch tube B QB, the switch tube C QC and the switch tube D QD.
  • the resonant capacitor Cr and the resonant inductor Lr together form a series connection Resonant unit
  • Figure 5(b) is a connection mode of the half-bridge conversion unit, using the switch tube A QA and the switch tube C QC to form the bridge arm, and the resonant capacitor Cr and the resonant inductance Lr together form a series resonant unit
  • the first The first resonant capacitor Cr1, the second resonant capacitor Cr2 and the resonant inductor Lr together form a series resonant unit.
  • the series relationship between the series resonant unit and the transformer coil, the connection sequence of the resonant capacitor Cr and the resonant inductance Lr in the series loop can all be adjusted.
  • the secondary transformation unit can be either a full-bridge transformation unit or a full-wave rectification converter.
  • Figure 6(a) is the circuit diagram of the full-bridge conversion unit, which uses switch tube A QA, switch tube B QB, switch tube C QC and switch tube D QD to form a full-bridge converter;
  • Figure 6(b) and Figure 6(c ) are two different connection modes of the full-wave rectifier converter, and the full-wave rectifier converter is also called a push-pull converter; in Fig. When it is used as a rectifier, it is called a full-wave rectifier;
  • the switch tube A QA and switch tube B QB in Figure (c) adopt the common source connection method, and the circuit function is the same as that shown in Figure 6(b).
  • FIG. 5 and FIG. 6 are well-known circuits, and those skilled in the art should understand their specific working principles, and will not be further analyzed in this article.
  • the present invention is not limited to the above implementation examples, and other combinations that can realize the functions of the present invention also belong to this category.
  • the power state setting circuit sampling or the external communication detection DC circuit device needs to output the voltage to determine whether the working state of the DC converter is the forward working state or the reverse working state;
  • the forward working state refers to the first DC
  • the second DC power supply DC2 is the output;
  • the reverse working state means that the second DC power supply DC2 is the input, and the first DC power supply DC1 is the output;
  • S200 Determine whether the working state of the primary bridge conversion unit and the secondary conversion unit is the inverter state or the rectification state, determine whether the resonance buffer unit is in the rectification buffer resonance state or the inverter resonance state; and perform corresponding sequential logic configuration and PWM drive configuration ;
  • the maximum duty cycle of the switching tube in the primary bridge conversion unit and the secondary conversion unit is not more than 0.5, and there is enough dead time;
  • the frequency of the driving signal applied to the buffer switch tube Q9 is the primary side bridge conversion unit or
  • the frequency of the switching tube driving signal in the secondary conversion unit is twice that of the primary side bridge conversion unit and the secondary conversion unit has the same operating frequency;
  • the stage conversion unit performs high-frequency pulse conversion, which is transmitted from the secondary side to the primary side through the isolation transformer Tra coupling, and then transmitted to the primary bridge conversion unit through the series resonant unit for high-frequency rectification conversion, and then the DC voltage is transmitted to the input filter capacitor and a first direct current power supply DC1;
  • the PWM drive applied to the secondary conversion unit will be Decrease the duty cycle adjustment, otherwise, adjust the duty cycle to increase;
  • the operating frequencies of the PWM driving signals of the primary bridge conversion unit and the switching tubes of the secondary bridge converter are the same, and the frequency range is 95%-115% of the resonant natural resonant frequency.
  • the operating frequency of the PWM driving signal of the switching tubes of the primary bridge conversion unit and the secondary conversion unit is 105% of the natural resonance frequency of the resonance.
  • the duty cycle is fixed. , adjust the operating frequency to the optimum operating frequency point to enter the boost mode, and increase the PWM drive to one of the switching tubes of the rectification and conduction bridge arm of the secondary conversion unit in the non-period of the rectification and conduction period before the next rectification and conduction period, and vice versa Gradually reduce the PWM driving duty cycle applied to the primary bridge conversion unit and exit the boost mode; when the DC converter is working and the DC converter is working in the reverse working state, if the PWM driving applied to the secondary If the duty ratio reaches the maximum limit value and still cannot meet the requirement of the voltage value of the first DC power supply DC1, the switch of the rectification and conduction bridge arm in the non-period period of the primary bridge conversion unit is performed before the next rectification conduction period is about to
  • the secondary conversion unit When the DC converter works in the forward working state, if the secondary conversion unit is a full-bridge conversion unit, only one of the switch tubes of the bridge arm is rectified and conducted in the off-cycle period before the next rectification conduction period is about to start. Apply PWM drive, or apply PWM drive to the two switching tubes of the rectification and conduction bridge arm in the off-cycle period; if the secondary conversion unit is a full-wave rectification converter, only this period will be used before the next rectification and conduction period.
  • PWM drive is applied to the internal non-rectification conduction switching tube; when the DC converter is working in the reverse working state, if it is in boost mode and the primary bridge conversion unit is a full bridge conversion unit, then the next rectification conduction Just before the cycle starts, only apply PWM drive to one of the switches of the rectifying and conducting bridge arm in the off-cycle period, or increase the PWM drive to both switches of the rectifying and conducting bridge arm in the off-cycle period; if the primary bridge transforms If the unit is a half-bridge conversion unit, then the PWM drive is only applied to the non-rectified conduction switch tube in this period before the next rectification conduction period is about to start.
  • the adjustment of the output voltage in a certain range in the corresponding conversion mode and the soft switching state of the secondary conversion unit are realized; when the DC converter works in the forward working state, the voltage applied to the buffer switch tube Q9 The PWM drive is delayed from the PWM drive of the primary bridge conversion unit, that is, there is a certain turn-on dead zone, and at the same time, the PWM drive applied to the buffer switch Q9 at the closing time is consistent with the PWM drive of the primary bridge conversion unit; when the DC converter is working In the reverse working state, the PWM drive applied to the buffer switch tube Q9 is delayed from the PWM drive of the secondary conversion unit, that is, there is a certain turn-on dead zone, and the minimum dead zone after the PWM drive applied to the buffer switch tube Q9 is turned off The time is consistent with the minimum dead time of the PWM drive of the secondary conversion unit; if the DC converter works in the boost mode of the reverse working state, the buffer switch tube Q9 must not be earlier than that connected to the
  • the driving signal applied to the switching tube of the primary bridge conversion unit for boosting work is earlier than the driving signal of the secondary conversion unit, and is applied to the primary bridge conversion unit.
  • the driving signal of the switching tube that acts as a boost in the conversion unit is the delayed signal of the synchronous rectification signal in the previous period, that is, the period of the delayed signal is the sum of the synchronous rectification duty cycle, the boost duty cycle and the dead time; if the DC The converter works in a non-boost mode, and the switch tubes in the primary bridge conversion unit and the secondary conversion unit apply a synchronous rectification drive signal.
  • the operating frequency of the bridge converter on the primary side is set as the series resonance unit resonant frequency of
  • lr is the inductance value of the resonant inductor Lr
  • cr is the capacitance value of the resonant capacitor Cr.
  • the switching frequency of the snubber switch tube Q9 of the resonant snubber unit is 2f 0 .
  • the function of the secondary conversion unit is a full-bridge rectifier.
  • a synchronous rectification signal can be applied to the secondary conversion unit.
  • no driving signal can be added, and the fifth switching tube Q5 and the sixth switching tube Q6 are regarded as diode rectification, and FIG. 4 can be simplified into the circuit diagram shown in FIG. 7 .
  • the bridge conversion driving duty cycle applied to the first switching tube Q1, the second switching tube Q2, the third switching tube Q3 and the fourth switching tube Q4 is about 50%, preferably 45%.
  • the full-bridge converter is equivalent to the LLC full-bridge converter.
  • the first switching tube Q1, the second switching tube Q2, the third switching tube Q3 and the fourth switching tube Q4 all realize soft switching.
  • the PWM drive applied to the snubber switch Q9 is slightly delayed, and the initial current flows through the anti-parallel diode of the snubber switch Q9 to charge the snubber capacitor Cs, and then the snubber switch Q9 is driven and turned on, which constitutes zero-voltage turn-on.
  • the secondary rectified current gradually increases and takes on a sinusoidal shape, while the current on the output side changes linearly due to the existence of the energy storage inductance L1.
  • the current charging the buffer capacitor Cs at this time is the secondary rectified current I-rec-sec minus the energy storage inductor L1 current I-L1.
  • the secondary rectified current gradually decreases, and the output current gradually increases. Large, so the buffer capacitor Cs starts to store energy, and the energy storage inductor L1 discharges.
  • the switching tube of the primary bridge conversion unit When the switching tube of the primary bridge conversion unit is turned off, it can be approximately regarded as the voltage in front of the energy storage inductor L1 is about to disappear. If the snubber switch Q9 is turned off immediately or with a slight delay, it means that the energy storage inductor L1 needs to pass through the secondary The conversion unit performs freewheeling. Before freewheeling, the energy storage inductor L1 will draw a current equivalent to the parasitic capacitance of the secondary conversion unit and gradually reduce it to zero voltage. Therefore, the snubber switch Q9 can be regarded as a zero-voltage turn-off.
  • the isolation transformer Tra is always clamped as Zero, providing preparation for the next zero-voltage turn-on.
  • the snubber switch tube Q9 and the snubber capacitor Cs assist the soft turn-on or soft turn-off of the secondary conversion unit, and absorb and buffer the excess current of the primary bridge conversion unit, allowing the energy storage inductor L1 to work at
  • the state of the applied pulse voltage is similar to the step-down state, which solves the shortcomings of the original series resonant converter that can only be adjusted by frequency conversion and the adjustment range is not large, and the duty cycle is nonlinear.
  • the embodiment of the present invention not only obtains the advantages of soft switching conversion, but also realizes the advantages of simple control of the step-down converter.
  • the related waveform schematic diagram is shown in FIG. 9 .
  • the snubber switch tube Q9 needs to cooperate with the series resonance unit on the primary side to adjust the duty cycle, so as to realize stable voltage regulation and soft switching. If the PWM drive applied to the primary bridge conversion unit is increased to the maximum limit value and still cannot meet the voltage requirement of the second direct current power supply DC2, then the PWM drive applied to the primary bridge conversion unit is fixed to the maximum duty cycle, And adjust the operating frequency of the primary bridge conversion unit to the optimum operating frequency point to enter the boost mode, and rectify and conduct one of the switch tubes of the bridge arm in the off-cycle period of the secondary conversion unit before the next rectification conduction period is about to start. The PWM drive is added, and the output voltage is adjusted by adjusting the duty cycle of the PWM drive. If the voltage requirement of the second DC power supply DC2 can be met without boosting, the boost mode is exited.
  • the primary bridge conversion unit mainly works in the rectification mode. If according to the calculation, the voltage of the second DC power supply DC2 is reversely converted through the turns ratio of the isolation transformer Tra, and there is no need to enter the boost mode, then Figure 4 can be simplified as shown in Figure 8, and the relevant principles are well known to those skilled in the art common sense. For ease of discussion, assuming that the embodiment of the present invention needs to enter the boost mode, even if it is only reverse rectification, it must be a converter with a bridge switch.
  • the driving duty cycle of the fifth switching tube Q5, the sixth switching tube Q6, the seventh switching tube Q7 and the eighth switching tube Q8 in the secondary conversion unit is applied to the maximum, and the non-primary bridge conversion unit
  • the PWM drive is added to one of the switching tubes of the rectifying and conducting bridge arm in the period, that is, the PWM driving is only added to the third switching tube Q3 or the fourth switching tube Q4 that is not rectified and conducting in the current period in Figure 4, and the driving signal applied at the same time It should be slightly earlier than the driving signal of the secondary conversion unit, generally at least by 2%-5% period.
  • the driving signal applied by the primary bridge conversion unit is 200 ns earlier than the driving signal of the secondary conversion unit.
  • the electromotive force When entering the next working cycle, since the electromotive force is reversed, it is similar to a short circuit, and the voltage that should be applied to the input port of the primary bridge conversion unit forms a return path on the third switching tube Q3 and the fourth switching tube Q4, Since the voltage of the resonant capacitor Cr of the series resonant unit cannot change abruptly, and the port voltage of the isolation transformer Tra is directly coupled, it is equivalent to applying energy storage to the resonant inductance Lr, and at the same time, the energy storage inductance L1 on the secondary side is also at In the energy storage state, the snubber switch Q9 is turned on due to the application of a driving signal, and the snubber switch Q9 acts as a voltage source to supply power to the isolation transformer Tra to compensate for the part of the current that the energy storage inductor L1 cannot supply.
  • the short-circuit state disappears, the current of the resonant inductor Lr cannot be reversed immediately, and the induced electromotive force of the resonant inductor Lr can only carry out freewheeling in the reverse direction, so
  • the coupling voltage on the secondary side of the isolation transformer Tra is superimposed on the voltage of the series resonant unit to turn on the primary bridge conversion unit, thereby completing the conversion process of the second DC source supplying power to the first DC source.
  • the on-off of the snubber switch Q9 is directly related to the discharge and supplementary energy of the snubber capacitor Cs.
  • the driving voltage of the snubber switch Q9 cannot be turned off before the rectification process of the primary bridge conversion unit is completed.
  • the buffer switch Q9 After the buffer capacitor Cs starts reverse charging, the buffer switch Q9 must be turned off before the turned-on bridge arms of the fifth switch Q5, the sixth switch Q6, the seventh switch Q7, and the eighth switch Q8 are turned off, so that obtain zero voltage turn-off.

Abstract

一种宽范围谐振式软开关双向直流变换器及其控制方法，宽范围谐振式软开关双向直流变换器包括第一直流电源(DC1)、输入储能滤波电容(C1)、初级桥式变换单元、串联谐振单元、隔离变压器(Tra)、次级变换单元、谐振缓冲单元、输出储能滤波单元和第二直流电源(DC2)；串联谐振单元包括串联连接的谐振电容(Cr)和谐振电感(Lr)；谐振缓冲单元包括串联连接缓冲开关管(Q9)和缓冲电容(Cs)。通过对初级桥式变换单元、次级变换单元及谐振缓冲单元中的开关管施加合适频率及合适时序的驱动信号，可实现直流电压的正方向或者反方向的宽范围软开关变换。相比传统的单向或双向的变换器，可达到目前的两级式稳压变换效果，适用于连接蓄电池等电压范围较宽的负载或者电源装置，可实现高功率密度和高效率。

Description

一种宽范围谐振式软开关双向直流变换器及其控制方法 技术领域
本申请涉及直流变换器技术领域，具体涉及一种宽范围谐振式软开关双向直流变换器及其控制方法。
背景技术
随着储能产品以及电池设备相关领域的快速发展，对可以进行双向变换的电源产品需求也越来越多。现在许多设备逐渐应用了电池，需要给电池充电或者放电，由于电池的天然宽电压范围特性，同时考虑到不同产品的兼容性，对应的电压范围也越来越宽，因此常规的采用两套电路分别进行充电和放电。如今，实现双向变换已经不具备成本优势，同时普通的单级电路在效率以及满足宽电压范围充电或者放电方面也有不足。
如图1所示，当前做低电压电池包的变换电路通常是两种方式：一种是采用两级，通常是先经过一级升压或者降压方案，再经过一级DC/DC稳压变换。两级方案成本较高，同时两级变换的效率会降低。另一种是采用切换开关变换变压器匝数比，通过改变变压器的匝数比，或者采用类似的电路增减变压器线圈，专利授权号CN107733236B的专利中提到的实现方法，如图2所示，其本质就是通过额外的变压器变换电路增减，从而实现不同的电压变比，控制原理简单直接，但是由于高匝比的变化会引起更高的开关管应力，同时还会改变原有主变压器的电感，漏感参数，引入新的电流环路干扰，包括突然间电压的突变从而可能会带来另外一系列的控制上的参数变化，阶跃性的占空比调整容易产生震荡等问题，另外两个变换器的软开关协同条件可实现性相对较差；因此必须另外增加变换电路及变压器，整个变换器结构复杂且难以推广应用。
发明内容
本发明的目的在于，提供一种宽范围谐振式软开关双向直流变换器及其控制方法，既可以实现软开关高效变换，又可以相对简单且满足宽范围电压的双向变换，解决现有技术存在需要两级变换器进行多次变换、导流通路器件多和采用不能实现全变换的软开关导致损耗大，从而不适宜在体积有限或者成本要求相对较高的场所进行应用的技术问题。
本发明采取的技术方案是：一种宽范围谐振式软开关双向直流变换器，包括第一直流电 源、输入储能滤波电容、初级桥式变换单元、串联谐振单元、隔离变压器、次级变换单元、谐振缓冲单元、输出储能滤波单元和第二直流电源；所述输入储能滤波电容与所述初级桥式变换单元并联，所述初级桥式变换单元还与所述第一直流电源连接；所述隔离变压器初级侧与所述串联谐振单元串联后，再与所述初级桥式变换单元连接，所述隔离变压器次级侧与所述次级变换单元连接；所述谐振缓冲单元与所述次级变换单元并联，所述输出储能滤波单元与所述谐振缓冲单元并联，所述第二直流电源与所述输出储能滤波单元连接；
所述初级桥式变换单元为全桥式变换单元或半桥式变换单元；所述次级变换单元为全桥式变换单元或全波整流变换器；所述串联谐振单元包括串联连接的谐振电容和谐振电感；所述谐振电容与所述初级桥式变换单元连接，所述谐振电感与所述隔离变压器的初级侧连接；所述谐振缓冲单元包括串联连接缓冲开关管和缓冲电容；所述输出储能滤波单元包括串联连接的储能电感和输出储能滤波电容；所述第二直流电源与所述输出储能滤波电容连接；
当初级桥式变换单元为全桥式变换单元时，所述初级桥式变换单元包括第一开关管、第二开关管、第三开关管和第四开关管；所述第一开关管和第三开关管串联成第一桥臂，所述第二开关管和第四开关管串联成第二桥臂，所述第一桥臂和第二桥臂并联连接；所述第一开关管和第二开关管的漏极与所述第一直流电源的正极和所述输入储能滤波电容一端连接，所述第三开关管和第四开关管的源极与所述第一直流电源的负极和所述输入储能滤波电容另一端连接；所述谐振电容与所述第三开关管的漏极连接，所述隔离变压器的初级侧与所述第四开关管的漏极连接；当初级桥式变换单元为半桥式变换单元时，所述初级桥式变换单元包括串联连接的第一开关管和第二开关管，所述第一开关管的漏极与所述第一直流电源的正极和所述输入储能滤波电容一端连接，所述第二开关管的源极与所述第一直流电源的负极和所述输入储能滤波电容另一端连接；所述谐振电容与所述第二开关管的漏极连接，所述隔离变压器的初级侧与所述第二开关管的源极连接；
当次级变换单元为全桥式变换单元时，所述次级变换单元包括第五开关管、第六开关管、第七开关管和第八开关管；所述第五开关管和第七开关管串联成第三桥臂，所述第六开关管和第八开关管串联成第四桥臂，所述第三桥臂和第四桥臂并联连接；所述隔离变压器的次级侧与所述第七开关管和第八开关管的漏极连接；当次级变换单元为全波整流变换器时，所述次级变换单元包括第五开关管和第六开关管；所述第五开关管的漏极与第六开关管的漏极连接后，再与所述缓冲开关管源极和储能电感的一端连接；所述第五开关管和第六开关管的源极与所述隔离变压器的次级侧连接，所述隔离变压器的次级侧还与所述缓冲电容的一端连接。
进一步地，所述第一直流电源和第二直流电源为直流电源、整流后的交流电源、有开关控制的阶跃性电源或可以提供电源电压的负载。
进一步地，所述初级桥式变换单元和次级变换单元在仅作单方向整流变换时，所述第一到第八开关管可以是二极管，也可以是设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管。
进一步地，所述输入储能滤波电容和输出储能滤波电容为无极性的电容或有极性的电容；当所述第一直流电源或第二直流电源是阶跃性变化的电源，所述输入储能滤波电容和输出储能滤波电容为可控开关与电容串联的等效电容；所述谐振电感为外置式电感、变压器内部的耦合漏感或者外置电感和变压器内部漏感的耦合电感。
上述宽范围谐振式软开关双向直流变换器的控制方法为：
S100：根据电源状态设定电路采样或者外界通讯检测直流电路器需要输出的电压判断直流变换器的工作状态为正向工作状态还是反向工作状态；所述正向工作状态是指第一直流电源为输入，第二直流电源为输出；所述反向工作状态是指第二直流电源为输入，第一直流电源为输出；
S200：判断初级桥式变换单元以及次级变换单元的工作状态为逆变状态还是整流状态，判定谐振缓冲单元是整流缓冲谐振状态或者逆变谐振状态；并进行相应的时序逻辑配置和PWM驱动配置；初级桥式变换单元以及次级变换单元中开关管占空比最大不超过0.5，且留有足够的死区时间；施加在缓冲开关管Q9上的驱动信号频率是初级侧桥式变换单元或者次级变换单元中开关管驱动信号频率的2倍，初级侧桥式变换单元与次级变换单元的工作频率相同；
S300：根据步骤S100和S200判定的工作状态，对初级桥式变换单元、次级变换单元及谐振缓冲单元的开关管施加PWM驱动控制信号；当判定为正向工作状态时，初级桥式变换单元做逆变变换，将第一直流电源的电压变换为高频脉冲通过串联谐振单元及隔离变压器耦合传向次级侧，经次级变换单元作高频整流后传向谐振缓冲单元及输出储能滤波单元和第二直流电源；当判定为反向工作状态时，次级变换单元做逆变变换，第二直流电源的电压经过输出储能滤波单元及谐振缓冲单元传至次级变换单元作高频脉冲变换，通过隔离变压器耦合由次级侧传向初级侧，再经过串联谐振单元传至初级桥式变换单元作高频整流变换，再将直流电压传向输入滤波电容和第一直流电源；
S400：直流变换器工作在正向工作状态时，若第一直流电源的电压值经隔离变压器耦合 后高于设定的第二直流电源的电压值，则对初级桥式变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比加大；
直流变换器工作在反向工作状态时，若第二直流电源的电压值经变压器耦合后高于设定的第一直流电源的电压值，则对次级变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比增大；
S500：在初级桥式变换单元和次级变换单元的开关管按照设定进行导通后，关断初级桥式变换单元和次级变换单元的所有驱动信号，并让输入储能滤波电容和输出储能滤波单元进行续流。
进一步地，在步骤S300～S500中，初级桥式变换单元和次级桥式变换器开关管的PWM驱动信号的工作频率相同，且频率区间是谐振固有谐振频率的95％～115％区间。
进一步地，在步骤S300～S500中，当直流变换器工作在正向工作状态时，若对初级桥式变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第二直流电源电压值的需求，则固定占空比，将工作频率调节至最佳工作频率点进入升压模式，在下一个整流导通周期即将开始前对次级变换单元非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，反之则逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式；当直流变换器工作当直流变换器工作在反向工作状态时，若对次级变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第一直流电源电压值的需求，则在下一个整流导通周期即将开始前对初级桥式变换单元中非本周期内整流导通桥臂的开关管增加PWM驱动进行升压，反之则根据控制逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式。
进一步地，在步骤S300～S500中，当直流变换器工作在正向工作状态时，若次级变换单元是全桥式变换单元，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都施加PWM驱动；若次级变换单元是全波整流变换器，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动；当直流变换器工作在反向工作状态时，如果是升压模式，且初级桥式变换单元是全桥式变换单元时，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；如果初级桥式变换单元是半桥式变换单元，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动。
进一步地，在步骤S300～S500中，通过调节缓冲开关管的占空比实现对应变换模式下输 出电压一定范围的调节和次级变换单元的软开关状态；当直流变换器工作在正向工作状态时，施加给缓冲开关管的PWM驱动延迟于初级桥式变换单元的PWM驱动，即留有一定的开通死区，同时关闭时刻施加给缓冲开关管的PWM驱动与初级桥式变换单元的PWM驱动一致；当直流变换器工作在反向工作状态时，施加给缓冲开关管的PWM驱动延迟于次级变换单元的PWM驱动，即留有一定的开通死区，施加给缓冲开关管的PWM驱动关闭后的最小死区时间与次级变换单元PWM驱动的最小死区时间一致；若直流变换器工作在反向工作状态的升压模式下，则缓冲开关管不得早于初级桥式变换单元异名端所连接的开关管关断。
进一步地，在步骤S300～S500中，当直流变换器工作在反向工作状态的升压模式时，施加给初级桥式变换单元的做升压工作的开关管的驱动信号要早于次级变换单元的驱动信号，施加给初级桥式变换单元中起升压作用的开关管驱动信号为上周期同步整流信号的延迟信号，即所述延迟信号的周期为同步整流占空比、升压占空比和死区时间的总和；若是直流变换器工作在非升压模式，初级桥式变换单元和次级变换单元中的开关管则施加同步整流驱动信号。
本发明的有益效果在于：
(1)从结构及性能上，克服了传统的需要两级稳压变换电路才能实现较宽范围双向直流变换，也简化了多级电路变换的复杂性；
(2)从控制上，改变了传统的串联谐振变换需要宽范围调频实现电压控制模式，本发明主要通过调节各变换单元开关管的占空比来实现调压，与传统的桥式变换器调压控制原理接近，相对简单；
(3)从软开关实现上，利用了串联谐振变换器和谐振缓冲单元的相互配合，实现了宽范围双向变换的软开关，实现了串联谐振的软开关和传统的桥式变换器宽范围的综合性能；避免了高的电压尖峰应力以及硬开关损耗。
(4)从适用性上，改变了传统的只能是较为稳定的直流源的限制，将输入储能滤波电容串联可控开关后，输入直流电源可是有开关控制的阶跃性电源。
(5)此外，由于从结构上的归一化控制，克服了多个变换器或者变压器线圈组合切换，使得性能更加稳定，综合性价比高。
附图说明
为了更清楚地说明本发明实施例中的技术方案，下面将对实施例中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通 技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其它的附图。
图1为现有的直流变换器方框示意图；
图2为现有的双向直流变换实现方案电路示意图；
图3为本发明实施例的方框示意图；
图4为本发明实施例的电路原理图；
图5为本发明实施例的初级桥式变换单元连接具体实施示意图；
图6是本发明实施例的次级变换单元连接具体实施示意图；
图7是本发明实施例处于正向整流工作状态的电路图；
图8是本发明实施例处于反向整流工作状态的电路图；
图9是本发明实施例处于正向整流工作状态的波形示意图；
图10是本发明实施例处于反向整流工作状态的波形示意图。
附图标记解释：D1-第一二极管，D2-第二二极管，D3-第三二极管，D4-第四二极管，Q1-第一开关管，Q2-第二开关管，Q3-第三开关管，Q4-第四开关管，Q5-第五开关管，Q6-第六开关管，Q7-第七开关管，Q8-第八开关管，Q9-缓冲开关管，QA-开关管A，QB-开关管B，QC-开关管C，QD-开关管D，L1-储能电感，Lr-谐振电感，Lm-主励磁电感，Tra-隔离变压器，Cr-谐振电容，Cr1-第一谐振电容，Cr2-第二谐振电容，Cs-缓冲电容，C1-输入储能滤波电容，C2-输出储能滤波电容，DC1-第一直流电源，DC2-第二直流电源。
具体实施方式
为了能够更清楚地理解本发明的上述目的、特征和优点，下面结合附图和具体实施方式对本发明进行进一步的详细描述。在下面的描述中阐述了很多具体细节以便于充分理解本发明，但是，本发明还可以采用其他不同于在此描述的其他方式来实施，因此，本发明并不限于下面公开的具体实施例的限制。
除非另作定义，此处使用的技术术语或者科学术语应当为本申请所述领域内具有一般技能的人士所理解的通常意义。本专利申请说明书以及权利要求书中使用的“第一”、“第二”以及类似的词语并不表示任何顺序、数量或者重要性，而只是用来区分不同的组成部分。同样，“一个”或者“一”等类似词语也不表示数量限制，而是表示存在至少一个。“连接”或者“相连”等类似的词语并非限定于物理的或者机械的连接，而是可以包括电性的连接，不管是直接的还是间接的。“上”、“下”、“左”、“右”等仅用于表示相对位置关系， 当被描述对象的绝对位置改变后，则该相对位置关系也相应地改变。
如图3～图4所示，一种宽范围谐振式软开关双向直流变换器，包括第一直流电源DC1、输入储能滤波电容C1、初级桥式变换单元、串联谐振单元、隔离变压器Tra、次级变换单元、谐振缓冲单元、输出储能滤波单元和第二直流电源DC2；所述输入储能滤波电容C1与所述初级桥式变换单元并联，所述初级桥式变换单元还与所述第一直流电源DC1连接；所述隔离变压器Tra初级侧与所述串联谐振单元串联后，再与所述初级桥式变换单元连接，所述隔离变压器Tra次级侧与所述次级变换单元连接；所述谐振缓冲单元与所述次级变换单元并联，所述输出储能滤波单元与所述谐振缓冲单元并联，所述第二直流电源DC2与所述输出储能滤波单元连接；
所述初级桥式变换单元为全桥式变换单元或半桥式变换单元；所述次级变换单元为全桥式变换单元或全波整流变换器；所述串联谐振单元包括串联连接的谐振电容Cr和谐振电感Lr；所述谐振电容Cr与所述初级桥式变换单元连接，所述谐振电感Lr与所述隔离变压器Tra的初级侧连接；所述谐振缓冲单元包括串联连接缓冲开关管Q9和缓冲电容Cs；所述输出储能滤波单元包括串联连接的储能电感L1和输出储能滤波电容C2；所述第二直流电源DC2与所述输出储能滤波电容C2连接；
当初级桥式变换单元为全桥式变换单元时，所述初级桥式变换单元包括第一开关管Q1、第二开关管Q2、第三开关管Q3和第四开关管Q4；所述第一开关管Q1和第三开关管Q3串联成第一桥臂，所述第二开关管Q2和第四开关管Q4串联成第二桥臂，所述第一桥臂和第二桥臂并联连接；所述第一开关管Q1和第二开关管Q2的漏极与所述第一直流电源DC1的正极和所述输入储能滤波电容C1一端连接，所述第三开关管Q3和第四开关管Q4的源极与所述第一直流电源DC1的负极和所述输入储能滤波电容C1另一端连接；所述谐振电容Cr与所述第三开关管Q3的漏极连接，所述隔离变压器Tra的初级侧与所述第四开关管Q4的漏极连接；当初级桥式变换单元为半桥式变换单元时，所述初级桥式变换单元包括串联连接的第一开关管Q1和第二开关管Q2，所述第一开关管Q1的漏极与所述第一直流电源DC1的正极和所述输入储能滤波电容C1一端连接，所述第二开关管Q2的源极与所述第一直流电源DC1的负极和所述输入储能滤波电容C1另一端连接；所述谐振电容Cr与所述第二开关管Q2的漏极连接，所述隔离变压器Tra的初级侧与所述第二开关管Q2的源极连接；
当次级变换单元为全桥式变换单元时，所述次级变换单元包括第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8；所述第五开关管Q5和第七开关管Q7串联成第三 桥臂，所述第六开关管Q6和第八开关管Q8串联成第四桥臂，所述第三桥臂和第四桥臂并联连接；所述隔离变压器Tra的次级侧与所述第七开关管Q7和第八开关管Q8的漏极连接；当次级变换单元为全波整流变换器时，所述次级变换单元包括第五开关管Q5和第六开关管Q6；所述第五开关管Q5的漏极与第六开关管Q6的漏极连接后，再与所述缓冲开关管Q9源极和储能电感L1的一端连接；所述第五开关管Q5和第六开关管Q6的源极与所述隔离变压器Tra的次级侧连接，所述隔离变压器Tra的次级侧还与所述缓冲电容Cs的一端连接。
在本发明实施例中，所述第一直流电源DC1和第二直流电源DC2为直流电源、整流后的交流电源、有开关控制的阶跃性电源或可以提供电源电压的负载。经过整流的三相交流作为直流源输入时，每相电压均不相同，串联开关进行控制切换时，经过整流的三相交流则会是阶跃性的输入电源，本发明实施例同样适用于有开关控制的阶跃性电源。所述输入储能滤波电容C1和输出储能滤波电容C2为无极性的电容或有极性的电容；当所述第一直流电源DC1或第二直流电源DC2是阶跃性变化的电源，所述输入储能滤波电容C1和输出储能滤波电容C2为可控开关与电容串联的等效电容；所述谐振电感Lr为外置式电感、变压器内部的耦合漏感或者外置电感和变压器内部漏感的耦合电感。
所述初级桥式变换单元和次级变换单元在仅作单方向整流变换时，所述第一到第八开关管Q8可以是二极管，也可以是设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管。
如图5所示，初级桥式变换单元既可以是全桥式变换单元，也可以是半桥式变换单元。图5(a)为全桥式变换单元的电路图，采用开关管A QA、开关管B QB、开关管C QC和开关管D QD组成全桥变换器，谐振电容Cr和谐振电感Lr一起构成串联谐振单元；图5(b)为半桥式变换单元的一种连接方式，采用开关管A QA和开关管C QC组成桥臂，谐振电容Cr和谐振电感Lr一起构成串联谐振单元；图5(c)为半桥式变换单元的另一种连接方式，采用开关管A QA和开关管C QC组成第一桥臂，采用第一谐振电容Cr1和第二谐振电容Cr2串联构成另一一个桥臂，并且cr1＝cr2＝1/2*cr，其中cr1为第一谐振电容Cr1电容值，cr2为第二谐振电容Cr2电容值，cr为图5(a)中谐振电容Cr的电容值，第一谐振电容Cr1、第二谐振电容Cr2和谐振电感Lr一起构成串联谐振单元。串联谐振单元与变压器线圈的串联关系、谐振电容Cr和谐振电感Lr在串联环路中的连接顺序是均是可以调动的。
如图6所示，次级变换单元既可以是全桥式变换单元，也可以是全波整流变换器。图6(a)为全桥式变换单元的电路图，采用开关管A QA、开关管B QB、开关管C QC和开关管 D QD组成全桥变换器；图6(b)与图6(c)为全波整流变换器的两种不同连接方式，全波整流变换器又称推挽式变换器；图6(b)中开关管C QC和开关管D QD采用共漏极接法，当用作整流时，称作全波整流器；图(c)中开关管A QA和开关管B QB则是采用共源极接法，电路功能与图6(b)中所示的电路一样。
如图5及图6所示的相关整流或者逆变电路，是大家熟知的电路，其具体工作原理本领域技术人员应该理解到，本文将不再深入分析。本发明亦不局限于上述实现案例，其他可实现本发明功能的组合方式亦都属于本范畴。
本发明实施例采取的控制方法包括如下步骤：
S100：根据电源状态设定电路采样或者外界通讯检测直流电路器需要输出的电压判断直流变换器的工作状态为正向工作状态还是反向工作状态；所述正向工作状态是指第一直流电源DC1为输入，第二直流电源DC2为输出；所述反向工作状态是指第二直流电源DC2为输入，第一直流电源DC1为输出；
S200：判断初级桥式变换单元以及次级变换单元的工作状态为逆变状态还是整流状态，判定谐振缓冲单元是整流缓冲谐振状态或者逆变谐振状态；并进行相应的时序逻辑配置和PWM驱动配置；初级桥式变换单元以及次级变换单元中开关管占空比最大不超过0.5，且留有足够的死区时间；施加在缓冲开关管Q9上的驱动信号频率是初级侧桥式变换单元或者次级变换单元中开关管驱动信号频率的2倍，初级侧桥式变换单元与次级变换单元的工作频率相同；
S300：根据步骤S100和S200判定的工作状态，对初级桥式变换单元、次级变换单元及谐振缓冲单元的开关管施加PWM驱动控制信号；当判定为正向工作状态时，初级桥式变换单元做逆变变换，将第一直流电源DC1的电压变换为高频脉冲通过串联谐振单元及隔离变压器Tra耦合传向次级侧，经次级变换单元作高频整流后传向谐振缓冲单元及输出储能滤波单元和第二直流电源DC2；当判定为反向工作状态时，次级变换单元做逆变变换，第二直流电源DC2的电压经过输出储能滤波单元及谐振缓冲单元传至次级变换单元作高频脉冲变换，通过隔离变压器Tra耦合由次级侧传向初级侧，再经过串联谐振单元传至初级桥式变换单元作高频整流变换，再将直流电压传向输入滤波电容和第一直流电源DC1；
S400：直流变换器工作在正向工作状态时，若第一直流电源DC1的电压值经隔离变压器Tra耦合后高于设定的第二直流电源DC2的电压值，则对初级桥式变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比加大；
直流变换器工作在反向工作状态时，若第二直流电源DC2的电压值经变压器耦合后高于设定的第一直流电源DC1的电压值，则对次级变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比增大；
S500：在初级桥式变换单元和次级变换单元的开关管按照设定进行导通后，关断初级桥式变换单元和次级变换单元的所有驱动信号，并让输入储能滤波电容和输出储能滤波单元进行续流。
在步骤S300～S500中，初级桥式变换单元和次级桥式变换器开关管的PWM驱动信号的工作频率相同，且频率区间是谐振固有谐振频率的95％～115％区间。在本发明实施例中，初级桥式变换单元和次级变换单元开关管的PWM驱动信号的工作频率为谐振固有谐振频率的105％。
当直流变换器工作在正向工作状态时，若对初级桥式变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第二直流电源DC2电压值的需求，则固定占空比，将工作频率调节至最佳工作频率点进入升压模式，在下一个整流导通周期即将开始前对次级变换单元非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，反之则逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式；当直流变换器工作当直流变换器工作在反向工作状态时，若对次级变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第一直流电源DC1电压值的需求，则在下一个整流导通周期即将开始前对初级桥式变换单元中非本周期内整流导通桥臂的开关管增加PWM驱动进行升压，反之则根据控制逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式。
当直流变换器工作在正向工作状态时，若次级变换单元是全桥式变换单元，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都施加PWM驱动；若次级变换单元是全波整流变换器，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动；当直流变换器工作在反向工作状态时，如果是升压模式，且初级桥式变换单元是全桥式变换单元时，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；如果初级桥式变换单元是半桥式变换单元，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动。
通过调节缓冲开关管Q9的占空比实现对应变换模式下输出电压一定范围的调节和次级 变换单元的软开关状态；当直流变换器工作在正向工作状态时，施加给缓冲开关管Q9的PWM驱动延迟于初级桥式变换单元的PWM驱动，即留有一定的开通死区，同时关闭时刻施加给缓冲开关管Q9的PWM驱动与初级桥式变换单元的PWM驱动一致；当直流变换器工作在反向工作状态时，施加给缓冲开关管Q9的PWM驱动延迟于次级变换单元的PWM驱动，即留有一定的开通死区，施加给缓冲开关管Q9的PWM驱动关闭后的最小死区时间与次级变换单元PWM驱动的最小死区时间一致；若直流变换器工作在反向工作状态的升压模式下，则缓冲开关管Q9不得早于初级桥式变换单元异名端所连接的开关管关断。
当直流变换器工作在反向工作状态的升压模式时，施加给初级桥式变换单元的做升压工作的开关管的驱动信号要早于次级变换单元的驱动信号，施加给初级桥式变换单元中起升压作用的开关管驱动信号为上周期同步整流信号的延迟信号，即所述延迟信号的周期为同步整流占空比、升压占空比和死区时间的总和；若是直流变换器工作在非升压模式，初级桥式变换单元和次级变换单元中的开关管则施加同步整流驱动信号。
若根据外部信号判定为正向工作模式，即将第一直流电源DC1电压变换为第二直流电源DC2电压，则按照上前述控制方法，设定初级侧桥式变换器的工作频率为串联谐振单元的谐振频率
Figure PCTCN2022117907-appb-000001
其中，lr为谐振电感Lr的电感值，cr为谐振电容Cr的电容值。谐振缓冲单元的缓冲开关管Q9的开关频率为2f 0，此时次级变换单元的功能是全桥整流器，为了获得高效率，可以对次级变换单元施加同步整流信号。为了讨论方便，也可以不加任何驱动信号，将第五开关管Q5和第六开关管Q6视作二极管整流，图4可简化成如图7所示的电路图。
根据计算，如果输出电压是第二直流电源DC2的最高电压点，经隔离变压器Tra匝数比折算到输入侧，折算后的输出电压略低于输入电压。因此，根据前述控制方法，对第一开关管Q1、第二开关管Q2、第三开关管Q3和第四开关管Q4所施加的桥式变换驱动占空比为50％左右，优选为45％，由于隔离变压器Tra初级侧回路中的串联谐振单元的存在，此时需对初级侧桥式变换器的工作频率进行调节，使初级侧桥式变换器的工作频率高于谐振频率，如105％*f 0。此时全桥变换器等同于LLC全桥变换器，第一开关管Q1、第二开关管Q2、第三开关管Q3和第四开关管Q4均实现软开关，同时为了实现缓冲开关管Q9的软开通，对缓冲开关管Q9施加的PWM驱动稍微延迟，开始电流经缓冲开关管Q9的反并联二极管对缓冲电容Cs充电，然后缓冲开关管Q9施加驱动开通，则构成了零电压开通。随着时间的推移，次级整流电流逐渐加大，并呈弦波形状，而输出侧的电流由于储能电感L1的存在，其呈线 性变化。因此，此时对缓冲电容Cs充电的电流为次级整流电流I-rec-sec减去储能电感L1电流I-L1，随着时间的推移，次级整流电流逐渐降低，且输出电流逐渐加大，因此缓冲电容Cs开始储能，储能电感L1放电。
当初级桥式变换单元的开关管关闭时，可近似的看作储能电感L1前的电压即将消失，如果缓冲开关管Q9立即或者稍作延迟关闭，则意味储能电感L1需要立即通过次级变换单元进行续流。在续流前，储能电感L1会抽取与次级变换单元等效寄生电容的电流并逐渐降低到零电压。因此，缓冲开关管Q9可以看作是零电压关断。在初级桥式变换单元下次开通且储能电感L1电流没有断流反向前，储能电感L1的电流都只能通过次级整流变换桥进行续流，因此隔离变压器Tra一直被箝位为零，为下次的零电压开通提供了准备。同时，在变换过程中，缓冲开关管Q9和缓冲电容Cs协助了次级变换单元的软开通或者软关断，并吸收和缓冲了初级桥式变换单元的多余电流，让储能电感L1工作在被施加脉冲电压的状态，类似于降压状态，很好的解决了原来串联谐振变换器只能靠变频调压且调节范围不大、调占空比调压非线性化的缺点。通过谐振缓冲单元的配合，使得本发明实施例既获得了软开关变换的优势，又实现了降压变换器降压的控制简单性优点，相关波形示意图如图9所示。
因此，当本发明实施例处于正向工作状态下，如需要调压，则需要缓冲开关管Q9配合初级侧的串联谐振单元一起调节占空比，从而实现稳定调压和软开关。若对初级桥式变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第二直流电源DC2的电压需求，则将初级桥式变换单元施加的PWM驱动固定为最大占空比，并将初级桥式变换单元的工作频率调节至最佳工作频率点进入升压模式，在下一个整流导通周期即将开始前对次级变换单元非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，通过调节PWM驱动的占空比大小实现输出电压的调节，如果不需要升压即可满足第二直流电源DC2的电压需求则退出该升压模式。
若判定本发明实施例需要工作在反向工作模式，即将第二直流电源DC2电压变换为第一直流电源DC1电压，此时初级桥式变换单元主要是工作在整流模式。如果根据计算，将第二直流电源DC2电压经过隔离变压器Tra匝数比反向折算后，不需进入升压模式，则图4可以简化如图8所示，相关原理为本领域技术人员的公知常识。为了便于讨论，假设本发明实施例需要进入升压模式，则即便只是反向整流，也必须是带桥式开关管的变换器。因此，将次级变换单元中第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8的驱动占空比施加到最大，并对初级桥式变换单元的非本周期内整流导通桥臂的其中一个开关管增加 PWM驱动，即只对图4中本周期内非整流导通的第三开关管Q3或者第四开关管Q4增加PWM驱动，同时施加的驱动信号应该要略微早于次级变换单元的驱动信号，一般至少提前以2％～5％周期。在本发明实施例中，初级桥式变换单元施加的驱动信号比次级变换单元的驱动信号早200ns。因为初级桥式变换单元的其中两个开关管为本周期内非整流导通，即在前周期内整流导通或者为变压器整流回路电动势的顺向通路，因此提前开通开关管的方式为零电压开通。等到进入下个工作周期，由于处于电动势反向，则类似短路，本应加在初级桥式变换单元输入端口上的电压在第三开关管Q3和第四开关管Q4上形成了回流通路，由于串联谐振单元的谐振电容Cr的电压不能突变，而隔离变压器Tra的端口电压是直接耦合，因此相当于给谐振电感Lr施加储能，同时次级侧的储能电感L1也因为电流通流处于储能状态，对缓冲开关管Q9因施加驱动信号导通，缓冲开关管Q9充当电压源给隔离变压器Tra供电，弥补储能电感L1所不能供给的那部分电流。
当施加在第三开关管Q3或者第四开关管Q4上的驱动电压结束后，短路状态消失，谐振电感Lr的电流不能立即反向，谐振电感Lr的感应电动势只能反向进行续流，因此隔离变压器Tra次级侧的耦合电压叠加串联谐振单元的电压，使初级桥式变换单元导通，以此完成了第二直流源向第一直流源供电的转换过程。同时，缓冲开关管Q9的通断直接关系到缓冲电容Cs的放电补充能量，因此，缓冲开关管Q9的驱动电压不能在初级桥式变换单元整流过程结束前关闭。待缓冲电容Cs开始反向充电后，缓冲开关管Q9必须在第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8的已开通桥臂关断前关闭，以便获得零电压关断。在第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8的已开通桥臂开始关闭的同时，由于隔离变压器Tra中的电流不能立即反向，只能通过第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8中的先前未开通桥臂的反并二极管进行续流，即隔离变压器Tra初级侧线圈中的电流被向第二直流电源DC2侧反向释放，由于储能电感L1的电流方向不能反转和突变，因此两种电流的汇合接点处导致电压不断提高，并最终通过缓冲开关管Q9的反并二极管被缓冲电容Cs箝位并吸收了电流。因此，若先前开通的是第五开关管Q5及第八开关管Q8，则进行续流的是第六开关管Q6和第七开关管Q7的反并二极管。因此在下一阶段开通时，第六开关管Q6和第七开关管Q7则实现了零电压开通。
以上所述仅为本发明的优选实施例而已，并不用于限制本发明，对于本领域的技术人员来说，本发明可以有各种更改和变化。凡在本发明的精神和原则之内，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (10)

  1. 一种宽范围谐振式软开关双向直流变换器，其特征在于，包括第一直流电源、输入储能滤波电容、初级桥式变换单元、串联谐振单元、隔离变压器、次级变换单元、谐振缓冲单元、输出储能滤波单元和第二直流电源；所述输入储能滤波电容与所述初级桥式变换单元并联，所述初级桥式变换单元还与所述第一直流电源连接；所述隔离变压器初级侧与所述串联谐振单元串联后，再与所述初级桥式变换单元连接，所述隔离变压器次级侧与所述次级变换单元连接；所述谐振缓冲单元与所述次级变换单元并联，所述输出储能滤波单元与所述谐振缓冲单元并联，所述第二直流电源与所述输出储能滤波单元连接；
    所述初级桥式变换单元为全桥式变换单元或半桥式变换单元；所述次级变换单元为全桥式变换单元或全波整流变换器；所述串联谐振单元包括串联连接的谐振电容和谐振电感；所述谐振电容与所述初级桥式变换单元连接，所述谐振电感与所述隔离变压器的初级侧连接；所述谐振缓冲单元包括串联连接缓冲开关管和缓冲电容；所述输出储能滤波单元包括串联连接的储能电感和输出储能滤波电容；所述第二直流电源与所述输出储能滤波电容连接；
    当初级桥式变换单元为全桥式变换单元时，所述初级桥式变换单元包括第一开关管、第二开关管、第三开关管和第四开关管；所述第一开关管和第三开关管串联成第一桥臂，所述第二开关管和第四开关管串联成第二桥臂，所述第一桥臂和第二桥臂并联连接；所述第一开关管和第二开关管的漏极与所述第一直流电源的正极和所述输入储能滤波电容一端连接，所述第三开关管和第四开关管的源极与所述第一直流电源的负极和所述输入储能滤波电容另一端连接；所述谐振电容与所述第三开关管的漏极连接，所述隔离变压器的初级侧与所述第四开关管的漏极连接；当初级桥式变换单元为半桥式变换单元时，所述初级桥式变换单元包括串联连接的第一开关管和第二开关管，所述第一开关管的漏极与所述第一直流电源的正极和所述输入储能滤波电容一端连接，所述第二开关管的源极与所述第一直流电源的负极和所述输入储能滤波电容另一端连接；所述谐振电容与所述第二开关管的漏极连接，所述隔离变压器的初级侧与所述第二开关管的源极连接；
    当次级变换单元为全桥式变换单元时，所述次级变换单元包括第五开关管、第六开关管、第七开关管和第八开关管；所述第五开关管和第七开关管串联成第三桥臂，所述第六开关管和第八开关管串联成第四桥臂，所述第三桥臂和第四桥臂并联连接；所述隔离变压器的次级侧与所述第七开关管和第八开关管的漏极连接；当次级变换单元为全波整流变换器时，所述次级变换单元包括第五开关管和第六开关管；所述第五开关管的漏极与第六开关管的漏极连接后，再与所述缓冲开关管源极和储能电感的一端连接；所述第五开关管和第六开关管的源 极与所述隔离变压器的次级侧连接，所述隔离变压器的次级侧还与所述缓冲电容的一端连接。
  2. 根据权利要求1所述的一种宽范围谐振式软开关双向直流变换器，其特征在于，所述第一直流电源和第二直流电源为直流电源、整流后的交流电源、有开关控制的阶跃性电源或可以提供电源电压的负载。
  3. 根据权利要求1所述的一种宽范围谐振式软开关双向直流变换器，其特征在于，所述初级桥式变换单元和次级变换单元在仅作单方向整流变换时，所述第一到第八开关管可以是二极管，也可以是设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管。
  4. 根据权利要求1所述的一种宽范围谐振式软开关双向直流变换器，其特征在于，所述输入储能滤波电容和输出储能滤波电容为无极性的电容或有极性的电容；当所述第一直流电源或第二直流电源是阶跃性变化的电源，所述输入储能滤波电容和输出储能滤波电容为可控开关与电容串联的等效电容；所述谐振电感为外置式电感、变压器内部的耦合漏感或者外置电感和变压器内部漏感的耦合电感。
  5. 一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，用于控制权利要求1～4任一权利要求所述的宽范围谐振式软开关双向直流变换器，包括如下步骤：
    S100：根据电源状态设定电路采样或者外界通讯检测直流电路器需要输出的电压判断直流变换器的工作状态为正向工作状态还是反向工作状态；所述正向工作状态是指第一直流电源为输入，第二直流电源为输出；所述反向工作状态是指第二直流电源为输入，第一直流电源为输出；
    S200：判断初级桥式变换单元以及次级变换单元的工作状态为逆变状态还是整流状态，判定谐振缓冲单元是整流缓冲谐振状态或者逆变谐振状态；并进行相应的时序逻辑配置和PWM驱动配置；初级桥式变换单元以及次级变换单元中开关管占空比最大不超过0.5，且留有足够的死区时间；施加在缓冲开关管Q9上的驱动信号频率是初级侧桥式变换单元或者次级变换单元中开关管驱动信号频率的2倍，初级侧桥式变换单元与次级变换单元的工作频率相同；
    S300：根据步骤S100和S200判定的工作状态，对初级桥式变换单元、次级变换单元及谐振缓冲单元的开关管施加PWM驱动控制信号；当判定为正向工作状态时，初级桥式变换单元做逆变变换，将第一直流电源的电压变换为高频脉冲通过串联谐振单元及隔离变压器耦合传向次级侧，经次级变换单元作高频整流后传向谐振缓冲单元及输出储能滤波单元和第二 直流电源；当判定为反向工作状态时，次级变换单元做逆变变换，第二直流电源的电压经过输出储能滤波单元及谐振缓冲单元传至次级变换单元作高频脉冲变换，通过隔离变压器耦合由次级侧传向初级侧，再经过串联谐振单元传至初级桥式变换单元作高频整流变换，再将直流电压传向输入滤波电容和第一直流电源；
    S400：直流变换器工作在正向工作状态时，若第一直流电源的电压值经隔离变压器耦合后高于设定的第二直流电源的电压值，则对初级桥式变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比加大；
    直流变换器工作在反向工作状态时，若第二直流电源的电压值经变压器耦合后高于设定的第一直流电源的电压值，则对次级变换单元施加的PWM驱动做缩小占空比调节，反之则调节占空比增大；
    S500：在初级桥式变换单元和次级变换单元的开关管按照设定进行导通后，关断初级桥式变换单元和次级变换单元的所有驱动信号，并让输入储能滤波电容和输出储能滤波单元进行续流。
  6. 根据权利要求5所述的一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，在步骤S300～S500中，初级桥式变换单元和次级桥式变换器开关管的PWM驱动信号的工作频率相同，且频率区间是谐振固有谐振频率的95％～115％区间。
  7. 根据权利要求5所述的一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，在步骤S300～S500中，当直流变换器工作在正向工作状态时，若对初级桥式变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第二直流电源电压值的需求，则固定占空比，将工作频率调节至最佳工作频率点进入升压模式，在下一个整流导通周期即将开始前对次级变换单元非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，反之则逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式；当直流变换器工作当直流变换器工作在反向工作状态时，若对次级变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到第一直流电源电压值的需求，则在下一个整流导通周期即将开始前对初级桥式变换单元中非本周期内整流导通桥臂的开关管增加PWM驱动进行升压，反之则根据控制逐渐减小对初级桥式变换单元施加的PWM驱动占空比并退出升压模式。
  8. 根据权利要求5所述的一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，在步骤S300～S500中，当直流变换器工作在正向工作状态时，若次级变换单元是全桥式变换单元，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个 开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都施加PWM驱动；若次级变换单元是全波整流变换器，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动；当直流变换器工作在反向工作状态时，如果是升压模式，且初级桥式变换单元是全桥式变换单元时，则在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，或对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；如果初级桥式变换单元是半桥式变换单元，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管施加PWM驱动。
  9. 根据权利要求5所述的一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，在步骤S300～S500中，通过调节缓冲开关管的占空比实现对应变换模式下输出电压一定范围的调节和次级变换单元的软开关状态；当直流变换器工作在正向工作状态时，施加给缓冲开关管的PWM驱动延迟于初级桥式变换单元的PWM驱动，即留有一定的开通死区，同时关闭时刻施加给缓冲开关管的PWM驱动与初级桥式变换单元的PWM驱动一致；当直流变换器工作在反向工作状态时，施加给缓冲开关管的PWM驱动延迟于次级变换单元的PWM驱动，即留有一定的开通死区，施加给缓冲开关管的PWM驱动关闭后的最小死区时间与次级变换单元PWM驱动的最小死区时间一致；若直流变换器工作在反向工作状态的升压模式下，则缓冲开关管不得早于初级桥式变换单元异名端所连接的开关管关断。
  10. 根据权利要求5所述的一种宽范围谐振式软开关双向直流变换器的控制方法，其特征在于，在步骤S300～S500中，当直流变换器工作在反向工作状态的升压模式时，施加给初级桥式变换单元的做升压工作的开关管的驱动信号要早于次级变换单元的驱动信号，施加给初级桥式变换单元中起升压作用的开关管驱动信号为上周期同步整流信号的延迟信号，即所述延迟信号的周期为同步整流占空比、升压占空比和死区时间的总和；若是直流变换器工作在非升压模式，初级桥式变换单元和次级变换单元中的开关管则施加同步整流驱动信号。
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