WO2023098216A1 - 一种无输入储能电感隔离谐振软开关型三相pfc变换器及其控制方法 - Google Patents

一种无输入储能电感隔离谐振软开关型三相pfc变换器及其控制方法 Download PDF

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Publication number
WO2023098216A1
WO2023098216A1 PCT/CN2022/117893 CN2022117893W WO2023098216A1 WO 2023098216 A1 WO2023098216 A1 WO 2023098216A1 CN 2022117893 W CN2022117893 W CN 2022117893W WO 2023098216 A1 WO2023098216 A1 WO 2023098216A1
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Prior art keywords
resonant
switch
switching
phase
rectification
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PCT/CN2022/117893
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English (en)
French (fr)
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刘斌
李伦全
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刘三英
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Publication of WO2023098216A1 publication Critical patent/WO2023098216A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/32—Means for protecting converters other than automatic disconnection
    • H02M1/34—Snubber circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/38—Means for preventing simultaneous conduction of switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
    • H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
    • H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the technical field of AC/DC converters, in particular to a three-phase PFC converter with no input energy storage inductance isolation resonant soft switch and a control method thereof.
  • the AC-DC converter circuit with three-phase AC input it is generally PFC+DC/DC two-stage circuit. Due to the multiple transformations of the two-stage circuit, more switching loss and conduction loss are caused, and the efficiency drops seriously.
  • the purpose of the present invention is to provide a three-phase PFC converter without input energy storage inductance isolation resonant soft switching and its control method, which can solve the limitation of application due to the complexity of the converter structure and control method in the above-mentioned prior art technical issues.
  • the technical solution adopted by the present invention is: a three-phase PFC converter with no input energy storage inductance isolation resonant soft switch, including an input switch type rectifier bridge arm group, an absorbing buffer unit, a bridge resonant conversion unit, a transformer and a secondary rectifier Output unit;
  • the input switching rectification bridge arm group includes a first switch rectification bridge arm, a second switch rectification bridge arm and a third switch rectification bridge arm, and each switch rectification bridge arm includes an AC input port, a positive output port and Negative output port, an equivalent controllable selection switch is set between the AC input port, the positive output port and the negative output port;
  • the absorbing buffer unit includes a fourth switch tube and a absorbing buffer capacitor connected in series; the fourth The sources of the switch tubes are respectively connected to the positive output ports of the first switch rectifier bridge arm, the second switch rectifier bridge arm and the third switch rectifier bridge arm, and the absorbing buffer capacitors are respectively connected to the first switch rectifier bridge arm and the second switch rectif
  • the negative output ports of the rectifier bridge arm and the third switch rectifier bridge arm are connected to a bridge resonant conversion unit;
  • the bridge resonant conversion unit includes a bridge inverter circuit and a first series resonant unit, and the bridge inverter circuit is a full In a bridge inverter circuit or a half bridge inverter circuit
  • the first series resonant unit includes a first resonant capacitor and a first resonant inductance connected in series
  • the input end of the bridge inverter circuit is connected to the absorbing buffer unit connected, the output terminal is connected to the first resonant capacitor and the primary side of the transformer, and the primary side of the transformer is also connected to the first resonant inductance;
  • the secondary side of the transformer is connected to the secondary rectification output unit;
  • the secondary The rectification output unit includes a rectification circuit and a filter capacitor.
  • the input end of the rectification circuit is connected to the secondary side of the transformer, and the output end is connected to the filter capacitor.
  • the rectification circuit is a voltage doubler rectification circuit, a full wave rectification circuit or a full wave rectification circuit.
  • Bridge rectification circuit, the rectification circuit is composed of all diodes, switching tubes with anti-parallel diodes or other forms of switching tubes with rectification function, or is composed of switching tubes and diodes mixed.
  • the equivalent controllable selection switch is a series combination of a diode and a high-frequency switching tube or two high-frequency switching tubes are connected in reverse series and then connected to a diode, and the equivalent controllable selection switch is conducted according to the AC rectification It is necessary to apply a high-frequency PWM drive signal to the high-frequency switch tube to control the turn-on and turn-off so as to achieve a direction-selective conduction connection, that is, to form a high-frequency pulse rectification conduction for the positive half-wave of the AC, or the negative half-wave of the AC Wave high frequency pulse rectification conduction.
  • the equivalent controllable selection switch consists of one switch tube and four diodes, or two switch tubes and two diodes;
  • the source of the switch tube is connected to the anodes of the eleventh diode and the twelfth diode, and the switch tube
  • the source is connected to the cathode of the third diode and the fourth diode;
  • the cathode of the eleventh diode is connected to the positive output port;
  • the cathode of the twelfth diode is connected to the third and second diodes
  • the anode of the pole tube is connected to the AC input port;
  • the anode of the fourth diode is connected to the negative output port;
  • the first connection mode is that after the first switch tube and the eleventh diode are connected in series to form a first branch circuit, one end is connected to the AC input port, the other end is connected to the positive output port, and the second switch tube and the first switch tube are connected to the first branch circuit.
  • twelve diodes are connected in series to form a second branch, one end is connected to the AC input port, and the other end is connected to the negative output port, and the first branch and the second branch are related to the AC input port symmetry;
  • the second connection mode is that after the first switch tube and the second switch tube are reversely connected in series, the first switch tube is connected to the AC input port, and the second switch tube is connected to the anode of the eleventh diode and the anode of the twelfth
  • the cathode of the pole tube is connected, the cathode of the eleventh diode is connected to the positive output port, and the anode of the twelfth diode is connected to the negative output port;
  • the third connection mode is that after the first switch tube and the second switch tube are reversely connected in series, the first switch tube is connected to the AC input port, and the anode of the eleventh diode is connected to the first switch tube and the second switch tube connected in series, the cathode of the eleventh diode is connected to the positive output port, the cathode of the twelfth diode is connected to the second switching tube, and the anode of the twelfth diode is connected to the negative output port connect.
  • the switching tubes in the input switching rectifier bridge arm group are high-frequency switching tubes provided with anti-parallel diodes, and the anti-parallel diodes are integrated diodes, parasitic diodes or external diodes;
  • the absorption and buffer capacitors are electrodeless a polarized capacitor or a polarized capacitor;
  • the first resonant inductance is an external inductor, a coupling leakage inductance inside the transformer, or a coupling inductor between an external inductor and an internal leakage inductance of the transformer.
  • the second series resonant unit includes a second resonant capacitor and a second resonant inductance connected in series, and the resonant frequency of the first resonant capacitor and the first resonant inductance is
  • lr1 is the inductance value of the first resonant inductor
  • cr1 is the capacitance value of the first resonant capacitor
  • an input filter is further included, and the input filter is connected to the AC input ports of the first switching rectification bridge arm, the second switching rectification bridge arm and the third switching rectification bridge arm.
  • S200 Analyze the instantaneous value of the voltage of each phase power supply in each of the interval segments according to the phase-locked phase in step S100;
  • S300 Detect the input and output conditions, judge whether the input and output conditions meet the working conditions required by the system, and continue to wait if the conditions are not met; if the conditions are met, the converter starts to work;
  • steps S300-S400 when the input switch-type rectifier bridge arm group and the fourth switch tube are in the PWM working state, the PWM switching frequency of the fourth switch tube and the control of the first switch rectifier bridge arm and the second switch rectifier
  • the PWM switching frequencies of the bridge arm and the third switching rectifier bridge arm are consistent, and are twice the operating frequency of the bridge resonant conversion unit.
  • the driving duty ratios of each drive are consistent and do not exceed 0.5, and the necessary dead time of the bridge inverter circuit is left;
  • the "high" mode PWM drive signal is a high-level signal or PWM drive signal that always exists, and the The high-level time of the drive voltage of the PWM drive signal in the "high” mode is greater than the high-level time of the PWM drive signal in the "middle” mode;
  • the "high" mode PWM driving signal of the bridge arm group is a PWM driving signal with a duty cycle not exceeding 50%.
  • the switching frequency of the fixed switching tube is near the resonance frequency;
  • PWM drive is only applied to one of the switching tubes of the rectifying and conducting bridge arm in the off-cycle period, and PWM drive can also be added to the two switching tubes of the rectifying and conducting bridge arm in the off-cycle period; if the secondary conversion unit is half
  • the PWM drive is only added to the non-rectification conduction switch in this period before the next rectification conduction period; when the inverter duty cycle of the bridge resonant conversion unit does not reach 47.5% or the maximum duty cycle
  • a duty ratio it is a step-down mode.
  • a differential drive is applied to the two switching tubes that are turned on each time.
  • the driving signal of one switching tube is driven by the duty cycle calculated according to the control, and the other
  • the driving signal of the switching tube is driven by a fixed maximum ratio.
  • the switching tube drive signal applied to the bridge resonant conversion unit for boosting work is earlier than the drive signal of the secondary rectification output unit, and is applied to the bridge resonant conversion unit
  • the driving signal of the switching tube for boosting work is the delay of the synchronous rectification signal in the previous cycle, that is, the sum of the synchronous rectification duty cycle plus the boost duty cycle and the dead time; in the non-boost mode, the bridge resonant conversion unit and the secondary A synchronous driving signal is applied to the switch tube of the rectification output unit.
  • the current conduction time of each phase is proportional to the instantaneous value of the phase voltage, and the current conduction time of the phase with the largest instantaneous value is equal to the sum of the current conduction time of the other two phases.
  • the present invention has changed the realization method that traditional AC/DC converters need PFC voltage stabilizing circuit plus DC isolation transformation circuit, by the topology of the present invention, can save the conventional AC/DC converter after the alternating current rectification Energy storage units, such as large inductors and large capacitors;
  • the present invention has better stability and can effectively deal with jumps in input voltage or sudden changes in polarity In some cases, the grid adaptability is stronger, the work stability is higher, and the equipment quality is more reliable.
  • FIG. 1 is a schematic block diagram of an existing AC-DC converter
  • Figure 2 is a schematic structural diagram of an existing isolated Swiss single-stage converter
  • Fig. 3 is a schematic diagram of a block structure of an embodiment of the present invention.
  • Fig. 4 is a schematic diagram of a three-phase voltage waveform and a schematic diagram of a definition of an intersection point according to an embodiment of the present invention
  • Fig. 5 is a specific connection schematic diagram of an embodiment of the present invention.
  • Fig. 6 is a schematic diagram of a specific embodiment of the input switching rectifier bridge arm group in the embodiment of the present invention.
  • Fig. 7 is a schematic diagram of the AB phase conduction circuit in the AC-BC section of the embodiment of the present invention.
  • Fig. 8 is a schematic diagram 1 of the freewheeling circuit of the BC phase in the AC-0 section of the embodiment of the present invention.
  • FIG. 9 is a schematic diagram of the current freewheeling circuit of the bridge-type resonant conversion unit in the AC-0 interval of the embodiment of the present invention.
  • Fig. 10 is a second schematic diagram of the AB-phase conduction loop in the AC-BC section of the embodiment of the present invention.
  • Fig. 11 is a second schematic diagram of the BC-phase freewheeling circuit in the AC-0 section of the embodiment of the present invention.
  • FIG. 12 is a second schematic diagram of the current freewheeling circuit of the bridge-type resonant conversion unit in the AC-0 interval of the embodiment of the present invention.
  • Fig. 13 is a schematic diagram of the AC phase freewheeling circuit in the 0-BC interval of the embodiment of the present invention.
  • Fig. 14 is a second schematic diagram of the AC phase freewheeling circuit in the 0-BC interval of the embodiment of the present invention.
  • Fig. 15 is a schematic diagram of a specific implementation key waveform at a certain moment in the AC-0 interval of the embodiment of the present invention.
  • Fig. 16 is a schematic diagram of the driving waveform relationship of each switch rectifier bridge arm in the power frequency cycle of the embodiment of the present invention.
  • Fig. 17 is a schematic diagram 1 of the specific implementation of the connection of the secondary rectification output unit according to the embodiment of the present invention.
  • Fig. 18 is a schematic diagram 2 of the specific implementation of the connection of the secondary rectification output unit according to the embodiment of the present invention.
  • Fig. 19 is a specific connection schematic diagram of another embodiment of the present invention.
  • Fig. 20 is a schematic diagram 1 of a specific implementation of the bridge-type resonant conversion unit as a half-bridge resonant conversion unit according to an embodiment of the present invention
  • FIG. 21 is a schematic diagram 2 of a specific implementation of the bridge-type resonant conversion unit as a half-bridge resonant conversion unit according to an embodiment of the present invention.
  • a three-phase PFC converter with no input energy storage inductance isolation resonant soft switching type includes input switching rectifier bridge arm group, absorption buffer unit, bridge resonant conversion unit, transformer Tra and secondary rectifier output unit;
  • the input switching rectification bridge arm group includes a first switch rectification bridge arm KB1, a second switch rectification bridge arm KB2 and a third switch rectification bridge arm KB3, and each switch rectification bridge arm includes an AC input port 1, a positive The output port 2 and the negative output port 3, an equivalent controllable selection switch is arranged between the AC input port 1, the positive output port 2 and the negative output port 3;
  • the absorbing buffer unit includes a fourth switch tube Q4 connected in series and absorbing snubber capacitor Cs; the source of the fourth switch tube Q4 is respectively connected to the positive output port 2 of the first switch rectifier bridge arm KB1, the second switch rectifier bridge arm KB2 and the third switch rectifier bridge arm KB3, the The absorbing buffer capacitor Cs is respectively connected to the negative output
  • a resonant inductance Lr1 is connected; the secondary side of the transformer Tra is connected to the secondary rectifier output unit; the secondary rectifier output unit includes a rectifier circuit and a filter capacitor C1, and the input terminal of the rectifier circuit is connected to the transformer Tra The secondary side is connected, and the output terminal is connected to the filter capacitor C1.
  • the rectification circuit is a voltage doubler rectification circuit, a full-wave rectification circuit or a full-bridge rectification circuit.
  • the rectification circuit is composed of all diodes, and a switch with an antiparallel diode tube or one of other forms of switching tubes with rectification function, or a mixture of switching tubes and diodes.
  • the bridge-type resonant conversion unit is a full-bridge resonant conversion unit
  • the rectification circuit of the secondary rectification output unit is a full-bridge rectification circuit composed of all diodes
  • the specific connection circuit of the embodiment of the present invention is shown in FIG. 5 .
  • the full-bridge resonant conversion unit includes a fifth switching tube Q5, a sixth switching tube Q6, a seventh switching tube Q7, and an eighth switching tube Q8, and the fifth switching tube Q5 and the seventh switching tube Q7 are connected in series to form a first bridge arm, the sixth switch tube Q6 and the eighth switch tube Q8 are connected in series to form a second bridge arm, and the first bridge arm and the second bridge arm are connected in parallel; the drains of the fifth switch tube Q5 and the sixth switch tube Q6 It is connected to the source of the fourth switch, the sources of the seventh switching tube Q7 and the eighth switching tube Q8 are connected to the snubber capacitor Cs; the source of the fifth switching tube Q5 is connected to the first resonant capacitor Cr1, and the sixth switching tube The source of Q6 is connected to the primary side of the transformer Tra.
  • the rectifier circuit includes a first diode D1, a second diode D2, a third diode D3 and a fourth diode D4, and the first diode D1 and the third diode D3 are connected in series to form The third bridge arm, the second diode D2 and the fourth diode D4 are connected in series to form the fourth bridge arm, the third bridge arm and the fourth bridge arm are connected in parallel; the first diode D1 and the first diode D1
  • the cathode of the diode D1 is connected to one end of the filter capacitor C1, and constitutes the positive port of the output power supply of the embodiment of the present invention; the anodes of the third diode D3 and the fourth diode D4 are connected to the other end of the filter capacitor C1, And constitute the negative port of the output power of the embodiment of the present invention.
  • the embodiment of the present invention also includes an input filter, and the input filter is connected to the AC input port of the first switch rectification bridge arm KB1, the second switch rectification bridge arm KB2 and the third switch rectification bridge arm KB3 1 connection, it can filter the input power supply, and can also filter and attenuate the internal clutter reflection to the input terminal.
  • the input filter at the three-phase three-wire input terminal of the embodiment of the present invention is an EMI filter, and setting the EMI filter can also effectively control the EMI signal generated by the equipment itself, preventing it from entering the power grid, polluting the electromagnetic environment, and endangering
  • Other devices may also be other types of filters in other alternative embodiments of the embodiments of the present invention.
  • the output side of the EMI filter is respectively connected to the first switch rectification bridge arm KB1, the second switch rectification bridge arm KB2 and the third switch rectification bridge arm KB3, and each switch rectification bridge arm includes an AC input port 1 and a positive output port 2 and the negative output port 3, an equivalent controllable selection switch is provided between the AC input port 1, the positive output port 2 and the negative output port 3, and the equivalent controllable selection switch can be a combination of a diode and a high-frequency switching tube
  • the series combination can also be that two high-frequency switching tubes are connected in reverse series and then connected with a diode.
  • the equivalent controllable selector switch can apply a high-frequency PWM drive signal to the high-frequency switch tube according to the needs of AC rectification and conduction, and control the turn-on and turn-off, so as to realize the direction-selective conduction connection, that is, to form a pair of AC positive Half-wave high-frequency pulse rectification conduction, or AC negative half-wave high-frequency pulse rectification conduction.
  • the equivalent controllable selection switch is composed of a switch tube and four diodes, or is composed of two switch tubes and two diodes; when the equivalent controllable selection switch is composed of a switch tube When composed of four diodes, the source of the switching tube is connected to the anodes of the first diode D1 and the second diode D2, and the source of the switching tube is connected to the third diode D3 and the fourth and second diodes
  • the cathode of the pole tube D4 is connected; the cathode of the first diode D1 is connected to the positive output port 2; the cathode of the second diode D2 and the anode of the third diode D3 are connected to the AC input
  • the port 1 is connected; the anode of the fourth diode D4 is connected to the negative output port 3 .
  • the first connection mode is that after the first switching tube Q1 and the eleventh diode D11 are connected in series to form a first branch, one end is connected to the AC input port 1, the other end is connected to the positive output port 2, and the second end is connected to the positive output port 2.
  • the switching tube Q2 and the twelfth diode D12 are connected in series to form a second branch, one end is connected to the AC input port 1, and the other end is connected to the negative output port 3, and the first branch is connected to the first branch.
  • the two branches are symmetrical about the AC input port 1;
  • the second connection mode is that after the first switching tube Q1 and the second switching tube Q2 are connected in reverse series, the first switching tube Q1 is connected to the AC input port 1, and the second switching tube Q2 is connected to the eleventh diode D11.
  • the anode is connected to the cathode of the twelfth diode D12, the cathode of the eleventh diode D11 is connected to the positive output port 2, and the anode of the twelfth diode D12 is connected to the negative output port 3;
  • the third connection mode is that after the first switching tube Q1 and the second switching tube Q2 are connected in reverse series, the first switching tube Q1 is connected to the AC input port 1, and the anode of the eleventh diode D11 is connected to the first switching tube
  • the series point of Q1 and the second switching tube Q2 is connected, the cathode of the eleventh diode D11 is connected to the positive output port 2, the cathode of the twelfth diode D12 is connected to the second switching tube Q2, and the cathode of the twelfth diode D11 is connected to the second switching tube Q2.
  • the anode of the diode D12 is connected to the negative output port 3 .
  • the switch tubes in the input switch-type rectifier bridge arm group are high-frequency switch tubes provided with anti-parallel diodes, and the anti-parallel diodes are integrated diodes, parasitic diodes or external diodes;
  • the absorbing buffer capacitor Cs is non-polar A capacitor or a capacitor with polarity;
  • the first resonant inductance Lr1 is an external inductance, a coupling leakage inductance inside the transformer Tra, or a coupling inductance between an external inductance and an internal leakage inductance in the transformer Tra.
  • the absorbing buffer unit is mainly used to absorb the peak of the input switching rectifier bridge arm group due to the stray inductance of the circuit during operation, and also absorb the reverse freewheeling current and energy generated by the bridge resonant conversion unit in the working process to avoid Generate high voltage spikes, and also provide energy supply to the rear-end bridge resonant conversion unit.
  • the bridge-type resonant conversion unit and transformer Tra will also provide necessary energy storage when the input voltage is high in the same switching cycle and release energy when the input is switched to a low voltage. conduction, so as to realize the conduction of the bridge resonant conversion unit under different input voltages.
  • the switch tubes not only realize the rectification function, but also function as a boost energy storage switch.
  • the resonant frequency of the first resonant capacitor Cr1 and the first resonant inductance Lr1 Where lr1 is the inductance value of the first resonant inductor Lr1, and cr1 is the capacitance value of the first resonant capacitor Cr1.
  • S200 Analyze the instantaneous value of the voltage of each phase power supply in each of the interval segments according to the phase-locked phase in step S100;
  • S300 Detect the input and output conditions, judge whether the input and output conditions meet the working conditions required by the system, and continue to wait if the conditions are not met; if the conditions are met, the converter starts to work;
  • steps S300-S400 when the input switch-type rectifier bridge arm group and the fourth switch tube Q4 are in the PWM working state, the PWM switching frequency of the fourth switch tube Q4 and the control of the first switch rectifier bridge arm KB1 and the second switch rectifier
  • the PWM switching frequency of the bridge arm KB2 and the third switching rectifier bridge arm KB3 is consistent, and is twice the operating frequency of the bridge resonant conversion unit.
  • the driving duty ratios of the resonant conversion units are consistent and do not exceed 0.5, and the necessary dead time of the bridge inverter circuit is left;
  • the "high" mode PWM driving signal is an always-existing high-level signal or PWM driving signal , the driving voltage high level time of the "high” mode PWM driving signal is greater than the high level time of the "middle” mode PWM driving signal;
  • the bridge resonant conversion unit is a half bridge resonant conversion unit, the input
  • the "high" mode PWM driving signal of the switch-type rectifier bridge arm group can only be a PWM driving signal with a duty cycle not exceeding 50%.
  • the switching frequency of the fixed switching tube is near the resonant frequency; if the secondary conversion unit is a full-bridge converter, it can only PWM drive is applied to one of the switching tubes of the rectifying and conducting bridge arm in the off-cycle period, and PWM drive can also be added to both switches of the rectifying and conducting bridge arm in the off-cycle period; if the secondary conversion unit is a half-bridge rectifier When the conversion is performed, the PWM drive is only added to the non-rectified conduction switch in this period before the next rectification conduction period; when the inverter duty cycle of the bridge resonant conversion unit does not reach 47.5% or the maximum duty cycle In this mode, differential drive is applied to the two switching tubes that are turned on each time.
  • the driving signal of one switching tube is driven by the duty ratio calculated according to the control, and the other switching tube
  • the drive signal is a fixed maximum ratio drive. In this way, the state of the full bridge in the freewheeling working mode is changed, so that the direct current converter is closer to the freewheeling characteristic of the traditional step-down converter in this working state.
  • the switching tube driving signal applied to the bridge resonant conversion unit for boosting work is earlier than the driving signal of the secondary rectification output unit, and is applied to the bridge resonant conversion unit for boosting
  • the working switching tube drive signal is the delay of the synchronous rectification signal in the previous cycle, that is, the sum of the synchronous rectification duty cycle plus the boost duty cycle and the dead time; in the non-boost mode, the bridge resonant conversion unit and the secondary rectification output unit
  • the switching tube is applied with a synchronous drive signal.
  • the current conduction time of each phase is proportional to the instantaneous value of the phase voltage, and the current conduction time of the phase with the largest instantaneous value is equal to the sum of the current conduction time of the other two phases.
  • the output voltage can be stabilized or adjusted by adjusting the operating frequency of the embodiment of the present invention.
  • the maximum operating frequency does not exceed 1.5 times the resonance frequency.
  • the duty cycle is adjusted.
  • the soft switching or lower switching loss of the input switching rectifier bridge arm group can be realized by adjusting the sequence of the input switching rectifier bridge arm group, the fourth switching tube Q4 and the switching tubes of the bridge resonant conversion unit.
  • the method of judging the magnitude of the instantaneous value is to compare the absolute value of the instantaneous value of each phase.
  • the three-phase three-wire power supply includes A phase, B phase, and C phase.
  • the voltage signals of the three phase lines are 120 degrees out of phase with each other. Since the actual input power signal may have a jump or a sudden change in polarity, Therefore, the voltage waveform shown in this embodiment is for the convenience of the following description, and the standard waveform is used as a reference.
  • FIG. 6 is a schematic diagram of various implementation circuits of the input switching rectifier bridge arm group that can realize this function.
  • the AC input port 1 is connected to the cathode of the twelfth diode D12 and the anode of the thirteenth diode D13
  • the source of the first switching tube Q1 is connected to the twelfth diode D12 and the anode of the eleventh diode D11
  • the drain of the first switching tube Q1 is connected to the cathodes of the thirteenth diode D13 and the fourteenth diode D14
  • the cathode of the eleventh diode D11 is a positive output Port 2
  • the anode of the fourteenth diode D14 is the negative output port 3.
  • the AC input port 1 is connected to the cathode of the eleventh diode D11 and the anode of the twelfth diode D12, and the source of the second switching tube Q2 is connected to the twelfth diode D12
  • the drain of the second switching tube Q2 is the negative output port 3
  • the drain of the first switching tube Q1 is connected to the cathode of the eleventh diode D11
  • the source of the first switching tube Q1 is the positive output port 2.
  • the AC input port 1 is connected to the drain of the first switching transistor Q1 and the source of the second switching transistor Q2, and the source of the first switching transistor Q1 is connected to the anode of the eleventh diode D11 , the cathode of the eleventh diode D11 is the positive output port 2 , the drain of the second switching tube Q2 is connected to the cathode of the twelfth diode D12 , and the anode of the twelfth diode D12 is the negative output port 3 .
  • the AC input port 1 is connected to the drain of the first switching tube Q1
  • the source of the first switching tube Q1 is connected to the source of the second switching tube Q2
  • the drain of the second switching tube Q2 Connect with the anode of the eleventh diode D11 and the cathode of the twelfth diode D12, the cathode of the eleventh diode D11 is the positive output port 2, and the anode of the twelfth diode D12 is the negative output port 3.
  • the AC input port 1 is connected to the source of the first switching tube Q1, the drain of the first switching tube Q1 is connected to the drain of the second switching tube Q2, and the source of the second switching tube Q2 Connect with the anode of the eleventh diode D11 and the cathode of the twelfth diode D12, the cathode of the eleventh diode D11 is the positive output port 2, and the anode of the twelfth diode D12 is the negative output port 3.
  • the AC input port 1 is connected to the drain of the first switching transistor Q1, the source of the first switching transistor Q1 is connected to the source of the second switching transistor Q2, and the eleventh diode D11
  • the anode is connected to the source of the first switching tube Q1
  • the drain of the second switching tube Q2 is connected to the cathode of the twelfth diode D12
  • the cathode of the eleventh diode D11 is the positive output port 2
  • the anode of the diode D12 is the negative output port 3 .
  • the entire switch bridge arm can also be equivalent to a connection from the negative output port 3 to the positive output port 2
  • the two diodes are connected in series, but the connection midpoint of the two diodes will be clamped by the AC voltage of the AC input port 1.
  • the present invention is not limited to the connection method of the above-mentioned high-frequency switching tube and diode to realize the connection between the AC input port 1 of the rectifier bridge arm and the positive output port 2 and the negative output port 3 respectively, such as replacing the diode in the example with
  • the above-mentioned functions can also be realized by the switch tube, which will not be discussed one by one here, and other combinations that can realize the functions of the controllable selector switch of the present invention also belong to this category.
  • connection between the AC input port 1 and the positive output port 2 is equivalent to the connection between the anode of the eleventh diode D11 and the AC input port 1, and the cathode of the eleventh diode D11 Connect the positive output port 2, so it can be used for positive rectification; conversely, assuming that when the AC port is applied with an AC negative half-wave, when it is necessary to perform negative rectification pulse conduction control, give the first switch tube Q1 or Figure 6(c) and Figure 6(d) the second switch tube Q2 applies the open PWM signal, the corresponding switch tube is turned on, in Figure 6(b) between the AC input port 1 and the negative output port 3 is the tenth
  • the second diode D12 is connected in series with the fourteenth diode D14, which is equivalent to a diode whose cathode is connected to the AC input port 1 and whose anode is connected to the negative output port 3, so it can be used for negative rectification.
  • input A represents phase A input to PhaseA
  • input B represents phase B input to PhaseB
  • input C represents phase C input to PhaseC.
  • the three-phase voltages have a difference of 120° and are sinusoidal voltages, and each cycle is 360°; considering the intuitive and convenient expression, a complete cycle is taken from 30° to 390°, that is, the 30° point of the next cycle, so As shown in Figure 4, each intersection point is defined as AC(30°), BC(90°), BA(150°), CA(210°), CB(270°), AB(330°), AC( 30°/390°); the zero-crossing point is marked as "0".
  • the relationship between the instantaneous voltage value of each phase of the three-phase three-wire power supply is that the instantaneous voltage value of phase A > the instantaneous voltage value of phase B > the instantaneous voltage value of phase C, so for the first
  • the first switching rectification bridge arm KB1 and the third switching rectification bridge arm KB3 apply a "high" PWM driving signal
  • the second switching rectifying bridge arm KB2 applies a "middle" PWM driving signal.
  • the "KB1 positive” path connected to A is turned on, and the voltage is recorded as Va;
  • the "KB2 negative” path connected to B is turned on, and the voltage is recorded as Vb;
  • the output terminal of the "KB3 positive” path connected to C Due to being reverse-biased by the voltage Va and unable to conduct, the current of the A phase can flow through the fifth switching tube Q5, the first resonant capacitor Cr1, the first resonant inductor Lr1, the transformer Tra and the eighth switching tube Q8 through the "KB1 positive” path , and then return to the B-phase AC source through the "KB2 negative” path.
  • the input voltage "Vab” is applied to the input end of the bridge inverter circuit, except for the voltage drop on the first resonant capacitor Cr1 and the first resonant inductance Lr1, the rest are all applied to the primary side of the transformer Tra, and then through the transformer Tra
  • the equivalent turns ratio is transmitted to the secondary side, and the secondary side rectification circuit of the transformer is formed through the second diode D2, the filter capacitor C1, the external load, and the third diode D3. Therefore, the excess voltage is dropped on the first resonant capacitor Cr1 and the first resonant inductor Lr1.
  • Vab also charges the snubber capacitor Cs at the initial stage after the PWM is turned on, and after a period of time, the snubber capacitor Cs and Vab supply power to the bridge resonant conversion unit.
  • the current of the bridge resonant conversion unit cannot be diverted immediately, and the freewheeling flow is carried out through the sixth switching tube Q6 and the seventh switching tube Q7, and the bridge
  • the time for the current conversion process of the type resonant conversion unit to complete is the minimum dead time of the duty ratio of the switching tube of the full-bridge inverter circuit.
  • the resonant current passes through the anti-parallel diodes of the sixth switching tube Q6 and the fourth switching tube Q4, and then returns to the seventh switching tube Q7 through the absorbing buffer capacitor Cs to form a loop, and the energy is stably absorbed, and will not be affected by load and input voltage.
  • Figures 10 to 12 show the conduction process of Vab and Vcb after switching the sixth switch tube Q6 and the seventh switch tube Q7 of the full-bridge inverter circuit. Explain and describe.
  • the relationship between the instantaneous value of the voltage of each phase of the three-phase three-wire power supply is that the instantaneous value of the B-phase voltage > the instantaneous value of the A-phase voltage > the instantaneous value of the C-phase voltage, so for the second
  • the switching rectification bridge arm KB2 and the third switching rectification bridge arm KB3 apply a "high" PWM driving signal, and a "medium” PWM driving signal is applied to the first switching rectification bridge arm KB1.
  • Vab conduction After the above-mentioned Vab conduction is completed, it is transformed into Vac conduction, as shown in Figures 13 and 14, the "KB1 positive" path of the first switching rectification bridge arm KB1 and the "KB3 negative" path of the third switching rectification bridge arm KB3 Form a flow loop.
  • the circuits of the embodiments of the present invention in each working mode can be transformed and simplified.
  • the AC source can be equivalent to a DC source after being rectified by a diode, or the AC source plus a diode can be converted into a DC source in an instant.
  • the circuit can be regarded as a DC source, so the circuit of the embodiment of the present invention can be regarded as a full-bridge LLC series resonant converter whose input is a DC source after performing the above equivalent.
  • the current of each phase can be turned on, and it is in the same phase as the voltage, so there will be no current interruption of a certain phase in the uncontrolled rectification.
  • the conduction current of each phase of AC per unit time is proportional to the instantaneous value of the phase voltage, and the current conduction value of the phase with the largest instantaneous value is equal to the sum of the current conduction of the other two phases. That is, the current waveform and the voltage waveform can be made consistent, so a higher PF value can be obtained, that is, the PFC correction function can be realized.
  • the embodiment of the present invention realizes the similar principle of the series resonant circuit.
  • the embodiment of the present invention Through the single-stage circuit structure and control method, various functions are realized, many power components are saved, and soft switching is realized at the same time, the loss of the switch is small, and it is suitable for occasions with high power density.
  • the embodiment of the present invention can use the similar principle of the series resonant circuit to change the operating frequency, so as to realize the stable voltage output of the secondary rectification output unit under different loads and different input voltages. When the input and load conditions deviate from a certain frequency change range , and then adjust by changing the duty cycle to ensure the maximum soft switching operation.
  • the driving signal of the B phase is always a "high" PWM driving signal; in the BC-0 interval, the driving of the C phase is closed before the AB two phases, The driving signal of the AB two-phase is the "high” PWM driving signal, and the C-phase "KB3 negative” channel is the "middle” PWM driving signal; in the 0-BA interval, the driving of the A phase is closed before the BC two-phase, and the BC two-phase The drive signal of the A-phase "KB1 positive” channel is the "middle” PWM drive signal.
  • the driving signal of the A phase is always a "high” PWM driving signal
  • the driving of the B phase is closed before the C phase, and the driving signals of the AC two phases are "high” PWM driving Signal
  • B phase "KB2 positive” channel is “middle” PWM drive signal
  • the drive of C phase is closed before B phase, and the drive signal of AB two phases is “high” PWM drive signal
  • C The phase "KB3 negative” path is the "middle” PWM drive signal.
  • the driving signal of the C phase is a "high" PWM driving signal; in the CA-0 interval, the driving of the A phase is closed before the B phase, and the driving signal of the BC two phases is a "high” PWM driving signal , A-phase "KB1 negative” channel is “middle” PWM drive signal; in the 0-CB interval, B-phase drive is closed before A-phase, AC two-phase drive signal is “high” PWM drive signal, B-phase " KB2 Positive” channel is the "Middle” PWM drive signal.
  • the driving signal of the B phase is a "high" PWM driving signal
  • the driving of the C phase is closed before the A phase
  • the driving signal of the AB two phases is a "high” PWM driving signal
  • the C-phase "KB3 positive” channel is the "middle” PWM drive signal
  • the A-phase drive is closed before the C-phase
  • the BC two-phase drive signal is a "high” PWM drive signal
  • the A-phase " KB1 negative" path is the "middle” PWM drive signal.
  • the driving signal of the A phase is a "high” PWM driving signal
  • the driving of the B phase is closed before the C phase
  • the driving signal of the AC two phases is a "high” PWM driving signal
  • the B-phase "KB2 negative” channel is the "middle” PWM driving signal
  • the driving of the C-phase is closed before the B-phase
  • the driving signal of the AB two-phase is a "high” PWM driving signal
  • the C-phase " KB3 Positive” channel is the "Middle” PWM drive signal.
  • the instantaneous waveform of each AC voltage in the interval should be used Judging by characteristics, rather than expressing it from an ideal perspective. According to the characteristics of the three-phase power supply signal, it can be divided into twelve intervals. According to the above principles, the driving of the first switching rectification bridge arm KB1, the second switching rectification bridge arm KB2 and the third switching rectification bridge arm KB3 The waveform logic table of the signal is shown in Table 1.
  • Low mode means that according to the control method described above, the same driving signal as that of the switching rectifier bridge arm of the phase with the maximum instantaneous value can be applied, or at the latest before the driving signal of the switching tube of the other phase in the same direction as the instantaneous value is turned off Then apply the switching tube that is in phase with the maximum instantaneous value to form a driving signal for freewheeling, and the duty cycle is recorded as "high-medium". Therefore, considering the simplification and normalization of control, the "low” mode can apply the same driving signal as the "high” mode without affecting the realization of the function. At this time, Table 1 can be simplified into a logic table of the driving state of the switching tube as shown in Table 2.
  • Detect the input AC voltage judge whether the various indicators of the input voltage meet the working conditions, and continue to wait if the conditions are not met; if the conditions are met, start working, and analyze the phase-locking judgment of the input three-phase three-wire power supply voltage signal.
  • the phase and interval segment at the current moment analyze the absolute value of the instantaneous value of the voltage of each phase power supply; and judge the initial work of the embodiment of the present invention according to the absolute value of the instantaneous value of the input phase-to-phase voltage and the output voltage setting value frequency.
  • the switching frequency can be adjusted and updated according to the result of the control operation.
  • the time of input conduction current of each phase has a relative relationship with the instantaneous value of the phase voltage, that is, the higher the instantaneous value, the longer the current conduction time and the larger the duty cycle, and each phase of AC conducts in unit time
  • the magnitude of the current is proportional to the instantaneous value of the phase voltage, and the current conduction value of the phase with the largest instantaneous value is equal to the sum of the current conduction of the other two phases.
  • the relevant waveform drive is shown in Figure 15.
  • the secondary rectification output unit can be in various forms such as a full-bridge converter or a full-wave synchronous rectifier.
  • Figure 17(a) is the circuit connection mode of the full-bridge converter, which can be used as a full-bridge synchronous rectifier;
  • Figure 17(b) and Figure 17(c) are two different connection modes of the full-wave synchronous rectifier, and
  • Figure 17(b) adopts the common anode connection mode
  • Figure 17(c) adopts the common cathode connection mode.
  • a second resonant unit is connected in series between the secondary rectified output unit and the secondary side of the transformer Tra, and the second series resonant unit includes a second resonant capacitor Cr2 and a second resonant inductance Lr2 connected in series .
  • the secondary rectification output unit can be a full bridge rectification unit, or a voltage doubler rectification unit, that is, one of the bridge arms of the full bridge is replaced by a resonant capacitor.
  • Figure 18(a) is a full-bridge converter with full synchronous rectification function
  • Figure 18(b) and Figure 18(c) are two different connection examples of hybrid partial synchronous rectification full-bridge converter
  • Figure 18(b) The bridge arm is composed of the tenth switching tube Q10 and the twelfth switching tube Q12 connected in series, or the bridge arm is formed by connecting the ninth switching tube Q9 and the eleventh switching tube Q11 in series, and another rectifying bridge arm is formed by connecting diodes in series;
  • the ninth switching tube Q9 and the tenth switching tube Q10 are connected in a common drain mode, or the eleventh switching tube Q11 and the twelfth switching tube Q12 are connected in a common source mode;
  • FIG. 18(c) the ninth switching tube Q9 and the tenth switching tube Q10 are connected in a common drain mode, or the eleventh switching tube Q11 and the twelfth switching tube Q12 are connected in a common source mode;
  • 18( d) is a half-bridge voltage doubler rectifier, replacing one of the bridge arms of the full-bridge converter with a resonant capacitor, using the ninth switching tube Q9 and the eleventh switching tube Q11 to form a bridge arm, and then resonating with the resonant capacitor Cr2a
  • the resonant capacitor Cr2a and the resonant capacitor Cr2b also form the second series resonant unit together with the second resonant inductance Lr2, and the second resonant inductance Lr2 is an external inductor, a coupling leakage inductance inside the transformer Tra or an external
  • Fig. 17 and Fig. 18 are well-known circuits, and those skilled in the art should understand the specific working principles, so this article will not further analyze them.
  • the present invention is not limited to the above implementation examples, and other combinations that can realize the functions of the present invention also belong to this category.
  • Fig. 19 is a specific connection diagram of another embodiment of the present invention.
  • a second resonance unit is connected in series between the secondary rectification output unit and the secondary side of the transformer Tra.
  • the secondary rectification output unit is a full-bridge rectification unit.
  • the electromotive force of the first resonant inductance Lr1 and the excitation inductance Lm will change because the current of the resonant circuit cannot change suddenly, and the current of the primary side of the transformer Tra will pass through the current that was not turned on before.
  • the anti-parallel diode of the switching tube performs freewheeling. If the duty cycle of the switch tubes is all turned off at this time, the freewheeling current will reversely charge the snubber capacitor Cs, and at the same time, the secondary side of the transformer Tra will also store the original energy in the second resonant inductor Lr2 and the excitation voltage due to the coupling voltage of the transformer Tra.
  • the difference between the two is that the latter method is closer to the traditional freewheeling characteristics of the buck converter.
  • the output voltage is relatively low, mainly because the effective duty ratio loss caused by the volt-second balance characteristics of the transformer Tra and the inductor during the freewheeling period;
  • the magnetic reset process to the filter capacitor C1 is faster, so after the next turn-on, the effective transfer time of the current is longer, and the return current transfers more energy to the secondary side, and the loss is also greater.
  • the bridge resonant conversion unit needs to work in the step-down mode, it is necessary to preferentially apply differential drive to the two switching tubes that are turned on each time, and the driving signal of one of the switching tubes is based on Control the calculated duty cycle drive, and the drive signal of the other switching tube is a fixed maximum ratio drive or a new high drive with a duty cycle of 47.5%, which can be applied to the full bridge common source or common source this time.
  • the duty cycle of the drain or the two switch tubes directly connected in series with the bridge arm is fixed at 47.5% or close to the maximum duty cycle.
  • the driving duty ratio is in the invalid duty ratio interval of the primary side of the corresponding transformer Tra
  • the drive of the secondary rectifier unit is only used for synchronous rectification before the current turns positive, that is, the step-up drive corresponds to the transformer Tra is the driving starting point of the primary side, and there is a short period of ineffective boost duty cycle.
  • the driving duty cycle of the fifth to eighth switching tubes Q5-Q8 in the bridge resonant conversion unit is applied to the maximum, and one of the switching tubes of the rectifying and conducting bridge arm in the secondary rectifying unit is increased in PWM driving, That is, only the boost PWM drive is added to the eleventh switch tube Q11 or the twelfth switch tube Q12 that is non-rectified and turned on in this period, or the synchronous rectification drive is additionally added.
  • the ninth switching tube Q9 and the tenth switching tube Q10 can be used as synchronous rectification or as a diode; at the same time, the driving signal applied by the secondary rectification unit is earlier than the driving signal of the bridge resonant conversion unit, generally at least 2% to 5% earlier than the period , in the embodiment of the present invention, the driving signal of the secondary rectification unit is 200 ns earlier than the driving signal of the bridge resonant conversion unit.
  • the boost drive can also be understood as a continuation of the PWM drive with synchronous rectification function, and the value is equal to the sum of synchronous rectification duty cycle plus boost duty cycle and dead time;
  • the port voltage reverses with the electromotive force of the inductor and the induced voltage of the transformer Tra, it enters into a short-circuit state on the output side, that is, the rectified voltage that should be added to the input port of the bridge resonant conversion unit is at the tenth
  • a path backflow is formed on the first switch tube Q11 and the twelfth switch tube Q12, because the voltage of the first resonant capacitor Cr1 of the first series resonant unit and the second resonant capacitor Cr2 of the second series resonant unit cannot change abruptly, and the transformer Tra port
  • the voltage is directly coupled, so it is equal to applying energy storage to the first series resonant inductance and the second series rectification
  • the bridge resonant conversion unit may also be a half-bridge resonant conversion unit, as shown in FIG. 20 .
  • the switch of the input switch-type rectifier bridge arm group can be reused as a bridge arm switch of the half-bridge resonant conversion unit, and the series resonant unit and the transformer Tra circuit They are respectively connected to the positive output port 2 and the negative output port 3 of the switching rectification bridge arm, the drain of the other switching tube of the half-bridge resonant conversion unit is connected to the positive output port 2, and the source of the switching tube is connected to the negative output port 3.
  • the upper half-bridge arm of the half-bridge resonant conversion unit in Figure 20 can also be replaced by an input controllable switch, thereby simplifying the figure 21.
  • the "high" PWM drive signal in the aforementioned full-bridge resonant conversion unit needs to be
  • the PWM signal adjusted to a duty cycle of no more than 50% cannot be long and high, and forms a complementary drive with the lower half-bridge arm of the half-bridge resonant conversion unit to avoid a direct short circuit.
  • the working principle of the half-bridge resonant conversion unit is the same as that of the full-bridge resonant conversion unit, and those skilled in the art can know it from the aforementioned principles, and will not repeat them here.

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Abstract

提供一种无输入储能电感隔离谐振软开关型三相PFC变换器及其控制方法，它包括输入开关型整流桥臂组、吸收缓冲单元、桥式谐振变换单元、变压器和次级整流输出单元；通过对输入开关型整流桥臂组中的开关管施加"中"、"高"模式PWM驱动信号，同时对桥式谐振变换单元施加LLC全桥PWM驱动信号，其中LLC全桥PWM驱动信号的频率是输入开关型整流桥臂组中的开关管的驱动信号的频率的二分之一，实现三相无输入储能电感隔离谐振软开关整流变换及功率因数矫正；方案结构简单，实现了交直流转换功能、功率因素校正功能、隔离变换、软开关以及调压功能，降低了功率器件的开关损耗，适合于高功率密度场合。

Description

一种无输入储能电感隔离谐振软开关型三相PFC变换器及其控制方法 技术领域
本申请涉及交直流变换器技术领域，具体涉及一种无输入储能电感隔离谐振软开关型三相PFC变换器及其控制方法。
背景技术
近年来单体设备用电负荷的容量越来越大，大多是采用三相供电，比如电动汽车充电站等，假如没有PFC矫正功能就会对电网的电能质量产生很大破坏，严重时甚至会导致电网的瘫痪。随着国家对电能质量法规的要求越来越严格，目前较大功率的AC/DC电源都必须采用PFC(功率因素校正)电路，包括升压型(Boost)和降压型(Buck)两种，为获得较为稳定和安全的的输出电压，一般都需要的在PFC变换器后级均需要增加一级隔离型直流变换器，所以长期以来，对于三相交流输入的交直流变换电路，一般为PFC+DC/DC两级电路。由于两级电路的多次变换，导致开关损耗及导通损耗较多，效率下降严重。因此，国外有相关学者提出过单级的三相交直流变换器，如文献1所提出的ZVSZCS变换器，某种程度实现了三相单级隔离软开关变换，同时为了解决变压器偏磁饱和等问题尝试了两种驱动方案和电路，但是第二种电路显得较为复杂以及在输入电压不同的区间内输入侧电路会面临较高的杂散电感引起的电压应力；文献2作者提出的一种单级隔离电路，由于是单向性变压器励磁，输入电压的导通角度和导通时间有限，因此对输入电流的滤波会有较高要求，对降低谐波等受到限制；文献3作者SilvaM等提出了一种谐振式的Swiss隔离变换器和一种桥式隔离变换器，但是其认为桥式变换器(如图1所示)较为复杂，而重点介绍了谐振式的正激隔离变换器，该变换器适用的功率较小，且电路中的开关管会因谐振复位承受较高电压应力。文献4中，SisiZhao等提出了一种改进型的ViennarectifierⅢ，但是仍然没有解决变压器励磁对称性的问题，且由于输入电压的变化以及负载的变化，导致桥式变换电路并不能保证一定实现零电压零电流开通关断，输入整流器相关部分会因为电路的杂散电感需要增加较多的吸收处理。前述文献中的方案以及其他作者所提出的软开关式矩阵式变换器，都是在比较理想输入条件 或者输出条件下配合其控制方法；但是由于实际设备的三相输入电压源并非理想，会有各种不同的瞬态电能质量问题，比如有电网暂降，突波，频率跳变等各种各样的实际工况，因此该拓扑装置对于电网的条件会太过敏感，或者说电网适应性较差，从而导致可靠性较差，无法大规模化量化生产，这也是至今不见该技术大范围推广应用到产品的原因。
参考文献：
1、K.Wang，F.Lee，D.Boroyevich，and X.Yan，“A new quasi-single-stage isolated three-phase ZVZCS buck PWM rectifier”，in Proc.of 27th Annual IEEE Power Electronics Specialists Conference(PESC)，1996，pp.449–455。
2、D.S.Greff and I.Barbi,，“A single-stage high-frequency isolated three-phase ac/dc converter”，in Proc.32nd IEEE Ind.Electron.Soc.Conf.，Nov.6–10，2006,pp.2648–2653。
3、Silva M，Hensgens N，Oliver J，et al.Isolated Swiss-Forward Three-Phase Rectifier with Resonant Reset[J].IEEE Transactions on Power Electronics，2015，31(7):4795-4808。
4、Sisi Zhao，Uros Borovic，Marcelo Silva，Oscar Garcia，Predrag Pejovic，“Modified VIENNA Rectifier III to Achieve ZVS in All Transitions:Analysis，Design and Validation”，IEEE Transactions on Power Electronics 2021/05/26。
发明内容
本发明的目的在于，提供一种无输入储能电感隔离谐振软开关型三相PFC变换器及其控制方法，可以解决上述现有技术中存在的因变换器结构及控制方法复杂导致应用受限的技术问题。
本发明采取的技术方案是：一种无输入储能电感隔离谐振软开关型三相PFC变换器，包括输入开关型整流桥臂组、吸收缓冲单元、桥式谐振变换单元、变压器和次级整流输出单元；所述输入开关型整流桥臂组包括第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂，每个开关整流桥臂均包括交流输入端口、正输出端口和负输出端口，所述交流输入端口、正输出端口和负输出端口之间设置有等效可控选择开关；所述吸收缓冲单元包括串联连接的第四开关管和吸收缓冲电容；所述第四开关管的源极分别与第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的正输出端口连接，所述吸收缓冲电容分别与第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的负输出端口连接桥式谐振变换单元；所述桥式谐振 变换单元包括桥式逆变电路和第一串联谐振单元，所述桥式逆变电路为全桥式逆变电路或半桥式逆变电路，所述第一串联谐振单元包括串联连接的第一谐振电容和第一谐振电感，所述桥式逆变电路的输入端与所述吸收缓冲单元连接，输出端与第一谐振电容和变压器初级侧连接，所述变压器初级侧还与所述第一谐振电感连接；所述变压器次级侧与所述次级整流输出单元连接；所述次级整流输出单元包括整流电路和滤波电容，所述整流电路的输入端与所述变压器的次级侧连接，输出端与滤波电容连接，所述整流电路为倍压整流电路、全波整流电路或全桥整流电路，所述整流电路由全二极管组成、带有反并联二极管的开关管或具备整流功能的其他形式开关管中的其中一种组成，或者由开关管和二极管混合组成。
进一步地，所述等效可控选择开关为二极管与高频开关管的串联组合或者两个高频开关管反向串联后再与二极管连接，所述等效可控选择开关根据交流整流导通的需要对高频开关管施加高频PWM驱动信号可以控制开通与关断从而实现有方向选择性的导通连接，即形成对交流正半波的高频脉冲式整流导通，或者交流负半波高频脉冲式整流导通。
进一步地，所述等效可控选择开关由一个开关管和四个二极管组成，或者由两个开关管和两个二极管组成；
当所述等效可控选择开关由一个开关管和四个二极管组成时，所述开关管的源极与第十一二极管和第十二二极管的阳极连接，所述开关管的源极与第三二极管和第四二极管的阴极连接；所述第十一二极管的阴极与所述正输出端口连接；所述第十二二极管的阴极和第三二极管的阳极与所述交流输入端口连接；所述第四二极管的阳极与所述负输出端口连接；
当所述等效可控选择开关由两个开关管和两个二极管组成时，具有三种连接方式：
第一种连接方式为第一开关管和第十一二极管串联成第一支路后，一端与所述交流输入端口连接，另一端与所述正输出端口连接，第二开关管和第十二二极管串联成第二支路后，一端与所述交流输入端口连接，另一端与所述负输出端口连接，所述第一支路与所述第二支路关于所述交流输入端口对称；
第二种连接方式为第一开关管和第二开关管反向串联后，第一开关管与所述交流输入端口连接，第二开关管与第十一二极管的阳极和第十二二极管的阴极连接，第十一二极管的阴极与所述正输出端口连接，第十二二极管的阳极与所述负输出端口连接；
第三种连接方式为第一开关管和第二开关管反向串联后，第一开关管与所述交流输入端口连接，第十一二极管的阳极与第一开关管和第二开关管的串联点连接，第十一二极管的阴极与所述正输出端口连接，第十二二极管的阴极与第二开关管连接，第十二二极管的阳极与 所述负输出端口连接。
进一步地，所述输入开关型整流桥臂组中的开关管为设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管；所述吸收缓冲电容为无极性的电容或有极性的电容；所述第一谐振电感为外置式电感、变压器内部的耦合漏感或者外置电感和变压器内部漏感的耦合电感。
进一步地，还包括第二串联谐振单元，所述第二串联谐振单元包括串联连接的第二谐振电容和第二谐振电感，所述第一谐振电容和第一谐振电感的的谐振频率
Figure PCTCN2022117893-appb-000001
其中lr1为第一谐振电感的电感值，cr1为第一谐振电容的电容值；所述第二谐振电容和第二谐振电感的的谐振频率
Figure PCTCN2022117893-appb-000002
其中lr2为第二谐振电感的电感值，cr2为第二谐振电容的电容值，并且f1 0＝f2 0。
进一步地，还包括输入滤波器，所述输入滤波器与所述第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的交流输入端口连接。
上述无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法为：
S100：根据输入的三相三线电源电压信号的锁相分析各相电源当前时刻所处的相位和区间段；
S200：根据步骤S100中的锁相相位分析出各个所述区间段中各相电源的电压的瞬时值大小；
S300：检测输入输出条件，判断输入输出条件是否满足系统需求的工作条件，不满足条件继续等待；如若满足条件，变换器开始工作；
S400：对当前区间段下的输入开关型整流桥臂组施加驱动信号进行PWM驱动控制使其中瞬时值较高的两相电流先导通；随之施加PWM驱动信号使第四开关管导通，然后将已导通的瞬时值次高相交流回路上的开关管通路及第四开关管关断，让瞬时值最高相和瞬时值最低相的电流继续导通；具体方法为：给瞬时值最高与最低的两相交流回路中对应开关管同时施加相同占空比大小的“高”模式PWM驱动信号，同时对幅值瞬时值次高的电流回路中对应的开关管施加“中”模式PWM驱动信号，使在各个区间段中，施加“高”模式PWM驱动信号的开关管后关断，施加“中”模式PWM驱动信号的开关管先关断；使各相电流在每个开关周期内都能够导通；同时对第四开关管施加延迟于“中”模式开通和同时关断的PWM驱动信号。
进一步地，在步骤S300～S400中，当输入开关型整流桥臂组和第四开关管处于PWM工作状态时，第四开关管的PWM开关频率与控制第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的PWM开关频率一致，并且是桥式谐振变换单元工作频率的两倍，第四开关管和桥式谐振变换单元定义的工作相对起点一致，桥式谐振变换单元的各驱动占空比一致且不超过0.5，并留有桥式逆变电路必要的死区时间；所述“高”模式PWM驱动信号为一直存在的高电平信号或PWM驱动信号，所述“高”模式PWM驱动信号的驱动电压高电平时间大于所述“中”模式PWM驱动信号高电平时间；当桥式谐振变换单元为半桥式谐振变换单元时，施加给输入开关型整流桥臂组的“高”模式PWM驱动信号为只能是不超过50％占空比的PWM驱动信号。
进一步地，当次级整流单元串联有第二谐振单元时，在步骤S300～S400中，若根据变压器变比计算需要升压，且桥式谐振变换单元的逆变占空比以及达到47.5％或者最大占空比时，变换器工作在升压模式，在该模式下，固定开关管的开关频率在谐振频率附近；若次级变换单元是全桥变换器，可以在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，也可以对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；若次级变换单元是半桥整流式变换时，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管增加PWM驱动；当桥式谐振变换单元的逆变占空比没有达到47.5％或者最大占空比时，则是降压模式，在该模式下，对每次导通的两个开关管施加差异化驱动，一个开关管的驱动信号为根据控制计算得出的占空比驱动，另外一个开关管驱动信号为固定的最大占比驱动。
进一步地，在步骤S300～S400中，若是升压模式，施加给桥式谐振变换单元做升压工作的开关管驱动信号要早于次级整流输出单元的驱动信号，施加给桥式谐振变换单元做升压工作的开关管驱动信号为上周期同步整流信号的延迟，即同步整流占空比加升压占空比及死区时间的和；非升压模式下桥式谐振变换单元和次级整流输出单元的开关管则施加同步驱动信号。
进一步地，每相导通电流的时间与相电压的瞬时值成正比关系，瞬时值最大相的电流导通时间等于其它两相电流导通时间的总和。
本发明的有益效果在于：
(1)从结构上，本发明改变了传统AC/DC变换器需要PFC稳压电路加直流隔离变换电路的实现方法，通过本发明的拓扑结构，可以节省常规交直流变换器的交流整流后的储能单 元，例如大电感和大电容；
(2)从器件占用体积上，由于整流储能单元的减少，大大减少器件占用面积，整个交直流变换电路简单，控制逻辑精简，效率高，适合于高效率及高功率密度需求场合；
(3)从效率上来说，由于输入开关型整流桥臂组直接与后端的桥式谐振变换单元连接，对输入电流实现了谐振软开关化，因此，相对于传统的带有功率因素校正功能的隔离型三相交直流变换器而言，通过一级变换电路就实现了交直流转换功能、功率因素校正功能、隔离变换、软开关以及调压功能，效率提高；
(4)此外，相对其他单级变换器，由于吸收缓冲电路的介入，以及从拓扑形态上更加接近两级变换，所以本发明稳定性较好，能有效应对输入电压出现跳变或者极性突变的情况，其电网适应性更强，工作稳定性更高，设备质量更加可靠。
附图说明
为了更清楚地说明本发明实施例中的技术方案，下面将对实施例中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其它的附图。
图1是现有的交直流变换器方框示意图；
图2是现有的隔离型Swiss单级变换器结构示意图；
图3是本发明实施例的方框结构示意图；
图4是本发明实施例的三相电压波形示意及交汇点定义示意图；
图5是本发明实施例的具体连接示意图；
图6是本发明实施例中输入开关型整流桥臂组的具体实施例示意图；
图7是本发明实施例的AC-BC区间AB相导通回路示意图一；
图8是本发明实施例的AC-0区间BC相续流回路示意图一；
图9是本发明实施例的AC-0区间桥式谐振变换单元电流续流回路示意图一；
图10是本发明实施例的AC-BC区间AB相导通回路示意图二；
图11是本发明实施例的AC-0区间BC相续流回路示意图二；
图12是本发明实施例的AC-0区间桥式谐振变换单元电流续流回路示意图二；
图13是本发明实施例的0-BC区间AC相续流回路示意图一；
图14是本发明实施例的0-BC区间AC相续流回路示意图二；
图15是本发明实施例的AC-0区间某一时刻的具体实施关键波形示意图；
图16是本发明实施例工频周期内各开关整流桥臂驱动波形关系示意；
图17是本发明实施例的次级整流输出单元的连接具体实施示意图1；
图18是本发明实施例的次级整流输出单元的连接具体实施示意图2；
图19是本发明另一实施例的具体连接示意图；
图20是本发明实施例的桥式谐振变换单元为半桥谐振变换单元具体实施示意图1；
图21是本发明实施例的桥式谐振变换单元为半桥谐振变换单元具体实施示意图2。
附图标记解释：KB1-第一开关整流桥臂，KB2-第二开关整流桥臂，KB3-第三开关整流桥臂，D1-第一二极管，D2-第二二极管，D3-第三二极管，D4-第四二极管，D11-第十一二极管，D12-第十二二极管，D13-第十三二极管，D14-第十四二极管，Q1-第一开关管，Q2-第二开关管，Q4-第四开关管，Q5-第五开关管，Q6-第六开关管，Q7-第七开关管，Q8-第八开关管，Q9-第九开关管，Q10-第十开关管，Q11-第十一开关管，Q12-第十二开关管，Lr1-第一谐振电感，Lr2-第二谐振电感，Lm-励磁电感，Tra-变压器，Cr1-第一谐振电容，Cr2-第二谐振电容，Cs-吸收缓冲电容，C1-滤波电容，PhaseA-A相输入，Phase B-B相输入，Phase C-C相输入，1-交流输入端口，2-正输出端口，3-负输出端口。
具体实施方式
为了能够更清楚地理解本发明的上述目的、特征和优点，下面结合附图和具体实施方式对本发明进行进一步的详细描述。在下面的描述中阐述了很多具体细节以便于充分理解本发明，但是，本发明还可以采用其他不同于在此描述的其他方式来实施，因此，本发明并不限于下面公开的具体实施例的限制。
除非另作定义，此处使用的技术术语或者科学术语应当为本申请所述领域内具有一般技能的人士所理解的通常意义。本专利申请说明书以及权利要求书中使用的“第一”、“第二”以及类似的词语并不表示任何顺序、数量或者重要性，而只是用来区分不同的组成部分。同样，“一个”或者“一”等类似词语也不表示数量限制，而是表示存在至少一个。“连接”或者“相连”等类似的词语并非限定于物理的或者机械的连接，而是可以包括电性的连接，不管是直接的还是间接的。“上”、“下”、“左”、“右”等仅用于表示相对位置关系，当被描述对象的绝对位置改变后，则该相对位置关系也相应地改变。
如图3所示，一种无输入储能电感隔离谐振软开关型三相PFC变换器，包括输入开关型 整流桥臂组、吸收缓冲单元、桥式谐振变换单元、变压器Tra和次级整流输出单元；所述输入开关型整流桥臂组包括第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3，每个开关整流桥臂均包括交流输入端口1、正输出端口2和负输出端口3，所述交流输入端口1、正输出端口2和负输出端口3之间设置有等效可控选择开关；所述吸收缓冲单元包括串联连接的第四开关管Q4和吸收缓冲电容Cs；所述第四开关管Q4的源极分别与第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3的正输出端口2连接，所述吸收缓冲电容Cs分别与第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3的负输出端口3连接桥式谐振变换单元；所述桥式谐振变换单元包括桥式逆变电路和第一串联谐振单元，所述桥式逆变电路为全桥式逆变电路或半桥式逆变电路，所述第一串联谐振单元包括串联连接的第一谐振电容Cr1和第一谐振电感Lr1，所述桥式逆变电路的输入端与所述吸收缓冲单元连接，输出端与第一谐振电容Cr1和变压器Tra初级侧连接，所述变压器Tra初级侧还与所述第一谐振电感Lr1连接；所述变压器Tra次级侧与所述次级整流输出单元连接；所述次级整流输出单元包括整流电路和滤波电容C1，所述整流电路的输入端与所述变压器Tra的次级侧连接，输出端与滤波电容C1连接，所述整流电路为倍压整流电路、全波整流电路或全桥整流电路，所述整流电路由全二极管组成、带有反并联二极管的开关管或具备整流功能的其他形式开关管中的其中一种组成，或者由开关管和二极管混合组成。
当所述桥式谐振变换单元为全桥谐振变换单元，次级整流输出单元的整流电路为由全二极管组成全桥整流电路时，本发明实施例的具体连接电路如图5所示。所述全桥谐振变换单元包括第五开关管Q5、第六开关管Q6、第七开关管Q7和第八开关管Q8，所述第五开关管Q5和第七开关管Q7串联成第一桥臂，所述第六开关管Q6和第八开关管Q8串联成第二桥臂，所述第一桥臂和第二桥臂并联连接；第五开关管Q5和第六开关管Q6的漏极与第四开关的源极连接，第七开关管Q7和第八开关管Q8的源极与吸收缓冲电容Cs连接；第五开关管Q5的源极与第一谐振电容Cr1连接，第六开关管Q6的源极与变压器Tra的初级侧连接。所述整流电路包括第一二极管D1、第二二极管D2、第三二极管D3和第四二极管D4，所述第一二极管D1和第三二极管D3串联成第三桥臂，所述第二二极管D2和第四二极管D4串联成第四桥臂，所述第三桥臂和第四桥臂并联连接；第一二极管D1和第一二极管D1的阴极与滤波电容C1的一端连接，并构成本发明实施例输出电源的正端口；第三二极管D3和第四二极管D4的阳极与滤波电容C1的另一端连接，并构成本发明实施例输出电源的负端口。在图 5中，本发明实施例还包括输入滤波器，所述输入滤波器与所述第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3的交流输入端口1连接,对输入电源起滤波作用，同时也可以对内部的杂波反射至输入端起滤波和衰减作用。
如图5所示，本发明实施例在三相三线输入端的输入滤波器为EMI滤波器，设置EMI滤波器也能有效地控制设备本身产生的EMI信号，防止它进入电网，污染电磁环境，危害其他设备，在本发明实施例其他变通实施例中，还可以是其他类型的滤波器。
EMI滤波器的输出侧分别连接着第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3，每个开关整流桥臂均包括交流输入端口1、正输出端口2和负输出端口3，所述交流输入端口1、正输出端口2和负输出端口3之间设置有等效可控选择开关，所述等效可控选择开关可以是二极管与高频开关管的串联组合，也可以是两个高频开关管反向串联后再与二极管连接。所述等效可控选择开关可根据交流整流导通的需要对高频开关管施加高频PWM驱动信号，控制开通与关断，从而实现有方向选择性的导通连接，即形成对交流正半波的高频脉冲式整流导通，或者交流负半波高频脉冲式整流导通。
在本发明实施例中，所述等效可控选择开关由一个开关管和四个二极管组成，或者由两个开关管和两个二极管组成；当所述等效可控选择开关由一个开关管和四个二极管组成时，所述开关管的源极与第一二极管D1和第二二极管D2的阳极连接，所述开关管的源极与第三二极管D3和第四二极管D4的阴极连接；所述第一二极管D1的阴极与所述正输出端口2连接；所述第二二极管D2的阴极和第三二极管D3的阳极与所述交流输入端口1连接；所述第四二极管D4的阳极与所述负输出端口3连接。
当所述等效可控选择开关由两个开关管和两个二极管组成时，具有三种连接方式：
第一种连接方式为第一开关管Q1和第十一二极管D11串联成第一支路后，一端与所述交流输入端口1连接，另一端与所述正输出端口2连接，第二开关管Q2和第十二二极管D12串联成第二支路后，一端与所述交流输入端口1连接，另一端与所述负输出端口3连接，所述第一支路与所述第二支路关于所述交流输入端口1对称；
第二种连接方式为第一开关管Q1和第二开关管Q2反向串联后，第一开关管Q1与所述交流输入端口1连接，第二开关管Q2与第十一二极管D11的阳极和第十二二极管D12的阴极连接，第十一二极管D11的阴极与所述正输出端口2连接，第十二二极管D12的阳极与所述负输出端口3连接；
第三种连接方式为第一开关管Q1和第二开关管Q2反向串联后，第一开关管Q1与所述 交流输入端口1连接，第十一二极管D11的阳极与第一开关管Q1和第二开关管Q2的串联点连接，第十一二极管D11的阴极与所述正输出端口2连接，第十二二极管D12的阴极与第二开关管Q2连接，第十二二极管D12的阳极与所述负输出端口3连接。
所述输入开关型整流桥臂组中的开关管为设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管；所述吸收缓冲电容Cs为无极性的电容或有极性的电容；所述第一谐振电感Lr1为外置式电感、变压器Tra内部的耦合漏感或者外置电感和变压器Tra内部漏感的耦合电感。
吸收缓冲单元主要是用来吸收输入开关型整流桥臂组因电路杂散电感在运行中产生的尖峰，同时也吸收桥式谐振变换单元在工作过程中产生的反向续流电流及能量，避免产生高的电压尖峰，此外还给后端桥式谐振变换单元提供能量供给。所述桥式谐振变换单元及变压器Tra除完成谐振变换的常见基本功能外，还会在同一开关周期内高输入电压时候提供必要的储能并在输入切换为低电压的时候释能续流导通，从而实现桥式谐振变换单元不同输入电压下的导通。当次级整流输出单元的整流器件全部为同步整流开关管或者由开关管和二极管混合整流时，开关管除了实现整流功能，还兼具升压储能开关作用。所述第一谐振电容Cr1和第一谐振电感Lr1的的谐振频率
Figure PCTCN2022117893-appb-000003
其中lr1为第一谐振电感Lr1的电感值，cr1为第一谐振电容Cr1的电容值。
上述无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法为：
S100：根据输入的三相三线电源电压信号的锁相分析各相电源当前时刻所处的相位和区间段；
S200：根据步骤S100中的锁相相位分析出各个所述区间段中各相电源的电压的瞬时值大小；
S300：检测输入输出条件，判断输入输出条件是否满足系统需求的工作条件，不满足条件继续等待；如若满足条件，变换器开始工作；
S400：对当前区间段下的输入开关型整流桥臂组施加驱动信号进行PWM驱动控制使其中瞬时值较高的两相电流先导通；随之桥式谐振变换单元施加PWM驱动信号使第四开关管Q4导通，然后将已导通的瞬时值次高相交流回路上的开关管通路及第四开关管Q4关断，让瞬时值最高相和瞬时值最低相的电流继续导通；具体方法为：给瞬时值最高与最低的两相交流回路中对应开关管同时施加相同占空比大小的“高”模式PWM驱动信号，同时对幅值瞬时值次高的电流回路中对应的开关管施加“中”模式PWM驱动信号，使在各个区间段中， 施加“高”模式PWM驱动信号的开关管后关断，施加“中”模式PWM驱动信号的开关管先关断；使各相电流在每个开关周期内都能够导通；同时对第四开关管Q4施加延迟于“中”模式开通和同时关断的PWM驱动信号。
在步骤S300～S400中，当输入开关型整流桥臂组和第四开关管Q4处于PWM工作状态时，第四开关管Q4的PWM开关频率与控制第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3的PWM开关频率一致，并且是桥式谐振变换单元工作频率的两倍，第四开关管Q4和桥式谐振变换单元定义的工作相对起点一致，桥式谐振变换单元的各驱动占空比一致且不超过0.5，并留有桥式逆变电路必要的死区时间；所述“高”模式PWM驱动信号为一直存在的高电平信号或PWM驱动信号，所述“高”模式PWM驱动信号的驱动电压高电平时间大于所述“中”模式PWM驱动信号高电平时间；当桥式谐振变换单元为半桥式谐振变换单元时，施加给输入开关型整流桥臂组的“高”模式PWM驱动信号为只能是不超过50％占空比的PWM驱动信号。
当次级整流单元串联有第二谐振单元时，在步骤S300～S400中，若根据变压器Tra变比计算需要升压，且桥式谐振变换单元的逆变占空比以及达到47.5％或者最大占空比时，变换器工作在升压模式，在该模式下，固定开关管的开关频率在谐振频率附近；若次级变换单元是全桥变换器，可以在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，也可以对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；若次级变换单元是半桥整流式变换时，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管增加PWM驱动；当桥式谐振变换单元的逆变占空比没有达到47.5％或者最大占空比时，则是降压模式，在该模式下，对每次导通的两个开关管施加差异化驱动，一个开关管的驱动信号为根据控制计算得出的占空比驱动，另外一个开关管驱动信号为固定的最大占比驱动。以此改变全桥在续流工作模式下的状态，使该直流变换器在此工作状态下更趋近于传统的降压变换器的续流特性。
在步骤S300～S400中，若是升压模式，施加给桥式谐振变换单元做升压工作的开关管驱动信号要早于次级整流输出单元的驱动信号，施加给桥式谐振变换单元做升压工作的开关管驱动信号为上周期同步整流信号的延迟，即同步整流占空比加升压占空比及死区时间的和；非升压模式下桥式谐振变换单元和次级整流输出单元的开关管则施加同步驱动信号。
每相导通电流的时间与相电压的瞬时值成正比关系，瞬时值最大相的电流导通时间等于其它两相电流导通时间的总和。通过调整本发明实施例的工作频率实现对输出电压的稳定或 者调节，工作频率最高不超过谐振频率的1.5倍，当超过最高频率时，通过调节输入开关型整流桥臂组及吸收缓冲单元的驱动占空比进行调节。通过输入开关型整流桥臂组和第四开关管Q4及桥式谐振变换单元的开关管的时序配合调节，可以实现输入开关型整流桥臂组的软开关或者更低的开关损耗。
判断瞬时值大小的方法为比较各相瞬时值的绝对值大小。如图4所示，三相三线电源包括A相、B相及C相，三个相线的电压信号彼此相差120度的相位，由于实际输入的电源信号可能存在跳变或者极性突变的，所以本实施例示出的电压波形为了便于后文叙述，以标准的波形作为参考。
图6为输入开关型整流桥臂组的多种可实现该功能的具体实施电路示意图。如图6(b)所示，交流输入端口1连接第十二二极管D12的阴极及第十三二极管D13的阳极，第一开关管Q1的源极连接第十二二极管D12和第十一二极管D11的阳极，第一开关管Q1的漏极连接第十三二极管D13和第十四二极管D14的阴极，第十一二极管D11的阴极为正输出端口2，第十四二极管D14的阳极为负输出端口3。如图6(c)所示，交流输入端口1连接第十一二极管D11的阴极和第十二二极管D12的阳极，第二开关管Q2的源极连接第十二二极管D12的阳极，第二开关管Q2的漏极为负输出端口3，第一开关管Q1的漏极连接第十一二极管D11的阴极，第一开关管Q1的源极为正输出端口2。如图6(d)所示，交流输入端口1连接第一开关管Q1的漏极和第二开关管Q2的源极，第一开关管Q1的源极连接第十一二极管D11的阳极，第十一二极管D11的阴极为正输出端口2，第二开关管Q2的漏极连接第十二二极管D12的阴极，第十二二极管D12的阳极为负输出端口3。如图6(e)所示，交流输入端口1连接第一开关管Q1的漏极，第一开关管Q1的源极与第二开关管Q2的源极连接，第二开关管Q2的漏极与第十一二极管D11的阳极和第十二二极管D12的阴极连接，第十一二极管D11的阴极为正输出端口2，第十二二极管D12的阳极为负输出端口3。如图6(f)所示，交流输入端口1连接第一开关管Q1的源极，第一开关管Q1的漏极与第二开关管Q2的漏极连接，第二开关管Q2的源极与第十一二极管D11的阳极和第十二二极管D12的阴极连接，第十一二极管D11的阴极为正输出端口2，第十二二极管D12的阳极为负输出端口3。如图6(g)所示，交流输入端口1连接第一开关管Q1的漏极，第一开关管Q1的源极与第二开关管Q2的源极连接，第十一二极管D11的阳极与第一开关管Q1的源极连接，第二开 关管Q2的漏极与第十二二极管D12的阴极连接，第十一二极管D11的阴极为正输出端口2，第十二二极管D12的阳极为负输出端口3。
此外，图6(b)、图6(c)或图6(d)在开关管被施加开通驱动信号后，整个开关桥臂也可以等效为一个从负输出端口3连接到正输出端口2的两个二极管串联，只是两个二极管的连接中点会被交流输入端口1的交流电压所箝位。此外，本发明不局限于上述高频开关管及二极管的连接方法来实现整流桥臂的交流输入端口1分别与正输出端口2及负输出端口3之间的连接，如将示例中的二极管换为开关管亦可以实现上述的功能，此处不一一展开讨论，其他可实现本发明可控选择开关的功能的组合方式亦都属于本范畴。
假设当交流端口施加交流正半波，需要做正向整流脉冲导通控制时，给图6中的第一开关管Q1施加开通的PWM信号，则第一开关管Q1导通。图6(b)中交流输入端口1与正输出端口2之间则为第十三二极管D13与第十一二极管D11串联，等效为一个阳极连接交流输入端口1，阴极连接正输出端口2的二极管，因此可做正向整流。图6(c)和图6(d)中交流输入端口1与正输出端口2之间则等效为第十一二极管D11的阳极连接交流输入端口1，第十一二极管D11阴极连接正输出端口2，因此可做正向整流；反之，假设当交流端口施加交流负半波，需要做负向整流脉冲导通控制时，给图6(b)中的第一开关管Q1或图6(c)与图6(d)的第二开关管Q2施加开通的PWM信号，对应开关管导通，图6(b)中交流输入端口1与负输出端口3之间则为第十二二极管D12与第十四二极管D14串联，等效为一个阴极连接交流输入端口1，阳极连接负输出端口3的二极管，因此可做负向整流。图6(c)和图6(d)中交流输入端口1与负输出端口3之间则等效为第十二二极管D12的阳极连接负输出端口3，第十二二极管D12的阴极连接交流输入端口1，因此可做负向整流。当多个开关整流桥臂的输出端并联在一起时，因通路的二极管等效性质，二极管的电压偏置效应小，因此会优先最高电压正向导通或者最低电压负向导通，而另外通路的电压会因等效二极管被截止而无法导通。因此，后续的案例讨论中，皆以开关整流桥臂正向整流导通或者负向整流导通表示上述工作原理和通路,并将对应通路记为“KB正”或者“KB负”。
如图4所示，输入A代表A相输入PhaseA，输入B代表B相输入PhaseB，输入C代表C相输入PhaseC。为了方便描述，设三相电压相差120°，且为正弦电压，每360°一个循环；考虑到表述直观方便，以30°到390°，即下一周期的30°点为一个完整周期，因此如 图4所示，各交汇点分别定义为AC(30°)、BC(90°)、BA(150°)、CA(210°)、CB(270°)、AB(330°)、AC(30°/390°)；过零点标为“0”点。
假设根据输入电压及输出电压的判断以及控制计算，得出某一工作频率。如图7所示，从AC点开始到BC点的AC-BC区间内，A相与B相电压瞬时值的绝对值高于C相，根据前述的控制方法及KB导通原理，如果对第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3同时施加PWM驱动开通信号。由于在AC-0区间内，即30°～60°区间内，三相三线电源每相的电压瞬时值关系为A相电压瞬时值>B相电压瞬时值>C相电压瞬时值，所以对第一开关整流桥臂KB1和第三开关整流桥臂KB3施加“高”PWM驱动信号，第二开关整流桥臂KB2施加“中”PWM驱动信号。
因此与A相连接的“KB1正”通路导通，电压记为Va；与B相连接的“KB2负”通路导通，电压记为Vb；与C相连接的“KB3正”通路的输出端因被电压Va反偏而无法导通，A相的电流可经由“KB1正”通路流经第五开关管Q5、第一谐振电容Cr1、第一谐振电感Lr1、变压器Tra和第八开关管Q8，再经过“KB2负”通路回到B相交流源。此时，输入电压“Vab”施加在桥式逆变电路的输入端，除第一谐振电容Cr1和第一谐振电感Lr1上的压降外，其余全部加在变压器Tra初级侧，再经变压器Tra等效匝数比传递到次级侧，经过第二二极管D2、滤波电容C1和外接负载、第三二极管D3形成变压器次级侧整流回路。因此多余的电压被降压在第一谐振电容Cr1和第一谐振电感Lr1上。此外在施加PWM开通后初期，Vab也对吸收缓冲电容Cs充电，经过一段时间，吸收缓冲电容Cs与Vab一起对桥式谐振变换单元供电。
当“KB2正”通路及第四开关管Q4的驱动被关断后，此时由于回路中有第一谐振电感Lr1的存在，所以电流无法立即反向。同时“KB3正”通路的偏置电压Va消失，“KB3正”通路可以导通，但是由于Vc比Va瞬时值低，所以电感电动势会发生反向，第一谐振电感Lr1释能续流；如图8所示，此时，输入电压“Vcb”施加在桥式逆变电路的输入端。变压器Tra初级侧的输入电压则变为输入电压“Vcb”加第一串联谐振单元的电压，以此实现C相与B相电流的导通。
如图9所示，当第五开关管Q5和第八开关管Q8关断后，桥式谐振变换单元的电流无法立即转向，通过第六开关管Q6和第七开关管Q7进行续流，桥式谐振变换单元电流转换过程 完成的时间即为全桥式逆变电路开关管的占空比最小死区时间。谐振电流经第六开关管Q6和第四开关管Q4的反并二极管，再经吸收缓冲电容Cs回到第七开关管Q7，形成环路，能量被稳定吸收，不会因为负载和输入电压的改变而不稳定，从而改善了背景技术参考文献中的变换器因反向电流引起电压尖峰问题。图10～12则是切换全桥式逆变电路第六开关管Q6、第七开关管Q7后Vab、Vcb导通过程，原理与前述一致，本领域的技术人员应该理解，所以不再做详细解释和描述。
由于0-BC区间内，即60°～90°区间内，三相三线电源每相的电压瞬时值关系为B相电压瞬时值>A相电压瞬时值>C相电压瞬时值，所以对第二开关整流桥臂KB2和第三开关整流桥臂KB3施加“高”PWM驱动信号，对第一开关整流桥臂KB1施加“中”PWM驱动信号。在前述Vab导通完成后，则变换为Vac导通，如图13和14所示，第一开关整流桥臂KB1的“KB1正”通路与第三开关整流桥臂KB3的“KB3负”通路形成通流回路。
由以上可见，施加在变压器Tra上的电压是在一个周期内是对称的，因此不会出现背景技术中文献1和文献4所描述的非对称电压情况，同时也不会出现文献2和文献3中的变压器Tra单向励磁需要另外的磁复位电路。
此外根据以上工作分析，在变换过程中，只有交流次高相的开关型整流桥臂和第四开关管Q4跟随全桥谐振变换单元做高频切换，同时开关型整流桥臂所通过的电流也跟随全桥谐振变换单元，开关损耗相对较低。相关各主要波形参考如图15的仿真图。所以本发明实施例从输入到输出的各主器件都实现谐振软开关。
根据前述工作原理分析，可以对本发明实施例在各工作模式下的电路进行变换简化，在各通路瞬态情况下交流源通过二极管整流后可等效为直流源，或者说交流源加二极管在瞬时电路中可以视为直流源，因此本发明实施例的电路进行上述等效后，实际可以看作是一个输入为直流源的全桥LLC串联谐振变换器。
此外，在0-BC区间内，各相的电流都可以导通，并跟电压同相，不会出现不控整流中的某相断流现象。且通过调节工作频率，使每相交流在单位时间内导通电流大小与相电压的瞬时值成正比关系，瞬时值最大相的电流导通值等于其它两相电流导通的总和。即可以使得电流波形与电压波形跟随一致，因此可以获得较高的PF值，即实现PFC矫正功能。
由此可见，本发明实施例实现了串联谐振电路的类似原理，同时，由于输入开关型整流 桥臂组和桥式谐振变换单元又起到的功率因素校正功能，也就是说，本发明实施例通过单级电路结构和控制方法便实现了多种功能，省去了很多功率元件，同时又实现了软开关，开关的损耗较小，适合高功率密度的场合。本发明实施例可利用串联谐振电路的类似原理，改变工作频率，从而可以实现次级整流输出单元在不同负载以及不同输入电压下的稳压输出，当输入及负载条件偏离一定的频率改变范围后，再利用改变占空比的方式来进行调节，以保证最大程度的软开关工作。
对于其他区间段而言，以此类推，在BC-BA区间内，B相的驱动信号一直为“高”PWM驱动信号；在BC-0区间内，C相的驱动先于AB两相关闭，AB两相的驱动信号为“高”PWM驱动信号，C相“KB3负”通路为“中”PWM驱动信号；在0-BA区间内，A相的驱动先于BC两相关闭，BC两相的驱动信号为“高”PWM驱动信号，A相“KB1正”通路为“中”PWM驱动信号。
在BA-CA区间内，A相的驱动信号一直为“高”PWM驱动信号，在BA-0区间内，B相的驱动先于C相关闭，AC两相的驱动信号为“高”PWM驱动信号，；B相“KB2正”通路为“中”PWM驱动信号；在0-CA区间内，C相的驱动先于B相关闭，AB两相的驱动信号为“高”PWM驱动信号，C相“KB3负”通路为“中”PWM驱动信号。
在CA-CB区间内，C相的驱动信号为“高”PWM驱动信号；在CA-0区间内，A相的驱动先于B相关闭，BC两相的驱动信号为“高”PWM驱动信号，A相“KB1负”通路为“中”PWM驱动信号；在0-CB区间内，B相的驱动先于A相关闭，AC两相的驱动信号为“高”PWM驱动信号，B相“KB2正”通路为“中”PWM驱动信号。
在CB-AB区间内，B相的驱动信号为“高”PWM驱动信号；在CB-0区间内，C相的驱动先于A相关闭，AB两相的驱动信号为“高”PWM驱动信号，C相“KB3正”通路为“中”PWM驱动信号；在0-AB区间内，A相的驱动先于C相关闭，BC两相的驱动信号为“高”PWM驱动信号，A相“KB1负”通路为“中”PWM驱动信号。
在AB-AC区间内，A相的驱动信号为“高”PWM驱动信号；在AB-0区间内，B相的驱动先于C相关闭，AC两相的驱动信号为“高”PWM驱动信号，B相“KB2负”通路为“中”PWM驱动信号；在0-AC区间内，C相的驱动先于B相关闭，AB两相的驱动信号为“高”PWM驱动信号，C相“KB3正”通路为“中”PWM驱动信号。
由于现实中三相电压并不一定完全理想，存在相位、幅值、方向的变化，只能根据实际锁相来判断产生各区间段的驱动波形，因此应该以区间段各交流电压的瞬时波形的特征来判断，而不以理想角度来表示。根据三相电源信号的特点，可以分成十二个区间段，十二个区间段根据上述原理，第一开关整流桥臂KB1、第二开关整流桥臂KB2和第三开关整流桥臂KB3的驱动信号的波形逻辑表如表1所示。
表1开关管驱动状态逻辑表
Figure PCTCN2022117893-appb-000004
“低”模式表示根据前文所述的控制方法，可施加与最大瞬时值相的开关整流桥臂同样的驱动信号，或者最迟在瞬时值同方向的另外一相的开关管的驱动信号关闭前再施加与最大瞬时值相的开关管构成续流的驱动信号，占空比记为“高-中”。因此，考虑到控制的简化和归一化，在不影响功能实现的基础上，“低”模式均可以施加同“高”模式一致的驱动信号。此时表1可简化成如表2所示的开关管驱动状态逻辑表。
表2简化后的开关管驱动状态逻辑表
Figure PCTCN2022117893-appb-000005
根据表2所示的开关管驱动状态逻辑表，将一个控制周期总分为12个区间段，并执行如下控制方法：
检测输入交流电压，判断输入电压的各项指标是否满足工作条件，不满足条件继续等待；如若满足条件，则开始工作，根据输入的三相三线电源电压信号的锁相判断，分析各相电源的当前时刻所处的相位和区间段；分析出各相电源的电压的瞬时值的绝对值大小；并根据输入相间电压瞬时值的绝对值大小与输出电压设定值判断本发明实施例工作的初始频率。同时对连接瞬时值次高相交流的开关整流桥臂中对应的开关管施加“中”模式PWM驱动信号，给另外两相开关整流桥臂中对应的开关管施加相同的“高”模式PWM驱动信号，这样使瞬时值较高的两相电源构成电流通路，同时桥式谐振变换单元施加驱动一起开始工作，稍作延迟后对第四开关管Q4施加“中”模式PWM驱动信号；待施加在交流回路和第四开关管Q4的“中”模式PWM驱动信号关闭后，原施加“高”模式PWM驱动信号的另外两相的开关管会给谐振电流提供续流通路而继续导通；待桥式谐振变换单元换相后，再重复上述交流侧的驱动施加，待一个周期完毕后，可以根据控制运算的结果调整和更新开关频率。总体来说，各相输入导通电流的时间与相电压的瞬时值成相对关系，即瞬时值越高的，电流导通时间越久、占空比越大，每相交流在单位时间内导通电流大小与相电压的瞬时值成正比关系，瞬时值最大相的电流导通值等于其它两相电流导通的总和。相关波形驱动如图15所示。
通过上述控制方法，有效保证在每个开关周期内，三相均有电流流通，同时根据实时控制将PWM驱动信号占空比调制好，就可以使得电流波形与电压波形跟随一致，因此可以获得较高的PF值，即实现PFC矫正功能。在高功率密度场合，优势十分明显，可满足高精尖产品需要。
如图17所示，次级整流输出单元可以是全桥变换器或全波同步整流器等多种形式。如图17(a)是全桥变换器的电路连接方式，可做全桥同步整流器；图17(b)和图17(c)是全波同步整流器的两种不同连接方式，图17(b)中的第十一开关管Q11和第十二开关管Q12采用共源极方式连接，图17(c)中的第九开关管Q9和第十开关管Q10采用共漏极方式连接；当使用二极管整流代替开关管同步整流时，图17(b)采用共阳极连接方式，图17(c)采用共阴极连接方式。
如图18所示，在次级整流输出单元与变压器Tra次级侧之间串入了第二谐振单元，所述第二串联谐振单元包括串联连接的第二谐振电容Cr2和第二谐振电感Lr2。次级整流输出单 元可以是全桥式整流单元，也可以是倍压整流单元，即将全桥的其中一个桥臂用谐振电容代替。图18(a)是具备完全同步整流功能的全桥变换器；图18(b)和图18(c)是混合式部分同步整流全桥变换器的两个不同连接实例，图18(b)中由第十开关管Q10和第十二开关管Q12串联组成桥臂，或是由第九开关管Q9和第十一开关管Q11串联组成桥臂，并由二极管串联组成另外一个整流桥臂；图18(c)中由第九开关管Q9和第十开关管Q10采用共漏极方式连接，或由第十一开关管Q11和第十二开关管Q12采用共源极方式连接；图18(d)是半桥式倍压整流器，将全桥变换器的其中一个桥臂用谐振电容代替，采用第九开关管Q9和第十一开关管Q11组成一个桥臂，然后由谐振电容Cr2a和谐振电容Cr2b串联构成另外一个倍压桥臂，且cr2a＝cr2b＝1/2*cr2，其中cr2a为谐振电容Cr2a的电容值，cr2b为谐振电容Cr2b的电容值，cr2为第二谐振电容Cr2Cr2的电容值；同时，谐振电容Cr2a和谐振电容Cr2b还与第二谐振电感Lr2一起构成第二串联谐振单元，第二谐振电感Lr2为外置式电感、变压器Tra内部的耦合漏感或者外置电感和变压器Tra内部漏感的耦合电感。第二谐振单元与变压器Tra次级侧的线圈串联关系、第二谐振电感Lr2和第二谐振电容Cr2在串联环路中的连接顺序均是可以调动的，且所述第二谐振电容Cr2和第二谐振电感Lr2的的谐振频率
Figure PCTCN2022117893-appb-000006
Figure PCTCN2022117893-appb-000007
其中lr2为第二谐振电感Lr2的电感值，cr2为第二谐振电容Cr2的电容值，并且f1 0＝f2 0。
如图17及图18所示的相关整流电路，是大家熟知的电路，具体工作原理本领域技术人员应该理解，本文将不再深入分析。本发明亦不局限于上述实现案例，其他可实现本发明功能的组合方式亦都属于本范畴。
图19为本发明另一实施例的具体连接示意图，在次级整流输出单元与变压器Tra次级侧之间串入了第二谐振单元次级整流输出单元是全桥式整流单元。按照前述的控制方法，假设输入电压经控制判断，需要将变压器Tra初级侧的第五至第八开关管Q5～Q8的驱动占空比减少，不能维持在最大占空比或者47.5％，同时也必须相应的缩减输入开关型整流桥臂组“中”模式PWM驱动的占空比值，此模式称之为降压模式。此时，当变压器Tra初级侧的占空比关闭后，由于谐振回路的电流不能突变，第一谐振电感Lr1及励磁电感Lm的电动势会发生变化，变压器Tra初级侧的电流则会通过先前未开通的开关管的反并联二极管进行续流。若此时开关管的占空比均已关闭，则续流电流反向对吸收缓冲电容Cs充电，同时变压器Tra 次级侧也会因为变压器Tra耦合电压将原来储存在第二谐振电感Lr2和励磁电感Lm中部分能量释而续流导通；但若将施加给桥式谐振变换单元共源极、共漏极或者是其中一个直接串联桥臂的两个开关管的占空比固定为47.5％，则续流电流通过开通的共源极或者共漏极的开关管，如第七开关管Q7和第八开关管Q8进行循环，原来第一谐振电感Lr1和第一谐振电容Cr1的串联电压则被施加在变压器Tra上，能量被耦合传递到变压器Tra次级侧，即通过次边整流输出。两者不同之处在于，后一种方法更趋近于传统的降压变换器续流特性。同时，在该变换特性下，输出电压相比较低，主要是因为在续流期间变压器Tra及电感的伏秒平衡特性引起的有效占空比丢失；相比来说，初级侧续流释放能量回到滤波电容C1的磁复位过程更快，因此在下一次的开通之后，电流的有效传递时间更长，则回流传递到次级侧的能量更多，损耗也更大。
因此，在本发明实施例中，桥式谐振变换单元如果是需要工作在降压模式，则需对每次导通的两个开关管优先施加差异化驱动，其中一个开关管的驱动信号为根据控制计算得出的占空比驱动，另外一个开关管的驱动信号为固定的最大占比驱动或者占空比为47.5％的驱动新高，即可将本次施加给该全桥共源极或者共漏极或者其中一个直接串联桥臂的两个开关管的占空比固定为47.5％或者接近最大占空比。
假设按照前述的控制方法，若对桥式谐振变换单元施加的PWM驱动加大占空比到最大限制值依然不能达到输出电压的需求，除将对桥式谐振变换单元施加的PWM驱动固定为最大占空比，对输入开关型整流桥臂组施加“中”模式PWM驱动占空比外，还需将工作频率调节至最佳工作频率点进入升压模式，在下一个整流导通周期即将开始前对次级整流单元非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，通过调节驱动的占空比大小实现输出电压的调节，如果不需要升压即可满足输出电压的需求，则退出该升压模式。同时驱动占空比在对应变压器Tra初级侧无效占空比区间时，在变压器Tra初级侧开关管开通后，电流未转正前次级整流单元的驱动仅作同步整流使用，即升压驱动对应变压器Tra初级侧的驱动起点，有较短时间的无效升压占空比。
由于桥式谐振变换单元中第五至第八开关管Q5～Q8的驱动占空比施加到最大，并次级整流单元中非本周期内整流导通桥臂的其中一个开关管增加PWM驱动，即只对本周期内非整流导通的第十一开关管Q11或第十二开关管Q12增加升压PWM驱动，或另外增加同步整流驱动。第九开关管Q9，第十开关管Q10可做同步整流或者视作二极管；同时次级整流单元施加的驱动信号要早于桥式谐振变换单元的驱动信号，一般至少提前2％～5％周期，在本发 明实施例里，次级整流单元的驱动信号要比桥式谐振变换单元驱动信号早200ns。由于桥式谐振变换单元的其中两个开关管在本周期内非整流导通，即在前周期内整流导通，或者为变压器Tra整流回路电动势顺向通路，因此提前开通开关管的方式则为零电压开通。同时根据前面的导通方式，结合同步整流，因此升压驱动也可以理解为是具有同步整流功能PWM驱动的延续，数值等于同步整流占空比加升压占空比及死区时间的和；进入下个工作周期后，由于端口电压随着电感电动势及变压器Tra感应电压的反向，则进入类似输出侧短路状态，即本应加在桥式谐振变换单元输入端口上的整流电压在第十一开关管Q11和第十二开关管Q12上形成了通路回流，由于第一串联谐振单元的第一谐振电容Cr1和第二串联谐振单元的第二谐振电容Cr2的电压不能突变，而变压器Tra端口电压是直接耦合，因此等于给第一串联谐振电感及第二串联谐振电感施加储能，当施加在第十一二极管D11或第十二二极管D12上的驱动电压结束后，短路状态消失，第一谐振电感Lr1、第一谐振电容Cr1、第二谐振电感Lr2和第二谐振电容Cr2续谐振并进行续流，因此在变压器Tra的次级侧上的耦合电压再叠加第二串联谐振单元的电压，使次级侧的次级整流单元导通，以此完成了初级向输出供电的升压转换过程。
此外，桥式谐振变换单元也可以是半桥式谐振变换单元，如图20所示。此外，由于输入开关型整流桥臂组也是可控型开关，因此可以复用输入开关型整流桥臂组的开关作为半桥式谐振变换单元的一个桥臂开关，将串联谐振单元与变压器Tra回路分别连接在开关整流桥臂的正输出端口2和负输出端口3，半桥式谐振变换单元的另外一个开关管的漏极连接正输出端口2，开关管的源极连接负输出端口3。因此图20的半桥式谐振变换单元的上半桥臂还可以由输入可控开关代替，从而简化为图21，此时前述的全桥式谐振变换单元中的“高”PWM驱动信号则需调整为不超过50％占空比的PWM信号，不能为长高，与半桥式谐振变换单元的下半桥臂形成互补的驱动，以免形成直通短路。半桥式谐振变换单元的工作原理与全桥式谐振变换单元的工作原理一致，本领域人员均可由前述原理得知，在此不再进行赘述。
以上所述仅为本发明的优选实施例而已，并不用于限制本发明，对于本领域的技术人员来说，本发明可以有各种更改和变化。凡在本发明的精神和原则之内，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (11)

  1. 一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，包括输入开关型整流桥臂组、吸收缓冲单元、桥式谐振变换单元、变压器和次级整流输出单元；所述输入开关型整流桥臂组包括第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂，每个开关整流桥臂均包括交流输入端口、正输出端口和负输出端口，所述交流输入端口、正输出端口和负输出端口之间设置有等效可控选择开关；所述吸收缓冲单元包括串联连接的第四开关管和吸收缓冲电容；所述第四开关管的源极分别与第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的正输出端口连接，所述吸收缓冲电容分别与第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的负输出端口连接；所述桥式谐振变换单元包括桥式逆变电路和第一串联谐振单元，所述桥式逆变电路为全桥式逆变电路或半桥式逆变电路，所述第一串联谐振单元包括串联连接的第一谐振电容和第一谐振电感，所述桥式逆变电路的输入端与所述吸收缓冲单元连接，输出端与第一谐振电容和变压器初级侧连接，所述变压器初级侧还与所述第一谐振电感连接；所述变压器次级侧与所述次级整流输出单元连接；所述次级整流输出单元包括整流电路和滤波电容，所述整流电路的输入端与所述变压器的次级侧连接，输出端与滤波电容连接，所述整流电路为倍压整流电路、全波整流电路或全桥整流电路，所述整流电路由全二极管组成、带有反并联二极管的开关管或具备整流功能的其他形式开关管中的其中一种组成，或者由开关管和二极管混合组成。
  2. 根据权利要求1所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，所述等效可控选择开关为二极管与高频开关管的串联组合或者两个高频开关管反向串联后再与二极管连接，所述等效可控选择开关根据交流整流导通的需要对高频开关管施加高频PWM驱动信号可以控制开通与关断从而实现有方向选择性的导通连接，即形成对交流正半波的高频脉冲式整流导通，或者交流负半波高频脉冲式整流导通。
  3. 根据权利要求2所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，所述等效可控选择开关由一个开关管和四个二极管组成，或者由两个开关管和两个二极管组成；
    当所述等效可控选择开关由一个开关管和四个二极管组成时，所述开关管的源极与第十一二极管和第十二二极管的阳极连接，所述开关管的源极与第三二极管和第四二极管的阴极连接；所述第十一二极管的阴极与所述正输出端口连接；所述第十二二极管的阴极和第三二极管的阳极与所述交流输入端口连接；所述第四二极管的阳极与所述负输出端口连接；
    当所述等效可控选择开关由两个开关管和两个二极管组成时，具有三种连接方式：
    第一种连接方式为第一开关管和第十一二极管串联成第一支路后，一端与所述交流输入端口连接，另一端与所述正输出端口连接，第二开关管和第十二二极管串联成第二支路后，一端与所述交流输入端口连接，另一端与所述负输出端口连接，所述第一支路与所述第二支路关于所述交流输入端口对称；
    第二种连接方式为第一开关管和第二开关管反向串联后，第一开关管与所述交流输入端口连接，第二开关管与第十一二极管的阳极和第十二二极管的阴极连接，第十一二极管的阴极与所述正输出端口连接，第十二二极管的阳极与所述负输出端口连接；
    第三种连接方式为第一开关管和第二开关管反向串联后，第一开关管与所述交流输入端口连接，第十一二极管的阳极与第一开关管和第二开关管的串联点连接，第十一二极管的阴极与所述正输出端口连接，第十二二极管的阴极与第二开关管连接，第十二二极管的阳极与所述负输出端口连接。
  4. 根据权利要求3所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，所述输入开关型整流桥臂组中的开关管为设置有反并二极管的高频开关管，所述反并二极管为集成二极管、寄生二极管或外加二极管；所述吸收缓冲电容为无极性的电容或有极性的电容；所述第一谐振电感为外置式电感、变压器内部的耦合漏感或者外置电感和变压器内部漏感的耦合电感。
  5. 根据权利要求1所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，还包括第二串联谐振单元，所述第二串联谐振单元包括串联连接的第二谐振电容和第二谐振电感，所述第一谐振电容和第一谐振电感的的谐振频率
    Figure PCTCN2022117893-appb-100001
    其中lr1为第一谐振电感的电感值，cr1为第一谐振电容的电容值；所述第二谐振电容和第二谐振电感的的谐振频率
    Figure PCTCN2022117893-appb-100002
    其中lr2为第二谐振电感的电感值，cr2为第二谐振电容的电容值，并且f1 0＝f2 0。
  6. 根据权利要求1所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器，其特征在于，还包括输入滤波器，所述输入滤波器与所述第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的交流输入端口连接。
  7. 一种无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法，其特征在于，用于控制权利要求1～6任一权利要求所述的无输入储能电感隔离谐振软开关型三相PFC变换器，包括如下步骤：
    S100：根据输入的三相三线电源电压信号的锁相分析各相电源当前时刻所处的相位和区间段；
    S200：根据步骤S100中的锁相相位分析出各个所述区间段中各相电源的电压的瞬时值大小；
    S300：检测输入输出条件，判断输入输出条件是否满足系统需求的工作条件，不满足条件继续等待；如若满足条件，变换器开始工作；
    S400：对当前区间段下的输入开关型整流桥臂组施加驱动信号进行PWM驱动控制使其中瞬时值较高的两相电流先导通；随之施加PWM驱动信号使第四开关管导通，然后将已导通的瞬时值次高相交流回路上的开关管通路及第四开关管关断，让瞬时值最高相和瞬时值最低相的电流继续导通；具体方法为：给瞬时值最高与最低的两相交流回路中对应开关管同时施加相同占空比大小的“高”模式PWM驱动信号，同时对幅值瞬时值次高的电流回路中对应的开关管施加“中”模式PWM驱动信号，使在各个区间段中，施加“高”模式PWM驱动信号的开关管后关断，施加“中”模式PWM驱动信号的开关管先关断；使各相电流在每个开关周期内都能够导通；同时对第四开关管施加延迟于“中”模式开通和同时关断的PWM驱动信号。
  8. 根据权利要求7所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法，其特征在于，在步骤S300～S400中，当输入开关型整流桥臂组和第四开关管处于PWM工作状态时，第四开关管的PWM开关频率与控制第一开关整流桥臂、第二开关整流桥臂和第三开关整流桥臂的PWM开关频率一致，并且是桥式谐振变换单元工作频率的两倍，第四开关管和桥式谐振变换单元定义的工作相对起点一致，桥式谐振变换单元的各驱动占空比一致且不超过0.5，并留有桥式逆变电路必要的死区时间；所述“高”模式PWM驱动信号为一直存在的高电平信号或PWM驱动信号，所述“高”模式PWM驱动信号的驱动电压高电平时间大于所述“中”模式PWM驱动信号高电平时间；当桥式谐振变换单元为半桥式谐振变换单元时，施加给输入开关型整流桥臂组的“高”模式PWM驱动信号为只能是不超过50％占空比的PWM驱动信号。
  9. 根据权利要求7所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法，其特征在于，当次级整流单元串联有第二谐振单元时，在步骤S300～S400中，若根据变压器变比计算需要升压，且桥式谐振变换单元的逆变占空比以及达到47.5％或者最大占空比时，变换器工作在升压模式，在该模式下，固定开关管的开关频率在谐振频率附近；若 次级变换单元是全桥变换器，可以在下一个整流导通周期即将开始前只对非本周期内整流导通桥臂的其中一个开关管施加PWM驱动，也可以对非本周期内整流导通桥臂的两个开关管都增加PWM驱动；若次级变换单元是半桥整流式变换时，则在下一个整流导通周期即将开始前只对本周期内非整流导通的开关管增加PWM驱动；当桥式谐振变换单元的逆变占空比没有达到47.5％或者最大占空比时，则是降压模式，在该模式下，对每次导通的两个开关管施加差异化驱动，一个开关管的驱动信号为根据控制计算得出的占空比驱动，另外一个开关管驱动信号为固定的最大占比驱动。
  10. 根据权利要求7所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法，其特征在于，在步骤S300～S400中，若是升压模式，施加给桥式谐振变换单元做升压工作的开关管驱动信号要早于次级整流输出单元的驱动信号，施加给桥式谐振变换单元做升压工作的开关管驱动信号为上周期同步整流信号的延迟，即同步整流占空比加升压占空比及死区时间的和；非升压模式下桥式谐振变换单元和次级整流输出单元的开关管则施加同步驱动信号。
  11. 根据权利要求7所述的一种无输入储能电感隔离谐振软开关型三相PFC变换器的控制方法，其特征在于，每相导通电流的时间与相电压的瞬时值成正比关系，瞬时值最大相的电流导通时间等于其它两相电流导通时间的总和。
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