WO2022127167A1 - 一种抑制单电阻采样永磁同步电机低速噪音的方法 - Google Patents

一种抑制单电阻采样永磁同步电机低速噪音的方法 Download PDF

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WO2022127167A1
WO2022127167A1 PCT/CN2021/113448 CN2021113448W WO2022127167A1 WO 2022127167 A1 WO2022127167 A1 WO 2022127167A1 CN 2021113448 W CN2021113448 W CN 2021113448W WO 2022127167 A1 WO2022127167 A1 WO 2022127167A1
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sampling
permanent magnet
synchronous motor
vector
magnet synchronous
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PCT/CN2021/113448
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French (fr)
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梁冠贤
童怀
李少钳
黄伟胜
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泛仕达机电股份有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H02P25/024—Synchronous motors controlled by supply frequency
    • H02P25/026—Synchronous motors controlled by supply frequency thereby detecting the rotor position
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation

Definitions

  • the invention relates to the field of motor control, in particular to a method for suppressing low-speed noise of a single-resistance sampling permanent magnet synchronous motor.
  • Permanent magnet synchronous motor has the advantages of simple structure, wide speed regulation range, high power density and high motor efficiency, and has been widely used in household appliances, fan and pump products and other fields.
  • the medium and small power permanent magnet synchronous motor drive system has high requirements on the cost performance of the product, and the proportion of the current sampling circuit in the hardware cost of the whole machine cannot be ignored.
  • the more commonly used current sampling method is to use two current sensors to measure the phase current, but This method is expensive.
  • the use of resistance sampling can simplify the system structure and reduce the cost of hardware circuits. At present, the more commonly used scheme is the double resistance sampling scheme.
  • the single-resistor sampling scheme of motor phase current can further simplify the hardware circuit layout and reduce the cost of hardware circuits. In recent years, this scheme has been paid more and more attention by manufacturers.
  • single-resistance sampling requires three-phase current reconstruction of the bus current of the drive system.
  • an unobservable area which can usually be divided into a low-modulation unobservable area, a sector transition unobservable area, and a high-modulation unobservable area.
  • the basic principle of three-phase current reconstruction of single-resistance sampling permanent magnet synchronous motor is to sample the bus current at different times in a PWM cycle, and obtain each phase current through phase current reconstruction.
  • the motor controller is controlled by SVPWM modulation.
  • Figure 2 shows the space voltage vector diagram corresponding to the 7-segment SVPWM of the permanent magnet synchronous motor vector control.
  • There are 8 switch working states including six non-zero voltage vectors V 1 ⁇ V 6 and two zero-voltage vectors V 0 , V 7 , which divide the voltage space plane into 6 sectors, in each period T s , any target voltage vector V ref in each sector can be determined by the two sectors of the sector.
  • a non-zero voltage vector and a zero vector are synthesized and generated together.
  • To analyze the unobservable region of current reconstruction it is first necessary to calculate the action time T 1 and T 2 of the non-zero voltage vector corresponding to each sector.
  • the relationship between the DC bus current and the three-phase current is determined by the state of the instantaneous switch value.
  • the two-phase current is sampled in one PWM cycle as shown in Figure 3.
  • the reference voltage vector is decomposed into the basic voltage vector V 1 (001) and V 3 (011), the bus current I dc sampled when the voltage vector V 1 acts corresponds to the W-phase current, and the bus current I dc sampled when the voltage vector V 3 acts corresponds to the U-phase current .
  • T min a minimum sampling time
  • T min consists of three parts: dead time T d , bus current settling time T set and AD conversion time T conv , and the size of T min is generally 3 ⁇ s to 5 ⁇ s.
  • the area where different phase currents cannot be sampled in a PWM cycle is called the unobservable area.
  • the space voltage vector hexagon is specifically divided into the observable area, the low modulation unobservable area, and the sector. Transition unobservable regions and high modulation unobservable regions.
  • the pulse phase shifting is the main reason for the abnormal sound in the single-resistor sampling scheme.
  • the low-speed operation load is very light, and the phase current is very small (such as 0.1A).
  • the sampling error relative to the phase current amplitude cannot be ignored.
  • the single-resistor sampling scheme is significantly more noisy than the dual-resistor sampling scheme.
  • the waveform distortion of the sampling current of the permanent magnet synchronous motor and the resulting low-speed noise problem limit the popularization and application of the single-resistor sampling scheme.
  • a single-resistor sampling permanent magnet synchronous motor phase current reconstruction strategy is proposed, which inserts measurement pulses in the unobservable region of low modulation and unobservable region of medium modulation, and adopts the method of voltage vector approximation in the unobservable region of high modulation, but this method exists.
  • Disadvantages of long execution time and complicated implementation process of the algorithm Saritha B et al. in "IEEE Trans on Industrial Electronics" (VOL. 54, NO. 5) used a sinusoidal fitting observer for the unobservable area to make the estimated current approach Reference is made to sinusoidal three-phase currents, but this method relies on motor parameters and does not account for low-speed errors.
  • the existing algorithm for improving the sampling accuracy of a single resistor is complex, and the processor takes a long time to execute the algorithm.
  • the scheme of suppressing the low-speed noise of a permanent magnet synchronous motor by optimizing the sampling current reconstruction algorithm of a single resistor is difficult to implement in a low-cost microprocessor control system.
  • the present invention provides a method with simple algorithm and high efficiency, which can effectively suppress the low-speed noise of a single-resistance sampling permanent magnet synchronous motor.
  • the technical solution adopted by the present invention is as follows: a method for suppressing low-speed noise of a single-resistance sampling permanent magnet synchronous motor, comprising the following steps:
  • Step 1 According to the parameters of the single-resistance sampling permanent magnet synchronous motor drive system, determine the minimum sampling time T min of single-resistance sampling;
  • Step 2 at the minimum set rotational speed n set_min , calculate the change curves of the two non-zero voltage vector action times T1 and T2 corresponding to each sector of the vector frequency conversion control, and calculate the size of the intersection point T cross of T1 and T2;
  • Step 3 Determine whether there is a low-modulation unobservable area reconstructed by a single-resistance three-phase current at the minimum set rotational speed n set_min ;
  • Step 4 If the single-resistor three-phase current reconstruction does not have an unobservable area with low modulation, go to Step 5 directly;
  • the PM synchronous motor back EMF coefficient is increased under the condition that the full rate of the motor slot is kept basically unchanged, and then return to step 1 to recalculate T min ;
  • Step 5 According to the change curves of T1 and T2, determine the maximum amplitude of T1 and T2 changes in each vector frequency conversion control iterative operation process, and limit the changes of T1 and T2 during operation.
  • T min T d +T set +T conv , wherein Td, Tset and Tconv are parameters in the system.
  • the permanent magnet synchronous motor vector control includes six non-zero voltage vectors V 1 -V 6 and two zero-voltage vectors V 0 and V 7 , six non-zero voltage vectors V 1 -V 6 and two zero-voltage vectors V 0 and V 7 divide the voltage space plane into 6 sectors.
  • any target voltage vector V ref in each sector is shared by the two non-zero voltage vectors and the zero vector of the sector.
  • T cross is the intersection of T 1 and T 2 curve functions
  • the Ts is a vector control operation cycle
  • V4 and V6 are the non-zero voltage vectors corresponding to the first sector
  • T0 is the sum of the action time of the zero voltage vectors V0 and V7
  • Vdc is the bus voltage
  • ⁇ ref is the voltage vector Vref and The angle between the non-zero voltage vectors V4.
  • the specific steps for determining the maximum magnitude of changes in T1 and T2 in each vector frequency conversion control iterative operation process are as follows:
  • Step 5-1 Calculate the electrical cycle of each phase current according to the minimum set speed n set_min
  • phase current cycle contains 6 sectors
  • Step 5-2 Calculate the maximum amplitude of the changes of T1 and T2 for each vector frequency conversion control iteration operation:
  • T 1_max and T 2_max are the maximum values of T1 and T2 respectively
  • T 1_min and T 2_min are the minimum values of T1 and T2 respectively
  • ⁇ is an amplification factor.
  • the specific steps for limiting the changes of T1 and T2 during operation are as follows:
  • Step 5-3 According to the situation that T1 and T2 are in the rising period or the falling period, limit the processing of T1 and T2:
  • T1 is in the rising period: T 1 (n) ⁇ T 1 (n-1)+ ⁇ T 1_max ;
  • T1 is in the falling period: T 1 (n)>T 1 (n-1)- ⁇ T 1_max ;
  • T2 is in the rising period: T 2 (n) ⁇ T 2 (n-1)+ ⁇ T 2_max ;
  • T2 is in the falling period: T 2 (n)>T 2 (n-1)- ⁇ T 2_max .
  • the present invention controls the law of the two non-zero voltage vector action times T1 and T2 curves corresponding to each sector by vector frequency conversion, and on the one hand, ensures the sampling of three-phase current by a single resistor by optimizing the back EMF coefficient of the motor.
  • the reconstruction does not enter the low-modulation unobservable area, and reduces the number of pulse shifts in the current sampling process, thereby eliminating the abnormal sound existing in the single-resistor sampling scheme;
  • the influence of small sampling error and the distortion of the motor phase current waveform thereby reducing the decibel value of the low-speed noise of the single-resistor sampling scheme.
  • Figure 1 is a schematic diagram of three-phase current reconstruction of a single-resistance sampling permanent magnet synchronous motor
  • Fig. 2 is the space voltage vector diagram of the permanent magnet synchronous motor
  • Fig. 3 is a three-phase current reconstruction observable area PWM waveform diagram
  • Figure 5-1 is a schematic diagram of the transition non-observation area of the three-phase current reconstruction sector
  • Figure 5-2 is a schematic diagram of the three-phase current reconstruction low modulation non-observation area
  • Fig. 6 is the T1, T2 curve simulation waveform diagram
  • FIG. 7 is a control block diagram of a single-resistance sampling permanent magnet synchronous motor system
  • Figure 8-1 shows the waveforms of T1 and T2 curves before Ke optimization
  • Figure 8-2 shows the T1 and T2 curve waveforms after Ke optimization
  • Figure 8-3 shows the T1 and T2 curve waveforms after limit processing
  • Figure 9-1 shows the optimized front-phase current waveform
  • Figure 9-2 shows the optimized phase current waveform
  • the present invention controls the law of two non-zero voltage vector action times T 1 and T 2 curves corresponding to each sector by vector frequency conversion.
  • the three-phase current reconstruction does not enter the low-modulation unobservable area, and reduces the number of pulse shifts in the current sampling process, thereby eliminating the abnormal sound existing in the single-resistor sampling scheme; on the other hand, through the optimization algorithm, T 1 , T 2 are reduced.
  • the burr of the waveform reduces the influence of sampling error and the distortion of the motor phase current waveform, thereby reducing the decibel value of the low-speed noise of the single-resistor sampling scheme, which includes the following steps:
  • Step 1 Determine the minimum sampling time T min of single-resistance sampling according to the single-resistance sampling permanent magnet synchronous motor drive system parameters
  • T min The size of T min is generally 3 ⁇ s to 5 ⁇ s, and the relationship is as follows:
  • T min T d +T set +T conv (5), Td, Tset and Tconv are parameters within the system;
  • Step 2 At the given minimum set speed n set_min , calculate the change curves of the two non-zero voltage vector action times T 1 and T 2 corresponding to each sector of the vector frequency conversion control, and calculate the difference between T 1 and T 2 The size of the intersection point T cross ;
  • the permanent magnet synchronous motor vector control includes six non-zero voltage vectors V 1 -V 6 and two zero-voltage vectors V 0 and V 7 , six non-zero voltage vectors V 1 -V 6 and two zero-voltage vectors V 0 and V 7 divide the voltage space plane into 6 sectors.
  • any target voltage vector V ref in each sector is shared by the two non-zero voltage vectors and the zero vector of the sector.
  • the other T1 and T2 of the second to sixth regions can be obtained from the non-zero voltage vectors V 1 to V 6 in their corresponding regions, two zero voltage vectors V 0 and V 7 and the vector control operation period Ts.
  • T cross is the intersection of T 1 and T 2 curve functions
  • the Ts is a vector control operation cycle
  • V4 and V6 are the non-zero voltage vectors corresponding to the first sector
  • T0 is the sum of the action time of the zero voltage vectors V0 and V7
  • Vdc is the bus voltage
  • ⁇ ref is the voltage vector Vref and The angle between the non-zero voltage vectors V4;
  • Step 3 Run the motor with no load at the lowest set speed n set_min , and judge whether there is a low-modulation unobservable area reconstructed by a single-resistor current: It can be seen from Figure 5-1 and Figure 5-2 that when , the reference voltage space vector is outside the low-modulation unobservable region of the single-resistor current reconstruction, and does not appear at any time At the same time, when it is less than T min , that is, there is no low-modulation unobservable region in the single-resistor current reconstruction.
  • Step 4 If the single-resistor current reconstruction does not enter the low-modulation unobservable area, go directly to Step 5. Otherwise, by increasing the number of motor winding turns and reducing the winding wire diameter accordingly, the full rate of the motor slot is kept basically unchanged. Next, increase the back EMF coefficient of the permanent magnet synchronous motor, and return to step 1 to recalculate T min .
  • the voltage vector Vref for calculating T1 and T2 can be calculated from the direct-axis voltage Vd and the quadrature-axis voltage Vq:
  • id and i q are the d and q axis currents, respectively;
  • R is the stator resistance;
  • L d and L q are the d and q axis inductances, respectively;
  • ⁇ is the electrical angular velocity;
  • Ke is the back- EMF coefficient of the permanent magnet synchronous motor.
  • T1, T2 and the size of the back EMF coefficient Ke are proportional, and T cross is the intersection of T1 and T2, so T cross is also proportional to Ke, increasing Ke can reach the goal of.
  • Step 5 Through the previous processing, the single-resistor current reconstruction does not enter the low modulation unobservable region when n set_min is no-load, then when the motor running speed n set > n set_min , the single-resistor current reconstruction will not enter the low modulation.
  • the unobservable area only exists in the unobservable area of the sector transition area.
  • T1 and T2 can be further limited to reduce the burrs of the T1 and T2 waveforms, reduce the influence of sampling errors and the distortion of the motor phase current waveform. Decrease the decibel value of the low-speed noise of the single-resistor sampling scheme.
  • the change curves of T1 and T2 are basically sawtooth waveforms, so it is much simpler to limit the amplitude of T1 and T2 than to limit the detected phase current.
  • Each phase current cycle contains 6 sectors, and the number of control iterations of each sector is further calculated where T s is the sampling period in FIG. 2 .
  • T 1_max and T 2_max are the maximum values of T1 and T2 respectively
  • T 1_min and T 2_min are the minimum values of T1 and T2 respectively
  • the minimum sampling time T min of single-resistor sampling is set to 4 ⁇ s.
  • the single-resistor sampling permanent magnet synchronous motor system adopts the position sensorless vector control, as shown in Fig. 7 is the system control block diagram, including current sampling, rotor position estimation, Clarke and PARK transformation, maximum torque current ratio control (MTPA), Speed loop, dq axis current loop, PARK inverse transformation, SVPWM calculation, three-phase PWM inverter and other units.
  • the back EMF coefficient ke 3.0V/krpm, as shown in Figure 8-1, for the waveforms of T1 and T2 before Ke optimization, the intersection point of T1 and T2 T cross ⁇ 7 ⁇ s, and the minimum sampling time Tmin is 4 ⁇ s, which does not meet the condition T cross /2>T min , which means that the single-resistor current reconstruction may enter either the low modulation unobservable area or the unobservable area of the sector transition area.
  • the control algorithm needs to perform frequent pulse phase shifting , which causes the PWM output waveform to be extremely asymmetrical, which greatly increases the harmonic components of the phase current and also increases the sampling error.
  • the intersection point T cross of T1 and T2 is increased from 7 ⁇ s to 10 ⁇ s.
  • the condition T cross /2>T min is satisfied, and the single-resistor current reconstruction does not enter the low modulation and unobservable region. And only enter the unobservable area of the transition area of the sector, through this processing can greatly reduce the number of pulse phase shift.
  • FIG. 9-1 shows the current waveform before optimization
  • Figure 9-2 shows the waveform after optimization. It can be seen from Figures 9-1 and 9-2 that the actual waveform of the phase current shows that the influence of sampling error is also greatly reduced, and the phase current of the motor is greatly reduced. The distortion of the waveform is greatly reduced.
  • Figure 10- 1 to Fig. 10-2 show that the present invention has a practical effect on suppressing motor noise.

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Abstract

本发明公开了一种抑制单电阻采样永磁同步电机低速噪音的方法，包括如下步骤：(1)根据单电阻采样永磁同步电机驱动系统参数，确定最小采样时间T min；(2)针对最低设定转速n set_min处电机空载运行工况，计算时间T 1、T 2的变化曲线；(3)判断在给定低速运行节点n set_min是否存在低调制不可观测区域；(4)如果不存在不可观测区域则直接转入下一个步骤(5)，否则返回步骤(1)重新计算；(5)根据T 1、T 2的变化曲线，确定每一次矢量变频控制迭代运算过程中T 1、T 2变化最大幅值，在运行过程中对T 1、T 2的变化进行限幅处理。本发明减小了T 1、T 2波形的毛刺，降低了单电阻采样方案低速噪音的分贝值。

Description

一种抑制单电阻采样永磁同步电机低速噪音的方法 技术领域
本发明涉及电机控制领域，具体涉及一种抑制单电阻采样永磁同步电机低速噪音的方法。
背景技术
永磁同步电机具有结构简单、调速范围宽、功率密度和电机效率高等优点，目前已被广泛应用于家电、风机泵类产品等领域。中小功率永磁同步电机驱动系统对产品性价比的要求较高，而电流采样电路在整机硬件成本中所占的比例不可忽视，较为常用的电流采样方法是采用两个电流传感器测量相电流，但这种方法成本较高。使用电阻采样可以简化系统结构，降低硬件电路成本，目前较常用的方案是双电阻采样方案。电机相电流单电阻采样方案可以进一步简化硬件电路布局、降低硬件电路成本，近几年来这种方案越来越受到生产厂家的重视，但是单电阻采样需要对驱动系统母线电流进行三相电流重构，电流重构过程中存在盲区，即不可观测区，通常可以这种盲区分为低调制不可观测区、扇区过渡不可观测区、高调制不可观测区。
单电阻采样永磁同步电机三相电流重构的基本原理就是利用一个PWM周期内在不同的时刻采样母线电流，通过相电流重构得到各个相电流，如图1为单电阻采样永磁同步电机三相电流重构示意图。电机控制器采用SVPWM调制方式控制，如图2所示为永磁同步电机矢量控制7段SVPWM对应的空间电压矢量图，有8种开关工作状态，包括六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7，其将电压空间平面分成6个扇区，在每一个周期T s内，每个扇区中任意目标电压矢量V ref均可以由该扇区的两个非零电压矢量和零矢量共同合成产生，分析电流重构不可观测区首先需要计算每个扇区对应的非零电压矢量的作用时间T 1、T 2。
直流母线电流与三相电流关系由瞬时开关量的状态决定，以第4扇区为例在一个PWM周期内采样两相电流如图3所示，图3中参考电压矢量分解成基本电压矢量V 1(001)与V 3(011)，在电压矢量V 1作用时采样的母线电流I dc对应的是W相电流，在电压矢量V 3作用时采样的母线电流I dc对应的是U相电流。
在实际系统中，考虑到母线电流的采样需要足够的采样窗口，这就要求非零电压矢量必须持续一个最小采样时间T min。当输出的电压矢量处于低调制区或非零电压矢量附近时，在一个PWM周期内可能存在非零电压矢量作用的时间小于T min，这种情况使采样的母线电流没有意义，如图4为实际情况中最短采样时间T min示意图。图4中T min由三部分组成：死区时间T d、母线电流建立时间T set与AD转换时间T conv，T min大小一般为3μs～5μs。
从图3可知完成单电阻采样电流重构必须保证非零电压矢量在前半个采样周期T s内的作用时间大于T min，也就是要求满足：
Figure PCTCN2021113448-appb-000001
在一个PWM周期内不能采样到不同相电流的区域称为不可观测区，在实际应用中为了便于处理，将其空间电压矢量六边形具体划分为可观测区、低调制不可观测区、扇区过渡不可观测区和高调制不可观测区。
前期生产实践过程中，对于一款用于室内空气净化的风机产品，发现在低速区(500转/分～800转/分)单电阻采样方案的噪音明显比双电阻采样方案大，且单电阻采样方案还存在异音。进一步对比研究发现，在低调制不可观测区进行单电阻采样电流重构时，某一基本电压矢量状态作用的时间很短，需要通过在PWM周期内平移脉冲使其错开以得到足够的单电阻采样时间，也就是需要进行脉冲移相，这将导致PWM输出波形不对称，增加相电流谐波成分，脉冲移相是单电阻采样方案存在异音的主要原因。另一方面，低速运行负载很轻，相电流很小(如0.1A)，这时的采样误差相对相电流幅值不可忽略，采样误差导致相电流产生波形畸变并伴有较多的毛刺，导致单电阻采样方案的噪音明显比双电阻采样方案大。目前永磁同步电机采样电流波形畸变以及由此导致的低速噪音问题，限制了单电阻采样方案的推广应用。
目前关于抑制单电阻采样永磁同步电机低速噪音的研究主要集中在单电阻采样电流重构方面，黄科元等人在《电力系统及其自动化学报》(VOL.30,NO.9)上提出了一种单电阻采样永磁同步电机相电流重构策略，在低调制不可观测区和中调制不可观测区插入测量脉冲，在高调制不可观测区采用电压矢量近似的方法，但这种方法存在算法执行时间长、实施过程复杂的缺点；Saritha B等人在《IEEE Trans on Industrial Electronics》(VOL.54,NO.5)针对不可观测区采用正弦曲线拟合观测器，使估计的电流趋近参考正弦三相电流，但这种方法依赖电机参数，且不能解决低速误差问题。
现有提高单电阻采样精度的算法复杂、处理器执行算法时间长，通过优化单电阻采样电流重构算法抑制永磁同步电机低速噪音的方案，在低成本微处理器控制系统中实施困难。
发明内容
为了解决上述技术问题，本发明的提供一种算法简单，效率高，可以有效抑制单电阻采样永磁同步电机低速噪音的方法。
为实现上述目的，本发明采取的技术方案如下：一种抑制单电阻采样永磁同步电机低速噪音的方法，包括以下步骤：
步骤1：根据单电阻采样永磁同步电机驱动系统的参数，确定单电阻采样的最小采样时间T min；
步骤2：在最低设定转速n set_min处，计算矢量变频控制每一个扇区对应的两个非零电压矢量作用时间T1、T2的变化曲线，并计算T1与T2的交点T cross的大小；
步骤3：判断最低设定转速n set_min处是否存在单电阻三相电流重构的低调制不可观测区域；
当
Figure PCTCN2021113448-appb-000002
时，判定参考电压空间矢量处于单电阻电流重构的低调制不可观察区域之外；
当
Figure PCTCN2021113448-appb-000003
时，判定单电阻电流重构存在低调制不可观测区域；
步骤4：若单电阻三相电流重构不存在低调制不可观测区域则直接转入步骤5；
否则，则通过增大电机绕组匝数并相应减小绕组线径，在维持电机槽满率基本不变的情况下增大永磁同步电机反电势系数，然后返回到步骤1重新计算T min；
步骤5：根据T1、T2的变化曲线，确定每一次矢量变频控制迭代运算过程中T1、T2变化最大幅值，在运行过程中对T1、T2的变化进行限幅处理。
优选地，所述步骤1中：T min＝T d+T set+T conv，其中Td、Tset和Tconv为系统内参数。
优选地，永磁同步电机矢量控制包括六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7，六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7将电压空间平面分成6个扇区，在 每一个周期T s内，每个扇区中任意目标电压矢量V ref均由该扇区的两个非零电压矢量和零矢量共同合成产生，其中第1扇区的T 1、T 2由以下公式获得：
Figure PCTCN2021113448-appb-000004
T s＝T 0+T 1+T 2          (2)
Figure PCTCN2021113448-appb-000005
Figure PCTCN2021113448-appb-000006
所述T cross为T 1、T 2曲线函数的交叉点；
所述Ts为一个矢量控制运算周期；V4、V6为第1扇区对应的非零电压矢量；T0为零电压矢量V0、V7的作用时间总和；Vdc为母线电压；θ ref为电压矢量Vref与非零电压矢量V4之间的夹角。
优选地，所述步骤5中确定每一次矢量变频控制迭代运算过程中T1、T2变化最大幅值的具体步骤如下：
步骤5-1：根据最低设定转速n set_min计算出每个相电流的电周期
Figure PCTCN2021113448-appb-000007
其中p n为电机极对数，每个相电流周期包含6个扇区；
进一步计算出每一个扇区控制迭代运算的次数
Figure PCTCN2021113448-appb-000008
其中T s为采样周期；
步骤5-2：计算每一次矢量变频控制迭代运算T1、T2变化的最大幅值：
Figure PCTCN2021113448-appb-000009
其中T 1_max、T 2_max分别为T1、T2的最大值，T 1_min、T 2_min分别为T1、T2的最小值，ξ为放大系数。
优选地，所述ξ的取值为：ξ＝1.3。
优选地，所述步骤4中在运行过程中对T1、T2的变化进行限幅处理的具体步骤如下：
步骤5-3：根据T1、T2处于上升期或下降期的情况，对T1、T2进行限幅处理：
若T1处于上升期：T 1(n)＜T 1(n-1)+ΔT 1_max；
若T1处于下降期：T 1(n)＞T 1(n-1)-ΔT 1_max；
若T2处于上升期：T 2(n)＜T 2(n-1)+ΔT 2_max；
若T2处于下降期：T 2(n)＞T 2(n-1)-ΔT 2_max。
本发明有益的技术效果：本发明通过研究矢量变频控制每一个扇区对应的两个非零电压矢量作用时间T1、T2曲线的规律，一方面通过电机反电势系数优化确保单电阻采样三相电流重构不进入低调制不可观测区，减小电流采样过程中脉冲移位的次数，从而消除单电阻采样方案存在的异音；另一方面通过优化算法，减小T1、T2波形的毛刺，减小采样误差的影响和电机相电流波形的畸变，从而降低单电阻采样方案低速噪音的分贝值。
附图说明
图1为单电阻采样永磁同步电机三相电流重构示意图；
图2为永磁同步电机空间电压矢量图；
图3为三相电流重构可观测区PWM波形图；
图4为三相电流重构最短采样时间Tmin示意图；
图5-1为三相电流重构扇区过渡非观测区示意图；
图5-2为三相电流重构低调制非观测区示意图；
图6为T1、T2曲线仿真波形图；
图7为单电阻采样永磁同步电机系统控制框图；
图8-1为Ke优化前T1、T2曲线波形；
图8-2为Ke优化后T1、T2曲线波形；
图8-3为限幅处理后T1、T2曲线波形；
图9-1为优化前相电流波形；
图9-2为优化后相电流波形；
图10-1为优化前电机在n set_min＝500rpm处噪音频谱；
图10-2为优化后电机在n set_min＝500rpm处噪音频谱。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合实施例对本发明进行进一步详细说明，但本发明要求保护的范围并不局限于下述具体实施例。
如图1-2所示，本发明通过研究矢量变频控制每一个扇区对应的两个非零电压矢量作用时间T 1、T 2曲线的规律，一方面通过电机反电势系数优化确保单电阻采样三相电流重构不进入低调制不可观测区，减小电流采样过程中脉冲移位的次数，从而消除单电阻采样方案存在的异音；另一方面通过优化算法，减小T 1、T 2波形的毛刺，减小采样误差的影响和电机相电流波形的畸变，从而降低单电阻采样方案低速噪音的分贝值，具体包括如下步骤：
步骤1：根据单电阻采样永磁同步电机驱动系统参数，确定单电阻采样的最小采样时间T min；
T min大小一般为3μs～5μs，其关系如下：
T min＝T d+T set+T conv    (5)，Td、Tset和Tconv为系统内参数；
从图3可知完成单电阻采样电流重构必须保证非零电压矢量在前半个采样周期T s内的作用时间大于T min，也就是要求满足：
Figure PCTCN2021113448-appb-000010
步骤2：在给定的最低设定转速n set_min处，计算矢量变频控制每一个扇区对应的两个非零电压矢量作用时间T 1、T 2的变化曲线，并计算T 1与T 2的交点T cross的大小；
具体地，永磁同步电机矢量控制包括六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7，六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7将电压空间平面分成6个扇区，在每一个周期T s内，每个扇区中任意目标电压矢量V ref均由该扇区的两个非零电压矢量和零矢量共同合成产生，其中第1扇区的T 1、T 2由以下公式获得：
Figure PCTCN2021113448-appb-000011
T s＝T 0+T 1+T 2        (2)
Figure PCTCN2021113448-appb-000012
Figure PCTCN2021113448-appb-000013
其他的第2到第6区的T1和T2由其相应区间的非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7以及矢量控制运算周期Ts可得。
所述T cross为T 1、T 2曲线函数的交叉点；
所述Ts为一个矢量控制运算周期；V4、V6为第1扇区对应的非零电压矢量；T0为零电压矢量V0、V7的作用时间总和；Vdc为母线电压；θ ref为电压矢量Vref与非零电压矢量V4之间的夹角；
步骤3：在最低设定转速n set_min处电机空载运行，判断是否存在单电阻电流重构的低调制不可观测区域：从图5-1和图5-2可知，当
Figure PCTCN2021113448-appb-000014
时，参考电压空间矢量处于单电阻电流重构的低调制不可观察区域之外，任何时刻都不会出现
Figure PCTCN2021113448-appb-000015
同时小于T min的情况，即单电阻电流重构不存在低调制不可观测区域。
当
Figure PCTCN2021113448-appb-000016
时，某些时刻可能会出现
Figure PCTCN2021113448-appb-000017
同时小于T min的情况，即单电阻电流重构存在低调制不可观测区域。
步骤4：如果单电阻电流重构不进入低调制不可观测区域则直接转入步骤5，否则通过增大电机绕组匝数并相应减小绕组线径，在维持电机槽满率基本不变的情况下增大永磁同步电机反电势系数，返回步骤1重新计算T min。
在公式(5)、(6)中计算T1、T2的电压矢量Vref可由直轴电压Vd和交轴电压Vq计算：
Figure PCTCN2021113448-appb-000018
而永磁同步电机的稳态电压方程：
Figure PCTCN2021113448-appb-000019
其中，i d、i q分别为d、q轴电流；R为定子电阻；L d、L q分别为d、q轴电感；ω为电角速度；K e为永磁同步电机反电势系数。
由上面公式(7)、(8)可推出V ref与K e的关系：
|V ref|∝K e      (9)
进一步由公式(3)、(4)可推出：
T 1∝K e，T 2∝K e      (10)
T1、T2和反电势系数K e的大小成正比，而T cross为T1与T2的交点，因此T cross也与K e成正比，增大K e可达到
Figure PCTCN2021113448-appb-000020
的目的。
由电机设计原理可知，增大电机绕组匝数并相应减小绕组线径可以增大永磁同步电机反电势系数K e，实际运行时为保证电机安匝数不变，电机的相电流会适当增大，针对低速运行且电机负载很轻的情况，相电流适当增大不会引起电机发热的问题。
步骤5：通过前面的处理在n set_min空载时单电阻电流重构不进入低调制不可观测区域，那么当电机的运行速度n set＞n set_min时，单电阻电流重构也不会进入低调制不可观测区域而只存在扇区过渡区不可观测区域，这样可以进一步对T1、T2进行限幅处理来减小T1、T2波形的毛刺，减小采样误差的影响和电机相电流波形的畸变，从而降低单电阻采样方案低速噪音的分贝值。
从图6可见T1、T2的变化曲线基本呈锯齿波形，这样对T1、T2进行限幅处理要比对所检测到的相电流进行限幅处理要简单得多。
1)根据最低设定转速n set计算出每个相电流的电周期
Figure PCTCN2021113448-appb-000021
其中p n为电机的极对数；
每个相电流周期包含6个扇区，进一步计算出每一个扇区控制迭代运算的次数
Figure PCTCN2021113448-appb-000022
其中T s为图2中的采样周期。
2)计算每一次矢量变频控制迭代运算T1、T2变化的最大幅值
Figure PCTCN2021113448-appb-000023
Figure PCTCN2021113448-appb-000024
其中T 1_max、T 2_max分别为T1、T2的最大值，T 1_min、T 2_min分别为T1、T2的最小值，ξ为放大系数，这里取ξ＝1.3。
3)根据T1、T2处于上升期或下降期的情况，对T1、T2进行限幅处理
若T1处于上升期：
T 1(n)＜T 1(n-1)+ΔT 1_max      (13)
若T1处于下降期：
T 1(n)＞T 1(n-1)-ΔT 1_max      (14)
若T2处于上升期：
T 2(n)＜T 2(n-1)+ΔT 2_max      (15)
若T2处于下降期：
T 2(n)＞T 2(n-1)-ΔT 2_max       (16)
本实施例实验验证所采用的永磁同步电机为一台应用于家用空气净化器的外转子风扇电机，其中永磁同步电机的参数为：额定电压DC 24V，最小运行转速n set_min＝500转/分(rpm)，最高运行转速n set_max＝3000转/分，极对数p n＝2，定子电阻R s＝1.6Ω，定子直轴电感L d＝1.0mH，交轴电感L q＝1.2mH，优化前反电势系数k e＝3.0V/krpm，矢量控制PWM频率为16KHz。根据系统硬件参数及功率管死区时间设定，单电阻采样的最小采样时间T min设为4μs。
本实施方案中单电阻采样永磁同步电机系统采用无位置传感器矢量控制，如图7为系统控制框图，包括电流采样、转子位置估算、Clarke和PARK变换、最大转矩电流比控制(MTPA)、速度环、dq轴电流环、PARK逆变换、SVPWM计算、三相PWM逆变器等单元。
优化前反电势系数k e＝3.0V/krpm，如图8-1为Ke优化前T1、T2波形，T1与T2的交点T cross≈7μs，而最小采样时间Tmin为4μs，不满足条件T cross/2＞T min，这意味着单电阻电流重构既可能进入低调制不可观测区域，也可能进入扇区过渡区不可观测区域，为了获得足够的采样时间，控制算法需要进行频繁的脉冲移相，这导致PWM输出波形极不对称，大大增加了相电流谐波成分，同时也增大了采样误差。
根据本发明的方案，将电机的反电势系数从k e＝3.0V/krpm提高到k e＝4.2V/krpm，同时实验校验表明电机在3000rpm同样可以安全运行，如图8-2为Ke优化后T1、T2的波形，如图所示T1与T2的交点T cross从7μs提高到了10μs，这时满足条件T cross/2＞T min，单电阻电流重构不进入低调制不可观测区域，而只进入扇区过渡区不可观测区域，通过这种处理可以大大减小脉冲移相的次数。
从图8-2可见，T1、T2波形还存在很多毛刺，尤其是在T1与T2的交点处，这可能会导致单电阻电流重构进入低调制不可观测区域，同时使脉冲移相的处理程序变得非常复杂，按本发明的方法对T1、T2波形进行限幅处理，图中取T 1_max＝T 2_max＝20μs，T 1_min＝T 2_min＝0，每个扇区迭代运算次数
Figure PCTCN2021113448-appb-000025
ξ＝1.3，如图8-3为按公式(13)-(16)计算进行限幅处理后T1、T2曲线波形，通过限幅处理前后T1、T2波形对比可见，限幅处理可以使T1、T2波形的毛刺大大减小。图9-1为优化前相电流波形，图9-2为优化后相电流波形，从图9-1和9-2可见，相电流实际波形表明采样误差的影响也大大减小，电机相电流波形的畸变大大减小。
图10-1到图10-2为本发明的优化前后，电机在n set_min＝500rpm处噪音频谱的对比，从图中可见优化前电机总的噪音达到22.8dB，而且由于脉冲移相次数频繁，在16KHz处有一个噪音达到14dB，这种频率超过10KHz的噪音工程上常常当作异音；优化处理后，电机总的噪音降为19.6dB，在16KHz处异音也降为4dB，图10-1到图10-2表明本发明对抑制电机噪音具有切实的效果。
根据上述说明书的揭示和教导，本发明所属领域的技术人员还可以对上述实施方式进行变更和修改。因此，本发明并不局限于上面揭示和描述的具体实施方式，对发明的一些修改和变更也应当落入本发明的权利要求的保护范围内。此外，尽管本说明书中使用了一些特定的术语，但这些术语只是为了方便说明，并不对发明构成任何限制。

Claims (6)

  1. 一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，包括以下步骤：
    步骤1：根据单电阻采样永磁同步电机驱动系统的参数，确定单电阻采样的最小采样时间T min；
    步骤2：在最低设定转速n set_min处，计算矢量变频控制每一个扇区对应的两个非零电压矢量作用时间T 1、T 2的变化曲线，并计算T 1与T 2的交点T cross的大小；
    步骤3：判断最低设定转速n set_min处是否存在单电阻三相电流重构的低调制不可观测区域；
    当
    Figure PCTCN2021113448-appb-100001
    时，判定参考电压空间矢量处于单电阻电流重构的低调制不可观察区域之外；
    当
    Figure PCTCN2021113448-appb-100002
    时，判定单电阻电流重构存在低调制不可观测区域；
    步骤4：若单电阻三相电流重构不存在低调制不可观测区域则直接转入步骤5；
    否则，则通过增大电机绕组匝数并相应减小绕组线径，在维持电机槽满率基本不变的情况下增大永磁同步电机反电势系数，然后返回到步骤1重新计算T min；
    步骤5：根据T 1、T 2的变化曲线，确定每一次矢量变频控制迭代运算过程中T 1、T 2变化最大幅值，在运行过程中对T 1、T 2的变化进行限幅处理。
  2. 如权利要求1所述的一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，所述步骤1中：T min＝T d+T set+T conv，其中Td、Tset和Tconv为系统内参数。
  3. 如权利要求1所述的一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，永磁同步电机矢量控制包括六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7，六个非零电压矢量V 1～V 6和两个零电压矢量V 0、V 7将电压空间平面分成6个扇区，在每一个周期T s内，每个扇区中任意目标电压矢量V ref均由该扇区的两个非零电压矢量和零矢量共同合成产生，其中第1扇区的T 1、T 2由以下公式获得：
    Figure PCTCN2021113448-appb-100003
    T s＝T 0+T 1+T 2  (2)
    Figure PCTCN2021113448-appb-100004
    Figure PCTCN2021113448-appb-100005
    所述T cross的大小为T 1、T 2曲线函数的交叉点；
    所述T s为一个矢量控制运算周期；V 4、V 6为第1扇区对应的非零电压矢量；T 0为零电压矢量V 0、V 7的作用时间总和；V dc为母线电压；θ ref为电压矢量V ref与非零电压矢量V 4之间的夹角。
  4. 如权利要求1所述的一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，所述步骤5中确定每一次矢量变频控制迭代运算过程中T 1、T 2变化最大幅值的具体步骤如下：
    步骤5-1：根据最低设定转速n set_min计算出每个相电流的电周期
    Figure PCTCN2021113448-appb-100006
    其中p n为电机极对数；
    进一步计算出每一个扇区控制迭代运算的次数
    Figure PCTCN2021113448-appb-100007
    其中T s为采样周期；
    步骤5-2：计算每一次矢量变频控制迭代运算T1、T2变化的最大幅值：
    Figure PCTCN2021113448-appb-100008
    其中T 1_max、T 2_max分别为T1、T2的最大值，T 1_min、T 2_min分别为T1、T2的最小值，ξ为放大系数。
  5. 如权利要求4所述的一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，所述ξ的取值为：ξ＝1.3。
  6. 如权利要求1所述的一种抑制单电阻采样永磁同步电机低速噪音的方法，其特征在于，所述步骤4中在运行过程中对T 1、T 2的变化进行限幅处理的具体步骤如下：
    步骤5-3：根据T 1、T 2处于上升期或下降期的情况，对T 1、T 2进行限幅处理：
    若T1处于上升期：T 1(n)＜T 1(n-1)+ΔT 1_max；
    若T1处于下降期：T 1(n)＞T 1(n-1)-ΔT 1_max；
    若T2处于上升期：T 2(n)＜T 2(n-1)+ΔT 2_max；
    若T2处于下降期：T 2(n)＞T 2(n-1)-ΔT 2_max。
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