WO2021233350A1 - 双向数字开关功率放大器及其多步电流预测控制方法 - Google Patents

双向数字开关功率放大器及其多步电流预测控制方法 Download PDF

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WO2021233350A1
WO2021233350A1 PCT/CN2021/094681 CN2021094681W WO2021233350A1 WO 2021233350 A1 WO2021233350 A1 WO 2021233350A1 CN 2021094681 W CN2021094681 W CN 2021094681W WO 2021233350 A1 WO2021233350 A1 WO 2021233350A1
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current
time
power amplifier
value
bidirectional digital
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PCT/CN2021/094681
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English (en)
French (fr)
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孙凤
宋高杰
陈丽
李强
徐方超
金俊杰
张明
佟玲
张晓友
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沈阳工业大学
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Priority to US17/926,693 priority Critical patent/US20230208292A1/en
Publication of WO2021233350A1 publication Critical patent/WO2021233350A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/157—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/02—Conversion of dc power input into dc power output without intermediate conversion into ac
    • H02M3/04—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters
    • H02M3/10—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/1555—Conversion of dc power input into dc power output without intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only for the generation of a regulated current to a load whose impedance is substantially inductive

Definitions

  • This application relates to the field of voltage-current digital switching power amplifier control, and in particular to a two-way digital switching power amplifier based on a magnetic levitation drive platform and a multi-step current predictive control method thereof.
  • the controller used in the two-way digital switching power amplifier for the magnetic levitation drive platform is the most widely used PI (Proportion Integration) controller.
  • the PI controller needs to repeatedly set Kp (proportional adjustment coefficient) and Ki (integral adjustment coefficient).
  • Kp proportional adjustment coefficient
  • Ki integral adjustment coefficient
  • Model predictive control is an optimal control method in a limited time domain. It predicts the state of the system in a limited time domain in the future and determines the control output of the system at the current moment.
  • traditional model predictive control requires adjustment of cumbersome weight coefficients and weight coefficients. The adjustment relies on experience, and the amount of online calculation is large, and it is difficult to apply in the high switching frequency of the switching power amplifier.
  • This application provides a two-way digital switch power amplifier suitable for a magnetic levitation drive platform and its multi-step current predictive control method. Its purpose is to solve the above-mentioned control technology that has long dynamic adjustment time, poor system robustness, and online operation The problem is large, and there is no need to tune complicated parameters, and there is no need to adjust the cumbersome weight coefficients.
  • the specific technical solutions are as follows:
  • an embodiment of the present application provides a multi-step current predictive control method for a bidirectional digital switch power amplifier, which includes the following steps:
  • Step 1 Establish a prediction model of the bidirectional digital switching power amplifier, and predict the current prediction values of the next two sampling moments (k+1, k+2) at the k-th sampling moment;
  • Step 2 Introduce a feedback correction term, closed-loop prediction, and obtain the current prediction values at k+1 and k+2 times after correction;
  • Step 3 According to the current reference value of the bidirectional digital switch power amplifier at the kth sampling time and the current predicted value at the k+1 and k+2th sampling time after the feedback correction term is introduced, the value function is used to obtain the optimal modulation Duty cycle
  • Step 4 According to the modulation duty ratio obtained in step 3, four PWM (Pulse Width Modulation) drive signals are generated by the pulse width modulation module to control the four switch tubes to realize current prediction control.
  • PWM Pulse Width Modulation
  • the prediction model described in step 1 is:
  • k) is the predicted current value (A) at time k at time k+1
  • i(k) is the output current sampling value at time k (A)
  • T S is the sampling period (s)
  • L is the inductance (H) of the load
  • R is the resistance ( ⁇ ) of the load
  • U o (k) is the output voltage (V) of the bidirectional digital switch power amplifier at time k.
  • the predicted current values of the two sampling moments (k+1, k+2) described in step 1 are:
  • k) is the current value (A) predicted at time k+1 at time k
  • k) is the current value predicted at time k+2 (A) at time k
  • i( k) is the output current sampling value (A) at time k
  • T S is the sampling period (s)
  • L is the inductance of the load (H)
  • R is the resistance of the load ( ⁇ )
  • U dc is the DC bus voltage (V)
  • D(k) is the duty cycle of the switch tube at k time.
  • the feedback correction term is introduced in step 2, and the current prediction values at k+1 and k+2 times after correction are obtained:
  • ⁇ 1 and ⁇ 2 are correction coefficients
  • k-1) is the predicted current value (A) at time k at time k-1
  • i(k) is the output current sampled value at time k (A)
  • k) is the predicted current value (A) at time k+1 at time k
  • k) is the predicted current value at time k+2 (A) at time k
  • T S is the sampling period (s)
  • L is the inductance (H) of the load
  • R is the resistance ( ⁇ ) of the load
  • U dc is the DC bus voltage (V)
  • D(k) is the duty cycle of the switch at k.
  • step 3 the value function described in step 3 is:
  • i*(k) is the current reference value (A) at time k
  • k) is the predicted current value (A) at time k+2 at time k.
  • step 3 the specific process of obtaining the optimal modulation duty ratio described in step 3 is:
  • the value range of the duty cycle D is: 0 ⁇ D ⁇ 1.
  • the embodiments of the present application provide a bidirectional digital switching power amplifier, which is composed of a current prediction controller, a PWM modulator, a phase shift circuit, an optoelectronic isolation circuit, a driving circuit, a power conversion circuit, a current Hall sensor and Load composition; the input terminal of the current prediction controller is connected to the current given signal and the current feedback signal in the load fed back by the current Hall sensor, and the output terminal of the current prediction controller is connected to the input terminal of the PWM modulator and the output terminal of the PWM modulator Connect the input end of the phase shift circuit, the output end of the phase shift circuit is connected to the input end of the photoelectric isolation signal, the output end of the photo isolation signal is connected to the input end of the drive circuit, and the four output ends of the drive circuit are respectively connected to the gate of the first switch tube S1 Terminal, the gate terminal of the second switching tube S2, the gate terminal of the third switching tube S3, the gate terminal of the fourth switching tube S4, the drain terminal of the first switching tube S1 is connected
  • the source terminal of a switching tube S1 is simultaneously connected to the drain terminal of the second switching tube S2 and one end of the current Hall sensor, the other terminal of the current Hall sensor is connected in series with one end of the load, and the source terminal of the third switching tube S3 is connected to the fourth switching tube S4 at the same time.
  • the drain terminal and the other terminal of the load, the source terminal of the second switch tube S2 is connected to the source terminal of the fourth switch tube S4 and the ground at the same time, and the output terminal of the current hall sensor is connected to the input terminal of the current prediction controller.
  • the power conversion circuit is composed of switch tubes S1, S2, S3, and S4, all of which are power MOS transistors (Metal Oxide Semiconductor Field Effect Transistor, metal oxide semiconductor field effect transistor) or IGBT (Insulated Field Effect Transistor).
  • MOS transistors Metal Oxide Semiconductor Field Effect Transistor, metal oxide semiconductor field effect transistor
  • IGBT Insulated Field Effect Transistor
  • Gate Bipolar Transistor, insulated gate bipolar transistor the driving signals of the switching tubes S1 and S2 are complementary, the driving signals of the switching tubes S3 and S4 are complementary, and the phase difference of the driving signals of the switching tubes S1 and S4 is 180 degrees, and the phase difference of the driving signals of the switching tubes S2 and S3 is 180 degrees.
  • the driving signal is 180 degrees out of phase.
  • the loads driven by the bidirectional digital switch power amplifier are all inductive loads.
  • the multi-step current predictive control method of the bidirectional digital switch power amplifier used in the magnetic levitation drive platform of this application is compared with the traditional PI control, which effectively improves the dynamic response speed of the system and reduces the dynamic adjustment time.
  • the multi-step current predictive control method of the bidirectional digital switch power amplifier used in the magnetic levitation drive platform of this application adopts two-step current predictive control to effectively compensate for the single-cycle delay caused by sampling conversion, algorithm calculation, and duty cycle update.
  • the multi-step current prediction control method of the bidirectional digital switch power amplifier used in the magnetic levitation drive platform of this application adopts closed-loop current prediction, which effectively improves the control accuracy of the system.
  • This application is used for the multi-step current predictive control method of the bidirectional digital switch power amplifier of the magnetic levitation drive platform.
  • the amount of online calculation is small, the algorithm is simple, and it is easy to implement digitally.
  • the multi-step current predictive control method for the bidirectional digital switch power amplifier of the magnetic levitation drive platform according to this application does not need to set complex parameters or adjust cumbersome weight coefficients.
  • Figure 1 is a schematic diagram of a two-way digital switch power amplifier
  • FIG. 2 is a block diagram of the multi-step current prediction control of the bidirectional digital switch power amplifier used in the magnetic levitation drive platform of the present application;
  • Figure 3 is a driving waveform diagram of the four switching tubes in the power conversion circuit of the bidirectional digital switch power amplifier
  • Figure 4 is a step simulation waveform diagram of a 1A current output from a PI control power amplifier
  • Fig. 5 is a step simulation waveform diagram of a power amplifier outputting a current of 1A using the control method of this application;
  • Figure 6 is a step simulation waveform diagram of a 2A current output from a PI control power amplifier
  • Fig. 7 is a step simulation waveform diagram of the power amplifier outputting 2A current using the control method of the present application
  • Figure 8 is a step simulation waveform diagram of a 5A current output from a PI control power amplifier
  • Fig. 9 is a step simulation waveform diagram of the power amplifier outputting a current of 5A using the control method of the present application.
  • This application provides a multi-step current predictive control method for a bidirectional digital switch power amplifier for a magnetic levitation drive platform, which includes the following steps:
  • Step 1 Establish a prediction model of the bidirectional digital switching power amplifier, and predict the current prediction values of the next two sampling moments (k+1, k+2) at the k-th sampling moment;
  • k is a positive integer.
  • the prediction model can be expressed by the following formula:
  • the current prediction values at the k+1 time and k+2 time are predicted.
  • the current prediction value at the k+1 time can be predicted first, and then the current prediction value at the k+2 time can be predicted using the current prediction value at the k+1 time.
  • Step 2 Introduce a feedback correction term, closed-loop prediction, and obtain the current prediction values at k+1 and k+2 times after correction;
  • the correction is to compensate for model mismatch and random interference
  • the feedback correction item can be introduced to realize the correction due to the influence of other factors.
  • Step 3 According to the current reference value of the bidirectional digital switch power amplifier at the kth sampling time and the current predicted value at the k+1 and k+2th sampling time after the feedback correction term is introduced, the value function is used to obtain the optimal modulation Duty cycle
  • the current reference value is the expected output current value, which can be customized according to the actual situation.
  • the current predicted value at the k+1 and k+2 sampling time after the feedback correction term is introduced is the corrected current predicted value at the k+1 and k+2 time obtained in step 2; the value function is used to express the prediction The error between the actual value and the actual value.
  • the cost function selects the square of the difference between the current reference value at time k and the predicted current value at time k+2.
  • Step 4 According to the modulation duty ratio obtained in step 3, four PWM drive signals are respectively generated by the pulse width modulation module to control the four switching tubes to realize the current predictive control.
  • the multi-step current predictive control method of the two-way digital switch power amplifier used in the magnetic levitation drive platform of this application effectively improves the dynamic response speed of the system and reduces the dynamic adjustment time compared with the traditional PI control.
  • the two-step current predictive control is adopted to effectively compensate for the sampling Single-cycle delay caused by conversion, algorithm calculation, and duty cycle update; closed-loop current prediction is used to effectively improve the system control accuracy; online calculation is small, the algorithm is simple, and easy to implement digitally; no need to set complex parameters, no need to adjust The cumbersome weight coefficient.
  • the bidirectional digital switching power amplifier in the embodiment of the application can be shown in Figure 1, and is composed of a current prediction controller, a PWM modulator, a phase shift circuit, an optoelectronic isolation circuit, a driving circuit, a power conversion circuit, a current Hall sensor, and a load.
  • the input terminal of the current prediction controller is connected to the current given signal (ie current reference value) and the current feedback signal in the load fed back by the current Hall sensor.
  • the output terminal of the current prediction controller is connected to the input terminal of the PWM modulator, PWM modulator
  • the output terminal of the phase shift circuit is connected to the input terminal of the phase shift circuit, the output terminal of the phase shift circuit is connected to the input terminal of the photoelectric isolation circuit, the output terminal of the photoelectric isolation circuit is connected to the input terminal of the drive circuit, and the four output terminals of the drive circuit are respectively connected to the first switch
  • the gate terminal of the tube S1, the gate terminal of the second switching tube S2, the gate terminal of the third switching tube S3, the gate terminal of the fourth switching tube S4, the drain terminal of the first switching tube S1 is connected to the drain terminal of the third switching tube S3 and the +U dc bus at the same time
  • the source terminal of the first switch tube S1 is connected to the drain terminal of the second switch tube S2 and one end of the current Hall sensor at the same time.
  • the other terminal of the current Hall sensor is connected in series with one end of the load.
  • the source terminal of the third switch tube S3 is connected to the fourth terminal at the same time.
  • the drain terminal of the switching tube S4 and the other end of the load, the source terminal of the second switching tube S2 are simultaneously connected to the source terminal of the fourth switching tube S4 and the ground, and the output terminal of the current hall sensor is connected to the input terminal of the current prediction controller.
  • the load is an inductive coil, the inductance is L and the resistance is R.
  • the current prediction controller calculates the optimal duty cycle according to the given signal of the circuit and the current feedback signal in the load fed back by the current Hall sensor, and then outputs the control signal after shifting
  • the phase circuit, photoelectric isolation circuit, and drive circuit control the conduction time and conduction sequence of the four switching tubes to realize current predictive control.
  • Fig. 2 is a block diagram of the multi-step current prediction control of the bidirectional digital switch power amplifier used in the maglev drive platform of this application.
  • the multi-step current prediction control method of the bidirectional digital switch power amplifier used in the maglev drive platform of this application specifically includes the following steps:
  • Step 1 Establish the prediction model of the two-way digital switching power amplifier. At the k-th sampling time, predict the current prediction values of the next two sampling times (k+1, k+2); the graph can be obtained from Kirchhoff's voltage law 2 The voltage relationship of the power conversion circuit is discretized and the prediction model of the two-way digital switch power amplifier is obtained:
  • k) is the predicted current value (A) at time k at time k+1
  • i(k) is the output current sampling value at time k (A)
  • T S is the sampling period (s)
  • L is the inductance (H) of the load
  • R is the resistance ( ⁇ ) of the load
  • U o (k) is the output voltage (V) of the bidirectional digital switch power amplifier at time k.
  • k) is the current value (A) predicted at time k+1 at time k
  • k) is the current value predicted at time k+2 (A) at time k
  • i( k) is the output current sampling value (A) at time k
  • T S is the sampling period (s)
  • L is the inductance of the load (H)
  • R is the resistance of the load ( ⁇ )
  • U dc is the DC bus voltage (V)
  • D(k) is the duty cycle of the switch tube at k time.
  • Step 2 Introduce feedback correction items to realize closed-loop prediction, correct the current prediction value in real time, reduce the prediction output error, compensate for the influence of factors such as model mismatch and random interference, enhance the robustness of the system, improve the control effect, and introduce
  • the feedback correction term is obtained by making the difference between the predicted current value at time k and the actual output current sampled value:
  • ⁇ is the correction coefficient
  • k-1) is the predicted current value (A) at time k at time k-1
  • i(k) is the output current sampled value (A) at time k.
  • the current predicted values at k+1 and k+2 times after correction are:
  • ⁇ 1 and ⁇ 2 are correction coefficients, i(k
  • the goal of control is to minimize the error between the current reference value and the actual output current value.
  • the actual output current value can track the current reference value stably.
  • the current reference value at k+2 can be approximately equal to k
  • the value function selects the difference between the current reference value at time k and the predicted current value at k+2 Squared to get:
  • i*(k) is the current reference value (A) at time k
  • k) is the predicted current value (A) at time k+2 at time k.
  • Step 3 According to the current given value of the two-way digital switch power amplifier at the current moment and the current predicted value after the feedback correction term is introduced, the optimal modulation duty cycle is obtained through the value function.
  • the specific process is as follows:
  • the value range of the duty cycle D is: 0 ⁇ D ⁇ 1.
  • Step 4 According to the optimal modulation duty ratio, four PWM drive signals are respectively generated by the pulse width modulation module to control the conduction time and conduction sequence of the four switching tubes, so as to realize the current predictive control.
  • FIG. 3 shows the driving waveforms of the four switching tubes in the power conversion circuit of the two-way digital switch power amplifier.
  • the optimal duty cycle obtained in step 4 is used as the duty cycle of the switching tube S1 to drive the switching tube S1.
  • the driving signal of the switching tube S2 is complementary to the driving signal of the switching tube S1.
  • the driving signals of the switching tube S3 and the switching tube S4 are obtained by shifting the phase of the driving signals of the switching tube S2 and the switching tube S1 by 180 degrees, which can also be understood as a phase shift of 1/ 2 cycles.
  • the loads driven by the two-way digital switching power amplifier used in the magnetic levitation drive platform are all inductive loads, and the switching frequency is fixed.
  • the dynamic adjustment time is about 0.6ms, and the output current is 0.997A when stable, and the steady-state error is 0.3%;
  • Figure 6 is the step response waveform of the power amplifier output 2A current under PI control, the dynamic adjustment time is about 1.5ms, the output current is 1.958A when stable, and the steady-state error is 2.1%;
  • Figure 7 is the control method of the application The step response waveform of the power amplifier output 2A current, the dynamic adjustment time is about 0.8ms, the output current is 1.994A when stable, the steady-state error is 0.3%;
  • Figure 8 is the step response waveform of the power amplifier output 5A current under PI control, dynamic The adjustment time is about 2.1ms, the output current is 4.89A when stable, and the steady-state error is 2.2%;
  • Figure 9 is the step response waveform of the power amplifier outputting a current of 5A under the control method of the application, and the dynamic adjustment time is about 1.6ms.
  • the output current is 4.984A, and the steady-state error is 0.32%; it can be seen that under the same conditions, the output current level is the same, and the control method of the present application has shorter dynamic adjustment time and higher steady-state accuracy than the traditional PI control.
  • This application is used for the multi-step current predictive control method of the bidirectional digital switching power amplifier of the magnetic levitation drive platform. Compared with the PI control adopted by the switching power amplifier in the prior art, the control method can effectively reduce the system dynamic adjustment time and increase the power amplifier current. Response speed, no need to tune complex parameters, avoiding the danger of PI controller parameter tuning and debugging in high-power applications.
  • the multi-step current predictive control method of the two-way digital switch power amplifier used in the magnetic levitation drive platform adopts closed-loop current prediction, which effectively improves the control accuracy of the system, and has small steady-state errors. It uses two-step current predictive control to effectively compensate for the sampling conversion, The single-cycle delay caused by the calculation of the algorithm and the update of the duty cycle, the amount of online calculation is small, the algorithm is simple and easy to implement digitally, and has good practical value and application prospects.

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Abstract

本申请涉及一种基于磁悬浮驱动平台的双向数字开关功率放大器及其多步电流预测控制方法，控制方法包括如下步骤：建立双向数字开关功率放大器的预测模型；引入反馈校正项，闭环预测；通过价值函数求取最优的调制占空比；根据得出的调制占空比，经脉宽调制模块分别产生四路PWM驱动信号来控制四个开关管，实现电流预测控制。本申请有效提高了系统控制精度，稳态误差小，在线计算量小，算法简便，易数字化实现，具有良好的实用价值和应用前景。

Description

双向数字开关功率放大器及其多步电流预测控制方法
本申请要求于2020年05月21日提交中国专利局、申请号为202010433364.4发明名称为“双向数字开关功率放大器及其多步电流预测控制方法”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电压-电流型数字开关功率放大器控制领域，具体涉及一种基于磁悬浮驱动平台用的双向数字开关功率放大器及其多步电流预测控制方法。
背景技术
磁悬浮驱动平台用的双向数字开关功率放大器所使用的控制器应用最广泛的为PI(Proportion Integration，比例积分)控制器，PI控制器需要反复整定Kp(比例调节系数)、Ki(积分调节系数)参数以适应当前的控制要求，整定过程困难且参数适应范围小，动态调节时间较长，系统的鲁棒性较差，大功率应用场合下，参数整定调试存在一定危险性。
模型预测控制是一种有限时域最优控制方式，对系统未来有限时间域内的状态进行预测并确定系统当前时刻的控制输出，但传统的模型预测控制，需要调节繁琐的权重系数、权重系数的调节依靠经验，且在线运算量大、在开关功率放大器这种开关频率较高的场合难以应用。
发明内容
本申请提供一种适用于磁悬浮驱动平台用的双向数字开关功率放大器及其多步电流预测控制方法，其目的为解决上述控制技术中存在动态调节时间较长、系统鲁棒性较差、在线运算量大的问题、且不需要整定复杂的参数、不需要调节繁琐的权重系数。具体技术方案如下：
第一方面，本申请实施例提供了一种双向数字开关功率放大器多步电流预测控制方法，包括如下步骤：
步骤1，建立双向数字开关功率放大器的预测模型，在第k个采样时刻， 预测未来两个采样时刻(k+1、k+2)的电流预测值；
步骤2，引入反馈校正项，闭环预测，得到校正后第k+1、k+2时刻的电流预测值；
步骤3，根据第k个采样时刻双向数字开关功率放大器的电流参考值以及引入反馈校正项后的第k+1、k+2个采样时刻的电流预测值，通过价值函数求取最优的调制占空比；
步骤4，根据步骤3得出的调制占空比，经脉宽调制模块分别产生四路PWM(Pulse Width Modulation，脉冲宽度调制)驱动信号来控制四个开关管，实现电流预测控制。
在一种可能的实施方式中，步骤1所述的预测模型为：
Figure PCTCN2021094681-appb-000001
其中，i(k+1|k)为k时刻预测k+1时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)，T S为采样周期(s)，L为负载的电感(H)，R为负载的电阻(Ω)，U o(k)为k时刻双向数字开关功率放大器的输出电压(V)。
在一种可能的实施方式中，步骤1所述的两个采样时刻(k+1、k+2)的电流预测值为：
Figure PCTCN2021094681-appb-000002
Figure PCTCN2021094681-appb-000003
其中，i(k+1|k)为k时刻预测k+1时刻的电流值(A)，i(k+2|k)为k时刻预测k+2时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)，T S为采样周期(s)，L为负载的电感(H)，R为负载的电阻(Ω)，U dc为直流母线电压(V)，D(k)为k时刻开关管的占空比。
在一种可能的实施方式中，步骤2所述的引入反馈校正项，得到校正后第k+1、k+2时刻的电流预测值为：
Figure PCTCN2021094681-appb-000004
Figure PCTCN2021094681-appb-000005
其中λ 1、λ 2为校正系数，i(k|k-1)为k-1时刻预测k时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)，i(k+1|k)为k时刻预测k+1时刻的电流值(A)，i(k+2|k)为k时刻预测k+2时刻的电流值(A)，T S为采样周期(s)，L为负载的电感(H)，R为负载的电阻(Ω)，U dc为直流母线电压(V)，D(k)为k时刻开关管的占空比。
在一种可能的实施方式中，步骤3所述的价值函数为：
J＝(i*(k)-i(k+2|k)) 2          (6)
其中i*(k)为k时刻的电流参考值(A)，i(k+2|k)为k时刻预测k+2时刻的电流值(A)。
在一种可能的实施方式中，得到步骤3所述的最优调制占空比具体过程为：
首先，将k+2时刻的电流预测值代入价值函数中，得到：
Figure PCTCN2021094681-appb-000006
其次，求价值函数J对D的导数：
Figure PCTCN2021094681-appb-000007
最后，令J对D的导数等于零，求出占空比D即为最优占空比：
Figure PCTCN2021094681-appb-000008
其中占空比D的取值范围为：0≤D≤1。
第二方面，本申请实施例提供了一种双向数字开关功率放大器，该放大器由电流预测控制器，PWM调制器，移相电路，光电隔离电路，驱动电路，功率转换电路，电流霍尔传感器和负载组成；电流预测控制器的输入端连接电流给定信号和电流霍尔传感器反馈的负载中的电流反馈信号，电流预测控制器的输出端连接PWM调制器的输入端，PWM调制器的输出端连接移相电路的输入端，移相电路的输出端连接光电隔离信号的输入端，光电隔离信号的输出端连接驱动电路的输入端，驱动电路的四路输出端分别连接第一开关 管S1栅极端、第二开关管S2栅极端、第三开关管S3栅极端、第四开关管S4栅极端，第一开关管S1漏极端同时连接第三开关管S3漏极端和+U dc母线电压，第一开关管S1源极端同时连接第二开关管S2漏极端和电流霍尔传感器的一端，电流霍尔传感器的另一端与负载的一端串联，第三开关管S3源极端同时连接第四开关管S4漏极端和负载的另一端，第二开关管S2源极端同时连接第四开关管S4源极端和接地，电流霍尔传感器的输出端连接电流预测控制器的输入端。
在一种可能的实施方式中，功率转换电路由开关管S1，S2，S3，S4组成，均为功率MOS管(Metal Oxide Semiconductor Field Effect Transistor,金属氧化物半导体型场效应管)或IGBT(Insulated Gate Bipolar Transistor,绝缘栅双极型晶体管)，开关管S1、S2的驱动信号互补，开关管S3、S4的驱动信号互补，开关管S1、S4的驱动信号相位差180度，开关管S2、S3的驱动信号相位差180度。
在一种可能的实施方式中，双向数字开关功率放大器驱动的负载皆为感性负载。
有益效果：
1、本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，相对于传统PI控制，有效改善系统动态响应速度，减小动态调节时间。
2、本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，采用两步电流预测控制，有效补偿由于采样转换、算法计算、占空比更新所导致的单周期延时。
3、本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，采用闭环电流预测，有效提高系统控制精度。
4、本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，在线计算量小，算法简便，易于数字化实现。
5、本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，不需要整定复杂的参数、不需要调节繁琐的权重系数。
附图说明
为了更清楚地说明本申请实施例和现有技术的技术方案，下面对实施例和现有技术中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1是双向数字开关功率放大器的原理图；
图2是本申请磁悬浮驱动平台用的双向数字开关功率放大器多步电流预测控制框图；
图3是双向数字开关功率放大器功率转换电路中四个开关管的驱动波形图；
图4是采用PI控制功放输出1A电流的阶跃仿真波形图；
图5是采用本申请控制方法功放输出1A电流的阶跃仿真波形图；
图6是采用PI控制功放输出2A电流的阶跃仿真波形图；
图7是采用本申请控制方法功放输出2A电流的阶跃仿真波形图；
图8是采用PI控制功放输出5A电流的阶跃仿真波形图；
图9是采用本申请控制方法功放输出5A电流的阶跃仿真波形图。
具体实施方式
为使本申请的目的、技术方案、及优点更加清楚明白，以下参照附图并举实施例，对本申请进一步详细说明。显然，所描述的实施例仅仅是本申请一部分实施例，而不是全部的实施例。基于本申请中的实施例，本领域普通技术人员在没有作出创造性劳动前提下所获得的所有其他实施例，都属于本申请保护的范围。
本申请提供一种基于磁悬浮驱动平台用的双向数字开关功率放大器多步电流预测控制方法，包括如下步骤：
步骤1，建立双向数字开关功率放大器的预测模型，在第k个采样时刻，预测未来两个采样时刻(k+1、k+2)的电流预测值；
其中，k为正整数。双向数字开关功率放大器的预测模型的建立方式可以 参见相关技术中预测模型的建立，一个例子中，预测模型可以通过如下公式表示：
Figure PCTCN2021094681-appb-000009
利用建立好的双向数字开关功率放大器的预测模型，根据第k个采样时刻采集的数据，预测第k+1时刻及第k+2时刻的电流预测值。
一个例子中，首先可以预测第k+1时刻的电流预测值，然后利用第k+1时刻的电流预测值预测第k+2时刻的电流预测值。
步骤2，引入反馈校正项，闭环预测，得到校正后第k+1、k+2时刻的电流预测值；
对第k+1时刻及第k+2时刻的电流预测值进行校正，得到校正后第k+1、k+2时刻的电流预测值，此处的校正是为了引补偿模型失配、随机干扰等因素的影响，可以引入反馈校正项，从而实现校正。
步骤3，根据第k个采样时刻双向数字开关功率放大器的电流参考值以及引入反馈校正项后的第k+1、k+2个采样时刻的电流预测值，通过价值函数求取最优的调制占空比；
电流参考值为期望输出的电流值，可以根据实际情况自定义设置。引入反馈校正项后的第k+1、k+2个采样时刻的电流预测值即为步骤2中得到的校正后第k+1、k+2时刻的电流预测值；价值函数用于表示预测值与实际值之间的误差，一个例子中，价值函数选取k时刻的电流参考值与k+2时刻的电流预测值差的平方。
步骤4，根据步骤3得出的调制占空比，经脉宽调制模块分别产生四路PWM驱动信号来控制四个开关管，实现电流预测控制。
本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，相对于传统PI控制，有效改善系统动态响应速度，减小动态调节时间；采用两步电流预测控制，有效补偿由于采样转换、算法计算、占空比更新所导致的单周期延时；采用闭环电流预测，有效提高系统控制精度；在线计算量小，算法简便，易于数字化实现；不需要整定复杂的参数、不需要调节繁琐的权重系数。
本申请实施例中的双向数字开关功率放大器可以如图1所示，由电流预 测控制器，PWM调制器，移相电路，光电隔离电路，驱动电路，功率转换电路，电流霍尔传感器和负载组成。电流预测控制器的输入端连接电流给定信号(即电流参考值)和电流霍尔传感器反馈的负载中的电流反馈信号，电流预测控制器的输出端连接PWM调制器的输入端，PWM调制器的输出端连接移相电路的输入端，移相电路的输出端连接光电隔离电路的输入端，光电隔离电路的输出端连接驱动电路的输入端，驱动电路的四路输出端分别连接第一开关管S1栅极端、第二开关管S2栅极端、第三开关管S3栅极端、第四开关管S4栅极端，第一开关管S1漏极端同时连接第三开关管S3漏极端和+U dc母线电压，第一开关管S1源极端同时连接第二开关管S2漏极端和电流霍尔传感器的一端，电流霍尔传感器的另一端与负载的一端串联，第三开关管S3源极端同时连接第四开关管S4漏极端和负载的另一端，第二开关管S2源极端同时连接第四开关管S4源极端和接地，电流霍尔传感器的输出端连接电流预测控制器的输入端。负载为电感线圈，其电感为L、电阻为R，电流预测控制器根据电路给定信号和电流霍尔传感器反馈的负载中的电流反馈信号计算出最优占空比，然后输出控制信号经移相电路、光电隔离电路、驱动电路，控制四个开关管的导通时间及导通顺序，实现电流预测控制。
如图2所示为本申请磁悬浮驱动平台用的双向数字开关功率放大器多步电流预测控制框图，本申请磁悬浮驱动平台用的双向数字开关功率放大器多步电流预测控制方法，具体包括以下步骤：
步骤1，建立双向数字开关功率放大器的预测模型，在第k个采样时刻，预测未来两个采样时刻(k+1、k+2)的电流预测值；由基尔霍夫电压定律可得到图2中功率转换电路的电压关系，经离散化后得到双向数字开关功率放大器的预测模型：
Figure PCTCN2021094681-appb-000010
其中，i(k+1|k)为k时刻预测k+1时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)，T S为采样周期(s)，L为负载的电感(H)，R为负载的电阻(Ω)，U o(k)为k时刻双向数字开关功率放大器的输出电压(V)。
根据预测模型，在当前第k个采样时刻，预测未来两个采样时刻(k+1、k+2)的电流预测值；
Figure PCTCN2021094681-appb-000011
Figure PCTCN2021094681-appb-000012
其中，i(k+1|k)为k时刻预测k+1时刻的电流值(A)，i(k+2|k)为k时刻预测k+2时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)，T S为采样周期(s)，L为负载的电感(H)，R为负载的电阻(Ω)，U dc为直流母线电压(V)，D(k)为k时刻开关管的占空比。
步骤2，引入反馈校正项，实现闭环预测，对电流预测值进行实时修正，减小预测输出误差，补偿模型失配、随机干扰等因素的影响，增强系统的鲁棒性，提高控制效果，引入的反馈校正项通过k时刻的电流预测值与实际输出电流采样值做差所得：
λ[i(k)-i(k|k-1)]
其中λ为校正系数，i(k|k-1)为k-1时刻预测k时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)。
引入反馈校正项后，得到校正后第k+1、k+2时刻的电流预测值为：
Figure PCTCN2021094681-appb-000013
Figure PCTCN2021094681-appb-000014
其中λ 1、λ 2为校正系数，i(k|k-1)为k-1时刻预测k时刻的电流值(A)，i(k)为k时刻的输出电流采样值(A)。
控制的目标是为使电流参考值与实际输出电流值之间的误差最小，实际输出电流值能够稳定跟踪电流参考值，当采样周期小时，可以令k+2时刻的电流参考值近似等于k时刻的电流参考值，当k+2时刻电流预测值等于k+2时刻实际输出电流值时，说明电流预测效果准确，故价值函数选取k时刻的电流参考值与k+2时刻的电流预测值差的平方，得到：
J＝(i*(k)-i(k+2|k)) 2          (6)
其中i*(k)为k时刻的电流参考值(A)，i(k+2|k)为k时刻预测k+2时刻的电流值(A)。
步骤3，根据当前时刻双向数字开关功率放大器的电流给定值以及引入反馈校正项后的电流预测值，通过价值函数求取最优的调制占空比，求取过程具体为：
首先，将k+2时刻的电流预测值代入价值函数中，得到：
Figure PCTCN2021094681-appb-000015
其次，求价值函数J对D的导数：
Figure PCTCN2021094681-appb-000016
最后，令价值函数J对D的导数等于零，求出占空比D，由于i(k+1|k)是关于D(k)的一次函数，D(k)是关于J的一次函数，令J对D的导数等于零必有唯一解，所求的D即为价值函数的最小值点，即为最优占空比：
Figure PCTCN2021094681-appb-000017
其中占空比D的取值范围为：0≤D≤1。
步骤4，根据最优的调制占空比，经脉宽调制模块分别产生四路PWM驱动信号来控制四个开关管的导通时间及导通顺序，实现电流预测控制。
如图3所示为双向数字开关功率放大器功率转换电路中四个开关管的驱动波形图，步骤4中求得的最优占空比作为开关管S1的占空比用来驱动开关管S1，开关管S2的驱动信号与开关管S1的驱动信号互补，开关管S3和开关管S4的驱动信号由开关管S2和开关管S1的驱动信号移相180度得到，也可以理解为相移1/2周期。
磁悬浮驱动平台用的双向数字开关功率放大器驱动的负载皆为感性负载，开关频率固定。
为了验证本申请控制方法的有益效果，在Matlab/simulink中分别搭建采 用本申请控制方法和采用传统PI控制方法的开关功率放大器仿真模型，仿真参数为：母线电压70V，开关管开关频率20KHz，线圈电感9mH，线圈内阻1.65Ω，在2ms时输入1A、2A、5A的阶跃信号，图4为PI控制下功放输出1A电流的阶跃响应波形，动态调节时间大约为1.2ms，稳定时输出电流为0.98A，稳态误差为2％；图5为本申请控制方法下功放输出1A电流的阶跃响应波形，动态调节时间大约为0.6ms，稳定时输出电流为0.997A，稳态误差为0.3％；图6为PI控制下功放输出2A电流的阶跃响应波形，动态调节时间大约为1.5ms，稳定时输出电流为1.958A，稳态误差为2.1％；图7为本申请控制方法下功放输出2A电流的阶跃响应波形，动态调节时间大约为0.8ms，稳定时输出电流为1.994A，稳态误差为0.3％；图8为PI控制下功放输出5A电流的阶跃响应波形，动态调节时间大约为2.1ms，稳定时输出电流为4.89A，稳态误差为2.2％；图9为本申请控制方法下功放输出5A电流的阶跃响应波形，动态调节时间大约为1.6ms，稳定时输出电流为4.984A，稳态误差为0.32％；可以看出在相同条件下输出相同的电流等级，本申请控制方法比传统PI控制的动态调节时间短，稳态精度高。
本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，该控制方法相比于现有技术中开关功放所采用的PI控制，能够有效减小系统动态调节时间，提高功放电流响应速度，不需要整定复杂的参数，避免了大功率应用场合下，PI控制器参数整定调试存在的危险性。
本申请用于磁悬浮驱动平台的双向数字开关功率放大器多步电流预测控制方法，采用闭环电流预测，有效提高了系统控制精度，稳态误差小，采用两步电流预测控制，有效补偿由于采样转换、算法计算、占空比更新所导致的单周期延时，在线计算量小，算法简便，易数字化实现，具有良好的实用价值和应用前景。
以上所述仅为本申请的较佳实施例而已，并不用以限制本申请，凡在本申请的精神和原则之内，所做的任何修改、等同替换、改进等，均应包含在本申请保护的范围之内。

Claims (9)

  1. 一种双向数字开关功率放大器多步电流预测控制方法，包括如下步骤：
    步骤1，建立双向数字开关功率放大器的预测模型，在第k个采样时刻，预测未来两个采样时刻k+1、k+2的电流预测值；
    步骤2，引入反馈校正项，闭环预测，得到校正后第k+1、k+2时刻的电流预测值；
    步骤3，根据第k个采样时刻双向数字开关功率放大器的电流参考值以及引入反馈校正项后的第k+1、k+2个采样时刻的电流预测值，通过价值函数求取最优的调制占空比；
    步骤4，根据步骤3得出的调制占空比，经脉宽调制模块分别产生四路脉冲宽度调制PWM驱动信号来控制四个开关管，实现电流预测控制。
  2. 根据权利要求1所述双向数字开关功率放大器多步电流预测控制方法，其中，步骤1所述的预测模型为：
    Figure PCTCN2021094681-appb-100001
    其中，i(k+1|k)为k时刻预测k+1时刻的电流值，i(k)为k时刻的输出电流采样值，T S为采样周期，L为负载的电感，R为负载的电阻，U o(k)为k时刻双向数字开关功率放大器的输出电压。
  3. 根据权利要求1所述双向数字开关功率放大器多步电流预测控制方法，其中，步骤1所述的两个采样时刻k+1、k+2的电流预测值为：
    Figure PCTCN2021094681-appb-100002
    Figure PCTCN2021094681-appb-100003
    其中，i(k+1|k)为k时刻预测k+1时刻的电流值，i(k+2|k)为k时刻预测k+2时刻的电流值，i(k)为k时刻的输出电流采样值，T S为采样周期，L为负载的电感，R为负载的电阻，U dc为直流母线电压，D(k)为k时刻开关管的占空比。
  4. 根据权利要求1所述双向数字开关功率放大器多步电流预测控制方法，其中，步骤2所述的引入反馈校正项，得到校正后第k+1、k+2时刻的电流预 测值为：
    Figure PCTCN2021094681-appb-100004
    Figure PCTCN2021094681-appb-100005
    其中λ 1、λ 2为校正系数，i(k|k-1)为k-1时刻预测k时刻的电流值，i(k)为k时刻的输出电流采样值，i(k+1|k)为k时刻预测k+1时刻的电流值，i(k+2|k)为k时刻预测k+2时刻的电流值，TS为采样周期，L为负载的电感，R为负载的电阻，U dc为直流母线电压，D(k)为k时刻开关管的占空比。
  5. 根据权利要求1所述双向数字开关功率放大器多步电流预测控制方法，其中，步骤3所述的价值函数为：
    J＝(i*(k)-i(k+2|k)) 2
    其中i*(k)为k时刻的电流参考值，i(k+2|k)为k时刻预测k+2时刻的电流值。
  6. 根据权利要求1所述双向数字开关功率放大器多步电流预测控制方法，其中，得到步骤3所述的最优调制占空比具体过程为：
    首先，将k+2时刻的电流预测值代入价值函数中，得到：
    Figure PCTCN2021094681-appb-100006
    其次，求价值函数J对D的导数：
    Figure PCTCN2021094681-appb-100007
    最后，令J对D的导数等于零，求出占空比D即为最优占空比：
    Figure PCTCN2021094681-appb-100008
    其中占空比D的取值范围为：0≤D≤1。
  7. 一种权利要求1所述的双向数字开关功率放大器，该放大器由电流预测控制器，PWM调制器，移相电路，光电隔离电路，驱动电路，功率转换电 路，电流霍尔传感器和负载组成；电流预测控制器的输入端连接电流给定信号和电流霍尔传感器反馈的负载中的电流反馈信号，电流预测控制器的输出端连接PWM调制器的输入端，PWM调制器的输出端连接移相电路的输入端，移相电路的输出端连接光电隔离信号的输入端，光电隔离信号的输出端连接驱动电路的输入端，驱动电路的四路输出端分别连接第一开关管S1栅极端、第二开关管S2栅极端、第三开关管S3栅极端、第四开关管S4栅极端，第一开关管S1漏极端同时连接第三开关管S3漏极端和+U dc母线电压，第一开关管S1源极端同时连接第二开关管S2漏极端和电流霍尔传感器的一端，电流霍尔传感器的另一端与负载的一端串联，第三开关管S3源极端同时连接第四开关管S4漏极端和负载的另一端，第二开关管S2源极端同时连接第四开关管S4源极端和接地，电流霍尔传感器的输出端连接电流预测控制器的输入端。
  8. 根据权利要求7所述双向数字开关功率放大器，其中，所述的功率转换电路由开关管S1，S2，S3，S4组成，均为功率MOS管或IGBT，开关管S1、S2的驱动信号互补，开关管S3、S4的驱动信号互补，开关管S1、S4的驱动信号相位差180度，开关管S2、S3的驱动信号相位差180度。
  9. 根据权利要求8所述双向数字开关功率放大器，其中：双向数字开关功率放大器驱动的负载皆为感性负载。
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