WO2021208925A1 - 压电滤波器及其带外抑制改善方法、多工器、通信设备 - Google Patents

压电滤波器及其带外抑制改善方法、多工器、通信设备 Download PDF

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WO2021208925A1
WO2021208925A1 PCT/CN2021/087003 CN2021087003W WO2021208925A1 WO 2021208925 A1 WO2021208925 A1 WO 2021208925A1 CN 2021087003 W CN2021087003 W CN 2021087003W WO 2021208925 A1 WO2021208925 A1 WO 2021208925A1
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resonator
series
resonance
parallel
resonators
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PCT/CN2021/087003
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English (en)
French (fr)
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郑云卓
庞慰
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诺思(天津)微系统有限责任公司
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/02—Details
    • H03H9/02007—Details of bulk acoustic wave devices
    • H03H9/02086—Means for compensation or elimination of undesirable effects
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H9/00—Networks comprising electromechanical or electro-acoustic devices; Electromechanical resonators
    • H03H9/15—Constructional features of resonators consisting of piezoelectric or electrostrictive material
    • H03H9/205—Constructional features of resonators consisting of piezoelectric or electrostrictive material having multiple resonators

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  • the present invention relates to the technical field of filters, in particular to a piezoelectric filter and its out-of-band suppression improvement method, multiplexers and communication equipment.
  • the small-size filter devices that can meet the use of communication terminals are mainly piezoelectric filters.
  • the resonators that constitute this type of filter mainly include: FBAR (Film Bulk Acoustic Resonator), SMR (Solidly Mounted Resonator) , Solid-state assembly resonator), SAW (Surface Acoustic Wave, surface acoustic wave resonator).
  • Piezoelectric filters compared with common filters based on the principle of electromagnetic waves, have the characteristics of small size and high Q value of the resonator.
  • FBAR and SMR are collectively referred to as BAW devices (Bulk Acoustic Wave, bulk acoustic wave).
  • the suppression characteristics of the BAW technology reduces the RF performance of the BAW technology in low-frequency filter devices and limits its application range. How to suppress the deterioration of the filter caused by the high-order parasitic resonance problem of BAW in low-frequency applications has become an urgent problem for design engineers.
  • the present invention proposes a piezoelectric filter and its out-of-band suppression improvement method, multiplexer, and communication equipment, which help suppress or eliminate the pseudo-passband phenomenon in the third-order parasitic resonance region, thereby increasing the band of the filter. External suppression performance.
  • a method for improving the out-of-band suppression of a piezoelectric filter is provided.
  • the piezoelectric filter is a multi-step ladder structure, and each step ladder structure includes a series resonator and a parallel resonator.
  • the method includes: adjusting The thickness of one or more of the series resonator and parallel resonator in the ladder structure of at least 1 order, and/or, adjust the materials of all series resonators and/or parallel resonators, so that the The anti-resonance frequency corresponding to the high-order parasitic resonance of the parallel resonator is greater than the resonance frequency corresponding to the high-order parasitic resonance of the series resonator, or the difference between the two is less than the set value.
  • the thickness of the layer in the resonator includes the thickness of the mass load of the layer.
  • the layers in the resonator include: upper and lower electrodes, a piezoelectric layer, and a passivation layer.
  • the thickness of the piezoelectric layer of the parallel resonator is greater than the thickness of the piezoelectric layer of the series resonator.
  • the thickness of the lower electrode is the same or different, and the thickness of the upper electrode is different.
  • the step of adjusting the materials of all series resonators and/or parallel resonators includes: making the materials of the electrodes of all series resonators different from the materials of the electrodes of all parallel resonators; or, adjusting all the series resonators and/or Or the step of the material of the parallel resonator includes: making the material of the piezoelectric layer of all the series resonators different from the material of the piezoelectric layer of all the parallel resonators.
  • the series resonator and the parallel resonator are respectively arranged in two wafers stacked one above the other.
  • a piezoelectric filter which is manufactured using the method of the present invention.
  • a multiplexer including the piezoelectric filter of the present invention.
  • a communication device including the piezoelectric filter of the present invention.
  • the high-order parasitic resonance position of the parallel resonator is moved from a position much lower than the high-order parasitic resonance of the series resonator to a position equivalent or even higher. , Thereby suppressing or eliminating the pseudo-passband phenomenon in the high-order parasitic resonance region, thereby improving the out-of-band suppression performance of the filter.
  • Figure 1(a) is the electrical symbol of the BAW resonator
  • Figure 1(b) is the equivalent circuit of the BAW resonator
  • Figure 2 is a schematic diagram of the relationship between the thickness of the piezoelectric layer of the BAW and the resonant stress field;
  • Figure 3 is a schematic diagram of the relationship between the thickness of each layer of BAW and the resonance stress field
  • Figure 4 is a broadband impedance amplitude curve of a low-frequency FBAR resonator (hereinafter referred to as resonator 10) of about 836MHz;
  • FIG. 5 is an enlarged view of the impedance curve of the resonator 10 in the first resonance region
  • FIG. 6 is an enlarged view of the impedance curve of the resonator 10 in the second resonance region
  • FIG. 7 is an enlarged view of the impedance curve of the resonator 10 in the third resonance region
  • FIG. 8 is a schematic diagram of a ladder structure unit 100 composed of resonators 110 and 120;
  • Figure 9(a) is a schematic diagram of the impedance of the two resonators 110 and 120 when the frequency difference between the series and parallel resonators generated by the mass load is small, only 3MHz;
  • Fig. 9(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Fig. 9(a);
  • Figure 10(a) is a schematic diagram of the impedance of the two resonators when the frequency difference between the series and parallel resonators generated by the mass load is moderate, about 32MHz;
  • FIG. 10(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of FIG. 10(a);
  • Figure 11(a) is a schematic diagram of the impedance of the two resonators when the frequency difference between the series and parallel resonators generated by the mass load is moderate, about 32MHz;
  • Fig. 11(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Fig. 11(a);
  • Figure 12(a) is a schematic diagram of the impedance of the two resonators when the frequency difference between the series and parallel position resonators generated by the mass load is negative, such as -80MHz;
  • Fig. 12(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Fig. 12(a);
  • Figure 13 shows the broadband impedance curves of three different laminated low-frequency resonators
  • FIG. 14 is a comparative enlarged view of impedance curves of the resonators in the first resonance region in FIG. 13;
  • FIG. 15 is a comparative enlarged view of impedance curves of the resonators in the second resonance region in FIG. 13;
  • FIG. 16 is a comparative enlarged view of the impedance curves of the resonators in the third resonance region in FIG. 13;
  • Figure 17 (a) and Figure 17 (b) are for the ladder structure shown in Figure 10, when the mass load is moderate, the first resonant region forms a better bandpass shape, and its impedance in the third resonant region and Amplitude-frequency curve
  • FIG. 18 is a circuit diagram of a Band5 transmitting filter 300 in the embodiment of the present invention.
  • 19(a) and 19(b) are the amplitude-frequency response curves of a Band5 transmitting filter with the same circuit structure as the filter 300 in the prior art;
  • Figure 20(a) and Figure 20(b) are the amplitude-frequency response curves of the filter 300;
  • Figure 21 (a) is a schematic diagram of superimposing and comparing Figure 19 (a) and Figure 20 (a);
  • Figure 21(b) is a schematic diagram of superimposing and comparing Figure 19(b) and Figure 20(b);
  • 22 is a schematic side view of the stacked arrangement of series and parallel resonators of the embodiment.
  • Fig. 23 is a side view of the stacked arrangement of series and parallel resonators of the comparative example
  • Fig. 24 is a schematic diagram of broadband impedance curve relationships of series and parallel resonators according to an embodiment of the present invention.
  • FIG. 25 is a circuit diagram of a filter 500 according to an embodiment of the present invention.
  • FIG. 26 is a schematic side view of the filter 500
  • FIG. 27 is a front view of the upper wafer and the lower wafer of the filter 500;
  • Fig. 28 is a circuit diagram of a Band 5 duplexer according to an embodiment of the present invention.
  • Figure 1(a) is the electrical symbol of the BAW resonator
  • Figure 1(b) is the equivalent circuit of the BAW resonator.
  • the electrical model of the BAW resonator is simplified to a resonant circuit composed of L m , C m and C 0. It includes a static capacitor C 0 connected between the input and output ports, and a resonant branch connected in parallel with C 0 , and the resonant branch is connected in series with L m and C m.
  • the resonant circuit has two resonant frequency points: one is f s when the impedance of the resonant circuit reaches the minimum value, and f s is defined as the series resonant frequency of the resonator, or resonant frequency; the other is when f p resonance circuit when the impedance value reaches the minimum value, the f p is defined as the parallel resonance frequency of the resonator, also known as anti-resonant frequency.
  • the BAW resonator In the frequency range far away from the resonance frequency, the BAW resonator generally appears as a static capacitance C 0 .
  • This model is also called the BVD model, but it simply reflects the electrical characteristics near the main resonance frequency of the resonator, and the actual electrical response of the BAW is more complicated.
  • Figure 2 is a schematic diagram of the relationship between the thickness of the piezoelectric layer of the BAW and the resonant stress field. Assuming that a 2d-thick piezoelectric material sheet is sandwiched between two infinitely thin electrodes like a sandwich. Outside the electrodes is an ideal air boundary. At this time, this approximate sandwich structure is a simple BAW device model. The direction is perpendicular to the piezoelectric plane sheet, so the condition for the structure to resonate is: the acoustic wave excited in the piezoelectric layer by the alternating voltage applied by the electrode by the piezoelectric effect can form a stable standing wave, which is similar to the piezoelectric material. The speed of sound is related to the wave number of the standing wave, namely:
  • K 2 is the electromechanical coupling coefficient related to the material characteristics
  • n is the serial number of the resonance frequency, which is sequentially taken as 0, 1, 2, and so on.
  • n represents the resonance with the lowest frequency, which is also called fundamental resonance.
  • n 1
  • n 2 ⁇ f p,n
  • f p,n 2 ⁇ f p,n .
  • f p,0 is the aforementioned f p
  • f s,0 is the aforementioned f s . It can be seen from the above formula that the parallel resonant frequency of the resonator when n takes different values is determined by the thickness of the stack.
  • Figure 3 is a schematic diagram of the relationship between the thickness of each layer of the BAW and the resonance stress field after considering the thickness of the electrode on the basis of the above structure.
  • the thickness of the upper electrode and the lower electrode are both t, and the thickness of the piezoelectric layer is still 2d.
  • the electromechanical coupling coefficient of the resonator can be calculated by the following formula:
  • the piezoelectric materials that can be used in BAW devices include aluminum nitride, zinc oxide, etc.
  • the electrode metal materials that can be used include molybdenum, tungsten, aluminum, copper, and gold.
  • molybdenum is described as the electrode material.
  • a thin passivation layer is usually formed on the upper electrode.
  • the material of the passivation layer can be silicon dioxide, aluminum nitride, etc.
  • Fig. 4 is a broadband impedance amplitude curve of a low-frequency FBAR resonator (hereinafter referred to as resonator 10) with f s about 836MHz.
  • the corresponding series resonance frequency and parallel resonance frequency can be determined by the minimum and maximum impedance values within the range, and the electromechanical coupling coefficient of the corresponding resonance can also be calculated, and the actual device used Perform a test to get a specific impedance value.
  • the first resonance region is located in the vicinity of 800MHz to 900MHz
  • the second resonance region is located in the vicinity of 2.3GHz
  • the third resonance region is located in the range of 2.75GHz to 2.9GHz.
  • the first resonance zone is the main resonance or fundamental resonance of the resonator.
  • the series resonance impedance of the first resonance zone is smaller, and the parallel resonance impedance is larger, so the loss of the resonator is Smaller, higher Q value, is the main interval used to make the filter.
  • the resonance of the second resonance zone is obviously weaker, and the influence on the filter is small.
  • the third resonator's third parasitic resonance that is, the third resonant region
  • the third resonator's third parasitic resonance may also form a pseudo-passband shape similar to the first resonant region, thereby deteriorating the filter.
  • Out-of-band suppression characteristics in the corresponding frequency band It should be pointed out that for the stacked arrangement in this example, the parasitic resonance strength of the second resonance region is significantly weaker than that of the third resonance region, so the main solution is the pseudo passband generated by the third parasitic resonance. If for another layered arrangement, the spurious resonance intensity of the second resonance region is stronger, the method described in the present invention can also be used to improve the suppression performance.
  • FIG. 5 is an enlarged view of the impedance curve of the resonator 10 in the first resonance region, and the series resonance frequency in this region is f s, 0 , 836 MHz.
  • the parallel resonance frequency is f p ,0,866MHz.
  • the electromechanical coupling coefficient of the resonator in this interval is It was 8.13%.
  • the distance between f s,0 and f p,0 is about 30 MHz.
  • FIG. 6 is an enlarged view of the impedance curve of the resonator 10 in the second resonance region.
  • the series resonance frequency in this region is f s, 1 and f p, and 0 is around 2292 MHz.
  • the electromechanical coupling coefficient of the resonator in this interval is It is 0.027%.
  • FIG. 7 is an enlarged view of the impedance curve of the resonator 10 in the third resonance region.
  • the series resonance frequency in this region is f s, 2 , 2787 MHz.
  • the parallel resonance frequency is f p,2 ,2830MHz.
  • the electromechanical coupling coefficient of the resonator in this interval is It is 3.65%. It can be seen that the electromechanical coupling coefficient of the third parasitic resonance region is reduced by more than half compared to the first resonance region, but since the frequency has also become about three times higher, the distance between f s,2 and f p,2 is about 46MHz.
  • FIG. 8 is a schematic diagram of a ladder structure unit 100 composed of resonators 110 and 120.
  • This ladder structure unit is arranged between an input port and an output port (shown by the black dots in the figure), thus forming a two The radio frequency network of the port.
  • the filter usually contains two or more of the above-mentioned ladder-type structures connected in series, for example, as shown in FIG. 18.
  • the resonator 110 is located in the series position between the input port and the output port.
  • One end of the resonator 120 is connected to the output port and the other end is grounded.
  • the resonator 120 is In parallel position.
  • each layer of the resonator 110 and the resonator 120 are the same, but a certain thickness of mass load is added to the 120 to make the frequency lower than 110. Therefore,
  • the electromechanical coupling coefficients of the two resonators 110 and 120 are also basically the same. According to the difference in frequency between 110 and 120 caused by the mass load, the basic principle of the band-pass filter formed by the ladder structure is explained below.
  • the area of the resonators of 110 and 120 is specified to be the same, and only the first resonance region is used as an example for analysis.
  • Figure 9(a) is a schematic diagram of the impedance of the two resonators 110 and 120 when the frequency difference between the series and parallel resonators generated by the mass load is small, only 3MHz.
  • the impedance curves of 110 and 120 basically overlap
  • the impedance of the series and parallel positions in the ladder structure is basically the same in the entire frequency band, so only a part of the energy can pass.
  • Figure 9(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Figure 9(a).
  • the insertion loss is about 3dB
  • f s and f of 110 Near p transmission zero points are formed respectively, that is, suppression pits.
  • Figure 10(a) is a schematic diagram of the impedance of the two resonators when the frequency difference between the series and parallel resonators generated by the mass load is moderate, about 32MHz.
  • the so-called moderate frequency difference means that the f s of 110 is similar to the f p of 120, and the impedance curve forms the misalignment distribution of Fig. 10(a).
  • the frequency can be analyzed by selecting three characteristic positions according to the characteristics of the series and parallel resonators: 1) Near the f s of 120, the parallel impedance is very small at this time, the series impedance is large, and the signal cannot pass through the ladder.
  • the network forms the transmission zero point, which is the stopband suppression on the left side of the filter; 2) Near the f p of 120 or f s of 110, the parallel impedance is extremely large, the series impedance is extremely small, and the signal completely passes through the ladder network to form The transmission pole, that is, the passband of the filter; 3) Near the f p of 110, the parallel impedance is small at this time, and the impedance of the series resonator is extremely large. The signal cannot pass through the ladder network, forming a transmission zero point, which is the right side of the filter. Side stop band suppression.
  • Fig. 10(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Fig. 10(a).
  • Figure 10(b) is the shape of a simple band-pass filter, using multi-stage ladder-type structural units cascaded, and adding auxiliary inductors at appropriate positions, that is, a complex ladder-type band-pass filter can be formed.
  • Figure 11(a) is a schematic diagram of the impedance of the two resonators when the frequency difference between the series and parallel resonators generated by the mass load is relatively large, about 108MHz.
  • the so-called large frequency difference means that the difference between f s of 110 and f p of 120 is much greater than the distance between f s and f p of a single resonator, and the impedance curve forms the misalignment distribution of Fig. 11(a).
  • the impedance relationship analysis for different specific intervals is the same as before, and will not be repeated.
  • Figure 11(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Figure 11(a).
  • Fig. 12(b) is the S21 curve of the ladder structure 100 as a two-port network corresponding to the situation of Fig. 12(a). As shown by m2 in Fig. 12(b), it can be formed in the frequency range of about 50MHz. Stop band with 10dB suppression.
  • FIG. 13 shows the broadband impedance curves of three different laminated low-frequency resonators, which are called low-frequency resonators 200, 201, and 202 (not shown in the figure) for the convenience of description.
  • low-frequency resonators 200, 201, and 202 use A, B, and C to mark the first resonance area, the second resonance area, and the third resonance area.
  • the characteristics of the curves 200-202 are that the f s of the three resonators in the first resonance region are basically the same, which is about 835MHz.
  • the piezoelectric layer thickness of the resonator 200 is 0.74um
  • the thickness of the bottom electrode is 1.2um
  • the thickness of the top electrode is about 835MHz.
  • the thickness of the piezoelectric layer of the resonator 201 is 0.9um, the thickness of the bottom electrode is 1.1um, and the thickness of the top electrode is 1.015um; the thickness of the piezoelectric layer of the resonator 202 is 1.0um, and the thickness of the bottom electrode is 1.05um.
  • the electrode thickness is 0.95um. From the resonator 200 to the resonator 202, the thickness of the piezoelectric layer sequentially increases, and at the same time, in order to ensure that their resonant frequencies are the same, the thickness of the electrode layer needs to be sequentially decreased.
  • FIG. 14 is an enlarged view of the impedance curve comparison of each resonator in the first resonant region in FIG. 13. Since the thickness of the piezoelectric layer of the resonators 200 to 202 increases sequentially, and the resonant frequency of each resonator is the same, the It also increased sequentially, respectively 8.1%, 9.2% and 9.7%.
  • Fig. 15 is an enlarged view of the impedance curve comparison of the resonators in the second resonance region in Fig. 13. It can be seen that as the proportion of the thickness of the piezoelectric layer increases, although the f s of the resonators 200-202 in the first resonance region Basically the same, the frequencies of their secondary parasitic resonances also increase sequentially, to 2280MHz, 2437MHz and 2525MHz respectively.
  • FIG. 16 is an enlarged view of the impedance curve comparison of the resonators in the third resonance region in FIG. 13. It can be seen that as the proportion of the thickness of the piezoelectric layer increases, even though the resonators 200-202 are in the first resonance region, the f s Basically the same, the frequencies of their three parasitic resonances also increase sequentially, which are 2775MHz, 2927MHz and 3025MHz, respectively. They are also 3.67%, 3.3% and 3.06% respectively.
  • the impedance and amplitude-frequency curves in the third resonant region are shown in Figure 17(a) and Figure 17.
  • the frequency difference between the series and parallel resonators in the first resonance region is 32MHz, but due to the effect of the mass load and the high-order parasitic resonance
  • the overall effect of becoming smaller is that in the third resonance region, an impedance relationship similar to that in the first resonance region of Fig. 11 is formed, thereby forming a pseudo passband at the position of the third parasitic resonance, and its insertion loss is about 0.8dB. It also causes the deterioration of the suppression of the ladder structure at the position of the third parasitic resonance.
  • FIG. 18 is a circuit diagram of a Band5 transmitting filter 300 in the embodiment of the present invention, and its passband frequency range is 824 MHz to 849 MHz.
  • four series resonators TS1 to TS4 are connected in series between the first port and the second port.
  • the connection points of adjacent series resonators and the second port are connected with parallel resonators TP1 to PT4, respectively.
  • One end of the parallel resonators TP1 and TP2 are grounded via an inductor LG1, and the other ends of the parallel resonators TP3 and TP4 are grounded via an inductor LG2.
  • the thickness of the piezoelectric layer of all series position resonators is 0.74um
  • the thickness of the bottom electrode is 1.2um
  • the thickness of the top electrode is 1.1um
  • the thickness of the piezoelectric layer of all parallel resonators is 0.935um
  • the thickness of the bottom electrode is 0.935um.
  • the thickness of the upper electrode is 0.95um
  • only 0.014um mass load is added to TP1 and TP4 on the parallel resonator to adjust the return loss characteristics of the filter.
  • the area of all resonators is shown in Table 1, and the unit is um 2 .
  • Ts1 27700 Ts2: 15000 Ts3: 15000 Ts4: 15600 Tp1: 55000 Tp2: 55000 Tp3: 55000 Tp4: 35800
  • Figure 19(a) and Figure 19(b) are the amplitude-frequency response curves of a Band5 transmitting filter with the same circuit structure as the filter 300 in the prior art.
  • the filter adopts the prior art design ,
  • the piezoelectric layer thickness of all series resonators and parallel resonators is 0.74um
  • the thickness of the lower electrode is 1.2um
  • the thickness of the upper electrode is 1.1um.
  • the area of all resonators is shown in Table 2, and the unit is um 2 .
  • the area of the two filters needs to be optimized separately. Only when the overall passband and stopband performance of the filter are basically the same can the high-order parasitic resonance be compared. Suppressive performance of the area.
  • Figure 19(b) is a wideband response curve from 0.5GHz to 3.5GHz
  • Figure 19(a) is a detailed display of the passband characteristics of the same curve in the range of 820MHz to 850MHz.
  • the area 301 in Fig. 19(b) is the suppression spike caused by the secondary parasitic resonance of the resonator.
  • the 302 area is the suppression peak formed by the three parasitic resonances of the resonator.
  • Figure 20(a) and Figure 20(b) are the amplitude-frequency response curves of the filter 300.
  • Figure 20(a) is the wideband response curve from 0.5GHz to 3.5GHz, and Figure 20(b) is the same curve at 820MHz.
  • the passband characteristics in the range of ⁇ 850MHz are shown in detail.
  • the area 401 in FIG. 20(b) is the suppression spike caused by the second parasitic resonance of the resonator; the area 402 is the suppression spike formed by the third parasitic resonance of the resonator.
  • the position of the third parasitic resonance of the parallel resonator is moved from a position much lower than the position of the third parasitic resonance of the series resonator to a position equivalent to or even higher.
  • the parallel resonator and the series resonator three parasitic resonances form an impedance cancellation relationship similar to the principle of Figure 9, and even similar to the principle of Figure 12 to form a band-stop filter located in the tertiary parasitic resonance region, effectively eliminating the third parasitic resonance.
  • the spike is weakened, leaving only a 403 suppressed spike of about 30dB and a 404 suppressed spike of about 25dB, which is a significant improvement compared to Figure 19.
  • 403 corresponds to the third parasitic resonance of the series resonator
  • 404 corresponds to the third parasitic resonance of the parallel resonator.
  • Figure 21 compares the curves of the example and the comparative example.
  • Figure 21(b) is the broadband response curve from 0.5GHz to 3.5GHz
  • Figure 21(a) is the same curve in the range of 820MHz to 850MHz.
  • the passband characteristics are shown in detail.
  • the thick solid line in the figure is the amplitude-frequency response curve of the filter of the embodiment
  • the thin dashed line is the amplitude-frequency response curve of the filter for comparison. It can be seen from Figure 21(b) that due to the increase in the proportion of the piezoelectric layer thickness of the parallel resonator in the embodiment, the resonance spike associated with the parallel resonator moves from the position of 303 to the position of 404, and both resonance peaks are obtained. Effective suppression.
  • the passband can be widened and the insertion loss can be improved, the relationship between the frequency difference between the series and parallel resonators at the third parasitic resonance has not been substantially changed, so the suppression of spikes at the third parasitic resonance It will only move in the direction of high frequency as the stack changes, but the height of the suppression peak will not be effectively weakened.
  • the relative frequency range of the secondary parasitic resonance of the resonator is relatively narrow, the intensity is not strong, and the overall suppression of the filter is not significant.
  • the second parasitic resonance is similar to the third parasitic resonance, which has an adverse effect on suppression, the same principle can be used to design and avoid.
  • the strength of the high-order (ie, secondary and above) parasitic resonance is sufficient to affect the performance of the filter, then the method provided by the embodiment of the present invention can be used to suppress it.
  • FIG. 22 is a schematic side view of the stacked arrangement of series and parallel resonators of the embodiment, the thickness of the lower electrode of both is 1.2um, the piezoelectric layer thickness t1 of the series resonator is 0.74um, and the piezoelectric layer thickness t2 of the parallel resonator is 0.935um At the same time, the thickness of the upper electrode of the two is different.
  • This method is different from the structure of the comparative example shown in FIG. 23.
  • FIG. 23 is a side view of the stacked arrangement of series and parallel resonators of the comparative example.
  • the f s of the series resonator is basically aligned with the f p of the parallel resonator to achieve better filtering. Pass-band characteristics of the device; 2) In the third resonance region, the frequency difference between f p,2 of the parallel resonator and f s,2 of the series resonator is within the range of -X to 3X, where X is a series resonance The difference between f p,2 and f s,2 of the filter to achieve better suppression characteristics of the tertiary parasitic resonance region.
  • the frequency difference between f p,2 of the parallel resonator and f s,2 of the series resonator is X.
  • FIG. 25 is a circuit diagram of a filter 500 according to an embodiment of the present invention
  • FIG. 26 is a schematic side view of the filter 500
  • FIG. 27 is the upper crystal of the filter 500 Front view of the circle and the lower wafer.
  • a parallel resonator is fabricated in the upper wafer 501 (the thin film bulk acoustic resonator FBAR is taken as an example in the figure), and a series resonator is fabricated in the lower wafer 502.
  • P1-P6 are parallel resonators
  • S1-S6 are series resonators
  • VIN is the input pin
  • VOUT is the output pin
  • VG1 and VG2 are ground pins
  • both wafers are arranged with VIN and VG1 , VG2, VOUT these four bonding areas.
  • a new bonding area is added, namely J1, J2, J3 in the figure. This new bonding area is only used to connect the upper wafer 501 and the lower wafer 502 together, without the need to pass the via to the outside of the chip. Connection, so its shape is different from the bonding area of the connection via.
  • Fig. 28 is a circuit diagram of a Band 5 duplexer adopting the present invention, in which the frequency of the transmitting filter Tx is 824 MHz to 849 MHz, and the frequency range of the receiving filter Rx is 869 MHz to 894 MHz. Both filters adopt the present invention to realize the improvement of high-order parasitic resonance suppression.
  • the effect of adjusting the thickness of each layer of the resonator to suppress the third-order spurious resonance is mainly explained, and it can also be used to suppress the high-order spurious resonance.
  • the series resonator and the parallel resonator may use different materials, so that their high-order parasitic resonances conform to the frequency relationship described above.
  • the adjustment of the thickness of each layer can also be combined.
  • the so-called different materials can be different electrode materials, such as molybdenum, ruthenium, gold, aluminum, magnesium, tungsten, copper, titanium, iridium, osmium, chromium, or composites or alloys of the above metals, or piezoelectric layers
  • the materials are different, such as aluminum nitride, zinc oxide, PZT (lead zirconate titanate series piezoelectric ceramics) and other materials and contain rare earth element doped materials with a certain atomic ratio of the above materials.
  • the electrode material used in the series-parallel resonator is molybdenum
  • the piezoelectric layer material is all scandium-doped aluminum nitride.
  • the doping concentration of the series resonator is different from that of the parallel resonator.
  • the impurity concentration is also different.
  • the scandium doping concentration of the series resonator can be set to about 7%.
  • the scandium doping concentration of the parallel resonator is about 10%.
  • the frequency difference X between f p,2 of the parallel resonator and f s,2 of the series resonator, and f p,2 of a series resonator and The frequency difference of f s,2 is equivalent, so its suppression near the third-order spurious resonance frequency is improved.
  • the position of the third parasitic resonance of the parallel resonator is moved from the position much lower than the position of the third parasitic resonance of the series resonator. To a position comparable or even higher, thereby suppressing or eliminating the pseudo-passband phenomenon in the third-order parasitic resonance region, thereby improving the out-of-band suppression performance of the filter.

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Abstract

一种改善压电滤波器带外抑制的方法，所述压电滤波器为梯形结构（100），每阶的梯形结构（100）包含1个串联谐振器（110）和1个并联谐振器（120），该方法包括：调节至少1阶的梯形结构（100）中的串联谐振器（110）和并联谐振器（120）中的一层或多层的厚度，并且/或者，调节所有串联谐振器（110）和/或并联谐振器（120）的材料，从而使该梯形结构（100）中的并联谐振器（120）的高次寄生谐振对应的反谐振频率大于串联谐振器（110）高次寄生谐振对应的谐振频率，或者二者之间的差值小于设定值。通过调整谐振器（110,120）中的层的厚度，使得并联谐振器（120）的高次寄生谐振位置从远低于串联谐振器（110）高次寄生谐振的位置，移动到与其相当甚至更高的位置，由此抑制或消除高次寄生谐振区的伪通带现象，从而提高滤波器的带外抑制性能。

Description

压电滤波器及其带外抑制改善方法、多工器、通信设备 技术领域
本发明涉及滤波器技术领域，特别地涉及一种压电滤波器及其带外抑制改善方法、多工器、通信设备。
背景技术
近年来，移动通信行业蓬勃发展，5G逐渐开始走入人们的生活，以手机为代表的移动通信终端成为人们日常生活中必不可少的通信工具。随着频谱资源的日益拥挤，对终端的数据吞吐量和功耗要求越来越高我，这给射频前端电路的设计带来了巨大挑战。
目前，能够满足通讯终端使用的小尺寸滤波类器件主要是压电滤波器，构成此类滤波器的谐振器主要包括：FBAR(Film Bulk Acoustic Resonator，薄膜体声波谐振器)，SMR(Solidly Mounted Resonator，固态装配谐振器)，SAW(Surface Acoustic Wave，表面声波谐振器)。压电滤波器，相比常见的基于电磁波原理的滤波器，具有尺寸小，谐振器Q值高的特点。其中，FBAR和SMR又合称为BAW器件(Bulk Acoustic Wave，体声波)。在2.5GHz～3.5GHz频率范围内，因为对制作SAW谐振器基础结构IDT(Inter digital transducer)的光刻精度较高导致制作困难，谐振器Q值不高，而在0.5GHz～1.5GHz范围内，SAW具有一定成本上的优势。相比之下，BAW滤波器则具有更高的频率应用范围(1GHz～10GHz)，更小的插入损耗，以及更好的静电释放(ESD)及功率耐受(Power handling)能力。但是当BAW的频率向低频扩展到800MHz甚至更低时，因为BAW结构特有三明治结构，会在3倍基础谐振频率的附近产生较强的高次寄生谐振，影响低频滤波器在高次寄生谐振区域的抑制特性，从而降低了BAW技术在低频滤波类器件中的射频性能，限制其应用范围。如何BAW在低频应用时因高次寄生谐 振问题产生的滤波器抑制恶化，成为设计工程师一个亟待解决的问题。
发明内容
有鉴于此，本发明提出一种压电滤波器及其带外抑制改善方法、多工器、通信设备，有助于抑制或消除三次寄生谐振区的伪通带现象，从而提高滤波器的带外抑制性能。
为实现上述目的，根据本发明的第一方面，提供了一种改善压电滤波器带外抑制的方法。
本发明的改善压电滤波器带外抑制的方法，所述压电滤波器为多阶的梯形结构，每阶的梯形结构包含1个串联谐振器和1个并联谐振器，该方法包括：调节至少1阶的梯形结构中的串联谐振器和并联谐振器中的一层或多层的厚度，并且/或者，调节所有串联谐振器和/或并联谐振器的材料，从而使该梯形结构中的并联谐振器的高次寄生谐振对应的反谐振频率大于串联谐振器高次寄生谐振对应的谐振频率，或者二者之间的差值小于设定值。
可选地，所述谐振器中的层的厚度包含该层的质量负载的厚度。
可选地，所述谐振器中的层包括：上、下电极，压电层，钝化层。
可选地，所述该梯形结构中，并联谐振器的压电层厚度大于串联谐振器的压电层厚度。
可选地，对于所述该梯形结构中的并联谐振器和串联谐振器，下电极厚度相同或不同，上电极厚度不同。
可选地，所述设定值的范围是-X～3X之间，其中X为所述1阶的梯形结构中的串联谐振器的高次寄生反谐振频率f p,n与相应的高次寄 生谐振频率f s,n之差，即X＝f p,n-f s,n。
可选地，调节所有串联谐振器和/或并联谐振器的材料的步骤包括：使所有串联谐振器的电极的材料与所有并联谐振器的电极的材料不同；或者，调节所有串联谐振器和/或并联谐振器的材料的步骤包括：使所有串联谐振器的压电层的材料与所有并联谐振器的压电层的材料不同。
可选地，所述滤波器中，串联谐振器和并联谐振器分别设置在上下层叠的2个晶圆中。
根据本发明的第二方面，提供了一种压电滤波器，该压电滤波器是使用本发明所述的方法制造。
根据本发明的第三方面，提供了一种多工器，其包含本发明中的压电滤波器。
根据本发明的第四方面，提供了一种通信设备，其包含本发明中的压电滤波器。
根据本发明的技术方案，通过调整谐振器中的层的厚度，使得并联谐振器的高次寄生谐振位置从远低于串联谐振器高次寄生谐振的位置，移动到与其相当甚至更高的位置，由此抑制或消除高次寄生谐振区的伪通带现象，从而提高滤波器的带外抑制性能。
附图说明
为了说明而非限制的目的，现在将根据本发明的优选实施例、特别是参考附图来描述本发明，其中：
图1(a)是BAW谐振器电学符号；
图1(b)是BAW谐振器的等效电路；
图2是BAW的压电层厚度与谐振应力场关系示意图；
图3是BAW的各层厚度与谐振应力场关系示意图；
图4是一个约为836MHz的低频FBAR谐振器(以下称作谐振器10)的宽频带阻抗幅值曲线；
图5是谐振器10在第一谐振区的阻抗曲线放大图；
图6是谐振器10在第二谐振区的阻抗曲线放大图；
图7是谐振器10在第三谐振区的阻抗曲线放大图；
图8是由谐振器110和120组成的梯型结构单元100的示意图；
图9(a)是当质量负载产生的串、并联谐振器的频率差较小，仅为3MHz时，两个谐振器110、120的阻抗示意图；
图9(b)是对应于图9(a)的情况的梯型结构100作为二端口网络的S21曲线；
图10(a)是当质量负载产生的串、并联谐振器的频率差适中，约为32MHz时，两个谐振器的阻抗示意图；
图10(b)是对应于图10(a)的情况的梯型结构100作为二端口网络的S21曲线；
图11(a)是当质量负载产生的串、并联谐振器的频率差适中，约为32MHz时，两个谐振器的阻抗示意图；
图11(b)是对应于图11(a)的情况的梯型结构100作为二端口网络的S21曲线；
图12(a)是当质量负载产生的串、并联位置谐振器的频率差为负值，如-80MHz时，两个谐振器的阻抗示意图；
图12(b)是对应于图12(a)的情况的梯型结构100作为二端口网络的S21曲线；
图13展示了三种不同层叠设置的低频谐振器的宽频带阻抗曲线；
图14是图13中各谐振器在第一谐振区的阻抗曲线对比放大图；
图15是图13中各谐振器在第二谐振区的阻抗曲线对比放大图；
图16是图13中各谐振器在第三谐振区的阻抗曲线对比放大图；
图17(a)和图17(b)是对于图10所示的梯型结构，在质量负载适中，第一谐振区形成较好带通波器形状时，其在第三谐振区的阻抗和幅频曲线；
图18是本发明的实施方式中的一个Band5发射滤波器300的电路图；
图19(a)和图19(b)是现有技术中，一个与滤波器300具有相同电路结构的Band5发射滤波器的幅频响应曲线；
图20(a)和图20(b)是滤波器300的幅频响应曲线；
图21(a)是将图19(a)和图20(a)进行叠加对比的示意图；
图21(b)是将图19(b)和图20(b)进行叠加对比的示意图；
图22是实施例的串、并联谐振器层叠设置的侧面示意图；
图23是对比例的串、并联谐振器层叠设置的侧面示意图；
图24根据本发明实施方式的串、并联谐振器的宽频阻抗曲线关系示意图；
图25是本发明实施方式涉及的一种滤波器500的电路图；
图26是滤波器500的侧面示意图；
图27是滤波器500的上晶圆和下晶圆的主视图；
图28是根据本发明实施方式的一个Band 5双工器的电路图。
具体实施方式
下面结合附图与实施例对本发明作进一步说明。
图1(a)是BAW谐振器电学符号，图1(b)是BAW谐振器的等效电路。在不考虑损耗项的情况下，BAW谐振器的电学模型简化为L m，C m和C 0组成的谐振电路。其中包含一个连接在输入和输出端口之间的静态电容C 0，还有一个与C 0并联的谐振支路，谐振支路由L m和C m串联。根据谐振条件可知，该谐振电路存在两个谐振频点：一个是谐振电路阻抗值达到最小值时的f s，将f s定义为该谐振器的串联谐振频率，或称谐振频率；另一个是当谐振电路阻抗值达到最小值时的f p，将f p定义为该谐振器的并联谐振频率，或称反谐振频率。其中，
Figure PCTCN2021087003-appb-000001
在远离谐振频点的频率范围内，BAW谐振器总体上表现为静态电 容C 0。此模型也被称作是BVD模型，但是它只是简单反映出谐振器主谐振频率附近的电学特性，实际的BAW的电学响应更为复杂。
图2是BAW的压电层厚度与谐振应力场关系示意图。假定厚度为2d的压电材料薄片，如同三明治一样被夹在上下两片无限薄的电极之间，电极以外是理想的空气边界，此时这个近似的三明治结构就是一个简单的BAW器件模型，电场方向为垂直于压电平面薄片，因此该结构发生谐振的条件是：由压电效应被电极施加的交变电压在压电层中激发出的声波能够形成稳定的驻波，这与压电材料的声速以及驻波的波数有关，即：
Figure PCTCN2021087003-appb-000002
其中，K 2是与材料特性相关的机电耦合系数，n为谐振次数的序号，依次取值为0，1，2，等等。n为0时，代表频率最低的谐振，也称作基础谐振，n为1时定义为二次寄生谐振，n为3时定义为三次寄生谐振，以此类推。ω a,n为n作相应取值时的反谐振角频率，单位为rad/s，其与频率的关系为ω a,n＝2·π·f p,n。f p,n即为对应n取值下的并联谐振频率。f p,0即为前面提到的f p，同理，f s,0也就是前面提到的f s。由上式可以看出，谐振器在n取不同值时的并联谐振频率，是由层叠厚度决定的。
图3是在上述结构的基础上，考虑了电极厚度之后，BAW的各层厚度与谐振应力场关系示意图。其中上电极和下电极的厚度均为t，压电层的厚度仍为2d。谐振器的机电耦合系数，可以通过下面的公式计算出来：
Figure PCTCN2021087003-appb-000003
并且，对于n＝0的情况，f s,0和f p,0以及谐振器的机电耦合系数
Figure PCTCN2021087003-appb-000004
满足以下关系：
Figure PCTCN2021087003-appb-000005
其中
Figure PCTCN2021087003-appb-000006
是n作相应取值时的谐振器的机电耦合系数，它与电极厚度与总层叠厚度的比值t/(d+t)有关，在谐振频率固定的条件下，压电层厚度越厚，谐振器的机电耦合系数也越大，f s,n和f p,n之间的相对距离也越远。从上面的公式也可以得出，当f s,0一定时，
Figure PCTCN2021087003-appb-000007
变大，会导致f p,0向高频移动，对于n取其它数值，也是同样的变化趋势。
需要说明的是，BAW器件可采用的压电材料有氮化铝，氧化锌等，可采用的电极金属材料有钼，钨，铝，铜，金等，本发明以下均以氮化铝作为压电材料，钼作为电极材料进行说明。为了保护上电极不受环境影响而发生氧化或腐蚀，通常会在上电极的上方再制作一层较薄的钝化层，钝化层的材料可以是二氧化硅，氮化铝等。
图4是一个f s约为836MHz的低频FBAR谐振器(以下称作谐振器10)的宽频带阻抗幅值曲线，图中以虚线框出的A区域，即为n＝0的第一谐振区；虚线框出的B区域，即为n＝1的第二谐振区；虚线框出的C区域，即为n＝2的第三谐振区。在这三个谐振区范围内，均可以通过范围内阻抗极小值和极大值确定相应的串联谐振频率与并联谐振频率，也可以计算出相应谐振的机电耦合系数，以及对实际采用的器件进行测试得出具体阻抗值。从图中可以看出，第一谐振区位于800MHz～900MHz范围附近，第二谐振区位于2.3GHz附近，而第三谐振区位于2.75GHz～2.9GHz范围内。第一谐振区是谐振器的主谐振或称基础谐振，与其他两个范围内的谐振阻抗相比，第一谐振区的串联谐振阻抗较小，并联谐振阻抗较大，因此其谐振器的损耗较小，Q值较高，是用来制作滤波器的主要区间。对于本例，第二谐振区的谐振明显较弱，对滤波器的影响较小。第三谐振区，虽然损耗相比第一谐振区损耗大了很多，但是总体上仍可以看作是一个频率在2.75GHz～2.9GHz的谐振器。因此当利用第一谐振区制作了梯型结构滤波器后，相应谐振器的三次寄生谐振，也就是第三谐振区，也可能会形成类似第一谐振区的伪通带形状，从而恶化滤波器在相应频段的带外抑制特性。需要指出的是，对于本例中的层叠设置，第二谐振区的 寄生谐振强度明显弱于第三谐振区，因此主要解决是的三次寄生谐振产生的伪通带。如果对于另外一种层叠设置，第二谐振区的寄生谐振强度更强，同样可以用本发明阐述的方法改善抑制性能。
图5是谐振器10在第一谐振区的阻抗曲线放大图，在此区域的串联谐振频率为f s,0，836MHz。并联谐振频率为f p,0，866MHz。此段区间的谐振器机电耦合系数为
Figure PCTCN2021087003-appb-000008
为8.13％。f s,0与f p,0之间的距离约为30MHz。
图6是谐振器10在第二谐振区的阻抗曲线放大图，在此区域的串联谐振频率为f s,1和f p,0均在2292MHz附近。此段区间的谐振器机电耦合系数为
Figure PCTCN2021087003-appb-000009
为0.027％。
图7是谐振器10在第三谐振区的阻抗曲线放大图，在此区域的串联谐振频率为f s,2，2787MHz。并联谐振频率为f p,2，2830MHz。此段区间的谐振器机电耦合系数为
Figure PCTCN2021087003-appb-000010
为3.65％。可以看到，三次寄生谐振区的机电耦合系数相比第一谐振区降低了超过一半，但是由于频率也变高到三倍左右，因此f s,2与f p,2之间的距离约为46MHz。
图8是由谐振器110和120组成的梯型结构单元100的示意图，这个梯型结构单元被设置在一个输入端口与一个输出端口(图中黑点所示)之间，因此形成了一个二端口的射频网络。滤波器中，通常包含2个或者更多的串联的上述梯型结构，例如图18所示。在图8中，谐振器110位于输入端口与输出端口之间的串联位置，谐振器120一端接在输出端口，另一端接地，在2个或更多的串联的梯形结构中，谐振器120则处于并联位置。
通常情况下，谐振器110和谐振器120的各层厚度，如上、下电极，压电层等，均相同，但120上会添加一定厚度的质量负载，以使其频率低于110，因此，110和120两个谐振器的机电耦合系数也基本相同。以下根据质量负载所产生的110和120之间频率差异的不同，阐述梯型结构形成带通滤波器的基本原理。为了分析方便，这里指定110和120的谐振器面积相同，并且仅以第一谐振区为例进行分析。
如图9(a)，是当质量负载产生的串、并联谐振器的频率差较小，仅为3MHz时，两个谐振器110、120的阻抗示意图，此时110和120 的阻抗曲线基本重叠，根据分压原理，梯型结构中串、并联位置的阻抗在整个频段都基本相同，因此只有一部分能量可以通过。图9(b)是对应于图9(a)的情况的梯型结构100作为二端口网络的S21曲线，在整个第一谐振区间，插入损耗均为3dB左右，同时在110的f s和f p附近，分别形成了传输零点，也就是抑制尖坑。
如图10(a)，是当质量负载产生的串、并联谐振器的频率差适中，约为32MHz时，两个谐振器的阻抗示意图。所谓频率差适中，是指110的f s和120的f p相差不多，阻抗曲线形成图10(a)的错位分布。同样根据分压原理，可以将频率根据串、并联谐振器的特点选取三个特征位置进行分析：1)120的f s附近，此时并联阻抗极小，串联阻抗较大，信号无法通过梯型网络，形成传输零点，也就是滤波器的左侧阻带抑制；2)120的f p或者110的f s附近，此时并联阻抗极大，串联阻抗极小，信号完全通过梯型网络，形成传输极点，也就是滤波器的通带；3)110的f p附近，此时并联阻抗较小，串联谐振器阻抗极大，信号无法通过梯型网络，形成传输零点，也就是滤波器的右侧阻带抑制。图10(b)是对应于图10(a)的情况的梯型结构100作为二端口网络的S21曲线，在整个第一谐振区间，插入损耗均为0.2dB左右，同时在120的f s和110的f p附近，分别形成了抑制尖坑。此时图10(b)就是一个简单带通滤波器的形状，采用多级梯型结构单元级联，并在适当位置添加辅助电感，即可以形成复杂的梯型带通滤波器。
如图11(a)，是当质量负载产生的串、并联谐振器的频率差较大，约为108MHz时，两个谐振器的阻抗示意图。所谓频率差较大，是指110的f s和120的f p相差远大于单个谐振器的f s和f p的距离，阻抗曲线形成图11(a)的错位分布。对于不同特定区间的阻抗关系分析与前面一样，不再赘述。图11(b)是对应于图11(a)的情况的梯型结构100作为二端口网络的S21曲线，由于串、并联谐振器的频率差(以质量负载衡量)，远大于单个谐振器的f s和f p的距离(以谐振器的机电耦合系数衡量)，因此会如图11(b)那样，在110的f s和120的f p位置分别形成一个类似通带的插入损耗尖峰，或者形象点描述，更像是在10(a)的基础上由于谐振器频率差的进一步增大，把原来的通带拉 伸为两个传输极点。
接下来讨论一种相对特殊的情况，即如图12(a)所示，是当质量负载产生的串、并联位置谐振器的频率差为负值，如-80MHz时两个谐振器的阻抗示意图。需要说明的是，由于前面设定，质量负载添加在并联谐振器120上，这样质量负载就变成了负数，而实际中并不能实现，以上条件实际上是通过将质量负载加在串联位置的谐振器来等效实现的。这里采用质量负载为负的表述，是为了理解方便。对此条件下的梯型结构进行阻抗分析，则情况与图10正好相反，在110的f s和120的f p之间的这段频率，并不会形成一个通带，而是会形成一个阻带。图12(b)是对应于图12(a)的情况的梯型结构100作为二端口网络的S21曲线，如图12(b)中的m2所示，可以在大概50MHz的频率范围内形成具有10dB抑制的阻带。
对比图12和图10可以得到一个初步的结论，对于梯型结构10，当串、并联谐振的频率差为正数且适中时，梯型结构表现为带通滤波器，当串、并联谐振的频率差为负数时，梯型结构会表现为带阻滤波器。
图13展示了三种不同层叠设置的低频谐振器的宽频带阻抗曲线，为描述方便，分别称作低频谐振器200，201和202(图中不示出)。同前面一样，分别用A，B，C标出了第一谐振区，第二谐振区，第三谐振区。曲线200～202的特点在于，三个谐振器在第一谐振区的f s基本相同，均为835MHz左右，谐振器200的压电层厚度为0.74um，下电极厚度为1.2um，上电极厚度为1.125um；谐振器201的压电层厚度为0.9um，下电极厚度为1.1um，上电极厚度为1.015um；谐振器202的压电层厚度为1.0um，下电极厚度为1.05um，上电极厚度为0.95um。从谐振器200到谐振器202，压电层的厚度依次增加，同时为了保证它们的谐振频率相同，需要将电极层的厚度依次降低。
图14是图13中各谐振器在第一谐振区的阻抗曲线对比放大图，由于谐振器200～202压电层厚度占比依次增加，且各谐振器的谐振频率相同，因此各谐振器的
Figure PCTCN2021087003-appb-000011
也依次增加，分别为8.1％，9.2％和9.7％。
图15是图13中各谐振器在第二谐振区的阻抗曲线对比放大图， 可以看到，随着压电层厚度占比的增加，尽管谐振器200～202在第一谐振区的f s基本相同，它们二次寄生谐振的频率也依次增加，分别为2280MHz，2437MHz和2525MHz。
图16是图13中各谐振器在第三谐振区的阻抗曲线对比放大图，可以看到，随着压电层厚度占比的增加，尽管谐振器200～202在第一谐振区的f s基本相同，它们三次寄生谐振的频率也依次增加，分别为2775MHz，2927MHz和3025MHz，
Figure PCTCN2021087003-appb-000012
也分别为3.67％，3.3％和3.06％。
从以上三张图的对比得知：可以通过增大压电层厚度的占比，使得谐振器在f s不变的前提下，其三次寄生谐振频率f s,2向高频移动。
对于图10所示的梯型结构，在质量负载适中，第一谐振区形成较好带通波器形状时，其在第三谐振区的阻抗和幅频曲线如图17(a)和图17(b)所示，此时串、并联谐振器在第一谐振区的频率差为32MHz，但由于质量负载的效应，以及高次寄生谐振
Figure PCTCN2021087003-appb-000013
变小的综合影响，其在第三谐振区，形成了类似于图11第一谐振区的阻抗关系，从而在三次寄生谐振的位置，形成一个伪通带，其插入损耗约为0.8dB，这也造成了梯型结构在三次寄生谐振位置抑制的恶化。
图18是本发明的实施方式中的一个Band5发射滤波器300的电路图，其通带频率范围是824MHz～849MHz。如图18所示，第1端口和第2端口之间串联有TS1至TS4这4个串联谐振器，相邻的串联谐振器的连接点和第2端口，分别连接有并联谐振器TP1至PT4的一端，并联谐振器TP1和TP2另一端经由电感LG1接地，并联谐振器TP3和TP4另一端经由电感LG2接地。滤波器300中，所有串联位置谐振器的压电层厚度为0.74um，下电极厚度均为1.2um，上电极厚度为1.1um；所有并联谐振器的压电层厚度为0.935um，下电极厚度均为1.2um，上电极厚度为0.95um，并联谐振器上只在TP1和TP4上添加0.014um的质量负载，用于调节滤波器的回波损耗特性。所有谐振器的面积如表1所示，单位为um 2。
表1
Ts1：27700 Ts2：15000 Ts3：15000 Ts4：15600
Tp1：55000 Tp2：55000 Tp3：55000 Tp4：35800
图19(a)和图19(b)是现有技术中，一个与滤波器300具有相同电路结构的Band5发射滤波器的幅频响应曲线，作为对比例，该滤波器采用了现有技术设计，所有串联谐振器和并联谐振器的压电层厚度均为0.74um，下电极厚度均为1.2um，上电极厚度为1.1um，同时，在所有并联位置的谐振器的上电极上，还添加有厚度为0.14um的质量负载层，质量负载与电极采用同样的金属材料。所有谐振器的面积如表2所示，单位为um 2。
表2
Ts1：31500 Ts2：15000 Ts3：15000 Ts4：16200
Tp1：48170 Tp2：44180 Tp3：46160 Tp4：25300
由于滤波器300与对比例的并联谐振器压电层厚度并不相同，因此两个滤波器的面积需要单独优化，在滤波器整体通带和阻带性能基本一致时，才能对比高次寄生谐振区域的抑制性能。
图19(b)是0.5GHz～3.5GHz的宽频带响应曲线，图19(a)则是同一曲线在820MHz～850MHz范围内的通带特性详细展示。图19(b)中的301区域，是由谐振器二次寄生谐振产生的抑制尖峰。302区域，是由谐振器三次寄生谐振形成的抑制尖峰。类似前面图17的原理说明，由于第三谐振区域内，串联谐振器的三次寄生谐振频率f s,2大于并联谐振器的f s,2，且二者频率差大于单个谐振器的f s,2和f p,2的间距，因此在第三谐振区会形成303和304两个抑制尖峰，303与并联谐振器的三次寄生谐振相对应，304与串联谐振器的三次寄生谐振相对应。这两个抑制恶化点恰好位于Band 41(2496MHz～2600MHz)范围内，因此有可能对系统中Band 41的通信产生不利影响。
图20(a)和图20(b)是滤波器300的幅频响应曲线，其中图20(a)是0.5GHz～3.5GHz的宽频带响应曲线，图20(b)则是同一曲线在820MHz～850MHz范围内的通带特性详细展示。图20(b)中的401区域，是由谐振器二次寄生谐振产生的抑制尖峰；402区域，是由谐振器三次寄生谐振形成的抑制尖峰。实施例通过增大并联谐振器压电层 厚度占比的方式，使得并联谐振器的三次寄生谐振位置从远低于串联谐振器三次寄生谐振的位置，移动到与其相当甚至更高的位置，若定义X为串联谐振器的三次寄生反谐振频率f p,3与三次寄生谐振频率f s,3之差，即X＝f p,3-f s,3，此时X的值在0附近，即满足此设定值的范围是-X～3X之间。通过以上设置，使得并联谐振器与串联谐振器三次寄生谐振形成类似图9原理的阻抗抵消关系，甚至类似图12原理形成位于三次寄生谐振区域的带阻滤波器，有效的将三次寄生谐振产生的尖峰削弱，仅留存一个403约30dB的抑制尖峰和一个404约25dB的抑制尖峰，相比图19有了明显的改善。403与串联谐振器的三次寄生谐振相对应，404与并联谐振器的三次寄生谐振相对应。
图21把实施例和对比例的曲线放在一起进行对比，其中图21(b)是0.5GHz～3.5GHz的宽频带响应曲线，图21(a)则是同一曲线在820MHz～850MHz范围内的通带特性详细展示。图中粗实线为实施例滤波器幅频响应曲线，细虚线为对比的滤波器幅频响应曲线。从图21(b)可以看出，由于实施例并联谐振器的压电层厚度占比增加，使得与并联谐振器关联的谐振尖峰由303的位置向404位置移动，并且两个谐振尖峰均得到了有效的抑制。
从图21(a)可以看出，在高次寄生谐振抑制得到改善的同时，由于并联谐振器压电层厚度占比增加同时还伴有
Figure PCTCN2021087003-appb-000014
的增加，因此滤波器通带的带宽也有所增加，通带的插入损耗也略有提升。需要说明的是，只有类似此实施例，把并联谐振器的压电层厚度占比增加，才能同时达到高次寄生谐振抑制改善和通带带宽增加并且插损改善的双重效果。如果是按现有技术，把串、并联的谐振器的压电层厚度占比同样增加到相同厚度，此时串并联谐振器的
Figure PCTCN2021087003-appb-000015
也同时变大，虽然也可以使通带变宽并且插损改善，但是由于串、并联谐振器在三次寄生谐振的频率差的关系没有得到本质上的改变，因此在三次寄生谐振处的抑制尖峰只会随层叠改变向高频方向移动，但是抑制尖峰的高度不会得到有效的削弱。
同时，本实施例中，谐振器的二次寄生谐振相对频率范围较窄，强度也不强，总体上对滤波器的抑制影响不大。但是如果在某些设计 中，二次寄生谐振也类似三次寄生谐振，产生了对抑制的不利影响，则可以利用同样的原理进行设计规避。一般来说，如果高次(即二次及以上)寄生谐振的强度足以影响滤波器的性能，那么都可采用本发明实施方式提供的方法加以抑制。
图22是实施例的串、并联谐振器层叠设置的侧面示意图，二者下电极厚度均为1.2um，串联谐振器压电层厚度t1为0.74um，并联谐振器压电层厚度t2为0.935um，同时二者的上电极厚度设置不同。这种方式区别于图23示出的对比例的结构，图23是对比例的串、并联谐振器层叠设置的侧面示意图。从图24所示的串、并联谐振器的宽频阻抗曲线关系能够看出：1)在第一谐振区，串联谐振器的f s与并联谐振器的f p基本对齐，以实现较好的滤波器通带特性；2)在第三谐振区，并联谐振器的f p,2与串联谐振器的f s,2的频率差，位于-X～3X区间之内，其中X是某个串联谐振器的f p,2与其f s,2之差，以实现较好的三次寄生谐振区域抑制特性。特别地，对于本实施例，使得并联谐振器的f p,2与串联谐振器的f s,2的频率差为X，事实上，只要符合前述频率关系，都能达到近似的技术效果。
考虑到上述方式涉及谐振器各层厚度的调节，为了便于加工制造，可以将串联谐振器和并联谐振器分别制作在不同的晶圆上。以下举例加以说明，请参见图25至图27，其中图25是本发明实施方式涉及的一种滤波器500的电路图，图26是滤波器500的侧面示意图，图27是滤波器500的上晶圆和下晶圆的主视图。其中上晶圆501中制作并联谐振器(图中以薄膜体声波谐振器FBAR为例)，下晶圆502中制作串联谐振器。图中，P1-P6为并联谐振器，S1-S6为串联谐振器，VIN为输入管脚，VOUT为输出管脚，VG1和VG2为接地管脚，并且两个晶圆都布置了VIN、VG1、VG2、VOUT这四个键合区。同时添加了新的键合区，即图中的J1，J2，J3，此新键合区只是用来将上晶圆501与下晶圆502连接在一起，而不需要通过过孔向芯片外部连接，因此其形状都与连接过孔的键合区不同。
图28是采用本发明的一个Band 5双工器电路图，其中发送滤波器Tx频率为824MHz～849MHz，接收滤波器Rx的频率范围为 869MHz～894MHz。两颗滤波器均采用本发明，实现了高次寄生谐振抑制的改善。
在上面的描述中，主要说明了谐振器各层厚度的调节对于抑制三次寄生谐振的作用，并且还可用于抑制高次寄生谐振。另外本发明实施方式中，串联谐振器与并联谐振器可以采用不同的材料，以使它们的高次寄生谐振符合上文所说明的频率关系。在调整串联谐振器与并联谐振器材料的情况下，也可以结合各层厚度的调整。所谓不同的材料，可以是电极的材料不同，如钼、钌、金、铝、镁、钨、铜，钛、铱、锇、铬或以上金属的复合或其合金等，也可以是压电层的材料不同，如氮化铝，氧化锌，PZT(锆钛酸铅系压电陶瓷)等材料并包含上述材料的一定原子比的稀土元素掺杂材料。例如，串并联谐振器采用的电极材料都是钼，压电层材料都是掺钪的氮化铝，电极和压电层二者除了厚度不同外，串联谐振掺杂浓度与并联谐振器的掺杂浓度也不同，通常情况下，钪的掺杂浓度越高，其三次寄生谐振的频率也越高，因此，在本发明实施方式中，可设置串联谐振器的钪掺杂浓度是7％左右，并联谐振器的钪掺杂浓度是10％左右，此时并联谐振器的f p,2与串联谐振器的f s,2的频率差X，与某个串联谐振器的f p,2与f s,2的频率差相当，因此其在三次寄生谐振频率附近的抑制得到了改善。
根据本发明实施方式的技术方案，通过调整谐振器中的层的厚度或材料，或二者同时调整，使得并联谐振器的三次寄生谐振位置从远低于串联谐振器三次寄生谐振的位置，移动到与其相当甚至更高的位置，由此抑制或消除三次寄生谐振区的伪通带现象，从而提高滤波器的带外抑制性能。
上述具体实施方式，并不构成对本发明保护范围的限制。本领域技术人员应该明白的是，取决于设计要求和其他因素，可以发生各种各样的修改、组合、子组合和替代。任何在本发明的精神和原则之内所作的修改、等同替换和改进等，均应包含在本发明保护范围之内。

Claims (11)

  1. 一种改善压电滤波器带外抑制的方法，所述压电滤波器为多阶的梯形结构，每阶的梯形结构包含1个串联谐振器和1个并联谐振器，其特征在于，该方法包括：
    调节至少1阶的梯形结构中的串联谐振器和并联谐振器中的一层或多层的厚度，并且/或者，调节所有串联谐振器和/或并联谐振器的材料，从而使该梯形结构中的并联谐振器的高次寄生谐振对应的反谐振频率大于串联谐振器高次寄生谐振对应的谐振频率，或者二者之间的差值小于设定值。
  2. 根据权利要求1所述的方法，其特征在于，所述谐振器中的层的厚度包含该层的质量负载的厚度。
  3. 根据权利要求1所述的方法，其特征在于，所述谐振器中的层包括：上、下电极，压电层，钝化层。
  4. 根据权利要求3所述的方法，其特征在于，所述该梯形结构中，并联谐振器的压电层厚度大于串联谐振器的压电层厚度。
  5. 根据权利要求4所述的方法，其特征在于，对于所述该梯形结构中的并联谐振器和串联谐振器，下电极厚度相同或不同，上电极厚度不同。
  6. 根据权利要求1所述的方法，其特征在于，所述设定值的范围是-X～3X之间，其中X为所述1阶的梯形结构中的串联谐振器的高次寄生反谐振频率f p,n与相应的高次寄生谐振频率f s,n之差，即X＝f p,n-f s,n。
  7. 根据权利要求1所述的方法，其特征在于，
    调节所有串联谐振器和/或并联谐振器的材料的步骤包括：使所有 串联谐振器的电极的材料与所有并联谐振器的电极的材料不同；
    或者，
    调节所有串联谐振器和/或并联谐振器的材料的步骤包括：使所有串联谐振器的压电层的材料与所有并联谐振器的压电层的材料不同。
  8. 根据权利要求1至7中任一项所述的方法，其特征在于，所述滤波器中，串联谐振器和并联谐振器分别设置在上下层叠的2个晶圆中。
  9. 一种压电滤波器，其特征在于，该压电滤波器是使用权利要求1至8中任一项所述的方法制造。
  10. 一种多工器，其特征在于，包含权利要求9所述的压电滤波器。
  11. 一种通信设备，其特征在于，包含权利要求9所述的压电滤波器。
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