WO2020093465A1 - 一种电力机车用大功率直驱永磁电传动系统 - Google Patents

一种电力机车用大功率直驱永磁电传动系统 Download PDF

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WO2020093465A1
WO2020093465A1 PCT/CN2018/117085 CN2018117085W WO2020093465A1 WO 2020093465 A1 WO2020093465 A1 WO 2020093465A1 CN 2018117085 W CN2018117085 W CN 2018117085W WO 2020093465 A1 WO2020093465 A1 WO 2020093465A1
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current
value
permanent magnet
voltage
rotor
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PCT/CN2018/117085
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English (en)
French (fr)
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王彬
詹哲军
张瑞峰
于森林
张宇龙
梁海刚
葸代其
柴璐军
王龙刚
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中车永济电机有限公司
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/04—Arrangements for controlling or regulating the speed or torque of more than one motor
    • B—PERFORMING OPERATIONS; TRANSPORTING
    • B60—VEHICLES IN GENERAL
    • B60L—PROPULSION OF ELECTRICALLY-PROPELLED VEHICLES; SUPPLYING ELECTRIC POWER FOR AUXILIARY EQUIPMENT OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRODYNAMIC BRAKE SYSTEMS FOR VEHICLES IN GENERAL; MAGNETIC SUSPENSION OR LEVITATION FOR VEHICLES; MONITORING OPERATING VARIABLES OF ELECTRICALLY-PROPELLED VEHICLES; ELECTRIC SAFETY DEVICES FOR ELECTRICALLY-PROPELLED VEHICLES
    • B60L15/00—Methods, circuits, or devices for controlling the traction-motor speed of electrically-propelled vehicles
    • B60L15/32—Control or regulation of multiple-unit electrically-propelled vehicles
    • B60L15/38—Control or regulation of multiple-unit electrically-propelled vehicles with automatic control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
    • H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
    • H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/40—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc
    • H02M5/42—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters
    • H02M5/44—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac
    • H02M5/453—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/458—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M5/4585—Conversion of ac power input into ac power output, e.g. for change of voltage, for change of frequency, for change of number of phases with intermediate conversion into dc by static converters using discharge tubes or semiconductor devices to convert the intermediate dc into ac using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only having a rectifier with controlled elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0003—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control
    • H02P21/0021—Control strategies in general, e.g. linear type, e.g. P, PI, PID, using robust control using different modes of control depending on a parameter, e.g. the speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0012—Control circuits using digital or numerical techniques
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/05—Synchronous machines, e.g. with permanent magnets or DC excitation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/72—Electric energy management in electromobility

Definitions

  • the invention relates to the technical field of train control, in particular to a high-power direct-drive permanent magnet electric drive system for electric locomotives.
  • the traction converter of the electric locomotive is an important part of the electric locomotive. It is used to convert the electric energy of the traction power supply network into the electric energy supplied to the traction motor to achieve the purpose of controlling the speed of the traction motor and manipulating the speed of the locomotive.
  • the design of the main circuit of the traction converter is one of the main factors of the working performance of the traction converter, which directly affects the weight, size, efficiency and related technical and economic indicators of the electric locomotive.
  • the electric locomotive in the prior art generally adopts the driving mode of AC asynchronous motor plus gear box.
  • the present invention uses a direct drive permanent magnet synchronous motor to be applied to the electric locomotive.
  • the direct-drive permanent magnet synchronous motor takes full advantage of the advantages of high efficiency, low loss, high power density and large starting torque of the permanent magnet synchronous motor.
  • the gear box is removed, and the permanent magnet synchronous motor and the machine are directly driven. The combination of the wheel pairs reduces the mass and the loss caused by the gear box, and further improves the overall efficiency of the electric locomotive.
  • the current traction converters in electric locomotives and the existing electric drive systems are not designed for direct drive permanent magnet synchronous motors, so there is no electric drive system that can be directly applied to the power using direct drive permanent magnet synchronous motors In the locomotive.
  • How to design a high-power direct-drive permanent-magnet electric transmission system for electric locomotives in electric locomotives using direct-drive permanent magnet synchronous motors is a technical problem that needs to be solved urgently.
  • the invention provides a high-power direct-drive permanent-magnet electric transmission system for electric locomotives, which can control the direct-drive permanent-magnet synchronous motors in electric locomotives using direct-drive permanent-magnet synchronous motors, thereby enriching high-power direct-drive locomotives for electric locomotives
  • the function of permanent magnet electric transmission system fills the gap of direct drive permanent magnet synchronous motor in electric locomotive.
  • the invention provides a high-power direct-drive permanent-magnet electric drive system for electric locomotives, which is used to control an electric locomotive using a direct-drive permanent-magnet synchronous motor.
  • the electric locomotive includes three direct-drive permanent-magnet synchronous motors;
  • the electric locomotive High-power direct-drive permanent magnet electric drive system includes: first four-quadrant rectifier, second four-quadrant rectifier, intermediate DC loop, first inverter module, second inverter module, third inverter module, and auxiliary converter ,
  • the first four-quadrant rectifier and the second four-quadrant rectifier are both connected to the main transformer of the electric locomotive and the intermediate DC circuit, and the intermediate DC circuit is respectively connected to the first inverter module and the second inverter Transformer module, third inverter module and the auxiliary converter;
  • the high-power direct-drive permanent magnet electric drive system for the electric locomotive is used for:
  • the auxiliary DC converter converts the received DC power into three-phase AC power and outputs it to the auxiliary load of the electric locomotive.
  • the first four-quadrant rectifier and the second four-quadrant rectifier convert the alternating current of the main transformer into direct current and output to the intermediate direct current loop, including :
  • the AC current includes a positive half-cycle current value and a negative half-cycle current value; wherein, according to the preset sampling frequency, the input
  • the AC current of the four-quadrant rectifier is sampled to obtain multiple sampling points, and the obtained multiple sampling points are plotted as a curve to obtain a sine or cosine curve;
  • the preset sampling frequency is N times the IGBT on-off frequency. N ⁇ 2;
  • the first PI controller Input the first difference between the current offset value and zero to the first PI controller to obtain the first output value output by the first PI controller; wherein, the DC offset value Q and zero are input to the first
  • the first PI controller constitutes a control deviation according to the DC offset value Q and zero, and linearly combines the proportionality and integral of the deviation to form a control amount, controls the AC current, and eliminates the DC offset of the AC current.
  • the control quantity is the first output value;
  • a pulse width modulation symbol is obtained according to the first output value and the second output value output by the PR controller, and the PR controller is used to perform static-free control of the alternating current so that the period and phase of the alternating current are The grid voltage is the same; where the AC current is input to the PR controller to ensure that the phase and cycle of the AC current are the same as the grid voltage, a stable output AC current is obtained, which is the second output value;
  • the on-off of the insulated gate bipolar transistor IGBT in the four-quadrant rectifier is controlled according to the pulse width modulation sign.
  • the method before sampling the alternating current of the input four-quadrant rectifier to obtain the alternating current in the sampling period, the method further includes:
  • the phase-locked loop is used to control the period and phase of the alternating current and the period and phase of the grid voltage to be consistent.
  • the AC current input to the four-quadrant rectifier is sampled to obtain the AC current within the sampling period, including:
  • the alternating current in the sampling period is obtained.
  • the method before obtaining the AC current in the sampling period according to the grid voltage phase determined by the phase-locked loop and the sampling current, the method further includes:
  • the sampling current through a first band-pass filter and a second band-pass filter to obtain a filtered sampling current; wherein, the first band-pass filter is used to obtain a main frequency signal of an alternating current, the The second band-pass filter is used to filter out interference harmonics.
  • the method before inputting the first difference between the current bias value and zero to the first PI controller and obtaining the first output value output by the first PI controller, the method further include:
  • obtaining the pulse width modulation symbol according to the first output value and the second output value output by the PR controller includes:
  • the pulse width modulation symbol is obtained according to the third sum value and the unipolar frequency doubling pulse modulation method.
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives further includes: a first chopping module and a second chopping module, the first chopping module is connected to the first four A quadrant rectifier and the intermediate DC circuit, and the second chopper module connects the second four-quadrant rectifier and the intermediate DC circuit;
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives is also used for:
  • DC power output from the first four-quadrant rectifier and the second four-quadrant rectifier is chopped by the first chopping module and the second chopping module, and then output to the intermediate DC loop;
  • control method further includes:
  • the intermediate DC bus voltage being the voltage on the DC bus on the electric locomotive
  • the P regulator When the detected intermediate DC bus voltage value is greater than the upper chopping threshold, the P regulator is used to adjust the intermediate DC bus voltage until the detected intermediate DC bus voltage value is less than the lower chopping threshold, the chopping The upper limit of the wave threshold is greater than the lower limit of the chopping threshold; wherein, the principle of the P regulator is to control the chopper tube to be turned on within a certain proportion of the detection cycle.
  • the adjusting the intermediate DC bus voltage with a P regulator includes:
  • the target detection period includes: from the detected intermediate DC bus voltage value greater than the upper chopping threshold, to the detected intermediate DC bus voltage value Less than the chopping lower threshold between the detected detection period;
  • the chopping duty ratio determine the opening time of the chopper tube within the target detection period
  • the turn-on or turn-off of the chopper tube is controlled so that the voltage value of the intermediate DC bus drops to less than the lower threshold of the chopper.
  • the above method further includes:
  • the chopper tube When it is detected that the voltage value of the intermediate DC bus is lower than the lower chopping threshold, the chopper tube is controlled to be turned off.
  • the method before using the P regulator to determine the chopping duty cycle within the target detection period, the method further includes:
  • Err represents the target parameter
  • U1 represents the intermediate DC bus voltage value detected in the target detection period
  • the use of the P regulator to determine the chopping duty cycle within the target detection period includes:
  • the chopping duty ratio is determined.
  • the acquiring the control coefficient of the P regulator includes:
  • Kp_chp 1 / (DC bus voltage overvoltage protection threshold-chopping lower threshold)
  • Kp_chp represents the control coefficient
  • the determining the chopper duty cycle according to the control coefficient and the target parameter includes:
  • C_duty represents the chopping duty ratio
  • Err represents the target parameter
  • Kp_chp represents the control coefficient
  • the method before determining the opening time of the chopper tube within the target detection period according to the chopping duty cycle, the method further includes:
  • the error prevention processing of the chopping duty cycle includes:
  • the value of the chopping duty ratio is set to 1;
  • the value of the chopping duty ratio is set to 0.
  • control method further includes:
  • the expected control phase angle of the direct drive permanent magnet synchronous motor to be controlled is determined according to the first control strategy.
  • the first mapping relationship includes:
  • the MTPA control strategy includes: determining the q-axis current reference and the d-axis current reference according to the torque current curve;
  • the d-axis voltage reference is obtained according to the first difference value through the first PI controller, and the q-axis voltage reference is obtained based on the second difference value through the second PI controller;
  • the feedforward voltage can be calculated by the following feedforward decoupled closed-loop transfer function matrix:
  • the closed-loop transfer function of the feedforward decoupling is obtained by the following voltage calculation equation of feedforward decoupling:
  • the field weakening control strategy includes: calculating, by the PI controller, the amount of d-axis current change in a given field weakening state according to the difference between the voltage limit value and the feedforward voltage amplitude;
  • the d-axis current reference after the field-weakening adjustment is obtained by giving the sum of the d-axis current change and the d-axis current under the given field weakening state;
  • the PI controller obtains the work angle ⁇ according to the difference between the q-axis current setting and the q-axis actual current;
  • Us is the voltage limit value
  • Ud is the actual d-axis voltage given
  • Uq is the actual q-axis voltage given.
  • the method further includes:
  • the voltage vector angle in the MTPA control strategy at the moment of switching is used as the initial power angle ⁇ in the field weakening control strategy;
  • the last beat power angle ⁇ in the instantaneous field weakening control strategy is passed through the formula by switching Calculate the actual q-axis voltage setting and actual d-axis voltage setting in the MTPA control strategy.
  • control method further includes:
  • the PWM carrier frequency of the direct drive permanent magnet synchronous motor is determined according to the first modulation strategy.
  • the second mapping relationship includes:
  • the frequency of the modulation wave When the frequency of the modulation wave is greater than the low-speed stage and lower than the high-speed stage, it corresponds to the middle 60-degree synchronous modulation strategy;
  • the frequency of the modulated wave corresponds to the square wave modulation strategy at the high-speed stage.
  • the method further includes:
  • the initial position angle of the rotor is obtained according to the d-axis target current and the q-axis target current, where the initial position angle is the initial position angle after compensation according to the magnetic pole polarity of the permanent magnet synchronous motor.
  • the acquiring the initial position angle of the rotor according to the d-axis target current and the q-axis target current includes:
  • the obtaining the first initial position angle of the rotor according to the q-axis target current includes:
  • the performing low-pass filtering on the q-axis target current to obtain the error input signal includes:
  • the acquiring the first initial position angle according to the error input signal includes:
  • the obtaining the magnetic pole compensation angle of the rotor according to the d-axis target current includes:
  • the pole compensation angle of the rotor is determined.
  • the determining the pole compensation angle of the rotor according to the plurality of response currents includes:
  • the rotor pole compensation is determined The angle is 0, and the first value is the maximum value of the amplitudes of the multiple response currents;
  • the rotor pole compensation is determined
  • the angle is ⁇
  • the second value is the minimum value of the amplitudes of the multiple response currents.
  • the high-frequency voltage signal is:
  • U mh is the amplitude of the high-frequency voltage signal
  • ⁇ h is the angular frequency of the high-frequency voltage signal
  • t is the time to inject the high-frequency voltage signal
  • the d-axis target current and the q-axis target current in the expected two-phase synchronous rotating coordinate system are obtained according to the three-phase stator winding current, and are calculated by the following formula:
  • the low-pass filtering process is performed on the q-axis target current to obtain an error input signal, which is calculated by the following formula:
  • LPF low-pass filtering
  • the first initial position angle is obtained and calculated by the following formula:
  • s represents Laplace operator
  • k p is the coefficient of proportional term
  • k i is the coefficient of integral term
  • the high-power direct-drive permanent-magnet electric drive system for electric locomotives provided in this embodiment further includes: according to a control interruption period, a modulated carrier period, and the current angular velocity of the rotor of the direct-drive permanent-magnet synchronous motor, obtaining the Compensation phase angle of rotor of direct drive permanent magnet synchronous motor;
  • the current actual control phase angle is corrected online.
  • the obtaining the compensation phase angle of the rotor of the direct-drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor includes:
  • the compensation phase angle of the direct-drive permanent magnet synchronous motor is obtained according to the first sub-compensation phase angle, the second sub-compensation phase angle, and the third sub-compensation phase angle.
  • the acquiring the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor includes:
  • the first sub-compensated phase angle is obtained according to the first phase angle delay and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor.
  • the obtaining the second sub-compensated phase angle according to the modulated carrier period and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor includes:
  • the second sub-compensated phase angle is obtained according to the second phase angle delay, the third phase angle delay, and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor.
  • the method before acquiring the third sub-compensated phase angle according to the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor, the method further includes:
  • a plurality of first d-axis currents, a plurality of first q-axis currents, and each of the first d-axis currents corresponding to the stable operating angular velocity range are acquired D-axis voltage and the q-axis voltage corresponding to each of the first q-axis currents.
  • the acquiring the third sub-compensated phase angle according to the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor includes:
  • the third sub-compensated phase angle is obtained according to the transmission error phase angle corresponding to each of the first angular speeds, the current angular speed of the rotor of the direct-drive permanent magnet synchronous motor, and the initial position phase angle of the rotor.
  • the obtaining the current actual control phase angle according to the compensated phase angle includes:
  • the current actual control phase angle is obtained according to the actual position phase angle and the modulation phase angle of the rotor, wherein the modulation phase angle is calculated by a modulation algorithm according to the given value of the d-axis voltage and the given value of the current q-axis voltage.
  • the online correction of the current actual control phase angle according to the proportional deviation and integral deviation of the current expected control phase angle and the current actual control phase angle includes:
  • the first sub-compensation phase angle is calculated by the following formula:
  • [omega] is the angular velocity of the current of the direct-drive permanent magnet synchronous motor rotor, a first phase angle ⁇ t1 to time delay, the first delay phase angle ⁇ t1 is calculated by the following equation:
  • ⁇ t1 A ⁇ T ctrl ⁇ 0.5T ctrl .
  • T ctrl is a control interruption cycle of the control algorithm
  • the second sub-compensation phase angle is calculated by the following formula:
  • ⁇ is the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor
  • ⁇ t2 is the time delay in the PWM pulse output process
  • the time delay ⁇ t2 in the PWM pulse output process is calculated by the following formula:
  • ⁇ t2 B ⁇ T PWM + C ⁇ T PWM ⁇ 0.75T PWM
  • T PWM is the PWM modulation carrier period
  • B is the modulation algorithm interrupt delay coefficient
  • C is the PWM pulse output delay coefficient
  • the current expected control phase angle is calculated by the following formula:
  • ⁇ ctrl represents the expected control phase angle
  • k p and k i are correction terms
  • ⁇ ctrl is the current expected phase angle
  • ⁇ PWM is the current actual phase angle
  • f ⁇ is the fundamental frequency compensation term
  • u d is the d-axis voltage corresponding to any first preset angular velocity
  • u q is the q-axis voltage corresponding to any first preset angular velocity
  • R s is the resistance of the rotor
  • L q is any first preset D-axis inductance corresponding to angular velocity
  • L d is the q-axis inductance corresponding to any first preset angular velocity
  • i d is the first d-axis current corresponding to the d-axis voltage
  • i q is the first q-axis current corresponding to the q-axis voltage
  • ⁇ f is the back-EMF of the permanent magnet flux linkage
  • phase angle ⁇ ⁇ of the transmission error is calculated by the following formula:
  • the electric locomotive further includes: at least four direct drive permanent magnet synchronous motors; the at least four direct drive permanent magnet synchronous motors include: a first motor, a second motor, a third motor and Fourth motor
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives is also used for:
  • the torque of the first motor is adjusted according to the torque reduction amount.
  • the method further includes:
  • a sanding control signal is generated, and the sanding control signal is used to indicate whether to perform sanding operation.
  • the torque reduction amount is determined according to the rotor frequency difference, the rotor frequency differential value, and the real-time torque of the first motor, including:
  • the rotor frequency difference of the first motor determines the idling coasting level corresponding to the rotor frequency difference of the first motor
  • the first torque reduction amount is determined according to the idling coasting level corresponding to the rotor frequency difference of the first motor and the real-time torque of the first motor;
  • the first motor rotor frequency differential value and the preset rotor frequency differential value classification rules determine the idling coasting level corresponding to the first motor rotor frequency differential value
  • the second torque reduction amount is determined according to the idling coasting level corresponding to the differential value of the rotor frequency of the first motor and the real-time torque of the first motor;
  • the first torque reduction amount is determined to be the torque reduction amount
  • the second torque reduction amount is determined as the torque reduction amount.
  • adjusting the torque of the first motor according to the torque reduction includes:
  • the recovery rate of the torque value of the first motor in the third preset time period is greater than the recovery rate of the torque value of the first motor in the fourth preset time period.
  • reducing the torque value of the first motor from the first value to the second value within the first preset time period includes:
  • the torque value of the first motor is gradually reduced according to the rate of decrease of the torque value of the first motor, and the torque value of the first motor is reduced from the first value to the second value.
  • determining the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors includes:
  • the rotor frequency difference and the rotor frequency differential value of the first motor are determined according to the rotor frequencies of the multiple motors after the limiting filtering and low-pass filtering.
  • the amplitude filtering and low-pass filtering processing of the collected multiple rotor frequencies includes:
  • Limiting filtering and low-pass filtering are performed on the compensated rotor frequencies of multiple motors.
  • a high-power direct-drive permanent magnet electric drive system for electric locomotives also includes an auxiliary converter control device, including: a digital signal processing DSP chip and a field programmable gate array FPGA chip, the DSP The chip is connected to the FPGA chip bus, wherein the high-power direct-drive permanent magnet electric drive system for electric locomotives is also used for:
  • the FPGA chip is used to obtain the analog quantity and digital quantity of the auxiliary converter through the analog quantity sampling board and the digital quantity sampling board, and perform logic operation processing on the analog quantity and digital quantity to obtain the logic operation processing The data;
  • the DSP chip is used to perform control operation processing on the data after the logic operation processing to obtain a pulse width
  • the FPGA chip is also used to perform modulation calculation processing according to the pulse width to obtain a driving pulse sequence of the auxiliary converter.
  • control operation processing includes: Park transformation processing, voltage and current double-loop decoupling control processing, IPark transformation processing, and zero sequence voltage injection processing;
  • the modulation operation processing includes: pulse generation processing;
  • the DSP chip is specifically used for sequentially performing Park transformation processing on the data after the logical operation processing, voltage and current double-loop decoupling control processing, IPark transformation processing and zero sequence voltage injection processing to obtain the zero sequence voltage injection processing data;
  • the FPGA chip is specifically used to perform pulse generation processing on the data after the zero-sequence voltage injection processing to obtain the data after the pulse generation processing.
  • the FPGA chip is also used to determine whether a component in the auxiliary converter fails according to the collected operating data of the auxiliary converter;
  • the FPGA chip determines that a component in the auxiliary converter is faulty, send the fault information to the DSP chip;
  • the FPGA chip determines that the components in the auxiliary converter have not failed, then send the operation data to the DSP chip;
  • the DSP chip is also used to determine that the component is faulty according to the fault information
  • the DSP chip is also used to determine whether the components in the auxiliary converter have failed according to the operation data.
  • the FPGA chip is further specifically configured to determine that the component is faulty when it is determined that the operating data is greater than the first preset threshold according to the collected operating data of the auxiliary converter.
  • the DSP chip is further specifically configured to determine that the component has failed when it is determined that the operating data is greater than a second preset threshold and the duration of the failure is greater than a preset duration, wherein, the The first preset threshold is greater than the second preset threshold.
  • the DSP chip is also used to determine the number of failures of the component after the auxiliary converter is restarted multiple times;
  • the DSP chip is also used to determine whether the number of faults is greater than a preset number of times. If yes, the fault is determined to be a permanent fault, and if not, the fault is determined to be a warning fault.
  • the DSP chip is also used to set the fault flag bit corresponding to the component to the permanent fault flag bit when determining that the fault of the component is a permanent fault.
  • it further includes: a host computer, the host computer is connected to the DSP chip;
  • the DSP chip is also used for splicing multiple fault flags and sending the spliced fault flags to the host computer.
  • the host computer communicates with the DSP chip through a CAN bus.
  • it further includes: a flash memory; a RAM space is provided in the DSP chip; the flash memory is connected to the DSP chip;
  • the DSP chip is also used to store the data processed by the logic operation into the RAM space;
  • the DSP chip is also used to determine that the component has failed, and store the data in the RAM space into the flash memory.
  • the alternating current of the main transformer is passed through the "AC-DC-AC"
  • the process is finally converted to three-phase AC power available for direct-drive permanent magnet synchronous motors. Therefore, the direct drive permanent magnet synchronous motor in the electric locomotive using the high power direct drive permanent magnet synchronous motor can be controlled, which enriches the functions of the high power direct drive permanent magnet electric drive system for the electric locomotive and fills the direct drive permanent magnet synchronous
  • the application of electric motors in electric locomotives is blank.
  • FIG. 1 is a schematic structural view of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention
  • FIG. 2 is a schematic flow chart of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention
  • FIG. 3 is a schematic structural view of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention
  • FIG. 4 is a partial circuit diagram of a four-quadrant rectifier provided by an embodiment of the present invention.
  • FIG. 5 is a schematic flow chart of a method for adjusting current bias of a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention
  • FIG. 6 is a schematic flowchart of a current bias adjustment method for a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention
  • FIG. 7 is a schematic flow chart of a method for adjusting current offset of a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention provided by this embodiment
  • Embodiment 8 is a schematic flowchart of Embodiment 1 of a chopping control method provided by the present invention.
  • FIG. 9 is a schematic structural view of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention.
  • Embodiment 10 is a schematic flowchart of Embodiment 2 of the chopper control method provided by the present invention.
  • FIG. 11 is another schematic flowchart of Embodiment 2 of the chopper control method provided by the present invention.
  • FIG. 12 is a schematic flowchart of Embodiment 3 of a chopper control method provided by the present invention.
  • FIG. 13 is a schematic flowchart of a control method for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention
  • FIG. 14 is a schematic structural view of a control system for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention
  • 15 is a schematic diagram of the system structure of the MTPA control system of the present invention.
  • 16 is a schematic diagram of the system structure of the front-end decoupling control of the present invention.
  • 17 is a schematic diagram of the system structure of the field weakening control of the present invention.
  • 19 is a schematic diagram of switching control of MTPA control and field weakening control of the present invention.
  • 20 is a schematic flowchart of a modulation method for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention
  • 21 is the relationship between the modulation angle and the modulation ratio under the intermediate 60 ° modulation method provided by the present invention.
  • 22 is a schematic diagram of a full speed range modulation strategy based on intermediate 60 ° modulation provided by the present invention
  • Embodiment 23 is a schematic flowchart of Embodiment 1 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention
  • 24 is a schematic diagram of the relationship between the two-phase synchronous rotating coordinate system, the two-phase stationary coordinate system and the expected two-phase synchronous rotating coordinate system provided by the present invention
  • Embodiment 25 is a schematic flowchart of Embodiment 2 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention
  • FIG. 26 is a schematic flowchart of Embodiment 3 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention
  • Figure 27 is a schematic diagram of signal changes of multiple channels during the operation of a permanent magnet synchronous motor
  • Figure 28 is a schematic diagram of the response current change law
  • 29 is a schematic structural diagram of a control system of a direct-drive permanent magnet synchronous motor corresponding to a high-power direct-drive permanent-magnet electric transmission system for an electric locomotive provided by the present invention
  • FIG. 30 is a first schematic flowchart of a control method of a direct drive permanent magnet synchronous motor provided by the present invention.
  • FIG. 31 is a second schematic flowchart of a control method of a direct drive permanent magnet synchronous motor provided by the present invention.
  • 32 is a schematic diagram of an interruption cycle of a control algorithm provided by the present invention.
  • Figure 34 is a schematic diagram of a multi-mode PWM modulation strategy
  • 35 is a third schematic flowchart of a control method of a direct drive permanent magnet synchronous motor provided by the present invention.
  • Figure 36A is a schematic diagram of the theoretical coordinate system and the actual coordinate system completely coincide;
  • Fig. 36B is a schematic diagram of the actual coordinate system leading the theoretical coordinate system
  • Figure 36C is a schematic diagram of the actual coordinate system lagging behind the theoretical coordinate system
  • 39 is a first schematic structural diagram of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention.
  • 40 is a schematic diagram of processing of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention
  • 41 is a second schematic structural diagram of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention.
  • FIG. 42 is a third schematic structural diagram of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention.
  • FIG. 1 is a schematic structural view of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention.
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives provided in this embodiment includes: a first four-quadrant rectifier, a second four-quadrant rectifier, an intermediate DC loop, a first inverter module, and a second inverter
  • the transformer module, the third inverter module and the auxiliary converter, the first four-quadrant rectifier and the second four-quadrant rectifier are connected to the main transformer of the electric locomotive and the intermediate DC circuit, and the intermediate DC circuit is respectively connected to the first inverter module and the second Inverter module, third inverter module and auxiliary converter.
  • the high-power direct-drive permanent-magnet electric transmission system for electric locomotives provided in this embodiment is used for an electric machine using a direct-drive permanent-magnet synchronous motor, and is used to control at least one direct-drive permanent-magnet synchronous motor on the electric locomotive.
  • the number of direct-drive permanent magnet synchronous motors is three as an example, and the main circuit provided in this embodiment can also be used to control direct-drive permanent magnet synchronous motors with fewer or more than three Of electric locomotives have the same principle and only increase or decrease in quantity.
  • FIG. 2 is a schematic flow chart of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention.
  • the method of using the high-power direct-drive permanent magnet electric drive system for electric locomotives as shown in FIG. 1 is described below with reference to FIG. 2, wherein the method of using the high-power direct-drive permanent magnet electric drive system for electric locomotives includes:
  • S101 Convert the alternating current of the main transformer into direct current through the first four-quadrant rectifier and the second four-quadrant rectifier, and then output to the intermediate DC circuit.
  • the method of this embodiment is used to control the high-power direct-drive permanent magnet electric drive system for electric locomotives as shown in FIG. 1 to convert the AC power of the converter into a three-phase variable frequency variable frequency converter that can be used by the direct-drive permanent magnet synchronous motor. Press alternating current.
  • the first four-quadrant rectifier and the second four-quadrant rectifier connected to the main transformer can be controlled to convert the AC power of the main transformer into DC power and input it into the intermediate DC circuit.
  • the input terminals of the first four-quadrant rectifier and the second four-quadrant rectifier can be regarded as the input terminals of the entire main circuit, and the input terminals of the first four-quadrant rectifier and the second four-quadrant rectifier can pass through the secondary side of the main transformer The traction winding is connected to obtain the alternating current provided by the main transformer.
  • the number of four-quadrant rectifiers is not specifically limited. For each four-quadrant rectifier arranged in parallel, each four-quadrant rectifier works independently and is used to receive the main transformer The supplied AC power is converted into DC power and output to the intermediate DC loop.
  • S102 Output the received DC power to the first inverter module, the second inverter module, the third inverter module, and the auxiliary converter through the intermediate DC loop, respectively.
  • the DC loop After the intermediate DC loop receives the DC power sent by the first four-quadrant rectifier and the second four-quadrant rectifier, the DC loop is controlled in S102 to direct the DC power to the first inverter module, the second inverter module, and the third Inverter module and auxiliary converter output.
  • the first four-quadrant rectifier and the second four-quadrant rectifier share the intermediate DC circuit, and the intermediate DC circuit transmits the received multiple DC power to the first inverter module, the second inverter module, and the third inverse Transformer module and auxiliary converter output.
  • S103 Convert the received DC power into three-phase AC power through the first inverter module, the second inverter module, and the third inverter module, and then output to three direct-drive permanent magnet synchronous motors, respectively.
  • the inverter module corresponds to the direct drive permanent magnet synchronous motor
  • the auxiliary converter corresponds to the auxiliary load.
  • the electric locomotive includes three direct-drive permanent magnet synchronous motors, so its main circuit also needs to be provided with three inverter modules.
  • the first inverter module is connected to the direct-drive permanent magnet synchronous motor 1 and converts the received DC power into the AC power available to the direct-drive permanent magnet synchronous motor 1 and outputs it to the second inverter module.
  • Direct drive permanent magnet synchronous motor 2 and convert the received DC power into direct drive permanent magnet synchronous motor 2 available AC power and output to it
  • the third inverter module is connected to direct drive permanent magnet synchronous motor 3, and the received The direct current is converted into direct current permanent magnet synchronous motor 3 available alternating current and output to it.
  • Each inverter module drives the direct-drive permanent-magnet synchronous motor through the alternating current sent to the direct-drive permanent-magnet synchronous motor connected to it, thereby realizing the drive control of the three direct-drive permanent-magnet synchronous motors in the electric locomotive.
  • S104 Convert the received DC power into three-phase AC power through an auxiliary converter and output it to the auxiliary load of the electric locomotive.
  • the auxiliary converter can also be connected to the intermediate DC circuit, and in S104, the auxiliary converter can be controlled
  • the DC power is converted into AC power available for the auxiliary load in the electric locomotive, and then output to the auxiliary load.
  • the auxiliary load described herein includes at least one or more of the following: a lighting system, a communication system, and an air conditioning system of an electric locomotive.
  • the alternating current of the main transformer is passed through the "AC-DC-AC"
  • the process is finally converted to three-phase AC power available for direct-drive permanent magnet synchronous motors. Therefore, the direct drive permanent magnet synchronous motor in the electric locomotive using the direct drive permanent magnet synchronous motor can be controlled, which enriches the functions of the high-power direct drive permanent magnet electric drive system for the electric locomotive and fills the direct drive permanent magnet synchronous motor.
  • FIG. 3 is a schematic structural diagram of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention.
  • the embodiment shown in FIG. 3 provides a specific circuit design and connection method of a high-power direct-drive permanent magnet electric drive system for electric locomotives on the basis of FIG. 1 to illustrate subsequent embodiments of the present invention High-power direct-drive permanent magnet electric drive system for electric locomotives in China.
  • a control method for a four-quadrant rectifier in S101 is provided to eliminate the influence of current bias during the control process of the four-quadrant rectifier.
  • FIG. 4 is a partial circuit diagram of a four-quadrant rectifier provided by an embodiment of the present invention.
  • the four-quadrant rectifier shown in FIG. 4 may be the first four-quadrant rectifier shown in FIGS. 1 and 3, or may be as shown in FIG. 1.
  • the working mode and principle of each four-quadrant rectifier provided in this embodiment are the same, and a four-quadrant rectifier will be specifically described below.
  • g1, g2, g3, and g4 are IGBT devices of four-quadrant rectifier, and g1, g2, g3, and g4 work together to realize the function of four-quadrant rectifier to convert AC voltage into DC voltage.
  • a method for adjusting the current offset of the high-power direct-drive permanent magnet electric drive system for electric locomotives is provided in S101.
  • the problem of DC bias can be solved without changing the hardware structure of Fig. 1 and Fig. 3. Detailed description will be given below with reference to FIG. 5.
  • FIG. 5 is a schematic flowchart of a current bias adjustment method for a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention. As shown in FIG. 5, the method includes:
  • the AC current input to the four-quadrant rectifier is sampled to obtain multiple sampling points, and the obtained multiple sampling points are drawn into a curve to obtain a sine or cosine curve.
  • the preset sampling frequency may be twice or even several times of the IGBT on-off frequency or other, as long as the complete sine or cosine curve can be sampled according to the preset sampling frequency, and the preset sampling frequency is not particularly limited here.
  • the preset sampling frequency may be twice the on-off frequency of the IGBT, and then a sine or cosine curve drawn from multiple sampling points obtained according to the preset sampling frequency is divided into positive half cycles according to the phase
  • the negative half cycle for example, the positive half cycle of the sine curve is 0 to ⁇ , and the negative half cycle is ⁇ to 2 ⁇ , then the values of the multiple sampling points of the positive half cycle are the value of the positive half cycle of the AC current, and the number of negative half cycles The value of each sampling point is the value of the negative half cycle of the AC current.
  • the values of the multiple sampling points in the positive half cycle are added to obtain the first sum P, and then the values of the multiple sampling points in the negative half cycle are added to obtain the second sum N, P and N
  • the absolute value of the value is calculated as the difference, and the resulting difference is Q. If the Q value is 0, the absolute values of the P value and the N value are also completely equal, the positive half cycle and the negative half cycle of the sine curve or cosine curve are completely symmetrical, and the AC current has no DC offset. If the Q value is not 0, the absolute value of the P value and the N value are not equal, then the positive half cycle and negative half cycle of the sine curve or cosine curve are asymmetric, the AC current has a DC offset, and the Q value is the DC offset Set value.
  • the DC offset value Q and zero are input to the first PI controller.
  • the first PI controller forms a control deviation according to the DC offset value Q and zero, and linearly combines the proportion and integral of the deviation to form a control amount.
  • the current is controlled to eliminate the DC bias of the AC current.
  • the controlled variable is the first output value.
  • a stable output AC current is obtained, which is the second output value.
  • the first output value and the second output value are summed to obtain a third sum value. That is, the control quantity obtained by the first PI controller regulates the stable output AC current, thereby suppressing the DC bias of the AC current.
  • the third sum value is modulated by a monopole frequency doubling pulse modulation method to obtain a pulse width modulation symbol.
  • the pulse width modulation symbol is used as an input of the insulated gate bipolar transistors IGBTs g1, g2, g3, and g4 in the four-quadrant rectifier to control the turning on and off of the bipolar transistor IGBT.
  • a method for adjusting the current bias of a high-power direct-drive permanent magnet electric drive system for electric locomotives is provided for high-power direct-drive permanent magnet electric drive systems for electric locomotives.
  • the current is sampled to obtain the AC current in the sampling period, which includes the current value of the positive half cycle and the current value of the negative half cycle; the first sum of the current value of the positive half cycle and the current value of the negative half cycle are obtained The second sum value, and obtain the current offset value according to the first sum value and the second sum value; input the first difference between the current offset value and zero to the first PI controller to obtain the first PI controller output
  • the first output value of the; the pulse width modulation symbol is obtained according to the first output value and the second output value output by the PR controller, the PR controller is used to control the AC current without static error, so that the period and phase of the AC current and the grid voltage The same; according to the pulse width modulation symbol to control the turning on and off of the insulated gate bipolar transistor IGBT in the four-quadrant rectif
  • the second output value is adjusted by the first output value output by the first PI controller to obtain a third sum value, thereby suppressing the DC bias of the alternating current, and modulating the third sum value by a unipolar frequency-doubled pulse modulation method
  • the pulse width modulation symbol is used to control the operation of the IGBT, which prevents the IGBT device from deviating from its rated operating area, thereby effectively suppressing and eliminating the current bias on the transformer side, thereby eliminating the effect of the current bias on the control of the four-quadrant rectifier.
  • FIG. 6 is a schematic flow chart of a method for adjusting current bias of a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by an embodiment of the present invention
  • FIG. 7 is a high-power direct-drive electric locomotive provided by an embodiment of the present invention provided by the embodiment.
  • Udc is the DC bus voltage
  • the trap is mainly to filter the fluctuation value of the DC bus voltage
  • Udc * is the command voltage
  • i To input the AC current of the four-quadrant rectifier
  • Us is the voltage of the AC current input to the four-quadrant rectifier.
  • this embodiment describes the specific implementation process of this embodiment on the basis of the embodiment of FIG. 5.
  • the method includes:
  • S601 provided in this embodiment is similar to S501 in the embodiment of FIG. 5, and details are not described herein again in this embodiment.
  • S602. Filter the sampling current by a first band-pass filter and a second band-pass filter to obtain a filtered sampling current; wherein, the first band-pass filter is used to obtain a main frequency signal of an alternating current, The second band-pass filter is used to filter out interference harmonics.
  • the passband frequency of the first bandpass filter is set between 40 Hz and 60 Hz, for example, in this embodiment, the passband frequency of the first bandpass filter 45-55 Hz, optionally, when the main frequency of the AC current is 50 Hz, the passband frequency of the first band-pass filter is set to 50 Hz, for acquiring the main frequency signal of the AC current.
  • the switching frequency of the four-quadrant rectifier is f, that is, the on-off frequency of the IGBT is f
  • the pass band frequency of the second band-pass filter is 2f / (50 ⁇ 5) Hz
  • the second band The pass filter is used to filter out high-order harmonic interference.
  • the first band-pass filter and the second band-pass filter are the filters in FIG. 5.
  • S603 Obtain a second difference between the DC bus voltage of the four-quadrant rectifier and the command voltage, and input the second difference to the second PI controller, so that the third output value output by the second PI controller Multiplied by the output value of the phase-locked loop, the phase-locked loop is used to obtain the grid voltage phase, thereby obtaining an alternating current with the same period and phase as the grid voltage.
  • the DC bus voltage Udc and the command voltage Udc * are input to the second PI controller.
  • the control amount is the third output value output by the second PI controller.
  • the third output value output by the second PI controller is multiplied by the output of the phase-locked loop to obtain an alternating current in the same phase as the grid voltage.
  • the phase-locked loop is the PLL in FIG. 5.
  • the phase-locked loop PLL is used to control the period and phase of the alternating current i and the period and phase of the grid voltage to be consistent.
  • the phase of the grid voltage is calculated according to the phase controlled by the phase-locked loop.
  • the second PI controller in S603 is the second PI in FIG. 7.
  • the phase of the grid voltage is calculated according to the phase controlled by the phase-locked loop PLL, the phase of the alternating current i is determined, and the phase of the sampling current is determined.
  • the sampling current is divided into a positive half cycle and a negative half cycle.
  • the positive half cycle of the sine curve is 0 to ⁇
  • the negative half cycle is ⁇ to 2 ⁇
  • the values of the multiple sampling points of the positive half cycle are the values of the positive half cycle of the AC current i
  • the values of the multiple sampling points of the negative half cycle The value is the value of the negative half cycle of the alternating current i.
  • S604 is the DC offset extraction calculation in FIG. 7.
  • S605 Acquire a first sum value of the current value of the positive half cycle and a second sum value of the current value of the negative half cycle, and obtain a current offset value according to the first sum value and the second sum value.
  • S605 provided in this embodiment is similar to S502 in the embodiment of FIG. 5, and S605 is also the calculation of the DC offset extraction in FIG. 7, which will not be repeated here in this embodiment.
  • the size of the Q value and the hysteresis loop width are calculated.
  • the hysteresis loop width can be ⁇ 5A, or any other value, as long as it can avoid becoming the first There is only an error in the difference Q.
  • the hysteresis loop width is ⁇ 5A; the absolute value of the first difference Q is greater than 5A, and the obtained judgment result is yes, that is, the AC has a DC bias.
  • the first difference Q is greater than 5A, the AC current has a positive DC bias, the first difference Q is less than -5A, and the AC current has a negative DC bias.
  • S607 Input the first difference between the current offset value and zero to the first PI controller, and obtain the first output value output by the first PI controller.
  • S607 provided in this embodiment is similar to S503 in the embodiment of FIG. 5, and the first PI controller in S607 is the first PI in FIG. 7, which will not be repeated here in this embodiment.
  • S608 Summing the first output value and the second output value of the PR control output to obtain a third sum value, the first output value is a current variable, and the second output value is a current value;
  • the pulse width modulation symbol is obtained according to the third sum value and the unipolar frequency doubling pulse modulation method.
  • S608 provided in this embodiment is similar to S504 in the embodiment of FIG. 5, and the PR controller in S608 is a PR in FIG. 7, which will not be repeated here in this embodiment.
  • S609 provided in this embodiment is similar to S505 in the embodiment of FIG. 5 and is also similar to the pulse modulation in FIG. 7, which will not be repeated here in this embodiment.
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives provided by the embodiments of the present invention samples the AC current to obtain the sampled current, and then inputs the second difference between the DC bus voltage and the command voltage to the second PI controller, A third output value output by the second PI controller is obtained, and the third output value is used to adjust the alternating current.
  • the phase of the AC current is determined according to the phase of the grid voltage calculated by the phase-locked loop, and then the phase of the sampled current is determined, and then the sampled current is divided into positive half periods and For the negative half cycle, calculate the current value of the positive half cycle and the negative half cycle, and then input the first difference between the current value of the positive half cycle and the current value of the negative half cycle to the first PI controller.
  • the first output value output by a PI controller adjusts the second output value output by the PR controller to obtain a third sum value, thereby suppressing the DC bias of the alternating current, and modulating the third sum value with a unipolar frequency-doubled pulse Modulation mode, the pulse width modulation symbol is used to control the operation of the IGBT, which avoids the IGBT device from deviating from its rated operating area, thereby effectively suppressing and eliminating the current bias on the transformer side, thereby eliminating the current bias control of the four-quadrant rectifier influences.
  • the current bias adjustment method of the high-power direct-drive permanent magnet electric drive system for electric locomotives improves the response speed of DC bias suppression, and adopts a software control algorithm to solve the DC bias, eliminating the need for
  • the hardware circuit design solves the problem that other DC bias suppression methods are not suitable for wide-band changes of grid voltage frequency.
  • a control method for the intermediate DC loop in S102 is provided, which specifically relates to a method for chopper control of the intermediate DC loop, so as to reduce The impact on the direct DC bus voltage in the power direct drive permanent magnet electric drive system.
  • the chopping control method of the intermediate DC circuit provided in this embodiment will be described below with reference to FIGS. 8 and 9.
  • FIG. 8 is a schematic flowchart of Embodiment 1 of the chopping control method provided by the present invention.
  • the chopping control method provided by this embodiment includes:
  • S801 Perform periodic detection on the intermediate DC bus voltage, where the intermediate DC bus voltage is the voltage on the DC bus on the AC-DC-AC electric locomotive.
  • FIG. 9 is a schematic structural view of an embodiment of a high-power direct-drive permanent magnet electric drive system for electric locomotives of the present invention.
  • the high-power direct-drive permanent magnet electric drive system for electric locomotives shown in FIG. 9 is a possible connection method based on FIG. 1.
  • the high-power direct-drive permanent magnet electric drive system for electric locomotive shown in Figure 9 includes four-quadrant rectifier module 1 and four-quadrant rectifier module 2, chopper module 1 and chopper module 2, ground detection module, inverter module 1, and inverse Transformer module 2 and inverter module 3, and auxiliary modules.
  • the four-quadrant rectifier module 1 is composed of eight switch tubes g1, g3, g2, g4, g5, g7, g6 and g8.
  • the four-quadrant rectifier module 2 and the four-quadrant rectifier module 1 have the same structure.
  • the chopper module 1 includes a chopper switch g9, a chopper current sensor A2, a reverse diode D1, and a chopper resistor R5.
  • the chopper module 2 and the chopper module 1 have the same structure.
  • the grounding detection module includes resistors R3 and R4, and the resistance value of R3 is equal to R4.
  • the resistors R3 and R4 are connected in series at both ends of the DC loop to form a grounding resistance detection loop.
  • the inverter module 1 includes a three-phase inverter circuit composed of six switch tubes g10, g11, g12, g13, g14, and g15.
  • the inverter module 2, the inverter module 3, and the inverter module 1 have the same structure.
  • K2 is a motor isolation contactor
  • M is a direct-drive permanent magnet motor
  • C1 and C3 are DC-side supporting capacitors
  • R2 is a slow discharge resistor
  • U1 is a DC bus voltage sensor.
  • the auxiliary module includes a three-phase inverter circuit composed of six switch tubes, g16, g17, g18, g19, g20 and g21, and an auxiliary filter cabinet.
  • the intermediate DC bus voltage mentioned in this embodiment refers to the voltage measured by U1.
  • the principle of the P regulator is to control the chopper tube to be in an open state within a certain time proportion of the detection cycle.
  • the specific time ratio is related to the detected intermediate DC bus voltage value. The larger the detected intermediate DC bus voltage value, the greater the time ratio.
  • the chopper tube is not always in the open state. Compared with the prior art, the intermediate DC bus is reduced. The impact of voltage.
  • the chopper tube is directly controlled to be turned off.
  • the chopping control method provided in this embodiment is applied to AC-DC-AC electric drive locomotives to periodically detect the intermediate DC bus voltage.
  • the P regulator is used to The intermediate DC bus voltage is adjusted; until the detected value of the intermediate DC bus voltage is less than the lower chopping threshold, the impact on the intermediate DC bus voltage is reduced.
  • S802 includes:
  • S1001 Using the P regulator, determine the chopping duty cycle within the target detection period.
  • the target detection period includes: the detected detection period between the detected intermediate DC bus voltage value being greater than the upper chopping threshold and the detected intermediate DC bus voltage value being less than the lower chopping threshold.
  • the detection period is 1min
  • the voltage value of the intermediate DC bus detected in the current detection period (1min) is greater than the upper chopping threshold
  • the P regulator will be used to adjust the intermediate DC bus voltage. If the middle DC bus voltage value is less than the lower chopping threshold in the fifth detection period from the current detection period, the current 1min, the second 1min, the third 1min, and the fourth 1min are the target detection period.
  • the chopping duty ratio refers to: the ratio of the time that the chopper tube is turned on to the detection period within one detection period.
  • the above achievable way of determining the chopping duty cycle within the target detection period is:
  • the target parameter is determined according to the following formula
  • Err represents the target parameter
  • U1 represents the intermediate DC bus voltage value detected in the target detection period
  • control coefficient corresponding to the P regulator is obtained, specifically:
  • Kp_chp 1 / (DC bus voltage overvoltage protection threshold-chopping lower threshold)
  • Kp_chp represents the control coefficient
  • the chopping duty ratio is determined, specifically:
  • C_duty represents the chopping duty ratio
  • Err represents the target parameter
  • Kp_chp represents the control coefficient
  • the upper chopping threshold is set to 3100V
  • the lower chopping threshold is set to 2900V
  • the DC bus voltage overvoltage protection threshold is 3200V.
  • the voltage measured by U1 in Figure 9 is the intermediate DC bus voltage. Assuming that the detected intermediate DC bus voltage value U1 in the current detection cycle is 3100V, since U1 is greater than the upper chopping threshold, a P regulator is used to adjust the intermediate DC bus voltage.
  • S1002 Determine the turn-on time of the chopper tube in the target detection period according to the chopper duty ratio.
  • the chopping duty ratio refers to: the ratio of the time that the chopper is turned on in the detection period within a detection period.
  • the opening time of the chopper tube in the current detection period can be controlled to be 0.66 min based on the opening time by controlling the opening or closing of the chopper tube.
  • the chopping control method provided in this embodiment describes a achievable way to determine the chopping duty ratio. Specifically, the target parameter Err is first determined, then the control coefficient of the P regulator is determined, and finally the target parameter and The control coefficient determines the chopping duty ratio, which provides a basis for subsequently controlling the opening time of the chopper tube according to the chopping duty ratio.
  • the chopping control method provided in this embodiment further includes: performing error prevention processing on the chopping duty ratio.
  • the implementation of the above error prevention processing is:
  • the upper chopping threshold is set to 3100V
  • the lower chopping threshold is set to 2900V
  • the DC bus voltage overvoltage protection threshold is 3200V.
  • the voltage measured by U1 in Figure 2 is the intermediate DC bus voltage. It is assumed that the voltage value of the intermediate DC bus detected in the current detection period is 3300V.
  • the control coefficient Kp_chp calculated according to S2012 1 / (3200V-2900V) ⁇ 0.0033
  • the chopping control method provided in this embodiment describes an implementable method of performing error prevention processing on the chopping duty ratio. Specifically, if the value of the chopping duty ratio is greater than 1, the chopping duty ratio is The value of the duty ratio is set to 1; if the value of the chopping duty ratio is less than 0, the value of the chopping duty ratio is set to 0. The ratio of the chopping duty cycle can be controlled in the range of 0 to 1.
  • an embodiment of the present invention also provides a method for controlling a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for electric locomotives, using a speed-based
  • the segmented vector control strategy completes current closed-loop control to meet the requirements for high-speed operating range, high torque performance, and high efficiency according to the operating conditions of the locomotive.
  • FIG. 13 is a schematic flowchart of a control method for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention.
  • the embodiment shown in FIG. 13 includes:
  • S1302 Determine a first control strategy according to a rotation speed and a first mapping relationship, where the first mapping relationship includes a one-to-one correspondence between at least one rotation speed range and at least one control strategy;
  • S1303 Determine the expected control phase angle of the direct-drive permanent magnet synchronous motor to be controlled according to the first control strategy.
  • the first mapping relationship in the foregoing embodiment includes at least: a correspondence relationship between the rated speed below and the MTPA control strategy; a correspondence relationship above the rated speed with the field weakening control strategy.
  • the direct-drive permanent magnet synchronous motor in this embodiment uses a speed-based segmented vector control strategy to complete the current closed-loop control.
  • the control strategy includes: maximum torque current ratio (MTPA) control in the low-speed region and weak in the high-speed region Magnetic control.
  • MTPA maximum torque current ratio
  • FIG. 14 is a schematic structural diagram of a control system for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention, and the above embodiment will be described below in conjunction with FIG. 14.
  • T_cmd is the input torque
  • T is the actual input torque after torque limiting
  • id * and iq * are the d-axis and q-axis current settings
  • id and iq are the d-axis and q-axis feedback current
  • ud * and uq * are given by d-axis and q-axis voltage
  • ua, ub, uc are input phase voltage of motor a-phase, b-phase and c-phase, respectively
  • ia, ib are motor a-phase, b-phase Current.
  • MTPA control is adopted, that is, a control method that uses the reluctance torque generated by the salient pole effect of a permanent magnet synchronous motor to obtain a higher torque-current ratio. It is also called maximum torque current ratio control, and its control implementation block diagram is shown in FIG. 15, which is a schematic diagram of the system structure of the MTPA control system of the present invention.
  • MTPA control is a control strategy adopted under non-weak magnetic field. Since the straight-axis inductance Ld of the salient pole motor is less than the cross-axis inductance Lq, the reluctance of the motor can be used when the motor is running below the rated speed. Torque to obtain a higher torque-current ratio.
  • the key of this strategy is to set the correct current operating point, and the dynamic response of the system is realized by the optimized current inner loop control.
  • the current current inner loop commonly has feedforward decoupling control, feedback decoupling control, and internal model decoupling control. And deviation decoupling control. Aiming at the problem that the system is under high acceleration and deceleration conditions, the d and q axis currents have serious dynamic coupling and affect the dynamic performance of the system.
  • An optimized feedforward decoupling control strategy is used to achieve optimal control of the current inner loop.
  • the MTPA control block diagram is shown in Figure 15. Among them, udf and uqf are the feedforward voltage of d axis and q axis respectively.
  • Feed-forward decoupling is to add decoupling voltage terms at the output signals u sd and u sq of the current controller, respectively with So as to cancel the coupling effect between excitation and torque current.
  • the MTPA control specifically includes the following steps: determining the q-axis current reference and the d-axis current reference according to the torque current curve; calculating the first difference between the q-axis current reference and the q-axis actual current and the d-axis current reference and The second difference value of the d-axis actual current; the d-axis voltage reference is obtained according to the first difference value through the first PI controller, and the q-axis voltage reference is obtained according to the second difference value through the second PI controller; The sum of the given and q-axis feedforward voltages gives the actual q-axis voltage reference, and the sum of the d-axis voltage reference and the d-axis feedforward voltage is calculated to get the actual d-axis voltage reference.
  • the given d-axis current given id * and q-axis current given iq * are determined according to the input and torque current curve, and then the id * and d-axis actual current id are subtracted and sent to PI
  • the controller subtracts iq * and the q-axis actual current iq and sends it to the PI controller.
  • the two PI controllers will calculate d-axis voltage given ud and q-axis voltage given uq.
  • the calculated d-axis voltage given ud is added to the d-axis feedforward voltage udf to obtain ud * as the actual output d-axis voltage given, and the calculated q-axis voltage given uq is added to the q-axis before The feed voltage uqf is given by uq * as the actual output q-axis voltage.
  • FIG. 16 is a schematic structural diagram of a system for front-end decoupling control of the present invention. As shown in Figure 16, assuming that the back EMF component has been cancelled, front-end decoupling control is required. Among them, according to the front-end structure control block diagram in FIG. 16, the voltage calculation equation of the front-end structure that can be written as a matrix is:
  • FIG. 17 is a schematic diagram of the system structure of the field weakening control of the present invention. Due to the limited capacity of the system converter, when the permanent magnet synchronous motor runs in steady state, the terminal voltage and stator current will be idle, and the voltage and current limit values cannot be exceeded. To further widen the speed regulation range, the field weakening control is adopted at the rated speed In the above, the permanent magnet synchronous motor enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current.
  • the control algorithm based on the above control strategy is used to calculate and obtain the current d-axis voltage given value and the current q-axis voltage given value, and further, obtain the current expected control according to the current d-axis voltage given value and the current q-axis voltage given value Phase angle.
  • the terminal voltage us and the stator current is limited, and cannot exceed the voltage and current limit values.
  • Field weakening control The permanent magnet synchronous motor above the rated speed enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current; the current loop adopts the power angle control strategy.
  • the voltage applied by the inverter on the motor is not controllable, only through The power angle ⁇ of the motor is controlled to adjust the excitation and torque of the motor.
  • the output of the PI regulator controls the power angle to realize the control of the power angle above the fundamental frequency of the permanent magnet motor.
  • Usmax and Ismax are voltage limit value and current limit value respectively
  • ⁇ id is the change of excitation current in a given field weakening state
  • id_wk * and iq_wk * are given d-axis and q-axis current after field-weakening adjustment
  • uf is the amplitude of the feedforward voltage
  • ⁇ is the power angle.
  • the field weakening control specifically includes the following steps: the PI controller calculates the d-axis current change amount in the given field weakening state according to the difference between the voltage limit value and the feedforward voltage amplitude; the d-axis current in the given field weakening state The sum of the amount of change and the d-axis current setting gives the d-axis current setting after the field weakening adjustment; the q-axis current setting after the field weakening adjustment is calculated according to the d-axis current setting and the torque formula; according to the q-axis through the PI controller The difference between the current setting and the q-axis actual current is the power angle ⁇ ; the actual q-axis voltage setting and the actual d-axis voltage setting are calculated by the following formula;
  • Us is the voltage limit value
  • Ud is the actual d-axis voltage given
  • Uq is the actual q-axis voltage given.
  • the difference between the q-axis current reference and the q-axis actual current iq is sent to the PI controller, and the PI controller obtains the power angle ⁇ .
  • the actual q-axis voltage reference and the actual d-axis voltage reference are calculated according to the above formula As output.
  • FIG. 18 is a schematic diagram of the trajectory of MTPA control and field weakening control in the full speed range of the present invention.
  • the OA segment is the MTPA control trajectory
  • the AB and BC segments are the field weakening control trajectory
  • ⁇ r1 is the rated speed
  • ⁇ r2 is the highest speed
  • - ⁇ f / Ld is the center of the voltage limit circle.
  • FIG. 19 is a schematic diagram of switching control of MTPA control and field weakening control of the present invention.
  • an embodiment of the present invention also provides a modulation method for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent magnet electric drive system for electric locomotives, by calculating the modulation phase Angle to achieve the actual control phase angle through PWM modulation.
  • the high power of the traction converter of the high-power traction drive system Due to the high power of the traction converter of the high-power traction drive system, affected by the heat dissipation of the switching device and the switching loss, it needs to work at a lower switching frequency, usually not exceeding 1000 Hz. On the one hand, the highest switching frequency is generally It is about 100 Hz. On the other hand, when the output reaches the rated value, it works in the square wave mode. Therefore, in the entire speed range, the variation range of the carrier ratio is very large.
  • this embodiment provides a multi-mode PWM modulation strategy, on the one hand, it can make full use of the allowable switching frequency of the inverter, and on the other hand, it can ensure a high DC voltage utilization rate after entering the field weakening control area.
  • 20 is a schematic flowchart of a modulation method for a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent magnet electric drive system for an electric locomotive provided by the present invention; as shown in FIG. 20, the method provided in this embodiment includes:
  • S2002 Determine the first modulation strategy according to the range of the frequency of the modulated wave and the second mapping relationship.
  • the second mapping relationship includes a one-to-one correspondence between the frequency range of at least one modulated wave and the at least one modulation strategy.
  • S2003 Determine the PWM carrier frequency of the direct drive permanent magnet synchronous motor according to the first modulation strategy.
  • the second mapping relationship at least includes: corresponding to the asynchronous modulation strategy when the frequency of the modulated wave is in the low-speed stage; corresponding to the synchronous modulation strategy of 60 degrees in the middle when the frequency of the modulated wave is greater than that in the low-speed stage; In the high-speed phase, it corresponds to the square wave modulation strategy.
  • the multi-mode PWM modulation strategy is mainly composed of asynchronous SPWM modulation, synchronous SPWM modulation and square wave modulation. among them,
  • Asynchronous modulation strategy is adopted in the low-speed phase; when the asynchronous modulation has a large carrier ratio, the positive and negative half-cycle asymmetry caused by the asynchronous modulation mode has less influence, and the introduced low-order harmonics can be ignored. 2.
  • the middle 60-degree synchronous modulation strategy is adopted; as the motor frequency rises and the carrier ratio decreases, the impact of this low-order harmonic is getting larger and larger, and synchronous modulation PWM is used at this time.
  • the conventional regular sampling synchronous modulation has a high content of low-order harmonics when the carrier ratio is relatively low, and the amplitude of the fundamental wave voltage obtained by sampling cannot meet the requirements of the command value, which is not conducive to entering the square wave.
  • a special modulation method should be used. , So that the current has better harmonic characteristics and symmetry, and smoothly enter the square wave. 3.
  • square wave modulation is used; the traction inverter outputs a higher fundamental wave voltage to increase the maximum output torque of the traction motor. It will operate in the square wave working condition in the high-speed section, and the modulation method uses square wave modulation .
  • the specific low speed and high speed in the process of acquiring the current modulation phase angle are both the angular speed of the rotor, and the specific division rule may be similar to the division rule in the prior art.
  • FIG. 21 is the relationship between the modulation angle and the modulation ratio under the intermediate 60 ° modulation mode provided by the present invention
  • FIG. 22 is a schematic diagram of the full speed range modulation strategy based on the intermediate 60 ° modulation provided by the present invention.
  • the asynchronous modulation strategy is used in the low-speed phase; when the speed increases, the regular sampling synchronous modulation and the intermediate 60-degree synchronous modulation strategy with different carrier ratios are used; the high-speed phase uses square wave modulation.
  • the switching process involved mainly includes the switching between asynchronous modulation to SVPWM synchronous modulation, the switching between synchronous modulation SVPWM and intermediate 60 ° modulation, and the internal 60 ° modulation.
  • the main difficulty in switching is the switching between synchronous modulation SVPWM and intermediate 60 ° modulation.
  • SVPWM synchronous modulation
  • intermediate 60 ° modulation there are 15 carriers per fundamental cycle, and the phase of the fundamental wave corresponding to each carrier is 24 °, while at the mid-seventh modulation of 60 °, the phase of the fundamental wave corresponding to each carrier cycle is 20 ° .
  • the phase at the switching point must be a common multiple of the phase corresponding to each carrier cycle before and after switching, 20 ° and 24 °
  • the common multiple of is 120 °, which means that only three points can be switched in a cycle, namely 0 °, 120 °, and 240 °, and each corresponds to one of the points during the switching process. If the leakage inductance of the motor is small, it may cause a certain impact during the switching process, and the other two switching processes can achieve shockless switching.
  • the abscissa in this embodiment is the frequency of the modulated wave obtained by the modulation algorithm in this embodiment.
  • the ordinate is the PWM carrier frequency.
  • the relationship between the modulation angle ⁇ and the modulation ratio at the middle 60 ° nineth frequency division, seventh frequency division, fifth frequency division, and third frequency division is shown. It shows that through the middle 60 ° modulation method in this embodiment, if the influence of the dead zone is not taken into account, it is possible to ensure that the actual output voltage and the reference value are completely coincident, with a very high voltage control accuracy.
  • the intermediate 60 ° synchronous modulation can achieve symmetry between the three phases of the output voltage waveform when the number of pulses is not a multiple of 3, and each phase Positive and negative half cycle and 1/4 cycle symmetry, so that the motor line voltage and current only contain 6k ⁇ 1 harmonic;
  • the switch angle under this modulation mode can be calculated online in real time, and the required calculation amount is very small .
  • the implementation process has relatively low hardware requirements, and the pulse is relatively easy to send; (3) Through digital control, the middle 60 ° modulation can accurately output the required fundamental voltage, and the maximum output voltage under different pulse numbers does not consider the minimum pulse width Can be directly transferred to the square wave; (4) When the number of pulses in the middle 60 ° modulation is greater than 9, the current harmonics cannot be significantly improved. Different pulse numbers have consistent low-order current harmonic characteristics, resulting in low-order torque ripples with stable and relatively large ripple amplitudes under different pulse numbers and modulation ratios; (5) Intermediate 60 ° modulation The trajectories of the stator flux linkage of the motor are all hexagonal trajectories. The increase in the number of pulses only increases the number of voltage zero vectors in each sector, that is, the number of pauses of the stator flux linkage.
  • a method for detecting the initial position angle of the rotor of a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for electric locomotives is also provided.
  • a method for detecting the initial position angle of the rotor of a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for electric locomotives is also provided.
  • FIG. 23 is a schematic flowchart of Embodiment 1 of a method for detecting a rotor initial position angle of a direct-drive permanent magnet synchronous motor provided by the present invention.
  • the main body of the method for detecting the initial position angle of the rotor of the direct drive permanent magnet synchronous motor provided in this embodiment is the device for detecting the initial position angle of the rotor of the direct drive permanent magnet synchronous motor provided by the present invention, for example, the device is a TCU control device .
  • the method of this embodiment includes:
  • S2301 Inject a high-frequency voltage signal to the stator winding of the direct-drive permanent magnet synchronous motor to be detected to obtain the three-phase stator winding current.
  • the coordinate system involved in the present invention includes a two-phase synchronous rotating coordinate system, a two-phase stationary coordinate system, and an expected two-phase synchronous coordinate system.
  • FIG. 24 is a schematic diagram of the relationship between the two-phase synchronous rotating coordinate system, the two-phase stationary coordinate system, and the expected two-phase synchronous rotating coordinate system provided by the present invention.
  • the ⁇ ⁇ coordinate system is a two-phase stationary coordinate system
  • the dq coordinate system is a two-phase synchronous rotating coordinate system.
  • the coordinate system is an expected two-phase synchronous rotating coordinate system.
  • ⁇ is the actual rotor position angle
  • ⁇ is the rotor position angle estimation error
  • a possible implementation is to inject a high-frequency voltage signal as shown in the following formula into the expected two-phase synchronous rotating coordinate system:
  • U mh is the amplitude of the high-frequency voltage signal
  • ⁇ h is the angular frequency of the high-frequency voltage signal
  • t represents the time when the high-frequency voltage signal is injected.
  • the two components of the high-frequency voltage signal injected into the stator winding of the direct drive permanent magnet synchronous motor are linearly independent, and thus the inductance parameters of the direct drive permanent magnet synchronous motor can be obtained.
  • the inductance parameters of the direct drive permanent magnet synchronous motor can be obtained according to the mathematical model and related calculation method of the direct drive permanent magnet synchronous motor established in the prior art.
  • the response current of the stator winding is obtained, and the response current is the three-phase stator winding current.
  • the three-phase stator winding current can be obtained through a current sensor.
  • the three-phase stator winding current can be represented by i a , i b and i c .
  • both the d-axis target current and the q-axis target current are injected high-frequency voltage signals, and the corresponding current components are excited on the stator winding according to the structure of the direct drive permanent magnet synchronous motor and the magnetic saturation characteristics.
  • the d-axis target current Both the q-axis target current and the rotor position angle estimation error are related. By performing signal processing on the d-axis target current and the q-axis target current, the initial rotor position angle can be obtained.
  • a possible implementation method is to first perform Clarke transformation on the three-phase stator winding currents i a , i b and i c to obtain the ⁇ -axis current i ⁇ and ⁇ -axis current i ⁇ in the two-phase stationary coordinate system, , And then Park transform the ⁇ -axis current and ⁇ -axis current to obtain the d-axis target current And q-axis target current
  • the d-axis target current And q-axis target current Both are related to the rotor position angle estimation error ⁇ .
  • the above-mentioned initial position angle is the initial position angle compensated according to the polarity of the magnetic pole of the direct drive permanent magnet synchronous motor.
  • the q-axis target current Contains the initial rotor position information, therefore, the q-axis target current can be signal processed to extract the initial rotor position angle.
  • the polarity information of the pole of the direct-drive permanent magnet synchronous motor is related to the d-axis inductance. Therefore, the polarity information of the pole can be obtained according to the nonlinear magnetization characteristic of the d-axis inductance of the direct-drive permanent magnet synchronous motor.
  • the initial position angle of the rotor is compensated according to the polarity of the magnetic pole, thereby obtaining the compensated initial position angle, and the compensated initial position angle is determined as the initial position angle of the rotor.
  • a high-frequency voltage signal is first injected into the stator winding of the direct-drive permanent magnet synchronous motor to be detected to obtain the three-phase stator winding current, and then the d under the expected two-phase synchronous rotating coordinate system is obtained according to the three-phase stator winding current Axis target current and q-axis target current, further, the initial position angle of the rotor is obtained according to the d-axis target current and the q-axis target current, where the initial position angle is the initial value after compensation according to the polarity of the magnetic pole of the direct drive permanent magnet synchronous motor Position angle.
  • the method provided by the present invention compensates the initial position angle of the rotor according to the polarity of the magnetic pole by considering the influence of the magnetic pole of the direct-drive permanent magnet synchronous motor.
  • the obtained initial position angle of the rotor is more accurate and improves the detection of the initial position angle Reliability.
  • the method provided by the present invention can also obtain high-accuracy detection results under the condition that the rotor is stationary, and has a wide application range.
  • the method provided by the present invention does not need to consider the parameters of the direct drive permanent magnet synchronous motor, and is easier to implement.
  • S2303. Obtaining the initial position angle of the rotor according to the d-axis target current and the q-axis target current may be implemented in the following ways:
  • the first initial position angle of the rotor is obtained according to the q-axis target current.
  • a possible implementation method when the rotor position angle estimation error ⁇ is zero, the q-axis target current Is zero, for the q-axis target current Signal processing is performed to obtain the error input signal of the rotor position angle, and the initial position angle of the rotor is obtained according to the error input signal.
  • the rotor pole compensation angle is obtained according to the d-axis target current.
  • the polarity information of the pole of the direct drive permanent magnet synchronous motor is related to the d-axis inductance. Therefore, the polarity information of the pole can be obtained according to the nonlinear magnetization characteristic of the d-axis inductance of the direct drive permanent magnet synchronous motor.
  • the initial position angle of the rotor is obtained.
  • the first initial position angle is compensated by using the magnetic pole compensation angle, and the compensated first initial position angle is determined as the initial position angle of the rotor.
  • FIG. 25 is a schematic flowchart of Embodiment 2 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention. As shown in FIG. 25, obtaining the first initial position angle of the rotor according to the q-axis target current may include:
  • S2501 Perform low-pass filtering on the q-axis target current to obtain an error input signal.
  • the error input signal is an error signal related to the initial position angle of the rotor.
  • One possible implementation method is to modulate the q-axis target current by using a modulation signal to obtain the modulated q-axis target current, and further, perform low-pass filtering on the modulated q-axis target current to obtain an error input signal.
  • the modulated q-axis target current is expressed as
  • the modulated q-axis target current is filtered by a low-pass filter to filter out the signal component of double frequency to obtain the error input signal f ( ⁇ ), where,
  • LPF stands for low-pass filtering
  • the error input signal includes the rotor position estimation error.
  • the process of low-pass filtering consider the effect of filter phase delay on the extracted signal, and consider adding delay compensation during implementation to ensure that the high-frequency voltage injection phase is consistent with the estimated angle phase.
  • the error input signal is used as the input of the PI regulator of the phase-locked loop.
  • the PI regulator obtains the proportional deviation and integral deviation of the error input signal according to the input error signal. Further, according to the linear combination of the proportional deviation and integral deviation, the The first initial position angle.
  • the first initial position angle can be obtained by the following formula:
  • s represents the Laplace operator
  • k p is the coefficient of proportional term
  • k i is the coefficient of integral term
  • Adjusting the proportional coefficient and integral coefficient of the PI regulator causes f ( ⁇ ) to converge, and the output term of the PI regulator is the rotor's first initial position angle ⁇ first .
  • the error input signal is obtained by modulating the q-axis target current and low-pass filtering, and further, a PI regulator is used to phase-lock the output of the error input signal to obtain the first initial position angle.
  • FIG. 26 is a schematic flowchart of Embodiment 3 of a method for detecting a rotor initial position angle of a permanent magnet synchronous motor provided by the present invention. As shown in FIG. 26, obtaining the rotor pole compensation angle according to the d-axis target current may include:
  • S2601 Inject a plurality of voltage pulse signals with the same voltage amplitude and different angles into the permanent magnet synchronous motor to obtain the response current of each voltage pulse signal.
  • the poles of permanent magnet synchronous motors have nonlinear saturation characteristics. Specifically, a voltage pulse signal is injected into the d-axis of the permanent magnet synchronous motor. When the angle of the voltage pulse signal is closer to the N pole of the permanent magnet synchronous motor, the magnitude of the response current is larger; when the angle of the voltage pulse signal is farther away from the permanent magnet For the N pole of a synchronous motor, the smaller the magnitude of the response current.
  • the d axis is the straight axis of the permanent magnet synchronous motor
  • the q axis is the intersection axis of the permanent magnet synchronous motor.
  • a possible implementation method is to inject a plurality of voltage pulse signals with a preset angle and equal amplitude into the permanent magnet synchronous motor, and to sample through the current sensor to obtain the response current of the multiple voltage pulses and further obtain the amplitude of the response current
  • the changing law For example, a permanent magnet synchronous motor is injected with a voltage pulse signal of equal amplitude every 5 °.
  • the preset angle may also be smaller or larger, which is not limited in the present invention. It should be noted that the smaller the preset angle, the more response current data is obtained, and the accuracy of the change law of the amplitude of the response current is higher. The larger the preset angle, the response current data is obtained. The less the accuracy of the change law of the amplitude of the response current is, the more appropriate the preset angle can be selected according to the actual situation in the actual application process.
  • Another possible implementation method is to inject a plurality of voltage pulse signals of equal angle and equal amplitude into the permanent magnet synchronous motor, and sample through the current sensor to obtain the response current of the multiple voltage pulses and further obtain the response current The law of amplitude change.
  • S2602 Determine the magnetic pole compensation angle of the rotor according to multiple response currents.
  • the pole compensation angle of the rotor is determined according to the magnitudes of multiple response currents.
  • the rotor pole compensation angle is 0, where the first The value is the maximum value of the magnitude of multiple response currents.
  • the d-axis direction is determined to be the magnetic pole N-pole direction.
  • the rotor pole compensation angle is ⁇ , where the second The value is the minimum value of the magnitude of multiple response currents.
  • the d-axis direction is determined as the S-pole direction.
  • the initial position angle of the rotor is the sum of the first initial position angle and the pole compensation angle. Specifically, when the d-axis direction is determined as the N-pole direction, the initial position angle of the rotor is equal to the first initial position angle, and when the d-axis direction is determined as the S-pole direction, the initial position angle of the rotor is equal to the first initial position angle and the magnetic pole The sum of the compensation angle ⁇ .
  • the accuracy of the identification of the magnetic pole polarity obtained based on the nonlinear saturation characteristics of the permanent magnet synchronous motor straight shaft inductance is high, and in the implementation process, it is not necessary to consider the influence of the motor parameters of the permanent magnet synchronous motor, reliability Higher and easier to implement.
  • the inverter switching frequency is 500Hz
  • the motor rated power is 1200kW
  • the motor rated torque is 32606N.m
  • the rated voltage is 2150V
  • the rated current is 375A
  • the rated speed is 350r / min
  • the number of motor pole pairs is 7
  • the motor d-axis inductance Ld is 0.008771 H
  • the motor q-axis inductance Lq is 0.012732H.
  • the amplitude of the high-frequency voltage signal injected into the permanent magnet synchronous motor is 180V
  • the angular frequency of the high-frequency voltage signal is 200 Hz
  • the inverter switching frequency is 500 Hz.
  • FIG. 27 is a schematic diagram of the signal changes of the multiple channels during the operation of the permanent magnet synchronous motor.
  • the channels from top to bottom are: permanent magnet synchronous motor UV phase line voltage signal, permanent magnet synchronous motor U phase upper tube pulse signal, bus voltage signal, permanent magnet synchronous motor U phase current signal, permanent magnet Synchronous motor V-phase current signal.
  • FIG. 28 is a schematic diagram of the response current variation rule. As shown in FIG. 28, when the angle of the injected voltage pulse signal is closer to the N pole of the permanent magnet synchronous motor, the magnitude of the response current is larger; when the angle of the injected voltage pulse signal The farther away from the N pole of the permanent magnet synchronous motor, the smaller the magnitude of the response current.
  • the actual position angle of the rotor obtained by detecting the resolver is compared with the expected position angle of the rotor calculated according to the control algorithm.
  • the calculation error is about ⁇ 1.2 °, and the error is small.
  • a method for actually controlling the phase angle of a direct-drive permanent magnet synchronous motor in a high-power direct-drive permanent-magnet electric drive system for electric locomotives is also provided, To improve the accuracy of the actual control phase angle of direct drive permanent magnet synchronous motor.
  • FIG. 29 is a schematic structural diagram of a control system of a direct-drive permanent magnet synchronous motor corresponding to a high-power direct-drive permanent-magnet electric drive system for an electric locomotive provided by the present invention.
  • the direct-drive permanent-magnet synchronous motor The control system includes: direct drive permanent magnet synchronous motor, tractor, traction controller TCU, and resolver.
  • control object of the control method of the direct drive permanent magnet synchronous motor provided by the present invention is the direct drive permanent magnet synchronous motor, wherein the direct drive permanent magnet synchronous motor includes a stator and a rotor.
  • the resolver is installed on the rotor of the direct-drive permanent magnet synchronous motor and is used to collect the rotor signal and input the collected signal to the traction controller.
  • the resolver is specifically used to detect the actual position of the rotor.
  • the dragging machine is connected to the tested direct-drive permanent magnet synchronous motor, which is used to drive the direct-drive permanent magnet synchronous motor.
  • the traction controller is connected to the direct-drive permanent magnet synchronous motor and is used to control the direct-drive permanent magnet synchronous motor.
  • the traction controller is used to implement a speed-based segmented vector control strategy for the direct-drive permanent magnet synchronous motor, and the speed-based segmented vector control strategy will be described in detail in subsequent embodiments.
  • the traction controller has functions of a control algorithm and a modulation algorithm, and functions of phase angle adjustment and speed monitoring.
  • the traction controller in the present invention includes a control algorithm unit, a modulation algorithm unit, a phase angle regulator, and a speed detector.
  • the control algorithm unit is used to obtain the expected control phase angle
  • the modulation algorithm unit is used to obtain the modulated phase angle, and then the actual control phase angle is realized by PWM modulation
  • the phase angle regulator is used to realize the expected control phase angle and the actual control phase angle Always keep the same
  • the speed detector is used to obtain the angular velocity of the rotor.
  • the above-mentioned control algorithm unit, modulation algorithm unit, phase angle regulator, and speed detector can be either software modules or physical modules, which are not limited by the present invention.
  • control method of the direct-drive permanent magnet synchronous motor provided by the present invention is implemented by using a traction controller as an executive body.
  • FIG. 30 is a first schematic flowchart of a control method of a direct-drive permanent magnet synchronous motor provided by the present invention.
  • the main body of the execution of the method flowchart shown in FIG. 30 is a traction controller, which can be implemented by any software and / or hardware.
  • the control method of the direct drive permanent magnet synchronous motor provided by this embodiment includes:
  • S3001 Acquire the compensation phase angle of the rotor of the direct drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the direct drive permanent magnet synchronous motor.
  • the compensation phase angle of the rotor of the direct-drive permanent magnet synchronous motor obtained in this embodiment is an offline compensation phase angle, that is, if the components in the control system of the direct-drive permanent magnet synchronous motor are set to obtain the compensated phase angle and operate normally, When changing time, the compensated phase angle obtained offline can be applied to the control system of the running direct drive permanent magnet synchronous motor. It is conceivable that when the settings of various components in the control system of the direct drive permanent magnet synchronous motor are changed, the new setting phase parameters can be obtained using the changed setting parameters.
  • the traction controller may use a control algorithm to process the voltage signal collected by the resolver to obtain the expected phase angle.
  • the traction controller may control the control algorithm unit to process the voltage signal collected by the resolver to obtain Expected phase angle.
  • the sampling period of the resolver can be the same as the control interruption period of the control algorithm.
  • the resolver samples at time t1 and inputs the collected voltage signal to the traction controller.
  • the control algorithm unit of the traction controller processes the voltage signal collected by the resolver at time t1, obtains the expected phase angle, and updates it at an indefinite time between the beginning of the next control interruption period and the end of the next control interruption period. That is, the expected phase angle is output to the modulation algorithm unit.
  • the rotor is still rotating, and the control algorithm interruption delay will be generated relative to the resolver sampling time. Further, according to the length of the interruption delay of the control algorithm and the angular velocity of the rotor, the error phase angle of the rotor in the process of the control algorithm is obtained.
  • control algorithm delay is half a control interrupt period.
  • the traction controller obtains the expected phase angle and uses a modulation algorithm to modulate and output the expected phase angle.
  • the modulation algorithm unit of the traction controller uses the modulation algorithm to modulate the expected phase angle and output PWM pulses.
  • the modulation sampling in this embodiment has periodicity, that is, the traction controller periodically acquires the expected phase angle and performs modulation processing.
  • the modulation carrier is a triangular PWM carrier, and the modulation sampling adopts an asymmetric regular sampling method, that is, sampling at the position of the symmetry axis of the vertex of each triangular PWM carrier cycle, and at the bottom of the triangular PWM carrier cycle
  • the point symmetry axis is sampled, that is, sampled twice per modulated carrier cycle.
  • the sampling of this PWM carrier cycle is performed, and the PWM command of this cycle is updated at the same time.
  • the interruption of modulation algorithm in double sampling mode is divided into sampling, modulation calculation, PWM update and PWM output process.
  • the traction controller obtains the expected phase angle at time t2, performs PWM modulation processing, and generates PWM pulses. After that, the PWM pulse is usually output when the carrier cycle count value is equal to the PWM comparison count value calculated by modulation.
  • the modulation update delay is caused.
  • the modulation update delay is half a modulated carrier period;
  • the continuous pulse counting method of the timer is generally used to output the PWM pulse, and the output delay will also be caused during the output.
  • the output delay is 1/4 modulated carrier period.
  • the error phase angle of the rotor in the process of the modulation algorithm can be obtained.
  • a delay is also generated during the process of sampling and signal transmission of the rotor by the resolver, which is referred to as resolver sampling and transmission delay.
  • the error phase angle corresponding to the sampling and transmission delay of the resolver is obtained according to the multiple d-axis voltages and multiple q-axis voltages in the current angular velocity and the preset angular velocity range of the direct-drive permanent magnet synchronous motor rotor.
  • the segmented vector control strategy includes maximum torque-current ratio control in the low-speed region and field weakening control in the high-speed region. Therefore, the preset angular speed range in this embodiment may be a speed range where the traction controller determines that the direct-drive permanent magnet synchronous motor does not enter the field weakening control stage and operates stably. Among them, according to the traction characteristics of the direct drive permanent magnet synchronous motor, the speed point corresponding to the constant voltage phase, the operating speed when the voltage reaches the maximum value, that is, the maximum stable operating speed without entering the field weakening control phase, which is the preset angular speed The maximum value of the range.
  • the sum of the error phase angles corresponding to the above control algorithm delay, modulation algorithm delay, and resolver acquisition and transmission delay respectively is the compensation phase angle of the rotor of the direct drive permanent magnet synchronous motor.
  • the magnetic field generated by the rotor poles corresponds to the stator magnetic field as the d-axis, and the 90-degree counterclockwise rotation is the q-axis.
  • the compensated phase angle obtained in step S3001 is an offline compensated phase angle, which is applied to the running direct drive permanent magnet synchronous motor.
  • the current actual control phase angle acquired in this step is the actual control phase angle after the offline correction of the rotor position angle of the direct-drive permanent magnet synchronous motor using the compensated phase angle acquired in step S3001.
  • the current voltage given value may include a current d-axis voltage given value and a current q-axis voltage given value.
  • the current d-axis voltage given value and the current q-axis voltage given value are calculated and obtained according to the speed-based segmented vector control strategy adopted by the direct drive permanent magnet synchronous motor and the corresponding control algorithm. Further, according to the current The d-axis voltage reference value and the current q-axis voltage reference value obtain the current expected control phase angle.
  • the current expected control phase angle and the current actual control phase angle may be deviated by the control algorithm, the modulation algorithm, and the delay in the acquisition and transmission process of the resolver, there may be a deviation between the current expected control phase angle and the current actual control phase angle.
  • the current actual control phase angle is corrected.
  • the linear combination of the proportional deviation between the current expected control phase angle and the current actual control phase angle and the integral deviation between the current expected control phase angle and the current actual control phase angle is used as the correction term to perform online correction on the current actual control phase angle .
  • This embodiment provides a control method of a direct-drive permanent magnet synchronous motor.
  • the method includes: acquiring the rotor of the direct-drive permanent magnet synchronous motor according to the control interruption period, the modulation carrier period, and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor Compensation phase angle; according to the compensation phase angle, obtain the current actual control phase angle; according to the current d-axis voltage given value and the current q-axis voltage given value, obtain the current expected control phase angle; further, according to the current expected control phase angle and The proportional deviation and integral deviation of the current actual control phase angle can be corrected online.
  • the present invention corrects the actual control phase angle by taking into account the time delay corresponding to the control interruption, the time delay corresponding to the carrier modulation, and the error phase angle caused by the corresponding time delay in the process of sampling and transmitting the rotor signal of the resolver. Ensure that the actual control phase angle and the expected control phase angle are always consistent, and the accuracy of the actual control phase angle is improved.
  • FIG. 31 is a schematic flowchart of Embodiment 2 of a control method of a direct-drive permanent magnet synchronous motor provided by the present invention. As shown in FIG. 31, on the basis of the embodiment shown in FIG. 30, step S3001 may include:
  • S3101 Acquire the first sub-compensated phase angle according to the control interruption period and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor.
  • FIG. 32 is a schematic diagram of control interruption of the control algorithm provided by the present invention. As shown in Fig. 32, the control interruption is divided into the processes of sampling, control calculation, and control variable update.
  • the resolver samples the rotor signal and inputs the collected voltage signal to the traction controller at time t1.
  • the traction controller performs control calculation on the received voltage signal, T ctrl is a control interruption cycle of the control algorithm, the control calculation is completed at t1 + T ctrl time, and then begins at the next control interruption cycle (time t1 + T ctrl ) to end (Time t1 + 2T ctrl ) At the indefinite time within this period, the control variable calculated by the control is output to the modulation algorithm unit.
  • the rotor is still rotating continuously, and the control algorithm interruption delay will be generated relative to the time when the control calculation is completed.
  • the first phase angle delay corresponding to the first sub-compensated phase angle is obtained according to the control interruption period of the control algorithm, where A is the control interruption delay coefficient and the value range is (0-1).
  • A 0.5.
  • the first phase angle delay ⁇ t1 can be expressed as follows:
  • ⁇ t1 A ⁇ T ctrl ⁇ 0.5T ctrl
  • the first sub-compensated phase angle is obtained according to the first phase angle delay and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor, and the first sub-compensated phase angle is the error phase angle corresponding to the control algorithm interruption delay.
  • the first sub-compensation phase angle ⁇ cmps1 can be expressed as follows:
  • ⁇ is the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor.
  • the modulation algorithm uses an asymmetric regular sampling method, that is, in The position of the symmetric axis of the vertex of each triangular PWM carrier cycle is sampled, and the position of the axis of symmetry of the bottom point of the triangular PWM carrier cycle is also sampled, that is, sampled twice per modulated carrier cycle.
  • the sampling of this PWM carrier cycle is performed, and the PWM command of this cycle is updated at the same time.
  • the interruption of modulation algorithm in double sampling mode is divided into sampling, modulation calculation, PWM update and PWM output process.
  • FIG. 33 is a schematic diagram of an interruption cycle of a modulation algorithm provided by the present invention.
  • the traction controller performs modulation sampling at time t, and obtains the control variables calculated by the control algorithm.
  • the control variable obtained by the traction controller is the expected phase angle
  • the modulation algorithm calculation is completed at t + 0.5T PWM time, and the PWM comparison count value update and the expected control phase angle sampling for the next modulation cycle are started.
  • the PWM carrier cycle count value is equal to the PWM comparison count value calculated by the modulation
  • T PWM is the PWM modulated carrier cycle.
  • the rotor is still rotating continuously.
  • the modulation algorithm interruption delay will be generated, which is the third phase angle delay B ⁇ T PWM , where B is the modulation algorithm interruption ⁇ efficient ⁇ Extension coefficient.
  • B 0.5.
  • the timer's continuous up and down counting method is generally used to output the PWM pulse.
  • the PWM pulse output delay is generated.
  • the PWM pulse output delay is C ⁇ T PWM , which is the second phase angle Delay.
  • C is the PWM pulse output delay coefficient, the value range is (0-0.5).
  • C 0.25.
  • the delay ⁇ t2 in the process of modulation calculation and PWM pulse output can be shown as the following formula:
  • ⁇ t2 B ⁇ T PWM + C ⁇ T PWM ⁇ 0.75T PWM
  • the second sub-compensated phase angle is obtained, and the second sub-compensated phase angle corresponds to the modulation algorithm time delay.
  • the phase angle of the error is obtained, and the second sub-compensated phase angle corresponds to the modulation algorithm time delay.
  • the second sub-compensation phase angle ⁇ cmps2 can be expressed as follows:
  • ⁇ is the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor.
  • the third sub-compensation phase angle is the error phase angle corresponding to the resolver sampling and transmission delay.
  • Obtain the d-axis voltage and q-axis voltage corresponding to each preset angular velocity in the range of stable operating angular velocity and obtain the corresponding to each preset angular velocity according to the d-axis voltage and q-axis voltage corresponding to each preset angular velocity Error phase angle, and then establish a curve with the preset angular velocity as the abscissa and the error phase angle as the ordinate, and determine the slope corresponding to the curve as the error coefficient; further, obtain it according to the angular velocity of the rotor and the error coefficient corresponding to the angular velocity
  • the error phase angle which is the error phase angle caused by the resolver sampling and transmission delay.
  • the magnetic field generated by the rotor poles corresponds to the stator magnetic field as the d-axis, and the 90-degree counterclockwise rotation is the q-axis.
  • the sum of the first compensation phase angle, the second compensation phase angle, and the third compensation phase angle is the compensation phase angle of the direct drive permanent magnet synchronous motor.
  • the current position phase angle of the rotor of the direct-drive permanent magnet synchronous motor is obtained, then the actual position phase angle of the rotor is obtained according to the current position phase angle, the initial position phase angle of the rotor, and the compensation phase angle, and further, according to the actual position phase of the rotor Angle and the current modulation phase angle to obtain the current actual control phase angle, where the modulation phase angle is calculated by using a modulation algorithm and according to the given value of the d-axis voltage and the given value of the current q-axis voltage.
  • the actual position phase angle of the rotor is obtained according to the current position phase angle of the rotor and the initial position phase angle of the rotor, and further, the above-mentioned compensated phase angle is used to modify the rotor position angle of the direct drive permanent magnet synchronous motor offline to correct the correction
  • the phase angle after the actual position is taken as the actual phase angle of the rotor.
  • the difference between the actual position phase angle of the rotor and the current modulation phase angle is determined as the current actual control phase angle.
  • the modulation algorithm unit adopts a multi-mode PWM modulation strategy.
  • the allowable switching frequency of the inverter can be fully utilized, and on the other hand, a high DC voltage utilization rate can be ensured after entering the field weakening control zone.
  • the multi-mode PWM modulation strategy is mainly composed of asynchronous SPWM modulation, regular sampling synchronous SPWM modulation and square wave modulation.
  • Figure 34 is a schematic diagram of the multi-mode PWM modulation strategy.
  • the asynchronous modulation strategy is used in the low speed stage; when the speed increases, the regular sampling synchronous modulation with different carrier ratios and the intermediate 60-degree synchronous modulation strategy are used;
  • the high-speed phase uses square wave modulation.
  • the abscissa is the frequency of the modulated wave obtained by the modulation algorithm in this embodiment.
  • the ordinate is the PWM carrier frequency.
  • the specific low speed and high speed in the process of acquiring the current modulation phase angle are both the angular speed of the rotor, and the specific division rule may be similar to the division rule in the prior art.
  • S3106 Obtain the current expected control phase angle according to the current d-axis voltage given value and the current q-axis voltage given value.
  • the direct-drive permanent magnet synchronous motor in this embodiment uses a speed-based segmented vector control strategy to complete the current closed-loop control.
  • the control strategy includes: maximum torque current ratio (MTPA) control in the low-speed region and weak in the high-speed region Magnetic control.
  • MTPA maximum torque current ratio
  • MTPA control is used, that is, a control method that uses the reluctance torque generated by the salient pole effect of a permanent magnet synchronous motor to obtain a higher torque-current ratio. Due to the limited capacity of the system converter, when the permanent magnet synchronous motor runs in steady state, the terminal voltage and stator current will be idle, and the voltage and current limit values cannot be exceeded. To further widen the speed regulation range, the field weakening control is adopted at the rated speed In the above, the permanent magnet synchronous motor enters the field weakening state, and the purpose of speed-up of the field weakening can be achieved by controlling the excitation current.
  • control algorithm based on the above control strategy is used to calculate and obtain the current d-axis voltage given value and the current q-axis voltage given value, and further, obtain the current expected control according to the current d-axis voltage given value and the current q-axis voltage given value Phase angle.
  • ⁇ ctrl represents the expected control phase angle
  • a possible implementation method first, obtain the proportional deviation and integral deviation according to the current expected control phase angle and the current actual control phase angle, and then obtain the correction item of the current actual control phase angle according to the linear combination of the proportional deviation and the integral deviation, Further, this correction item is used to perform online correction on the current actual control phase angle.
  • k p and k i are correction terms
  • ⁇ ctrl is the current expected phase angle
  • ⁇ PWM is the current actual phase angle
  • f ⁇ is the fundamental frequency compensation term, which is a known quantity.
  • the online adjustment of the correction items enables the current actual control phase angle to track the expected control phase angle quickly and error-free, thereby realizing the online correction of the actual control phase angle.
  • the closed-loop PI control is adopted for the control of the phase angle, which can realize the control of the control phase angle accurately and without static error, thereby improving the control performance.
  • the current actual control phase angle Online correction is carried out to make the actual control phase angle consistent with the expected control phase angle, which improves the accuracy of the actual control phase angle, reduces the probability of direct drive permanent magnet synchronous motor operation failures, and thus improves the direct drive permanent magnet synchronous motor. Control performance of traction system.
  • FIG. 35 is a third schematic flowchart of a control method of a direct drive permanent magnet synchronous motor provided by the present invention. As shown in FIG. 35, on the basis of the embodiment implemented in FIG. 31, optionally, the following steps are included before step S3103:
  • the stable operating angular velocity range of the direct drive permanent magnet synchronous motor is first obtained, that is, the direct drive permanent magnet synchronous motor is not entered into the field weakening control stage,
  • the speed range of stable operation where the speed point corresponding to the constant voltage stage is reached and the voltage reaches the maximum value, which is the highest stable operation speed without entering the field weakening control stage.
  • a possible implementation manner is to obtain multiple first presets corresponding to the preset angular velocity interval when the rotor of the direct-drive permanent magnet synchronous motor is within the range of stable operating angular velocity according to the preset angular velocity interval Angular velocity
  • the d-axis current corresponding to each first preset angular velocity meets the preset error threshold, and the given values of the q-axis current and the q-axis current corresponding to each first preset angular velocity satisfy the preset error
  • the d-axis current corresponding to each first preset angular velocity is determined as the first d-axis current
  • the q-axis current corresponding to each first preset angular velocity is determined as the first q-axis current
  • the d-axis voltage corresponding to each first d-axis current is obtained according to each first d-axis current
  • the q-axis voltage corresponding to each first q-axis current is obtained according to each first q-axis current.
  • each first d-axis current and each first q-axis current acquired by the traction controller are the d-axis current and the q-axis current in the steady state of the direct-drive permanent magnet synchronous motor.
  • u d is the d-axis voltage corresponding to any first preset angular velocity
  • u q is the q-axis voltage corresponding to any first preset angular velocity
  • R s is the resistance of the rotor
  • L q is any first preset D-axis inductance corresponding to angular velocity
  • L d is the q-axis inductance corresponding to any first preset angular velocity
  • i d is the first d-axis current corresponding to the d-axis voltage
  • i q is the first q-axis current corresponding to the q-axis voltage
  • ⁇ f is the back electromotive force of the permanent magnet flux linkage.
  • FIG. 36A is a schematic diagram in which the theoretical coordinate system and the actual coordinate system completely coincide
  • FIG. 36B is a schematic diagram in which the actual coordinate system leads the theoretical coordinate system
  • FIG. 36C is a schematic diagram in which the actual coordinate system lags the theoretical coordinate system.
  • step S3103 can be implemented in the following manner:
  • the d-axis voltage corresponding to each first d-axis current and the q-axis voltage corresponding to each first q-axis current are used to obtain the transmission error phase angle corresponding to each first preset angular velocity.
  • the specific phase angle ⁇ ⁇ of transmission error can be obtained by the following formula:
  • the transmission error phase angle coefficient k can be obtained from the product of the transmission error phase angle coefficient and the current angular velocity of the rotor of the direct drive permanent magnet synchronous motor
  • the specific sub-compensation phase angle ⁇ cmps3 can be obtained as follows:
  • the stable operating angular velocity range of the direct-drive permanent magnet synchronous motor is obtained, and the stable operation is obtained according to the given value of the d-axis current and the given value of the q-axis current.
  • a plurality of first d-axis currents in the angular velocity range, a plurality of first q-axis currents, a d-axis voltage corresponding to each of the first d-axis currents, and a q-axis voltage corresponding to each of the first q-axis currents Obtain the transmission error phase angle corresponding to each first angular velocity according to the d-axis voltage corresponding to each first d-axis current and the q-axis voltage corresponding to each first q-axis current, and according to the transmission error corresponding to each first angular velocity
  • the phase angle, and the current angular velocity of the rotor of the direct-drive permanent magnet synchronous motor obtain the third sub-compensated phase angle.
  • a method for adhesion control of a direct-drive permanent magnet synchronous motor in the main circuit is also provided, so as to reduce the idling and sliding degrees in time, and effectively improve the adhesion utilization rate So that the traction of the locomotive can be stably exerted, reduce the abnormal load of the wheelset, and reduce the wheel scraping and peeling damage.
  • adhesion control is performed by at least four direct-drive permanent magnet synchronous motors on the electric locomotive; the at least four direct-drive permanent
  • the magnetic synchronous motor includes: a first motor, a second motor, a third motor, and a fourth motor.
  • six direct drive permanent magnet synchronous motors are provided on the motor locomotive, and two direct drive permanent magnet motor locomotive converters as shown in the foregoing embodiments are used
  • the main circuit separately controls six direct-drive permanent magnet synchronous motors.
  • the four direct drive permanent magnet synchronous motors involved in the calculation in the control method of this embodiment may be any four of the six direct drive permanent magnet synchronous motors of the electric locomotive, and the first motor and the second motor are provided on the electric locomotive.
  • the shaft motor on the first bogie, the third motor and the fourth motor are shaft motors provided on the second bogie of the electric locomotive.
  • FIG. 37 is a flowchart of an embodiment of the adhesion control method provided by the present invention.
  • the method provided in this embodiment can be applied to a direct drive permanent magnet traction system. As shown in FIG. 37, the method provided in this embodiment may include:
  • S3701 Collect the rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor to obtain the real-time torque of the first motor.
  • the first motor and the second motor are the axle motors of the first bogie and the third motor
  • the fourth motor is a shaft motor of the second bogie, and the first bogie is adjacent to the second bogie.
  • the four motors in this embodiment are located on adjacent bogies.
  • the operating conditions of the locomotive can be determined according to the real-time torque of the first motor.
  • the rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor may be collected according to a preset sampling period or a preset sampling frequency.
  • S3702 Determine the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors.
  • the smallest rotor frequency among the first motor, the second motor, the third motor, and the fourth motor is used as the rotor frequency reference.
  • the rotor frequency difference of the first electric machine is the difference between the rotor frequency of the first electric machine and the rotor frequency reference.
  • the differential value of the rotor frequency of the first motor in this embodiment may be the difference between the rotor frequency of the first motor at the current sampling time and the rotor frequency of the first motor at the previous sampling time divided by the sampling time interval.
  • the torque reduction amount can be determined according to the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor.
  • the torque reduction amount is used to indicate the amount of torque that the first motor needs to be unloaded.
  • the torque corresponding to the torque reduction amount of the first motor is unloaded to eliminate the idling phenomenon.
  • the adhesion control method provided in this embodiment collects the rotor frequencies of the first motor, the second motor, the third motor, and the fourth motor on adjacent bogies, and the real-time torque of the first motor, according to the collected Rotor frequency of multiple motors, determine the rotor frequency difference and rotor frequency differential value of the first motor, determine the torque reduction amount based on the rotor frequency difference, rotor frequency differential value and real-time torque of the first motor, and reduce the torque according to the torque To adjust the torque of the first motor.
  • the torque reduction is determined according to the rotor frequency for adhesion control, with low noise and strong resistance to external interference; according to the rotor frequency difference and rotor frequency differential value, it can quickly and accurately determine whether the locomotive is in the idling state, and reduce the idling and coasting degree in time, effectively Improve the adhesion utilization rate, make the traction of the locomotive stable, reduce the abnormal load of the wheel set, and reduce the wheel scraping and peeling damage.
  • the method provided in this embodiment may further include:
  • a sanding control signal is generated, and the sanding control signal is used to indicate whether to perform sanding operation.
  • Sanding can increase the adhesion coefficient between the wheels and rails, and reduce the idling and sliding of the locomotive. If it is determined according to the rotor frequency difference, the rotor frequency differential value and the real-time torque of the first motor that the idling coasting level of the locomotive satisfies the preset condition, the sanding operation is performed.
  • determining the amount of torque reduction according to the rotor frequency difference, rotor frequency differential value, and real-time torque of the first motor may include:
  • the rotor frequency difference of the first motor and the preset rotor frequency difference level rules determine the idling coasting level corresponding to the rotor frequency difference of the first motor, according to the idling coasting level corresponding to the rotor frequency difference of the first motor, and the first motor
  • the real-time torque determines the first torque reduction.
  • the preset rotor frequency differential level rule may include a mapping relationship between the rotor frequency difference and the idling coasting level. Different idling coasting levels correspond to different torque reduction coefficients. For example, the torque reduction coefficient corresponding to a higher idling coasting level may be set The bigger.
  • the first torque reduction amount may be equal to the real-time torque of the first motor multiplied by the torque reduction factor corresponding to the difference in the rotor frequency of the first motor.
  • the rotor frequency differential value of the first motor determines the idling coasting level corresponding to the rotor frequency differential value of the first motor, and according to the idling coasting level corresponding to the rotor frequency differential value of the first motor, As well as the real-time torque of the first motor, the second torque reduction amount is determined.
  • the preset grading rules of the rotor frequency differential value can include the mapping relationship between the rotor frequency differential value and the idling coasting level. Different idling coasting levels correspond to different torque reduction coefficients. For example, the torque corresponding to the higher idling coasting level can be set The greater the reduction factor.
  • the second torque reduction amount may be equal to the real-time torque of the first motor multiplied by the torque reduction coefficient corresponding to the rotor frequency differential value of the first motor.
  • the first torque reduction amount is determined as the torque reduction amount; if the first torque reduction amount is less than the second torque reduction amount, the second rotation is determined
  • the amount of torque reduction is the amount of torque reduction. That is, the larger of the first torque reduction amount and the second torque reduction amount is selected as the torque reduction amount, and the torque of the first motor is adjusted.
  • this embodiment describes in detail the process of adjusting the torque of the first motor according to the amount of torque reduction.
  • adjusting the torque of the first motor according to the torque reduction amount may include:
  • the torque value of the first motor is reduced from the first value to the second value, and the difference between the first value and the second value is the torque reduction amount.
  • the torque value of the first motor is gradually reduced from the first value to the second value according to the decreasing rate of the torque value of the first motor. That is, the unloading of the torque value of the first motor is from fast to slow, which is beneficial to the search for the best sticking point and avoids a sudden drop in torque.
  • the torque value of the first motor is kept unchanged at the second value.
  • the torque value of the first motor is increased from the second value to a preset percentage of the preset torque value, for example, it can be increased to 90% of the preset torque value.
  • the torque value of the first motor is increased to the preset torque value.
  • the recovery rate of the torque value of the first motor in the third preset time period is greater than the recovery rate of the torque value of the first motor in the fourth preset time period. That is to say, for the recovery of the torque value of the first motor, the segment recovery is adopted, and the recovery is performed first quickly and then slowly, which can effectively avoid the occurrence of idling coasting again.
  • the specific durations of the first preset time period, the second preset time period, the third preset time period, and the fourth preset time period in this embodiment can be set as needed, and this embodiment does not limit this.
  • the first preset time period, the second preset time period, the third preset time period, and the fourth preset time period constitute a torque adjustment period, and adjust the torque of the first motor when idling occurs.
  • 38 is a schematic diagram of an adhesion control process provided by an embodiment of the present invention. 38 is a schematic diagram of the process of adjusting the torque of the first motor by the adhesion control method provided by an embodiment of the present invention when idling occurs.
  • the T1, T2, T3 and T4 sub-tables represent the first preset time period, the second preset time period, the third preset time period and the fourth preset time period, and T1, T2, T3 and T4 constitutes a torque adjustment cycle.
  • the reference frequency curve of the locomotive represents the changing trend that the rotor frequency of the first motor should follow when the locomotive is in the traction mode, and the rotor frequency curve represents the actual rotor frequency of the first motor.
  • Stage T1 is the stage of torque unloading.
  • Point a is the moment when the locomotive is idling.
  • Figure 38 once the idling is detected, the torque is quickly unloaded immediately.
  • the unloading amount is from large to small, as shown in the figure.
  • the torque unloading curve shown in section ab in 38 can be fitted as an inverse proportional function curve, and then continue to unload with two small slopes, as shown in section bc and cd in Figure 2, where the unloading rate of section bc is greater than that of section cd Unloading rate until the torque unloading amount is equal to the determined torque reduction amount, that is, the torque difference between point a and point d is equal to the torque reduction amount.
  • the T2 stage is a stage where the torque is kept constant. When the torque unloading amount reaches the torque reduction amount, the locomotive does not run idle and maintains a low torque output, as shown in paragraphs d-e in FIG. 38.
  • the T3 phase is the first recovery phase of torque. After maintaining the low torque output for a period of T2, that is, after idling disappears for a period of T2, the torque is restored to 90% of the preset torque at a preset rate, as shown in FIG. 38 As shown in paragraph ef.
  • the T4 stage is the complete recovery stage of the torque, and the torque is restored to the preset torque, as shown in paragraph f-g in FIG. 38.
  • the lifting rate of the f-g torque is smaller than that of the e-f torque.
  • the preset torque may be the torque at the moment of idling, that is, the preset torque may be set equal to the torque at point a in the figure.
  • the torque unloading is from fast to slow, which is beneficial to the search for the best sticking point and avoids a sudden drop in torque.
  • segment recovery is adopted, which can effectively avoid the idling again. It is understandable that the process of gliding is similar and will not be repeated here.
  • determining the rotor frequency difference and the rotor frequency differential value of the first motor according to the collected rotor frequencies of the multiple motors may include:
  • the operating conditions of the locomotive may be determined according to the real-time torque of the first electric machine, and the operating conditions of the locomotive may include idle running conditions, traction operating conditions, and braking operating conditions.
  • the first torque threshold and the second torque threshold are set, where the first torque threshold is greater than zero and the second torque threshold is less than zero.
  • This embodiment is specific to the first torque threshold and the second torque threshold The value is not limited and can be set according to actual needs.
  • the locomotive If the real-time torque of the first motor is greater than or equal to the first torque threshold, the locomotive is in traction mode; if the real-time torque of the first motor is less than or equal to the second torque threshold, the locomotive is in braking mode; if the first If the real-time torque of the motor is greater than the second torque threshold and less than the first torque threshold, the locomotive is in an idle mode.
  • amplitude limiting filtering and low-pass filtering on the collected multiple rotor frequencies may include:
  • the rotor frequency compensation coefficient is determined and compensated for each motor, which improves the rotor frequency acquisition accuracy and thus the accuracy of adhesion control.
  • an embodiment of the present invention further provides an auxiliary converter control device, which is used to implement the auxiliary converter as S104 in the foregoing FIG. 1 embodiment. Control and related functions.
  • FIG. 39 is a schematic structural diagram 1 of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention.
  • the device includes: Digital Signal Processing (DSP) Chip 101 and Field Programmable Gate Array (FPGA) chip 102, DSP chip 101 is connected to FPGA chip 102 bus, of which DSP chip 101 is connected to FPGA chip 102 bus, of which DSP chip 101 is connected to FPGA chip 102 bus, of which
  • DSP Digital Signal Processing
  • FPGA Field Programmable Gate Array
  • the FPGA chip 102 is used to obtain the analog quantity and digital quantity of the auxiliary converter through the analog quantity sampling board 103 and the digital quantity sampling board 104, and perform logic operation processing on the analog quantity and digital quantity to obtain data after logic operation processing ;
  • the DSP chip 101 is used to perform control operation processing on the data after logic operation processing to obtain the pulse width;
  • the FPGA chip 102 is also used for pulse width modulation processing, to obtain the driving pulse sequence of the auxiliary converter.
  • the DSP chip 101 is a device that processes a large amount of information with digital signals. Its working principle is to receive an analog signal and convert it to a digital signal of 0 or 1, and then process the digital signal and use it in other system chips. The digital signal data is converted into the required data type in order to fulfill the corresponding requirements.
  • the DSP chip has the advantages of embeddable type, good stability and programmability, but at the same time because the DSP chip completes the work by serially executing instructions Therefore, the efficiency of DSP chips in data acquisition and data processing is low.
  • FPGA chip 102 refers to a chip that uses FPGA to implement functions.
  • An FPGA is a device that contains a reconfigurable gate array logic circuit matrix, which is connected in a certain way by configuring the internal circuits of FPGA, thereby creating software
  • FPGA uses special hardware for logic processing, so FPGA is faster in data collection and data processing.
  • FPGA is completed by executing instructions in parallel, so FPGA chip 102 is used for cooperation
  • the DSP chip 101 realizes the control function of the auxiliary converter, which can effectively make up for the shortcomings of the DSP chip and improve the working efficiency.
  • the FPGA chip 102 is used for data sampling to obtain the analog and digital quantities of the auxiliary converter.
  • the data sampling operation is performed by the analog sampling board 103 and the digital sampling board 104.
  • the sampling board is mainly used for To collect data, the analog sampling board 103 can collect the analog quantity of the auxiliary converter, and the digital sampling board 104 can acquire the digital quantity of the auxiliary converter. Among them, the analog quantity and the digital quantity are both in the auxiliary converter.
  • the analog quantity refers to the quantity of the variable continuously changing in a certain range, that is, the quantity that can always correspond to a value within the definition domain, such as the voltage value measured by the voltage sensor and the current sensor measurement
  • the obtained current value, etc. and the digital value means that the change is discontinuous in time, and always occurs at a series of discrete moments.
  • the digital value is scattered, and there is no intermediate value. For example, it can be a switching signal , The switch is turned on at a certain moment, and the switch is turned off at another moment, which is discontinuous in time. You can use "0" and "1" to represent the digital quantity
  • the logical operation processing can be, for example, AND operations on data, or, for example, AND operations on data.
  • the embodiment does not particularly limit the logical operation processing, as long as the processing related to the logical operation is called logical operation processing, the data after the logical operation processing is obtained, and the data after the logical operation processing is transferred to the DSP chip 101.
  • the data after the logic operation processing is sent to the DSP chip 101, and the DSP chip 101 performs control operation processing on the data.
  • the specific processing method is to process the data after the logic operation processing transmitted from the FPGA chip 102 according to the set control algorithm .
  • the control calculation process is the core of the auxiliary converter control function, which refers to specific data processing and related judgment according to the received data, and instructs the auxiliary converter what kind of operation to perform.
  • the control calculation process can be, for example, current and voltage control , Conversion, etc., there is no particular limitation on the control arithmetic processing here, the DSP chip 102 performs the control arithmetic processing to obtain the pulse width, and sends the obtained pulse width to the FPGA chip 102.
  • the FPGA chip 102 receives the pulse width transmitted by the DSP chip 101, and the FPGA chip 102 performs modulation arithmetic processing.
  • the so-called modulation arithmetic processing refers to using the relevant modulation algorithm to convert the initial signal amount into the required target signal amount.
  • the modulation algorithm is used to process the data after the control arithmetic processing in the modulation algorithm.
  • the modulation algorithm may be to use the modulation algorithm to convert the pulse width into a pulse sequence.
  • the embodiment of the present invention does not specifically limit the modulation algorithm.
  • the auxiliary change is generated.
  • the drive pulse sequence of the flow generator reaches the target component.
  • the target component can be, for example, the drive board 105, and then the target component begins to perform related operations, for example, the drive board 105 can drive the conduction and shutdown of the relevant pipeline to obtain the output needs.
  • the voltage of the auxiliary power converter realizes the control function of the auxiliary converter. Part of the logic operation processing and related modulation operation processing functions are implemented by the FPGA chip 102, which can save more resources for the DSP chip 101 to complete other functions and better realize the allocation of chip resources, thereby improving the work of the auxiliary converter effectiveness.
  • the control device of the auxiliary converter of the direct-drive permanent magnet electric locomotive includes: a DSP chip 101 and an FPGA chip 102, the DSP chip 101 is connected to the FPGA chip 102 bus, wherein the FPGA chip 102 is used for sampling by analog quantity Board and digital sampling boards acquire the analog and digital quantities of the auxiliary converter, and perform logic operation processing on the analog and digital quantities to obtain the data after the logic operation processing; DSP chip 101 is used to process the logic operation The data of the data is subjected to control calculation processing to obtain a pulse width; the FPGA chip 102 is also used to perform modulation calculation processing according to the pulse width to obtain a driving pulse sequence of the auxiliary converter.
  • the control function of the auxiliary converter is realized, and the DSP chip 101 and the FPGA chip 102 are used to complete the control of the auxiliary converter, which makes up for the defects of low sampling speed and unreasonable allocation of chip resources when the separate DSP chip implements the control function .
  • control arithmetic processing of the DSP chip 101 and the modulation arithmetic processing of the FPGA chip 102 include multiple processing operations.
  • the control arithmetic processing of the DSP chip 101 and the modulation arithmetic processing of the FPGA chip will be described in detail in conjunction with FIG. 40 below. Introduction.
  • FIG. 40 is a schematic diagram of the processing of the control device of the auxiliary converter of the direct-drive permanent magnet electric locomotive provided by the embodiment of the present invention.
  • the control calculation processing includes: Park transformation processing 202, voltage and current double-loop decoupling control Processing 203, IPark transformation process 204, and zero-sequence voltage injection process 205;
  • modulation operation process includes: pulse generation process 206;
  • DSP chip is specifically used to sequentially perform Park transformation process 202 on the data after the logic operation process 201, voltage and current double closed loop Decoupling control process 203, IPark transformation process 204, and zero sequence voltage injection process 205 to obtain data after zero sequence voltage injection process 205;
  • FPGA chip is specifically used to perform pulse generation process 206 on the data after zero sequence voltage injection process 205, The data after the pulse generation processing 205 is obtained.
  • the DSP chip first receives the data after the FPGA chip performs the logical operation process 201.
  • the voltage data is first subjected to the Park transformation process 201, and the Park transformation process 201 is Refers to the transformation of three-phase AC A, B, and C phases to a rotating dq coordinate system, where three-phase AC refers to the power of three AC circuits with the same frequency, equal potential amplitude, and phase difference of 120 degrees.
  • the system currently produces and distributes three-phase alternating current in our country, so in this embodiment, the voltage data is also three-phase alternating current.
  • the three phases of the three-phase alternating current are phase A, phase B, and phase C, respectively.
  • Park transformation 202 is required to convert the three-phase AC A , B, and C phases are transformed into a rotating dq coordinate system. As shown in FIG. 2, three-phase alternating currents U A , U B , and U C are subjected to Park transformation processing 202 to obtain U d and U q .
  • the dq coordinate system is obtained in the following way: because the three-phase alternating current is three-phase symmetrical, then the two-phase alternating current can be used to achieve the same magnetic field effect as the three-phase alternating current, and the three-phase alternating current is projected to the two-phase interval of 90 On the two-phase alternating current of degree, the transformation from three-phase static alternating current to two-phase static alternating current is realized. If the two-phase static alternating current rotates at a certain angular velocity, then the rotating dq coordinate system is obtained for a long time. The three-phase AC power is converted into a DC quantity in the dq coordinate system. The control of the DC quantity is simple and there is no coupling between them. Therefore, the control of the DC quantity in the dq coordinate system realizes the decoupling control of the three-phase AC power.
  • the DC quantity in the dq coordinate system is obtained, and then the DC quantity is subjected to a double closed-loop decoupling control process 203, wherein the double closed-loop control refers to the voltage outer loop and current inner loop control, which is through closed loop Voltage and closed-loop current to control, so as to keep the current and voltage constant, to achieve double closed-loop control in the rotating dp coordinate system, which is the voltage and current double-loop decoupling control process 203 in Figure 2, as shown in Figure 40, park to obtain direct current U d U q and the dq coordinate system after the conversion process 202, through the voltage and current double closed loop control process 203 decoupling and U qout U dout obtained, wherein, U dout and U qout is a constant value, thereby avoiding The voltage fluctuation caused by the load switching of the auxiliary converter causes the control to fail, and the three-phase AC power in the stationary coordinate system is converted into a DC quantity in the dq coordinate
  • the voltage in the rotating dq coordinate system needs to be transformed into three-phase alternating current in the stationary coordinate system through the IPark transformation process 204, so that the subsequent normal power supply, as shown in Figure 2 , U dout and U qout are transformed into U p , V p and W p through the IPark transform process 204 to obtain the output voltage of the IPark transform process 204.
  • the zero-sequence voltage injection process 205 is performed on the output voltage of the IPark conversion process 204 in the DSP chip, and the zero is calculated according to the output voltages U p , V p and W p of the control operation process first Sequence voltage, in which, when the three-phase alternating current is analyzed using the symmetrical component method, the three components of the same size and the same phase are the zero sequence voltage.
  • the zero-sequence voltage is combined with the output voltages U p , V p and W p of the control arithmetic processing to obtain the output voltage of the modulation arithmetic processing and generate pulses, as shown in FIG.
  • the control arithmetic processing The output voltages U p , V p and W p are subjected to zero-sequence voltage injection processing 205 to obtain the output voltage of the control arithmetic processing, that is, pulse width modulation (Pulse Width Modulation, PWM) pulse width variables U out , V out and W out , After obtaining the output voltage of the control operation process, that is, the PWM pulse width variable, the output voltage of the control operation process is sent to the FPGA chip.
  • PWM Pulse Width Modulation
  • the FPGA chip receives the pulse width variable of the output voltage PWM of the control operation process transmitted by the DSP chip, and performs the modulation operation process. Specifically, the modulation operation process performs the pulse generation process 206. After the pulse generation process 206, the effective voltage value can reach the input The size of the DC voltage can effectively improve the utilization rate of the input DC voltage.
  • the control operation processing provided by the embodiment of the present invention includes: Park transformation processing 202, voltage and current double-closed loop decoupling control processing 203, IPark transformation processing 204, and zero sequence voltage injection processing 205; modulation operation processing includes: pulse generation processing 206; DSP
  • the chip is specifically used to sequentially perform the Park transformation process 202, the voltage and current double-closed-loop decoupling control process 203, the IPark transformation process 204, and the zero sequence voltage injection process 205 to obtain the data after the zero sequence voltage injection process.
  • the FPGA chip is specifically used to perform pulse generation processing 206 on the data after the zero-sequence voltage injection processing to obtain the data after the pulse generation processing. It can effectively realize the voltage decoupling control, and the current and voltage are constant during the control process, thereby avoiding the control failure caused by the voltage fluctuation.
  • the zero sequence voltage injection process can improve the utilization rate of the input DC voltage.
  • the DSP chip and the FPGA chip are not only used in logic operation processing, control operation processing, and modulation operation processing, but also in failure detection.
  • the failure detection of the DSP chip and the FPGA chip will be described in detail in conjunction with FIG. 3 below.
  • FIG. 41 is a second schematic structural diagram of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention.
  • the FPGA chip is also used to collect the operating data of the auxiliary converter according to the collected data. Determine if the component in the auxiliary converter has failed; if the FPGA chip determines that the component in the auxiliary converter has failed, send the failure information to the DSP chip; if the FPGA chip determines that the component in the auxiliary converter has not failed, Then the operation data is sent to the DSP chip; the DSP chip is also used to determine that the component has failed according to the failure information; the DSP chip also uses the operation data to determine whether the component in the auxiliary converter has failed.
  • the FPGA chip is also specifically used to determine that the component is faulty when it is determined that the operating data is greater than the first preset threshold according to the collected operating data of the auxiliary converter.
  • the DSP chip is also specifically used to determine that the component has failed when it is determined that the operating data is greater than the second preset threshold and the fault duration is greater than the preset duration, where the first preset threshold is greater than the second preset threshold.
  • a data acquisition module 301 and a fault judgment module 302 may be provided in the FPGA chip; a detection module 303, a counting module 304, and a flag generation module 305 are provided in the DSP chip.
  • the specific implementation process of the above-mentioned FPGA chip and DSP chip is specifically described by means of modules.
  • the data collection module 301 is used to collect the operation data of the auxiliary converter
  • the fault judgment module 302 is used to determine whether a component of the auxiliary converter fails, specifically, according to the collected operation of the auxiliary converter
  • the data is compared with the first preset threshold. If the operating data is greater than the first preset threshold, it is determined that the corresponding component is faulty in the FPGA chip, corresponding fault handling is performed in the FPGA chip, and the fault information is sent to the DSP chip
  • the fault information contains the signal of the detected fault, and the DSP chip receives the fault information to determine that the component of the auxiliary converter is faulty.
  • the FPGA chip determines that the component in the auxiliary converter has not failed, then this time The FPGA chip does not send the fault information to the DSP chip, but only sends the operating data to the DSP chip. It should be noted that the FPGA chip will transmit the related analog quantity and data amount to the DSP chip regardless of whether the FPGA chip determines a failure.
  • the operation data sent by the FPGA chip received by the DSP chip first enters the detection module 303.
  • the detection module 303 is used to determine whether a component in the auxiliary converter has failed according to the operation data, specifically, determine whether the operation data is greater than The second preset threshold, if the operating data is greater than the second preset threshold, it is further determined whether the fault duration is greater than the preset duration, if the operating data is greater than the second preset threshold, and the fault duration is greater than the preset duration, then It is determined in the DSP chip that the component corresponding to the auxiliary converter has failed.
  • the first preset threshold in the fault judgment module 302 is greater than the second preset threshold in the detection module 303, because data collection is completed in the FPGA, and the data processing speed of the FPGA chip is fast, while the data processing speed of the DSP chip is relatively high Slow, so the DSP chip cannot receive all the data transmitted by the FPGA chip within the set data transmission time.
  • the purpose of setting the first preset threshold to be greater than the second preset threshold is that if a more serious failure occurs, the FPGA The chip can quickly detect and make corresponding fault operation processing, and the protection response time of the DSP chip is longer, the protection threshold is low and the duration detection is set, which can ensure the accuracy of fault detection and improve the efficiency of fault handling.
  • the FPGA chip can quickly detect major faults and perform logic protection actions.
  • the fault detection in the DSP chip can ensure the accuracy of fault detection, thereby ensuring the safe operation of the auxiliary converter and improving the reliability of the system .
  • the DSP chip is also used to determine the number of component failures after multiple restarts of the auxiliary converter
  • the DSP chip is also used to determine whether the number of failures is greater than the preset number of times. If so, the failure is determined to be a permanent failure, and if not, the failure is determined to be a warning failure.
  • the counting module 304 in the DSP chip is used to perform multiple restart operations after determining that the component has failed, and use the counter provided in the DSP chip to record the number of restarts to determine the number of component failures, and then determine the failure Whether the number of times is greater than the preset number of times, if the number of failures is greater than the preset number of times, it is determined that the failure of the component is a permanent failure, that is to say, the failure cannot be resolved even after multiple restarts, if the number of failures is less than or the preset number It is determined that the fault is a warning fault, that is, the fault can be repaired after multiple restarts, and the warning fault is repaired after multiple restart operations, saving resources for the auxiliary converter to perform other operations.
  • the DSP chip is also used to set the fault flag bit corresponding to the component as the permanent fault flag bit when determining that the fault of the component is a permanent fault.
  • the DSP chip is also used to splice multiple fault flags and send the spliced fault flags to the host computer.
  • the flag bit generation module 305 is used to determine the fault flag bit corresponding to the component as the permanent fault flag bit when the fault of the component is a permanent fault.
  • the fault flag bit corresponding to the component is set as the warning fault flag bit, the processing method for the permanent fault flag bit and the warning fault flag bit is different, and the DSP chip will splice all the fault flag bits, and the spliced fault
  • the flag is sent to the host computer, where the host computer refers to a computer that can directly issue control commands.
  • the host computer is connected to the DSP chip, and the fault feedback module 306 of the embodiment of the present invention is used to receive After the failure flag bit sent by the DSP chip, corresponding failure processing is performed.
  • the DSP chip When the DSP chip needs to send the fault flag to the upper computer, it needs to communicate with the upper computer.
  • the communication method between the DSP chip and the upper computer will be described below in conjunction with FIG. 42.
  • FIG. 42 is a schematic structural diagram 3 of a control device for an auxiliary converter of a direct-drive permanent magnet electric locomotive provided by an embodiment of the present invention. As shown in FIG. 42, the host computer 404 and the DSP chip 403 communicate via a CAN bus.
  • the DSP chip 403 needs to communicate with the locomotive control unit 401.
  • the communication between the DSP chip 403 and the locomotive control unit 401 adopts the multifunction vehicle bus (Multifunction Vehicle, MVB) communication, so in the control device of the auxiliary converter Equipped with MVB board 402, and the communication between DSP chip 403 and MVB board 402 adopts Controller Area Network (Controller Area Network, CAN) bus communication
  • CAN bus is a serial communication that effectively supports distributed control or real-time control
  • the bus, and the communication between the DSP chip 403 and the upper computer 404 also uses CAN bus communication, which can effectively realize the communication and improve the communication efficiency.
  • the control device of the auxiliary converter of the direct drive permanent magnet electric locomotive provided by the embodiment of the present invention effectively communicates with the control device and the locomotive control unit and the host computer by setting the host computer and the DSP chip to communicate through the CAN bus, which improves Communication efficiency.
  • the data After performing the related operations on the DSP chip and the FPGA chip described above, for example, after determining that the auxiliary converter has failed, the data needs to be stored.
  • the data storage will be described below in conjunction with specific embodiments.
  • the control device of the auxiliary converter of the direct-drive permanent magnet electric locomotive provided by the embodiment of the present invention further includes: a flash memory; a random access memory (random access memory (RAM) space) is provided in the DSP chip; the flash memory is connected to the DSP chip;
  • a flash memory random access memory (random access memory (RAM) space) is provided in the DSP chip; the flash memory is connected to the DSP chip;
  • RAM random access memory
  • the DSP chip is also used to store the data processed by the logic operation in the RAM space;
  • the DSP chip is also used to determine the failure of the component and store the data in the RAM space to the flash memory.
  • the flash memory is a non-volatile memory, and data will not be lost even if the power is turned off.
  • a flash memory is provided in the control device of the auxiliary converter, and the flash memory is connected to the DSP chip, and is also provided inside the DSP chip.
  • Array space after the related logic operation processing is performed on the data collected by the FPGA chip in the working process of the auxiliary converter, the data after the logic operation processing is stored in the array space set in the DSP chip.
  • the DSP chip After the DSP chip determines that the fault occurs, it calls related functions to store the data in the array space into the flash memory.
  • the flash memory used for data storage has a larger storage space, which can meet the needs of higher frequency data collection and data storage, and In the later maintenance process, it is sufficient to read the data at the time of failure from the flash memory to improve the working efficiency of the auxiliary converter.

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Abstract

一种电力机车用大功率直驱永磁电传动系统，包括：第一四象限整流器、第二四象限整流器、中间直流回路、第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器，第一四象限整流器和第二四象限整流器均连接电力机车的主变压器和中间直流回路，中间直流回路分别连接第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器。该传动系统能够对使用大功率直驱永磁同步电机的电力机车中的直驱永磁同步电机进行控制，填补了直驱永磁同步电机在电力机车中应用的空白。

Description

一种电力机车用大功率直驱永磁电传动系统 技术领域
本发明涉及列车控制技术领域，尤其涉及一种电力机车用大功率直驱永磁电传动系统。
背景技术
电力机车的牵引变流器是电力机车的重要组成部分，用于将牵引供电网的电能转换为供给牵引电动机电能，以达到控制牵引电动机的转速，操纵机车速度的目的。牵引变流器的主电路的设计是牵引变流器的工作性能的主要因素之一，直接影响电力机车的重量、尺寸、效率以及相关技术经济指标。
现有技术中的电力机车普遍采用交流异步电机加齿轮箱的驱动方式，为了提升电力机车的效率，减少损耗，本发明采用了直驱永磁同步电机应用到电力机车中。直驱永磁同步电机一方面充分利用了永磁同步电机高效、低损耗、高功率密度和启动转矩大的优点，一方面将齿轮箱去掉，采用直接驱动的方式将永磁同步电机和机车轮对结合在一起，减少了质量以及齿轮箱带来的损耗，更进一步的提高了电力机车的整体效率。
当前电力机车中的牵引变流器以及现有的电传动系统并没有针对直驱永磁同步电机进行设计的，因此并没有一种电传动系统能够直接应用于使用直驱永磁同步电机的电力机车中。而如何设计使用直驱永磁同步电机的电力机车中的电力机车用大功率直驱永磁电传动系统是目前亟待解决的技术问题。
发明内容
本发明提供一种电力机车用大功率直驱永磁电传动系统，能够对使用直驱永磁同步电机的电力机车中的直驱永磁同步电机进行控制，丰富了电力机车用大功率直驱永磁电传动系统的功能，填补了直驱永磁同步 电机在电力机车中应用的空白。
本发明提供一种电力机车用大功率直驱永磁电传动系统，用于控制使用直驱永磁同步电机的电力机车，所述电力机车包括三台直驱永磁同步电机；所述电力机车用大功率直驱永磁电传动系统包括：第一四象限整流器、第二四象限整流器、中间直流回路、第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器，所述第一四象限整流器和所述第二四象限整流器均连接所述电力机车的主变压器和所述中间直流回路，所述中间直流回路分别连接所述第一逆变模块、第二逆变模块、第三逆变模块和所述辅助变流器；所述电力机车用大功率直驱永磁电传动系统用于：
通过所述第一四象限整流器和所述第二四象限整流器将所述主变压器的交流电转换为直流电后输出至所述中间直流回路；
通过所述中间直流回路将接收到的直流电分别输出至所述第一逆变模块、第二逆变模块、第三逆变模块和所述辅助变流器；
通过所述第一逆变模块、第二逆变模块和第三逆变模块将接收到的直流电转换为三相交流电后分别输出至所述三台直驱永磁同步电机；
通过所述辅助变流器将接收到的直流电转换为三相交流电后输出至所述电力机车的辅助负载。
可选地，在本发明一实施例中，所述通过所述第一四象限整流器和所述第二四象限整流器将所述主变压器的交流电转换为直流电后输出至所述中间直流回路，包括：
对输入四象限整流器的交流电流进行采样，得到采样周期内的交流电流，所述交流电流包括正半周期的电流值和负半周期的电流值；其中，根据预设采样频率，对输入所述四象限整流器的交流电流进行采样，得到多个采样点，将得到的多个采样点绘制成曲线，得到一个正弦或者余弦曲线；所述预设采样频率为IGBT通断频率的N倍，所述N≥2；
获取正半周期的电流值的第一和值与负半周期的电流值的第二和值，并根据所述第一和值和所述第二和值，获取电流偏置值；其中，将正半周期的多个采样点的值进行加和得到第一和值P，再将负半周期的多 个采样点的值进行加和得到第二和值N，P值与N值的绝对值进行做差计算，所得到的差值为Q；
将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值；其中，直流偏置值Q与零输入至第一PI控制器，第一PI控制器根据直流偏置值Q与零构成控制偏差，将偏差的比例和积分通过线性组合构成控制量，对交流电流进行控制，消除交流电流的直流偏置。控制量即为第一输出值；
根据所述第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，所述PR控制器用于对所述交流电流进行无静差控制，使所述交流电流的周期和相位与电网电压相同；其中，将交流电流输入到PR控制器，保证交流电流的相位和和周期与电网电压相同后，得到稳定的输出交流电流，即为第二输出值；
根据所述脉冲宽度调制符号控制所述四象限整流器中的绝缘栅双极型晶体管IGBT的通断。
在一种可能的设计中，对所述输入四象限整流器的交流电流进行采样，得到采样周期内的交流电流之前，还包括：
获取所述四象限整流器的直流母线电压与指令电压的第二差值；
将所述第二差值输入至第二PI控制器，使得所述第二PI控制器输出的第三输出值与锁相环输出值相乘，得到与电网电压同相位的交流电流，所述锁相环用于控制所述交流电流的周期与相位和电网电压的周期与相位保持一致。
在一种可能的设计中，对输入所述四象限整流器的交流电流进行采样，得到采样周期内的交流电流，包括：
根据预设采样频率对输入四象限整流器的交流电流进行采样，得到采样电流，所述预设采样频率为所述IGBT通断频率的两倍；
根据所述锁相环确定的电网电压相位和所述采样电流，得到采样周期内的交流电流。
在一种可能的设计中，根据所述锁相环确定的电网电压相位和所述采样电流，得到采样周期内的交流电流之前，所述方法还包括：
通过第一带通滤波器和第二带通滤波器对所述采样电流进行滤波， 得到滤波后的采样电流；其中，所述第一带通滤波器用于获取交流电流的主频信号，所述第二带通滤波器用于滤除干扰谐波。
在一种可能的设计中，将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值之前，所述方法还包括：
判断所述第一差值的绝对值是否大于所述电流环宽的绝对值，得到的判断结果为是。
在一种可能的设计中，根据所述第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，包括：
对所述第一输出值和所述第二输出值进行求和，得到第三和值，所述第一输出值为电流变量，所述第二输出值为电流值；
根据所述第三和值和单极倍频脉冲调制方式，得到所述脉冲宽度调制符号。
在本发明一实施例中，所述电力机车用大功率直驱永磁电传动系统还包括：第一斩波模块和第二斩波模块，所述第一斩波模块连接所述第一四象限整流器和所述中间直流回路，所述第二斩波模块连接所述第二四象限整流器和所述中间直流回路；
所述电力机车用大功率直驱永磁电传动系统还用于：
通过第一斩波模块和第二斩波模块分别将所述第一四象限整流器和所述第二四象限整流器输出的直流电进行斩波处理后输出至所述中间直流回路；
具体地，对于所述第一斩波模块和所述第二斩波模块中的任一斩波模块，所述控制方法还包括：
对中间直流母线电压进行周期性检测，所述中间直流母线电压为所述电力机车上直流母线上的电压；
当检测到的中间直流母线电压值大于斩波上限阈值时，采用P调节器对所述中间直流母线电压进行调节，直至检测到的所述中间直流母线电压值小于斩波下限阈值，所述斩波上限阈值大于所述斩波下限阈值；其中，所述P调节器的原理为：控制斩波管在检测周期的特定时间比例内处于开通状态。
可选地，所述采用P调节器对所述中间直流母线电压进行调节，包括：
采用所述P调节器，确定目标检测周期内的斩波占空比；所述目标检测周期包括：从检测到的中间直流母线电压值大于斩波上限阈值，到检测到的中间直流母线电压值小于斩波下限阈值之间的经历的检测周期；
根据所述斩波占空比，确定斩波管在目标检测周期内的开通时间；
根据所述开通时间，控制所述斩波管的开通或关断，以使所述中间直流母线电压值下降至小于所述斩波下限阈值。
可选地，上述方法，还包括：
当检测到中间直流母线电压值小于斩波下限阈值时，控制斩波管关断。
可选地，所述采用所述P调节器，确定目标检测周期内的斩波占空比之前，还包括：
根据以下公式确定目标参数；
Err＝U1斩波下限阈值
其中，Err表示目标参数，U1表示目标检测周期内检测到的中间直流母线电压值；
相应的，所述采用所述P调节器，确定目标检测周期内的斩波占空比，包括：
获取所述P调节器对应的控制系数；
根据所述控制系数和所述目标参数，确定所述斩波占空比。
可选地，所述获取所述P调节器的控制系数，包括：
根据如下公式确定所述控制系数；
Kp_chp＝1/(直流母线电压过压保护值阈值-斩波下限阈值)
其中，Kp_chp表示控制系数。
可选地，所述根据所述控制系数和所述目标参数，确定所述斩波占空比，包括：
根据如下公式确定所述斩波占空比；
C_duty＝Err*Kp_chp
其中，C_duty表示斩波占空比，Err表示目标参数，Kp_chp表示控 制系数。
可选地，所述根据所述斩波占空比，确定斩波管在目标检测周期内的开通时间之前，还包括：
对所述斩波占空比进行防错处理。
可选地，所述对所述斩波占空比进行防错处理，包括：
若所述斩波占空比的值大于1，则将所述斩波占空比的值设为1；
若所述斩波占空比的值小于0，则将所述斩波占空比的值设为0。
可选地，在本实施例一种可能的实现方式中，控制方法还包括：
确定待控制直驱永磁同步电机的转速；
根据所述转速与第一映射关系确定第一控制策略，所述第一映射关系包括至少一个转速的范围和至少一个控制策略的一一对应关系；
根据所述第一控制策略确定所述待控制直驱永磁同步电机的预期控制相角。
可选地，所述第一映射关系包括：
额定转速以下的转速与MTPA控制策略的对应关系；
额定转速以上的转速与弱磁控制策略的对应关系。
可选地，所述MTPA控制策略包括：根据转矩电流曲线确定q轴电流给定和d轴电流给定；
计算所述q轴电流给定与q轴实际电流的第一差值和所述d轴电流给定与d轴实际电流的第二差值；
通过第一PI控制器根据所述第一差值得到d轴电压给定、通过第二PI控制器根据所述第二差值得到q轴电压给定；
计算所述q轴电压给定与q轴前馈电压之和得到实际q轴电压给定、计算所述d轴电压给定与d轴前馈电压之和得到实际d轴电压给定；其中，所述前馈电压可通过如下前馈解耦的闭环传递函数矩阵计算：
Figure PCTCN2018117085-appb-000001
其中，所述前馈解耦的闭环传递函数通过如下前馈解耦的电压计算方程得到：
Figure PCTCN2018117085-appb-000002
Figure PCTCN2018117085-appb-000003
可选地，所述弱磁控制策略包括：通过PI控制器根据电压极限值与前馈电压幅值之差计算给定弱磁状态下d轴电流变化量；
通过给定弱磁状态下d轴电流变化量和d轴电流给定之和得到弱磁调节后的d轴电流给定；
根据所述d轴电流给定和所述转矩公式计算弱磁调节后的q轴电流给定；
通过PI控制器根据所述q轴电流给定与q轴实际电流之差得到功角β；
通过如下公式计算实际q轴电压给定和实际d轴电压给定；
U d＝U s cosβ
U q＝U s cosβ
其中，Us为电压极限值，Ud为实际d轴电压给定，Uq为实际q轴电压给定。
可选地，在本实施例一种可能的实现方式中，还包括：
当控制策略从所述MTPA控制策略切换至所述弱磁控制策略时，将切换瞬间MTPA控制策略中的电压矢量角度作为所述弱磁控制策略中初始功角β；
当控制策略从所述弱磁控制策略切换至所述MTPA控制策略时，通过切换瞬间弱磁控制策略中的最后一拍功角β通过公式
Figure PCTCN2018117085-appb-000004
计算出MTPA控制策略中的实际q轴电压给定和实际d轴电压给定。
可选地，在本实施例一种可能的实现方式中，控制方法还包括：
获取待调制直驱永磁同步电机的调制波的频率；
根据所述调制波的频率所在范围与第二映射关系确定第一调制策略，所述第二映射关系包括至少一个调制波的频率范围和至少一个调制策略的一一对应关系；
根据所述第一调制策略确定所述直驱永磁同步电机的PWM载波频率。
可选地，所述第二映射关系包括：
调制波的频率为低速阶段时对应异步调制策略；
调制波的频率大于低速阶段低于高速阶段时对应中间60度同步调制策略；
调制波的频率为高速阶段时对应方波调制策略。
可选地，在本实施例一种可能的实现方式中，还包括：
向待检测永磁同步电机的定子绕组注入高频电压信号，获取三相定子绕组电流；
根据所述三相定子绕组电流获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流；
根据所述d轴目标电流和所述q轴目标电流获取转子的初始位置角，其中，所述初始位置角为根据所述永磁同步电机的磁极极性进行补偿后的初始位置角。
进一步地，所述根据所述d轴目标电流和所述q轴目标电流获取转子的初始位置角，包括：
根据所述q轴目标电流获取转子的第一初始位置角；
根据所述d轴目标电流获取转子的磁极补偿角；
根据所述第一初始位置角以及所述磁极补偿角，获取所述转子的初始位置角。
进一步地，所述根据所述q轴目标电流获取转子的第一初始位置角，包括：
对所述q轴目标电流进行低通滤波处理，获取误差输入信号；
根据所述误差输入信号，获取所述第一初始位置角。
进一步地，所述对所述q轴目标电流进行低通滤波处理，获取误差输入信号，包括：
采用调制信号对所述q轴目标电流进行调制，获取调制后的q轴目标电流；
对所述调制后的q轴目标电流进行低通滤波处理，获取所述误差输入信号。
进一步地，所述根据所述误差输入信号，获取所述第一初始位置角，包括：
根据所述输入误差信号获取所述误差输入信号的比例偏差和积分偏差；
根据所述比例偏差和所述积分偏差的线性组合，获取所述第一初始位置角。
进一步地，所述根据所述d轴目标电流获取转子的磁极补偿角，包括：
向所述永磁同步电机注入多个电压幅值相等、角度不同的电压脉冲信号，获取每个所述电压脉冲信号的响应电流；
根据多个所述响应电流，确定所述转子的磁极补偿角。
进一步地，所述根据多个所述响应电流，确定所述转子的磁极补偿角，包括：
当注入的所述电压脉冲信号的角度与所述第一初始位置角之差满足预设误差范围，所述电压脉冲信号的响应电流的幅值大于第一值，则确定所述转子的磁极补偿角为0，所述第一值为多个所述响应电流的幅值的最大值；
当注入的所述电压脉冲信号的角度与所述第一初始位置角之差满足预设误差范围，所述电压脉冲信号的响应电流的幅值小于第二值，则确定所述转子的磁极补偿角为π，所述第二值为多个所述响应电流的幅值的最小值。
可选地，在本实施例一种可能的实现方式中，所述高频电压信号为：
Figure PCTCN2018117085-appb-000005
其中，U mh为高频电压信号的幅值，ω h为高频电压信号的角频率，t为注入高频电压信号的时间；
所述根据三相定子绕组电流获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流，通过如下公式计算：
Figure PCTCN2018117085-appb-000006
其中，L为平均电感L＝(L d+L q)/2，△L为半差电感△L＝(L d-L q)/2；
所述对q轴目标电流进行低通滤波处理，获取误差输入信号，通过如下公式计算：
Figure PCTCN2018117085-appb-000007
其中，LPF表示低通滤波；当转子位置估计误差足够小，极限等效线性化后该误差输入信号为：
Figure PCTCN2018117085-appb-000008
所述获取第一初始位置角，通过以下公式计算：
Figure PCTCN2018117085-appb-000009
其中，s表示拉普拉斯算子，k p为比例项系数，k i为积分项系数。
可选地，本实施例提供的电力机车用大功率直驱永磁电传动系统还包括：根据控制中断周期、调制载波周期，以及所述直驱永磁同步电机的转子当前角速度，获取所述直驱永磁同步电机的转子的补偿相角；
根据所述补偿相角，获取当前实际控制相角；
根据当前d轴电压给定值和当前q轴电压给定值，获取当前预期控制相角；
根据所述当前预期控制相角与所述当前实际控制相角的比例偏差和积分偏差，对所述当前实际控制相角进行在线修正。
进一步地，所述根据控制中断周期、调制载波周期，以及所述直驱永磁同步电机的转子的当前角速度，获取所述直驱永磁同步电机的转子的补偿相角，包括：
根据所述控制中断周期和所述直驱永磁同步电机的转子的当前角速度，获取第一子补偿相角；
根据所述调制载波周期和所述直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角；
根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角；
根据所述第一子补偿相角、所述第二子补偿相角和所述第三子补偿相角，获取所述直驱永磁同步电机的补偿相角。
进一步地，所述根据所述控制中断周期和所述直驱永磁同步电机的 转子的当前角速度，获取第一子补偿相角，包括：
根据所述控制中断周期，获取第一子补偿相角对应的第一相角时延；
根据所述第一相角时延和所述直驱永磁同步电机的转子的当前角速度，获取所述第一子补偿相角。
进一步地，所述根据所述调制载波周期和所述直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角，包括：
根据所述调制载波周期，获取调制输出对应的第二相角时延；
根据调制算法的调制中断周期，获取调制计算对应的第三相角时延；
根据所述第二相角时延、所述第三相角时延和所述直驱永磁同步电机的转子的当前角速度，获取所述第二子补偿相角。
进一步地，所述根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角之前，还包括：
根据所述直驱永磁同步电机的矢量控制策略，获取所述直驱永磁同步电机的稳定运行角速度范围；
根据d轴电流给定值和q轴电流给定值，获取所述稳定运行角速度范围内的多个第一d轴电流、多个第一q轴电流、每个所述第一d轴电流对应的d轴电压以及每个所述第一q轴电流对应的q轴电压。
进一步地，所述根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角，包括：
根据每个所述第一d轴电流对应的d轴电压以及每个所述第一q轴电流对应的q轴电压，获取每个第一角速度对应的传输误差相角；
根据每个所述第一角速度对应的传输误差相角、所述直驱永磁同步电机的转子的当前角速度以及所述转子的初始位置相角，获取所述第三子补偿相角。
进一步地，所述根据所述补偿相角，获取当前实际控制相角，包括：
获取所述直驱永磁同步电机的转子的当前位置相角；
根据所述当前位置相角、所述转子的初始位置相角以及所述补偿相 角，获取所述转子的实际位置相角；
根据所述转子的实际位置相角以及调制相角，获取当前实际控制相角，其中，所述调制相角为根据d轴电压给定值和当前q轴电压给定值经过调制算法计算得到。
进一步地，所述根据所述当前预期控制相角与所述当前实际控制相角的比例偏差和积分偏差，对所述当前实际控制相角进行在线修正，包括：
根据所述当前预期控制相角与所述当前实际控制相角获取所述比例偏差、所述积分偏差；
根据所述比例偏差以及所述积分偏差的线性组合，获取当前实际控制相角的修正项；
根据所述修正项对所述当前实际控制相角进行在线修正。
可选地，在本实施例一种可能的实现方式中，
所述获取第一子补偿相角，通过如下公式计算：
θ cmps1＝Δ t1·ω
其中，ω为直驱永磁同步电机的转子的当前角速度，Δ t1为第一相角时延，第一相角时延Δ t1通过如下公式计算：
Δ t1＝A·T ctrl≈0.5T ctrl。
其中，T ctrl为控制算法的一个控制中断周期；
所述获取第二子补偿相角，通过如下公式计算：
θ cmps2＝Δ t2·ω
其中，ω为直驱永磁同步电机的转子的当前角速度，Δ t2为PWM脉冲输出过程中的时延，PWM脉冲输出过程中的时延Δ t2通过如下公式计算：
Δ t2＝B·T PWM+C·T PWM≈0.75T PWM
其中，T PWM为PWM的调制载波周期，B为调制算法中断时延系数，C为PWM脉冲输出时延系数；
所述获取当前预期控制相角，通过如下公式计算：
Figure PCTCN2018117085-appb-000010
其中，θ ctrl表示预期控制相角，
Figure PCTCN2018117085-appb-000011
表示q轴电压给定值，
Figure PCTCN2018117085-appb-000012
表示d轴电压给定值；
所述对当前实际控制相角进行在线修正，通过如下公式计算：
Figure PCTCN2018117085-appb-000013
其中，k p和k i为修正项，θ ctrl为当前预期相角，θ PWM为当前实际相角，f Δ为基波频率补偿项；
所述获取所述直驱永磁同步电机的稳定运行角速度范围，通过如下公式计算：
Figure PCTCN2018117085-appb-000014
其中，u d为任一第一预设角速度对应的d轴电压，u q为任一第一预设角速度对应的q轴电压，R s为转子的电阻，L q为任一第一预设角速度对应的d轴电感，L d为任一第一预设角速度对应的q轴电感，i d为d轴电压对应的第一d轴电流，i q为q轴电压对应的第一q轴电流，ψ f为永磁体磁链的反电势；
所述获取传输误差相角θ Δ，通过如下公式计算：
θ Δ＝tan -1(u d/u q)
所述获取第三子补偿相角θ cmps3，通过如下公式计算：
θ cmps3＝k·ω。
在本发明一实施例中，所述电力机车还包括：至少四个直驱永磁同步电机；所述至少四个直驱永磁同步电机包括：第一电机、第二电机、第三电机和第四电机；
所述电力机车用大功率直驱永磁电传动系统还用于：
采集第一电机、第二电机、第三电机和第四电机的转子频率，获取所述第一电机的实时转矩，所述第一电机和所述第二电机为第一转向架的轴电机，所述第三电机和所述第四电机为第二转向架的轴电机，所述第一转向架与所述第二转向架相邻；
根据所采集的多个电机的转子频率，确定所述第一电机的转子频率差和转子频率微分值；
根据所述第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量；
根据所述转矩削减量对所述第一电机的转矩进行调整。
在一种可能的实现方式中，所述方法还包括：
根据第一电机的转子频率差、转子频率微分值和实时转矩，生成撒砂控制信号，撒砂控制信号用于指示是否进行撒砂操作。
在一种可能的实现方式中，根据第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量，包括：
根据第一电机的转子频率差以及预设的转子频率差分级规则，确定第一电机的转子频率差对应的空转滑行等级；
根据第一电机的转子频率差对应的空转滑行等级，以及第一电机的实时转矩，确定第一转矩削减量；
根据第一电机的转子频率微分值以及预设的转子频率微分值分级规则，确定第一电机的转子频率微分值对应的空转滑行等级；
根据第一电机的转子频率微分值对应的空转滑行等级，以及第一电机的实时转矩，确定第二转矩削减量；
若第一转矩削减量大于等于第二转矩削减量，则确定第一转矩削减量为转矩削减量；
若第一转矩削减量小于第二转矩削减量，则确定第二转矩削减量为转矩削减量。
在一种可能的实现方式中，根据转矩削减量对第一电机的转矩进行调整，包括：
在第一预设时间段内，将第一电机的转矩值由第一值降低至第二值，第一值与第二值的差值为转矩削减量；
在第二预设时间段内，保持第一电机的转矩值为第二值不变；
在第三预设时间段内，将第一电机的转矩值由第二值提高至预设转矩值的预设百分比；
在第四预设时间段内，将第一电机的转矩值提高至预设转矩值；
其中，第一电机的转矩值在第三预设时间段内的恢复速率，大于第一电机的转矩值在第四预设时间段内的恢复速率。
在一种可能的实现方式中，在第一预设时间段内，将第一电机的转矩值由第一值降低至第二值，包括：
在第一预设时间段内，根据第一电机的转矩值的降低速率逐渐减 小，将第一电机的转矩值由第一值降低至第二值。
在一种可能的实现方式中，根据所采集的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值，包括：
对所采集的多个电机的转子频率进行限幅滤波和低通滤波处理；
根据限幅滤波和低通滤波处理后的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值。
在一种可能的实现方式中，若机车处于惰行工况，则对所采集的多个转子频率进行限幅滤波和低通滤波处理，包括：
获取第一电机的电流值；
根据第一电机的电流值和每个电机的转子频率，确定每个电机的转子频率补偿系数；
根据每个电机的转子频率补偿系数对每个电机的转子频率进行补偿；
对补偿后的多个电机的转子频率进行限幅滤波和低通滤波处理。
本发明一实施例中，电力机车用大功率直驱永磁电传动系统还包括一种辅助变流器的控制设备，包括：数字信号处理DSP芯片以及现场可编程门阵列FPGA芯片，所述DSP芯片与所述FPGA芯片总线连接，其中，电力机车用大功率直驱永磁电传动系统还用于：
通过所述FPGA芯片用于通过模拟量采样板卡和数字量采样板卡获取辅助变流器的模拟量和数字量，并对所述模拟量和数字量进行逻辑运算处理，得到逻辑运算处理后的数据；
通过所述DSP芯片用于对所述逻辑运算处理后的数据进行控制运算处理，得到脉冲宽度；
通过所述FPGA芯片还用于根据所述脉冲宽度进行调制运算处理，得到所述辅助变流器的驱动脉冲序列。
在一种可能的设计中，所述控制运算处理包括：Park变换处理，电压、电流双闭环解耦控制处理、IPark变换处理以及零序电压注入处理；
所述调制运算处理包括：脉冲生成处理；
所述DSP芯片具体用于对所述逻辑运算处理后的数据依次进行Park变换处理，电压、电流双闭环解耦控制处理、IPark变换处理以及零序电 压注入处理，得到零序电压注入处理后的数据；
所述FPGA芯片具体用于对所述零序电压注入处理后的数据进行脉冲生成处理，得到脉冲生成处理后的数据。
在一种可能的设计中，所述FPGA芯片还用于根据采集到的所述辅助变流器的运行数据，确定所述辅助变流器中的部件是否发生故障；
若所述FPGA芯片确定所述辅助变流器中的部件发生故障，则将故障信息发送给所述DSP芯片；
若所述FPGA芯片确定所述辅助变流器中的部件未发生故障，则将所述运行数据发送至所述DSP芯片；
所述DSP芯片还用于根据所述故障信息确定所述部件发生故障；
所述DSP芯片还用根据所述运行数据判断所述辅助变流器内的部件是否发生故障。
在一种可能的设计中，所述FPGA芯片还具体用于根据采集到的所述辅助变流器的运行数据，在确定运行数据大于第一预设阈值时，确定所述部件发生故障。
在一种可能的设计中，所述DSP芯片还具体用于在确定所述运行数据大于第二预设阈值，并且故障持续时间大于预设时长时，确定所述部件发生故障，其中，所述第一预设阈值大于所述第二预设阈值。
在一种可能的设计中，所述DSP芯片还用于在所述辅助变流器多次重启后，确定所述部件的故障次数；
所述DSP芯片还用于判断所述故障次数是否大于预设次数，若是，则确定所述故障为永久故障，若否，则确定所述故障为警告故障。
在一种可能的设计中，所述DSP芯片还用于在确定所述部件的故障为永久故障时，将所述部件对应的故障标志位设置为永久故障标志位。
在一种可能的设计中，还包括：上位机，所述上位机与所述DSP芯片连接；
所述DSP芯片还用于将多个故障标志位进行拼接处理，并将拼接处理后的故障标志位发送给所述上位机。
在一种可能的设计中，所述上位机与所述DSP芯片通过CAN总线通信。
在一种可能的设计中，还包括：闪存；所述DSP芯片内设置有RAM空间；所述闪存与所述DSP芯片连接；
所述DSP芯片还用于将所述逻辑运算处理后的数据存储至所述RAM空间；
所述DSP芯片还用在确定所述部件发生故障，将所述RAM空间中的数据存储至所述闪存中。
综上，本实施例提供的电力机车用大功率直驱永磁电传动系统中，依次通过四象限整流器、中间直流回路和逆变模块，将主变压器的交流电通过“交-直-交”的流程最终转换为直驱永磁同步电机可用的三相交流电。从而能够对使用大功率直驱永磁同步电机的电力机车中的直驱永磁同步电机进行控制，丰富了电力机车用大功率直驱永磁电传动系统的功能，填补了直驱永磁同步电机的电力机车中应用的空白。
附图说明
为了更清楚地说明本发明实施例或现有技术中的技术方案，下面将对实施例或现有技术描述中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动性的前提下，还可以根据这些附图获得其他的附图。
图1为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图；
图2为本发明电力机车用大功率直驱永磁电传动系统一实施例的流程示意图；
图3为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图；
图4为本发明实施例提供的四象限整流器的局部电路图；
图5为本发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法的流程示意图；
图6为本发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法的流程示意图；
图7为本实施例提供的本发明实施例提供的电力机车用大功率直驱永磁 电传动系统电流偏置调节方法的流程示意图
图8为本发明提供的斩波控制方法的实施例一的流程示意图；
图9为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图；
图10为本发明提供的斩波控制方法的实施例二的流程示意图；
图11为本发明提供的斩波控制方法的实施例二的另一流程示意图；
图12为本发明提供的斩波控制方法的实施例三的流程示意图；
图13为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的控制方法流程示意图；
图14为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的控制系统的结构示意图；
图15为本发明MTPA控制的系统结构示意图；
图16为本发明前端解耦控制的系统结构示意图；
图17为本发明弱磁控制的系统结构示意图；
图18为本发明全速度范围内MTPA控制和弱磁控制的轨迹示意图；
图19为本发明MTPA控制和弱磁控制切换控制示意图；
图20为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的调制方法流程示意图；
图21为本发明提供的中间60°调制方式下调制角度与调制比的关系；
图22为本发明提供的基于中间60°调制的全速度范围调制策略示意图；
图23为本发明提供的永磁同步电机转子初始位置角检测方法实施例一的流程示意图；
图24为本发明提供的两相同步旋转坐标系、两相静止坐标系以及预期两相同步旋转坐标系关系示意图；
图25为本发明提供的永磁同步电机转子初始位置角检测方法实施例二的流程示意图；
图26为本发明提供的永磁同步电机转子初始位置角检测方法实施例三的流程示意图；
图27为永磁同步电机运行过程中多个通道的信号变化示意图；
图28为响应电流变化规律示意图；
图29为本发明提供的电力机车用大功率直驱永磁电传动系统对应的直驱永磁同步电机的控制系统的结构示意图；
图30为本发明提供的直驱永磁同步电机的控制方法的流程示意图一；
图31为本发明提供的直驱永磁同步电机的控制方法的流程示意图二；
图32为本发明提供的控制算法的中断周期示意图；
图33为本发明提供的调制算法的中断周期示意图；
图34为多模式PWM调制策略的示意图；
图35为本发明提供的直驱永磁同步电机的控制方法的流程示意图三；
图36A为理论坐标系与实际坐标系完全重合的示意图；
图36B为实际坐标系超前理论坐标系的示意图；
图36C为实际坐标系滞后理论坐标系的示意图；
图37为本发明提供的粘着控制方法一实施例的流程图；
图38为本发明一实施例提供的粘着控制过程的示意图；
图39为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图一；
图40为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的处理示意图；
图41为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图二；
图42为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图三。
具体实施方式
下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。
本发明的说明书和权利要求书及上述附图中的术语“第一”、“第二”、“第三”、“第四”等(如果存在)是用于区别类似的对象，而不必用于 描述特定的顺序或先后次序。应该理解这样使用的数据在适当情况下可以互换，以便这里描述的本发明的实施例例如能够以除了在这里图示或描述的那些以外的顺序实施。此外，术语“包括”和“具有”以及他们的任何变形，意图在于覆盖不排他的包含，例如，包含了一系列步骤或单元的过程、方法、系统、产品或设备不必限于清楚地列出的那些步骤或单元，而是可包括没有清楚地列出的或对于这些过程、方法、产品或设备固有的其它步骤或单元。
下面以具体地实施例对本发明的技术方案进行详细说明。下面这几个具体的实施例可以相互结合，对于相同或相似的概念或过程可能在某些实施例不再赘述。
图1为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图。如图1所示，本实施例提供的电力机车用大功率直驱永磁电传动系统包括：第一四象限整流器、第二四象限整流器、中间直流回路、第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器，第一四象限整流器和第二四象限整流器均连接电力机车的主变压器和中间直流回路，中间直流回路分别连接第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器。
具体地，本实施例提供的电力机车用大功率直驱永磁电传动系统用于使用直驱永磁同步电机的电力机，用于控制电力机车上的至少一个直驱永磁同步电机。需要说明的是，本发明各实施例中以直驱永磁同步电机数量为三个作为示例，本实施例提供的主电路还可用于控制具有少于或者多于三个直驱永磁同步电机的电力机车，原理相同且仅为数量上的增减。
图2为本发明电力机车用大功率直驱永磁电传动系统一实施例的流程示意图。下面结合图2对如图1所示的电力机车用大功率直驱永磁电传动系统的使用方法进行说明，其中，该电力机车用大功率直驱永磁电传动系统的使用方法包括：
S101：通过第一四象限整流器和第二四象限整流器将主变压器的交流电转换为直流电后输出至中间直流回路。
其中，本实施例的方法用于控制如图1所示的电力机车用大功率直驱永磁电传动系统将变流器的交流电转换为直驱永磁同步电机所能够使用 的三相变频变压交流电。则在S101中，可以控制接入主变压器的第一四象限整流器和第二四象限整流器，将主变压器的交流电转换为直流电后输入中间直流回路。可选地，第一四象限整流器和第二四象限整流器的输入端可视为整个主电路的输入端，第一四象限整流器和第二四象限整流器的输入端可通过与主变压器的次边牵引绕组连接的方式获取主变压器所提供的交流电。可选地，在本发明相同或相似的主电路替代方案中，四象限整流器的数目不作具体限定，对于并列设置的每个四象限整流器，每个四象限整流器独立工作，均用于接收主变压器提供的交流电并转换为直流电后向中间直流回路输出。
S102：通过中间直流回路将接收到的直流电分别输出至第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器。
当中间直流回路接收到第一四象限整流器和第二四象限整流器发送的直流电后，在S102中控制直流回路将直流电分别向其所连接的第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器输出。其中，第一四象限整流器和第二四象限整流器共用中间直流回路，中间直流回路将收到的多路直流电经过汇总传输后，分别向第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器输出。
S103：通过第一逆变模块、第二逆变模块和第三逆变模块将接收到的直流电转换为三相交流电后分别输出至三台直驱永磁同步电机。
则在S103中，当接收到中间回路发送的直流电后，需要控制第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器分别向其所连接的负载供电。其中，逆变模块与直驱永磁同步电机一一对应，辅助变流器与辅助负载相对应。例如在图1所示的主电路的实施例中，电力机车包括了三个直驱永磁同步电机，因此其主电路也需要相应设置三个逆变模块。如图中的连接关系，第一逆变模块连接直驱永磁同步电机1，并将接收到的直流电转换为直驱永磁同步电机1可用的交流电后向其输出、第二逆变模块连接直驱永磁同步电机2，并将接收到的直流电转换为直驱永磁同步电机2可用的交流电后向其输出、第三逆变模块连接直驱永磁同步电机3，并将接收到的直流电转换为直驱永磁同步电机3可用的交流电后向其输出。每个逆变模块均通过向其连接的直驱永磁同步电机发送的交流电驱动直 驱永磁同步电机，从而实现电力机车中的三个直驱永磁同步电机的驱动控制。
S104：通过辅助变流器将接收到的直流电转换为三相交流电后输出至电力机车的辅助负载。
同时，本实施例提供的电力机车用大功率直驱永磁电传动系统中，辅助变流器也可以连接中间直流回路，并可以在S104中控制辅助变流器将从中间直流回路接收到的直流电，转换为电力机车中辅助负载可用的交流电后，向辅助负载输出。可选地，这里所述的辅助负载至少包括但不限于以下的一项或多项：电力机车的照明系统、通信系统和空调系统。
综上，本实施例提供的电力机车用大功率直驱永磁电传动系统中，依次通过四象限整流器、中间直流回路和逆变模块，将主变压器的交流电通过“交-直-交”的流程最终转换为直驱永磁同步电机可用的三相交流电。从而能够对使用直驱永磁同步电机的电力机车中的直驱永磁同步电机进行控制，丰富了电力机车用大功率直驱永磁电传动系统的功能，填补了直驱永磁同步电机的电力机车中对该类型电机的电力机车用大功率直驱永磁电传动系统的空白。
进一步地，图3为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图。如图3所示的实施例在图1所示的基础上，提供的一种电力机车用大功率直驱永磁电传动系统具体的电路设计以及连接方式，用以说明本发明后续各实施例中的电力机车用大功率直驱永磁电传动系统。
可选地，在本发明控制方法的一种具体实现方式中，提供一种S101中对于四象限整流器的控制方式，以消除在四象限整流器控制过程中电流偏置的影响。
具体地，图4为本发明实施例提供的四象限整流器的局部电路图，如图4所示的四象限整流器可以是如图1和图3中的第一四象限整流器，也可以是如图1和图3中的第二四象限整流器。本实施例提供的每个四象限整流器的工作方式以及原理相同，下面以一个四象限整流器进行具体说明。如图所示，g1、g2、g3和g4为四象限整流器的IGBT器件，g1、g2、 g3和g4协同工作，实现四象限整流器将交流电压转换成直流电压的作用。但是在现有技术中，当四象限整流器因器件、控制等因素出现电压偏置时，四象限整流器将不稳定，IGBT器件偏离其额定工作区，会在变压器上产生较大的直流偏置，基于该问题，本发明电力机车用大功率直驱永磁电传动系统的一实施例中，在S101提供一种电力机车用大功率直驱永磁电传动系统电流偏置调节方法，该方法在不改变图1和图3硬件结构的基础上能够解决直流偏置的问题。下面结合图5进行详细说明。
图5为本发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法的流程示意图，如图5所示，该方法包括：
S501、对输入四象限整流器的交流电流进行采样，得到采样周期内的交流电流，所述交流电流包括正半周期的电流值和负半周期的电流值。
具体地，根据预设采样频率，对输入四象限整流器的交流电流进行采样，得到多个采样点，将得到的多个采样点绘制成曲线，得到一个正弦或者余弦曲线。预设采样频率可以为IGBT通断频率的两倍甚至数倍或者其他，只要能根据预设采样频率采样得到完整的正弦或者余弦曲线即可，在此对预设采样频率不做特别限制。例如，在本实施例中，预设采样频率可以为IGBT通断频率的两倍，再将根据预设采样频率得到的多个采样点绘制成的正弦或者余弦曲线，根据相位分为正半周期和负半周期，例如正弦曲线的正半周期为0到π，负半周为π到2π，则正半周期的多个采样点的值即为交流电流正半周期的值，负半周期的多个采样点的值即为交流电流负半周期的值。
S502、获取正半周期的电流值的第一和值与负半周期的电流值的第二和值，并根据所述第一和值和所述第二和值，获取电流偏置值。
具体地，将正半周期的多个采样点的值进行加和得到第一和值P，再将负半周期的多个采样点的值进行加和得到第二和值N，P值与N值的绝对值进行做差计算，所得到的差值为Q。如果Q值为0，认为P值和N值的绝对值也完全相等，正弦曲线或者余弦曲线的正半周期和负半周期完全对称，交流电流没有直流偏置。若Q值不为0，则认为P值和N值的绝对值不相等，则正弦曲线或者余弦曲线的正半周期和负半周期不对称，交流电流存在直流偏置，Q值即为直流偏置值。
S503、将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值。
具体地，直流偏置值Q与零输入至第一PI控制器，第一PI控制器根据直流偏置值Q与零构成控制偏差，将偏差的比例和积分通过线性组合构成控制量，对交流电流进行控制，消除交流电流的直流偏置。控制量即为第一输出值。
S504、根据所述第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，所述PR控制器用于对所述交流电流的无静差控制，使所述交流电流的周期和相位与电网电压相同。
具体地，将交流电流输入到PR控制器，保证交流电流的相位和和周期与电网电压相同后，得到稳定的输出交流电流，即为第二输出值。再将第一输出值和第二输出值进行求和，得到第三和值。即第一PI控制器得到的控制量调节稳定的输出交流电流，从而抑制交流电流的直流偏置。再将第三和值用单极倍频脉冲调制方式进行调制，得到脉冲宽度调制符号。
S505、根据所述脉冲宽度调制符号控制所述四象限整流器中的绝缘栅双极型晶体管IGBT的通断。
具体地，结合图4所示，脉冲宽度调制符号作为四象限整流器中的绝缘栅双极型晶体管IGBT g1、g2、g3和g4的输入，来控制双极型晶体管IGBT的通断。
因此，在本实施例中，为电力机车用大功率直驱永磁电传动系统提供了一种电力机车用大功率直驱永磁电传动系统电流偏置调节方法，对输入四象限整流器的交流电流进行采样，得到采样周期内的交流电流，该交流电流包括正半周期的电流值和负半周期的电流值；获取正半周期的电流值的第一和值与负半周期的电流值的第二和值，并根据第一和值和第二和值，获取电流偏置值；将电流偏置值与零的第一差值输入至第一PI控制器，获取第一PI控制器输出的第一输出值；根据第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，PR控制器用于对交流电流进行无静差控制，使交流电流的周期和相位与电网电压相同；根据脉冲宽度调制符号控制四象限整流器中的绝缘栅双极型晶体管IGBT的通断。通过第一PI控制器输出的第一输出值来调节第二输出值，得到第三和值， 从而抑制交流电流的直流偏置，将该第三和值用单极倍频脉冲调制方式进行调制，得到脉冲宽度调制符号控制IGBT的工作，避免了IGBT器件偏离其额定工作区，从而有效的对变压器侧电流偏置进行根本抑制和消除，进而消除电流偏置对四象限整流器控制的影响。
图6为本发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法的流程示意图，图7为本实施例提供的本发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法的流程示意图，如图7所示，Udc为直流母线电压，陷波器主要是滤除直流母线电压Udc上的波动值，Udc*为指令电压，i为输入四象限整流器的交流电流，Us为输入四象限整流器的交流电流的电压，结合图7，本实施例在图5实施例的基础上，对本实施例的具体实现过程进行了详细说明。如图6所示，该方法包括：
S601、根据预设采样频率对输入四象限整流器的交流电流进行采样，得到采样电流，所述预设采样频率为所述IGBT通断频率的两倍。
本实施例提供的S601与图5实施例中的S501类似，本实施例此处不再赘述。
S602、通过第一带通滤波器和第二带通滤波器对所述采样电流进行滤波，得到滤波后的采样电流；其中，所述第一带通滤波器用于获取交流电流的主频信号，所述第二带通滤波器用于滤除干扰谐波。
具体地，考虑到不同地域交流电流主频存在的差异，第一带通滤波器的通带频率设置在40Hz-60Hz之间，例如在本实施例中，第一个带通滤波器通带频率为45-55Hz，可选地，当交流电流的主频为50Hz时，将该第一带通滤波器的通带频率设置为50Hz，用于获取交流电流的主频信号。同样的，在本实施例中，四象限整流器的开关频率为f，即IGBT的通断频率为f，第二个带通滤波器通带频率为2f/(50±5)Hz，第二带通滤波器用于滤除高次谐波干扰。第一带通滤波器和第二带通滤波器即为图5中的滤波器。
S603、获取所述四象限整流器的直流母线电压与指令电压的第二差值，将所述第二差值输入至第二PI控制器，使得所述第二PI控制器输出的第三输出值与锁相环的输出值相乘，所述锁相环用于得到电网电压相位，从而得到与所述电网电压同周期与相位的交流电流。
具体地，直流母线电压Udc与指令电压Udc*输入至第二PI控制器，第 二PI控制器根据直流母线电压Udc与指令电压Udc*偏差，将偏差的比例和积分通过线性组合构成控制量，控制量即为第二PI控制器输出的第三输出值。再将第二PI控制器输出的第三输出值与锁相环输出相乘，得到与电网电压同相位的交流电流。锁相环即图5中的PLL，该锁相环PLL用于控制交流电流i的周期与相位和电网电压的周期与相位保持一致。根据锁相环所控制的相位计算出电网电压的相位。S603中的第二PI控制器即为图7中的第二PI。
S604、根据所述锁相环确定的电网电压相位和所述采样电流，得到采样周期内的交流电流，所述交流电流包括正半周期的电流值和负半周期的电流值。
具体地，根据锁相环PLL所控制的相位计算出电网电压的相位，确定交流电流i的相位，也就确定了采样电流的相位，根据相位将采样电流分为正半周期和负半周期，例如正弦曲线的正半周期为0到π，负半周为π到2π，则正半周期的多个采样点的值即为交流电流i正半周期的值，负半周期的多个采样点的值即为交流电流i负半周期的值。S604即为图7中的直流偏置提取计算。
S605、获取正半周期的电流值的第一和值与负半周期的电流值的第二和值，并根据所述第一和值和所述第二和值，获取电流偏置值。
本实施例提供的S605与图5实施例中的S502类似，S605也为图7中的直流偏置提取计算，本实施例此处不再赘述。
S606、判断所述第一差值的绝对值是否大于所述电流环宽的绝对值，得到的判断结果为是。
具体地，为避免采样误差造成第一差值Q存在误差，将Q值大小和滞环环宽进行计算，滞环环宽可以为±5A，也可以为任意其他值，只要能避免成第一差值Q存在误差即可。例如，在本实施例中，滞环环宽为±5A；第一差值Q的绝对值大于5A，得到的判断结果为是，即交流存在直流偏置。具体地，第一差值Q大于5A，交流电流存在正直流偏置，第一差值Q小于-5A，交流电流存在负直流偏置。
S607、将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值。
本实施例提供的S607与图5实施例中的S503类似，S607中的第一PI 控制器即为图7中的第一PI，本实施例此处不再赘述。
S608、对所述第一输出值和所述PR控制输出的第二输出值进行求和，得到第三和值，所述第一输出值为电流变量，所述第二输出值为电流值；根据所述第三和值和单极倍频脉冲调制方式，得到所述脉冲宽度调制符号。
本实施例提供的S608与图5实施例中的S504类似，S608中的PR控制器即为图7中一PR，本实施例此处不再赘述。
S609、根据所述脉冲宽度调制符号控制所述四象限整流器中的绝缘栅双极型晶体管IGBT的通断。
本实施例提供的S609与图5实施例中的S505类似，同时与图7脉冲调制类似，本实施例此处不再赘述。
本发明实施例提供的电力机车用大功率直驱永磁电传动系统，将交流电流进行采样，得到采样电流，再将直流母线电压和指令电压的第二差值输入到第二PI控制器，得到第二PI控制器输出的第三输出值，第三输出值用于对交流电流进行调整。再将第三输出值与到锁相环输出值相乘后，根据锁相环计算出的电网电压相位，确定交流电流相位，进而确定采样电流的相位，再将采样电流分为正半周期和负半周期，计算出正半周期的电流值和负半周期的电流值，再将正半周期的电流值和负半周期的电流值的第一差值输入到第一PI控制器，通过第一PI控制器输出的第一输出值来调节PR控制器输出的第二输出值，得到第三和值，从而抑制交流电流的直流偏置，将该第三和值用单极倍频脉冲调制方式进行调制，得到脉冲宽度调制符号控制IGBT的工作，避免了IGBT器件偏离其额定工作区，从而有效的对变压器侧电流偏置进行根本抑制和消除，进而消除电流偏置对四象限整流器控制的影响。
进一步地，发明实施例提供的电力机车用大功率直驱永磁电传动系统电流偏置调节方法，提高了直流偏置抑制的响应速度，同时采用软件控制算法来解决直流偏置，省去了硬件电路设计，解决了其他直流偏置抑制方法不适用于电网电压频率宽频变化的问题。
可选地，在本发明控制方法的一种具体实现方式中，提供一种S102中对于中间直流回路的控制方式，具体涉及对于中间直流回路的斩波控制方法，以减小在电力机车用大功率直驱永磁电传动系统中对中间直流母线电压的冲 击。下面结合附图8和附图9对本实施例提供的中间直流回路的斩波控制方法进行说明。
具体地，图8为本发明提供的斩波控制方法的实施例一的流程示意图，如图8所示，本实施例提供的斩波控制方法，包括：
S801、对中间直流母线电压进行周期性检测，所述中间直流母线电压为所述交直交电力传动机车上直流母线上的电压。
图9为本发明电力机车用大功率直驱永磁电传动系统一实施例的结构示意图。如图9所示的电力机车用大功率直驱永磁电传动系统是在如图1基础上一种可能的连接方式。图9所示的电力机车用大功率直驱永磁电传动系统包含四象限整流模块1和四象限整流模块2，斩波模块1和斩波模块2，接地检测模块，逆变模块1、逆变模块2和逆变模块3，以及辅助模块。
其中，四象限整流模块1由g1、g3、g2、g4、g5、g7、g6和g8八个开关管组成，四象限整流模块2和四象限整流模块1结构相同。斩波模块1包括斩波开关管g9、斩波电流传感器A2、反向二极管D1和斩波电阻R5，斩波模块2和斩波模块1结构相同。接地检测模块包括电阻R3和R4，且R3阻值等于R4，电阻R3和R4串联在直流回路的两端组成了接地电阻检测回路。逆变模块1包括g10、g11、g12、g13、g14、g15六个开关管组成的三相逆变电路，逆变模块2、逆变模块3和逆变模块1结构相同。K2为电机隔离接触器，M为直驱永磁电动机，C1和C3为直流侧支撑电容，R2为慢放电阻，U1为直流母线电压传感器。辅助模块包括g16、g17、g18、g19、g20和g21六个开关管组成的三相逆变电路和一个辅助滤波柜组成。其中，在图9所示主电路拓扑图中，本实施例提及的中间直流母线电压指的是U1所测电压。
S802、当检测到的中间直流母线电压值大于斩波上限阈值时，采用P调节器对所述中间直流母线电压进行调节；直至检测到的所述中间直流母线电压值小于斩波下限阈值，所述斩波上限阈值大于所述斩波下限阈值。
其中，P调节器的原理为：控制斩波管在检测周期的特定时间比例内处于开通状态。该特定时间比例和检测到的中间直流母线电压值有关，检测到的中间直流母线电压值越大时，该时间比例越大。
由于，中间直流母线电压值从大于斩波上限阈值下降至小于斩波下限阈值的若干检测周期内，斩波管并不是始终处于开通状态，和现有技术相比， 减小了对中间直流母线电压的冲击。
需要说明的是，在采用P调节器对所述中间直流母线电压进行调节后，当检测到中间直流母线电压值小于斩波下限阈值时，直接控制斩波管关断。
本实施例提供的斩波控制方法，应用于交直交电力传动机车，对中间直流母线电压进行周期性检测，当检测到的中间直流母线电压值大于斩波上限阈值时，采用P调节器对所述中间直流母线电压进行调节；直至检测到的所述中间直流母线电压值小于斩波下限阈值，减小了对中间直流母线电压的冲击。
图10为本发明提供的斩波控制方法的实施例二的流程示意图。本实施例是进一步对上述实施例中S802的可实现方式的描述，如图10所示，S802包括：
S1001、采用所述P调节器，确定目标检测周期内的斩波占空比。
其中，所述目标检测周期包括：从检测到的中间直流母线电压值大于斩波上限阈值，到检测到的中间直流母线电压值小于斩波下限阈值之间的经历的检测周期。
举例来说，假设检测周期为1min，若在当前检测周期(1min)内检测到的中间直流母线电压值大于斩波上限阈值，则开始使用P调节器对中间直流母线电压进行调节，若经过调节在距离当前检测周期的第五个检测周期内检测到中间直流母线电压值小于斩波下限阈值，则当前的1min、第二个1min、第三个1min、第四个1min为目标检测周期。
其中，斩波占空比指的是：一个检测周期内，斩波管开通的时间占检测周期的比例。
可选地，参见图11所示，上述确定目标检测周期内的斩波占空比的可实现的方式为：
首先，确定目标参数，具体为：
S2011、在根据以下公式确定目标参数；
Err＝U1斩波下限阈值
其中，Err表示目标参数，U1表示目标检测周期内检测到的中间直流母线电压值；
其次，获取所述P调节器对应的控制系数，具体为：
S2012、根据如下公式确定所述控制系数；
Kp_chp＝1/(直流母线电压过压保护值阈值-斩波下限阈值)
其中，Kp_chp表示控制系数。
最后，根据所述控制系数和所述目标参数，确定所述斩波占空比，具体为：
S2013、根据如下公式确定所述斩波占空比；
C_duty＝Err*Kp_chp
其中，C_duty表示斩波占空比，Err表示目标参数，Kp_chp表示控制系数。
以图9所示拓扑图为例进行说明：设定斩波上限阈值为3100V，斩波下限阈值为2900V，直流母线电压过压保护值阈值为3200V。图9中U1所测的电压为中间直流母线电压。假设当前检测周期内检测到的中间直流母线电压值U1为3100V，由于U1大于斩波上限阈值，采用P调节器对中间直流母线电压进行调节，首先，根据S2011计算得到目标参数Err为：3100V-2900V＝200V；其次，根据S2012计算得到控制系数K p_ch p为：1/(3200V-2900V)≈0.0033；最后，根据S2013计算得到斩波占空比为：200V*0.0033＝0.66。则在当前检测周期内斩波占空比为0.66。
S1002、根据所述斩波占空比，确定斩波管在目标检测周期内的开通时间。
S1003、根据所述开通时间，控制所述斩波管的开通或关断，以使所述中间直流母线电压值下降至小于所述斩波下限阈值。
由于，斩波占空比指的是：一个检测周期内，斩波管开通的时间占检测周期的比例。继续以S201中的例子进行说明：假设检测周期为1min，在确定当前检测周期内斩波占空比为0.66的基础上，可以计算得到当前检测周期内斩波管的开通时间为1min*0.66＝0.66min。
具体的，在得到上述开通时间后，可基于该开通时间，通过控制斩波管的开通或关断来控制在当前检测周期内斩波管的开通时间为0.66min。
本实施例提供的斩波控制方法，描述了确定斩波占空比的一种可实现的方式，具体为，首先确定目标参数Err，然后确定P调节器的控制系 数，最后根据该目标参数和控制系数，确定斩波占空比，为后续根据该斩波占空比控制斩波管的开通时间提供了依据。
图12为本发明提供的斩波控制方法的实施例三的流程示意图。在上述实施例的基础上，如图12所示，本实施例提供的斩波控制方法，还包括：对所述斩波占空比进行防错处理。
可选地，上述防错处理的实现方式为：
S1201、若所述斩波占空比的值大于1，则将所述斩波占空比的值设为1；若所述斩波占空比的值小于0，则将所述斩波占空比的值设为0。
以图9所示拓扑图为例进行说明：设定斩波上限阈值为3100V，斩波下限阈值为2900V，直流母线电压过压保护值阈值为3200V。图2中U1所测的电压为中间直流母线电压。假设当前检测周期内检测到的中间直流母线电压值为3300V。则根据S2011计算得到目标参数Err为：3300V-2900V＝400V；其次，根据S2012计算得到控制系数Kp_chp为：1/(3200V-2900V)≈0.0033；最后，根据S2013计算得到斩波占空比为：400V*0.0033＝1.32。计算得到的该斩波占空比的值大于1，则将斩波占空比的值设为1。同理，当计算得到的斩波占空比的值小于0时，则将斩波占空比的值设为0。
本实施例提供的斩波控制方法，描述了对斩波占空比进行防错处理的可实现方式，具体为，若所述斩波占空比的值大于1，则将所述斩波占空比的值设为1；若所述斩波占空比的值小于0，则将所述斩波占空比的值设为0。可控制斩波占空比的比例在0到1的范围内。
可选地，在前述实施例的基础上，本发明一实施例中还提供一种电力机车用大功率直驱永磁电传动系统中对于直驱永磁同步电机的控制方法，采用基于速度的分段矢量控制策略完成电流闭环控制，以根据机车的运行条件，满足对高速度运行范围、高转矩性能、高效率的要求。
具体地，图13为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的控制方法流程示意图，如图13所示实施例中包括：
S1301：确定待控制直驱永磁同步电机的转速；
S1302：根据转速与第一映射关系确定第一控制策略，第一映射关系包括至少一个转速的范围和至少一个控制策略的一一对应关系；
S1303：根据第一控制策略确定待控制直驱永磁同步电机的预期控制相角。
可选地，上述实施例中第一映射关系至少包括：额定转速以下与MTPA控制策略的对应关系；额定转速以上与弱磁控制策略的对应关系。
具体地，本实施例中的直驱永磁同步电机采用基于速度的分段矢量控制策略完成电流闭环控制，该控制策略包括：低速区的最大转矩电流比(MTPA)控制和高速区的弱磁控制。图14为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的控制系统的结构示意图，下面结合图14对上述实施例进行说明。如图14所示，其中，T_cmd为输入转矩，T为经过转矩限幅后的实际输入转矩，id*和iq*为d轴和q轴电流给定，id和iq为d轴和q轴反馈电流，ud*和uq*为d轴和q轴电压给定，ua、ub、uc分别为电机a相、b相和c相输入相电压，ia、ib为电机a相、b相电流。
对于在额定转速以下，采用的MTPA控制，即利用永磁同步电机凸极效应产生的磁阻转矩，来获得较高转矩电流比值的一种控制方法。又被称为最大转矩电流比控制，其控制实现框图如图15所示，图15为本发明MTPA控制的系统结构示意图。其中，MTPA控制是非弱磁下所采用的控制策略，由于凸极电机直轴电感Ld小于交轴电感Lq，电机在额定转速以下范围内运行时，可以利用电机的凸极效应而产生的磁阻转矩来获得较高的转矩电流比值。该策略的关键是设定正确的电流工作点，而系统的动态响应由优化的电流内环控制实现，目前常用的电流内环有前馈解耦控制、反馈解耦控制、内模解耦控制和偏差解耦控制等。针对系统在高加、减速工况下，d、q轴电流存在严重动态耦合影响系统动态性能的问题，采用一种优化的前馈解耦控制策略实现对电流内环的优化控制。MTPA控制框图如图15所示。其中，udf和uqf分别为d轴和q轴的前馈电压。前馈解耦是在电流控制器的输出信号u sd、u sq处，分别加上解耦电压项
Figure PCTCN2018117085-appb-000015
和
Figure PCTCN2018117085-appb-000016
从而抵消励磁、转矩电流间的耦合作用。其中，MTPA控制具体包括如下步骤：根据转矩电流曲线确定q轴电流给定和d轴电流给定；计算q轴电流给定与q轴实际电流的第一差值和d 轴电流给定与d轴实际电流的第二差值；通过第一PI控制器根据第一差值得到d轴电压给定、通过第二PI控制器根据第二差值得到q轴电压给定；计算q轴电压给定与q轴前馈电压之和得到实际q轴电压给定、计算d轴电压给定与d轴前馈电压之和得到实际d轴电压给定。如图15所示，首先根据输入以及转矩电流曲线确定给定的d轴电流给定id*和q轴电流给定iq*，随后将id*和d轴实际电流id相减后送入PI控制器、将iq*和q轴实际电流iq相减后送入PI控制器。如图中两个PI控制器会分别计算得到d轴电压给定ud和q轴电压给定uq。随后，将所计算的d轴电压给定ud加上d轴前馈电压udf得到ud*为实际所输出的d轴电压给定，并将所计算的q轴电压给定uq加上q轴前馈电压uqf得到uq*为实际所输出的q轴电压给定。
特别地，图16为本发明前端解耦控制的系统结构示意图。如图16所示，假设反电势分量已抵消，则需要进行前端解耦控制。其中，根据图16中的前端结构控制框图，可以写为矩阵形式的前端结构的电压计算方程为：
根据上图可以写为矩阵形式，于是前馈解耦的电压计算方程为
Figure PCTCN2018117085-appb-000017
进一步地，该前端结构的电压计算方程可以写为矩阵表示的形式
Figure PCTCN2018117085-appb-000018
相应的可求得前馈解耦的闭环传递函数矩阵
Figure PCTCN2018117085-appb-000019
图17为本发明弱磁控制的系统结构示意图。由于受系统变流器容量限制，永磁同步电机稳态运行时，端电压和定子电流都会受到闲置，不能超出电压、电流极限值，为进一步拓宽调速范围，采用弱磁控制，在额定转速上，永磁同步电机进入弱磁状态，通过控制励磁电流可以达到弱磁升速的目的。因此，采用基于上述控制策略的控制算法计算获取当前d轴电压给定值和当前q轴电压给定值，进一步，根据当前d轴电压给定值和当前q轴电压给定值获取当前预期控制相角。如图17所示，受变流器容量限制，永磁同步电机稳态运行时，端电压us和定子电流is都要受到限制，不能超出电压、电流极限值，为进一步拓宽调速范围，采用弱磁控制。在额定转速以上永磁同步电机进入弱磁状态，通过控制励磁电流可以达到弱磁升速的目的；电流环采用功角控制策略，此时逆变器施加在电机上的电压不可控，只有通过控制电机 的功角β来调节电机的励磁和扭矩，这时只控制电机d轴电流，其PI调节器的输出控制功角，实现对永磁电机基频以上的功角控制。其中，Usmax、Ismax分别为电压极限值和电流极限值，Δid为给定弱磁状态下励磁电流的变化量，id_wk*、iq_wk*分别为弱磁调节后的d轴和q轴电流给定，uf为前馈电压幅值，β为功角。具体地，弱磁控制具体包括如下步骤：通过PI控制器根据电压极限值与前馈电压幅值之差计算给定弱磁状态下d轴电流变化量；通过给定弱磁状态下d轴电流变化量和d轴电流给定之和得到弱磁调节后的d轴电流给定；根据d轴电流给定和转矩公式计算弱磁调节后的q轴电流给定；通过PI控制器根据q轴电流给定与q轴实际电流之差得到功角β；通过如下公式计算实际q轴电压给定和实际d轴电压给定；
U d＝U s cosβ
U q＝U s cosβ
其中，Us为电压极限值，Ud为实际d轴电压给定，Uq为实际q轴电压给定。具体地，如图17所示，在弱磁控制中，首先需要将电压极限值Us与前馈电压幅值uf作差相减处理，并通过PI控制器得到给定弱磁状态下励磁电流的变化量Δid，将Δid与d轴电流给定之和作为得到弱磁调节后的d轴电流给定id_wk*送入转矩公式，根据转矩公式反推出得到弱磁调节后的q轴电流给定iq_wk*。随后将q轴电流给定与q轴实际电流iq作差后送入PI控制器，由PI控制器得到功角β，最后根据上述公式计算出实际q轴电压给定和实际d轴电压给定作为输出。可选地，在如图17所示的实施例中，计算前馈电压幅值uf时，需要先通过PI控制器得到前馈的Δid，将Δid与d轴电流给定之和作为id_wk*，并通过根据转矩公式反推出iq_wk*，将id_wk*和iq_wk*送入电压方程计算出d轴电压给定udf和q轴电压给定uqf后，通过公式
Figure PCTCN2018117085-appb-000020
计算出前馈电压幅值uf。
此外，图18为本发明全速度范围内MTPA控制和弱磁控制的轨迹示意图。如图18所示的全速度范围内的控制轨迹中，在以id和iq为坐标轴的坐标系下，OA段为MTPA控制轨迹，AB和BC段为弱磁控制轨迹；ωr1为额定转速，ωr2为最高转速。-ψf/Ld为电压极限圆的圆心。
进一步地，图19为本发明MTPA控制和弱磁控制切换控制示意图。如图19示出了两种控制策略切换的框图，由于MTPA控制策略和弱磁控制策略之间要能平滑、可靠的过度，当逆变器输出的电压达到电压极限圆附近时， 切换到弱磁控制状态，此时切换瞬间的电压矢量角度作为弱磁控制的初始相位角β0；当从弱磁控制切换到MTPA控制时，电压Usd和Usq由最后一拍的功角计算得出。其中，饱和电压为：Usat＝2*Udc/pi。
可选地，在前述实施例的基础上，本发明一实施例中还提供一种电力机车用大功率直驱永磁电传动系统中对于直驱永磁同步电机的调制方法，通过计算调制相角，以通过PWM调制实现实际控制相角。
由于大功率牵引传动系统其牵引变流器通常功率较大，受开关器件散热以及开关损耗的影响，需要工作在较低的开关频率下，通常不超过1000Hz，一方面其最高开关频率一般在几百赫兹左右，另一方面其输出达到额定值时工作在方波工况，因此在整个速度范围内，载波比的变化范围非常大。
因此，本实施例提供一种多模式PWM调制策略，一方面可以充分利用逆变器的允许开关频率，另一方面保证进入弱磁控制区后能够有较高的直流电压利用率。图20为本发明提供的电力机车用大功率直驱永磁电传动系统中，对于直驱永磁同步电机的调制方法流程示意图；如图20所示，本实施例提供的方法包括：
S2001：获取待调制直驱永磁同步电机的调制波的频率；
S2002：根据调制波的频率所在范围与第二映射关系确定第一调制策略，第二映射关系包括至少一个调制波的频率范围和至少一个调制策略的一一对应关系。
S2003：根据第一调制策略确定直驱永磁同步电机的PWM载波频率。
其中，可选地，第二映射关系至少包括：调制波的频率为低速阶段时对应异步调制策略；调制波的频率大于低速阶段低于高速阶段时对应中间60度同步调制策略；调制波的频率为高速阶段时对应方波调制策略。
具体地，多模式PWM调制策略主要由异步SPWM调制、同步SPWM调制和方波调制组成。其中，
1、在低速阶段采用异步调制策略；异步调制在载波比比较大时，由异步调制方式造成的正负半周不对称的影响较小，引入的低次谐波可以忽略。2、当转速升高后，采用中间60度同步调制策略；随着电机频率的上升，载波比的下降，这种低次谐波的影响越来越大，此时采用同步调制PWM。但是常规 的规则采样同步调制在载波比比较低时，低次谐波含量高，采样得到的基波电压幅值达不到指令值的要求，不利于进入方波，此时应当采用特殊调制方法，使电流具有较好的谐波特性和对称性，顺利进入方波。3、在高速阶段则采用方波调制；牵引逆变器为输出更高的基波电压，提升牵引电机最大输出转矩，其在高速段将运行于方波工况，调制方式采用方波调制。
本实施例中在获取当前调制相角的过程中的具体的低速、高速均为转子的角速度，具体的划分规则可与现有技术中的划分规则相似。
图21为本发明提供的中间60°调制方式下调制角度与调制比的关系；图22为本发明提供的基于中间60°调制的全速度范围调制策略示意图。如图22所示，在低速阶段采用异步调制策略；当转速升高后，采用不同载波比的规则采样同步调制和中间60度同步调制策略；高速阶段则采用方波调制。其中涉及到的切换过程主要包括异步调制到SVPWM同步调制之间的切换，同步调制SVPWM与中间60°调制之间的切换，以及中间60°调制内部之间的切换。其中主要的切换难点在于同步调制SVPWM与中间60°调制之间的切换。在15分频下，每个基波周期有15个载波，每个载波对应的基波相位为24°，而中间60°七分频调制下，每个载波周期对应的基波相位为20°。在载波型PWM中，必须要等到一个载波周期结束后才能进行切换，所以为了保证基波相位的连续，切换点处的相位必须为切换前后每个载波周期对应相位的公倍数，20°和24°的公倍数为120°，这意味着在一个周期中只有三个点可以进行切换，分别为0°，120°和240°，切换过程中每一相对应其中一个点。如果电机漏感较小，那么在切换过程中可能会引起一定的冲击，而另外两种切换过程可以做到无冲击切换。此外，需要说明的是，本实施例中横坐标为本实施例中由调制算法获取的调制波的频率。纵坐标为PWM载波频率。
特别地，如图21中示出了中间60°九分频，七分频，五分频和三分频下调制角度β和调制比的关系。示出了通过本实施例中的中间60°的调制方法，如果不考虑死区的影响，可以保证实际输出电压和参考值的完全吻合，具有非常高的电压控制精度。此外，本实施例所采用的中间60°调制的特点可总结为：(1)中间60°同步调制能够在脉冲数不是3的倍数时实现输出电压波形三相之间的对称性，每一相正负半周以及1/4周期的对称性，从而 使得电机线电压和电流中只含有6k±1次谐波；(2)该调制方式下的开关角度能够在线实时计算，所需计算量很小。实现过程对硬件要求比较低，脉冲的发出比较容易；(3)通过数字控制，中间60°调制能够准确的输出所需的基波电压，不同脉冲数下的最大输出电压如果不考虑最小脉宽的限制都可以直接过渡到方波；(4)中间60°调制在脉冲数大于9时，电流谐波不能得到明显改善。不同脉冲数下具有比较一致的低次电流谐波特性，造成低次转矩脉动在不同脉冲数和调制比下都具有稳定的相对较大的脉动幅值；(5)中间60°调制下的电机定子磁链轨迹全部为六边形轨迹，脉冲数的增多只是在每个扇区中增加了电压零矢量的数量，即增加了定子磁链的停顿次数。
可选地，在本发明控制方法的一种具体实现方式中，还提供一种对于电力机车用大功率直驱永磁电传动系统中的直驱永磁同步电机转子初始位置角进行检测的方法，提高对于直驱永磁同步电机转子初始位置角检测的可靠性，以在永磁同步电机的矢量控制中，减少转子的初始位置角的检测不准确对于矢量控制性能的影响。
具体地，图23为本发明提供的直驱永磁同步电机转子初始位置角检测方法实施例一的流程示意图。本实施例中所提供给的直驱永磁同步电机转子初始位置角检测方法的执行主体为本发明所提供的直驱永磁同步电机转子初始位置角检测装置，例如，该装置为TCU控制装置。如图23所示，本实施例的方法包括：
S2301、向待检测直驱永磁同步电机的定子绕组注入高频电压信号，获取三相定子绕组电流。
为使本实施例中的技术方案更加清楚，这里首先对本发明中所涉及的相关的几个坐标系进行介绍。
具体地，本发明所涉及的坐标系包括：两相同步旋转坐标系、两相静止坐标系以及预期两相同步坐标系。其中，图24为本发明提供的两相同步旋转坐标系、两相静止坐标系以及预期两相同步旋转坐标系关系示意图。如图1B所示，α β坐标系为两相静止坐标系，dq坐标系为两相同步旋转坐标系，
Figure PCTCN2018117085-appb-000021
坐标系为预期两相同步旋转坐标系。
由于直驱永磁同步电机在运行的过程中，预期转子位置角与实际转 子位置角之间可能存在误差，因此，定义转子位置角估计误差为：
Figure PCTCN2018117085-appb-000022
其中，
Figure PCTCN2018117085-appb-000023
为预期转子位置角，θ为实际转子位置角，Δθ为转子位置角估计误差。
在上述预期两相同步旋转坐标系下，向直驱永磁同步电机的定子绕组注入高频电压信号。
一种可能的实现方式，向预期两相同步旋转坐标系的注入如下公式所示的高频电压信号：
Figure PCTCN2018117085-appb-000024
其中，U mh为高频电压信号的幅值，ω h为高频电压信号的角频率，t表示注入高频电压信号的时间。
由上述公式可知，向直驱永磁同步电机的定子绕组中注入的高频电压信号的两个分量是线性无关的，由此可获取直驱永磁同步电机的电感参数。具体地，可根据现有技术中所建立的直驱永磁同步电机的数学模型以及相关的计算方法获取直驱永磁同步电机的电感参数。
注入高频电压信号后，获取定子绕组的响应电流，该响应电流即为三相定子绕组电流。一种可能的实现方式，可通过电流传感器获取三相定子绕组电流。
其中，三相定子绕组电流可采用i a，i b和i c表示。
S2302、根据三相定子绕组电流获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流。
需要说明的是，d轴目标电流和q轴目标电流均为注入的高频电压信号根据直驱永磁同步电机结构以及磁饱和特性在定子绕组上激励出的相应的电流分量，d轴目标电流和q轴目标电流均与转子位置角估计误差有关，通过对d轴目标电流和q轴目标电流进行信号处理，可获取转子初始位置角。
因此，根据预期两相同步旋转坐标系以及两相静止坐标系之间的关 系，对三相定子绕组电流进行坐标转换，从而获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流。
一种可能的实现方式，首先对三相定子绕组电流i a，i b和i c进行克拉克(Clarke)变换，获取两相静止坐标系下的α轴电流i α和β轴电流i β，之后，再对α轴电流和β轴电流进行派克(Park)变换，从而获取d轴目标电流
Figure PCTCN2018117085-appb-000025
和q轴目标电流
Figure PCTCN2018117085-appb-000026
进一步，d轴目标电流
Figure PCTCN2018117085-appb-000027
和q轴目标电流
Figure PCTCN2018117085-appb-000028
如下公式所示：
Figure PCTCN2018117085-appb-000029
其中，L为平均电感L＝(L d+L q)/2，△L为半差电感△L＝(L d-L q)/2。
由上述公式可知，d轴目标电流
Figure PCTCN2018117085-appb-000030
和q轴目标电流
Figure PCTCN2018117085-appb-000031
均与转子位置角估计误差Δθ有关。
S2303、根据d轴目标电流和q轴目标电流获取转子的初始位置角。
其中，上述初始位置角为根据直驱永磁同步电机的磁极极性进行补偿后的初始位置角。
具体地，根据上述公式可知，q轴目标电流
Figure PCTCN2018117085-appb-000032
中包含转子初始位置信息，因此，可对q轴目标电流进行信号处理，提取转子的初始位置角。
而直驱永磁同步电机磁极的极性信息与d轴电感有关，因此，可根据直驱永磁同步电机的d轴电感的非线性磁化特性获取磁极的极性信息
进一步，根据磁极极性对转子的初始位置角进行补偿，从而得到补偿后的初始位置角，并将补偿后的初始位置角确定为转子的初始位置角。
本实施例中，首先向待检测的直驱永磁同步电机的定子绕组注入高频电压信号，获取三相定子绕组电流，之后根据三相定子绕组电流获取 预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流，进一步，根据d轴目标电流和q轴目标电流获取转子的初始位置角，其中，初始位置角为根据直驱永磁同步电机的磁极极性进行补偿后的初始位置角。本发明所提供的方法通过考虑直驱永磁同步电机的磁极的影响，根据磁极的极性对转子的初始位置角进行补偿，得到的转子初始位置角准确度更高，提高了初始位置角检测的可靠性。另外，本发明所提供的方法在转子静止的工况，也能够得到准确度较高的检测结果，适用范围较广。另外，本发明所提供的方法无需考虑直驱永磁同步电机的参数，更易于实现。
在图23所示实施例的基础上，一些实施例中，S2303、根据d轴目标电流和q轴目标电流获取转子的初始位置角，可通过以下方式实现：
首先，根据q轴目标电流获取转子的第一初始位置角。
一种可能的实现方式，当转子位置角估计误差Δθ为零时，q轴目标电流
Figure PCTCN2018117085-appb-000033
为零，对q轴目标电流
Figure PCTCN2018117085-appb-000034
进行信号处理，获取转子的位置角的误差输入信号，并根据误差输入信号获取转子的初始位置角。
进一步，根据d轴目标电流获取转子的磁极补偿角。
直驱永磁同步电机磁极的极性信息与d轴电感有关，因此，可根据直驱永磁同步电机的d轴电感的非线性磁化特性获取磁极的极性信息。
进一步，根据第一初始位置角以及磁极补偿角，获取转子的初始位置角。
该实施例采用磁极补偿角对第一初始位置角进行补偿，将补偿后的第一初始位置角确定为转子的初始位置角。
接下来，对根据q轴目标电流获取转子的第一初始位置角的具体实现方式进行介绍。
图25为本发明提供的永磁同步电机转子初始位置角检测方法实施例二的流程示意图。如图25所示，根据q轴目标电流获取转子的第一初始位置角，可以包括：
S2501、对q轴目标电流进行低通滤波处理，获取误差输入信号。
其中，误差输入信号为与转子的初始位置角相关的误差信号。
一种可能的实现方式，采用调制信号对q轴目标电流进行调制，获取 调制后的q轴目标电流，进一步，对调制后的q轴目标电流进行低通滤波处理，获取误差输入信号。
具体地，对q轴目标电流
Figure PCTCN2018117085-appb-000035
与调制信号2sin(ω ht)相乘，得到调制后的q轴目标电流。
其中，调制后的q轴目标电流表示为
Figure PCTCN2018117085-appb-000036
进一步地，通过一个低通滤波器对调制后的q轴目标电流进行滤波处理，滤除2倍频的信号分量，得到误差输入信号f(△θ)，其中，
Figure PCTCN2018117085-appb-000037
其中，LPF表示低通滤波。
由上述公式可知，该误差输入信号中包括转子位置估计误差。在低通滤波过程中，考虑滤波器相位延迟对提取信号影响，在实现时考虑增加延时补偿，保证高频电压注入相位和估计角度相位一致。
进一步地，当转子位置估计误差足够小，极限等效线性化后该误差输入信号，即：
Figure PCTCN2018117085-appb-000038
S2502、根据误差输入信号，获取第一初始位置角。
该步骤中，将误差输入信号作为锁相环的PI调节器的输入，PI调节器根据输入误差信号获取误差输入信号的比例偏差和积分偏差，进一步，根据比例偏差和积分偏差的线性组合，获取第一初始位置角。
具体地，可通过以下公式获取第一初始位置角：
Figure PCTCN2018117085-appb-000039
其中，s表示拉普拉斯算子，k p为比例项系数，k i为积分项系数；
调节PI调节器的比例项系数和积分项系数使得f(△θ)收敛，PI调节器的输出项即为转子第一初始位置角θ first。
本实施例中，通过对q轴目标电流进行调制以及低通滤波处理，获取 误差输入信号，进一步，采用PI调节器对误差输入信号进行锁相输出，从而得到第一初始位置角。
接下来，对根据d轴目标电流获取转子的磁极补偿角的具体实现方式进行介绍。
图26为本发明提供的永磁同步电机转子初始位置角检测方法实施例三的流程示意图。如图26所示，根据d轴目标电流获取转子的磁极补偿角可以包括：
S2601、向永磁同步电机注入多个电压幅值相等、角度不同的电压脉冲信号，获取每个电压脉冲信号的响应电流。
永磁同步电机的磁极具有非线性饱和特征。具体地，向永磁同步电机的d轴注入电压脉冲信号，当电压脉冲信号的角度越接近永磁同步电机的N极，响应电流的幅值越大；当电压脉冲信号的角度越远离永磁同步电机的N极，响应电流的幅值越小。需要说明的是，d轴即为永磁同步电机的直轴，q轴即为永磁同步电机的交轴。
因此，向永磁同步电机注入多个电压幅值相等、角度不同的电压脉冲信号，获取每个电压脉冲信号的响应电流，从而获取响应电流的幅值的变化规律。
一种可能的实现方式，向永磁同步电机注入间隔预设角度、幅值相等的多个电压脉冲信号，通过电流传感器进行采样，获取多个电压脉冲的响应电流，进一步获取响应电流的幅值的变化规律。例如，向永磁同步电机注入每隔5°、幅值相等的电压脉冲信号。可以理解的是，预设角度也可以更小或者更大，本发明对此不做限制。需要说明的是，预设角度越小，获取的响应电流的数据越多，得到的关于响应电流的幅值的变化规律的准确度也更高，预设角度越大，获取的响应电流的数据越少，得到的关于响应电流的幅值的变化规律的准确度较低，因此，在实际的应用过程中，可根据实际情况选择合适的预设角度。
另一种可能的实现方式，向永磁同步电机注入多个特殊角度的、幅值相等的多个电压脉冲信号，通过电流传感器进行采样，获取多个电压脉冲的响应电流，进一步获取响应电流的幅值的变化规律。
S2602、根据多个响应电流，确定转子的磁极补偿角。
具体地，根据多个响应电流的幅值，来确定转子的磁极补偿角。
当注入的电压脉冲信号的角度与第一初始位置角之差满足预设误差范围，电压脉冲信号的响应电流的幅值大于第一值，则确定转子的磁极补偿角为0，其中，第一值为多个响应电流的幅值的最大值。也就是说，确定d轴方向即为磁极N极方向。
当注入的电压脉冲信号的角度与第一初始位置角之差满足预设误差范围，电压脉冲信号的响应电流的幅值小于第二值，则确定转子的磁极补偿角为π，其中，第二值为多个响应电流的幅值的最小值。也就是说，确定d轴方向即为S极方向。
相应地，转子的初始位置角即为第一初始位置角与磁极补偿角之和。具体地，当确定d轴方向为N极方向时，转子的初始位置角等于第一初始位置角，当确定d轴方向为S极方向时，转子的初始位置角等于第一初始位置角与磁极补偿角π之和。
本实施例中，通过根据永磁同步电机直轴电感非线性饱和特性获取的磁极极性辨识的准确性较高，且在实现的过程中无需考虑永磁同步电机的电机参数的影响，可靠性较高，且更易于实现。
接着，以一台1200kW的永磁同步电机为例对本发明的方法在实施的过程中，一些具体参数的设置进行说明：
逆变器开关频率为500Hz，电机额定功率1200kW，电机额定转矩为32606N.m，额定电压2150V，额定电流375A，额定转速为350r/min，电机极对数7，电机d轴电感Ld为0.008771H，电机q轴电感Lq为0.012732H。
在向该永磁同步电机注入高频电压信号的幅值为180V，高频电压信号的角频率为200Hz，逆变器开关频率为500Hz。
永磁同步电机在运行过程中，采集多个通道的信号变化，其中，图27为永磁同步电机运行过程中多个通道的信号变化示意图。如图27所示，由上至下通道依次为：永磁同步电机UV相线电压信号，永磁同步电机U相上管脉冲信号，母线电压信号，永磁同步电机U相电流信号，永磁同步电机V相电流信号。
进一步，采用本发明实施例所提供的方法向上述永磁同步电机注入 电压幅值相等、角度不同的电压脉冲信号，获取电压脉冲信号对应的响应电流。其中，图28为响应电流变化规律示意图，如图28所示，当注入的电压脉冲信号的角度越靠近永磁同步电机的N极，响应电流幅值越大；当注入的电压脉冲信号的角度越远离永磁同步电机的N极，响应电流幅值越小。
进一步地，将通过检测旋转变压器获取的转子实际位置角和根据控制算法计算获取的转子预期位置角进行比较，通过多组数据对比，可知计算误差在±1.2°左右，误差较小。
表1
Figure PCTCN2018117085-appb-000040
可选地，在本发明控制方法的一种具体实现方式中，还提供一种对于电力机车用大功率直驱永磁电传动系统中的直驱永磁同步电机实际控制相角的方法，以实现提高直驱永磁同步电机实际控制相角的准确性。
具体地，图29为本发明提供的电力机车用大功率直驱永磁电传动系统对应的直驱永磁同步电机的控制系统的结构示意图，如图29所示，该直驱永磁同步电机的控制系统包括：直驱永磁同步电机、拖动机、牵引控制器TCU、和旋转变压器。
其中，本发明提供的直驱永磁同步电机的控制方法的控制对象即为直驱永磁同步电机，其中，直驱永磁同步电机包括定子和转子。
旋转变压器安装于直驱永磁同步电机的转子上，用于采集转子信号，并将采集到的信号输入至牵引控制器。在本发明中，旋转变压器具体用于检测转子的实际位置。
拖动机与被测直驱永磁同步电机连接，用于拖动直驱永磁同步电机运转。
牵引控制器与直驱永磁同步电机连接，用于对直驱永磁同步电机进行控制。在本发明中，牵引控制器用于对直驱永磁同步电机进行基于速度的分段矢量控制策略，其中，对于基于速度的分段矢量控制策略在后续实施例中再进行详细说明。具体地，牵引控制器具有控制算法、调制算法的功能，且具有相角调节、转速监测的功能。
可选地，本发明中的牵引控制器包括控制算法单元、调制算法单元、相角调节器和转速检测器。其中，控制算法单元用于获取预期控制相角；调制算法单元用于获取调制相角，之后通过PWM调制实现实际控制相角；相角调节器，用于实现预期控制相角和实际控制相角始终保持一致；转速检测器，用于获取转子的角速度。需说明的是，上述提及的控制算法单元、调制算法单元、相角调节器和转速检测器等既可以为软件模块，也可以为实体模块，本发明不对其进行限制。
下述实施例中均是以牵引控制器作为执行主体实施本发明所提供的直驱永磁同步电机的控制方法。
图30为本发明提供的直驱永磁同步电机的控制方法的流程示意图一，图30所示方法流程的执行主体为牵引控制器，该牵引控制器可由任意的软件和/或硬件实现。如图30所示，本实施例提供的直驱永磁同步电机的控制方法包括：
S3001、根据控制中断周期、调制载波周期，以及直驱永磁同步电机的转子当前角速度，获取直驱永磁同步电机的转子的补偿相角。
本实施例中获取的直驱永磁同步电机的转子的补偿相角为离线补偿相角，即若直驱永磁同步电机的控制系统中各部件在获取补偿相角和正常运行的设置保持不变时，可以将离线获取的补偿相角应用于正在运行 的直驱永磁同步电机的控制系统中。可以想到的是，当直驱永磁同步电机的控制系统中各部件的设置发生改变时，可采用改变后的设置参数获取新的补偿相角。
具体的，牵引控制器可采用控制算法对旋转变压器采集到的电压信号进行处理，获取预期相角，具体的，牵引控制器可控制其中的控制算法单元对旋转变压器采集的电压信号进行处理，获取预期相角。其中，旋转变压器的采样周期可以与控制算法的控制中断周期相同。
示例性地，旋转变压器在t1时刻进行采样，并将采集到的电压信号输入至牵引控制器。牵引控制器的控制算法单元在t1时刻对旋转变压器采集的电压信号进行处理，获取预期相角，并在下一个控制中断周期开始至下一个控制中断周期结束的这段时间内的不定时刻进行更新，也就是，将预期相角输出给调制算法单元。而在这个过程中，转子仍在不停地旋转，相对于旋转变压器采样时刻，会产生控制算法中断时延。进一步，根据控制算法中断时延的时长和转子的角速度，获取在控制算法过程中转子的误差相角。
优选地，控制算法时延为半个控制中断周期。
牵引控制器获取预期相角，并采用调制算法对该预期相角进行调制输出处理。具体的，牵引控制器的调制算法单元采用调制算法对预期相角进行调制，输出PWM脉冲。本实施例中的调制采样具有周期性，即牵引控制器周期性地获取预期相角，并进行调制处理。示例性地，本实施例中调制载波为三角PWM载波，调制采样采用一种不对称的规则采样法，即在每个三角PWM载波周期的顶点对称轴位置采样，又在三角PWM载波周期的底点对称轴位置采样，也就是每个调制载波周期采样两次。每个调制载波周期开始和中间时刻进行本PWM载波周期的釆样，同时进行本周期的PWM指令更新。双采样模式的调制算法中断分为采样、调制计算、PWM更新和PWM输出过程。
示例性地，牵引控制器在t2时刻获取预期相角，进行PWM调制处理，生成PWM脉冲，之后，通常会在载波周期计数值与调制计算得到的PWM比较计数值相等时进行输出PWM脉冲。而在上述过程中，转子仍在不停地旋转，因此，造成调制更新时延。优选地，调制更新时延为半 个调制载波周期；
另外，PWM计算值更新后一般采用定时器的连续增减计数方式来输出PWM脉冲，输出时也会造成输出时延。优选地，输出时延为1/4个调制载波周期。
根据在调制算法中获取的调制更新时延和输出时延以及转子的当前角速度，可以获取在调制算法过程中的转子的误差相角。
另外，在旋转变压器对转子的位置进行采样和信号传输过程中也会产生时延，这里称为旋转变压器采样和传输时延。具体地，本实施中根据直驱永磁同步电机转子的当前角速度和预设角速度范围内的多个d轴电压和多个q轴电压获取旋转变压器采样和传输时延对应的误差相角。
接下来，对预设角速度范围进行详细的介绍。
由于本申请中对于直驱永磁同步电机传动系统，采用的是基于速度的分段矢量控制策略，分段矢量控制策略包括低速区的最大转矩电流比控制和高速区的弱磁控制。因此，本实施例中的预设角速度范围可以是牵引控制器确定直驱永磁同步电机在不进入弱磁控制阶段、且稳定运行的速度范围。其中，根据直驱永磁同步电机的牵引特性，进入恒压阶段对应的速度点，电压达到最大值时的运行速度，即为不进入弱磁控制阶段、最高稳定运行速度，也就是预设角速度范围的最大值。
在该预设角速度范围获取多个预设角速度中每个预设角速度对应的d轴电压和q轴电压，根据每个预设角速度对应的d轴电压和q轴电压，获取每个预设角速度对应的误差相角，再建立以预设角速度为横坐标，以误差相角为纵坐标的曲线，将该曲线对应的斜率确定为误差系数；进一步，根据转子的角速度以及该角速度对应的误差系数获取误差相角，该误差相角即为旋转变压器采样和传输时延造成的误差相角。
可选地，由上述控制算法时延、调制算法时延、以及旋转变压器采集和传输时延分别对应的误差相角之和即为直驱永磁同步电机的转子的补偿相角。
还需要补充说明的是，在两相同步旋转(d、q)坐标系中，转子磁极产生的磁场与定子磁场相对应时为d轴，逆时针旋转90度为q轴。
S3002、根据补偿相角，获取当前实际控制相角。
步骤S3001中获取的补偿相角为离线补偿相角，将其应用于正在运行的直驱永磁同步电机中。
因此，本步骤中获取的当前实际控制相角为采用步骤S3001中获取的补偿相角对直驱永磁同步电机的转子位置角进行离线修正后的实际控制相角。
S3003、根据当前d轴电压给定值和当前q轴电压给定值，获取当前预期控制相角。
当前电压给定值可以包括当前d轴电压给定值和当前q轴电压给定值。本实施例中根据直驱永磁同步电机所采用的基于速度的分段矢量控制策略以及相应的控制算法，计算获取当前d轴电压给定值和当前q轴电压给定值，进一步，根据当前d轴电压给定值和当前q轴电压给定值获取当前预期控制相角。
S3004、根据当前预期控制相角和当前实际控制相角的比例偏差和积分偏差，对当前实际控制相角进行在线修正。
由于当前预期控制相角和当前实际控制相角由于控制算法、调制算法以及旋转变压器采集和传输过程中的时延，造成当前预期控制相角和当前实际控制相角可能存在偏差，因此，需要对当前实际控制相角进行修正。
该步骤中，将当前预期控制相角和当前实际控制相角的比例偏差以及当前预期控制相角和当前实际控制相角的积分偏差的线性组合作为修正项，对当前实际控制相角进行在线修正。
本实施例提供一种直驱永磁同步电机的控制方法，该方法包括：根据控制中断周期、调制载波周期，以及直驱永磁同步电机的转子当前角速度，获取直驱永磁同步电机的转子的补偿相角；根据补偿相角，获取当前实际控制相角；根据当前d轴电压给定值和当前q轴电压给定值，获取当前预期控制相角；进一步，根据当前预期控制相角与当前实际控制相角的比例偏差和积分偏差，对当前实际控制相角进行在线修正。本发明通过将控制中断对应的时延、载波调制对应的时延以及旋转变压器采样及传输转子信号过程中对应的时延所造成的误差相角考虑在内，对实际控制相角进行在线修正，保证实际控制相角和预期控制相角始终保持 一致，提高了实际控制相角的准确性。
图31对本发明提供的直驱永磁同步电机的控制方法实施例二的流程示意图。如图31所示，在图30所示实施例的基础上，步骤S3001可以包括：
S3101、根据控制中断周期和直驱永磁同步电机的转子的当前角速度，获取第一子补偿相角。
为使本实施例所提供的控制方法更加清楚，这里，对本申请所涉及的控制中断进行详细说明。图32为本发明所提供的控制算法的控制中断示意图。如图32所示，控制中断分为采样、控制计算、控制变量更新的过程。旋转变压器对转子信号进行采样，并在t1时刻将采集到的电压信号输入至牵引控制器。牵引控制器对接收到的电压信号进行控制计算，T ctrl为控制算法的一个控制中断周期，t1+T ctrl时刻完成控制计算，之后会在下一个控制中断周期开始(t1+T ctrl时刻)至结束(t1+2T ctrl时刻)这段时间内的不定时刻将控制计算得到的控制变量输出给调制算法单元。
在这个过程中，转子仍在不停地旋转，相对于控制计算完成的时刻，会产生控制算法中断时延。本实施例中，根据控制算法的控制中断周期，获取第一子补偿相角对应的第一相角时延，其中，A为控制中断时延系数，取值范围为(0-1)。优选地，A＝0.5。
因此，第一相角时延Δ t1可如下公式所示：
Δ t1＝A·T ctrl≈0.5T ctrl
进一步，根据第一相角时延和直驱永磁同步电机的转子的当前角速度，获取第一子补偿相角，第一子补偿相角即为控制算法中断时延对应的误差相角。
具体地，第一子补偿相角θ cmps1可如下公式所示：
θ cmps1＝Δ t1·ω
其中，ω为直驱永磁同步电机的转子的当前角速度。
S3102、根据调制载波周期和直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角。
示例性地，以本实施例中调制载波为三角PWM载波为例进行说明， 为提高直驱永磁同步电机的控制系统的动态相应，调制算法所采用的是不对称的规则采样法，即在每个三角PWM载波周期的顶点对称轴位置采样，又在三角PWM载波周期的底点对称轴位置采样，也就是每个调制载波周期采样两次。每个调制载波周期开始和中间时刻进行本PWM载波周期的釆样，同时进行本周期的PWM指令更新。双采样模式的调制算法中断分为采样、调制计算、PWM更新和PWM输出过程。
其中，图33为本发明提供的调制算法的中断周期示意图。如图33所示，牵引控制器在t时刻进行调制采样，获取的是由控制算法计算的控制变量。具体地牵引控制器获取的控制变量为预期相角，并在t+0.5T PWM时刻完成调制算法计算，并开始进行PWM比较计数值更新和下一个调制周期的预期控制相角采样，通常会在PWM载波周期计数值与调制计算得到的PWM比较计数值相等时输出PWM脉冲，T PWM为PWM的调制载波周期。
在这个过程中，转子仍在不停地旋转，相对于调制计算完成的时刻，会产生调制算法中断时延，即为第三相角时延B·T PWM，其中，B为调制算法中断时延系数。可选地，B＝0.5。
PWM比较计算值更新后一般采用定时器的连续增减计数方式来输出PWM脉冲，在这个过程中会产生PWM脉冲输出时延，PWM脉冲输出时延为C·T PWM，即为第二相角时延。其中，C为PWM脉冲输出时延系数，取值范围为(0-0.5)。可选地，C＝0.25。
具体的，进行调制计算和PWM脉冲输出过程中的时延Δ t2可如下公式所示：
Δ t2＝B·T PWM+C·T PWM≈0.75T PWM
进一步，根据第二相角时延、第三相角时延和直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角，第二子补偿相角即为调制算法时延对应的误差相角。
具体地，第二子补偿相角θ cmps2可如下公式所示：
θ cmps2＝Δ t2·ω
其中，ω为直驱永磁同步电机的转子的当前角速度。
S3103、根据直驱永磁同步电机的转子当前角速度，获取第三子补偿相角。
其中，第三子补偿相角为旋转变压器采样和传输时延对应的误差相角。在稳定运行角速度范围获取多个预设角速度中每个预设角速度对应的d轴电压和q轴电压，根据每个预设角速度对应的d轴电压和q轴电压，获取每个预设角速度对应的误差相角，再建立以预设角速度为横坐标，以误差相角为纵坐标的曲线，将该曲线对应的斜率确定为误差系数；进一步，根据转子的角速度以及该角速度对应的误差系数获取误差相角，该误差相角即为旋转变压器采样和传输时延造成的误差相角。
还需要补充说明的是，在两相同步旋转(d、q)坐标系中，转子磁极产生的磁场与定子磁场相对应时为d轴，逆时针旋转90度为q轴。
S3104、根据第一子补偿相角、第二子补偿相角和第三子补偿相角，获取直驱永磁同步电机的补偿相角。
可选地，第一补偿相角、第二补偿相角和第三补偿相角之和即为直驱永磁同步电机的补偿相角。
S3105、根据补偿相角，获取当前实际控制相角。
首先获取直驱永磁同步电机的转子的当前位置相角，接着根据当前位置相角、转子的初始位置相角以及补偿相角，获取转子的实际位置相角，进一步，根据转子的实际位置相角以及当前调制相角，获取当前实际控制相角，其中，调制相角为采用调制算法并根据d轴电压给定值和当前q轴电压给定值计算得到。
具体地，根据转子的当前位置相角以及转子的初始位置相角获取转子的实际位置相角，进一步，采用上述补偿相角对直驱永磁同步电机的转子位置角进行离线修正，从而将修正后的实际位置相角作为转子的实际位置相角。之后，将转子的实际位置相角以及当前调制相角的差值确定为当前实际控制相角。
一种可能的实现方式，调制算法单元采用多模式PWM调制策略，一方面可以充分利用逆变器的允许开关频率，另一方面保证进入弱磁控制区后能够有较高的直流电压利用率。具体地，多模式PWM调制策略主要 由异步SPWM调制、规则采样同步SPWM调制和方波调制组成。
其中，图34为多模式PWM调制策略的示意图，如图34所示，在低速阶段采用异步调制策略；当转速升高后，采用不同载波比的规则采样同步调制和中间60度同步调制策略；高速阶段则采用方波调制。其中，横坐标为本实施例中由调制算法获取的调制波的频率。纵坐标为PWM载波频率。
本实施例中在获取当前调制相角的过程中的具体的低速、高速均为转子的角速度，具体的划分规则可与现有技术中的划分规则相似。
S3106、根据当前d轴电压给定值和当前q轴电压给定值，获取当前预期控制相角。
具体地，本实施例中的直驱永磁同步电机采用基于速度的分段矢量控制策略完成电流闭环控制，该控制策略包括：低速区的最大转矩电流比(MTPA)控制和高速区的弱磁控制。
在额定转速以下，采用MTPA控制，即利用永磁同步电机凸极效应产生的磁阻转矩，来获得较高转矩电流比值的一种控制方法。由于受系统变流器容量限制，永磁同步电机稳态运行时，端电压和定子电流都会受到闲置，不能超出电压、电流极限值，为进一步拓宽调速范围，采用弱磁控制，在额定转速上，永磁同步电机进入弱磁状态，通过控制励磁电流可以达到弱磁升速的目的。
因此，采用基于上述控制策略的控制算法计算获取当前d轴电压给定值和当前q轴电压给定值，进一步，根据当前d轴电压给定值和当前q轴电压给定值获取当前预期控制相角。
具体地，可根据如下公式进行计算：
Figure PCTCN2018117085-appb-000041
其中，θ ctrl表示预期控制相角，
Figure PCTCN2018117085-appb-000042
表示q轴电压给定值，
Figure PCTCN2018117085-appb-000043
表示d轴电压给定值。
S3107、根据当前预期控制相角和当前实际控制相角的比例偏差和积分偏差，对当前实际控制相角进行在线修正。
一种可能的实现方式，首先，根据当前预期控制相角和当前实际控制相角获取比例偏差、积分偏差，之后再根据比例偏差和积分偏差的线 性组合，获取当前实际控制相角的修正项，进一步，采用该修正项对当前实际控制相角进行在线修正。
可选地，采用如下公式获取修正项：
Figure PCTCN2018117085-appb-000044
其中，k p和k i为修正项，θ ctrl为当前预期相角，θ PWM为当前实际相角，f Δ为基波频率补偿项，为已知量。
牵引控制器获取修正项k p和k i后，通过在线调节修正项，使得当前的实际控制相角快速、无差的跟踪预期控制相角，实现实际控制相角的在线修正。
该步骤中，对相角的控制采用的是闭环PI控制，能够实现对控制相角准确地、无静差的控制，从而提升控制性能。
本实施例中，通过将控制算法、调制算法以及旋转变压器采集及传输造成的时延考虑在内，并根据实际控制相角和预期控制相角的比例偏差和积分偏差，对当前实际控制相角进行在线修正，使得实际控制相角与预期控制相角始终保持一致，提高了实际控制相角的准确性，降低直驱永磁同步电机运行故障的发生概率，从而提高了直驱永磁同步电机牵引系统的控制性能。
图35为本发明提供的直驱永磁同步电机的控制方法流程示意图三。如图35所示，在图31所实施实施例的基础上，可选地，步骤S3103之前包括以下步骤：
S3501、根据直驱永磁同步电机的矢量控制策略，获取直驱永磁同步电机的稳定运行角速度范围。
本实施例中，在上述基于速度的分段矢量控制策略的基础上，首先获取直驱永磁同步电机的稳定运行角速度范围，也就是获取直驱永磁同步电机在不进入弱磁控制阶段、稳定运行的速度范围，其中，进入恒压阶段对应的速度点，电压达到最大值，即为不进入弱磁控制阶段的最高稳定运行速度。
S3502、根据d轴电流给定值和q轴电流给定值，获取所述稳定运行角速度范围内的多个第一d轴电流、多个第一q轴电流、每个所述第一d轴电流对应的d轴电压以及每个所述第一q轴电流对应的q轴电压。
一种可能的实现方式，根据预设角速度间隔，获取所述直驱永磁同步电机的转子处于所述稳定运行角速度范围内时，每隔所述预设角速度间隔对应的多个第一预设角速度；
当每个第一预设角速度对应的d轴电流与d轴电流给定值满足预设误差阈值，且每个第一预设角速度对应的q轴电流与q轴电流给定值满足预设误差阈值时，将每个第一预设角速度对应的d轴电流确定为第一d轴电流、将每个第一预设角速度对应的q轴电流确定为第一q轴电流；
根据每个第一d轴电流获取每个第一d轴电流对应的d轴电压，根据每个第一q轴电流获取每个第一q轴电流对应的q轴电压。
本实施例中，牵引控制器获取的每个第一d轴电流和每个第一q轴电流均为直驱永磁同步电机稳态下的d轴电流和q轴电流。
在稳态条件下，忽略直驱永磁同步电机的微分项，因此，直驱永磁同步电机稳态方程可如下公式所示：
Figure PCTCN2018117085-appb-000045
其中，u d为任一第一预设角速度对应的d轴电压，u q为任一第一预设角速度对应的q轴电压，R s为转子的电阻，L q为任一第一预设角速度对应的d轴电感，L d为任一第一预设角速度对应的q轴电感，i d为d轴电压对应的第一d轴电流，i q为q轴电压对应的第一q轴电流，ψ f为永磁体磁链的反电势。
从上述直驱永磁同步电机稳态方程可以看出，当直驱永磁同步电机的d、q轴电流都为0时，此时的d轴电压为0，q轴电压全部由永磁体磁链的反电势产生。
其中，图36A为理论坐标系与实际坐标系完全重合的示意图，图36B为实际坐标系超前理论坐标系的示意图，图36C为实际坐标系滞后理论坐标系的示意图。
如图36A-36C所示，首先定义控制算法采用的dq坐标系为理论dq坐标系，调制算法实际输出PWM脉冲所采用的dq坐标系为实际
Figure PCTCN2018117085-appb-000046
坐标系。当转子位置定位准确、理想情况下，理论dq坐标系与实际
Figure PCTCN2018117085-appb-000047
坐标系完全重合，u d等于0，u q等于ωψ f，如图36A所示；当转子位置定位超前 情况下，实际
Figure PCTCN2018117085-appb-000048
坐标系超前理论dq坐标系一定角度θ cmps3，u d为正值，u q为正值，如图36B所示；当转子位置定位滞后情况下，实际
Figure PCTCN2018117085-appb-000049
坐标系滞后理论dq坐标系一定角度θ cmps3，u d为负值，u d为正值，如图36C所示。
相应地，步骤S3103可通过以下方式实现：
S3503、根据每个第一d轴电流对应的d轴电压以及每个第一q轴电流对应的q轴电压，获取每个第一角速度对应的传输误差相角。
本实施例中，每个第一d轴电流对应的d轴电压以及每个第一q轴电流对应的q轴电压，获取每个第一预设角速度对应的传输误差相角。获取传输误差相角θ Δ具体可如下公式所示：
θ Δ＝tan -1(u d/u q)
S3504、根据每个第一角速度对应的传输误差相角，以及，所述直驱永磁同步电机的转子的当前角速度，获取所述第三子补偿相角。
以第一预设角速度作为横坐标，以传输误差相角作为纵坐标，可以获取传输误差相角系数k，由传输误差相角系数和直驱永磁同步电机的转子的当前角速度的乘积可获取第三子补偿相角。具体获取第三子补偿相角θ cmps3可如下公式所示：
θ cmps3＝k·ω
本实施例中，根据直驱永磁同步电机的矢量控制策略，获取直驱永磁同步电机的稳定运行角速度范围，根据d轴电流给定值和q轴电流给定值，获取所述稳定运行角速度范围内的多个第一d轴电流、多个第一q轴电流、每个所述第一d轴电流对应的d轴电压以及每个所述第一q轴电流对应的q轴电压，根据每个第一d轴电流对应的d轴电压以及每个第一q轴电流对应的q轴电压，获取每个第一角速度对应的传输误差相角，根据每个第一角速度对应的传输误差相角，以及，所述直驱永磁同步电机的转子的当前角速度，获取所述第三子补偿相角。通过预先获取稳定运行速度范围内多个第一角速度对应的传输误差相角，之后再根据直驱永磁同步电机的转子的当前角速度快速获取第三子补偿相角，并采用第三子补偿相角对实际控制相角进行准确地在线修正，提高了在线修正的效率。
可选地，在本发明控制方法的一种具体实现方式中，还提供一种对于主电路中的直驱永磁同步电机粘着控制的方法，以及时减轻空转和滑行程度，有效提高粘着利用率，使机车牵引力稳定发挥，减少轮对异常负载，降低车轮擦伤、剥离损伤。
其中，当本实施例提供的粘着控制方法应用于如图1所示的电力机车时，通过电力机车上至少四个直驱永磁同步电机进行粘着控制；这里记所述至少四个直驱永磁同步电机包括：第一电机、第二电机、第三电机和第四电机进行说明。
可选地，在本实施例一种可能的实现方式中，电机机车上设置六个直驱永磁同步电机，并通过两个如前述实施例中所示的直驱永磁电机机车变流器主电路对六个直驱永磁同步电机分别进行控制。本实施例的控制方法中参与计算的四个直驱永磁同步电机可以是电力机车的六个直驱永磁同步电机中的任意四个，并且第一电机和第二电机为设置在电力机车上第一转向架的轴电机，第三电机和第四电机为设置在电力机车第二转向架的轴电机。
图37为本发明提供的粘着控制方法一实施例的流程图。本实施例提供的方法可以应用与直驱永磁牵引系统。如图37所示，本实施例提供的方法可以包括：
S3701、采集第一电机、第二电机、第三电机和第四电机的转子频率，获取第一电机的实时转矩，第一电机和第二电机为第一转向架的轴电机，第三电机和第四电机为第二转向架的轴电机，第一转向架与第二转向架相邻。
本实施例中的四个电机位于相邻的转向架上。可以根据第一电机的实时转矩确定机车的运行工况。可以按照预设的采样周期或者预设的采样频率，采集第一电机、第二电机、第三电机和第四电机的转子频率。
S3702、根据所采集的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值。
可选地，本实施例中以第一电机、第二电机、第三电机和第四电机中最小的转子频率作为转子频率基准。第一电机的转子频率差为第一电机的转子频率与转子频率基准之间的差值。
可选地，本实施例中第一电机的转子频率微分值可以为，当前采样时刻第一电机的转子频率与当前采样时刻的前一采样时刻第一电机的转子频率的差值除以采样时间间隔。
S3703、根据第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量。
根据第一电机的转子频率差、转子频率微分值可以快速准确的确定出机车是否处于空转滑行状态。一旦机车发生空转滑行，便可以根据第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量，转矩削减量用于指示第一电机需要卸载的转矩量。
S3704、根据转矩削减量对第一电机的转矩进行调整。
将第一电机的转矩卸载转矩削减量对应的数值，以消除空转滑行现象。
本实施例提供的粘着控制方法，通过采集位于相邻转向架上的第一电机、第二电机、第三电机和第四电机的转子频率，及第一电机的实时转矩，根据所采集的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值，根据第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量，并根据转矩削减量对第一电机的转矩进行调整。根据转子频率确定转矩消减量进行粘着控制，噪声小且抗外部干扰能力强；根据转子频率差以及转子频率微分值能够快速准确的确定机车是否处于空转滑行状态，及时减轻空转和滑行程度，有效提高粘着利用率，使机车牵引力稳定发挥，减少轮对异常负载，降低车轮擦伤、剥离损伤。
可选地，为了进一步提高粘着利用率，在上述实施例的基础上，本实施例提供的方法还可以包括：
根据第一电机的转子频率差、转子频率微分值和实时转矩，生成撒砂控制信号，撒砂控制信号用于指示是否进行撒砂操作。撒砂可以增大轮轨之间的粘着系数，减轻机车的空转和滑行程度。若根据第一电机的转子频率差、转子频率微分值和实时转矩，确定机车的空转滑行等级满足预设条件，则进行撒砂操作。
可选地，根据第一电机的转子频率差、转子频率微分值和实时转 矩，确定转矩削减量，可以包括：
根据第一电机的转子频率差以及预设的转子频率差分级规则，确定第一电机的转子频率差对应的空转滑行等级，根据第一电机的转子频率差对应的空转滑行等级，以及第一电机的实时转矩，确定第一转矩削减量。
预设的转子频率差分级规则可以包括转子频率差与空转滑行等级之间的映射关系，不同的空转滑行等级对应不同的转矩削减系数，例如可以设置空转滑行等级越高对应的转矩削减系数越大。第一转矩削减量可以等于第一电机的实时转矩乘以第一电机的转子频率差对应的转矩削减系数。
根据第一电机的转子频率微分值以及预设的转子频率微分值分级规则，确定第一电机的转子频率微分值对应的空转滑行等级，根据第一电机的转子频率微分值对应的空转滑行等级，以及第一电机的实时转矩，确定第二转矩削减量。
预设的转子频率微分值分级规则可以包括转子频率微分值与空转滑行等级之间的映射关系，不同的空转滑行等级对应不同的转矩削减系数，例如可以设置空转滑行等级越高对应的转矩削减系数越大。第二转矩削减量可以等于第一电机的实时转矩乘以第一电机的转子频率微分值对应的转矩削减系数。
若第一转矩削减量大于等于第二转矩削减量，则确定第一转矩削减量为转矩削减量；若第一转矩削减量小于第二转矩削减量，则确定第二转矩削减量为转矩削减量。即选取第一转矩削减量和第二转矩削减量中较大者作为转矩削减量，对第一电机的转矩进行调整。
在上述任一实施例的基础上，本实施例针对根据转矩削减量对第一电机的转矩进行调整的过程进行详细说明。本实施例中根据转矩削减量对第一电机的转矩进行调整，可以包括：
在第一预设时间段内，将第一电机的转矩值由第一值降低至第二值，第一值与第二值的差值为转矩削减量。
可选地，在第一预设时间段内，根据第一电机的转矩值的降低速率逐渐减小，将第一电机的转矩值由第一值降低至第二值。即对于第一电 机的转矩值的卸载由快至慢，有利于最佳粘着点的搜寻，避免转矩突降。
在第二预设时间段内，保持第一电机的转矩值为第二值不变。
在第三预设时间段内，将第一电机的转矩值由第二值提高至预设转矩值的预设百分比，如可以提高至预设转矩值的90％。
在第四预设时间段内，将第一电机的转矩值提高至预设转矩值。
其中，第一电机的转矩值在第三预设时间段内的恢复速率，大于第一电机的转矩值在第四预设时间段内的恢复速率。即对于第一电机的转矩值的恢复，采用了分段恢复，且先快速恢复后缓慢恢复，可以有效避免再次发生空转滑行。
本实施例中的第一预设时间段、第二预设时间段、第三预设时间段和第四预设时间段的具体时长可以根据需要进行设置，本实施例对此不做限制。第一预设时间段、第二预设时间段、第三预设时间段和第四预设时间段构成一个转矩调整周期，在发生空转滑行时，对第一电机的转矩进行调整。
图38为本发明一实施例提供的粘着控制过程的示意图。图38为发生空转时，本发明一实施例提供的粘着控制方法对于第一电机的转矩的调整过程的示意图。如图38所示，T1、T2、T3和T4分表表示第一预设时间段、第二预设时间段、第三预设时间段和第四预设时间段，T1、T2、T3和T4构成了一个转矩调整周期。其中机车基准参考频率曲线表示机车处于牵引工况下，第一电机的转子频率应该遵循的变化趋势，转子频率曲线表示第一电机的实际的转子频率。
T1阶段为进行转矩卸载的阶段，a点为机车发生空转的时刻点，如图38所示，一旦检测到发生空转，便立即进行转矩的快速卸载，卸载量由大到小，如图38中a-b段所示转矩卸载曲线可以拟合为反比例函数曲线，然后分别以两种小斜率继续卸载，如图2中b-c段和c-d段所示，其中b-c段的卸载速率大于c-d段的卸载速率，直至转矩卸载量等于所确定的转矩削减量，即a点与d点的转矩差值等于转矩削减量。T2阶段为保持转矩不变的阶段，转矩卸载量达到转矩削减量时，机车不发生空转，维持低转矩输出，如图38中d-e段所示。T3阶段为转矩的第一恢复阶段，在维持 低转矩输出T2时间段后，即空转消失T2时间段后，按照预设速率将转矩恢复至预设转矩的90％，如图38中e-f段所示。T4阶段为转矩的完全恢复阶段，将转矩恢复至预设转矩，如图38中f-g段所示。f-g段转矩的提升速率小于e-f段转矩的提升速率。其中，预设转矩可以为发生空转时刻的转矩，即可以设置预设转矩等于图中a点处的转矩。
本实施例中转矩卸载由快到慢，有利于最佳粘着点的搜寻，避免转矩突降。后期转矩恢复过程，采用了分段恢复，可有效避免再次发生空转。可以理解的是，对于发生滑行的过程类似，此处不再赘述。
可选地，根据所采集的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值，可以包括：
对所采集的多个电机的转子频率进行限幅滤波和低通滤波处理，根据限幅滤波和低通滤波处理后的多个电机的转子频率，确定第一电机的转子频率差和转子频率微分值。通过对转子频率进行限幅滤波和低通滤波处理，能够消除因外界干扰引起的噪声信号，提高转子频率的精度，进而可以提高粘着控制的精度。
可选地，可以根据第一电机的实时转矩确定机车的运行工况，机车的运行工况可以包括惰行工况、牵引工况和制动工况。例如，设置第一转矩阈值和第二转矩阈值，其中，第一转矩阈值大于零，第二转矩阈值小于零，本实施例对于第一转矩阈值和第二转矩阈值的具体数值不作限制，可以根据实际需要进行设定。若第一电机的实时转矩大于等于第一转矩阈值，则机车处于牵引工况；若第一电机的实时转矩小于等于第二转矩阈值，则机车处于制动工况；若第一电机的实时转矩大于第二转矩阈值，且小于第一转矩阈值，则机车处于惰行工况。
可选地，若机车处于惰行工况，则对所采集的多个转子频率进行限幅滤波和低通滤波处理，可以包括：
获取第一电机的电流值，根据第一电机的电流值和每个电机的转子频率，确定每个电机的转子频率补偿系数，根据每个电机的转子频率补偿系数对每个电机的转子频率进行补偿，对补偿后的多个电机的转子频率进行限幅滤波和低通滤波处理。
本实施例中根据第一电机的电流值和每个电机的转子频率，为各个 电机确定转子频率补偿系数进行补偿，提高了转子频率采集精度，进而提高了粘着控制的精度。
进一步可选地，在本发明前述各实施例的基础上，本发明一实施例还提供一种辅助变流器的控制设备，用以实现如前述图1实施例中S104对辅助变流器进行的控制及其相关功能。
具体地，图39为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图一，如图39所示，该设备包括：数字信号处理(Digital Signal Processing，DSP)芯片101以及现场可编程门阵列(Field Programmable Gate Array，FPGA)芯片102，DSP芯片101与FPGA芯片102总线连接，其中
FPGA芯片102用于通过模拟量采样板卡103和数字量采样板卡104获取辅助变流器的模拟量和数字量，并对模拟量和数字量进行逻辑运算处理，得到逻辑运算处理后的数据；
DSP芯片101用于对逻辑运算处理后的数据进行控制运算处理，得到脉冲宽度；
FPGA芯片102还用于脉冲宽度进行调制运算处理，得到辅助变流器的驱动脉冲序列。
其中，DSP芯片101是一种以数字信号来处理大量信息的器件，其工作原理是接收模拟信号，并将其转换为0或1的数字信号，再对数字信号进行处理，并在其他系统芯片中把数字信号数据转换为需要的数据类型从而完成相应的需求，DSP芯片具有可嵌入型、稳定性好以及可编程性等优点，但是同时由于DSP芯片是通过串行执行指令的方式来完成工作的，因此DSP芯片在数据采集和数据处理方面的效率较低。
FPGA芯片102是指采用FPGA来实现功能的芯片，一个FPGA是一种包含有一个可重配置的门阵列逻辑电路矩阵的设备，通过配置FPGA的内部电路以一定的方式相连接，从而创建了软件应用的一个硬件实现，FPGA采用专门的硬件进行逻辑处理，所以FPGA在数据采集以及数据处理方面速度较快，进一步地，FPGA是通过并行执行指令的方式来完成工作的，因此采用FPGA芯片102配合DSP芯片101实现辅助变流器的控 制功能，能够有效地弥补DSP芯片的不足之处，提高工作效率。
首先利用FPGA芯片102进行数据采样，获取辅助变流器的模拟量和数字量，数据采样操作由模拟量采样板卡103和数字量采样板卡104来完成，具体地，采样板卡主要用于采集数据，模拟量采样板卡103可采集得到辅助变流器的模拟量，数字量采样板卡104可采集得到辅助变流器的数字量，其中模拟量和数字量都是辅助变流器在工作时生成的相关数据，其中模拟量是指变量在一定范围连续变化的量，也就是在定义域范围内总可以对应一个值的量，例如可以为电压传感器测量得到的电压值、电流传感器测量得到的电流值等，而数字量是指变化在时间上是不连续的，总是发生在一系列离散的瞬间，数字量是分散开来的，不存在中间值的量，例如可以为开关信号，开关在某一时刻打开，在另一时刻开关关闭，在时间上是不连续的，可以通过“0”、“1”来表示数字量。
在FPGA芯片102采集得到相关数字量和模拟量之后，首先对数字量和模拟量进行逻辑运算处理，逻辑运算处理例如可以为对数据进行与运算，还例如可以为对数据进行和运算等，本实施例对逻辑运算处理不进行特别限定，只要是和逻辑运算相关的处理都称为逻辑运算处理，之后得到逻辑运算处理后的数据，将逻辑运算处理之后的数据传送给DSP芯片101。
逻辑运算处理后的数据送到DSP芯片101，DSP芯片101对该数据进行控制运算处理，具体的处理方式为根据设定的控制算法，对FPGA芯片102传送来的逻辑运算处理之后的数据进行处理，控制运算处理是辅助变流器控制功能的核心，是指根据接收到的数据进行具体的数据处理以及相关判断，指示辅助变流器要进行何种操作，控制运算处理例如可以为电流电压控制、变换等，此处对控制运算处理不做特别限制，DSP芯片102进行控制运算处理后得到脉冲宽度，将得到的脉冲宽度送至FPGA芯片102。
FPGA芯片102接收到DSP芯片101传送的脉冲宽度，FPGA芯片102进行调制运算处理，所谓的调制运算处理指的是，利用相关调制算法将初始信号量转换为需要的目标信号量，在FPGA芯片102中利用调制算法对控制运算处理后的数据进行处理，调制算法例如可以为利用调制算法 将脉冲宽度转化为脉冲序列，本发明实施例对调制算法不作特别限制，进行调制运算处理之后，产生辅助变流器的驱动脉冲序列，驱动脉冲序列到达目标部件，目标部件例如可以为驱动板105，之后目标部件开始进行相关的操作，例如可以为驱动板105驱动相关管道的导通与关闭从而得到输出需要的电压，也就实现了辅助变流器的控制功能。部分逻辑运算处理和相关调制运算处理的功能由FPGA芯片102实现，可以为DSP芯片101完成其他方面的功能节约出更多的资源，更好地实现芯片资源分配，从而提高辅助变流器的工作效率。
本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备，包括：DSP芯片101以及FPGA芯片102，DSP芯片101与FPGA芯片102总线连接，其中FPGA芯片102用于通过模拟量采样板卡和数字量采样板卡获取辅助变流器的模拟量和数字量，并对模拟量和数字量进行逻辑运算处理，得到逻辑运算处理后的数据；DSP芯片101用于对逻辑运算处理后的数据进行控制运算处理，得到脉冲宽度；FPGA芯片102还用于根据脉冲宽度进行调制运算处理，得到所述辅助变流器的驱动脉冲序列。实现了辅助变流器的控制功能，并且采用DSP芯片101和FPGA芯片102共同完成辅助变流器的控制，弥补了单独的DSP芯片在实现控制功能时采样速度低和芯片资源分配不合理的缺陷。
在上述实施例的基础上，DSP芯片101的控制运算处理以及FPGA芯片102的调制运算处理包含多个处理操作，下面结合图40对DSP芯片101的控制运算处理和FPGA芯片的调制运算处理进行详细介绍。
图40为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的处理示意图，如图40所示，控制运算处理包括：Park变换处理202，电压、电流双闭环解耦控制处理203、IPark变换处理204以及零序电压注入处理205；调制运算处理包括：脉冲生成处理206；DSP芯片具体用于对逻辑运算处理201后的数据依次进行Park变换处理202，电压、电流双闭环解耦控制处理203、IPark变换处理204以及零序电压注入处理205，得到零序电压注入处理205后的数据；FPGA芯片具体用于对零序电压注入处理205后的数据进行脉冲生成处理206，得到脉冲生成处理处理205后的数据。
在本实施例中，DSP芯片首先接收FPGA芯片进行逻辑运算处理201之后的数据，此处以电压数据为例，在接收到电压数据之后，首先对电压数据进行Park变换处理201，Park变换处理201是指将三相交流电A相、B相、C相变换到旋转的dq坐标系下，其中三相交流电是指三个频率相同、电势振幅相等、相位差互差120度角的交流电路组成的电力系统，目前我国生产、配送的都是三相交流电，因此在本实施例中电压数据也是三相交流电，三相交流电的三相分别是A相、B相和C相，在图2中对应U A、U B、U C，因为三相交流电的数学方程非常复杂，并且很多控制参数在方程式中耦合在一起，想要实现单独的控制非常困难，因此需要进行Park变换处理202将三相交流电A、B、C相变换到旋转的dq坐标系下，如图2所示，三相交流电U A、U B、U C经过Park变换处理202得到U d和U q。
其中dq坐标系是通过如下的方式得到：因为三相交流电是三相对称的，那么就可以利用两相交流电来实现与三相交流电相同的磁场效应，将三相交流电投影到两相间隔为90度的两相交流电上，就实现了由三相静止交流电到两相静止交流电的变换，如果两相静止交流电以某个角速度旋转起来，那么久得到了旋转的dq坐标系，静止坐标系下的三相交流电转换到dq坐标系下变为直流量，直流量的控制简单并且互相之间不存在耦合，因此控制dq坐标系下的直流量就实现了三相交流电的解耦控制。
在park变换处理202之后得到dq坐标系下的直流量，然后对该直流量进行双闭环解耦控制处理203，其中双闭环控制，指的是电压外环和电流内环控制，它是通过闭环电压和闭环电流来进行控制，从而保持电流和电压恒定，在旋转的dp坐标系下实现双闭环控制也就是图2中的电压、电流双闭环解耦控制处理203，如图40所示，park变换处理202之后得到dq坐标系下的直流量U d和U q，经过电压、电流双闭环解耦控制处理203得到U dout和U qout，其中，U dout和U qout是恒定值，从而避免了辅助变流器的负载投切而引起的电压波动导致控制失效，并且静止坐标系下的三相交流电转换到dq坐标系下变为直流量，直流量之间不存在耦合，也就实现了解耦处理。
在电压、电流双闭环解耦控制处理203之后，需要再通过IPark变换处理204将旋转的dq坐标系下的电压变换为静止坐标系下的三相交流电，以便后续正常供电，如图2所示，U dout和U qout经过IPark变换处理204变为U p、V p和W p，得到IPark变换处理204的输出电压。
在得到IPark变换处理204的输出电压后，在DSP芯片中对IPark变换处理204的输出电压进行零序电压注入处理205，首先根据控制运算处理的输出电压U p、V p和W p计算得到零序电压，其中，在使用对称分量法对三相交流电进行分析时，三个相同大小相同相位的分量为零序电压。在计算得到零序电压之后，将零序电压与控制运算处理的输出电压U p、V p和W p结合，得到调制运算处理的输出电压，并生成脉冲，如图2所示，控制运算处理的输出电压U p、V p和W p经过零序电压注入处理205得到控制运算处理的输出电压，即脉冲宽度调制(Pulse Width Modulation，PWM)的脉冲宽度变量U out、V out和W out，得到控制运算处理的输出电压，也就是PWM的脉冲宽度变量后，将控制运算处理的输出电压发送给FPGA芯片。
FPGA芯片接收到DSP芯片传送的控制运算处理的输出电压PWM的脉冲宽度变量，进行调制运算处理，具体地，调制运算处理进行脉冲生成处理206，经过脉冲生成处理206后，电压有效值能够达到输入直流电压的大小，可以有效提高输入直流电压的利用率。
本发明实施例提供的控制运算处理包括：Park变换处理202，电压、电流双闭环解耦控制处理203、IPark变换处理204以及零序电压注入处理205；调制运算处理包括：脉冲生成处理206；DSP芯片具体用于对逻辑运算处理后的数据依次进行Park变换处理202，电压、电流双闭环解耦控制处理203、IPark变换处理204以及零序电压注入处理205，得到零序电压注入处理后的数据；FPGA芯片具体用于对零序电压注入处理后的数据进行脉冲生成处理206，得到脉冲生成处理后的数据。能够有效实现电压的解耦控制，并且在控制过程中电流电压恒定，从而避免了因电压波动导致的控制失效，零序电压注入处理能够提高输入直流电压的利用率。
DSP芯片和FPGA芯片除了应用于逻辑运算处理、控制运算处理和调制运算处理之外，还应用于故障检测中，下面结合图3对DSP芯片和 FPGA芯片进行故障检测进行详细介绍。
图41为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图二，如图41所示，FPGA芯片还用于根据采集到的辅助变流器的运行数据，确定辅助变流器中的部件是否发生故障；若FPGA芯片确定辅助变流器中的部件发生故障，则将故障信息发送给DSP芯片；若FPGA芯片确定辅助变流器中的部件未发生故障，则将运行数据发送至DSP芯片；DSP芯片还用于根据故障信息确定部件发生故障；DSP芯片还用根据运行数据判断辅助变流器内的部件是否发生故障。
FPGA芯片还具体用于根据采集到的辅助变流器的运行数据，在确定运行数据大于第一预设阈值时，确定部件发生故障。
DSP芯片还具体用于在确定运行数据大于第二预设阈值，并且故障持续时间大于预设时长时，确定部件发生故障，其中，第一预设阈值大于第二预设阈值。
可选地，可以在FPGA芯片中设置数据采集模块301、故障判断模块302；在DSP芯片中设置检测模块303、计数模块304以及标志位生成模块305。通过模块的方式来具体描述上述FPGA芯片和DSP芯片的具体实现过程。
具体地，数据采集模块301，用于采集得到辅助变流器的运行数据，故障判断模块302用于确定辅助变流器的部件是否发生故障，具体地，根据采集到的辅助变流器的运行数据与第一预设阈值进行比较，若运行数据大于第一预设阈值，则在FPGA芯片中确定对应的部件发生故障，在FPGA芯片中进行相应的故障处理，并将故障信息发送给DSP芯片，故障信息中包含检测到故障的信号，DSP芯片接收到故障信息确定辅助变流器的部件发生故障。
若根据采集到的辅助变流器的运行数据与第一预设阈值进行比较，运行数据小于或者等于第一预设阈值，则FPGA芯片确定辅助变流器中的部件未发生故障，则此时FPGA芯片不向DSP芯片发送故障信息，仅将运行数据发送给DSP芯片，需要说明的是，无论FPGA芯片是否确定发生故障，FPGA芯片都会将相关模拟量和数据量传送给DSP芯片。
DSP芯片接收到的FPGA芯片发送的运行数据，首先进入检测模块 303，具体地，检测模块303用于根据运行数据判断辅助变流器内的部件是否发生故障，具体地，判断该运行数据是否大于第二预设阈值，若该运行数据大于第二预设阈值，则进一步判断故障持续时间是否大于预设时长，若运行数据大于第二预设阈值，并且故障持续时间大于预设时长，则在DSP芯片中确定辅助变流器对应的部件发生故障。
其中，故障判断模块302中的第一预设阈值大于检测模块303中的第二预设阈值，因为数据采集在FPGA中完成，并且FPGA芯片的数据处理速度快，而DSP芯片数据处理的速度较慢，因此DSP芯片在设定的数据传输时间内无法接收到FPGA芯片传送的全部数据，设定第一预设阈值大于第二预设阈值的目的是，若发生了较为严重的故障，则FPGA芯片能快速检测到并且做出相应的故障操作处理，而DSP芯片的保护反应时间较长，保护阈值低并且设置有持续时间检测，可以确保故障检测的准确性，提高故障处理的效率。
采用两级故障检测机制，FPGA芯片可以快速检测到重大故障并且进行逻辑保护动作，DSP芯片中的故障检测可以确保故障检测的准确性，从而保证辅助变流器的安全运行，提高系统的可靠性。
DSP芯片还用于在辅助变流器多次重启后，确定部件的故障次数；
DSP芯片还用于判断故障次数是否大于预设次数，若是，则确定故障为永久故障，若否，则确定故障为警告故障。
进一步地，DSP芯片中的计数模块304用于在确定部件发生故障之后，进行多次重启操作，并且利用DSP芯片内部设置的计数器对重启的次数进行记录，从而确定部件的故障次数，然后判断故障次数是否大于预设次数，若故障次数大于预设次数，则确定该部件的故障为永久故障，也就是说即使进行了多次重启也无法解决该故障，若故障次数小于或者预设次数，则确定该故障为警告故障，即判定该故障在经过多次重启后可以被修复，并且多次重启操作之后警告故障被修复，为辅助变流器进行其他的操作节省了资源。
DSP芯片还用于在确定部件的故障为永久故障时，将部件对应的故障标志位设置为永久故障标志位。
DSP芯片还用于将多个故障标志位进行拼接处理，并将拼接处理后 的故障标志位发送给上位机。
在上述实施例的基础上，标志位生成模块305，用于确定部件的故障为永久故障时，将部件对应的故障标志位设置为永久故障标志位，相应的，在确定部件的故障为警告故障时，将部件对应的故障标志位设置为警告故障标志位，对于永久故障标志位和警告故障标志位的处理方式不相同，并且DSP芯片将所有故障标志位进行拼接处理，并将拼接后的故障标志位发送给上位机，其中上位机是指可以直接发出操控命令的计算机，在本发明实施例中，上位机与DSP芯片连接，本发明实施例的故障反馈模块306用于在上位机接收到DSP芯片发送的故障标志位之后，进行相应地故障处理。
通过标识永久故障标志位与警告故障标志位，并将故障标志位进行拼接处理发送给上位机，可以提高故障处理的效率。
在DSP芯片需要将故障标志位发送给上位机时，需要与上位机进行通信，下面结合图42对DSP芯片与上位机的通信方式进行说明。
图42为本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备的结构示意图三，如图42所示，上位机404与DSP芯片403通过CAN总线通信。
其中DSP芯片403需要与机车控制单元401通信，DSP芯片403和机车控制单元401之间的通信采用的是多功能车辆总线(Multifunction Vehicle Bus，MVB)通信，因此在辅助变流器的控制设备中配备有MVB板卡402，而DSP芯片403与MVB板卡402的通信采用控制器局域网络(Controller Area Network，CAN)总线通信，CAN总线是一种有效支持分布式控制或实时控制的串行通信总线，并且DSP芯片403与上位机404的通信同样采用CAN总线通信，可以有效实现通信并且提高通信效率。
本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备，通过设置上位机与DSP芯片通过CAN总线通信，有效实现了控制设备和机车控制单元以及与上位机的通信，提高了通信效率。
上述介绍的DSP芯片以及FPGA芯片在进行相关操作之后，例如在确定辅助变流器发生故障之后，需要对数据进行存储，下面结合具体的实施例对数据存储进行介绍。
本发明实施例提供的直驱永磁电力机车辅助变流器的控制设备还包括：闪存；DSP芯片内设置有随机存取存储器(random access memory，RAM)空间；闪存与DSP芯片连接；
DSP芯片还用于将逻辑运算处理后的数据存储至RAM空间；
DSP芯片还用在确定部件发生故障，将RAM空间中的数据存储至闪存中。
具体地，闪存是一种非易失性的存储器，即使断电数据也不会丢失，在辅助变流器的控制设备中设置有闪存，并且闪存与DSP芯片连接，在DSP芯片内部还设置有数组空间，辅助变流器的工作过程中对于FPGA芯片采集的数据进行相关逻辑运算处理后，将逻辑运算处理后数据存储在DSP芯片内设置的数组空间中。
在DSP芯片确定故障发生后，调用相关函数将数组空间中的数据存储至闪存中，用于数据存储的闪存有较大的存储空间，可以满足较高频率的数据采集以及数据存储的需要，并且在后期维护过程中，从闪存中读取故障发生时的数据即可，提高辅助变流器的工作效率。
本领域普通技术人员可以理解：实现上述各方法实施例的全部或部分步骤可以通过程序指令相关的硬件来完成。前述的程序可以存储于一计算机可读取存储介质中。该程序在执行时，执行包括上述各方法实施例的步骤；而前述的存储介质包括：ROM、RAM、磁碟或者光盘等各种可以存储程序代码的介质。
最后应说明的是：以上各实施例仅用以说明本发明的技术方案，而非对其限制；尽管参照前述各实施例对本发明进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分或者全部技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本发明各实施例技术方案的范围。

Claims (25)

  1. 一种电力机车用大功率直驱永磁电传动系统，用于控制使用直驱永磁同步电机的电力机车，所述电力机车包括三台直驱永磁同步电机；其特征在于，所述电力机车用大功率直驱永磁电传动系统包括：第一四象限整流器、第二四象限整流器、中间直流回路、第一逆变模块、第二逆变模块、第三逆变模块和辅助变流器，所述第一四象限整流器和所述第二四象限整流器均连接所述电力机车的主变压器和所述中间直流回路，所述中间直流回路分别连接所述第一逆变模块、第二逆变模块、第三逆变模块和所述辅助变流器；所述电力机车用大功率直驱永磁电传动系统用于：
    通过所述第一四象限整流器和所述第二四象限整流器将所述主变压器的交流电转换为直流电后输出至所述中间直流回路；
    通过所述中间直流回路将接收到的直流电分别输出至所述第一逆变模块、第二逆变模块、第三逆变模块和所述辅助变流器；
    通过所述第一逆变模块、第二逆变模块和第三逆变模块将接收到的直流电转换为三相交流电后分别输出至所述三台直驱永磁同步电机；
    通过所述辅助变流器将接收到的直流电转换为三相交流电后输出至所述电力机车的辅助负载。
  2. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，对于所述第一四象限整流器和所述第二四象限整流器中的任一四象限整流器，所述通过所述第一四象限整流器和所述第二四象限整流器将所述主变压器的交流电转换为直流电后输出至所述中间直流回路，具体包括：
    对输入所述四象限整流器的交流电流进行采样，得到采样周期内的交流电流，所述交流电流包括正半周期的电流值和负半周期的电流值；其中，根据预设采样频率，对输入所述四象限整流器的交流电流进行采样，得到多个采样点，将得到的多个采样点绘制成曲线，得到一个正弦或者余弦曲线；所述预设采样频率为IGBT通断频率的N倍，所述N≥2；
    获取正半周期的电流值的第一和值与负半周期的电流值的第二和值，并根据所述第一和值和所述第二和值，获取电流偏置值；其中，将正半周期的多个采样点的值进行加和得到第一和值P，再将负半周期的多个采样点的值进行加和得到第二和值N，P值与N值的绝对值进行做差计算，所得到 的差值为Q；
    将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值；其中，直流偏置值Q与零输入至第一PI控制器，第一PI控制器根据直流偏置值Q与零构成控制偏差，将偏差的比例和积分通过线性组合构成控制量，对交流电流进行控制，消除交流电流的直流偏置，控制量即为第一输出值；
    根据所述第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，所述PR控制器用于对所述交流电流进行无静差控制，使所述交流电流的周期和相位与电网电压相同；其中，将交流电流输入到PR控制器，保证交流电流的相位和和周期与电网电压相同后，得到稳定的输出交流电流，即为第二输出值；
    根据所述脉冲宽度调制符号控制所述四象限整流器中的绝缘栅双极型晶体管IGBT的通断，以控制所述四象限整流器将所述主变压器的交流电转换为直流电。
  3. 根据权利要求2所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述对所述输入四象限整流器的交流电流进行采样，得到采样周期内的交流电流之前，还包括：
    获取所述四象限整流器的直流母线电压与指令电压的第二差值；
    将所述第二差值输入至第二PI控制器，使得所述第二PI控制器输出的第三输出值与锁相环输出值相乘，得到与所述电网电压同相位的交流电流，所述锁相环用于控制所述交流电流的周期与相位和所述电网电压的周期与相位保持一致；
    所述对输入所述四象限整流器的交流电流进行采样，得到采样周期内的交流电流，包括：
    根据预设采样频率对输入四象限整流器的交流电流进行采样，得到采样电流，所述预设采样频率为所述IGBT通断频率的两倍；
    根据所述锁相环确定的电网电压相位和所述采样电流，得到采样周期内的交流电流；
    所述根据所述锁相环确定的电网电压相位和所述采样电流，得到采样周期内的交流电流之前，还包括：
    通过第一带通滤波器和第二带通滤波器对所述采样电流进行滤波，得到滤波后的采样电流；其中，所述第一带通滤波器用于获取交流电流的主频信号，所述第二带通滤波器用于滤除干扰谐波。
  4. 根据权利要求2所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述将所述电流偏置值与零的第一差值输入至第一PI控制器，获取所述第一PI控制器输出的第一输出值之前，还包括：
    判断所述第一差值的绝对值是否大于所述电流环宽的绝对值，得到的判断结果为是；
    所述根据所述第一输出值以及PR控制器输出的第二输出值得到脉冲宽度调制符号，包括：
    对所述第一输出值和所述第二输出值进行求和，得到第三和值，所述第一输出值为电流变量，所述第二输出值为电流值；
    根据所述第三和值和单极倍频脉冲调制方式，得到所述脉冲宽度调制符号。
  5. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，
    所述电力机车用大功率直驱永磁电传动系统还包括：第一斩波模块和第二斩波模块，所述第一斩波模块连接所述第一四象限整流器和所述中间直流回路，所述第二斩波模块连接所述第二四象限整流器和所述中间直流回路；
    所述电力机车用大功率直驱永磁电传动系统还用于：
    通过第一斩波模块和第二斩波模块分别将所述第一四象限整流器和所述第二四象限整流器输出的直流电进行斩波处理后输出至所述中间直流回路；
    具体地，对于所述第一斩波模块和所述第二斩波模块中的任一斩波模块，所述电力机车用大功率直驱永磁电传动系统还用于：
    对中间直流母线电压进行周期性检测，所述中间直流母线电压为所述电力机车上直流母线上的电压；
    当检测到的中间直流母线电压值大于斩波上限阈值时，采用P调节器对所述中间直流母线电压进行调节，直至检测到的所述中间直流母线电压 值小于斩波下限阈值，所述斩波上限阈值大于所述斩波下限阈值；其中，所述P调节器的原理为：控制斩波管在检测周期的特定时间比例内处于开通状态。
  6. 根据权利要求5所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述采用P调节器对所述中间直流母线电压进行调节，包括：
    采用所述P调节器，确定目标检测周期内的斩波占空比；所述目标检测周期包括：从检测到的中间直流母线电压值大于斩波上限阈值，到检测到的中间直流母线电压值小于斩波下限阈值之间的经历的检测周期；
    根据所述斩波占空比，确定斩波管在目标检测周期内的开通时间；
    根据所述开通时间，控制所述斩波管的开通或关断，以使所述中间直流母线电压值下降至小于所述斩波下限阈值；
    所述控制方法还包括：
    当检测到中间直流母线电压值小于所述斩波下限阈值时，控制斩波管关断。
  7. 根据权利要求6所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述采用所述P调节器，确定目标检测周期内的斩波占空比之前，还包括：
    根据以下公式确定目标参数；
    Err＝U1-斩波下限阈值
    其中，Err表示目标参数，U1表示目标检测周期内检测到的中间直流母线电压值；
    相应的，所述采用所述P调节器，确定目标检测周期内的斩波占空比，包括：
    获取所述P调节器对应的控制系数；
    根据所述控制系数和所述目标参数，确定所述斩波占空比；
    所述获取所述P调节器的控制系数，包括：
    根据如下公式确定所述控制系数；
    Kp_chp＝1/(直流母线电压过压保护值阈值-斩波下限阈值)
    其中，Kp_chp表示控制系数；
    所述根据所述控制系数和所述目标参数，确定所述斩波占空比，包括：
    根据如下公式确定所述斩波占空比；
    C_duty＝Err*Kp_chp
    其中，C_duty表示斩波占空比，Err表示目标参数，Kp_chp表示控制系数；
    所述根据所述斩波占空比，确定斩波管在目标检测周期内的开通时间之前，还包括：
    对所述斩波占空比进行防错处理；
    其中，所述对所述斩波占空比进行防错处理，包括：
    若所述斩波占空比的值大于1，则将所述斩波占空比的值设为1；
    若所述斩波占空比的值小于0，则将所述斩波占空比的值设为0。
  8. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，还包括：
    确定待控制直驱永磁同步电机的转速；
    根据所述转速与第一映射关系确定第一控制策略，所述第一映射关系包括至少一个转速的范围和至少一个控制策略的一一对应关系；
    根据所述第一控制策略确定所述待控制直驱永磁同步电机的预期控制相角。
  9. 根据权利要求8所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述第一映射关系包括：额定转速以下的转速与MTPA控制策略的对应关系；
    额定转速以上的转速与弱磁控制策略的对应关系。
  10. 根据权利要求9所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述MTPA控制策略包括：
    根据转矩电流曲线确定q轴电流给定和d轴电流给定；
    计算所述q轴电流给定与q轴实际电流的第一差值和所述d轴电流给定与d轴实际电流的第二差值；
    通过第一PI控制器根据所述第一差值得到d轴电压给定、通过第二PI控制器根据所述第二差值得到q轴电压给定；
    计算所述q轴电压给定与q轴前馈电压之和得到实际q轴电压给定、计算所述d轴电压给定与d轴前馈电压之和得到实际d轴电压给定；其中，所述前馈电压可通过如下前馈解耦的闭环传递函数矩阵计算：
    Figure PCTCN2018117085-appb-100001
    其中，所述前馈解耦的闭环传递函数通过如下前馈解耦的电压计算方程得到：
    Figure PCTCN2018117085-appb-100002
  11. 根据权利要求9所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述弱磁控制策略包括：
    通过PI控制器根据电压极限值与前馈电压幅值之差计算给定弱磁状态下d轴电流变化量；
    通过给定弱磁状态下d轴电流变化量和d轴电流给定之和得到弱磁调节后的d轴电流给定；
    根据所述d轴电流给定和转矩公式计算弱磁调节后的q轴电流给定；
    通过PI控制器根据所述q轴电流给定与q轴实际电流之差得到功角β；
    通过如下公式计算实际q轴电压给定和实际d轴电压给定；
    U d＝U scosβ
    U q＝U scosβ
    其中，Us为电压极限值，Ud为实际d轴电压给定，Uq为实际q轴电压给定。
  12. 根据权利要求9-11任一项所述的电力机车用大功率直驱永磁电传动系统，其特征在于，还包括：
    当控制策略从所述MTPA控制策略切换至所述弱磁控制策略时，将切换瞬间MTPA控制策略中的电压矢量角度作为所述弱磁控制策略中初始功角β；
    当控制策略从所述弱磁控制策略切换至所述MTPA控制策略时，通过切换瞬间弱磁控制策略中的最后一拍功角β通过公式
    Figure PCTCN2018117085-appb-100003
    计算出MTPA控制策略中的实际q轴电压给定和实际d轴电压给定。
  13. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，还包括：
    获取待调制直驱永磁同步电机的调制波的频率；
    根据所述调制波的频率所在范围与第二映射关系确定第一调制策略，所述第二映射关系包括至少一个调制波的频率范围和至少一个调制策略的一一对应关系；
    根据所述第一调制策略确定所述直驱永磁同步电机的PWM载波频率。
  14. 根据权利要求13所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述第二映射关系包括：
    调制波的频率为低速阶段时对应异步调制策略；
    调制波的频率大于低速阶段低于高速阶段时对应同步调制策略；
    调制波的频率为高速阶段时对应方波调制策略。
  15. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，
    向所述直驱永磁同步电机的定子绕组注入高频电压信号，获取三相定子绕组电流；
    根据所述三相定子绕组电流获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流；
    根据所述d轴目标电流和所述q轴目标电流获取转子的初始位置角，其中，所述初始位置角为根据所述直驱永磁同步电机的磁极极性进行补偿后的初始位置角。
  16. 根据权利要求15所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述根据所述d轴目标电流和所述q轴目标电流获取转子的初始位置角，包括：
    根据所述q轴目标电流获取转子的第一初始位置角；
    根据所述d轴目标电流获取转子的磁极补偿角；
    根据所述第一初始位置角以及所述磁极补偿角，获取所述转子的初始位置角；
    所述根据所述q轴目标电流获取转子的第一初始位置角，包括：
    对所述q轴目标电流进行低通滤波处理，获取误差输入信号；
    根据所述误差输入信号，获取所述第一初始位置角；
    所述对所述q轴目标电流进行低通滤波处理，获取误差输入信号，包括：
    采用调制信号对所述q轴目标电流进行调制，获取调制后的q轴目标电流；
    对所述调制后的q轴目标电流进行低通滤波处理，获取所述误差输入信号；
    所述根据所述误差输入信号，获取所述第一初始位置角，包括：
    根据所述输入误差信号获取所述误差输入信号的比例偏差和积分偏差；
    根据所述比例偏差和所述积分偏差的线性组合，获取所述第一初始位置角；
    所述根据所述d轴目标电流获取转子的磁极补偿角，包括：
    向所述永磁同步电机注入多个电压幅值相等、角度不同的电压脉冲信号，获取每个所述电压脉冲信号的响应电流；
    根据多个所述响应电流，确定所述转子的磁极补偿角；
    所述根据多个所述响应电流，确定所述转子的磁极补偿角，包括：
    当注入的所述电压脉冲信号的角度与所述第一初始位置角之差满足预设误差范围，所述电压脉冲信号的响应电流的幅值大于第一值，则确定所述转子的磁极补偿角为0，所述第一值为多个所述响应电流的幅值的最大值；
    当注入的所述电压脉冲信号的角度与所述第一初始位置角之差满足预设误差范围，所述电压脉冲信号的响应电流的幅值小于第二值，则确定所述转子的磁极补偿角为π，所述第二值为多个所述响应电流的幅值的最小值。
  17. 根据权利要求16所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述高频电压信号为：
    Figure PCTCN2018117085-appb-100004
    其中，U mh为高频电压信号的幅值，ω h为高频电压信号的角频率，t为注入高频电压信号的时间；
    所述根据三相定子绕组电流获取预期两相同步旋转坐标系下的d轴目标电流和q轴目标电流，通过如下公式计算：
    Figure PCTCN2018117085-appb-100005
    其中，L为平均电感L＝(L d+L q)/2，△L为半差电感△L＝(L d-L q)/2；
    所述对所述q轴目标电流进行低通滤波处理，获取误差输入信号，通过如下公式计算：
    Figure PCTCN2018117085-appb-100006
    其中，LPF表示低通滤波；当转子位置估计误差足够小，极限等效线性化后该误差输入信号为：
    Figure PCTCN2018117085-appb-100007
    所述获取第一初始位置角，通过以下公式计算：
    Figure PCTCN2018117085-appb-100008
    其中，s表示拉普拉斯算子，k p为比例项系数，k i为积分项系数。
  18. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，还包括：
    根据控制中断周期、调制载波周期，以及所述直驱永磁同步电机的转子当前角速度，获取所述直驱永磁同步电机的转子的补偿相角；
    根据所述补偿相角，获取当前实际控制相角；
    根据当前d轴电压给定值和当前q轴电压给定值，获取当前预期控制相角；
    根据所述当前预期控制相角与所述当前实际控制相角的比例偏差和积分偏差，对所述当前实际控制相角进行在线修正。
  19. 根据权利要求18所述的电力机车用大功率直驱永磁电传动系 统，其特征在于，所述根据控制中断周期、调制载波周期，以及所述直驱永磁同步电机的转子的当前角速度，获取所述直驱永磁同步电机的转子的补偿相角，包括：
    根据所述控制中断周期和所述直驱永磁同步电机的转子的当前角速度，获取第一子补偿相角；
    根据所述调制载波周期和所述直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角；
    根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角；
    根据所述第一子补偿相角、所述第二子补偿相角和所述第三子补偿相角，获取所述直驱永磁同步电机的补偿相角；
    所述根据所述控制中断周期和所述直驱永磁同步电机的转子的当前角速度，获取第一子补偿相角，包括：
    根据所述控制中断周期，获取第一子补偿相角对应的第一相角时延；
    根据所述第一相角时延和所述直驱永磁同步电机的转子的当前角速度，获取所述第一子补偿相角；
    所述根据所述调制载波周期和所述直驱永磁同步电机的转子的当前角速度，获取第二子补偿相角，包括：
    根据所述调制载波周期，获取调制输出对应的第二相角时延；
    根据调制算法的调制中断周期，获取调制计算对应的第三相角时延；
    根据所述第二相角时延、所述第三相角时延和所述直驱永磁同步电机的转子的当前角速度，获取所述第二子补偿相角；
    所述根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角之前，还包括：
    根据所述直驱永磁同步电机的矢量控制策略，获取所述直驱永磁同步电机的稳定运行角速度范围；
    根据d轴电流给定值和q轴电流给定值，获取所述稳定运行角速度范围内的多个第一d轴电流、多个第一q轴电流、每个所述第一d轴电流对 应的d轴电压以及每个所述第一q轴电流对应的q轴电压；
    所述根据所述直驱永磁同步电机的转子当前角速度，获取第三子补偿相角，包括：
    根据每个所述第一d轴电流对应的d轴电压以及每个所述第一q轴电流对应的q轴电压，获取每个第一角速度对应的传输误差相角；
    根据每个所述第一角速度对应的传输误差相角，以及，所述直驱永磁同步电机的转子的当前角速度，获取所述第三子补偿相角；
    所述根据所述补偿相角，获取当前实际控制相角，包括：
    获取所述直驱永磁同步电机的转子的当前位置相角；
    根据所述当前位置相角、所述转子的初始位置相角以及所述补偿相角，获取所述转子的实际位置相角；
    根据所述转子的实际位置相角以及调制相角，获取当前实际控制相角，其中，所述调制相角为根据d轴电压给定值和当前q轴电压给定值经过调制算法计算得到；
    所述根据所述当前预期控制相角与所述当前实际控制相角的比例偏差和积分偏差，对所述当前实际控制相角进行在线修正，包括：
    根据所述当前预期控制相角与所述当前实际控制相角获取所述比例偏差、所述积分偏差；
    根据所述比例偏差以及所述积分偏差的线性组合，获取当前实际控制相角的修正项；
    根据所述修正项对所述当前实际控制相角进行在线修正。
  20. 根据权利要求19所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述获取第一子补偿相角，通过如下公式计算：
    θ cmps1＝Δ t1·ω
    其中，ω为直驱永磁同步电机的转子的当前角速度，Δ t1为第一相角时延，第一相角时延Δ t1通过如下公式计算：
    Δ t1＝A·T ctrl≈0.5T ctrl
    其中，T ctrl为控制算法的一个控制中断周期；
    所述获取第二子补偿相角，通过如下公式计算：
    θ cmps2＝Δ t2·ω
    其中，ω为直驱永磁同步电机的转子的当前角速度，Δ t2为PWM脉冲输出过程中的时延，PWM脉冲输出过程中的时延Δ t2通过如下公式计算：
    Δ t2＝B·T PWM+C·T PWM≈0.75T PWM
    其中，T PWM为PWM的调制载波周期，B为调制算法中断时延系数，C为PWM脉冲输出时延系数；
    所述获取当前预期控制相角，通过如下公式计算：
    Figure PCTCN2018117085-appb-100009
    其中，θ ctrl表示预期控制相角，
    Figure PCTCN2018117085-appb-100010
    表示q轴电压给定值，
    Figure PCTCN2018117085-appb-100011
    表示d轴电压给定值；
    所述对所述当前实际控制相角进行在线修正，通过如下公式计算：
    Figure PCTCN2018117085-appb-100012
    其中，k p和k i为修正项，θ ctrl为当前预期相角，θ PWM为当前实际相角，f Δ为基波频率补偿项；
    所述获取所述直驱永磁同步电机的稳定运行角速度范围，通过如下公式计算：
    Figure PCTCN2018117085-appb-100013
    其中，u d为任一第一预设角速度对应的d轴电压，u q为任一第一预设角速度对应的q轴电压，R s为转子的电阻，L q为任一第一预设角速度对应的d轴电感，L d为任一第一预设角速度对应的q轴电感，i d为d轴电压对应的第一d轴电流，i q为q轴电压对应的第一q轴电流，ψ f为永磁体磁链的反电势；
    所述获取传输误差相角θ Δ，通过如下公式计算：
    θ Δ＝tan -1(u d/u q)
    所述获取第三子补偿相角θ cmps3，通过如下公式计算：
    θ cmps3＝k·ω。
  21. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，
    所述电力机车还包括：至少四个直驱永磁同步电机；所述至少四个直驱 永磁同步电机包括：第一电机、第二电机、第三电机和第四电机；
    所述电力机车用大功率直驱永磁电传动系统还用于：
    采集第一电机、第二电机、第三电机和第四电机的转子频率，获取所述第一电机的实时转矩，所述第一电机和所述第二电机为第一转向架的轴电机，所述第三电机和所述第四电机为第二转向架的轴电机，所述第一转向架与所述第二转向架相邻；
    根据所采集的多个电机的转子频率，确定所述第一电机的转子频率差和转子频率微分值；
    根据所述第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量；
    根据所述转矩削减量对所述第一电机的转矩进行调整。
  22. 根据权利要求21所述的电力机车用大功率直驱永磁电传动系统，其特征在于，还包括：
    根据所述第一电机的转子频率差、转子频率微分值和实时转矩，生成撒砂控制信号，所述撒砂控制信号用于指示是否进行撒砂操作；
    所述根据所述第一电机的转子频率差、转子频率微分值和实时转矩，确定转矩削减量，包括：
    根据所述第一电机的转子频率差以及预设的转子频率差分级规则，确定所述第一电机的转子频率差对应的空转滑行等级；
    根据所述第一电机的转子频率差对应的空转滑行等级，以及所述第一电机的实时转矩，确定第一转矩削减量；
    根据所述第一电机的转子频率微分值以及预设的转子频率微分值分级规则，确定所述第一电机的转子频率微分值对应的空转滑行等级；
    根据所述第一电机的转子频率微分值对应的空转滑行等级，以及所述第一电机的实时转矩，确定第二转矩削减量；
    若所述第一转矩削减量大于等于所述第二转矩削减量，则确定所述第一转矩削减量为所述转矩削减量；
    若所述第一转矩削减量小于所述第二转矩削减量，则确定所述第二转矩削减量为所述转矩削减量；
    所述根据所述转矩削减量对所述第一电机的转矩进行调整，包括：
    在第一预设时间段内，将所述第一电机的转矩值由第一值降低至第二值，所述第一值与所述第二值的差值为所述转矩削减量；
    在第二预设时间段内，保持所述第一电机的转矩值为所述第二值不变；
    在第三预设时间段内，将所述第一电机的转矩值由所述第二值提高至预设转矩值的预设百分比；
    在第四预设时间段内，将所述第一电机的转矩值提高至所述预设转矩值；
    其中，所述第一电机的转矩值在所述第三预设时间段内的恢复速率，大于所述第一电机的转矩值在所述第四预设时间段内的恢复速率；
    所述在第一预设时间段内，将所述第一电机的转矩值由第一值降低至第二值，包括：
    在第一预设时间段内，根据所述第一电机的转矩值的降低速率逐渐减小，将所述第一电机的转矩值由第一值降低至第二值；
    根据所采集的多个电机的转子频率，确定所述第一电机的转子频率差和转子频率微分值，包括：
    对所采集的多个电机的转子频率进行限幅滤波和低通滤波处理；
    根据限幅滤波和低通滤波处理后的所述多个电机的转子频率，确定所述第一电机的转子频率差和转子频率微分值；
    若机车处于惰行工况，则所述对所采集的多个转子频率进行限幅滤波和低通滤波处理，包括：
    获取所述第一电机的电流值；
    根据所述第一电机的电流值和每个电机的转子频率，确定所述每个电机的转子频率补偿系数；
    根据所述每个电机的转子频率补偿系数对所述每个电机的转子频率进行补偿；
    对补偿后的所述多个电机的转子频率进行限幅滤波和低通滤波处理。
  23. 根据权利要求1所述的电力机车用大功率直驱永磁电传动系统，其特征在于，
    所述电力机车用大功率直驱永磁电传动系统还包括：数字信号处理DSP芯片以及现场可编程门阵列FPGA芯片，所述DSP芯片与所述FPGA芯片总线连接；
    所述电力机车用大功率直驱永磁电传动系统还用于：
    通过所述FPGA芯片通过模拟量采样板卡和数字量采样板卡获取辅助变流器的模拟量和数字量，并对所述模拟量和数字量进行逻辑运算处理，得到逻辑运算处理后的数据；
    通过所述DSP芯片对所述逻辑运算处理后的数据进行控制运算处理，得到脉冲宽度；
    通过所述FPGA芯片根据所述脉冲宽度进行调制运算处理，得到所述辅助变流器的驱动脉冲序列，以控制所述辅助变流器将所述中间直流回路的直流电转换为交流电。
  24. 根据权利要求23所述的电力机车用大功率直驱永磁电传动系统，其特征在于，所述控制运算处理包括：Park变换处理，电压、电流双闭环解耦控制处理、IPark变换处理以及零序电压注入处理；
    所述调制运算处理包括：脉冲生成处理；
    所述DSP芯片具体用于对所述逻辑运算处理后的数据依次进行Park变换处理，电压、电流双闭环解耦控制处理、IPark变换处理以及零序电压注入处理，得到零序电压注入处理后数据；
    所述FPGA芯片具体用于对所述零序电压注入处理后的数据进行脉冲生成处理，得到脉冲生成处理后的数据；
    所述FPGA芯片还用于根据采集到的所述辅助变流器的运行数据，确定所述辅助变流器中的部件是否发生故障；
    若所述FPGA芯片确定所述辅助变流器中的部件发生故障，则将故障信息发送给所述DSP芯片；
    若所述FPGA芯片确定所述辅助变流器中的部件未发生故障，则将所述运行数据发送至所述DSP芯片；
    所述DSP芯片还用于根据所述故障信息确定所述部件发生故障；
    所述DSP芯片还用根据所述运行数据判断所述辅助变流器内的部件是否发生故障。
  25. 根据权利要求24所述的电力机车用大功率直驱永磁电传动系统，其特征在于，
    所述FPGA芯片还具体用于根据采集到的所述辅助变流器的运行数据， 在确定运行数据大于第一预设阈值时，确定所述部件发生故障；
    所述DSP芯片还具体用于在确定所述运行数据大于第二预设阈值，并且故障持续时间大于预设时长时，确定所述部件发生故障，其中，所述第一预设阈值大于所述第二预设阈值；
    所述DSP芯片还用于在所述辅助变流器多次重启后，确定所述部件的故障次数；
    所述DSP芯片还用于判断所述故障次数是否大于预设次数，若是，则确定所述故障为永久故障，若否，则确定所述故障为警告故障；
    所述DSP芯片还用于在确定所述部件的故障为永久故障时，将所述部件对应的故障标志位设置为永久故障标志位；
    还包括：上位机，所述上位机与所述DSP芯片连接；
    所述DSP芯片还用于将多个故障标志位进行拼接处理，并将拼接处理后的故障标志位发送给所述上位机；
    所述上位机与所述DSP芯片通过CAN总线通信；
    还包括：闪存；所述DSP芯片内设置有RAM空间；所述闪存与所述DSP芯片连接；
    所述DSP芯片还用于将所述逻辑运算处理后的数据存储至所述RAM空间；
    所述DSP芯片还用在确定所述部件发生故障，将所述RAM空间中的数据存储至所述闪存中。
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Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111831225A (zh) * 2020-07-05 2020-10-27 中车永济电机有限公司 具有数据存储及远程传输扩展设计的辅助微机测控装置
CN112100945A (zh) * 2020-08-05 2020-12-18 苏州汇川联合动力系统有限公司 对地短路检测方法、电机控制器及计算机可读存储介质
CN114285303A (zh) * 2021-11-11 2022-04-05 中车永济电机有限公司 一种分段控制的四象限变流器
CN115441787A (zh) * 2022-09-30 2022-12-06 东风商用车有限公司 一种电机前馈解耦控制方法
CN116176530A (zh) * 2022-12-23 2023-05-30 吉林大学 一种集成式线控液压制动系统液压力控制方法
EP4322394A1 (en) * 2022-08-04 2024-02-14 Milwaukee Electric Tool Corporation Power tool including current-based field weakening
CN117749025A (zh) * 2023-12-15 2024-03-22 荆州市三焱火炉金属制品有限公司 宽转速自适应调节的永磁同步电机控制方法
CN117749025B (zh) * 2023-12-15 2024-06-07 上海能环实业有限公司 宽转速自适应调节的永磁同步电机控制方法

Families Citing this family (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111740650B (zh) * 2020-07-07 2022-06-10 深圳市兆威机电股份有限公司 电机同步控制方法、装置、控制器、系统及存储介质
CN111923742A (zh) * 2020-07-20 2020-11-13 中车永济电机有限公司 直驱永磁牵引电传动系统
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CN112713829A (zh) * 2020-12-15 2021-04-27 中车永济电机有限公司 一种基于安全导向的轨道交通永磁牵引系统
CN112713749A (zh) * 2020-12-18 2021-04-27 中车永济电机有限公司 一种机车变流器上用于安装矩形连接器的异型支撑架
CN112976838B (zh) * 2021-02-02 2022-02-18 昆山大世界油墨涂料有限公司 喷墨印刷机纸张传输机构的迭代学习速度同步控制方法
CN116979850B (zh) * 2023-09-25 2023-11-28 苏州利氪科技有限公司 电机转动控制方法及装置
CN117270487B (zh) * 2023-11-17 2024-01-23 北京芯驰半导体科技有限公司 模拟信号的采样控制系统、方法及芯片

Citations (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH11136807A (ja) * 1997-10-31 1999-05-21 Hitachi Ltd 内燃機関付車両の制御方式
CN103296897A (zh) * 2012-03-05 2013-09-11 永济新时速电机电器有限责任公司 电力机车牵引变流器及电传动系统
CN105720831A (zh) * 2014-12-02 2016-06-29 永济新时速电机电器有限责任公司 带双电压传感器母线电压检测电路的牵引变流器
CN105720893A (zh) * 2014-12-02 2016-06-29 永济新时速电机电器有限责任公司 带二次滤波电路的牵引变流器
CN105790597A (zh) * 2016-03-29 2016-07-20 中车永济电机有限公司 一种动车用永磁同步电机牵引变流器主电路
CN105811814A (zh) * 2016-03-29 2016-07-27 中车永济电机有限公司 一种高速动车组牵引变流器主电路
CN105835703A (zh) * 2016-03-29 2016-08-10 中车永济电机有限公司 一种地铁用牵引变流器主电路
CN108696149A (zh) * 2018-05-25 2018-10-23 中车青岛四方车辆研究所有限公司 牵引变流器及其控制、故障处理和载波移相方法

Family Cites Families (13)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP2194643B1 (en) * 2007-09-25 2014-03-05 Mitsubishi Electric Corporation Controller for electric motor
CN101630938A (zh) * 2009-07-28 2010-01-20 哈尔滨工业大学 无位置传感器永磁同步电机转子初始位置辨识方法
CN102386836B (zh) * 2010-08-27 2014-04-23 永济新时速电机电器有限责任公司 永磁同步电机传动控制方法及装置
JP5243651B2 (ja) * 2011-10-12 2013-07-24 ファナック株式会社 永久磁石同期電動機のｄ軸電流を制御するモータ制御装置
CN202918232U (zh) * 2012-08-01 2013-05-01 北京海斯德电机技术有限公司 一种高速无刷直流电动机软起动控制系统
CN103560540A (zh) * 2013-11-18 2014-02-05 国家电网公司 基于三电平变换器直驱式风电机组低电压穿越控制系统
CN103684027B (zh) * 2013-11-22 2016-05-18 中南大学 基于纹波功率转移的单相光伏并网逆变器及调制控制方法
CN103684179B (zh) * 2013-12-17 2017-01-18 清华大学 一种永磁同步电机电流滤波及死区补偿装置与补偿方法
CN204374697U (zh) * 2015-01-26 2015-06-03 北京纵横机电技术开发公司 辅助变流器控制设备
CN105449639B (zh) * 2015-12-28 2018-05-25 广东威灵电机制造有限公司 基于永磁同步电机驱动器的电压保护控制方法及装置
CN105811836B (zh) * 2016-05-30 2018-03-16 中国铁路总公司 一种大功率表面式永磁同步电机优化控制方法
CN107171535B (zh) * 2017-05-17 2024-04-16 中国铁道科学研究院 一种分布式控制器
CN108390608A (zh) * 2018-04-20 2018-08-10 哈尔滨理工大学 一种具有谐波抑制功能的无位置传感器永磁同步电机控制系统及其方法

Patent Citations (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH11136807A (ja) * 1997-10-31 1999-05-21 Hitachi Ltd 内燃機関付車両の制御方式
CN103296897A (zh) * 2012-03-05 2013-09-11 永济新时速电机电器有限责任公司 电力机车牵引变流器及电传动系统
CN105720831A (zh) * 2014-12-02 2016-06-29 永济新时速电机电器有限责任公司 带双电压传感器母线电压检测电路的牵引变流器
CN105720893A (zh) * 2014-12-02 2016-06-29 永济新时速电机电器有限责任公司 带二次滤波电路的牵引变流器
CN105790597A (zh) * 2016-03-29 2016-07-20 中车永济电机有限公司 一种动车用永磁同步电机牵引变流器主电路
CN105811814A (zh) * 2016-03-29 2016-07-27 中车永济电机有限公司 一种高速动车组牵引变流器主电路
CN105835703A (zh) * 2016-03-29 2016-08-10 中车永济电机有限公司 一种地铁用牵引变流器主电路
CN108696149A (zh) * 2018-05-25 2018-10-23 中车青岛四方车辆研究所有限公司 牵引变流器及其控制、故障处理和载波移相方法

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN111831225A (zh) * 2020-07-05 2020-10-27 中车永济电机有限公司 具有数据存储及远程传输扩展设计的辅助微机测控装置
CN112100945A (zh) * 2020-08-05 2020-12-18 苏州汇川联合动力系统有限公司 对地短路检测方法、电机控制器及计算机可读存储介质
CN112100945B (zh) * 2020-08-05 2023-12-26 苏州汇川联合动力系统股份有限公司 对地短路检测方法、电机控制器及计算机可读存储介质
CN114285303A (zh) * 2021-11-11 2022-04-05 中车永济电机有限公司 一种分段控制的四象限变流器
CN114285303B (zh) * 2021-11-11 2024-04-19 中车永济电机有限公司 一种四象限变流器的分段控制方法
EP4322394A1 (en) * 2022-08-04 2024-02-14 Milwaukee Electric Tool Corporation Power tool including current-based field weakening
CN115441787A (zh) * 2022-09-30 2022-12-06 东风商用车有限公司 一种电机前馈解耦控制方法
CN116176530A (zh) * 2022-12-23 2023-05-30 吉林大学 一种集成式线控液压制动系统液压力控制方法
CN117749025A (zh) * 2023-12-15 2024-03-22 荆州市三焱火炉金属制品有限公司 宽转速自适应调节的永磁同步电机控制方法
CN117749025B (zh) * 2023-12-15 2024-06-07 上海能环实业有限公司 宽转速自适应调节的永磁同步电机控制方法

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