WO2018227769A1 - 一种无位置传感器的bldc电机的启动方法 - Google Patents

一种无位置传感器的bldc电机的启动方法 Download PDF

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Publication number
WO2018227769A1
WO2018227769A1 PCT/CN2017/099662 CN2017099662W WO2018227769A1 WO 2018227769 A1 WO2018227769 A1 WO 2018227769A1 CN 2017099662 W CN2017099662 W CN 2017099662W WO 2018227769 A1 WO2018227769 A1 WO 2018227769A1
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Prior art keywords
torque
starting
motor
flux
current
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Ceased
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PCT/CN2017/099662
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English (en)
French (fr)
Inventor
王继忠
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Zhongshan Broad Ocean Motor Co Ltd
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Zhongshan Broad Ocean Motor Co Ltd
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Publication of WO2018227769A1 publication Critical patent/WO2018227769A1/zh
Priority to US16/254,543 priority Critical patent/US10840843B2/en
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/34—Arrangements for starting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/20—Arrangements for starting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/20—Arrangements for starting
    • H02P6/21—Open loop start

Definitions

  • the invention relates to a method for starting a BLDC motor without a position sensor.
  • BLDC motors without position sensor are controlled by vector control method. Since the rotor angle/position ⁇ is the key to voltage and current conversion in vector control, there are few key elements before the actual operation of the previous position sensorless brushless motor.
  • the position sensor brushless motor the usual method is to first give energy to the motor, and use the DC current Id to send to the d-axis (straight axis of the rotor rotating coordinate system) while maintaining the q-axis (the axis of the rotating coordinate system of the rotor). The current Iq is zero. The motor will lock to its original position and define the position at zero.
  • the sensorless motor starting strategy is poorly adapted for different load conditions, especially if the initial loading is unknown, for example, in a washing machine, the daily clothes and water volume are different. That is, the load is indefinite, sometimes larger, sometimes lighter, and the load can vary greatly. To compensate for this weakness, it is common practice to increase the initial startup phase to maximize the startup current to maximize the starting torque and accommodate different startup loads. The disadvantage of this method is that you must design for the worst case, increase the torque with a large current, but this may lead to inrush current power components and noise, the life of electronic components will be shortened, and the motor is prone to failure. Because of the large load, a large starting current is used to cause a large impact on the motor.
  • a method for starting a BLDC motor without a position sensor characterized in that it comprises the following steps:
  • the ds and qs axes are the two axes in the stator flux rotation coordinate system. ;
  • Step B Blocking state Torque lifting phase: When the motor is locked firmly (ie, when there is no rotation), a magnetic flux ⁇ is given to the motor, and the voltage of the ds axis is used during the ramping up of the current Iqs of the qs axis.
  • the magnetic flux closed loop circuit performs flux control to keep the magnetic flux constant, and uses the qs axis torque/current closed loop circuit to compare the preset starting torque T0 with the measured torque T1 and perform closed loop control until the measured torque T1 reaches the preset starting torque T0;
  • Step C Start the running phase: on the basis of the starting torque T0, continue to increase the torque by the torque/current closed loop to rotate the load, compare the preset starting speed V0 with the measured speed V1 and perform closed-loop control until the measured speed V1 reaches The preset start speed V0 is completed, and the normal start is completed; if the measured speed V1 is less than the preset start speed V0, the start failure is considered.
  • the starting torque T0 described above is a critical torque value, that is, when the measured torque T1 is greater than the starting torque T0, the motor can rotate.
  • the above-mentioned motor drives a fixed load
  • the starting torque T0 is a preset value, which is measured by experimental means, that is, how much torque value is required to start the load.
  • the motor described above drives a variable load, and the starting torque T0 is a preset value measured by experimental means, that is, the torque value required to start the minimum load.
  • Step C Start the operation phase, use all closed-loop PID control to control the motor, and increase the torque ⁇ T every interval until the rotation speed reaches the required speed.
  • the invention has the following effects:
  • the magnetic flux closed loop circuit performs flux control to keep the magnetic flux constant, and uses the qs axis torque/current closed loop circuit to compare the preset starting torque T0 with the measured torque T1 and perform closed loop control until the measured torque T1 reaches the preset starting torque T0; On the basis of the starting torque T0, the torque/current closed loop is used to continue the lifting torque to rotate the load, and the preset starting speed V0 and the measured speed V1 are compared and closed-loop control is performed until the measured speed V1 reaches the preset starting speed V0, and the completion is completed. Normal start; if the measured speed V1 is less than the preset start speed V0, the start failure is considered.
  • the starting torque T0 of the invention is adapted to the starting load, adopts full-range closed-loop control, and the starting smoothing is even and stable, and does not give too much starting torque like the existing open-loop control mode, thereby avoiding large inrush current during the starting phase of the motor.
  • the utility model causes the impact current power component and generates noise; therefore, the starting current is small, the impact on the current power component is small, the noise generating noise is small, the normal operation of the motor is effectively guaranteed, and the product life is prolonged.
  • a flux ⁇ value for a given motor is in the range of 60% saturation flux to 100% saturation flux.
  • a magnetic flux ⁇ of a given motor is preferably a motor saturation magnetic flux value, and the torque increase is the largest when the torque is raised, and the torque is increased. Time can be further shortened.
  • the motor is driven by a fixed load.
  • the starting torque T0 is a preset value, which is measured by experimental means, that is, how much torque value is required to start the load, and the operation is strong and easy to implement;
  • Step C Start the running phase: at the starting torque T0 Based on the torque/current closed loop, the torque is continuously increased to allow the load to rotate to accommodate load changes.
  • FIG. 1 is a schematic view of a BLDC motor without a position sensor according to the present invention
  • FIG. 2 is a circuit block diagram of a motor controller of a position sensorless BLDC motor of the present invention
  • FIG. 3 is a structural view of a position sensorless BLDC motor of the present invention using a salient pole inner rotor
  • Figure 4 is a coordinate system diagram of direct stator flux vector control of the present invention.
  • Figure 5 is a block diagram of direct stator flux vector control of the direct torque input of the present invention.
  • Figure 6 is a block diagram of direct stator flux vector control of the speed input of the present invention.
  • Figure 7 is a development view of the direct flux vector control block of Figure 5 of the present invention.
  • FIG 8 is a development view of the stator flux observer of Figure 5 of the present invention.
  • Figure 9 is a flow chart showing the generation of a reference magnetic flux vector control reference magnetic flux of the present invention.
  • Figure 10 is a flow chart showing the generation of the maximum qs-axis current of the direct stator flux vector control of the present invention.
  • Figure 11 is a vector control reference diagram of the control method of the present invention.
  • Figure 12a is a partial software control flow chart of the microprocessor of the present invention.
  • Figure 12b is another software control flow chart in the microprocessor of the present invention.
  • Figure 13 is a flowchart showing the startup of the position sensorless BLDC motor of the present invention.
  • Figure 14 is a schematic diagram of the comparison of the starting line of the present invention with a conventional starting line.
  • the invention is based on the patent number CN201210179372.0, the patent name is: a control method of the variable speed fan system, and the patent number of the publication number: CN103452883A, some of which are the same as CN201210179372.0, and the person skilled in the art completely It is possible to understand the technical solution of the present invention with reference to the contents of the technical solution of CN201210179372.0.
  • Embodiment 1 is a diagrammatic representation of Embodiment 1:
  • the position sensorless BLDC motor includes a stator assembly, a rotor assembly 2 and a motor controller.
  • the stator assembly includes a stator core 1 and coil windings, and the stator core 1 is provided with teeth 12 and slots. 11.
  • the coil winding is wound on the tooth 12, the rotor assembly is a salient pole inner rotor, and the motor controller adopts a vector control method without a position sensor.
  • the motor controller includes a microprocessor, a frequency converter and a detecting unit, and the detecting unit uses a phase current. DC bus current and phase voltage are input to the microprocessor, and the microprocessor is input.
  • the output signal controls the frequency converter, and the frequency converter is connected with the winding of the stator assembly.
  • the salient pole of the salient rotor has a convex polarity Lq/Ld ranging from 1.3 to 1.7, and the salient pole rotor surface air gap magnetic density is in the range of 0.6 Tesla. 0.8 Tesla, the processor controls the inverter through the output signal of the drive circuit, and the inverter is connected to the winding of the stator assembly.
  • the salient pole inner rotor 2 includes a rotor core 21 and a permanent magnet 22, and the rotor core 21 includes an annular ring 210 provided with a central shaft hole and a plurality of magnetic conductive blocks 211 protruding from the outer side of the annular ring 210.
  • a radial groove 212 for mounting the permanent magnet 22 is formed between the adjacent two magnetic conductive blocks 211, and a stopper hook 213 is protruded on the magnetic conductive block 211 on both sides of the opening portion of the radial groove 212, and the magnetic conductive block 211 is protruded
  • the cross-sectional shape of the outer side surface 214 is a circular arc line, and the outer side surface 214 is centered on a point A offset from the center o of the central shaft hole by a certain distance H. As shown in FIG.
  • the peripheral dotted line 6 represents the inner wall of the stator
  • the inner dotted line 7 represents the outer edge of the rotor core 21, and the outer dotted line 6 and the inner dotted line 7 form a radial gap therebetween, which we call the air gap L1
  • the permanent magnet 22 may also be referred to as magnetic steel
  • the thickness of the magnetic steel is indicated by H
  • the ratio of the air gap L1 to the thickness H of the magnetic steel is 0.03-0.065, so that the salient pole of the salient rotor has a convexity Lq/Ld ranging from 1.3 to 1.7
  • the ratio of the pole arc length L2 of the iron core 21 to the length L3 of the magnet steel is 0.8-1.0
  • the surface of the same polarity of the two permanent magnets is formed by the iron core to form a magnetic collecting effect, so that the surface of the salient pole rotor can be magnetically dense.
  • the motor torque density is improved, and the motor cost is reduced by using the salient pole structure to increase the magnetic flux density.
  • the magnet steel can replace the original neodymium iron crucible with ferrite, further reducing the cost, and the number of magnetic poles of the rotor. It is 8 poles or 10 poles or 12 poles.
  • Fig. 4 defines the reference coordinates of the permanent magnet synchronous motor vector, the stationary coordinate system ⁇ - ⁇ , the rotor rotating coordinate system dq, the stator flux rotating coordinate system ds-qs, the ⁇ rotor speed, and ⁇ is the load angle. (ie the angle between the stator flux rotation coordinate system and the rotor rotation coordinate system), angle Is the angle between the stationary coordinate system ⁇ - ⁇ and the rotor rotating coordinate system dq, the stator flux vector ⁇ ; angle It is the angle between the stationary coordinate system ⁇ - ⁇ and the stator flux rotation coordinate system ds-qs.
  • Rs is the stator resistance and p is the pole number.
  • the motor's control mode will be coupled to the flux through the flux and, therefore, converted to electromagnetic flux control.
  • the flux is usually expressed in the rotor coordinates d, q:
  • Ld is the motor d-axis inductance
  • Lq is the motor q-axis inductance
  • stator flux coordinates ds, qs, voltage model and torque are as follows:
  • the load angle and moment can be controlled directly by the qs axis voltage control formula 7 to control the qs axis current Iqs.
  • Vqs is related to Iqs
  • Fig. 5 it has a torque control mode flux control combination module diagram, including direct flux vector control DFVC, stator flux observer and dead time compensation module.
  • the torque command is introduced via the torque gain and re-adjusted as the required torque reference.
  • the block diagram of the speed control mode flux control combination includes a direct flux vector control DFVC, a stator flux observer, and a dead time compensation module.
  • the speed command is used as a reference for the proportional integral (PI) controller, and then the speed loop controller generates a torque command.
  • PI proportional integral
  • FIG. 7 is a block diagram of the entire direct flux vector control DFVC scheme implemented in a stator flux reference frame, with feedback information and output from the flux observer being fed to the direct flux vector control DFVC control strategy,
  • the torque command is used as the control reference variable.
  • the proportional integral controller is used for the two control loops.
  • This direct flux vector control DFVC The advantage of the solution is that the drive voltage, current limit and load angle limits are considered when adjusting the flux and current.
  • the first PID control is: torque current loop control The PI D (torque controller), the second PID control is the voltage flux loop control PID (flux controller).
  • the torque current loop control PI D is based on the input torque requirement T.
  • the formula (7) is used to calculate the magnitude of Iqs, and then compared with the measured current of the qs axis, and then realized by the qs axis current integral controller, thereby obtaining the control qs axis.
  • the second PID control is the voltage flux loop control PID, according to the input torque requirement T and ds axis measured current, according to the MTPA control method to find the best control flux ⁇ (ie reference flux), use the reference
  • the magnetic flux ⁇ is compared with the measured magnetic flux, and the voltage Vds of the ds axis is obtained by using the magnetic flux integral controller and the formula (8);
  • the stationary coordinate voltage V ⁇ is obtained by the coordinate transformation method by the voltage Vqs of the qs axis and the voltage Vds of the ds axis, V ⁇ , using the stationary coordinate voltage V ⁇ , V ⁇ is converted into three-phase voltages Va, Vb, Vc;
  • FIG 8 is a block diagram of a stator flux observer.
  • the observer is a key entity that provides stator flux values, stator flux position and load angle.
  • the output from this stator flux observer is the direct flux vector control DFVC.
  • the amount of input is the stator flux observer.
  • the stator flux observer is a hybrid of two different models. It is properly processed in two different situations based on whether the inverter is saturated or not. In low speed, it operates in current mode, that is, only controls the current according to the input torque. Control, which is the torque current control module mentioned above; on the other hand, it operates in voltage mode at high speed, only controls the voltage, which is the direct stator flux vector control module mentioned above, and the cross angle frequency is low speed and high speed.
  • the transition mode state between modes can be defined by the observer gain (rad/s rad/sec).
  • FIG. 9 is a block diagram of magnetic flux reference generation, illustrating low-speed MTPA control and field weakening control of the lifting torque, and the reference magnetic flux generating module provides a reference magnetic flux based on the inverter saturation state or speed range, as shown in FIG. Below the basic speed, the reference flux is given by the best control method, named Maximum Torque per Ampere MTPA.
  • This MTPA generation uses command torque as input. For most built-in permanent magnet motors, this MTPA method is a non-linear curve. This curve can be obtained by characteristic test, or finite element analysis can simulate this curve. Then, the look-up table method is implemented more effectively.
  • the back electromotive force of the motor increases and the frequency converter begins to saturate, which makes the voltage limiting condition effective and the MTPA condition is no longer valid.
  • the highest voltage is dependent on the PWM pulse width modulation strategy and the instantaneous DC link voltage, which is achieved by limiting the stator flux reference value from the field weakening limit module.
  • the transition between the field weakening control and the MTPA control is automatically performed based on the actual effective maximum DC bus voltage and the required qs axis current. As shown in Figure 10, the voltage limiting acts as if the flux and current are present. Controller output.
  • the voltage limit is based on the following formula:
  • Vsmax relies on the PWM strategy and the instantaneous maximum DC bus voltage Vdc. It is known from equation (9) that the operation of the voltage constraint is to constrain the stator flux.
  • FIG. 10 is a block diagram of maximum qs-axis current generation illustrating the current and load angle limits in the MTPV---maximum torque per volt-control strategy for lift torque control.
  • the qs axis current is calculated from the torque/current generation module in Figure 10, however, this qs axis current is limited by the maximum inverter current.
  • the positive and negative values of the current required by the current limiter for the qs axis are bidirectional.
  • the qs axis current is constrained by the maximum inverter current, and the qs axis maximum reference current is defined:
  • Ids is the ds-axis stator current in high-speed torque boost control.
  • the best control strategy is to maximize the effectiveness of the available phase voltage, resulting in a minimum current.
  • the motor runs either on or off.
  • the maximum load angle condition called the maximum torque per volt MTPV operation, is obtained by using the load angle analysis process, including simulation and acceleration implementation evaluation tests. This maximum load angle is established as the load angle limit ensures motor stability. As shown in Figure 10, the load angle limit is performed using a PI proportional-integral controller to reduce the maximum allowable current.
  • the motor control operation is marked in the MTPA mode, as the curve segment (0, W1), the current vector ISW1, as the speed increases, the inverter saturates, and the motor works in the MTPV mode curve. That is, the line segment (W1, W2), the current vector ISW2, reaches the maximum torque and speed, and the control is very efficient and saves energy.
  • the current vector IWN is the current over-vector of point W1 to point W2. As can be seen from the figure, the path is very short and efficient, saving energy loss during the transition phase.
  • the BLDC motor includes a stator assembly, a rotor assembly and a motor controller.
  • the rotor assembly is embedded in the rotor core.
  • the salient pole inner rotor of steel, the motor controller adopts the vector control mode without position sensor, the motor controller includes a microprocessor, a frequency converter and a detecting unit, and the detecting unit inputs the phase current, the phase voltage and the DC bus voltage to the microprocessor.
  • the microprocessor output signal controls the frequency converter, the frequency converter is connected with the winding of the stator component, the microprocessor sets the torque current control module, the direct stator flux vector control module (ie the flux voltage control module) and the stator flux observer,
  • the method comprises the following steps:
  • Step 2) the stator flux observer outputs the stator flux, the flux angle and the load angle;
  • Step 3 calculates a reference magnetic flux based on the MTPA working mode, and the MTPA working mode is an operating mode that outputs the maximum torque per ampere;
  • Step 4) calculating the limiting magnetic flux based on the MTPV working mode, and the working mode of the MTPV working mode is output according to the maximum torque per volt;
  • Step 5 limiting whether the magnetic flux is greater than the reference magnetic flux
  • Step 6 If the limiting magnetic flux is greater than the reference magnetic flux, the frequency converter is not saturated, the voltage Vqs is calculated according to the torque demand, and the voltage Vds is calculated in the MTPA working mode; if the limiting magnetic flux is not greater than the reference magnetic flux, the voltage Vqs is calculated according to the torque demand, Calculating the voltage Vds in the MTPV working mode;
  • Step 7) Converting the voltages Vds and Vqs into voltages V ⁇ and V ⁇ in the stationary coordinates, and converting the voltages V ⁇ and V ⁇ in the stationary coordinates into three-phase voltages Va, Vb, and Vc, and performing the three-phase voltages Va, Vb, and Vc. PWM modulation.
  • the ds and qs axes are the two axes in the stator flux rotation coordinate system. ;
  • Step B Blocking state Torque lifting phase: When the motor is locked firmly (ie, when there is no rotation), a magnetic flux ⁇ is given to the motor, and the voltage of the ds axis is used during the ramping up of the current Iqs of the qs axis.
  • the magnetic flux closed loop circuit performs flux control to keep the magnetic flux constant, and uses the qs axis torque/current closed loop circuit to compare the preset starting torque T0 with the measured torque T1 and perform closed loop control until the measured torque T1 reaches the preset starting torque T0;
  • Step C Start the running phase: on the basis of the starting torque T0, continue to increase the torque by the torque/current closed loop to rotate the load, compare the preset starting speed V0 with the measured speed V1 and perform closed-loop control until the measured speed V1 reaches The preset start speed V0 is completed, and the normal start is completed; if the measured speed V1 is less than the preset start speed V0, the start failure is considered.
  • the starting torque T0 is a critical torque value, that is, when the measured torque T1 is greater than the starting torque T0, the motor can rotate.
  • the above motor drives a fixed load
  • the starting torque T0 is a preset value, which is measured by experimental means, that is, how much torque value is required to start the load.
  • the above motor drives a variable load
  • the starting torque T0 is a preset value measured by experimental means, that is, the torque value required to start the minimum load.
  • Step C Start the operation phase, use all closed-loop PID control to control the motor, and increase the torque ⁇ T every interval, until the rotation speed reaches the requirement, ⁇ T may be in the range of 5%T0 to 10%T0.
  • the principle of the position sensorless BLDC motor starting method of the present invention is based on the two loop PID controls of FIG.
  • the first PID control is: torque current loop control PI D (torque controller), and the second PID control is voltage flux loop control PID (flux controller).
  • torque current loop control PI D torque controller
  • voltage flux loop control PID flux controller
  • the starting torque T0 is a preset value, which is measured by experimental means, that is, how much torque value is required to start the load, and the starting torque T0 is regarded as
  • An external input torque demand a given motor a flux ⁇ is the saturation flux value (fixed at the start-up phase, the fastest speed can be increased), using the formula 7 to gradually climb the qs axis current Iqs to gradually increase the torque,
  • the torque current loop control PI D torque controller
  • the second PID control is a voltage flux loop control PID (flux controller) for closed-loop control, so that the magnetic flux is in saturation magnetic flux.
  • Step B The lock state torque boost phase is achieved by the above principle. Since the starting torque T0 is a critical torque value, step C: start the running phase: on the basis of the starting torque T0, the torque/current closed loop is used to continue the lifting torque to rotate the load, and the preset starting speed V0 and the measured speed V1 are compared. And the closed-loop control is performed until the measured rotational speed V1 reaches the preset starting rotational speed V0, and the normal starting is completed; if the measured rotational speed V1 is less than the preset starting rotational speed V0, the starting failure is considered.
  • a magnetic flux ⁇ of a given motor is not necessarily a saturation magnetic flux value, and may be a 60% saturation magnetic flux value or an 80% saturation magnetic flux value, as long as It is a fixed value.
  • the second PID control is a voltage flux loop control PID (flux controller) for closed-loop control, so that the magnetic flux remains unchanged, and then the torque value can be quantitatively calculated by formula (7).
  • the torque current loop control PI D torque controller
  • the load When the motor is driven by a variable load, such as a washing machine, the load is variable, but the variable load has a range, that is, the range from the minimum load to the maximum load.
  • the starting torque T0 is a preset value, which is experimentally Measured, that is, the torque value required to start the minimum load.
  • step B proceed to step C to start the operation phase: based on the starting torque T0, continue to increase the torque with the torque/current closed loop to rotate the load. Comparing the preset starting speed V0 with the measured speed V1 and performing closed-loop control until the measured speed V1 reaches the preset starting speed V0, and the normal starting is completed; if the measured speed V1 is less than the preset starting speed V0, the starting failure is considered.
  • the motor is controlled by all closed-loop PID controls, and the torque ⁇ T is increased every interval until the rotation speed reaches the requirement, and ⁇ T can be in the range of 5% T0 to 10% T0.
  • the step of speed closed-loop control is introduced in step C, which can better adapt to the start of variable load.
  • Figure 14 is a comparison diagram of the starting line of the position sensorless BLDC motor starting method of the present invention and the starting line of the conventional starting method, wherein the line A is the starting line of the conventional BLDC motor starting method without position sensor, B The line is the starting line of the position sensorless BLDC motor starting method of the present invention.
  • the start line of the position sensorless BLDC motor starting method of the present invention pays attention to the actual load requirement from the beginning, so the start is soft, the range is wide, the whole process is closed-loop control, and the start smoothness is even and stable, thereby avoiding the inrush current of the motor starting stage. Large, causing inrush current power components and generating noise; therefore, the starting current is small, the impact on the current power components is small, the noise generating noise is small, effectively ensuring the normal operation of the motor and prolonging the life of the product.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)
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Abstract

一种无位置传感器的BLDC电机的启动方法，包括步骤A：复位：在ds轴选择一个电压Vds，同时保持qs轴的电压Vqs＝0，使电机回到预设的位置；步骤B：在电机被牢牢锁定的状态，给定电机一个磁通λ，在qs轴的电流Iqs进行爬坡上升过程中利用ds轴的电压/磁通闭环回路进行磁通控制使磁通保持一定，利用qs轴的力矩/电流闭环回路，比较预设启动力矩T0与实测力矩T1并进行闭环控制，直到实测力矩T1达到预设启动力矩T0；步骤C：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制。该方法全程闭环控制，启动顺滑均匀平稳，启动电流小，对电流功率组件冲击小，产生噪音的噪音小。

Description

一种无位置传感器的BLDC电机的启动方法 技术领域：
本发明涉及一种无位置传感器的BLDC电机的启动方法。
背景技术：
无位置传感器的BLDC电机多数采用矢量控制方法进行控制，由于转子角/位置θ是矢量控制中电压和电流转换的关键,在先前无位置传感器无刷电动机实际运行前中确少关键要素,启动无位置传感器器无刷电动机时，通常的方法是先给能量给电机，将用直流电流Id送到d轴(转子旋转坐标系的直轴),同时保持q轴(转子旋转坐标系的交轴)电流Iq为零。电动机将锁到它原来的位置,定义为零时的位置。然后,它雇佣了一个开环控制的模式控制电机运行,在到达预定义速度时实现正常启动，然后进入正常工作状态并切换到常规的闭环控制模式。因为传统的无位置传感器的BLDC电机采用反电动势(BEMF信号)去估算转子位置，当转子启动阶段的低速状态无法测量计算反电动势，故一开始只能开环控制。如美国专利号：US7667423(B2)所述描述的电机启动方法。
由于在初始阶段没有反馈信号,针对不同的负载条件下,无位置传感器电动机启动策略适应性很差,尤其是在初始加载是未知的情况下，比如：在一个洗衣机,每天的衣服和水量都不同，也就是负载是不定的，有时候比较大，有时候比较轻，负载可以大幅变化。为了弥补这个弱点,常见的做法是增加初始启动阶段尽量增大启动电流以尽量增加启动转矩，适应不同的启动负载。这种方法的缺点是,你必须为最坏的情况设计,以大电流启动增大力矩，但这样可能导致冲击电流功率组件和产生噪音，电子元器件的寿命会缩短，电机容易发生故障。因为当轻的负载时，用大的启动电流给电机造成较大的冲击。
发明内容：
本发明的目的是提供一种无位置传感器的BLDC电机的启动方法，解决现有 技术中电机启动阶段冲击电流大，造成冲击电流功率组件和产生噪音，影响电机工作寿命的技术问题。
本发明的目的是通过下述技术方案予以实现的：
一种无位置传感器的BLDC电机的启动方法，其特征在于：它包括如下步骤：
步骤A：复位：在ds轴选择一个的电压Vds,同时保持qs轴的电压Vqs＝0,使电机回到预设的位置,ds轴和qs轴是定子磁通旋转坐标系里面的两个轴；
步骤B：闭锁状态力矩提升阶段：在电机被牢牢锁定的状态(即没有转动时),给定电机一个磁通λ，在qs轴的电流Iqs进行爬坡上升过程中利用ds轴的电压/磁通闭环回路进行磁通控制使磁通保持一定，利用qs轴的力矩/电流闭环回路，比较预设启动力矩T0与实测力矩T1并进行闭环控制，直到实测力矩T1达到预设启动力矩T0；
步骤C：启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。
上述所述的步骤B：闭锁状态力矩提升阶段：给定电机一个磁通λ值是在60％的饱和磁通至100％的饱和磁通的范围。
上述所述的步骤B：闭锁力矩提升阶段：给定电机一个磁通λ是一个电机饱和磁通值。
上述所述的启动力矩T0是一个临界力矩值，即当实测力矩T1大于启动力矩T0时，电机才能转动。
上述所述的电机驱动的是一个固定负载，启动力矩T0是一个预设值，是通过实验手段测量得到，即要启动负载需要多大的力矩值。
上述所述的电机驱动的是一个可变负载，启动力矩T0是一个预设值，是通过实验手段测量的，即要启动最小负载时需要的力矩值。
上述所述的步骤C：启动运转阶段，利用所有的闭环PID控制来控制电机，每间隔一段时间提升力矩△T，直到转速度达到要求。
本发明与现有技术相比，具有如下效果：
1)在闭锁状态力矩提升阶段：在电机被牢牢锁定的状态(即没有转动时),给定电机一个磁通λ，在qs轴的电流Iqs进行爬坡上升过程中利用ds轴的电压/磁通闭环回路进行磁通控制使磁通保持一定，利用qs轴的力矩/电流闭环回路，比较预设启动力矩T0与实测力矩T1并进行闭环控制，直到实测力矩T1达到预设启动力矩T0；在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。本发明的启动力矩T0与启动负载相适应，采用全程闭环控制，启动顺滑均匀平稳，不会象现有的开环控制方式给与太大的启动力矩，从而避免电机启动阶段冲击电流大，造成冲击电流功率组件和产生噪音；因此，启动电流小，对电流功率组件冲击小，产生噪音的噪音小，有效保障电机的正常运作，延长产品寿命。
2)当闭锁状态力矩提升阶段：给定电机一个磁通λ值是在60％的饱和磁通至100％的饱和磁通的范围。当磁通λ为一定值时，根据力矩计算公式计算力矩的就方便，给定电机一个磁通λ最好是一个电机饱和磁通值，提升力矩时力矩的增量是最大的，提升力矩的时间可以进一步缩短。
3)电机驱动的是一个固定负载，启动力矩T0是一个预设值，是通过实验手段测量得到，即要启动负载需要多大的力矩值，操作性强，容易实施；
4)电机驱动的是一个可变负载，启动力矩T0是一个预设值，是通过实验手段测量的，即要启动最小负载时需要的力矩值，靠步骤C：启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，来适应负载的变化。
附图说明：
图1是本本发明无位置传感器的BLDC电机的示意图；。
图2是本发明无位置传感器的BLDC电机的电机控制器的电路方框图；
图3是本发明无位置传感器的BLDC电机采用凸极内转子的结构图；
图4是本发明的直接定子磁通矢量控制的坐标系图；
图5是本发明的直接力矩输入的直接定子磁通矢量控制的方框图；
图6是本发明的速度输入的直接定子磁通矢量控制的方框图；
图7是本发明的图5中直接磁通矢量控制方块的展开图；
图8是本发明的图5中定子磁通观测器的的展开图；
图9是本发明的直接定子磁通矢量控制参考磁通的产生流程图；
图10是本发明的直接定子磁通矢量控制最大qs轴电流的产生流程图；
图11是本发明的控制方法的矢量控制参考图；
图12a本发明微处理器里面的部分软件控制流程图；
图12b本发明微处理器里面的另一部软件控制流程图；
图13是本发明的无位置传感器的BLDC电机的启动流程图；
图14是本发明的启动线路与传统启动线路的对照示意图。
具体实施方式：
下面通过具体实施例并结合附图对本发明作进一步详细的描述。
本发明是借鉴专利号为CN201210179372.0、专利名称为：一种变速风机系统的控制方法、公告号为：CN103452883A的专利文献，其中部分内容与CN201210179372.0是相同的，本领域普通技术人员完全可能参考CN201210179372.0的技术方案的内容，理解本发明的技术方案。
实施例一：
如图1、图2、图3所示，无位置传感器的BLDC电机包括定子组件、转子组件2和电机控制器，定子组件包括定子铁芯1和线圈绕组，定子铁芯1设置齿12和槽11，线圈绕组绕在齿12上，转子组件是凸极内转子，电机控制器采用无位置传感器的矢量控制方式，电机控制器包括微处理器、变频器和检测单元，检测单元将相电流,DC总线电流及相电压输入到微处理器，微处理器输 出信号控制变频器，变频器与定子组件的绕组连接，所述的凸极转子的凸极性Lq/Ld的范围1.3—1.7，凸极转子表面气隙磁密的范围在0.6特斯拉—0.8特斯拉，处理器通过驱动电路输出信号控制变频器，变频器与定子组件的绕组连接。
所述的凸极内转子2包括转子铁芯21和永磁体22，转子铁芯21包括设置有中心轴孔的环形圈210和从环形圈210外侧凸出的若干块导磁块211,在相邻两导磁块211之间形成用于安装永磁体22的径向凹槽212，在径向凹槽212的开口部的两侧导磁块211上凸出挡钩213，导磁块211的外侧面214的截面形状是圆弧线，外侧面214是以偏移中心轴孔的中心o一定距离H的点A为圆心。如图17所示，外围虚线6代表定子内壁，内围虚线7代表转子铁芯21的外边缘，外围虚线6与内围虚线7之间径向形成间隙，我们称为气隙L1,永磁体22也可以称为磁钢，磁钢厚度用H标示，气隙L1与磁钢厚度H的比值0.03-0.065可使所述的凸极转子的凸极性Lq/Ld的范围1.3—1.7，转子铁芯21的极弧长度L2与磁钢长度L3的比值0.8-1.0,依靠两片永磁铁相同极性的那一面通过铁芯形成聚磁效果,可使凸极转子表面气隙磁密的范围在0.6-0.8特斯拉，改进电机力矩密度，而且通过利用凸极结构提高磁通密度来降低电机成本，磁钢可以采用铁氧体取代原来的钕铁錋，进一步减低成本，转子的磁极数为8极或者10极或者12极。
如图4所示，图4定义了永磁同步电机矢量的参考坐标，静止坐标系α-β,转子旋转坐标系d-q，定子磁通旋转坐标系ds-qs，ω转子速度，δ是负载角(即定子磁通旋转坐标系与转子旋转坐标系的夹角)，角度
Figure PCTCN2017099662-appb-000001
是静止坐标系α-β与转子旋转坐标系d-q的夹角，定子磁通矢量λ；角度
Figure PCTCN2017099662-appb-000002
是静止坐标系α-β与定子磁通旋转坐标系ds-qs的夹角。
在静止坐标α,β下内置永磁电机的电压与力矩的关系如下：e:
Figure PCTCN2017099662-appb-000003
Figure PCTCN2017099662-appb-000004
Rs是定子电阻，p是磁极对数.
电机的控制模式将通过磁通耦合电流，因此，转化为电磁磁通控制，对于内置永磁电机的磁通通常在转子坐标d,q来表达数学式：
Figure PCTCN2017099662-appb-000005
λm是磁链
如果是磁通未饱和状态，上述公式可以简化为：
Figure PCTCN2017099662-appb-000006
Ld是电机d轴电感，Lq是电机q轴电感.
如果转子位置是
Figure PCTCN2017099662-appb-000007
然后磁域在静止坐标α,β下，该公式表示为
Figure PCTCN2017099662-appb-000008
Figure PCTCN2017099662-appb-000009
在定子磁通坐标ds,qs，电压模型和力矩关系如下：
Figure PCTCN2017099662-appb-000010
Te＝(3/2)·p·λ·iqs＝kT·iqs   (7)
ω转子速度，δ是负载角
从公式6知道,我们得到定子磁通矢量λ，ds轴电压被直接修正，而负
载角和力矩能被qs轴电压控制公式7控制qs轴电流Iqs直接控制力矩。
而Vqs与Iqs相关；
Vqs＝Iqs*Rs+Ls*dIqs/dt Rs是定子电阻,故控制qs轴的电压Vqs可以控制qs轴电流Iqs；
从公式式(6)中立即发现，定子磁通矢量值λ可直接通过ds-轴电压Vds来调节，而负载角(因而和转矩)可通过qs-轴电压Vqs来控制。然而，公式(7)显示，调节qs-轴电流Iqs可直截了当地调节转矩。因此，(ds，qs)坐标系的电压模型具有qs轴电流Iqs作为状态变量来替代在[7]中用到的负载角。
如在[6]中证实的那样，在(ds，qs)构架中的电压模型就变成：
Figure PCTCN2017099662-appb-000011
如图5所示，它力矩控制模式磁通控制控制组合模块图，包括直接磁通矢量控制DFVC、定子磁通观测器和死区时间补偿模块。经由力矩增益导入力矩指令，作为要求的力矩基准而重新调节。
如图6所示，它速度控制模式磁通控制组合的模块图包括直接磁通矢量控制DFVC、定子磁通观测器和死区时间补偿模块。速度指令用作比例积分(PI)控制器的基准，然后，速度环控制器生成力矩指令。
图7是整个直接磁通矢量控制DFVC方案的模块图，方案在定子磁通基准构架中实施，从磁通观测器传感的反馈信息和输出量输送给这个直接磁通矢量控制DFVC控制策略，力矩指令是作为控制参照变量，在方案内有两个控制环路，定子磁通环路和q-轴电流环路，比例积分控制器用于这两个控制环路，这个直接磁通矢量控制DFVC方案的优点是，当调节磁通和电流时，变频器电压、电流限制和负载角限制予以相关考虑。
图7中本方案中有两个环路PID控制。第一个PID控制是：力矩电流环控 制PI D(力矩控制器)，第二个PID控制是电压磁通环控制PID(磁通控制器)。力矩电流环控制PI D就是根据输入力矩要求T利用公式(7)来计算出Iqs的大小，再与qs轴实测电流进行比较，然后通过qs轴电流积分控制器来实现，从而得到控制qs轴的电压Vqs；第二个PID控制是电压磁通环控制PID,根据输入力矩要求T和ds轴实测电流,根据MTPA控制方式查表法获得最佳控制磁通λ(即参考磁通)，利用参考磁通λ与实测磁通进行比较，利用磁通积分控制器和公式(8)得到ds轴的电压Vds；通过qs轴的电压Vqs和ds轴的电压Vds利用坐标变换方式得到静止坐标电压Vα,Vβ，利用静止坐标电压Vα,Vβ再转成三相电压Va、Vb、Vc；
图8是定子磁通观测器的模块图，观测器是关键实体，提供定子磁通值、定子磁通位置和负载角，来自这个定子磁通观测器的输出量是对直接磁通矢量控制DFVC的输入量。自然，定子磁通观测器是两个不同模型的混杂体，基于变频器是否饱和以两种不同的情形适当地处理，在低速时，以电流模式运行，即根据输入力矩只控制电流，便完成控制，也就是上面提到的力矩电流控制模块；另一方面，在高速时以电压模式工作，只控制电压，也就是上面提到的直接定子磁通矢量控制模块，交叉角频率在低速与高速模式之间转换模式状态，可用观测器增益(rad/s拉德/秒)来定义。
图9是磁通基准生成的模块图，图示说明低速MTPA控制和提升力矩的弱磁控制，基准磁通生成模块基于变频器饱和状态或速度范围提供基准磁通，如图6所示，在基本转速之下时，基准磁通由如下最佳控制方式给出，命名为最大力矩每安培MTPA方式。这种MTPA的生成使用指令力矩作为输入，对于大多数内置永磁电机来说，这种MTPA方式是非线性曲线，进行特性测试可以得到此曲线，或有限元分析模拟这种曲线。然后，查表法较为有效地实施。当转速上升时，电机的反电动势增大，变频器开始饱和，这就使得电压限制条件有效，同时MTPA条件不再有效。最高电压取决于PWM脉宽调制策略以及瞬时DC链路电压，电压限制运行通过限制定子磁通基准值而得以实现，此基准值来自弱磁限制模块。 按照这种方法，弱磁控制与MTPA控制之间转换自动执行，这基于实际的有效的最高DC总线电压和所需要的qs轴电流，如图10所示，电压限制作用犹如对磁通和电流控制器输出。
电压限制根据如下公式：
Figure PCTCN2017099662-appb-000012
Vsmax依靠PWM策略和即时的最高DC总线电压Vdc.从公式(9)知道电压约束的的操作是使定子磁通被约束
Figure PCTCN2017099662-appb-000013
Figure PCTCN2017099662-appb-000014
如图10所示，图10是最大qs轴电流生成的模块图，图示说明在提升力矩控制的MTPV---最大力矩每伏特-控制策略中，电流和负载角的限制。为了传送所要求的力矩，qs轴电流从图10中的力矩/电流生成模块计算出来，然而，这一qs轴电流为最大变频器电流所限制。电流限制器对于qs轴所要求电流的正负值是双向的。
qs轴电流被约束被最大的变频器电流，qs轴最大参考电流被定义：
Figure PCTCN2017099662-appb-000015
ids是ds轴定子电流在高速时力矩提高控制中，最佳控制策略是使可供利用的相电压的有效性最大化，导致最小电流，为了实施这项策略，电机运行或是开启或是关闭最大负载角条件，称之为最大力矩每伏特MTPV运行，这一最大负载角通过利用负载角分析过程可以得到，包括模拟和加速实施评估测试。这一最大负载角的确立如同负载角限制那样，确保电机稳定性。如图10所示，负载角限制是利用PI比例积分控制器来执行的，降低最大许可电流。
如图11所示，在低速范围,电机控制操作在MTPA工作模式下,标示为曲线段(0,W1)，电流矢量ISW1,随着速度增加，变频器饱和,电机工作在MTPV模式的曲线,即线段(W1,W2),电流矢量ISW2，达到最大力矩和转速,控制非常高效和节约电能。电流矢量IWN是点W1到点W2的电流过度矢量，从图中可以看过，路径是非常短的，是高效的，节约过渡阶段的能量损耗。
如图12a、图12b所示是一种无位置传感器的BLDC电机启动成功后的控制方法，所述的BLDC电机包括定子组件、转子组件和电机控制器，转子组件是在转子铁芯内嵌入磁钢的凸极内转子，电机控制器采用无位置传感器的矢量控制方式，电机控制器包括微处理器、变频器和检测单元，检测单元将相电流、相电压及DC总线电压输入到微处理器，微处理器输出信号控制变频器，变频器与定子组件的绕组连接，微处理器设置力矩电流控制模块、直接定子磁通矢量控制模块(即磁通电压控制模块)和定子磁通观测器，该方法包括如下步骤：
步骤1)读取需求力矩；
步骤2)定子磁通观测器输出定子磁通、磁通角和负载角；
步骤3基于MTPA工作模式下计算参考磁通，所述的MTPA工作模式是按每安培最大力矩输出的工作模式；
步骤4)基于MTPV工作模式下计算限制磁通，所述的MTPV工作模式按每伏特最大力矩输出的工作模式；
步骤5)限制磁通是否大于参考磁通；
步骤6)若限制磁通大于参考磁通，变频器未饱和,根据力矩需求计算电压Vqs,在MTPA工作模式下计算电压Vds；若限制磁通不大于参考磁通，根据力矩需求计算电压Vqs,在MTPV工作模式下计算电压Vds；
步骤7)将电压Vds、Vqs转换成静止坐标下的电压Vα、Vβ，再将静止坐标下的电压Vα、Vβ变换成三相电压Va、Vb、Vc，利用三相电压Va、Vb、Vc进行PWM调制。
以上是申请人对本发明的无位置传感器的BLDC电机启动后的控制方法及相 关理论作了详细的描述，以便本领域普通技术人员对本发明有一个大概的了解，有了这个基础，下面着重介绍无位置传感器的BLDC电机启动方法，如图13所示，它包括如下步骤：
步骤A：复位：在ds轴选择一个的电压Vds,同时保持qs轴的电压Vqs＝0,使电机回到预设的位置,ds轴和qs轴是定子磁通旋转坐标系里面的两个轴；
步骤B：闭锁状态力矩提升阶段：在电机被牢牢锁定的状态(即没有转动时),给定电机一个磁通λ，在qs轴的电流Iqs进行爬坡上升过程中利用ds轴的电压/磁通闭环回路进行磁通控制使磁通保持一定，利用qs轴的力矩/电流闭环回路，比较预设启动力矩T0与实测力矩T1并进行闭环控制，直到实测力矩T1达到预设启动力矩T0；
步骤C：启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。
上述步骤B：闭锁状态力矩提升阶段：给定电机一个磁通λ值是在60％的饱和磁通至100％的饱和磁通的范围。上述步骤B：闭锁力矩提升阶段：给定电机一个磁通λ是一个电机饱和磁通值。
上述启动力矩T0是一个临界力矩值，即当实测力矩T1大于启动力矩T0时，电机才能转动。
上述电机驱动的是一个固定负载，启动力矩T0是一个预设值，是通过实验手段测量得到，即要启动负载需要多大的力矩值。
上述电机驱动的是一个可变负载，启动力矩T0是一个预设值，是通过实验手段测量的，即要启动最小负载时需要的力矩值。
上述所述的步骤C：启动运转阶段，利用所有的闭环PID控制来控制电机，每间隔一段时间提升力矩△T，直到转速度达到要求，△T可以是5％T0至10％T0的范围。
本发明的无位置传感器的BLDC电机启动方法的原理是：基于图7的两个环路PID控制。第一个PID控制是：力矩电流环控制PI D(力矩控制器)，第二个PID控制是电压磁通环控制PID(磁通控制器)。并利用如下的公式：
Figure PCTCN2017099662-appb-000016
Te＝(3/2)·p·λ·iqs＝kT·iqs   (7)
Figure PCTCN2017099662-appb-000017
Vqs＝Iqs*Rs+Ls*dIqs/dt  (12)
启动阶段没有外部输入的力矩需求，假设电机驱动的是一个固定负载，启动力矩T0是一个预设值，是通过实验手段测量得到，即要启动负载需要多大的力矩值，将启动力矩T0视为一个外部输入的力矩需求，给定电机一个磁通λ是饱和磁通值(在启动阶段固定不变，可最快速度提升力矩)，利用公式7逐步爬升qs轴的电流Iqs以逐步提升力矩，并利用力矩电流环控制PI D(力矩控制器)进行闭环控制，同时利用第二个PID控制是电压磁通环控制PID(磁通控制器)进行闭环控制，使磁通处于饱和磁通。步骤B：闭锁状态力矩提升阶段就是利用上述的原理实现。因为启动力矩T0是一个临界力矩值，步骤C：启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。
上述的无位置传感器的BLDC电机启动方法的原理中，给定电机一个磁通λ不一定是饱和磁通值，也可以是60％饱和磁通值，也可以是80％饱和磁通值，只要 是一个定值就可以，利用第二个PID控制是电压磁通环控制PID(磁通控制器)进行闭环控制，使磁通保持不变，然后通过公式(7)就可以定量计算出力矩值，实现力矩电流环控制PI D(力矩控制器)进行闭环控制。
当电机驱动的是一个可变负载，例如洗衣机，其负载是可变的，但可变负载有一个范围，即最小负载至最大负载的范围，启动力矩T0是一个预设值，是通过实验手段测量的，即要启动最小负载时需要的力矩值，当完成步骤B后，进入步骤C，启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。利用所有的闭环PID控制来控制电机，每间隔一段时间提升力矩△T，直到转速度达到要求，△T可以是5％T0至10％T0的范围。步骤C中引入了速度闭环控制的环节，可以更加好适应可变负载的启动。
图14所示，是本发明的无位置传感器的BLDC电机启动方法的启动线路与传统启动方法的启动线路的对比图，图中A线是传统无位置传感器的BLDC电机启动方法的启动线路，B线是本发明的无位置传感器的BLDC电机启动方法的启动线路，很明显传统的无位置传感器的BLDC电机启动方法的启动线路一开始就给与很大的电流和力矩，而没有估算实际负载需要；而本发明的的无位置传感器的BLDC电机启动方法的启动线路一开始就注意实际负载需要，因而启动柔和，适应范围广，全程闭环控制，启动顺滑均匀平稳，从而避免电机启动阶段冲击电流大，造成冲击电流功率组件和产生噪音；因此，启动电流小，对电流功率组件冲击小，产生噪音的噪音小，有效保障电机的正常运作，延长产品寿命。
以上实施例为本发明的较佳实施方式，但本发明的实施方式不限于此，其他任何未背离本发明的精神实质与原理下所作的改变、修饰、替代、组合、简化，均为等效的置换方式，都包含在本发明的保护范围之内。

Claims (7)

  1. 一种无位置传感器的BLDC电机的启动方法，其特征在于：它包括如下步骤：
    步骤A：复位：在ds轴选择一个的电压Vds,同时保持qs轴的电压Vqs＝0,使电机回到预设的位置,ds轴和qs轴是定子磁通旋转坐标系里面的两个轴；
    步骤B：闭锁状态力矩提升阶段：在电机被牢牢锁定的状态(即没有转动时),给定电机一个磁通λ，在qs轴的电流Iqs进行爬坡上升过程中利用ds轴的电压/磁通闭环回路进行磁通控制使磁通保持一定，利用qs轴的力矩/电流闭环回路，比较预设启动力矩T0与实测力矩T1并进行闭环控制，直到实测力矩T1达到预设启动力矩T0；
    步骤C：启动运转阶段：在启动力矩T0的基础上，利用力矩/电流闭环回路继续提升力矩以便使负载转动起来，比较预设启动转速V0与实测转速V1并进行闭环控制，直到实测转速V1达到预设启动转速V0，完成正常启动；若实测转速V1小于预设启动转速V0，则认为启动失败。
  2. 根据权利要求1所述的一种无位置传感器的BLDC电机的启动方法，其特征在于：步骤B：闭锁状态力矩提升阶段：给定电机一个磁通λ值是在60％的饱和磁通至100％的饱和磁通的范围。
  3. 根据权利要求2所述的一种无位置传感器的BLDC电机的启动方法，其特征在于：步骤B：闭锁力矩提升阶段：给定电机一个磁通λ是一个电机饱和磁通值。
  4. 根据权利要求3所述的一种无位置传感器的BLDC电机的启动方法，其特征在于：启动力矩T0是一个临界力矩值，即当实测力矩T1大于启动力矩T0时，电机才能转动。
  5. 根据权利要求4所述的一种无位置传感器的BLDC电机的启动方法，其特征在于：电机驱动的是一个固定负载，启动力矩T0是一个预设值，是通过实验手段测量得到，即要启动负载需要多大的力矩值。
  6. 根据权利要求4所述的一种无位置传感器的BLDC电机的启动方法，其特征 在于：电机驱动的是一个可变负载，启动力矩T0是一个预设值，是通过实验手段测量的，即要启动最小负载时需要的力矩值。
  7. 根据权利要求1至6任何一项所述的一种无位置传感器的BLDC电机的启动方法，其特征在于：步骤C：启动运转阶段，利用所有的闭环PID控制来控制电机，每间隔一段时间提升力矩△T，直到转速度达到要求。
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