WO2018133071A1 - 一种双频天线 - Google Patents

一种双频天线 Download PDF

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Publication number
WO2018133071A1
WO2018133071A1 PCT/CN2017/072085 CN2017072085W WO2018133071A1 WO 2018133071 A1 WO2018133071 A1 WO 2018133071A1 CN 2017072085 W CN2017072085 W CN 2017072085W WO 2018133071 A1 WO2018133071 A1 WO 2018133071A1
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WIPO (PCT)
Prior art keywords
waveguide
electromagnetic wave
frequency
wall
frequency feed
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Application number
PCT/CN2017/072085
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English (en)
French (fr)
Inventor
罗昕
林红勇
郭智力
Original Assignee
华为技术有限公司
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
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Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to PCT/CN2017/072085 priority Critical patent/WO2018133071A1/zh
Priority to JP2019506088A priority patent/JP6707269B2/ja
Priority to PL17893209T priority patent/PL3419113T3/pl
Priority to CN202110008312.7A priority patent/CN112821076A/zh
Priority to CN201780014019.3A priority patent/CN108701900B/zh
Priority to EP20187178.7A priority patent/EP3790113B1/en
Priority to EP17893209.1A priority patent/EP3419113B1/en
Publication of WO2018133071A1 publication Critical patent/WO2018133071A1/zh
Priority to US16/134,519 priority patent/US10916849B2/en
Priority to US17/101,826 priority patent/US11652294B2/en

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/50—Feeding or matching arrangements for broad-band or multi-band operation
    • H01Q5/55—Feeding or matching arrangements for broad-band or multi-band operation for horn or waveguide antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/02—Waveguide horns
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/06—Waveguide mouths
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q19/00—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic
    • H01Q19/10—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q19/00—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic
    • H01Q19/10—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces
    • H01Q19/18—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces having two or more spaced reflecting surfaces
    • H01Q19/19—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using reflecting surfaces having two or more spaced reflecting surfaces comprising one main concave reflecting surface associated with an auxiliary reflecting surface
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/40—Imbricated or interleaved structures; Combined or electromagnetically coupled arrangements, e.g. comprising two or more non-connected fed radiating elements
    • H01Q5/45—Imbricated or interleaved structures; Combined or electromagnetically coupled arrangements, e.g. comprising two or more non-connected fed radiating elements using two or more feeds in association with a common reflecting, diffracting or refracting device
    • H01Q5/47—Imbricated or interleaved structures; Combined or electromagnetically coupled arrangements, e.g. comprising two or more non-connected fed radiating elements using two or more feeds in association with a common reflecting, diffracting or refracting device with a coaxial arrangement of the feeds

Definitions

  • the present application relates to the field of wireless communications, and in particular, to a coaxial dual-band antenna that can be used for a dual-frequency parabolic antenna.
  • the microwave equipment in the Eband (71-76 GHz, 81-86 GHz) frequency band plays an increasingly important role in the base station backhaul network.
  • the electromagnetic wave "rain decay" of the Eband band is particularly serious, the Eband microwave single-hop distance is usually less than 3 kilometers.
  • one solution is to use the Eband band microwave device in combination with other low-frequency microwave devices. When there is a large rainfall, the Eband microwave device cannot work normally, but the low-frequency microwave The device is still working.
  • the solution uses a dual-frequency parabolic antenna, the structure of which is shown in Figure 1.
  • the dual-frequency parabolic antenna includes a primary reflecting surface, a secondary reflecting surface, a low frequency feed and a high frequency feed, wherein the high frequency feed is inserted in the low frequency feed.
  • the two are coaxial, forming a coaxial dual-frequency antenna; the two feeds of the coaxial dual-frequency antenna share a primary reflective surface and a secondary reflective surface, and the phase centers of the two feeds coincide with the secondary reflective surface. Focus, thus achieving dual-frequency multiplexing.
  • the low frequency feed of the coaxial dual-frequency antenna is usually a large-opening horn shape, and the high-frequency feed needs to be inserted with a medium needle, and the radiation efficiency is low regardless of the high-frequency feed and the low-frequency feed.
  • the gain does not reach the single-frequency antenna gain level.
  • the embodiment of the present application provides a coaxial dual-frequency antenna, which uses a circular waveguide with a constant diameter or a micro-opening circular waveguide to replace the large-opening horn waveguide as a low-frequency feed, thereby solving the problem of the existing coaxial dual-frequency antenna.
  • Both the frequency feed and the low frequency feed have low radiation efficiency and the gain does not reach the gain level of the single frequency antenna.
  • a coaxial dual-frequency antenna comprising: a waveguide, an annular groove, a high frequency feed, and a dielectric ring, the waveguide being a cylindrical structure for transmitting a first electromagnetic wave, the waveguide
  • the pipe wall has an annular groove whose opening direction is the same as the output direction of the first electromagnetic wave, wherein the frequency of the first electromagnetic wave is lower than the frequency of the electromagnetic wave emitted by the high frequency feed; the high frequency feed is located at the In the waveguide, coaxial with the waveguide, wherein the first electromagnetic wave excites a transverse electric mode TE 11 in the waveguide; the dielectric ring is filled in the waveguide and the high frequency feed
  • the medium ring is a multi-layer structure coaxial with the waveguide, and an area of a plane of each layer of the medium ring perpendicular to the axis alternates, wherein the height of the medium ring is smaller than the The height of the waveguide.
  • the coaxial dual-frequency antenna provided by the embodiment of the present application will excite the TE 11 mode of the first electromagnetic wave of low frequency, and does not generate a high-order mode inside the waveguide, avoiding the transmission loss of the high-order mode inside the waveguide, and improving the low-frequency radiation efficiency of the dual-frequency antenna. Moreover, the high-order mode is not generated inside the waveguide, and there is no need to worry that the high-frequency feed in the waveguide affects the electromagnetic field distribution of the high-order mode, and the medium needle can be omitted to improve the high-frequency radiation efficiency of the dual-frequency antenna.
  • the height of the high frequency feed is the same as the height of the waveguide.
  • a sum of a radius of the inner wall of the waveguide and a radius of the outer wall of the high frequency feed is greater than 1/ ⁇ of the wavelength of the first electromagnetic wave, the two The difference in radii is less than 1/2 of the wavelength of the first electromagnetic wave.
  • the radius of the annular groove and the radius of the inner wall of the waveguide The difference is 1/8 of the wavelength of the first electromagnetic wave.
  • the depth of the annular groove is between 1/5 and 1/4 of the wavelength of the first electromagnetic wave
  • the width is 1/8 of the wavelength of the first electromagnetic wave.
  • the above two embodiments provide the size requirements of the annular groove, and the high-order mode excited by the annular groove satisfying the above-mentioned size requirement can be superimposed with the TE 11 mode, so that the beam width of the first electromagnetic wave on the E surface and the H surface is uniform, so that The radiation efficiency of an electromagnetic wave is maximized.
  • the adjacent two-layer media ring Only one outer wall of the medium ring is connected to the inner wall of the waveguide, and the inner wall is connected to the outer wall of the high frequency feed, which can function as a sealing, waterproof and fixed high frequency feed.
  • the layer of media that is furthest from the output plane is not connected to the waveguide and the high frequency feed at the same time, which can reduce the reflection of the first electromagnetic wave on the dielectric ring and improve the radiation efficiency.
  • the height of each layer of the dielectric ring is 1/4 of the wavelength of the first electromagnetic wave.
  • the dielectric ring has a relative dielectric constant between 2 and 4.
  • the above two embodiments describe the height and relative dielectric constant of each layer of the dielectric ring, and can realize the matching of the characteristic impedance of the coaxial dual-frequency antenna and the wave impedance of the free space to improve the radiation efficiency.
  • the coaxial dual-frequency antenna provided by the present application will excite the TE 11 mode of the first electromagnetic wave of low frequency, and does not generate a high-order mode inside the waveguide, avoiding the transmission loss of the high-order mode inside the waveguide, and improving the low-frequency radiation efficiency of the dual-frequency antenna;
  • the high-order mode is not generated inside the waveguide, so there is no need to worry that the high-frequency feed in the waveguide affects the electromagnetic field distribution of the high-order mode, and the medium needle can be omitted to improve the high-frequency radiation efficiency of the dual-frequency antenna.
  • FIG. 1 is a schematic structural view of a conventional dual-frequency parabolic antenna
  • FIG. 2 is a schematic structural view of a conventional coaxial dual-frequency antenna
  • FIG. 3(a) is a schematic structural diagram of a coaxial dual-frequency antenna according to an embodiment of the present application.
  • FIG. 3(b) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 3(c) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • 4(a) is a diagram showing electric field distribution of a TE 11 mode in a coaxial dual-frequency antenna according to an embodiment of the present application
  • FIG. 4 (b) an electric field distribution in the TM 11 mode coaxial dual-band antenna according to an embodiment of the present application
  • 4(c) is a diagram showing an electric field distribution of a TE 11 and a TM 11 mode in a coaxial dual-frequency antenna according to an embodiment of the present application;
  • FIG. 5(a) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 5(b) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 6(a) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 6(b) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 7(a) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 7(b) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 8(a) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • FIG. 8(b) is a schematic structural diagram of a coaxial dual-frequency antenna according to another embodiment of the present application.
  • the existing coaxial dual-frequency antenna structure is shown in FIG. 2, and the low-frequency feed 201 of the coaxial dual-frequency antenna is a large-opening horn waveguide, and includes a high-frequency feed 202 inside the waveguide, and the high-frequency feed 202 A medium needle 203 is inserted.
  • the use of a horn-shaped waveguide facilitates the matching of the characteristic impedance of the waveguide and the wave impedance of the free space, and reduces the reflection; as the radius of the waveguide increases, a higher-order mode is excited, and the higher-order mode and the transverse electric mode TE 11 act to cause the output electromagnetic wave to be in the E
  • the beam width on the surface and the H plane is the same, which has the greatest gain effect.
  • the E plane refers to the plane composed of the direction of the electric field and the maximum direction of the radiation
  • the H plane refers to the direction of the magnetic field and the maximum direction of the radiation. flat.
  • the high-order mode is generated inside the large-opening horn waveguide, and the transmission loss inside the waveguide is large, so the low-frequency radiation efficiency of the dual-frequency antenna is low.
  • the high-frequency feed Since the high-frequency feed is made of metal, it will affect the electromagnetic field distribution of the high-order mode. Therefore, the high-frequency feed cannot directly extend to the mouth of the large-opening horn waveguide. A medium needle is needed to lead the phase center of the high-frequency feed to the large opening. The mouth surface of the horn waveguide, but the medium needle is not easy to process and the loss is large, so that the high frequency gain of the dual frequency antenna can not reach the level of the single frequency antenna.
  • the embodiment of the present application provides a coaxial dual-frequency antenna, as shown in FIG. 3( a ), including: a waveguide 301 , an annular groove 302 , a high frequency feed 303 , and a medium ring 304 .
  • the waveguide 301 is a cylindrical structure for transmitting a first electromagnetic wave, and the wall of the waveguide 301 has an annular groove 302 whose opening direction is the same as the output direction of the first electromagnetic wave, wherein the frequency of the first electromagnetic wave is lower than the high frequency The frequency of the electromagnetic wave emitted by the feed 303;
  • the high frequency feed 303 is located in the waveguide 301 coaxial with the waveguide 301, wherein the first electromagnetic wave excites the transverse electric mode TE 11 in the waveguide 301;
  • the dielectric ring 304 is filled between the waveguide 301 and the high frequency feed 303.
  • the dielectric ring 304 has a multi-layer structure coaxial with the waveguide 301, and the area of the plane of each layer of the dielectric ring 304 perpendicular to the axis alternates.
  • the height of the dielectric ring 304 is smaller than the height of the waveguide 301.
  • the height of the high frequency feed 303 is the same as the height of the waveguide 301. It should be understood that it is also feasible that the height of the high frequency feed is slightly less than the height of the waveguide.
  • the waveguide will excite the TE 11 mode of the first electromagnetic wave of low frequency, and no high-order mode is generated inside the waveguide, which avoids the transmission loss of the high-order mode inside the waveguide and improves the low-frequency radiation efficiency of the dual-frequency antenna;
  • the high-order mode is not generated inside the waveguide, so there is no need to worry that the high-frequency feed in the waveguide affects the electromagnetic field distribution of the high-order mode, and the medium needle can be omitted to improve the high-frequency radiation efficiency of the dual-frequency antenna.
  • the coaxial dual-frequency antenna shown in FIG. 3( a ) is a case where the inner wall of the dielectric ring 304 is connected to the outer wall of the high-frequency feed 303 , which is only one possible structure of the coaxial dual-frequency antenna provided by the present application.
  • the antenna may also be connected to the inner wall of the waveguide 301 as the outer wall of the dielectric ring 304, as shown in FIG. 3(b); or the inner wall of the one or more dielectric rings 304 may be connected to the outer wall of the high frequency feed 303.
  • the outer walls of the remaining layers of the dielectric ring are connected to the inner wall of the waveguide 301 as shown in FIG. 3(c); it is only necessary to alternately change the plane area of each of the dielectric rings 304 perpendicular to the axis.
  • the electromagnetic field distribution of the waveguide cross section is called the propagation mode of the waveguide.
  • Different propagation modes have different cutoff wavelengths.
  • the mode with no cutoff wavelength or maximum cutoff wavelength is called the main mode or the fundamental mode.
  • Other modes with smaller cutoff wavelengths are collectively called higher order modes.
  • the higher the order of the propagation mode the smaller the cutoff wavelength. .
  • the TE 11 mode is regarded as a fundamental mode, and other modes whose cutoff wavelength is smaller than the TE 11 mode are collectively referred to as a high-order mode.
  • the waveguide provided by the embodiment of the present application may be in the shape of a cylinder, a square cylinder, or the like, or may be slightly enlarged in the opening of the first electromagnetic wave, and only needs to be satisfied by the waveguide, the high frequency feed, and the annular groove.
  • the coaxial dual-frequency antenna composed of the medium ring only the fundamental mode of the first electromagnetic wave is excited; wherein the wall of the tube is usually made of metal.
  • the sum of the radius of the inner wall of the waveguide 302 and the radius of the outer wall of the high frequency feed 303 is greater than 1/ ⁇ of the wavelength of the first electromagnetic wave, and the difference between the two radii is less than 1/2 of the wavelength of the first electromagnetic wave, wherein The frequency of the first electromagnetic wave is lower than the frequency of the electromagnetic wave emitted by the high frequency feed 303.
  • the cutoff wavelength of the different modes of the first electromagnetic wave and the outer diameter of the inner waveguide of the coaxial waveguide a (high frequency feed)
  • the radius of the outer wall of 303 is related to the inner diameter b of the outer waveguide (the radius of the inner wall of the waveguide 301), and the corresponding relationship is shown in Table 1.
  • the wavelength of the first electromagnetic wave is ⁇
  • the first electromagnetic wave may exist in the TE 11 mode. If b of the coaxial waveguide becomes large, such that (ba)> ⁇ /2, (b+a) ⁇ 2 ⁇ / ⁇ , the first electromagnetic wave can theoretically exist in TE 11 , TM m1 , TE 01, etc., but Since the tangential component is continuous when the electromagnetic field mode changes, that is, m must be consistent. Therefore, only TE 11 and TM 11 modes exist in practice; as the outer diameter b of the outer waveguide of the coaxial waveguide increases, the existing mode exists. Will gradually increase.
  • the embodiment of the present application excavates an annular groove 302 having the same opening direction as the first electromagnetic wave output direction on the wall of the waveguide 301, and the discontinuity of the tube wall of the waveguide 301 is used to excite the high-order mode.
  • the electric field distribution of the TE 11 mode is made uniform by the higher order mode, wherein the depth and width of the annular groove 302 and the distance between the annular groove 302 and the inner wall of the waveguide 301 affect the order and amplitude of the higher order mode.
  • the difference between the radius of the annular groove 302 and the radius of the inner wall of the waveguide 301 is 1/8 of the wavelength of the first electromagnetic wave.
  • the annular groove 302 has a depth between 1/5 and 1/4 of the wavelength of the first electromagnetic wave and a width of 1/8 of the wavelength of the first electromagnetic wave.
  • a wall having a width and a depth satisfying the above requirements is excavated to form an annular groove 302, and the annular groove is formed.
  • 302 creates a discontinuity in the surface of the tube wall that excites higher order modes.
  • the position, width and depth of the annular groove 302 satisfy the above requirements, and a high-order mode TM 11 of a suitable amplitude can be generated.
  • the electric field distribution is as shown in FIG. 4(b), and the TE 11 mode and the TM 11 mode are superimposed to make the first
  • the electric field distribution of an electromagnetic wave becomes uniform, as shown in Fig. 4(c), so that the beam widths of the first electromagnetic wave on the E plane and the H plane are uniform, and the gain effect is maximized.
  • the horn-shaped waveguide with a large opening is omitted, and the characteristic impedance is gradually changed by gradually increasing the diameter of the waveguide at the output end of the waveguide, so that the characteristic impedance and freedom of the coaxial dual-frequency antenna cannot be realized.
  • the wave impedances of the space are matched with each other.
  • impedance matching can be implemented in the following two ways:
  • Impedance matching is achieved using a dielectric ring 304 filled between the waveguide 301 and the high frequency feed 303.
  • the medium ring 304 has a multi-layer structure coaxial with the waveguide 301.
  • the area of the plane of each layer of the dielectric ring 304 perpendicular to the axis alternates, and the height of the dielectric ring 304 is smaller than the height of the waveguide 301, and the structure thereof can be It is any of FIG. 3 (a), FIG. 3 (b), and FIG. 3 (c).
  • the characteristic impedance of the waveguide is equal to the load impedance after the matching segment transformation:
  • R 0 is the characteristic impedance of the waveguide and R L is the load impedance.
  • the load impedance is the wave impedance of the free space
  • the characteristic impedance of the waveguide is the characteristic impedance of the coaxial dual-frequency antenna
  • the dielectric ring 304 structure used in the present application does not completely fill the gap between the waveguide 301 and the high frequency feed 303, but uses a multi-layer structure coaxial with the waveguide 301, and the vertical of each layer of the dielectric ring 304
  • the plane area of the axis alternates to form a mixture of medium and air, so the equivalent relative dielectric constant is no longer equal to the relative dielectric constant of the material itself, but can be controlled to change, and the target of control change is
  • the characteristic impedance of the matching segment is such that the value calculated by the above formula is obtained.
  • the height of each layer of the dielectric ring 304 is 1/4 of the wavelength of the first electromagnetic wave, and the first electromagnetic wave is a low frequency electromagnetic wave emitted by the coaxial dual frequency antenna.
  • the inner wall of the multilayer dielectric ring 304 is connected to the outer wall of the high frequency feed 303, and the outer wall is connected to the inner wall of the waveguide 301, and can function as a gas-tight, waterproof and fixed intermediate high-frequency feed 303.
  • the coaxial dual-frequency antenna be applied to the ground, not just for satellite communications.
  • the spacing between the inner and outer walls of the other dielectric rings 304 is optimized in accordance with the principle of equivalent dielectric constant described above.
  • the layer of dielectric ring 304 that is the farthest from the output plane of the waveguide 301 is not connected to the waveguide 301 and the high frequency feed 303 at the same time, reducing the reflection of the first electromagnetic wave, wherein the distance from the output plane is the most
  • the far layer of media is the lowermost dielectric ring in Figures 5(a) and 5(b).
  • the medium ring of the embodiment of the present application may be a dielectric material having a relative dielectric constant of 2-4, such as polycarbonate, polystyrene, polytetrafluoroethylene, etc., and the specific materials used in the embodiments of the present application are not limited. .
  • the spacing between the inner wall and the outer wall of each layer of dielectric ring 304 is also related to the wavelength of the first electromagnetic wave.
  • a specific example is given below when the frequency of the first electromagnetic wave is 18 GHz, assuming a relative dielectric constant of 2.8.
  • Polycarbonate is used to make the medium ring.
  • the radius of the inner wall of the waveguide is R.
  • the first layer of medium The radius of the outer wall of the ring, the third dielectric ring and the fifth dielectric ring are both R, the radius of the outer wall of the second dielectric ring is 0.78R, the radius of the outer wall of the fourth dielectric ring is 0.7R, and the outer wall of the sixth dielectric ring The radius is 0.7R.
  • Impedance matching is achieved by providing a plurality of metal rings 601 in the waveguide.
  • the metal ring is a matching section.
  • One possible structure is shown in Fig. 6(a).
  • the inner wall of each metal ring 601 is connected to the outer wall of the high frequency feed 303, and the radius of each metal ring 601 can be changed.
  • the spacing between the metal rings 601 to change the equivalent inductance and equivalent capacitance of the metal ring 601, so that the characteristic impedance of the matching segment reaches the value calculated by the formula (1).
  • a dielectric layer 602 may also be filled inside the waveguide 301 near the output plane, the inner wall of the dielectric layer 602 being connected to the outer wall of the high frequency feed 303, the outer wall of the dielectric layer 602 and the waveguide 301 The inner walls are connected, as shown in Figure 6(b), which acts as a seal, waterproof and fixed high frequency feed.
  • the dielectric layer 602 may be a hard material, and the specific material is not limited in the present application.
  • FIG. 6(a) and FIG. 6(b) are also only one possible structure of the embodiment of the present application, and the outer wall of the metal ring 601 may be connected to the inner wall of the waveguide 301 to form a matching segment, as shown in FIG. 7. (a) and FIG. 7(b); or the outer wall of the partial metal ring 601 is connected to the inner wall of the waveguide 301, and the inner wall of the other part of the metal ring 601 is connected to the outer wall of the high-frequency feed 303 to form a matching section, as shown in the figure. 8(a) and Figure 8(b).
  • the specific implementation manner of the embodiment of the present application is not limited.
  • the coaxial dual-frequency antenna provided by the present application has the following advantages: the waveguide 301 will excite the TE 11 mode of the low-frequency first electromagnetic wave, and no high-order mode is generated inside the waveguide 301, thereby avoiding the transmission loss of the high-order mode in the waveguide 301.
  • the low-frequency radiation efficiency of the dual-frequency antenna is improved; further, the high-order mode is not generated inside the waveguide 301, and there is no need to worry that the high-frequency feed 303 located in the waveguide 301 affects the electromagnetic field distribution of the high-order mode, and the medium needle can be omitted to improve the dual-frequency antenna. High frequency radiation efficiency.
  • the beam widths of the first electromagnetic wave on the E surface and the H surface can be ensured, and the characteristic impedance of the coaxial dual frequency antenna and the wave impedance of the free space can be matched with each other.

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Abstract

本申请公开一种同轴双频天线，包括：波导管，环形槽，高频馈源和介质环。波导管为筒形结构，用于传输第一电磁波，波导管的管壁具有开口方向与第一电磁波的输出方向相同的环形槽，其中，第一电磁波的频率低于高频馈源发出的电磁波的频率；高频馈源位于波导管中，与波导管同轴，其中，第一电磁波在波导管中激发横电模TE11；介质环填充在波导管和高频馈源之间，介质环为多层结构，与波导管同轴，各层介质环的垂直于该轴的平面的面积大小呈交替变化，介质环的高度小于波导管的高度。本申请相对于现有技术，可以避免高阶模在波导管内的损耗，并省略介质环，提高同轴双频天线的辐射效率。

Description

一种双频天线 技术领域
本申请涉及无线通信领域，尤其涉及一种可用于双频抛物面天线的同轴双频天线。
背景技术
随着无线通信技术的迅速发展，微波点到点通信的传输容量不断增大，Eband(71-76GHz，81-86GHz)频段的微波设备在基站回传网络中发挥越来越重要的作用。但是，因为Eband频段的电磁波“雨衰”特别严重，所以Eband微波单跳距离通常小于3公里。为了增大Eband微波的单跳距离，降低建站成本，有一种解决方案是将Eband频段微波设备和其他低频微波设备组合使用，当出现较大降雨时，Eband微波设备虽然无法正常工作，但是低频微波设备依然可以正常工作。
该解决方案采用双频抛物面天线，其结构如图1所示，该双频抛物面天线包括主反射面，副反射面，低频馈源和高频馈源，其中，高频馈源插在低频馈源内部，二者共轴，形成同轴双频天线；该同轴双频天线的两个馈源共用一个主反射面和一个副反射面，两个馈源的相位中心重合于副反射面的焦点，从而实现双频复用的功能。
在现有技术中，同轴双频天线的低频馈源通常为大开口的喇叭形状，高频馈源需插有介质针，无论高频馈源和低频馈源，均存在辐射效率偏低，增益达不到单频天线增益水平的问题。
发明内容
本申请实施例提供一种同轴双频天线，采用直径不变的圆波导或者微张口圆波导来代替大开口的喇叭形波导做低频馈源，解决了现有的同轴双频天线的高频馈源和低频馈源均存在辐射效率偏低，增益达不到单频天线增益水平的问题。
第一方面，提供一种同轴双频天线，包括：波导管，环形槽，高频馈源和介质环，所述波导管为筒形结构，用于传输第一电磁波，所述波导管的管壁具有开口方向与所述第一电磁波的输出方向相同的环形槽，其中，所述第一电磁波的频率低于所述高频馈源发出的电磁波的频率；所述高频馈源位于所述波导管中，与所述波导管同轴，其中，所述第一电磁波在所述波导管中激发横电模TE11；所述介质环填充在所述波导管和所述高频馈源之间，所述介质环为多层结构，与所述波导管同轴，各层介质环的垂直于所述轴的平面的面积大小呈交替变化，其中，所述介质环的高度小于所述波导管的高度。
本申请实施例提供的同轴双频天线将激发低频的第一电磁波的TE11模，在波导管内部不会产生高阶模，避免了高阶模在波导内部的传输损耗，提高双频天线的低频辐射效率；而且，波导管内部不产生高阶模，也就无需担心位于波导管内的高频馈源会影响高阶模的电磁场分布，可以省略介质针，提高双频天线的高频辐射效率。
结合第一方面，在第一方面的第一种可能的实现方式，所述高频馈源的高度与所述波导管的高度相同。
结合第一方面，在第一方面的第二种可能的实现方式，所述波导管内壁的半径和高频馈源外壁的半径之和大于所述第一电磁波波长的1/π，上述两个半径之差小于所述第一电磁波波长的1/2。本实施例可以保证在天线内只会激发TE11模，不会有更高阶的模式存在，避免了高阶模在波导内的传输损耗。
结合第一方面或第一方面的第一种或第二种可能的实现方式，在第一方面的第三种可能的实现方式中，所述环形槽的半径与所述波导管内壁的半径之差为所述第一电磁波波长的1/8。
结合第一方面的第三种可能的实现方式，在第一方面的第四种可能的实现方式中，所述环形槽的深度在所述第一电磁波波长的1/5至1/4之间，宽度为所述第一电磁波波长的1/8。
上述两个实施例提供了环形槽的尺寸要求，满足上述尺寸要求的环形槽激励起的高阶模，可以通过和TE11模叠加，使第一电磁波在E面和H面的波束宽度一致，使第一电磁波的辐射效率最大化。
结合第一方面或第一方面的第一种至第四种可能的实现方式中的任一种可能的实现方式，在第一方面的第五种可能的实现方式中，相邻两层介质环中只有一个介质环的外壁与所述波导管的内壁相连，且内壁与所述高频馈源的外壁相连，可以起到密封、防水以及固定高频馈源的作用。
结合第一方面或第一方面的第一种至第五种可能的实现方式中的任一种可能的实现方式，在第一方面的第六种可能的实现方式中，与所述波导管的输出平面距离最远的那层介质环不同时与所述波导管和所述高频馈源相连，可以降低第一电磁波在介质环上的反射，提高辐射效率。
结合第一方面的第六种可能的实现方式，在第一方面的第七种可能的实现方式中，每层介质环的高度为所述第一电磁波波长的1/4。
结合第一方面的第六种或第七种可能的实现方式，在第一方面的第八种可能的实现方式中，所述介质环的相对介电常数在2至4之间。
上述两个实施例描述了每层介质环的高度和相对介电常数，可以实现同轴双频天线的特征阻抗和自由空间的波阻抗相互匹配，提高辐射效率。
本申请提供的同轴双频天线将激发低频的第一电磁波的TE11模，在波导管内部不会产生高阶模，避免了高阶模在波导内部的传输损耗，提高双频天线的低频辐射效率；而且，波导管内部不产生高阶模，也就无需担心位于波导管内的高频馈源会影响高阶模的电磁场分布，可以省略介质针，提高双频天线的高频辐射效率。
附图说明
图1为一种现有的双频抛物面天线的结构示意图；
图2为现有的同轴双频天线的结构示意图；
图3(a)为本申请一实施例提供的同轴双频天线的结构示意图；
图3(b)为本申请另一实施例提供的同轴双频天线的结构示意图；
图3(c)为本申请另一实施例提供的同轴双频天线的结构示意图；
图4(a)为本申请一实施例提供的同轴双频天线内TE11模的电场分布图；
图4(b)为本申请一实施例提供的同轴双频天线内TM11模的电场分布图；
图4(c)为本申请一实施例提供的同轴双频天线内TE11和TM11模叠加后的电场分布图；
图5(a)为本申请另一实施例提供的同轴双频天线的结构示意图；
图5(b)为本申请另一实施例提供的同轴双频天线的结构示意图；
图6(a)为本申请另一实施例提供的同轴双频天线的结构示意图；
图6(b)为本申请另一实施例提供的同轴双频天线的结构示意图；
图7(a)为本申请另一实施例提供的同轴双频天线的结构示意图；
图7(b)为本申请另一实施例提供的同轴双频天线的结构示意图；
图8(a)为本申请另一实施例提供的同轴双频天线的结构示意图；
图8(b)为本申请另一实施例提供的同轴双频天线的结构示意图。
具体实施方式
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行描述。
现有的同轴双频天线结构如图2所示，该同轴双频天线的低频馈源201为大开口的喇叭形波导，在波导内部包括高频馈源202，在高频馈源202上插有介质针203。采用喇叭形波导便于实现波导的特征阻抗和自由空间的波阻抗相互匹配，减少反射；随着波导半径的增大，会激发出高阶模，该高阶模和横电模TE11作用，使输出电磁波在E面和H面上的波束宽度一致，起到最大的增益效果，其中，E面指的是电场所在方向和辐射最大方向组成的平面,而H面指的是磁场所在方向和辐射最大方向组成的平面。然而，高阶模产生于大开口喇叭形波导的内部，在波导内部的传输损耗较大，故该双频天线的低频辐射效率较低。
由于高频馈源是金属材质，会影响高阶模的电磁场分布，故高频馈源不能直接伸到大开口喇叭形波导的口面，需要一个介质针将高频馈源的相位中心引出到大开口喇叭形波导的口面，但介质针不易加工且损耗较大，导致双频天线的高频增益也达不到单频天线的水平。
本申请实施例提供一种同轴双频天线，如图3(a)所示，包括：波导管301，环形槽302、高频馈源303和介质环304，
波导管301为筒形结构，用于传输第一电磁波，波导管301的管壁具有开口方向与该第一电磁波的输出方向相同的环形槽302，其中，该第一电磁波的频率低于高频馈源303发出的电磁波的频率；
高频馈源303位于波导管301中，与波导管301同轴，其中，该第一电磁波在波导管301中激发横电模TE11；
介质环304填充在波导管301和高频馈源303之间，介质环304为多层结构，与波导管301同轴，每层介质环304的垂直于该轴的平面面积大小呈交替变化，其中，介质环304的高度小于波导管301的高度。
可选地，高频馈源303的高度与波导管301的高度相同。应理解，高频馈源的高度略小于波导管的高度也是可行的。
在本申请实施例中，波导管将激发低频的第一电磁波的TE11模，在波导管内部 不会产生高阶模，避免了高阶模在波导内部的传输损耗，提高双频天线的低频辐射效率；而且，波导管内部不产生高阶模，也就无需担心位于波导管内的高频馈源会影响高阶模的电磁场分布，可以省略介质针，提高双频天线的高频辐射效率。
应理解，图3(a)所示的同轴双频天线是介质环304的内壁与高频馈源303的外壁相连的情况，这只是本申请提供的同轴双频天线的一种可能结构，该天线还可以为介质环304的外壁与波导管301的内壁相连，如图3(b)所示；还可以为一层或多层介质环304的内壁与高频馈源303的外壁相连，剩下的各层介质环的外壁与波导管301的内壁相连，如图3(c)所示；只需满足每层介质环304的垂直于该轴的平面面积大小呈交替变化即可。
需要说明的是，波导横截面的电磁场分布，称为波导的传播模式。不同的传播模式有不同的截止波长，没有截止波长或截止波长最大的模式称为主模或者基模，其他截止波长更小的模式统称为高阶模，传播模式的阶数越高，截止波长越小。在本申请实施例中，将TE11模当做基模，截止波长小于TE11模的其他模式统称高阶模。
应理解，本申请实施例提供的波导管可以为圆筒、方形筒等形状，也可以在输出第一电磁波的开口稍微扩大一些，只需满足在由该波导管、高频馈源、环形槽和介质环构成的同轴双频天线中，只会激发第一电磁波的基模即可；其中，管壁通常为金属材质。
可选地，波导管302内壁的半径和高频馈源303外壁的半径之和大于第一电磁波波长的1/π，上述两个半径之差小于该第一电磁波波长的1/2，其中，该第一电磁波的频率低于高频馈源303发出的电磁波的频率。
具体的，以本申请描述的由高频馈源303与波导管301形成的同轴波导为例，第一电磁波的不同模式的截止波长与同轴波导的内波导外径a(高频馈源303外壁的半径)和外波导内径b(波导管301内壁的半径)有关，对应关系如表1所示。
表1
传播模式 截止波长
TEM 没有截止波长
TE11 π×(b+a)
TMm1(m＝0,1,2……)、TE01 2×(b-a)
TE21 π×(b+a)/2
TEm1(m＝3,4,5……) π×(b+a)/m
TMm2、TE02 b-a
TMmn(n＝3,4,5……)、TE0n(n＝3,4,5……) 2×(b-a)/n
假设第一电磁波的波长为λ，则从表1可以知道，在同轴波导满足(b+a)>λ/π，(b-a)<λ/2的情况下，第一电磁波可以存在TE11模；如果同轴波导的b变大，使得(b-a)>λ/2，(b+a)<2λ/π时，第一电磁波在理论上可以存在TE11、TMm1、TE01等模式，但是由于电磁场模式变化时要保证切向分量连续，也就是m要一致，因此，实际只会存在TE11和TM11两种模式；随着同轴波导的外波导内径b的增大，存在的模式会逐渐增多。
需要说明的是，在同轴波导中还会存在横电磁模TEM，这种模式不存在截止波长， 或者说它的截止波长无穷大，但是TEM模在进入同轴双频天线之前，就已经通过对称馈电的方式进行抑制，故本申请实施例不考虑该模式。
进一步地，由于波导管内部只存在TE11模，而波导管内的TE11模的电场分布是不均匀的，如图4(a)所示，也就是说，第一电磁波的电场分布不均匀，从而导致第一电磁波在E面和H面上的波束宽度不一致。针对上述问题，本申请实施例通过在波导管301的管壁上挖一个具有与该第一电磁波输出方向相同的开口方向的环形槽302，利用该波导管301管壁的不连续性激励起高阶模，利用高阶模使TE11模的电场分布变均匀，其中，环形槽302的深度和宽度以及环形槽302与波导管301内壁的距离，均会影响高阶模的阶数和幅度。
可选地，环形槽302的半径与波导管301内壁的半径之差为该第一电磁波波长的1/8。环形槽302的深度在该第一电磁波波长的1/5至1/4之间，宽度为该第一电磁波波长的1/8。具体的，在波导管输出端的管壁平面上，离波导管内壁距离1/8个第一电磁波波长的位置，挖掉一圈宽度和深度满足上述要求的管壁，形成环形槽302，环形槽302使管壁表面产生了不连续性，会激励起高阶模。而环形槽302的位置、宽度以及深度满足上述要求，可以产生合适幅度的高阶模TM11，其电场分布如图4(b)所示，将TE11模和TM11模叠加在一起，会使第一电磁波的电场分布变得均匀，如图4(c)所示，从而使第一电磁波在E面和H面上的波束宽度一致，让增益效果最大化。
另外，本申请实施例省掉了大开口的喇叭形波导，没有在波导管的输出端通过逐渐增大波导管的直径来逐渐改变特征阻抗，也就无法实现同轴双频天线的特征阻抗与自由空间的波阻抗相互匹配，本申请实施例可以采用如下两种方式实现阻抗匹配：
(1)采用填充在波导管301和高频馈源303之间的介质环304来实现阻抗匹配。介质环304为多层结构，与波导管301同轴，各层介质环304的垂直于该轴的平面的面积大小呈交替变化，而且介质环304的高度小于波导管301的高度，其结构可为图3(a)、图3(b)及图3(c)中的任意一种。
根据阻抗匹配原理，当负载阻抗和波导的特征阻抗不一致时，为了使能量能够传递给负载，而不反射回去，需要在负载和波导之间有一段匹配段，当匹配段的特征阻抗Z0需要满足下面公式时，波导的特征阻抗经过匹配段变换后和负载阻抗相等：
Figure PCTCN2017072085-appb-000001
其中，R0为波导的特征阻抗，RL为负载阻抗。
在本申请实施例中，负载阻抗即为自由空间的波阻抗，波导特征阻抗即为同轴双频天线的特征阻抗；在波导管内填充介质可以改变波导管的特征阻抗，也就是说，填充的介质环形成了匹配段。但是如果全部填满介质的话，在波导管内，介质和空气的界面又形成了特征阻抗突变的情况，会产生强烈的反射。
本申请采用的介质环304结构，并不是完全填充了波导管301和高频馈源303之间的空隙，而是采用了与波导管301同轴的多层结构，每层介质环304的垂直于该轴的平面面积大小呈交替变化，形成介质和空气的混合物，所以等效的相对介电常数不再等于材料自身的相对介电常数，而是可以控制改变的，控制改变的目标就 是使得匹配段的特征阻抗达到上面公式计算得到的值。
可选地，每层介质环304的高度为该第一电磁波波长的1/4，该第一电磁波为同轴双频天线发射的低频电磁波。
可选地，相邻两层介质环304中只有一层介质环304的外壁与波导管301的内壁相连，且内壁与高频馈源303的外壁相连，如图5(a)或图5(b)所示的结构。这样，会有多层介质环304内壁与高频馈源303的外壁相连，且外壁与波导管301的内壁相连，可以起到气密、防水以及固定中间的高频馈源303的功能，可以让该同轴双频天线适用于地面，而不仅仅用于卫星通信中。除了与波导管301和高频馈源303均相连的那几层介质环，其他层介质环304的内外壁之间的间距，需根据前面所述的等效介电常数原则进行设计优化。
可选地，与波导管301的输出平面距离最远的那层介质环304不同时与波导管301和高频馈源303相连，降低对第一电磁波的反射，其中，与该输出平面距离最远的那层介质环即为图5(a)和图5(b)中最下面的那层介质环。
本申请实施例的介质环可以采用相对介电常数在2-4之间的介质材料，例如聚碳酸脂、聚苯乙烯、聚四氟乙烯等，本申请实施例对采用的具体材料并不限定。
确定了材料之后，每层介质环304内壁和外壁的间距还与第一电磁波的波长有关，下面给出在第一电磁波的频率为18GHz时的具体实施例，假设采用相对介电常数为2.8的聚碳酸脂制作介质环，波导管内壁的半径为R，我们采用六层介质环，如图5(a)所示，从上到下，各层介质环的半径长短交替变化，第一层介质环、第三层介质环和第五层介质环外壁的半径均为R，第二层介质环外壁的半径为0.78R，第四层介质环外壁的半径为0.7R，第六层介质环外壁的半径为0.7R。采用满足上述尺寸的介质环，可以让匹配段的特征阻抗满足公式(1)，从而实现同轴双频天线的特征阻抗和自由空间的波阻抗相互匹配，减少电磁波的反射，提高辐射效率。
(2)通过在波导管内设置多个金属环601来实现阻抗匹配。该金属环形成了匹配段，一种可能的结构如图6(a)所示，每个金属环601的内壁均与高频馈源303的外壁相连，可以通过改变每个金属环601的半径以及金属环601之间的间距，来改变金属环601的等效电感和等效电容，使得匹配段的特征阻抗达到公式(1)计算得到的值。
可选地，还可以在波导管301内部，靠近输出平面的位置填充介质层602，该介质层602的的内壁与高频馈源303的外壁相连，该介质层602的外壁与波导管301的内壁相连，如图6(b)所示，起到密封，防水和固定高频馈源的作用。该介质层602采用硬质材料即可，本申请对具体的材料并不做限定。
应理解，图6(a)和图6(b)也仅为本申请实施例的一种可能结构，还可以金属环601的外壁与波导管301的内壁相连，来形成匹配段，如图7(a)和图7(b)所示；或者部分金属环601的外壁与波导管301的内壁相连，另一部分金属环601的内壁与高频馈源303的外壁相连，形成匹配段，如图8(a)和图8(b)所示。本申请实施例对具体的实现方式不做限定。
本申请提供的同轴双频天线具有如下优势：波导管301将激发低频的第一电磁波的TE11模，在波导管301内部不会产生高阶模，避免了高阶模在波导管301内的 传输损耗，提高双频天线的低频辐射效率；而且，波导管301内部不产生高阶模，也就无需担心位于波导管301内的高频馈源303会影响高阶模的电磁场分布，可以省略介质针，提高双频天线的高频辐射效率。另外，通过环形槽302和介质环304的设计，可以保证第一电磁波在E面和H面上的波束宽度一致，并实现同轴双频天线的特征阻抗和自由空间的波阻抗相互匹配。
以上所述，仅为本申请的具体实施方式，但本申请的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本申请揭露的技术范围内，可轻易想到变化或替换，都应涵盖在本申请的保护范围之内。因此，本申请的保护范围应以所述权利要求的保护范围为准。

Claims (9)

  1. 一种同轴双频天线，其特征在于，包括：波导管，环形槽，高频馈源和介质环，
    所述波导管为筒形结构，用于传输第一电磁波，所述波导管的管壁具有开口方向与所述第一电磁波的输出方向相同的环形槽，其中，所述第一电磁波的频率低于所述高频馈源发出的电磁波的频率；
    所述高频馈源位于所述波导管中，与所述波导管同轴，其中，所述第一电磁波在所述波导管中激发横电模TE11；
    所述介质环填充在所述波导管和所述高频馈源之间，所述介质环为多层结构，与所述波导管同轴，各层介质环的垂直于所述轴的平面的面积大小呈交替变化，其中，所述介质环的高度小于所述波导管的高度。
  2. 根据权利要求1所述的天线，其特征在于，所述高频馈源的高度与所述波导管的高度相同。
  3. 根据权利要求1所述的天线，其特征在于，所述波导管内壁的半径和高频馈源外壁的半径之和大于所述第一电磁波波长的1/π，上述两个半径之差小于所述第一电磁波波长的1/2。
  4. 根据权利要求1至3中任一项所述的天线，其特征在于，所述环形槽的半径与所述波导管内壁的半径之差为所述第一电磁波波长的1/8。
  5. 根据权利要求4所述的天线，其特征在于，所述环形槽的深度在所述第一电磁波波长的1/5至1/4之间，宽度为所述第一电磁波波长的1/8。
  6. 根据权利要求1至3中任一项所述的天线，其特征在于，相邻两层介质环中只有一个介质环的外壁与所述波导管的内壁相连，且内壁与所述高频馈源的外壁相连。
  7. 根据权利要求6所述的天线，其特征在于，与所述波导管的输出平面距离最远的那层介质环不同时与所述波导管和所述高频馈源相连。
  8. 根据权利要求6所述的天线，其特征在于，每层介质环的高度为所述第一电磁波波长的1/4。
  9. 根据权利要求6所述的天线，其特征在于，所述介质环的相对介电常数在2至4之间。
PCT/CN2017/072085 2017-01-22 2017-01-22 一种双频天线 WO2018133071A1 (zh)

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EP17893209.1A EP3419113B1 (en) 2017-01-22 2017-01-22 Dual-frequency antenna
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